General Description
The MAX16818 pulse-width modulation (PWM) LED dri-
ver controller provides high-output-current capability in
a compact package with a minimum number of external
components. The MAX16818 is suitable for use in syn-
chronous and nonsynchronous step-down (buck)
topologies, as well as in boost, buck-boost, SEPIC, and
Cuk LED drivers. The MAX16818 is the first LED driver
controller that enables Maxim’s technology for fast LED
current transients of up to 20A/µs and 30kHz dimming
frequency.
This device utilizes average-current-mode control that
enables optimal use of MOSFETs with optimal charge
and on-resistance characteristics. This results in the
minimized need for external heatsinking even when
delivering up to 30A of LED current. True differential
sensing enables accurate control of the LED current. A
wide dimming range is easily implemented to accom-
modate an external PWM signal. An internal regulator
enables operation over a wide input voltage range:
4.75V to 5.5V or 7V to 28V and above with a simple
external biasing device. The wide switching frequency
range, up to 1.5MHz, allows for the use of small induc-
tors and capacitors.
The MAX16818 features a clock output with 180° phase
delay to control a second out-of-phase LED driver to
reduce input and output filter capacitors size or to mini-
mize ripple currents. The MAX16818 offers programma-
ble hiccup, overvoltage, and overtemperature protection.
The MAX16818ETI+ is rated for the extended tempera-
ture range (-40°C to +85°C) and the MAX16818ATI+ is
rated for the automotive temperature range (-40°C to
+125°C). This LED driver controller is available in a
lead-free, 0.8mm high, 5mm x 5mm 28-pin TQFN pack-
age with exposed paddle.
Applications
Front Projectors/Rear Projection TVs
Portable and Pocket Projectors
Automotive, Bus/Truck Exterior Lighting
LCD TVs and Display Backlight
Automotive Emergency Lighting and Signage
Features
♦High-Current LED Driver Controller IC, Up to 30A
Output Current
♦Average-Current-Mode Control
♦True-Differential Remote-Sense Input
♦4.75V to 5.5V or 7V to 28V Input Voltage Range
♦Programmable Switching Frequency or External
Synchronization from 125kHz to 1.5MHz
♦Clock Output for 180° Out-of-Phase Operation
♦Integrated 4A Gate Drivers
♦Output Overvoltage and Hiccup Mode
Overcurrent Protection
♦Thermal Shutdown
♦Thermally Enhanced 28-Pin Thin QFN Package
♦-40°C to +125°C Operating Temperature Range
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
________________________________________________________________
Maxim Integrated Products
1
19-0666; Rev 2; 3/09
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
EVALUATION KIT
AVAILABLE
Pin Configuration appears at end of data sheet.
Ordering Information
+
Denotes a lead(Pb)-free/RoHS-compliant package.
*
EP = Exposed pad.
PART TEMP RANGE PIN-PACKAGE
MAX16818ATI+ -40°C to +125°C 28 TQFN-EP*
MAX16818ETI+ -40°C to +85°C 28 TQFN-EP*
Q1
HIGH-FREQUENCY
PULSE TRAIN
C2
Q3
L1
R1
V
LED
C1
7V TO 28V
CSP
DL
DH
NOTE: MAXIM TOPOLOGY
PGND
EN
IN
ILIM
OVI
CLP
Q2
MAX16818
Simplified Diagram
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VCC = 5V, VDD = VCC, TA= TJ= TMIN to TMAX, unless otherwise noted. Typical specifications are at TA= +25°C.) (Note 1)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
SYSTEM SPECIFICATIONS
728
Input Voltage Range VIN Short IN and VCC together for 5V input
operation 4.75 5.50 V
Quiescent Supply Current IQEN = VCC or SGND, not switching 2.7 5.5 mA
LED CURRENT REGULATOR
No load, VIN = 4.75V to 5.5V, fSW = 500kHz 0.594 0.6 0.606
SENSE+ to SENSE- Accuracy
(Note 2) No load, VIN = 7V to 28V, fSW = 500kHz 0.594 0.6 0.606 V
Soft-Start Time tSS 1024 Clock
Cycles
STARTUP/INTERNAL REGULATOR
VCC Undervoltage Lockout UVLO VCC rising 4.1 4.3 4.5 V
VCC Undervoltage Hysteresis 200 mV
VCC Output Voltage VIN = 7V to 28V, ISOURCE = 0 to 60mA 4.85 5.1 5.30 V
MOSFET DRIVERS
Output Driver Impedance RON Low or high output, ISOURCE/SINK = 20mA 1.1 3.0 Ω
Output Driver Source/Sink Current IDH,IDL 4A
Nonoverlap Time tNO CDH/DL = 5nF 35 ns
OSCILLATOR
Switching Frequency Range 125 1500 kHz
Switching Frequency RT = 500kΩ121 125 129
Switching Frequency RT = 120kΩ495 521 547
Switching Frequency
fSW
RT = 39.9kΩ1515 1620 1725
kHz
120kΩ ≤ RT ≤ 500kΩ-5 +5
Switching Frequency Accuracy 40kΩ ≤ RT ≤ 120kΩ-8 +8 %
IN to SGND.............................................................-0.3V to +30V
BST to SGND..........................................................-0.3V to +35V
BST to LX..................................................................-0.3V to +6V
DH to LX .......................................-0.3V to [(VBST - VLX_) + 0.3V]
DL to PGND................................................-0.3V to (VDD + 0.3V)
VCC to SGND............................................................-0.3V to +6V
VCC, VDD to PGND ...................................................-0.3V to +6V
SGND to PGND .....................................................-0.3V to +0.3V
All Other Pins to SGND...............................-0.3V to (VCC + 0.3V)
Continuous Power Dissipation (TA= +70°C)
28-Pin TQFN (derate 34.5mW/°C above +70°C) .......2758mW
Operating Temperature Range
MAX16818ATI+..............................................-40°C to +125°C
MAX16818ETI+................................................-40°C to +85°C
Maximum Junction Temperature .....................................+150°C
Storage Temperature Range .............................-60°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(VCC = 5V, VDD = VCC, TA= TJ= TMIN to TMAX, unless otherwise noted. Typical specifications are at TA= +25°C.) (Note 1)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
CLKOUT Phase Shift φ_CLKOUT With respect to DH, fSW = 125kHz 180 D eg r ees
CLKOUT Output Low Level VCLKOUTL ISINK = 2mA 0.4 V
CLKOUT Output High Level VCLKOUTH ISOURCE = 2mA 4.5 V
SYNC Input-High Pulse Width tSYNC 200 ns
SYNC Input Clock High Threshold VSYNCH 2.0 V
SYNC Input Clock Low Threshold VSYNCL 0.4 V
SYNC Pullup Current ISYNC_OUT VRT/SYNC = 0V 250 750 µA
SYNC Power-Off Level VSYNC_OFF 0.4 V
INDUCTOR CURRENT LIMIT
Average Current-Limit Threshold VCL CSP to CSN 24.0 26.9 28.2 mV
Reverse Current-Limit Threshold VCLR CSP to CSN -3.2 -2.3 -0.1 mV
Cycle-by-Cycle Current Limit CSP to CSN 60 mV
Cycle-by-Cycle Overload VCSP to VCSN = 75mV 260 ns
Hiccup Divider Ratio LIM to VCM, no switching 0.547 0.558 0.565 V/V
Hiccup Reset Delay 200 ms
LIM Input Impedance LIM to SGND 55.9 kΩ
CURRENT-SENSE AMPLIFIER
CSP or CSN Input Resistance RCS 4kΩ
Common-Mode Range VCMR
(
CS
)
VIN = 7V to 28V 0 5.5 V
Input Offset Voltage VOS
(
CS
)
0.1 mV
Amplifier Gain AV(CS) 34.5 V/V
3dB Bandwidth f3dB 4 MHz
CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance gm550 µS
Open-Loop Gain AVOL
(
CE
)
No load 50 dB
DIFFERENTIAL VOLTAGE AMPLIFIER FOR LED CURRENT (DIFF)
Common-Mode Voltage Range VCMR
(
DIFF
)
0 +1.0 V
DIFF Output Voltage VCM VSENSE+ = VSENSE- = 0V 0.6 V
Input Offset Voltage VOS
(
DIFF
)
-1 +1 mV
Amplifier Gain AV
(
DIFF
)
0.994 1 1.006 V/V
3dB Bandwidth f3dB CDIFF = 20pF 3 MHz
Minimum Output-Current Drive IOUT
(
DIFF
)
4mA
SENSE+ to SENSE- Input RVS VSENSE- = 0V 50 100 kΩ
V_IOUT AMPLIFIER
Gain-Bandwidth Product VV_IOUT = 2.0V 4 MHz
3dB Bandwidth VV_IOUT = 2.0V 1 MHz
Output Sink Current 30 µA
Output Source Current 90 µA
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
4 _______________________________________________________________________________________
Note 1: Specifications at TA= +25°C are 100% tested. Specifications over the temperature range are guaranteed by design.
Note 2: Does not include an error due to finite error amplifier gain. See the
Voltage-Error Amplifier (EAOUT)
section.
ELECTRICAL CHARACTERISTICS (continued)
(VCC = 5V, VDD = VCC, TA= TJ= TMIN to TMAX, unless otherwise noted. Typical specifications are at TA= +25°C.) (Note 1)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Maximum Load Capacitance 50 pF
V_IOUT Output to IOUT Transfer
Function RSENSE = 1mΩ, 100mV ≤ V_IOUT ≤ 5.5V 132.3 135 137.7 mV/A
Offset Voltage 1mV
VOLTAGE-ERROR AMPLIFIER (EAOUT)
Open-Loop Gain AVOLEA 70 dB
Unity-Gain Bandwidth fGBW 3 MHz
EAN Input Bias Current IB(EA) VEAN = 2.0V -0.2 +0.03 +0.2 µA
Error Amplifier Output Clamping
Voltage VCLAMP
(
EA
)
With respect to VCM 883 930 976 mV
POWER-GOOD AND OVERVOLTAGE PROTECTION
PGOOD Trip Level VUV PGOOD goes low when VOUT is below this
threshold 87.5 90 92.5 %VOUT
PGOOD Output Low Level VPGLO ISINK = 4mA 0.4 V
PGOOD Output Leakage Current IPG PGOOD = VCC 1µA
OVI Trip Threshold OVPTH With respect to SGND 1.244 1.276 1.308 V
OVI Input Bias Current IOVI 0.2 µA
ENABLE INPUT
EN Input High Voltage VEN EN rising 2.437 2.5 2.562 V
EN Input Hysteresis 0.28 V
EN Pullup Current IEN 13.5 15 16.5 µA
THERMAL SHUTDOWN
Thermal Shutdown Temperature rising 150 °C
Thermal Shutdown Hysteresis 30 °C
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
_______________________________________________________________________________________ 5
SUPPLY CURRENT (IQ) vs. FREQUENCY
MAX16818 toc01
FREQUENCY (kHz)
SUPPLY CURRENT (mA)
13001100900700500300
10
20
30
40
50
60
0
100 1500
EXTERNAL CLOCK
NO DRIVER LOAD
VIN = 12V
VIN = 24V
VIN = 5V
SUPPLY CURRENT vs. TEMPERATURE
MAX16818 toc02
TEMPERATURE (°C)
SUPPLY CURRENT (mA)
603510-15
62
64
66
68
70
60
-40 85
VIN = 12V
fSW = 250kHz
CDL/CDH = 22nF
CURRENT-SENSE THRESHOLD
vs. OUTPUT VOLTAGE
MAX16818 toc03
VOUT (V)
(VCSP - VCSN) (mV)
4321
26.5
27.0
27.5
28.0
28.5
29.0
26.0
05
VIN = 12V
fSW = 250kHz
HICCUP CURRENT LIMIT vs. REXT
MAX16818 toc04
REXT (MΩ)
CURRENT LIMIT (A)
161284
23.5
24.0
24.5
25.0
25.5
26.0
23.0
020
VIN = 12V
fSW = 250kHz
R1 = 1mΩ
VOUT = 1.5V
VCC LOAD REGULATION
vs. INPUT VOLTAGE
MAX16818 toc05
VCC LOAD CURRENT (mA)
VCC (V)
125100755025
4.85
4.95
5.05
5.15
5.25
4.75
0 150
VIN = 12V
VIN = 5V
VIN = 24V
DRIVER RISE TIME
vs. DRIVER LOAD CAPACITANCE
MAX16818 toc06
CAPACITANCE (nF)
tR (ns)
2116116
20
40
60
80
100
0
1
VIN = 12V
fSW = 250kHz
DH
DL
DRIVER FALL TIME
vs. DRIVER LOAD CAPACITANCE
MAX16818 toc07
CAPACITANCE (nF)
tF (ns)
2116116
20
40
60
80
100
0
1
VIN = 12V
fSW = 250kHz
DH
DL
HIGH-SIDE DRIVER (DH) SINK
AND SOURCE CURRENT
MAX16818 toc08
2A/div
100ns/div
CLOAD = 22nF
VIN = 12V
LOW-SIDE DRIVER (DL) SINK
AND SOURCE CURRENT
MAX16818 toc09
3A/div
100ns/div
CLOAD = 22nF
VIN = 12V
Typical Operating Characteristics
(TA= +25°C, using Figure 5
,
unless otherwise noted.)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
6 _______________________________________________________________________________________
HIGH-SIDE DRIVER (DH) RISE TIME
MAX16818 toc10
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
HIGH-SIDE DRIVER (DH) FALL TIME
MAX16818 toc11
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
LOW-SIDE DRIVER (DL) RISE TIME
MAX16818 toc12
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
LOW-SIDE DRIVER (DL) FALL TIME
MAX16818 toc13
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
FREQUENCY vs. RT
MAX16818 toc14
R
T
(
kΩ
)
fSW (kHz)
470
430
390
350
310
270
230
190
150
110
70
1000
10,000
100
30 510
VIN = 12V
FREQUENCY vs. TEMPERATURE
MAX16818 toc15
TEMPERATURE (°C)
fSW (kHz)
603510-15
242
244
246
248
250
252
254
256
258
260
240
-40 85
VIN = 12V
SYNC, CLKOUT, AND LX WAVEFORM
MAX16818 toc16
1µs/div
CLKOUT
5V/div
VIN = 12V
fSW = 250kHz
LX
10V/div
SYNC
5V/div
Typical Operating Characteristics (continued)
(TA= +25°C, using Figure 5
,
unless otherwise noted.)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
_______________________________________________________________________________________ 7
Pin Description
PIN NAME FUNCTION
1 PGND Power-Supply Ground
2, 7 N.C. No Connection. Not internally connected.
3 DL Low-Side Gate Driver Output
4 BST Boost Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver
supply. Connect a ceramic capacitor between BST and LX.
5 LX Source connection for the high-side MOSFET.
6 DH High-Side Gate Driver Output. Drives the gate of the high-side MOSFET.
8, 22, 25 SGND Signal Ground. Ground connection for the internal control circuitry. Connect SGND and PGND
together at one point near the IC.
9 CLKOUT Oscillator Output. Rising edge of CLKOUT is phase-shifted from the rising edge of DH by 180°.
10 PGOOD Power-Good Output
11 EN
Output Enable. Drive high or leave unconnected for normal operation. Drive low to shut down the
power drivers. EN has an internal 15µA pullup current. Connect a capacitor from EN to SGND to
program the hiccup-mode duty cycle.
12 RT/SYNC
Switching Frequency Programming and Chip-Enable Input. Connect a resistor from RT/SYNC to
SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency
with external clock.
13 V_IOUT Voltage Source Output Proportional to the Inductor Current. The voltage at V_IOUT = 135 x ILED x RS.
14 LIM Current-Limit Setting Input. Connect a resistor from LIM to SGND to set the hiccup current-limit
threshold. Connect a capacitor from LIM to SGND to ignore short output overcurrent pulses.
15 OVI
Overvoltage Protection. Connect OVI to DIFF. When OVI exceeds 12.7% above the programmed
output voltage, DH is latched low and DL is latched high. Toggle EN or recycle the input power to
reset the latch.
16 CLP Current-Error Amplifier Output. Compensate the current loop by connecting an RC network to ground.
17 EAOUT Voltage-Error Amplifier Output. Connect to the external compensation network.
18 EAN Voltage-Error Amplifier Inverting Input
19 DIFF Differential Remote-Sense Amplifier Output. DIFF is the output of a precision unity-gain amplifier
whose inputs are SENSE+ and SENSE-.
20 CSN Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
8 _______________________________________________________________________________________
Pin Description (continued)
PIN NAME FUNCTION
21 CSP Current-Sense Differential Amplifier Positive Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
23 SENSE- Differential LED Current-Sensing Negative Input. SENSE- is used to sense the LED current. Connect
SENSE- to the negative side of the LED current-sense resistor.
24 SENSE+ Differential LED Current-Sensing Positive Input. SENSE+ is used to sense the LED current. Connect
SENSE+ to the positive side of the LED current-sense resistor.
26 IN Supply Voltage Connection. Connect IN to VCC for a +5V system.
27 VCC Internal +5V Regulator Output. VCC is derived from the IN voltage. Bypass VCC to SGND with 4.7µF
and 0.1µF ceramic capacitors.
28 VDD
Supply Voltage for Low-Side and High-Side Drivers. Connect a parallel combination of 0.1µF and 1µF
ceramic capacitors to PGND and a 1Ω resistor to VCC to filter out the high peak currents of the driver
from internal circuitry.
—EP
Exposed Paddle. Connect the exposed paddle to a copper pad (SGND) to improve power
dissipation.
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
_______________________________________________________________________________________ 9
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
L1
VLED
C1 LED
STRING
R8
VIN
C2
D1
VIN
7V TO 28V
C10
C9
C8
C7
R12
R7
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C3
C6 C5 C4
R1
R2
R3
Typical Application Circuits
Figure 1. Typical Application Circuit for a Boost LED Driver (Nonsynchronous)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
10 ______________________________________________________________________________________
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
VLED
C1
LED
STRING
1 TO 6
LEDS
R8
VIN
C2
L1
D1
VIN
7V TO 28V
C10
C9
C8
C7
R12
R7
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C3
C6 C5 C4
R1
R2
R3
VCC
VCC
RS+
RS- OUT
MAX4073T
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
Typical Application Circuits (continued)
Figure 2. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (Input: 7V to 28V, Output: 1 to 6 LEDs in Series)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 11
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
L1
L2
VLED
C2 LED
STRING
R8
VIN
C3
D1
VIN
7V TO 28V
C11
C10
C9
C8
R12
R7
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C4
C7 C6 C5
R1
R2
R3
C1
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
Typical Application Circuits (continued)
Figure 3. Typical Application Circuit for a SEPIC LED Driver
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
12 ______________________________________________________________________________________
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q2
Q3
Q1
L1
VLED
D1
C1
LED
STRING
R8
VIN
C2
C4
VIN
7V TO 18V
C11
C10
C9
C8
R12
R7
D2
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C3
C7 C6 C5
R1
R2
R3
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
Typical Application Circuits (continued)
Figure 4. Application Circuit for a Ground-Referred Buck-Boost LED Driver
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 13
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
L1
C1
LED
STRING
R6
VIN
C2
C4
VIN
7V TO 28V
C11
C10
C9
C8
R10
R5
D1
R3
R4
C3
VCC
ON/OFF
R9
R8
PGND
N.C.
N.C.
DL
BST
LX
DH
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
C7 C6 C5
R1
R2
Typical Application Circuits (continued)
Figure 5. Application Circuit for a Buck LED Driver
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
14 ______________________________________________________________________________________
Functional Diagram
2 x fS (V/s)
RAMP
RT/SYNC
CSP
CSN
SGND
SENSE-
SENSE+
CLP
LIM
IN
EN
VDD
BST
DH
LX
DL
PGND
PGOOD
AV = 34.5
AV = 4
100kΩ
126.7kΩ
PWM
COMPARATOR
0.5V x VCC
TO INTERNAL
CIRCUITS HICCUP MODE
CURRENT LIMIT
S
R
Q
Q
V_IOUT
gm = 500µS
DIFF
CLKOUT
CLK
CPWM
CEA
VCLAMP
HIGH
VCLAMP
LOW
CA
VCC
0.1 x VREF
N
+0.6V
VREF = 0.6V
VCM (0.6V)
OVP COMP
0.12 x VREF
LATCH
RAMP
GENERATOR
SOFT-
START
OSCILLATOR
CLEAR ON UVLO RESET OR
ENABLE LOW
OVP LATCH
DIFF
AMP
EAN
EAOUT
OVI
VEA
ERROR AMP
0.5 x VCLAMP
VCM
VCM
IS
VCC
Ct
RT
S
R
Q
Q
UVLO
POR
TEMP SENSOR
5V
LDO
REGULATOR
MAX16818
Figure 6. MAX16818 Functional Diagram
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 15
Detailed Description
The MAX16818 is a high-performance average-current-
mode PWM controller for high-power, high-brightness
LEDs (HBLEDs). Average current-mode control is the
ideal method for driving HBLEDs. This technique offers
inherently stable operation, reduces component derat-
ing and size by accurately controlling the inductor cur-
rent. The device achieves high efficiency at high
current (up to 30A) with a minimum number of external
components. The high- and low-side drivers source
and sink up to 4A for lower switching losses while dri-
ving high-gate-charge MOSFETs. The MAX16818’s
CLKOUT output is 180° out-of-phase with respect to the
high-side driver. CLKOUT drives a second MAX16818
LED driver out of phase, reducing the input-capacitor
ripple current.
The MAX16818 consists of an inner average current loop
representing inductor current and an outer voltage loop
voltage-error amplifier (VEA) that directly controls LED
current. The combined action of the two loops results in
a tightly regulated LED current. The inductor current is
sensed across a current-sense resistor. The differential
amplifier senses LED current through a sense resistor in
series with the LEDs and the resulting sensed voltage is
compared against an internal 0.6V reference at the error-
amplifier input. The MAX16818 will adjust the LED cur-
rent to within 1% accuracy to maintain emitted spectrum
of the light in HBLEDs.
IN, VCC, and VDD
The MAX16818 accepts either a 4.75V to 5.5V or 7V to
28V input voltage range. All internal control circuitry
operates from an internally regulated nominal voltage of
5V (VCC). For input voltages of 7V or greater, the inter-
nal VCC regulator steps the voltage down to 5V. The
VCC output voltage is a regulated 5V output capable of
sourcing up to 60mA. Bypass the VCC to SGND with
4.7µF and 0.1µF low-ESR ceramic capacitors for high-
frequency noise rejection and stable operation.
The MAX16818 uses VDD to power the low-side and
high-side drivers. Isolate VDD from VCC with a 1Ωresis-
tor and put a 1µF capacitor in parallel with a 0.1µF
capacitor to ground to prevent high-current noise spikes
created by the driver from disrupting internal circuitry.
The TQFN is a thermally enhanced package and can
dissipate up to 2.7W. The high-power packages allow
the high-frequency, high-current converter to operate
from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
total VCC regulator output current (ICC). ICC includes qui-
escent current (IQ) and gate-drive current (IDD):
PD= VIN x ICC
ICC = IQ+ [fSW x (QG1 + QG2)]
where QG1 and QG2 are the total gate charge of the
low-side and high-side external MOSFETs at VGATE =
5V, IQis 3.5mA (typ), and fSW is the switching frequen-
cy of the converter.
Undervoltage Lockout (UVLO)
The MAX16818 includes an undervoltage lockout with
hysteresis and a power-on-reset circuit for converter
turn-on. The UVLO rising threshold is internally set at
4.35V with a 200mV hysteresis. Hysteresis at UVLO
eliminates chattering during startup.
Most of the internal circuitry, including the oscillator,
turns on when the input voltage reaches 4V. The
MAX16818 draws up to 3.5mA of current before the
input voltage reaches the UVLO threshold.
Soft-Start
The MAX16818 has an internal digital soft-start for a
monotonic, glitch-free rise of the output current. Soft-
start is achieved by the controlled rise of the error
amplifier dominant input in steps using a 5-bit counter
and a 5-bit DAC. The soft-start DAC generates a linear
ramp from 0 to 0.7V. This voltage is applied to the error
amplifier at a third (noninverting) input. As long as the
soft-start voltage is lower than the reference voltage,
the system converges to that lower reference value.
Once the soft-start DAC output reaches 0.6V, the refer-
ence takes over and the DAC output continues to climb
to 0.7V, assuring that it does not interfere with the refer-
ence voltage.
Internal Oscillator
The internal oscillator generates a clock with the fre-
quency proportional to the inverse of RT. The oscillator
frequency is adjustable from 125kHz to 1.5MHz with
better than 8% accuracy using a single resistor con-
nected from RT/SYNC to SGND. The frequency accura-
cy avoids the over-design, size, and cost of passive
filter components like inductors and capacitors. Use
the following equation to calculate the oscillator fre-
quency:
For 120kΩ≤RT≤500kΩ:
For 40kΩ≤RT≤120kΩ:
Rx
f
TSW
.
=640 10
10
Rx
f
TSW
.
=625 10
10
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
16 ______________________________________________________________________________________
The oscillator also generates a 2VP-P voltage-ramp sig-
nal for the PWM comparator and a 180° out-of-phase
clock signal for CLKOUT to drive a second LED regula-
tor out-of-phase.
Synchronization
The MAX16818 can be easily synchronized by connect-
ing an external clock to RT/SYNC. If an external clock is
present, then the internal oscillator is disabled and the
external clock is used to run the device. If the external
clock is removed, the absence of clock for 32µs is
detected and the circuit starts switching from the inter-
nal oscillator. Pulling RT/SYNC to ground for at least
50µs disables the converter. Use an open-collector
transistor to synchronize the MAX16818 with the exter-
nal system clock.
Control Loop
The MAX16818 uses an average-current-mode control
scheme to regulate the output current (Figure 7). The
main control loop consists of an inner current loop for
controlling the inductor current and an outer current
loop for regulating the LED current. The inner current
loop absorbs the inductor pole reducing the order of the
outer current loop to that of a single-pole system. The
current loop consists of a current-sense resistor (RS), a
current-sense amplifier (CA), a current-error amplifier
(CEA), an oscillator providing the carrier ramp, and a
PWM comparator (CPWM) (Figure 7). The precision CA
amplifies the sense voltage across RSby a factor of
34.5. The inverting input to the CEA senses the CA out-
put. The CEA output is the difference between the volt-
age-error amplifier output (EAOUT) and the amplified
voltage from the CA. The RC compensation network
connected to CLP provides external frequency compen-
sation for the CEA. The start of every clock cycle
enables the high-side drivers and initiates a PWM on-
cycle. Comparator CPWM compares the output voltage
from the CEA with a 0V to 2V ramp from the oscillator.
The PWM on-cycle terminates when the ramp voltage
exceeds the error voltage. Compensation for the outer
LED current loop varies based upon the topology.
The MAX16818 outer LED current control loop consists
of the differential amplifier (DIFF AMP), reference volt-
age, and VEA. The unity-gain differential amplifier pro-
vides true differential remote sensing of the voltage
across the LED current set resistor, RLS. The differential
amplifier output connects to the inverting input (EAN) of
the VEA. The DIFF AMP is bypassed and the inverting
input is available to the pin for direct feedback. The
noninverting input of the VEA is internally connected to
an internal precision reference voltage, set to 0.6V. The
VEA controls the inner current loop (Figure 6). A feed-
back network compensates the outer loop using the
EAOUT and EAIN pins.
DRIVE
Z COMP
CPWM
CEA
CA
CLP
VREF + VCM = 1.2V
CSPCSN
CCFF
VIN
COUT
RLS
RS
LED
STRING
IL
CCF RCF
VEA
600mV
SENSE+
EAOUT
SENSE-
EAN
DIFF
MAX16818
DIFF
AMP
Figure 7. MAX16818 Control Loop
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 17
Inductor Current-Sense Amplifier
The differential current-sense amplifier (CA) provides a
DC gain of 34.5. The maximum input offset voltage of
the current-sense amplifier is 1mV and the common-
mode voltage range is 0 to 5.5V (IN = 7V to 28V). The
current-sense amplifier senses the voltage across a
current-sense resistor. The maximum common-mode
voltage is 3.6V when VIN = 5V.
Inductor Peak-Current Comparator
The peak-current comparator provides a path for fast
cycle-by-cycle current limit during extreme fault condi-
tions, such as an inductor malfunction (Figure 8). Note
the average current-limit threshold of 26.9mV still limits
the output current during short-circuit conditions. To
prevent inductor saturation, select an inductor with a
saturation current specification greater than the average
current limit. Proper inductor selection ensures that only
the extreme conditions trip the peak-current comparator,
such as an inductor with a shorted turn. The 60mV
threshold for triggering the peak-current limit is twice the
full-scale average current-limit voltage threshold. The
peak-current comparator has only a 260ns delay.
Current-Error Amplifier
(For Inductor Currents)
The MAX16818 has a transconductance current-error
amplifier (CEA) with a typical gmof 550µS and 320µA
output sink- and source-current capability. The current-
error amplifier output CLP serves as the inverting input
to the PWM comparator. CLP is externally accessible to
provide frequency compensation for the inner current
loops (Figure 7). Compensate (CEA) so the inductor
current negative slope, which becomes the positive
slope to the inverting input of the PWM comparator, is
less than the slope of the internally generated voltage
ramp (see the
Compensation
section).
PWM Comparator and R-S Flip-Flop
The PWM comparator (CPWM) sets the duty cycle for
each cycle by comparing the output of the current-error
amplifier to a 2VP-P ramp. At the start of each clock
cycle, an R-S flip-flop resets and the high-side driver
(DH) goes high. The comparator sets the flip-flop as
soon as the ramp voltage exceeds the CLP voltage,
thus terminating the on-cycle (Figure 8).
Figure 8. MAX16818 Phase Circuit
2 x fS (V/s)
RAMP
CLK
CSP
CSN
EAN
EAOUT
SHDN
CLP
VDD
BST
DH
LX
DL
PGND
AV = 34.5
PEAK-CURRENT
COMPARATOR
60mV
S
R
Q
Q
gm = 550µS
CPWM
SET
CLR
CEA
CA
MAX16818
VEA
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
18 ______________________________________________________________________________________
Differential Amplifier
The DIFF AMP facilitates remote sensing at the load
(Figure 7). It provides true differential LED current
(through the RLS sense resistor) sensing while rejecting
the common-mode voltage errors due to high-current
ground paths. The VEA provides the difference
between the differential amplifier output (DIFF) and the
desired LED current-sense voltage. The differential
amplifier has a bandwidth of 3MHz. The difference
between SENSE+ and SENSE- is regulated to 0.6V.
Connect SENSE+ to the positive side of the LED current-
sense resistor and SENSE- to the negative side of the
LED current-sense resistor (which is often PGND).
MOSFET Gate Drivers (DH, DL)
The high-side (DH) and low-side (DL) drivers drive the
gates of external n-channel MOSFETs (Figures 1–5).
The drivers’ 4A peak sink- and source-current capabili-
ty provides ample drive for the fast rise and fall times of
the switching MOSFETs. Faster rise and fall times result
in reduced cross-conduction losses. Due to physical
realities, extremely low gate charges and RDS(ON)
resistance of MOSFETs are typically exclusive of each
other. MOSFETs with very low RDS(ON) will have a high-
er gate charge and vice versa. Choosing the high-side
MOSFET (Q1) becomes a trade-off between these two
attributes. Applications where the input voltage is much
higher than the output voltage result in a low duty cycle
where conduction losses are less important than
switching losses. In this case, choose a MOSFET with
very low gate charge and a moderate RDS(ON).
Conversely, for applications where the output voltage is
near the input voltage resulting in duty cycles much
greater than 50%, the RDS(ON) losses become at least
equal, or even more important than the switching losses.
In this case, choose a MOSFET with very low RDS(ON)
and moderate gate charge. Finally, for the applications
where the duty cycle is near 50%, the two loss compo-
nents are nearly equal, and a balanced MOSFET with
moderate gate charge and RDS(ON) work best.
In a buck topology, the low-side MOSFET (Q2) typically
operates in a zero voltage switching mode, thus it does
not have switching losses. Choose a MOSFET with very
low RDS(ON) and moderate gate charge.
Size both the high-side and low-side MOSFETs to han-
dle the peak and RMS currents during overload condi-
tions. The driver block also includes a logic circuit that
provides an adaptive nonoverlap time to prevent shoot-
through currents during transition. The typical nonover-
lap time between the high-side and low-side MOSFETs
is 35ns.
BST
The MAX16818 uses VDD to power the low- and high-
side MOSFET drivers. The high-side driver derives its
power through a bootstrap capacitor and VDD supplies
power internally to the low-side driver. Connect a
0.47µF low-ESR ceramic capacitor between BST and
LX. Connect a Schottky rectifier from BST to VDD. Keep
the loop formed by the boost capacitor, rectifier, and IC
small on the PCB.
Protection
The MAX16818 includes output overvoltage protection
(OVP). During fault conditions when the load goes to
high impedance (opens), the controller attempts to
maintain LED current. The OVP protection disables the
MAX16818 whenever the voltage exceeds the thresh-
old, protecting the external circuits from undesirable
voltages.
Current Limit
The VEA output is clamped to 930mV with respect to
the common-mode voltage (VCM). Average-current-
mode control has the ability to limit the average current
sourced by the converter during a fault condition. When
a fault condition occurs, the VEA output clamps to
930mV with respect to the common-mode voltage
(0.6V) to limit the maximum current sourced by the con-
verter to ILIMIT = 26.9mV / RS. The hiccup current limit
overrides the average current limit. The MAX16818
includes hiccup current-limit protection to reduce the
power dissipation during a fault condition. The hiccup
current-limit circuit derives inductor current information
from the output of the current amplifier. This signal is
compared against one half of VCLAMP(EA). With no
resistor connected from the LIM pin to ground, the hic-
cup current limit is set at 90% of the full-load average
current limit. Use REXT to increase the hiccup current
limit from 90% to 100% of the full load average limit.
The hiccup current limit can be disabled by connecting
LIM to SGND. In this case, the circuit follows the aver-
age current-limit action during overload conditions.
Overvoltage Protection
The OVP comparator compares the OVI input to the
overvoltage threshold. A detected overvoltage event
latches the comparator output forcing the power stage
into the OVP state. In the OVP state, the high-side
MOSFET turns off and the low-side MOSFET latches on.
Connect OVI to the center tap of a resistor-divider from
VLED to SGND. In this case, the center tap is compared
against 1.276V. Add an RC delay to reduce the sensitivity
of the overvoltage circuit and avoid nuisance tripping of
the converter. Disable the overvoltage function by con-
necting OVI to SGND.
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 19
Applications Information
Application Circuit Descriptions
This section provides some detail regarding the appli-
cation circuits in the
Simplified Diagram
and Figures
1–5. The discussion includes some description of the
topology as well as basic attributes.
High-Frequency LED Current Pulser
The
Simplified Diagram
shows the MAX16818 providing
high-frequency, high-current pulses to the LEDs. The
basic topology must be a buck, since the inductor
always connects to the load in that configuration (in all
other topologies, the inductor disconnects from the
load at one time or another). The design minimizes the
current ripple by oversizing the inductor, which allows
for a very small (0.01µF) output capacitor. When MOS-
FET Q3 turns on, it diverts the current around the LEDs
at a very fast rate. Q3 also discharges the output
capacitor, but since the capacitor is so small, it does
not stress the MOSFET. Resistor R1 senses the LED/Q3
current and there is no reaction to the short that Q3
places across the LEDs. This design is superior in that
it does not attempt to actually change the inductor cur-
rent at high frequencies and yet the current in the LEDs
varies from zero to full in very small periods of time. The
efficiency of this technique is very high. Q3 must be
able to dissipate the LED current applied to its RDS(ON)
at some maximum duty cycle. If the circuit needs to
control extremely high currents, use paralleled
MOSFETs. PGOOD is low during LED pulsed-current
operation.
Boost LED Driver
In Figure 1, the external components configure the
MAX16818 as a boost converter. The circuit applies the
input voltage to the inductor during the on-time, and
then during the off-time the inductor, which is in series
with the input capacitor, charges the output capacitor.
Because of the series connection between the input
voltage and the inductor, the output voltage can never
go lower than the input voltage. The design is nonsyn-
chronous, and since the current-sense resistor con-
nects to ground, the power supply can go to any output
voltage (above the input) as long as the components are
rated appropriately. R2 again provides the sense voltage
the MAX16818 uses to regulate the LED current.
Input-Referenced LED Driver
The circuit in Figure 2 shows a step-up/step-down reg-
ulator. It is similar to the boost converter in Figure 1 in
that the inductor is connected to the input and the
MOSFET is essentially connected to ground. However,
rather than going from the output to ground, the LEDs
span from the output to the input. This effectively
removes the boost-only restriction of the regulator in
Figure 1, allowing the voltage across the LEDs to be
greater than or less than the input voltage. LED current
sensing is not ground-referenced, so a high-side cur-
rent-sense amplifier is used to measure current.
SEPIC LED Driver
Figure 3 shows the MAX16818 configured as a SEPIC
LED driver. While buck topologies require the output to
be lesser than the input, and boost topologies require
the output to be greater than the input, a SEPIC topolo-
gy allows the output voltage to be greater than, equal
to, or less than the input. In a SEPIC topology, the volt-
age across C1 is the same as the input voltage, and L1
and L2 are the same inductance. Therefore, when Q1
conducts (on-time), both inductors ramp up current at
the same rate. The output capacitor supports the output
voltage during this time. During the off-time, L1 current
recharges C1 and combines with L2 to provide current
to recharge C2 and supply the load current. Since the
voltage waveform across L1 and L2 are exactly the
same, it is possible to wind both inductors on the same
core (a coupled inductor). Although voltages on L1 and
L2 are the same, RMS currents can be quite different
so the windings may have a different gauge wire.
Because of the dual inductors and segmented energy
transfer, the efficiency of a SEPIC converter is some-
what lower than standard bucks or boosts. As in the
boost driver, the current-sense resistor connects to
ground, allowing the output voltage of the LED driver to
exceed the rated maximum voltage of the MAX16818.
Ground-Referenced Buck/Boost LED Driver
Figure 4 depicts a buck/boost topology. During the on-
time with this circuit, the current flows from the input
capacitor, through Q1, L1, and Q3 and back to the
input capacitor. During the off-time, current flows up
through Q2, L1, D1, and to the output capacitor C1.
This topology resembles a boost in that the inductor sits
between the input and ground during the on-time.
However, during the off-time the inductor resides
between ground and the output capacitor (instead of
between the input and output capacitors in boost
topologies), so the output voltage can be any voltage
less than, equal to, or greater than the input voltage. As
compared to the SEPIC topology, the buck/boost does
not require two inductors or a series capacitor, but it
does require two additional MOSFETs.
Buck Driver with Synchronous Rectification
In Figure 5, the input voltage can go from 7V to 28V and,
because of the ground-based current-sense resistor, the
output voltage can be as high as the input. The synchro-
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
20 ______________________________________________________________________________________
nous MOSFET keeps the power dissipation to a minimum,
especially when the input voltage is large when com-
pared to the voltage on the LED string. It is important to
keep the current-sense resistor, R1, inside the LC loop,
so that ripple current is available. To regulate the LED
current, R2 creates a voltage that the differential amplifier
compares to 0.6V. If power dissipation is a problem in R2,
add a noninverting amplifier and reduce the value of the
sense resistor accordingly.
Inductor Selection
The switching frequencies, peak inductor current, and
allowable ripple at the output determine the value and
size of the inductor. Selecting higher switching frequen-
cies reduces the inductance requirement, but at the
cost of lower efficiency. The charge/discharge cycle of
the gate and drain capacitances in the switching
MOSFETs create switching losses. The situation wors-
ens at higher input voltages, since switching losses are
proportional to the square of the input voltage. The
MAX16818 can operate up to 1.5MHz, however for
VIN > +12V, use lower switching frequencies to limit the
switching losses.
The following discussion is for buck or continuous
boost-mode topologies. Discontinuous boost, buck-
boost, and SEPIC topologies are quite different in
regards to component selection.
Use the following equations to determine the minimum
inductance value:
Buck regulators:
Boost regulators:
where VLED is the total voltage across the LED string.
As a first approximation choose the ripple current, ∆IL,
equal to approximately 40% of the output current.
Higher ripple current allows for smaller inductors, but it
also increases the output capacitance for a given volt-
age ripple requirement. Conversely, lower ripple cur-
rent increases the inductance value, but allows the
output capacitor to reduce in size. This trade-off can be
altered once standard inductance and capacitance val-
ues are chosen. Choose inductors from the standard
surface-mount inductor series available from various
manufacturers.
For example, for a buck regulator and 2 LEDs in series,
calculate the minimum inductance at VIN(MAX) = 13.2V,
VLED = 7.8V, ∆IL= 400mA, and fSW = 330kHz:
Buck regulators:
For a boost regulator with four LEDs in series, calculate
the minimum inductance at VIN(MAX) = 13.2V, VLED =
15.6V, ∆IL=400mA, and fSW = 330kHz:
Boost regulators:
The average-current-mode control feature of the
MAX16818 limits the maximum peak inductor current
and prevents the inductor from saturating. Choose an
inductor with a saturating current greater than the
worst-case peak inductor current. Use the following
equation to determine the worst-case inductor current:
where RSis the inductor sense resistor and VCL =
0.0282V.
Switching MOSFETs
When choosing a MOSFET for voltage regulators, con-
sider the total gate charge, RDS(ON), power dissipation,
and package thermal impedance. The product of the
MOSFET gate charge and on-resistance is a figure of
merit, with a lower number signifying better perfor-
mance. Choose MOSFETs optimized for high-frequency
switching applications.
The average current from the MAX16818 gate-drive
output is proportional to the total capacitance it drives
at DH and DL. The power dissipated in the MAX16818
is proportional to the input voltage and the average
drive current. See the
IN, V
CC
, and V
DD
section to
determine the maximum total gate charge allowed from
the combined driver outputs. The gate-charge and
drain-capacitance (CV2) loss, the cross-conduction loss
in the upper MOSFET due to finite rise/fall times, and
the I2R loss due to RMS current in the MOSFET
RDS(ON) account for the total losses in the MOSFET.
ILPEAK CL
S
L
V
R
I
=+
∆
2
Lx
xkx H
MIN
(. .) .
. . .=−=µ
15 6 13 2 13 2
15 6 330 0 4 15 3
Lx
xkx H
MIN
(. .) .
. . .=−=µ
13 2 7 8 7 8
13 2 330 0 4 24 2
LVV xV
VxfxI
MIN LED INMAX INMAX
LED SW L
()
=−
∆
LVVxV
VxfxI
MIN INMAX LED LED
INMAX SW L
( )
=−
∆
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 21
Buck Regulator
Estimate the power loss (PDMOS_) caused by the high-side
and low-side MOSFETs using the following equations:
where QG, RDS(ON), tR, and tFare the upper-switching
MOSFET’s total gate charge, on-resistance at maximum
operating temperature, rise time, and fall time, respectively.
For the buck regulator, D = VLEDs / VIN, IVALLEY =
(IOUT - ∆IL/ 2) and IPK = (IOUT + ∆IL/ 2).
For example, from the typical specifications in the
Applications Information
section with VOUT = 7.8V, the
high-side and low-side MOSFET RMS currents are
0.77A and 0.63A, respectively, for a 1A buck regulator.
Ensure that the thermal impedance of the MOSFET
package keeps the junction temperature at least +25°C
below the absolute maximum rating. Use the following
equation to calculate the maximum junction tempera-
ture: TJ= (PDMOS x θJA) + TA, where θJA and TAare
the junction-to-ambient thermal impedance and ambi-
ent temperature, respectively.
To guarantee that there is no shoot-through from VIN to
PGND, the MAX16818 produces a nonoverlap time of
35ns. During this time, neither high- nor low-side MOS-
FET is conducting, and since the output inductor must
maintain current flow, the intrinsic body diode of the
low-side MOSFET becomes the conduction path. Since
this diode has a fairly large forward voltage, a Schottky
diode (in parallel to the low-side MOSFET) diverts current
flow from the MOSFET body diode because of its lower
forward voltage, which, in turn, increases efficiency.
Boost Regulator
Estimate the power loss (PDMOS_) caused by the MOS-
FET using the following equations:
For a boost regulator in continuous mode, D = VLEDs /
(VIN + VLEDs), IVALLEY = (IOUT - ∆IL/ 2) and IPK = (IOUT
+ ∆IL/ 2).
The voltage across the MOSFET:
VMOSFET = VLED + VF
where VFis the maximum forward voltage of the diode.
The output diode on a boost regulator must be rated to
handle the LED series voltage, VLED. It should also
have fast reverse-recovery characteristics and should
handle the average forward current that is equal to the
LED current.
Input Capacitors
For buck regulator designs, the discontinuous input
current waveform of the buck converter causes large
ripple currents in the input capacitor. The switching fre-
quency, peak inductor current, and the allowable peak-
to-peak voltage ripple reflected back to the source
dictate the capacitance requirement. Increasing switch-
ing frequency or paralleling out-of-phase converters
lowers the peak-to-average current ratio, yielding a
lower input capacitance requirement for the same LED
current. The input ripple is comprised of ∆VQ(caused
by the capacitor discharge) and ∆VESR (caused by the
ESR of the capacitor). Use low-ESR ceramic capacitors
with high-ripple-current capability at the input. Assume
the contributions from the ESR and capacitor discharge
are equal to 30% and 70%, respectively. Calculate the
input capacitance and ESR required for a specified ripple
using the following equation:
ESR V
II
IN ESR
OUT L
=
+
⎛
⎝
⎜⎞
⎠
⎟
∆
∆
2
PD Q x V x f
VxI xt txf RxI
IIIIxIx
D
FET G DD SW
IN OUT R F SW DS ON RMS HI
RMS HI VALLEY PK VALLEY PK
( )
( ) ( )
( )
()
=+
+
⎛
⎝
⎜⎞
⎠
⎟+
=++
−
−
2
3
2
22
( )
( )
()
()
RxI
IIIIxIx
D
DS ON RMS LO
RMS LO VALLEY PK VALLEY PK
−
−=++ −
2
22 1
3
PD Q x V x f
MOS LO G DD SW−=+ ( )
IIIIxIx
D
RMS HI VALLEY PK VALLEY PK−=++ ( )
22
3
VxI xt txf
RxI
IN OUT R F SW
DS ON RMS HI
( )
( )
()
+
⎛
⎝
⎜⎞
⎠
⎟
+−
2
2
PD Q x V x f
MOS HI G DD SW−=+ ( )
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
22 ______________________________________________________________________________________
Buck:
where IOUT is the output current of the converter. For
example, at VIN = 13.2V, VLED = 7.8V, IOUT = 1A, ∆IL=
0.4A, and fSW = 330kHz, the ESR and input capaci-
tance are calculated for the input peak-to-peak ripple of
100mV or less yielding an ESR and capacitance value
of 25mΩand 10µF.
For boost regulator designs, the input-capacitor current
waveform is dominated by the inductor, a triangle wave
a magnitude of ∆IL. For simplicity’s sake, the current
waveform can be approximated by a square wave with
a magnitude that is half that of the triangle wave.
Calculate the input capacitance and ESR required for a
specified ripple using the following equation:
Boost:
Duty cycle, D, for a boost regulator is equal to (VOUT -
VIN) / VOUT. As an example, at VIN = 13.2V, VLED =
15.6V, IOUT = 1A, ∆IL= 0.4A, and fSW = 330kHz, the
ESR and input capacitance are calculated for the input
peak-to-peak ripple of 100mV or less yielding an ESR
and capacitance value of 250mΩand 1µF, respectively.
Output Capacitor
For buck converters, the inductor always connects to
the load, so the inductance controls the ripple current.
The output capacitance shunts a fraction of this ripple
current and the LED string absorbs the rest. The
capacitor reactance (which includes the capacitance
and ESR) and the dynamic impedance of the LED
diode string form a conductance divider that splits the
ripple current between the LEDs and the capacitor. In
many cases, the capacitor is very large as compared to
the ESR, and this divider reduces to the ESR and the
LED resistance.
Boost converters place a harsher requirement on the
output capacitors as they must sustain the full load dur-
ing the on-time of the MOSFET and are replenished
during the off-time. The ripple current in this case is the
full load current, and the holdup time is equal to the
duty cycle times the switching period.
Current Limit
In addition to the average current limit, the MAX16818
also has hiccup current limit. The hiccup current limit is
set to 10% below the average current limit to ensure that
the circuit goes in hiccup mode during continuous out-
put short circuit. Connecting a resistor from LIM to
ground increases the hiccup current limit, while shorting
LIM to ground disables the hiccup current-limit circuit.
Average Current Limit
The average-current-mode control technique of the
MAX16818 accurately limits the maximum output current.
The MAX16818 senses the voltage across the sense
resistor and limit the peak inductor current (IL-PK)
accordingly. The on-cycle terminates when the current-
sense voltage reaches 25.5mV (min). Use the following
equation to calculate the maximum current-sense resis-
tor value:
where PDRis the dissipation in the series resistors.
Select a 5% lower value of RSto compensate for any
parasitics associated with the PCB. Also, select a non-
inductive resistor with the appropriate power rating.
Hiccup Current Limit
The hiccup current-limit value is always 10% lower than
the average current-limit threshold, when LIM is left
unconnected. Connect a resistor from LIM to SGND to
increase the hiccup current-limit value from 90% to
100% of the average current-limit value. The average
current-limit architecture accurately limits the average
output current to its current-limit threshold. If the hiccup
current limit is programmed to be equal or above the
average current-limit value, the output current does not
reach the point where the hiccup current limit can trig-
ger. Program the hiccup current limit at least 5% below
the average current limit to ensure that the hiccup cur-
rent-limit circuit triggers during overload. See the
Hiccup Current Limit vs. REXT graph in the
Typical
Operating Characteristics
.
RI
PD x
R
SOUT
RS
.
.
=
=−
0 0255
075 10 3
C
IxD
Vxf
IN
L
QSW
=
∆
∆
2
ESR V
I
IN ESR
L
=∆
∆
CIxDD
Vxf
IN OUT
QSW
()
=−1
∆
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
______________________________________________________________________________________ 23
Compensation
The main control loop consists of an inner current loop
(inductor current) and an outer LED current loop. The
MAX16818 uses an average current-mode control
scheme to regulate the LED current (Figure 7). The VEA
output provides the controlling voltage for the current
source. The inner current loop absorbs the inductor pole
reducing the order of the LED current loop to that of a
single-pole system. The major consideration when
designing the current control loop is making certain that
the inductor downslope (which becomes an upslope at
the output of the CEA) does not exceed the internal
ramp slope. This is a necessary condition to avoid sub-
harmonic oscillations similar to those in peak current
mode with insufficient slope compensation. This requires
that the resistance, RCF, at the output of the CEA be lim-
ited, based on the following equation (Figure 6):
Buck:
where VRAMP = 2V, gm= 550µS, and AV= 34.5.
Boost:
The crossover frequency of the inner current loop is
expressed as:
Buck:
When AV = 34.5, gm = 550µS, and VRAMP = 2V, this
becomes:
Boost:
which becomes:
For adequate phase margin, place the zero formed by
RCF and CCZ not more than 1/3 to 1/5 of the crossover
frequency. The pole formed by RCF and CCP may not
be required in most applications but can be added to
minimize noise at a frequency at or above the switching
frequency.
Power Dissipation
The TQFN is a thermally enhanced package and can dis-
sipate about 2.7W. The high-power package makes the
high-frequency, high-current LED driver possible to oper-
ate from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
total VCC regulator output current (ICC). ICC includes qui-
escent current (IQ) and gate drive current (IDD):
where QG1 and QG2 are the total gate charge of the low-
side and high-side external MOSFETs at VGATE = 5V, IQ
is estimated from the Supply Current (IQ) vs. Frequency
graph in the
Typical Operating Characteristics
, and fSW
is the switching frequency of the LED driver. For boost
drivers, only consider one gate charge, QG1.
Use the following equation to calculate the maximum
power dissipation (PDMAX) in the chip at a given ambi-
ent temperature (TA):
PDMAX = 34.5 x (150 - TA) mW.
PVxI
IIfxQQ
DINCC
CC Q SW G G
( )
=
=+ +
[]
12
fmS V R V R
L
C boost SLED CF
_
.
=
()
×× ×
×
9 488
2π
fAgRV R
VL
C boost VmSLEDCF
RAMP
_=××× ×
××2π
fmS V R V R
L
C buck SIN CF
_
.
=
()
×× ×
×
9 488
2π
fAgRVR
VL
C buck VmSINCF
RAMP
_=××××
××2π
RVfL
AgR V V
R
CF RAMP SW
VmS LEDIN
CF
≤××
××× −
()
≤1005 ××
×−
()
fL
RV V
SW
SLEDIN
RfL
RV
CF SW
SLED
≤× ×
×
105
RVfL
AgRV
CF RAMP SW
VmSLED
≤××
×××
PCB Layout Guidelines
Use the following guidelines to layout the switching volt-
age regulator:
1) Place the IN, VCC, and VDD bypass capacitors
close to the MAX16818.
2) Minimize the area and length of the high current
loops from the input capacitor, upper switching
MOSFET, inductor, and output capacitor back to
the input capacitor negative terminal.
3) Keep short the current loop formed by the lower
switching MOSFET, inductor, and output capacitor.
4) Place the Schottky diodes close to the lower
MOSFETs and on the same side of the PCB.
5) Keep the SGND and PGND isolated and connect
them at one single point.
6) Run the current-sense lines CSP and CSN very
close to each other to minimize the loop area.
Similarly, run the remote voltage-sense lines
SENSE+ and SENSE- close to each other. Do not
cross these critical signal lines through power cir-
cuitry. Sense the current right at the pads of the
current-sense resistors.
7) Avoid long traces between the VDD bypass capaci-
tors, the driver output of the MAX16818, the MOS-
FET gates, and PGND. Minimize the loop formed by
the VCC bypass capacitors, bootstrap diode, boot-
strap capacitor, the MAX16818, and the upper
MOSFET gate.
8) Distribute the power components evenly across the
board for proper heat dissipation.
9) Provide enough copper area at and around the
switching MOSFETs, inductor, and sense resistors
to aid in thermal dissipation.
10) Use wide copper traces (2oz) to keep trace induc-
tance and resistance low to maximize efficiency.
Wide traces also cool heat-generating components.
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
24 ______________________________________________________________________________________
Chip Information
TRANSISTOR COUNT: 5654
PROCESS: BiCMOS
28
27
26
25
24
23
22
8
9
10
11
12
13
14
15
16
17
18
19
20
21
7
6
5
432
1
MAX16818
TQFN
+
TOP VIEW
N.C.
PGND
DL
BST
LX
DH
N.C.
VDD
VCC
IN
SGND
SENSE+
SENSE-
SGND
CSP
CSN
DIFF
EAN
EAOUT
CLP
* EXPOSED PAD
OVI
LIM
V_IOUT
RT/SYNC
EN
PGOOD
CLKOUT
SGND
Pin Configuration
Package Information
For the latest package outline information and land patterns, go
to www.maxim-ic.com/packages.
PACKAGE TYPE PACKAGE CODE DOCUMENT NO.
28 TQFN T2855-3 21-0140
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________
25
© 2009 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.
Revision History
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
0 10/06 Initial release —
1 6/08 Replaced Compensation section and corrected Figure 4. 12, 23
2 3/09 Updated formula in Inductor Selection section. 20
