MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Typical Application Circuit
19-5673; Rev 1; 10/13
Ordering Information appears at end of data sheet.
General Description
The MAX16975 is a 1.2A current-mode step-down con-
verter with an integrated high-side switch. The device
operates with input voltages from 3.5V to 28V while using
only 45FA quiescent current at no load. The switching
frequency is adjustable from 220kHz to 1.0MHz by using
an external resistor, and can be synchronized to an exter-
nal clock. The device’s output voltage is pin-selectable to
a fixed 5V or adjustable from 1V to 10V using external
resistors. The wide input voltage range makes the device
ideal for automotive and industrial applications.
The device operates in skip mode for reduced current
consumption in light-load conditions. An adjustable
reset threshold helps keep microcontrollers alive down
to the lowest specified input voltage. Protection features
include cycle-by-cycle current limit, soft-start, overvolt-
age, and thermal shutdown with automatic recovery.
The device also features a power-good monitor to ease
power-supply sequencing.
The device is available in 16-pin QSOP and thermally
enhanced QSOP-EP packages. It operates over the
-40°C to +125°C automotive temperature range.
Features
S Wide 3.5V to 28V Input Voltage Range
S 42V Input Transient Tolerance
S 5V Fixed or 1V to 10V Adjustable Output Voltage
S Integrated 1.2A High-Side Switch
S 220kHz to 1.0MHz Adjustable Switching Frequency
S Frequency Synchronization Input
S Internal Boost Diode
S 45µA Skip-Mode Operating Current
S Less than 10µA Shutdown Current
S Adjustable Power-Good Output Level and Timing
S 3.3V Logic Level to 42V Compatible Enable Input
S Current-Limit, Thermal Shutdown, and
Overvoltage Protection
S -40°C to +125°C Automotive Temperature Range
Applications
Automotive
Industrial
For related parts and recommended products to use with this part, refer to: www.maximintegrated.com/MAX16975.related
D1 RFB1
25kI
RFB2
100kI
COUT1
47µF
COUT2
47µF
CIN2
4.7µF
CCRES
1nF
RCOMP
12kI
RRES
10kI
RFOSC
61.9kI
L1
10µH
VOUT = 1.25V AT
1.2A AT 400kHz
CBST
0.1µF
LX
BST
VOUT
OUT
3.5V TO 28V
FB
RESETI
VBIAS
RES
FOSC
CRES
CBIAS
1µF
CCOMP2
OPEN
BIAS
CCOMP1
5600pF
COMP
FSYNC
EN
SUPSWSUP
GND
CIN1
47µF
CIN3
0.1µF
MAX16975
PLACE CIN3 (0.1µF) RIGHT NEXT TO SUP.
For pricing, delivery, and ordering information, please contact Maxim Direct
at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com.
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
2Maxim Integrated
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional opera-
tion of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.)
SUP, SUPSW, LX, EN to GND ...............................-0.3V to +45V
BST to GND ..........................................................-0.3V to +47V
BST to LX ...............................................................-0.3V to +6V
OUT to GND ..........................................................-0.3V to +12V
SUP to SUPSW .....................................................-0.3V to +0.3V
RESETI, FOSC, COMP, BIAS,
FSYNC, CRES, RES, FB to GND .........................-0.3V to +6V
Output Short-Circuit Duration .................................... Continuous
Continuous Power Dissipation (TA = +70NC)
QSOP (derate 9.6mW/NC above +70NC) .................. 771.5mW
QSOP-EP (derate 22.7mW/NC above +70NC) ......1818.20mW
Operating Temperature Range ........................ -40NC to +125NC
Junction Temperature .....................................................+150NC
Storage Temperature Range ............................ -65NC to +150NC
Lead Temperature (soldering, 10s) ................................+300NC
Soldering Temperature (reflow) ......................................+260NC
ABSOLUTE MAXIMUM RATINGS
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-
layer board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
QSOP
Junction-to-Ambient Thermal Resistance (qJA) .....103.7°C/W
Junction-to-Case Thermal Resistance (qJC) ...............37°C/W
QSOP-EP
Junction-to-Ambient Thermal Resistance (qJA) ..........44°C/W
Junction-to-Case Thermal Resistance (qJC) .................6°C/W
PACKAGE THERMAL CHARACTERISTICS (Note 1)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Supply Voltage VSUP,
VSUPSW Normal operation 3.5 28 V
Supply Current ISUP
Normal operation, no switching 2.9 mA
Skip mode, no load, VOUT = 5V 45 FA
Shutdown Supply Current VEN = 0V 9FA
BIAS Regulator Voltage VBIAS VSUP = VSUPSW = 6V to 42V, VOUT < 3V or
VOUT > 5.5V, ILOAD = 0A (Note 2) 4.7 5.0 5.3 V
BIAS Undervoltage Lockout VUVBIAS VBIAS rising 2.95 3.15 3.35 V
BIAS Undervoltage Hysteresis 550 mV
Thermal-Shutdown Threshold +175 NC
Thermal-Shutdown Threshold
Hysteresis +15 NC
OUTPUT VOLTAGE (OUT)
Output Voltage VOUT
Normal operation, VFB = VBIAS, ILOAD = 1A,
TA = +25°C4.95 5 5.05
V
Normal operation, VFB = VBIAS, ILOAD = 1A,
-40°C P TA P +125°C4.9 5 5.1
Skip-Mode Output Voltage VOUT_SKIP No load, VFB = VBIAS (Note 3) 4.9 5.05 5.2 V
Load Regulation VOUT = 5V, VFB = VBIAS, 30mA < ILOAD < 1A 0.3 %
Line Regulation 6V < VSUP < 28V 0.02 %/V
*The parametric values (min, typ, max limits) shown in the Electrical Characteristics table supersede values quoted elsewhere in this data sheet.
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
3Maxim Integrated
ELECTRICAL CHARACTERISTICS* (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
BST Input Current IBST VBST - VLX = 5V 1.7 2.5 mA
LX Current Limit ILX
VSUP = 4.5V to 28V, VSUPSW = 14V,
TA = +25°C1.5 1.8 2.0 A
VSUP = 4.5V to 28V, VSUPSW = 14V 1.5 1.8
Skip-Mode Current Threshold ISKIP_TH 200 mA
Power-Switch On-Resistance RON RON measured between SUPSW and LX, ILX =
1A, VSUP = 4.5V to 28V, VBST - VLX = 4.5V 300 550 mI
LX Leakage Current ILX,LEAK VSUPSW = 28V, VLX = 0V, TA = +25°C0.01 1 FA
TRANSCONDUCTANCE AMPLIFIER (COMP)
FB Input Current IFB 20 nA
FB Regulation Voltage VFB
FB connected to an external resistive divider,
TA = +25°C0.99 1.0 1.01
V
FB connected to an external resistive divider,
-40°C P TA P +125°C 0.985 1.0 1.015
FB Line Regulation DVLINE 4.5V < VSUP < 28V 0.02 %/V
Transconductance (from FB to
COMP) gmVFB = 1V, VBIAS = 5V 1000 FS
Minimum On-Time tON 110 ns
Cold-Crank Event Duty Cycle DCCC 94 %
OSCILLATOR FREQUENCY
Oscillator Frequency
RFOSC = 25.5kI, VSUP = 4.5V to 28V 1.0 MHz
RFOSC = 61.9kI, VSUP = 4.5V to 28V 348 400 452 kHz
RFOSC = 120kI, VSUP = 4.5V to 28V (Note 3) 191 220 249 kHz
Oscillator Frequency Range fOSC (Note 3) 220 1000 kHz
EXTERNAL CLOCK INPUT (FSYNC)
External Input Clock Acquisition
Time tFSYNC 1 Cycles
External Input Clock Frequency (Note 3) fOSC +
10% Hz
External Input Clock High Threshold VFSYNC_HI VFSYNC rising 1.4 V
External Input Clock Low Threshold VFSYNC_LO VFSYNC falling 0.4 V
FSYNC Pulldown Resistance RFSYNC 500 kI
Soft-Start Time tSS
fSW = 400kHz 4ms
fSW = 1.0MHz 1.6 ms
ENABLE INPUT (EN)
Enable On Threshold Voltage Low VEN_LO 0.8 V
*The parametric values (min, typ, max limits) shown in the Electrical Characteristics table supersede values quoted elsewhere in this data sheet.
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
4Maxim Integrated
ELECTRICAL CHARACTERISTICS* (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.)
Note 2: When 3V < VOUT < 5.5V, the bias regulator is connected to the output to save quiescent current, VBIAS = VOUT.
Note 3: Guaranteed by design; not production tested.
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Enable On Threshold Voltage High VEN_HI 2.2 V
Enable Threshold Voltage
Hysteresis VEN,HYS 0.2 V
Enable Input Current IEN 10 nA
RESET
Reset Internal Switching Level VTH_RISING VFB rising, VRESETI = 0V 93 95 96.5 %VFB
VTH_FALLING VFB falling, VRESETI = 0V 91 93 95
RESETI Threshold Voltage VRESETI_HI VRESETI falling 1.05 1.25 1.4 V
CRES Threshold Voltage VCRES_HI VCRES rising 1.07 1.13 1.19 V
CRES Threshold Hysteresis VCRES_HYS 0.05 V
RESETI Input Current IRESET VRESETI = 0V 0.02 FA
CRES Source Current ICRES VOUT in regulation 9.5 10 10.5 FA
CRES Pulldown Current ICRES_PD VOUT out of regulation 1 mA
RES Output Low Voltage ISINK = 5mA 0.4 V
RES Leakage Current (Open-
Drain Output) VOUT in regulation TA = +25°C 1 FA
TA = +125°C 20 nA
Reset Debounce Time tRES_DEB VRESETI falling 25 Fs
*The parametric values (min, typ, max limits) shown in the Electrical Characteristics table supersede values quoted elsewhere in this data sheet.
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
5Maxim Integrated
Typical Operating Characteristics
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = +25NC, unless otherwise noted.)
STARTUP WITH FULL LOAD
(OUT = 1.25V, fSW = 400kHz)
MAX16975 toc01
0V
0V
0V
0V
OUT
1V/div
RES
5V/div
EN
5V/div
SUP
5V/div
2ms/div
ILOAD = 1.2A
EFFICIENCY vs. LOAD CURRENT
MAX16975 toc02
ILOAD (mA)
400 800 1200
10
20
30
40
50
60
70
80
90
100
0
0
8V/400kHz
5V/400kHz
3.3V/400kHz
1.25V/400kHz
EFFICIENCY (%)
SWITCHING FREQUENCY
vs. LOAD CURRENT (1.25V/400kHz)
MAX16975 toc03
ILOAD (mA)
SWITCHING FREQUENCY (kHz)
950700450
400.4
400.8
401.2
401.6
402.0
400.0
200 1200
PWM MODE
SWITCHING FREQUENCY vs. RFOSC
MAX16975 toc04
RFOSC (kI)
SWITCHING FREQUENCY (kHz)
98765432
400
600
800
1000
1200
200
10 120
5V OUTPUT
SWITCHING FREQUENCY vs. TEMPERATURE
(1.25V/400kHz, 5V/400kHz)
MAX16975 toc05
SWITCHING FREQUENCY (kHz)
370
390
410
430
450
350
TEMPERATURE (°C)
1109580655035205-10-25-40 125
ILOAD = 1.2A, RFOSC = 64.87kI
1.25V/400kHz
5V/400kHz
LOAD-STEP RESPONSE
(1.25V/400kHz)
MAX16975 toc06
0
0
VOUT
100mV/div
ILOAD
1A/div
4ms/div
0 TO 1.25A LOAD STEP
VOUT
AC-COUPLED
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
6Maxim Integrated
Typical Operating Characteristics (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = +25NC, unless otherwise noted.)
COLD-CRANK PULSE (1.25V/400kHz)
MAX16975 toc07
0V
0V
0V
0V
VSUPSW
10V/div
VOUT
1V/div
VRES
5V/div
VLX
10V/div
10ms/div
DIPS AND DROPS TEST (1.25V/400kHz)
MAX16975 toc08
0V
0V
0V
0V
VSUPSW
10V/div
VOUT
1V/div
VRES
5V/div
VLX
10V/div
10ms/div
SLOW VIN RAMP-UP TEST
MAX16975 toc09
VSUP/SUPSW
10V/div
VOUT
5V/div
VLX
10V/div
10s/div
OUTPUT SHORT-CIRCUIT TEST
(1.25V/400kHz)
MAX16975 toc10
0V
0V
0A
VOUT
2V/div
ILOAD
2A/div
VLX
10V/div
1ms/div
RLOAD = 0.3I
QUIESCENT CURRENT
vs. INPUT VOLTAGE
MAX16975 toc11
INPUT VOLTAGE (V)
QUIESCENT CURRENT (µA)
242016128
10
20
30
40
50
60
70
80
90
0
42
8
5V/400kHz
VOUT vs. TEMPERATURE IN PWM MODE
(5V/400kHz)
MAX16975 toc12
OUTPUT VOLTAGE CHANGE (%)
-1
0
1
2
-2
TEMPERATURE (°C)
1109580655035205-10-25-40 125
ILOAD = 1.2A
5V/400kHz
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
7Maxim Integrated
Typical Operating Characteristics (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = +25NC, unless otherwise noted.)
VOUT vs. TEMPERATURE IN PWM MODE
(1.25V/400kHz)
MAX16975 toc13
OUTPUT VOLTAGE CHANGE (%)
-1
0
1
2
-2
TEMPERATURE (°C)
1109580655035205-10-25-40 125
ILOAD = 1.2A
VOUT vs. TEMPERATURE IN SKIP MODE
(5V/400kHz)
MAX16975 toc14
OUTPUT VOLTAGE CHANGE (%)
-1
0
1
2
-2
TEMPERATURE (°C)
1109580655035205-10-25-40 125
ILOAD = 0A, SKIP MODE
5V/400kHz
VOUT vs. TEMPERATURE IN SKIP MODE
(1.25V/400kHz)
MAX16975 toc15
OUTPUT VOLTAGE CHANGE (%)
-1
0
1
2
-2
TEMPERATURE (°C)
1109580655035205-10-25-40 125
ILOAD = 0A, SKIP MODE
LINE REGULATION
MAX16975 toc16
VOUT (V)
4.95
5.00
5.05
5.10
4.90
VSUPSW (V)
2624222018161412108
62
8
5V/400kHz
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
8Maxim Integrated
Pin Configurations
Pin Description
PIN NAME FUNCTION
1 CRES
Analog Reset Timer. CRES sources 10FA (typ) of current into an external capacitor to set the reset timeout
period. Reset timeout period is defined as the time between the start of output regulation and RES switch-
ing to high impedance. Leave CRES unconnected for minimum delay time.
2 FOSC Resistor-Programmable Switching Frequency Control Input. Connect a resistor from FOSC to GND to set
the switching frequency (see the Internal Oscillator section).
3 FSYNC Synchronization Input. The device synchronizes to an external signal applied to FSYNC. The external signal
period must be 10% shorter than the internal clock period for proper operation.
4 I.C. Internally Connected. Connect to GND.
5 COMP Error-Amplifier Output. Connect a compensation network from COMP to GND for stable operation. See the
Compensation Network section.
6 FB
Feedback Input. Connect an external resistive divider from FB to OUT and GND to set the output voltage
between 1V and 10V. Connect FB directly to BIAS to set the output voltage to 5V. See the Applications
Information section.
7 OUT
Connect OUT to the output of the converter. OUT provides power to the internal circuitry when the output
voltage of the converter is set between 3V and 5.6V. During shutdown, OUT is pulled to GND with a 50I
resistor.
8 GND Ground
9 BIAS Linear Regulator Output. BIAS powers the internal circuitry. Bypass BIAS with a 1FF capacitor to ground as
close as possible to the device. During shutdown, BIAS is actively discharged through a 32kI resistor.
10 BST High-Side Driver Supply. Connect a 0.1FF capacitor between LX and BST for proper operation.
11 SUP Voltage Supply Input. SUP powers the internal linear regulator. Connect a 4.7FF capacitor from SUP to
ground. Connect SUP to SUPSW.
12 LX Inductor Connection. Connect a rectifying Schottky diode between LX and GND. Connect an inductor from
LX to the output.
EN
SUPSWI.C.
1
2
16
15
RESETI
RESFOSC
FSYNC
CRES
TOP VIEW
3
4
14
13
SUP
BSTOUT
512LXCOMP
FB 6
7
11
10
BIASGND 89
MAX16975A
+
QSOP
EN
SUPSWI.C.
1
2
16
15
RESETI
RESFOSC
FSYNC
CRES
TOP VIEW
3
4
14
13
SUP
BSTOUT
512LXCOMP
FB 6
7
11
10
BIASGND 89
MAX16975B
+
QSOP
EP
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
9Maxim Integrated
Pin Description (continued)
Functional Diagram
PIN NAME FUNCTION
13 SUPSW Internal High-Side Switch Supply Input. SUPSW provides power to the internal switch. Connect a 4.7FF
capacitor from SUPSW to ground. Connect SUP to SUPSW. See the Input Capacitor section.
14 EN Battery-Compatible Enable Input. Drive EN low to disable the device. Drive EN high to enable the device.
15 RES Open-Drain Active-Low Reset Output. RES asserts when VOUT is below the reset threshold set by RESETI.
16 RESETI Reset Threshold Level Input. Connect to a resistive divider to set the reset threshold for RES. Connect
RESETI to GND to enable the internal reset threshold.
— EP Exposed Pad (MAX16975A Only). Connect EP to a large-area contiguous copper ground plane for effec-
tive power dissipation. Do not use as the only IC ground connection. EP must be connected to GND.
10µA
COMP
COMP
B.G.
REF
SOFT-
START
UVLO
LDO
STANDBY
SUPPLY
REF EA
LOGIC FOR
100% DUTY-CYCLE
OPERATION
RES
VBIAS
FB
RESETI
CRES
PWM
COMP
LEVEL
SHIFT
ILIM
LOGIC
EN
OSC SUM
ISENSE
FSYNC
FOSCBIAS
SUP
OUT
COMP
DRV
LX
BST SUPSW
GND
MAX16975
MUX
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
10Maxim Integrated
Detailed Description
The MAX16975 is a constant-frequency, current-mode
automotive buck converter with an integrated high-side
switch. The device operates with input voltages from
3.5V to 28V and tolerates input transients up to 42V.
During undervoltage events, such as cold-crank condi-
tions, the internal pass device maintains 94% duty cycle
for a short time.
An open-drain, active-low reset output helps to monitor
the output voltage. The device offers an adjustable reset
threshold that helps to keep microcontrollers alive down
to the lowest specified input voltage and a capacitor-
programmable reset timeout to ensure proper startup.
The switching frequency is resistor-programmable from
220kHz to 1.0MHz to allow optimization for efficiency,
noise, and board space. A clock input, FSYNC, allows
the device to synchronize to an external clock.
During light-load conditions, the device enters skip
mode that reduces the quiescent current down to 45FA
and increases light-load efficiency. The 5V fixed output
voltage eliminates the need for external resistors and
reduces the supply current by up to 50FA.
Linear Regulator Output (BIAS)
The device includes a 5V linear regulator, VBIAS, that
provides power to the internal circuitry. Connect a 1FF
ceramic capacitor from BIAS to GND. When the output
voltage is set between 3V and 5.5V, the internal linear
regulator only provides power until the output is in regula-
tion. The internal linear regulator turns off once the output
is in regulation and allows OUT to provide power to the
device. The internal regulator turns back on once the
external load on the output of the device is higher than
100mA. In addition, the linear regulator turns on anytime
the output voltage is outside the 3V to 5.5V range.
External Clock Input (FSYNC)
The device synchronizes to an external clock signal
applied at FSYNC. The signal at FSYNC must have a fre-
quency of 10% higher than the internal clock frequency
for proper synchronization.
Adjustable Reset Level
The device features a programmable reset threshold
using a resistive divider between OUT, RESETI, and
GND. Connect RESETI to GND for the internal threshold.
RES asserts low when the output voltage falls to 93% of
the programmed level. RES deasserts when the output
voltage goes above 95% of the set voltage.
Some microprocessors accept a wide input voltage range
(3.3V to 5V, for example) and can operate during dropout
of the device. Use a resistive divider at RESETI to adjust
the reset activation level (RES goes low) to lower levels.
The reference voltage at RESETI is 1.25V (typ).
The device also offers a capacitor-programmable reset
timeout period. Connect a capacitor from CRES to GND
to adjust the reset timeout period. When the output volt-
age goes out of regulation, RES asserts low and the
reset timing capacitor discharges with a 1mA pulldown
current. Once the output is back in regulation the reset
timing capacitor recharges with 10FA (typ) current. RES
stays low until the voltage at CRES reaches 1.13V (typ).
Dropout Operation
The device features an effective maximum duty cycle to
help refresh the BST capacitor when continuously oper-
ated in dropout. When the high-side switch is on for three
consecutive clock cycles, the device forces the high-side
switch off during the final 35% of the fourth clock cycle.
When the high-side switch is off, the LX node is pulled low
by the current flowing through the inductor. This increases
the voltage across the BST capacitor. To ensure that the
inductor has enough current to pull LX to ground, an
internal load sinks current from VOUT when the device is
close to dropout and external load is small. Once the input
voltage is increased above the dropout region, the device
continues to regulate at the set output voltage.
The device operates with no load and no external clock
at an effective maximum duty cycle of 94% in deep drop-
out. This effective maximum duty cycle is influenced by
the external load and by the optional external synchro-
nized clock.
System Enable (EN)
An enable-control input (EN) activates the device from
the low-power shutdown mode. EN is compatible with
inputs from the automotive battery level down to 3.3V.
The high-voltage compatibility allows EN to be con-
nected to SUP, KEY/KL30, or the INH inputs of a CAN
transceiver.
EN turns on the internal regulator. Once VBIAS is above
the internal lockout level, VUVL = 3.15V (typ), the control-
ler activates and the output voltage ramps up within 2048
cycles of the switching frequency.
A logic-low at EN shuts down the device. During shut-
down, the internal linear regulator and gate drivers turn
off. Shutdown mode reduces the quiescent current to
9FA (typ). Drive EN high to turn on the device.
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
11Maxim Integrated
Overvoltage Protection
The device includes overvoltage protection circuitry that
protects the device when there is an overvoltage condi-
tion at the output. If the output voltage increases by more
than 12% of its set voltage, the device stops switching.
The device resumes regulation once the overvoltage
condition is removed.
Overload Protection
The overload protection circuitry is activated when the
device is in current limit and VOUT is below the reset
threshold. Under these conditions, the device enters
a soft-start mode. When the overcurrent condition is
removed before the soft-start mode is over, the device
regulates the output voltage to the set value. Otherwise,
the soft-start cycle repeats until the overcurrent condition
is removed.
Skip Mode
During light-load operation, IINDUCTOR P 200mA, the
device enters skip-mode operation. Skip mode turns off
the internal switch and allows the output to drop below
regulation voltage before the switch is turned on again.
The lower the load current, the longer it takes for the
regulator to initiate a new cycle effectively increasing
light-load efficiency. During skip mode, the device qui-
escent current drops to as low as 45FA.
Overtemperature Protection
Thermal-overload protection limits the total power dis-
sipation in the device. When the junction temperature
exceeds +175NC (typ), an internal thermal sensor shuts
down the step-down controller, allowing the device to
cool. The thermal sensor turns on the device again after
the junction temperature cools by +15NC.
Applications Information
Output Voltage/Reset Threshold
Resistive Divider Network
Although the device’s output voltage and reset threshold
can be set individually, Figure 1 shows a combined resis-
tive divider network to set the desired output voltage and
the reset threshold using three resistors. Use the follow-
ing formula to determine the RFB3 of the resistive divider
network:
TOTAL REF
FB3 OUT
RV
R V
×
=
where VREF = 1V, RTOTAL = selected total resistance of
RFB1, RFB2, and RFB3 in ohms, and VOUT is the desired
output voltage in volts.
Use the following formula to calculate the value of RFB2
of the resistive divider network:
TOTAL REF_RES
FB2 FB3
RES
RV
R R
V
×
= −
where VREF_RES is 1.25V (see the Electrical Characteristics
table) and VRES is the desired reset threshold in volts.
The precision of the reset threshold function is depen-
dent on the tolerance of the resistors used for the divider.
BST Capacitor Selection
for Dropout Operation
The device includes an internal boost capacitor refresh
algorithm for dropout operation. This is required to ensure
proper boost capacitor voltage that delivers power to the
gate-drive circuitry. When the HSFET is on consecutively
for 3.65 clock cycles, the internal counter detects this
and turns off the HSFET for 0.35 clock cycles. This is of
particular concern when VIN is falling and approaching
VOUT at the minimum switching frequency (220kHz).
The worst-case condition for boost capacitor refresh time
is with no load on the output. For the boost capacitor
to recharge completely, the LX node must be pulled to
ground. If there is no current through the inductor then
the LX node does not go to ground. To solve this issue,
an internal load of about 100mA turns on at the sixth
clock cycle, which is determined by a separate counter.
In the worst-case condition with no load, the LX node
does not go below ground during the first detect of the
Figure 1. Output Voltage/Reset Threshold Resistive Divider
Network
RFB3
RFB2
RFB1
VOUT
RESETI
FB
MAX16975
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
12Maxim Integrated
3.65 clock cycles. The device waits for the next 3.65
clock cycles to finish. As a result, the soonest the LX
node can go below ground is 4 + 3.65 = 7.65 clock
cycles. This time does not factor in the size of the induc-
tor and the time it takes for the inductor current to build
up to 100mA (internal load).
No load minimum time before refresh is:
∆T (no load) = 7.65 clock cycles = 7.65 x 4.54μs (at
220kHz) = 34.73μs
Assuming a full 100mA is needed to refresh the BST capac-
itor and depending on the size of the inductor, the time it
takes to build up full 100mA in the inductor is given by:
∆T (inductor) = L x ∆I/∆V (current build-up starts from the
sixth clock cycle)
L = inductor value chosen in the design guide.
∆I is the required current = 100mA.
∆V = voltage across the inductor (assume this to be
0.5V), which means VIN is greater than VOUT by 0.5V.
If ∆T (inductor) < 7.65 – 6 (clock cycles) then the BST
capacitor is sized as follows:
BST_CAP ≥ I_BST(dropout) x ∆T (no load)/∆V (BST
capacitor)
∆T (no load) = 7.65 clock cycles = 34.73μs.
∆V (BST capacitor), for (3.3V to 5V) output = VOUT – 2.7V
(2.7V is the minimum voltage allowed on the BST capaci-
tor).
If ∆T (inductor) > 7.65 - 6 clock cycles then we need to
wait for the next count of 3.65 clock cycles making ∆T (no
load) = 11.65 clock cycles.
Assume ∆T (no load) to be 16 clock cycles when design-
ing the BST capacitor with a typical inductor value for
220kHz operation.
The final BST_CAP equation is:
BST_CAP = I_BST (dropout) x ∆T (no load)/∆V (BST
capacitor)
where:
I_BST (dropout) = 2.5mA (worst case)
∆T (no load) = 16 clock cycles
∆V (BST capacitor) = VOUT - 2.7V
Reset Timeout Period
The device offers a capacitor-adjustable reset timeout
period. CRES can source 10FA of current. Use the fol-
lowing formula to set the timeout period.
1.13V C
RESET_TIMEOUT (s)
10 A
×
=µ
where C is the capacitor from CRES to GND in Farads.
Internal Oscillator
The device’s internal oscillator is programmable from
220kHz to 1.0MHz using a single resistor at FOSC. Use
the following formula to calculate the switching frequency:
9
OSC 26.4 10 ( x Hz)
f (Hz) R
×Ω
≈
where R is the resistor from FOSC to GND in ohms.
For example, a 220kHz switching frequency is set with
RFOSC = 120kI. Higher frequencies allow designs with
lower inductor values and less output capacitance.
Consequently, peak currents and I2R losses are lower
at higher switching frequencies, but core losses, gate-
charge currents, and switching losses increase.
Inductor Selection
Three key inductor parameters must be specified for
operation with the device: inductance value (L), inductor
saturation current (ISAT), and DC resistance (RDCR). To
select inductance value, the ratio of inductor peak-to-
peak AC current to DC average current (LIR) must be
selected first. A good compromise between size and loss
is a 30% peak-to-peak ripple current to average-current
ratio (LIR = 0.3). The switching frequency, input voltage,
output voltage, and selected LIR then determine the
inductor value as follows:
OUT SUPSW OUT
SUPSW SW OUT
V(V -V)
LV f I LIR
=
where VSUPSW, VOUT, and IOUT are typical values (so
that efficiency is optimum for typical conditions). The
switching frequency is set by RFOSC. The exact inductor
value is not critical and can be adjusted to make trade-
offs among size, cost, efficiency, and transient response
requirements. Table 1 shows a comparison between
small and large inductor sizes.
Table 1. Inductor Size Comparison
INDUCTOR SIZE
SMALLER LARGER
Lower price Smaller ripple
Smaller form-factor Higher efficiency
Faster load response Larger fixed-frequency
range in skip mode
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
13Maxim Integrated
The inductor value must be chosen so that the maximum
inductor current does not reach the minimum current limit
of the device. The optimum operating point is usually
found between 15% and 35% ripple current. When pulse
skipping (light loads), the inductor value also determines
the load-current value at which PFM/PWM switchover
occurs.
Find a low-loss inductor having the lowest possible
DC resistance that fits in the allotted dimensions. Most
inductor manufacturers provide inductors in standard
values, such as 1.0FH, 1.5FH, 2.2FH, 3.3FH, etc. Also
look for nonstandard values, which can provide a better
compromise in LIR across the input voltage range. If
using a swinging inductor (where the no-load inductance
decreases linearly with increasing current), evaluate
the LIR with properly scaled inductance values. For
the selected inductance value, the actual peak-to-peak
inductor ripple current (DIINDUCTOR) is defined by:
OUT SUPSW OUT
INDUCTOR SUPSW SW
V(V -V)
IV fL
∆= ××
where DIINDUCTOR is in A, L is in H, and fSW is in Hz.
Ferrite cores are often the best choices, although pow-
dered iron is inexpensive and can work well at 220kHz.
The core must be large enough not to saturate at the
peak inductor current (IPEAK):
INDUCTOR
PEAK LOAD(MAX)
I
II 2
∆
= +
Input Capacitor
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor RMS current requirement (IRMS) is
defined by the following equation:
OUT SUPSW OUT
RMS LOAD(MAX) SUPSW
V(V -V)
II V
=
IRMS is at a maximum value when the input voltage
equals twice the output voltage (VSUPSW = 2VOUT), so
IRMS(MAX) = ILOAD(MAX)/2.
Choose an input capacitor that exhibits less than +10NC
self-heating temperature rise at the RMS input current for
optimal long-term reliability.
The input-voltage ripple comprises DVQ (caused by the
capacitor discharge) and DVESR (caused by the ESR
of the capacitor). Use low-ESR ceramic capacitors with
high ripple-current capability at the input. Assume the
contribution from DVQ and DVESR to be 50%. Calculate
the input capacitance and ESR required for a specified
input-voltage ripple using the following equations:
ESR
IN L
OUT
V
ESR I
I
2
∆
=∆
+
where:
SUPSW OUT OUT
LSUPSW SW
(V - V ) V
IV fL
×
∆= ××
and
OUT
IN Q SW
I D(1- D)
CVf
×
=∆×
and
OUT
SUPSW
V
DV
=
IOUT is the maximum output current and D is the duty
cycle.
Output Capacitor
The output filter capacitor must have low enough equiva-
lent series resistance (ESR) to meet output ripple and
load transient requirements, yet have high enough ESR
to satisfy stability requirements. The output capacitance
must be high enough to absorb the inductor energy while
transitioning from full-load to no-load conditions without
tripping the overvoltage fault protection. When using
high-capacitance, low-ESR capacitors, the filter capaci-
tor’s ESR dominates the output voltage ripple. So the size
of the output capacitor depends on the maximum ESR
required to meet the output voltage ripple (VRIPPLE(P-P))
specifications:
RIPPLE(P P) LOAD(MAX)
V ESR I LIR
−
=××
The actual capacitance value required relates to the
physical size needed to achieve low ESR, as well as
to the chemistry of the capacitor technology. Thus, the
capacitor is usually selected by ESR and voltage rating
rather than by capacitance value.
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
14Maxim Integrated
When using low-capacity filter capacitors, such as
ceramic capacitors, size is usually determined by the
capacity needed to prevent VSAG and VSOAR from caus-
ing problems during load transients. Generally, once
enough capacitance is added to meet the overshoot
requirement, undershoot at the rising load edge is no
longer a problem. However, low-capacity filter capacitors
typically have high-ESR zeros that can affect the overall
stability.
Rectifier Selection
The device requires an external Schottky diode recti-
fier as a freewheeling diode. Connect this rectifier close
to the device using short leads and short PCB traces.
Choose a rectifier with a continuous current rating higher
than the highest output current-limit threshold (1.5A) and
with a voltage rating higher than the maximum expected
input voltage, VSUPSW. Use a low forward-voltage-drop
Schottky rectifier to limit the negative voltage at LX. Avoid
higher than necessary reverse-voltage Schottky rectifiers
that have higher forward-voltage drops.
Compensation Network
The device uses an internal transconductance error
amplifier with its inverting input and its output available for
external frequency compensation. The output capacitor
and compensation network determine the loop stability.
The inductor and the output capacitor are chosen based
on performance, size, and cost. Additionally, the compen-
sation network optimizes the control-loop stability.
The controller uses a current-mode control scheme that
regulates the output voltage by forcing the required cur-
rent through the external inductor, so the device uses
the voltage drop across the high-side MOSFET. Current-
mode control eliminates the double pole in the feedback
loop caused by the inductor and output capacitor result-
ing in a smaller phase shift and requiring less elaborate
error-amplifier compensation than voltage-mode control.
A simple single series resistor (RC) and capacitor (CC)
are all that is required to have a stable, high-bandwidth
loop in applications where ceramic capacitors are used
for output filtering (Figure 2). For other types of capaci-
tors, due to the higher capacitance and ESR, the fre-
quency of the zero created by the capacitance and
ESR is lower than the desired closed-loop crossover fre-
quency. To stabilize a nonceramic output capacitor loop,
add another compensation capacitor (CF) from COMP to
GND to cancel this ESR zero.
The basic regulator loop is modeled as a power modula-
tor, output feedback divider, and an error amplifier. The
power modulator has a DC gain set by gMC O RLOAD,
with a pole and zero pair set by RLOAD, the output
capacitor (COUT), and its ESR. The following equations
allow to approximate the value for the gain of the power
modulator (GAINMOD(DC)), neglecting the effect of the
ramp stabilization. Ramp stabilization is necessary when
the duty cycle is above 50% and is internally done for
the device.
LOAD SW
MOD(dc) MC LOAD SW
R fL
GAIN g R (f L)
××
= ×
+×
where RLOAD = VOUT/ILOUT(MAX) in I, fSW is the switch-
ing frequency in MHz, L is the output inductance in FH,
and gMC = 3S.
In a current-mode step-down converter, the output
capacitor, its ESR, and the load resistance introduce a
pole at the following frequency:
pMOD LOAD SW
OUT LOAD SW
1
fR fL
2 C ESR
R (f L)
=

××
π× × +

+×

The output capacitor and its ESR also introduce a zero at:
zMOD OUT
1
f2 ESR C
=π× ×
When COUT is composed of “n” identical capacitors in
parallel, the resulting COUT = n O COUT(EACH) and ESR
= ESR(EACH)/n. Note that the capacitor zero for a paral-
lel combination of alike capacitors is the same as for an
individual capacitor.
Figure 2. Compensation Network
R2
R1
VREF
VOUT
RC
CC
CF
COMP
gm
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
15Maxim Integrated
The feedback voltage-divider has a gain of GAINFB =
VFB/VOUT, where VFB is 1V (typ). The transconductance
error amplifier has a DC gain of GAINEA(DC) = gm,EA O
ROUT,EA, where gm,EA is the error-amplifier transcon-
ductance, which is 1000FS (typ), and ROUT,EA is the
output resistance of the 50MI error amplifier.
A dominant pole (fdpEA) is set by the compensa-
tion capacitor (CC) and the amplifier output resistance
(ROUT,EA). A zero (fzEA) is set by the compensation resis-
tor (RC) and the compensation capacitor (CC). There is
an optional pole (fpEA) set by CF and RC to cancel the
output capacitor ESR zero if it occurs near the crossover
frequency (fC, where the loop gain equals 1 (0dB)).
Thus:
dpEA C OUT,EA C
1
f2 C (R R )
=π× × +
zEA CC
1
f2C R
=π× ×
pEA FC
1
f2CR
=π× ×
The loop-gain crossover frequency (fC) is set below
1/5th the switching frequency and much higher than the
power-modulator pole (fpMOD):
SW
pMOD C
f
ff
5
<< ≤
The total loop gain as the product of the modulator gain,
the feedback voltage-divider gain, and the error-amplifier
gain at fC is equal to 1. So:
C
FB
MOD(fC) EA(f )
OUT
V
GAIN GAIN 1
V
×× =
For the case where fzMOD is greater than fC:
EA(fC) m,EA C
GAIN g R= ×
pMOD
MOD(fC) MOD(dc) C
f
GAIN GAIN f
= ×
Therefore:
FB
MOD(fC) m,EA C
OUT
V
GAIN g R 1
V
× × ×=
Solving for RC:
OUT
Cm,EA FB MOD(fC)
V
Rg V GAIN
=××
Set the error-amplifier compensation zero formed by RC
and CC (fzEA) at the fpMOD. Calculate the value of CC
as follows:
CpMOD C
1
C2f R
=π× ×
If fzMOD is lower than 5 x fC, add a second capacitor,
CF, from COMP to GND and set the compensation pole
formed by RC and CF (fpEA) at the fzMOD. Calculate the
value of CF as follows:
FzMOD C
1
C2f R
=π× ×
As the load current decreases, the modulator pole
also decreases; however, the modulator gain increases
accordingly and the crossover frequency remains the
same. For the case where fzMOD is less than fC:
The power-modulator gain at fC is:
C
pMOD
MOD(f ) MOD(dc) zMOD
f
GAIN GAIN f
= ×
The error-amplifier gain at fC is:
C
zMOD
EA(f ) m,EA C C
f
GAIN g R f
= ××
Therefore:
C
zMOD
FB
MOD(f ) m,EA C
OUT C
f
V
GAIN g R 1
Vf
× × ×× =
Solving for RC:
C
OUT C
Cm,EA FB MOD(f ) zMOD
Vf
Rg V GAIN f
×
=×× ×
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
16Maxim Integrated
Set the error-amplifier compensation zero formed by RC
and CC at the fpMOD (fzEA = fpMOD):
CpMOD C
1
C2f R
=π× ×
If fzMOD is less than 5 O fC, add a second capacitor, CF,
from COMP to GND. Set fpEA = fzMOD and calculate CF
as follows:
FzMOD C
1
C2f R
=π× ×
PCB Layout Guidelines
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer board
whenever possible for better noise immunity and power
dissipation. Follow these guidelines for good PCB layout:
1) Use a large contiguous copper plane under the
device package. Ensure that all heat-dissipating com-
ponents have adequate cooling.
2) Isolate the power components and high-current path
from the sensitive analog circuitry. This is essential to
prevent any noise coupling into the analog signals.
3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation. Make the high-current path com-
prising of an input capacitor, high-side FET, inductor,
and the output capacitor as short as possible.
4) Keep the power traces and load connections short. This
practice is essential for high efficiency. Use thick copper
PCBs (2oz vs. 1oz) to enhance full-load efficiency.
5) Route the analog signal lines away from the high-
frequency planes. This ensures integrity of sensitive
signals feeding back into the device.
6) Make the ground connection for the analog and
power section close to the device. This keeps the
ground current loops to a minimum. In cases where
only one ground is used, enough isolation between
analog return signals and high power signals must be
maintained.
Ordering Information
Chip Information
PROCESS: BiCMOS
Package Information
For the latest package outline information and land patterns (foot-
prints), go to www.maximintegrated.com/packages. Note that
a “+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
/V denotes an automotive qualified part.
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
PART TEMP RANGE PIN-PACKAGE
MAX16975AAEE/V+ -40°C to +125°C 16 QSOP-EP*
MAX16975BAEE/V+ -40°C to +125°C 16 QSOP
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
16 QSOP E16+4 21-0055 90-0167
16 QSOP-EP E16E+10 21-0112 90-0239
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent
licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and
max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated 160 Rio Robles, San Jose, CA 95134 USA 1-408-601-1000 17
© 2013 Maxim Integrated Products, Inc. Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
Revision History
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
0 3/11 Initial release —
1 10/13 Updated the General Description section and added thermal characteristics of the
QSOP-EP package 1, 2