FN9310 Rev.0.00 Page 1 of 47
Sep 11, 2018
FN9310
Rev.0.00
Sep 11, 2018
ISL81401, ISL81401A
40V 4-Switch Synchronous Buck-Boost Controller
DATASHEET
The ISL81401 and ISL81401A are 4-switch synchronous
buck-boost controller with peak and average current
sensing and monitoring at both ends. The ISL81401 is a
bidirectional device that can conduct current in both
directions while the ISL81401A is an unidirectional
device.
The ISL81401 and ISL81401A use the proprietary
buck-boost control algorithm with valley current
modulation for Boost mode and peak current modulation
for Buck mode control.
The ISL81401 and ISL81401A have four independent
control loops for input and output voltages and currents.
Inherent peak current sensing at both ends and
cycle-by-cycle current limit of this family of products
ensures high operational reliability by providing instant
current limit in fast transient conditions at either ends and
in both directions. It also has two current monitoring pins
at both input and output to facilitate Constant Current
(CC) limit and other system management functions. CC
operation down to low voltages avoids any runaway
condition at over load or short-circuit conditions. In
addition to multilayer overcurrent protection, it also
provides full protection features such as OVP, UVP, OTP,
and average and peak current limit on both input and
output to ensure high reliability in both unidirectional
and bidirectional operation.
The IC is packaged in a space conscious 32 Ld
5mmx5mm QFN package to improve thermal dissipation
and noise immunity. The unique DE/Burst mode at
light-load dramatically lowers standby power
consumption with consistent output ripple over different
load levels.
Related Literature
For a full list of related documents, visit our website:
•ISL81401 and ISL81401A product pages
Features
• Single inductor 4-switch buck-boost controller
• On-the-fly bidirectional operation with independent
control of voltage and current on both ends
• Proprietary algorithm for smoothest mode transition
• MOSFET drivers with adaptive shoot-through
protection
• Wide input voltage range: 4.5V to 40V
• Wide output voltage range: 0.8V to 40V
• Supports pre-biased output with SR soft-start
• Programmable frequency: 100kHz to 600kHz
• Supports parallel operation current sharing with
cascade phase interleaving
• External sync with clock out or frequency dithering
• External bias for higher efficiency supports 5V - 36V
input
• Output and input current monitor
• Selectable PWM mode operation between
PWM/DE/Burst modes
• Accurate EN/UVLO and PGOOD indicator
• Low shut down current: 2.7µA
• Complete protection: OCP, SCP, OVP, OTP, and UVP
• Dual-level OCP protection with average current
and pulse-by-pulse peak current limit
• Selectable OCP response with either hiccup or
constant current mode
• Negative pulse-by-pulse peak current limit
Applications
• Battery backup
• Type-C power supply and chargers
• Battery powered industrial applications
• Aftermarket automotive
• Redundant power supplies
• Robot and drones
• Medical equipment
• Security surveillance
Figure 1. Buck-Boost Power Train Topology
Q1
Q3
VIN VOUT
Q2
Q4
Rs_in Rs_out
L
UG1
LG1
UG2
LG2
PH1 PH2
FN9310 Rev.0.00 Page 2 of 47
Sep 11, 2018
ISL81401, ISL81401A
Figure 2. Typical Application Diagram
Figure 3. Efficiency (VOUT = 12V, CCM)
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82
84
86
88
90
92
94
96
98
100
0123456789
Efficiency (%)
IOUT(A)
VIN = 6V VIN = 9V
VIN = 12V VIN = 24V
VIN = 36V
FN9310 Rev.0.00 Page 3 of 47
Sep 11, 2018
ISL81401, ISL81401A
Contents
1. Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
1.1 Typical Application Schematics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
1.2 Block Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.3 Ordering Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
1.4 Pin Configurations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
1.5 Pin Descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
2. Specifications. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.1 Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.2 Thermal Information. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.3 Recommended Operating Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
2.4 Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
3. Typical Performance Curves. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
4. Functional Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
4.1 General Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
4.2 Internal 5.3V Linear Regulator (VDD), External Bias Supply (EXTBIAS),
and 5V Linear Regulator (VCC5V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
4.3 Enable (EN/UVLO) and Soft-Start Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
4.4 Tracking Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
4.5 Control Loops . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
4.6 Buck-Boost Conversion Topology and Control Algorithm . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
4.7 Light-Load Efficiency Enhancement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
4.8 Prebiased Power-Up . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
4.9 Frequency Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
4.10 Phase Lock Loop (PLL) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
4.11 Frequency Synchronization and Dithering. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
4.12 Gate Drivers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
4.13 Power-Good Indicator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
5. Protection Circuits. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
5.1 Input Undervoltage Lockout. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
5.2 VCC5V Power-On Reset (POR) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
5.3 Overcurrent Protection (OCP) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
5.4 Overvoltage Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
5.5 Over-Temperature Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
6. Layout Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
6.1 Layout Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
6.2 General EPAD Design Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
FN9310 Rev.0.00 Page 4 of 47
Sep 11, 2018
ISL81401, ISL81401A
7. Component Selection Guideline. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
7.1 MOSFET Considerations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
7.2 Inductor Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
7.3 Output Capacitor Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
7.4 Input Capacitor Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
8. Revision History. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
9. Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
FN9310 Rev.0.00 Page 5 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
1. Overview
1.1 Typical Application Schematics
Figure 4. ISL81401EVAL1Z (VIN = 6V to 40V, VOUT = 12V, IOUT = 8A) Evaluation Board Schematic
9,1 9287
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FN9310 Rev.0.00 Page 6 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
1.2 Block Diagram
Figure 5. Block Diagram
OVP
HICCUPOCP FB_OUT
PGOOD EN/UVLO VIN VDD EXTBIAS
FB_OUT
SS/TRK
CS+
2µA
SS/TRK
SS/TRK
+
_
+
_0.8V
REF
Slope-Comp Ramp
Generator Clock
RT
SYNC
EPAD
ZCD
PWM
VOUTVIN
IMON_OUT
VCC5V
LDOLDO
LDO
VDD
BOOT1
UG1
PHASE1
LG1
PGND
Adaptive Dead Time
Boot Refresh
Logic
VDD
BOOT2
UG2
PHASE2
LG2
OC_MODE
PGND
Adaptive Dead Time
Boot Refresh
Logic
GM1
COMP
CS-
A1 Slope
Compensation
Buck-Boost Logic
PWM Logic
DEM Burst Logic
PWM
Comparator
+
_
COMP
COMP
ISEN+
ISEN-
A2
+
_
VOUT
Pulse-by-Pulse Peak Current
Limit
+
_
PWM
Buck
PWM
Boost
FB_IN
CLKOUT
DITHER
+
_
+
_0.8V
REF
GM2
1.2V
GM3 GM4
IMON_IN
PLL_COMP
SGND
MODE
MODE
Pulse-by-Pulse Negative Peak
Current Limit
PGND
+
_
Enable POR
Bias Supplies
Reference
Fault Latch
FN9310 Rev.0.00 Page 7 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
1.3 Ordering Information
Part Number
(Notes 2, 3) Part Marking Temp. Range (°C)
Tape and Reel
(Units) (Note 1)
Package
(RoHS Compliant) Pkg. Dwg. #
ISL81401FRZ 81401 FRZ -40 to +125 - 32 Ld 5x5 QFN L32.5x5B
ISL81401FRZ-T 81401 FRZ -40 to +125 6k 32 Ld 5x5 QFN L32.5x5B
ISL81401FRZ-T7A 81401 FRZ -40 to +125 250 32 Ld 5x5 QFN L32.5x5B
ISL81401AFRZ 81401A FRZ -40 to +125 - 32 Ld 5x5 QFN L32.5x5B
ISL81401AFRZ-T 81401A FRZ -40 to +125 6k 32 Ld 5x5 QFN L32.5x5B
ISL81401AFRZ-T7A 81401A FRZ -40 to +125 250 32 Ld 5x5 QFN L32.5x5B
ISL81401EVAL1Z ISL81401 Evaluation Board
ISL81401AEVAL1Z ISL81401A Evaluation Board
Notes:
1. Refer to TB347 for details about reel specifications.
2. These Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and
100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free
soldering operations). Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), refer to the ISL81401 and ISL81401A product information pages. For more information
about MSL, see TB363.
Table 1. Summary of Key Differences
Device
VIN
Op/Maximum
(V) VDD (V)
Current
Direction Parallelable Dithering Removed Pins Pin 12
ISL81401 40/45 5.3 Bidirectional Yes Yes BSTEN, CLKEN MODE
ISL81401A 40/45 5.3 Unidirectional No No BSTEN, CLKEN,
FIB_IN, CLKOUT
MODE
ISL81601 60/70 8 Bidirectional Yes Yes None OV
FN9310 Rev.0.00 Page 8 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
1.4 Pin Configurations
ISL81401
32 Ld 5x5 QFN
Top View
EXTBIAS
VDD
RT/SYNC
FB_OUT
PGOOD
PHASE2
PGND
LG1
PHASE1
UG1
BOOT1
VIN
COMP
EPAD
VCC5V
PLL_COMP
ISEN-
ISEN+
CS+
CS-
UG2
BOOT2
LG2/OC_MODE
   
24
22
23
15 16
17
18
19
20
21
1
2
3
4
5
6
7
8
91011121314
FB_IN
IMON_OUT
CLKOUT/DITHER
EN/UVLO
IMON_IN
29303132
SS/TRK
NC
SGND
MODE
NC
NC
FN9310 Rev.0.00 Page 9 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
1.5 Pin Descriptions
ISL81401A
32 Ld 5x5 QFN
Top View
Pin
Number
Pin Name
(ISL81401)
Pin Name
(ISL81401A) Function
1 NC NC No connection pin.
2 FB_IN NC Input voltage feedback pin for reverse direction operation. Use a resistor divider to feed the
input voltage back to this pin. When the input voltage drops to the pin voltage below 0.8V, the
internal control loop reduces the duty cycle to sink in current from output to input to keep the pin
voltage regulated at 0.8V.
Keep the pin voltage below 0.3V to disable the reverse direction operation.
When the reverse operation function is not used, tie this pin to VCC5V or SGND to set up the
phase shift for the interleaving parallel operation.
3 VCC5V VCC5V Output of the internal 5V linear regulator. This output supplies bias for the IC. The VCC5V pin
must always be decoupled to SGND with a minimum of 4.7µF ceramic capacitor placed very
close to the pin.
4 NC NC No connection pin.
FN9310 Rev.0.00 Page 10 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
5 RT/SYNC RT/SYNC A resistor from this pin to ground adjusts the default switching frequency from 100kHz to
600kHz. The default switching frequency of the PWM controller is determined by the resistor RT
as shown in Equation 1.
where fSW is the switching frequency in MHz.
When this pin is open or tied to VCC5V, the fSW is set to 100kHz. When this pin is tied to GND,
the fSW is set to 550kHz.
When an external clock signal is applied to this pin, the internal frequency is synchronized to the
external clock frequency.
6 PLL_COMP PLL_COMP Compensation pin for the internal PLL circuit. A compensation network shown in the “Typical
Application Diagram” on page 2 is required. RPLL(2.7kΩ), CPLL1 (10nF), and CPLL2 (820pF) are
recommended.
7CLKOUT/
DITHER
NC Dual function pin. When a capacitor is not connected to this pin, it provides a clock signal to
synchronize the other ISL81401. The phase shift of the clock signal is set by the FB_IN and
IMON_IN pin voltages.
When a capacitor is connected to this pin, the clock out function is disabled and the frequency
dither function is enabled before the soft-start. The capacitor is charged and discharged by
internal current sources. As the voltage on the pin ramps up and down, the oscillator frequency
is modulated between –15% and +15% of the nominal frequency set by the RT resistor. The
frequency dither function is disabled in the external Sync mode or if the RT pin is open or
shorted.
8 SS/TRK SS/TRK Dual function pin. When used for soft-starting control, a soft-start capacitor is connected from
this pin to ground. A regulated 2µA soft-starting current charges up the soft-start capacitor and
ramps up the pin voltage. The pin voltage sets the reference of the output voltage control loop
during soft-start. The value of the soft-start capacitor sets the output voltage ramp up slope.
When used for tracking control, an external supply rail is configured as the master and the
output voltage of the master supply is applied to this pin using a resistor divider. The output
voltage tracks the master supply voltage.
9 COMP COMP Voltage error GM amplifier output. It sets the reference of the inner current loop. The feedback
compensation network is connected between the COMP and SGND pins. When the COMP pin
is pulled below 1V, the PWM duty cycle reduces to 0%.
10 FB_OUT FB_OUT Output voltage feedback input. Connect FB_OUT to a resistive voltage divider between the
output and SGND to adjust the output voltage. The FB_OUT pin voltage is regulated to the
internal 0.8V reference.
11 IMON_OUT IMON_OUT Output current monitor. The current from this pin is proportional to the differential voltage
between the ISEN+ and ISEN- pins. Connect a resistor and capacitor network between the pin
and SGND to make the pin voltage proportional to the average output current. When the pin
voltage reaches 1.2V, the internal average current limit loop reduces the output voltage to keep
the output current constant when constant current OCP mode is set or the converter shuts down
when hiccup OCP mode is set.
In DEM Burst mode, when this pin voltage is less than 840mV, the controller runs in Burst mode.
When this pin voltage is higher than 880mV, the controller exits Burst mode. When a higher
resistance on this pin is used to set its voltage higher than 880mV at no load condition, the
controller runs in DEM mode with no burst operation.
12 MODE MODE Input pin. Forces CCM when low or floating and allows DEM when high.
13 ISEN- ISEN- Output current sense signal negative input pin.
14 ISEN+ ISEN+ Output current sense signal positive input pin.
15 PGOOD PGOOD Open-drain logic output used to indicate the status of output voltage. This pin is pulled low when
the output is not within ±10% of the nominal voltage or the EN pin is pulled low.
16 UG2 UG2 High-side MOSFET gate driver output controlled by the boost PWM signal.
17 PHASE2 PHASE2 Phase node connection of the boost converter. This pin is connected to the junction of the upper
MOSFET’s source, filter inductor, and lower MOSFET’s drain of the boost converter.
Pin
Number
Pin Name
(ISL81401)
Pin Name
(ISL81401A) Function
RT
34.7
fSW
----------- 4.78–


k=
(EQ. 1)
FN9310 Rev.0.00 Page 11 of 47
Sep 11, 2018
ISL81401, ISL81401A 1. Overview
18 BOOT2 BOOT2 Bootstrap pin to provide bias for the boost high-side driver. The positive terminal of the
bootstrap capacitor connects to this pin. Connect a bootstrap diode between this pin and VDD
to create the bias for the high-side driver. The BOOT2 to PHASE2 voltage is monitored
internally. When the voltage drops to 3.9V at no switching condition, a minimum off-time pulse is
issued to turn off UG2 and turn on LG2 to refresh the bootstrap capacitor and maintain the
high-side driver bias voltage.
19 LG2/
OC_MODE
LG2/
OC_MODE
Low-side MOSFET gate driver output controlled by the boost PWM signal and OCP mode set
pin. The OCP mode is set by a resistor connected between the pin and ground during the
initiation stage before soft-start. During the initiation stage, the pin sources out 10µA current to
set the voltage on the pin. If the pin voltage is less than 0.3V, the OCP is set to constant current
mode. If the pin voltage is higher than 0.3V, the OCP is set to Hiccup mode.
20 VDD VDD Output of the internal 5.3V linear regulator supplied by either VIN or EXTBIAS. This output
supplies bias for the IC low-side drivers and the boot circuitries for the high-side drivers. The
VDD pin must always be decoupled to the PGND pin with a minimum 4.7µF ceramic capacitor
placed very close to the pin.
21 PGND PGND Power ground connection. This pin should be connected to the sources of the lower MOSFETs
and the (-) terminals of the VDD decoupling capacitors.
22 LG1 LG1 Low-side MOSFET gate driver output controlled by the PWM Buck signal.
23 BOOT1 BOOT1 Bootstrap pin to provide bias for the buck high-side driver. The positive terminal of the bootstrap
capacitor connects to this pin. Connect a bootstrap diode between this pin and VDD to create
the bias for the high-side driver. The BOOT1 to PHASE1 voltage is monitored internally. When
the voltage drops to 3.9V at no switching condition, a minimum off-time pulse is issued to turn
off UG1 and turn on LG1 to refresh the bootstrap capacitor and maintain the high-side driver
bias voltage.
24 PHASE1 PHASE1 Phase node connection of the buck converter. This pin is connected to the junction of the upper
MOSFET’s source, filter inductor, and lower MOSFET’s drain of the buck converter.
25 UG1 UG1 High-side MOSFET gate driver output controlled by the buck PWM signal.
26 EXTBIAS EXTBIAS External bias input for the optional VDD LDO. There is an internal switch to disconnect the VIN
LDO when EXTBIAS voltage is higher than typical 4.8V. Decouple this pin to ground with a
10µF ceramic capacitor when it is in use, otherwise tie this pin to ground. DO NOT float this pin.
27 VIN VIN Tie this pin to the input rail using a 5Ω to 10Ω resistor. It provides power to the internal LDO for
VDD. Decouple this pin with a small ceramic capacitor (10nF to 1µF) to ground.
28 CS+ CS+ Input current sense signal positive input pin.
29 CS- CS- Input current sense signal negative input pin.
30 EN/
UVLO
EN/
UVLO
This pin provides enable/disable and accurate UVLO functions. The output is disabled when the
pin is pulled to ground. When the voltage on the pin reaches 1.3V, the VDD and VCC5V LDOs
become active. When the voltage on the pin reaches 1.8V, the PWM modulator is enabled.
When the pin is floating, it is enabled in default by internal pull-up.
31 IMON_IN IMON_IN Input current monitor. The current from this pin is proportional to the differential voltage between
the CS+ and CS- pins. Connect a resistor and capacitor network between the pin and SGND to
make the pin voltage proportional to the average input current. When the pin voltage reaches
1.2V, the internal average current limit loop reduces the output voltage to keep the input current
constant when constant current OCP mode is set or the converter shuts down when hiccup
OCP mode is set.
When the input current monitor function is not used, tie this pin to VCC5V or SGND to set up the
phase shift for interleaving parallel operation.
32 NC NC No connection pin.
-SGND
EPAD
SGND
EPAD
Small-signal ground common to all control circuitry. Route this pin separately from the high
current ground (PGND). SGND and PGND can be tied together if there is one solid ground
plane with no noisy currents around the chip. All voltage levels are measured with respect to this
pin.
EPAD at ground potential. EPAD is connected to SGND internally. However, it is highly
recommended to solder it directly to the ground plane for better thermal performance and noise
immunity.
Pin
Number
Pin Name
(ISL81401)
Pin Name
(ISL81401A) Function
FN9310 Rev.0.00 Page 12 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
2. Specifications
2.1 Absolute Maximum Ratings
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions can
adversely impact product reliability and result in failures not covered by warranty.
2.2 Thermal Information
Notes:
4. JA is measured in free air with the component mounted on a high-effective thermal conductivity test board with “direct attach”
features. See TB379.
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Parameter Minimum Maximum Unit
VCC5V, EN/UVLO, FB_IN to GND -0.3 +5.9 V
VDD to GND -0.3 +9 V
EXTBIAS to GND -0.3 +40 V
VIN, CS+, CS-, ISEN+, ISEN- to GND -0.3 +45 V
BOOT1, 2/UG1, 2 to PHASE1, 2 -0.3 VDD + 0.3 V
PHASE1, 2 to GND -5 (<20ns)/
-0.3 (DC)
+45 V
MODE, FB_OUT, SS/TRK, COMP, RT/SYNC, PLL_COMP,
CLKOUT/DITHER, PGOOD, IMON_IN, IMON_OUT to GND
-0.3 VCC5V + 0.3 V
LG1, LG2/OC_MODE to GND -0.3 VDD + 0.3 V
CS+ to CS- and ISEN+ to ISEN- -0.3 +0.3 V
VCC5V, VDD Short-Circuit to GND Duration 1 s
ESD Ratings Value Unit
Human Body Model (Tested per JS-001-2017) 2 kV
Machine Model (Tested per JESD22-A115C) 150 V
Charge Device Model (Tested per JS-002-2014) 1.5 kV
Latch-Up (Tested per JESD78E; Class II, Level A, +125°C (TJ)) 100 mA
Thermal Resistance (Typical) JA (°C/W) JC (°C/W)
32 Ld QFN Package (Notes 4, 5)291.2
Parameter Minimum Maximum Unit
Junction Temperature -55 +150 °C
Operating Temperature -40 +125 °C
Storage Temperature Range -65 +150 °C
Pb-Free Reflow Profile see TB493
FN9310 Rev.0.00 Page 13 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
2.3 Recommended Operating Conditions
2.4 Electrical Specifications
Parameter Minimum Maximum Unit
Temperature -40 +125 °C
VIN to GND 4.5 40 V
VCC5V, EN/UVLO, FB_IN to GND 0 5.25 V
VDD to GND 0 5.6 V
EXTBIAS to GND 0 36 V
Recommended operating conditions unless otherwise noted. Refer to “Block Diagram” on page 6 and “Typical Application Schematics”
on page 5. VIN = 4.5V to 40V, or VDD = 5.3V ±10%, C_VCC5V = 4.7µF, TA = -40°C to +125°C, Typical values are at TA= +25°C, unless
otherwise specified. Boldface limits apply across the operating temperature range, -40°C to +125°C.
Parameter Symbol Test Conditions
Min
(Note 6)Typ
Max
(Note 6)Unit
VIN Supply
Input Voltage Range VIN 4.5 40.0 V
VIN Supply Current
Shutdown Current (Note 7)I
VINQ EN = 0V, PGOOD is floating 2.7 6.0 µA
Operating Current (Note 8)I
VINOP PGOOD is floating 4.7 6.0 mA
VCC5V Supply
Internal LDO Output Voltage VCC5V VIN = 8V, IL = 0mA 4.70 5.10 5.25 V
VIN = 40V, IL = 0mA 4.70 5.10 5.25 V
VIN = 4.5V, IL = 5mA 4.00 4.25 V
VIN > 5.6V, IL = 10mA 4.65 5.00 V
Maximum Supply Current of
Internal LDO
IVCC_MAX VVCC5V = 0V, VIN = 8V 120 mA
VDD Supply
Internal LDO Output Voltage VDD VIN = 12V, EXTBIAS = 0V, IL = 0mA 55.3 5.6 V
VIN = 40V, EXTBIAS = 0V, IL = 0mA 55.3 5.6 V
VIN = 4.5V, EXTBIAS = 12V, IL = 0mA 55.3 5.6 V
VIN = 40V, EXTBIAS = 12V, IL = 0mA 55.3 5.6 V
VIN = 4.5V, EXTBIAS = 0V, IL = 30mA 3.9 4.3 V
VIN = 4.5V, EXTBIAS = 7.8V, IL = 30mA 4.8 5.2 V
VIN > 8.6V, EXTBIAS = 0V, IL = 75mA 4.8 5.2 V
VIN = 4.5V, EXTBIAS > 9.0V, IL = 75mA 4.8 5.2 V
Maximum Supply Current of
Internal LDO
IVDD_MAX VVDD = 0V, EXTBIAS = 0V, VIN = 12V 120 mA
VVDD = 4.5V, EXTBIAS = 12V, VIN = 4.5V 160 mA
EXTBIAS Supply
Switch Over Threshold Voltage,
Rising
VEXT_THR EXTBIAS voltage 4.5 4.8 5V
Switch Over Threshold Voltage,
Falling
VEXT_THF EXTBIAS voltage 4.25 4.45 4.7 V
FN9310 Rev.0.00 Page 14 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
VIN UVLO
VIN Rising UVLO Threshold (Note 10)V
UVLOTHR VIN voltage, 0mA on VCC5V and VDD 3.20 3.50 3.85 V
VIN Falling UVLO Threshold VUVLOTHF VIN voltage, 0mA on VCC5V and VDD 3.0 3.2 3.4 V
VCC5V Power-On Reset
VCC5V Rising POR Threshold VPORTHR VCC5V voltage, 0mA on VCC5V and VDD 3.7 4.0 4.3 V
VCC5V Falling POR Threshold VPORTHF VCC5V voltage, 0mA on VCC5V and VDD 3.30 3.55 3.75 V
EN/UVLO Threshold
EN Rise Threshold VENSS_THR VIN > 5.6V 0.75 1.05 1.30 V
EN Fall Threshold VENSS_THF VIN > 5.6V 0.60 0.90 1.10 V
EN Hysteresis VENSS_HYST VIN > 5.6V 70 150 300 mV
UVLO Rise Threshold VUVLO_THR VIN > 5.6V 1.77 1.80 1.83 V
UVLO Hysteresis Current IUVLO_HYST VIN = 12V, EN/UVLO = 1.815V 2.5 4.2 5.5 µA
Soft-Start Current
SS/TRK Soft-Start Charge Current ISS SS/TRK = 0V 2.00 µA
Default Internal Minimum Soft-Starting
Default Internal Output Ramping Time tSS_MIN SS/TRK open 1.7 ms
Power-Good Monitors
PGOOD Upper Threshold VPGOV 107 109 112 %
PGOOD Lower Threshold VPGUV 87 90 92 %
PGOOD Low Level Voltage VPGLOW I_SINK = 2mA 0.35 V
PGOOD Leakage Current IPGLKG PGOOD = 5V 0 150 nA
PGOOD Timing
VOUT Rising Threshold to PGOOD
Rising (Note 9)
tPGR 1.1 5ms
VOUT Falling Threshold to PGOOD
Falling
tPGF 80 µs
Reference Section
Internal Voltage Loop Reference
Voltage
VREFV 0.800 V
Reference Voltage Accuracy TA = 0°C to +85°C -0.75 +0.75 %
TA = -40°C to +125°C -1.00 +1.00 %
Internal Current Loop Reference
Voltage
VREFI 1.200 V
Reference Voltage Accuracy TA = 0°C to +85°C -0.75 +0.75 %
TA = -40°C to +125°C -1.00 +1.00 %
PWM Controller Error Amplifiers
FB_OUT Pin Bias Current IFBOUTLKG -50 0+50 nA
FB_OUT Error Amp GM Gm1 1.75 mS
FB_OUT Error Amp Voltage Gain AV1 82 dB
FB_OUT Error Amp Gain-BW Product GBW1 8 MHz
Recommended operating conditions unless otherwise noted. Refer to “Block Diagram” on page 6 and “Typical Application Schematics”
on page 5. VIN = 4.5V to 40V, or VDD = 5.3V ±10%, C_VCC5V = 4.7µF, TA = -40°C to +125°C, Typical values are at TA= +25°C, unless
otherwise specified. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
Parameter Symbol Test Conditions
Min
(Note 6)Typ
Max
(Note 6)Unit
FN9310 Rev.0.00 Page 15 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
FB_OUT Error Amp Output Current
Capability
±310 µA
COMP Max High Voltage VCOMP_HIGH FB_OUT = 0V 3.8 V
COMP Min Low Voltage VCOMP_LOW FB_OUT = 1V 0.01 V
FB_IN Pin Bias Current IFBINLKG -50 0+50 nA
FB_IN Error Amp GM Gm2 12 µS
FB_IN Error Amp Voltage Gain AV2 72 dB
FB_IN Error Amp Gain-BW Product GBW2 5 MHz
FB_IN Active Range (Note 10)V
FB_IN_ACT VCC5V = 5V 04.3V
FB_IN Logic Low Threshold (Note 10)V
FB_IN_L 0.2 V
FB_IN Logic High Threshold (Note 10)V
FB_IN_H VCC5V = 5V 4.7 V
PWM Regulator
Buck Mode Minimum Off-Time tOFF_MIN1 220 ns
Buck Mode Minimum On-Time tON_MIN1 100 ns
Boost Mode Minimum Off-Time tOFF_MIN2 180 ns
Boost Mode Minimum On-Time tON_MIN2 140 ns
Buck Mode Peak-to-Peak Sawtooth
Amplitude
DVRAMP1 VIN = VOUT = 12V, fSW = 300kHz 1.0 V
Boost Mode Peak-to-Peak Sawtooth
Amplitude
DVRAMP2 VIN = VOUT = 12V, fSW = 300kHz 0.93 V
Buck Mode Ramp Offset VROFFSET1 0.88 0.95 1.11 V
Boost Mode Ramp Offset VROFFSET2 2.84 3.15 3.7 V
Current Sense, Current Monitors, and Average Current Loop
Input Current Sense Differential
Voltage Range
VCS+ - VCS- -80 +150 mV
Input Current Sense Common-Mode
Voltage Range
CMIRCS 040V
IMON_IN Offset Current ICSOFFSET CS+ = CS- = 12V 17.5 20 22 µA
Input Current Sense Voltage to
IMON_IN Current Source Gain
GmCS 12V common-mode voltage applied to
CS+/- pins, 0 to 40mV differential voltage
170 200 220 µS
IMON_IN Error Amp GM Gm3 12 µS
IMON_IN Error Amp Voltage Gain AV3 72 dB
IMON_IN Active Range (Note 10)V
IMON_IN_ACT VCC5V = 5V 04.3V
IMON_IN Logic High Threshold
(Note 10)
VIMON_IN_H VCC5V = 5V 4.7 V
IMON_IN Error Amp Gain-BW
Product
GBW3 5 MHz
Output Current Sense Differential
Voltage Range
VISEN+ -V
ISEN- -80 +150 mV
Output Current Sense
Common-Mode Voltage Range
CMIRISEN 040V
IMON_OUT Offset Current IISENOFFSET ISEN+ = ISEN- = 12V 17.5 20 22 µA
Recommended operating conditions unless otherwise noted. Refer to “Block Diagram” on page 6 and “Typical Application Schematics”
on page 5. VIN = 4.5V to 40V, or VDD = 5.3V ±10%, C_VCC5V = 4.7µF, TA = -40°C to +125°C, Typical values are at TA= +25°C, unless
otherwise specified. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
Parameter Symbol Test Conditions
Min
(Note 6)Typ
Max
(Note 6)Unit
FN9310 Rev.0.00 Page 16 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
IMON_OUT Current ISEN+ = 12V. ISEN- = 11.96V 25.5 27.6 29 µA
Output Current Sense Voltage to
IMON_OUT Current Source Gain
GmISEN 12V common-mode voltage applied to
ISEN+/- pins, 0mV to 40mV differential
voltage
170 200 220 µS
IMON_OUT Error Amp GM Gm4 12 µS
IMON_OUT Error Amp Voltage Gain AV4 72 dB
IMON_OUT Error Amp Gain-BW
Product
GBW4 5 MHz
Switching Frequency and Synchronization
Switching Frequency fSW RT = 144kΩ 220 245 265 kHz
RT = 72kΩ 420 450 485 kHz
RT Open or to VCC5V 90 120 145 kHz
RT = 0V 470 575 650 kHz
RT Voltage VRT RT = 72kΩ 580 mV
SYNC Synchronization Range fSYNC 140 600 kHz
SYNC Input Logic High VSYNCH 3.2 V
SYNC Input Logic Low VSYNCL 0.5 V
Clock Output and Frequency Dither
CLKOUT Output High VCLKH ISOURCE = 1mA, VCC5V = 5V 4.55 V
CLKOUT Output Low VCLKL ISINK = 1mA 0.3 V
CLKOUT Frequency fCLK RT = 72kΩ 420 450 485 kHz
Dither Mode Setting Current Source IDITHER_MODE_SO 10 µA
Dither Mode Setting Threshold Low VDITHER_MODE_L 0.26 V
Dither Mode Setting Threshold High VDITHER_MODE_H 0.34 V
Dither Source Current IDITHERSO 8µA
Dither Sink Current IDITHERSI 10 µA
Dither High Threshold Voltage VDITHERH 2.2 V
Dither Low Threshold Voltage VDITHERL 1.05 V
Diode Emulation Mode Detection
MODE Input Logic High 3.2 V
MODE Input Logic High 1V
Buck Mode Diode Emulation Phase
Threshold (Note 11)
VCROSS1 VIN = 12V 2 mV
Boost Mode Diode Emulation Shunt
Threshold (Note 12)
VCROSS2 VIN = 12V -2 mV
Diode Emulation Burst Mode
Burst Mode Enter Threshold VIMONOUTBSTEN IMON_OUT pin voltage 0.815 0.84 0.855 V
Burst Mode Exit Threshold VMONOUTBSTEX IMON_OUT pin voltage 0.86 0.88 0.89 V
Burst Mode Peak Current Limit Input
Shunt Set Point
VBST-CS VCS+ - VCS-, 12V common-mode voltage
applied to CS+/- pins
16 27 39 mV
Recommended operating conditions unless otherwise noted. Refer to “Block Diagram” on page 6 and “Typical Application Schematics”
on page 5. VIN = 4.5V to 40V, or VDD = 5.3V ±10%, C_VCC5V = 4.7µF, TA = -40°C to +125°C, Typical values are at TA= +25°C, unless
otherwise specified. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
Parameter Symbol Test Conditions
Min
(Note 6)Typ
Max
(Note 6)Unit
FN9310 Rev.0.00 Page 17 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
Burst Mode Peak FB Voltage Limit
Set Point
VBST-VFB-UTH 0.82 V
Burst Mode Exit FB Voltage Set Point VBST-VFB-LTH 0.78 V
PWM Gate Drivers
Driver 1, 2 BOOT Refresh Trip
Voltage
VBOOTRF1,2 BOOT voltage - PHASE voltage 3.45 3.9 4.35 V
Driver 1, 2 Source and Upper Sink
Current
IGSRC1,2 2000 mA
Driver 1, 2 Lower Sink Current IGSNK1,2 3000 mA
Driver 1, 2 Upper Drive Pull-Up RUG_UP1,2 2.2 Ω
Driver 1, 2 Upper Drive Pull-Down RUG_DN1,2 1.7 Ω
Driver 1, 2 Lower Drive Pull-Up RLG_UP1,2 3Ω
Driver 1, 2 Lower Drive Pull-Down RLG_DN 2Ω
Driver 1, 2 Upper Drive Rise Time tGR_UP COUT = 1000pF 10 ns
Driver 1, 2 Upper Drive Fall Time tGF_UP COUT = 1000pF 10 ns
Driver 1, 2 Lower Drive Rise Time tGR_DN COUT = 1000pF 10 ns
Driver 1, 2 Lower Drive Fall Time tGF_DN COUT = 1000pF 10 ns
Overvoltage Protection
Output OVP Threshold VOVTH_OUT 112 114 116 %
Overcurrent Protection
LG2/OC_MODE Current Source IMODELG2 7.5 10 12.5 µA
LG2/OC_MODE Threshold Low VMODETHLOC 0.26 V
LG2/OC_MODE Threshold High VMODETHHOC 0.34 V
Pulse-by-Pulse Peak Current Limit
Input Shunt Set Point
VOCSET-CS VCS+ - VCS-, 12V common-mode voltage
applied to CS+/- pins
73 83 93 mV
Hiccup Peak Current Limit Input
Shunt Set Point
VOCSET-CS-HIC VCS+ - VCS- 100 mV
Pulse-by-Pulse Negative Peak
Current Limit Output Shunt Set Point
VOCSET-ISEN VISEN+ - VISEN-, 12V common-mode
voltage applied to ISEN+/- pins
-70 -59 -48 mV
Hiccup and Current Input Constant
Limit Set Point
VIMONINCC IMON_IN Pin Voltage 1.185 1.2 1.215 V
Input Constant and Hiccup Current
Limit Set Point at CS+/- Input
VAVOCP_CS VCS+ - VCS-, 12V common-mode applied
to CS+/- pins, RIMON_IN = 40.2k,
TJ= -40°C to +125°C
44 51 64 mV
VCS+ - VCS-, 12V common-mode applied
to CS+/- pins, RIMON_IN = 40.2k,
TJ= -40°C to +85°C
44 51 60 mV
Output Constant and Hiccup Current
Limit Set Point
VIMONOUTCC IMON_OUT Pin Voltage 1.185 1.2 1.215 V
Recommended operating conditions unless otherwise noted. Refer to “Block Diagram” on page 6 and “Typical Application Schematics”
on page 5. VIN = 4.5V to 40V, or VDD = 5.3V ±10%, C_VCC5V = 4.7µF, TA = -40°C to +125°C, Typical values are at TA= +25°C, unless
otherwise specified. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
Parameter Symbol Test Conditions
Min
(Note 6)Typ
Max
(Note 6)Unit
FN9310 Rev.0.00 Page 18 of 47
Sep 11, 2018
ISL81401, ISL81401A 2. Specifications
Output Constant and Hiccup Current
Limit Set Point at ISEN+/- Input
VAVOCP_ISEN VISEN+ - VISEN-, 12V common-mode
applied to ISEN+/- pins,
RIMON_OUT = 40.2k, TJ = -40°C to +125°C
44 51 64 mV
VISEN+ - VISEN-, 12V common-mode
applied to ISEN+/- pins,
RIMON_OUT = 40.2k, TJ = -40°C to +85°C
44 51 60 mV
Hiccup OCP Off-Time tHICC_OFF 50 ms
Over-Temperature
Over-Temperature Shutdown TOT-TH 160 °C
Over-Temperature Hysteresis TOT-HYS 15 °C
Notes:
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested.
7. This is the total shutdown current with VIN = 5.6V and 40V.
8. Operating current is the supply current consumed when the device is active but not switching. It does not include gate drive
current.
9. When soft-start time is less than 4.5ms, tPGR increases. With internal soft-start (the fastest soft-start time), tPGR increases close
to its max limit 5ms.
10. Compliance to datasheet limits is assured by one or more methods: production test, characterization, and/or design.
11. Threshold voltage at the PHASE1 pin for turning off the buck bottom MOSFET during DE mode.
12. Threshold voltage between the ISEN+ and ISEN- pins for turning off the boost top MOSFET during DE mode.
Recommended operating conditions unless otherwise noted. Refer to “Block Diagram” on page 6 and “Typical Application Schematics”
on page 5. VIN = 4.5V to 40V, or VDD = 5.3V ±10%, C_VCC5V = 4.7µF, TA = -40°C to +125°C, Typical values are at TA= +25°C, unless
otherwise specified. Boldface limits apply across the operating temperature range, -40°C to +125°C. (Continued)
Parameter Symbol Test Conditions
Min
(Note 6)Typ
Max
(Note 6)Unit
FN9310 Rev.0.00 Page 19 of 47
Sep 11, 2018
ISL81401, ISL81401A 3. Typical Performance Curves
3. Typical Performance Curves
Oscilloscope plots are taken using the ISL81401EVAL1Z evaluation board, VIN = 9V to 40V, VOUT = 12V, IOUT = 10A, unless otherwise
noted.
Figure 6. Shutdown Current vs Temperature Figure 7. Quiescent Current vs Temperature
Figure 8. VDD Load Regulation at 12V Input Figure 9. VDD Line Regulation at 20mA Load
Figure 10. VCC5V Load Regulation at 12VIN Figure 11. VCC5V Line Regulation at 20mA Load
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
-60 -40 -20 0 20 40 60 80 100 120 140
Shutdown Current (μA)
Temperature (°C)
Vin = 12V Vin = 40V
Vin = 4.5V Vin = 5.6V
4.0
4.1
4.2
4.3
4.4
4.5
4.6
4.7
-60 -40 -20 0 20 40 60 80 100 120 140
Quiescent Current (mA)
Temperature (°V)
Vin = 12V Vin = 40V
Vin = 4.5V Vin = 5.6V
0
1
2
3
4
5
6
020406080100120140
VDD (V)
Load Current (mA)
VIN = 12V, Vextbias = 0
VIN = 4.5V, Vextbias = 12V
4.0
4.2
4.4
4.6
4.8
5.0
5.2
5.4
0 10203040
50
V
DD
(V)
V
IN
, Vextbias (V)
VDD vs VIN
VDD Vs Vextbias
0
1
2
3
4
5
6
0 20406080100120
VCC5V (V)
Load Current (mA)
4.4
4.5
4.6
4.7
4.8
4.9
5.0
5.1
4.4 4.6 4.8 5.0 5.2 5.4
5.6
VCC5V (V)
VDD (V)
FN9310 Rev.0.00 Page 20 of 47
Sep 11, 2018
ISL81401, ISL81401A 3. Typical Performance Curves
Figure 12. Switching Frequency vs Temperature Figure 13. Switching Frequency vs VIN, RT = 144k
Figure 14. 0.8V Reference Voltage vs Temperature Figure 15. 1.2V Reference Voltage vs Temperature
Figure 16. Normalized Output Voltage vs Voltage on
Soft-Start Pin
Figure 17. Input Current IIN (DC) vs IMON_IN Pin Voltage,
RS_IN = 3mΩ, RIM_IN = 37.4k
Oscilloscope plots are taken using the ISL81401EVAL1Z evaluation board, VIN = 9V to 40V, VOUT = 12V, IOUT = 10A, unless otherwise
noted. (Continued)
200
250
300
350
400
450
500
-50 0 50 100 150
Switching Frequency (kHz)
Temperature (°C)
RT= 72kΩ
RT= 144kΩ
231
232
233
234
235
236
237
238
239
240
241
01020304050
fSW (kHz)
VIN (V)
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
-50 0 50 100 150
0.8V Reference Voltage (V)
Temperature (°C)
1.195
1.196
1.197
1.198
1.199
1.200
1.201
1.202
1.203
1.204
1.205
-50 0 50 100 150
1.2V Reference Voltage (V)
Temperature (°C)
0
20
40
60
80
100
120
0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5
Normalized Output Voltate (% )
Soft-Start Pin Voltage (V)
0.60
0.65
0.70
0.75
0.80
0.85
0.90
0.95
1.00
02468
10
V_IMON_IN (V)
IIN
㸦
A)
TA = +25°C TA = +125°C
TA = -40°C
FN9310 Rev.0.00 Page 21 of 47
Sep 11, 2018
ISL81401, ISL81401A 3. Typical Performance Curves
Figure 18. Output Current IOUT (DC) vs IMON_OUT
Pin Voltage, RS_OUT = 4mΩ, RIM_OUT = 43.2k
Figure 19. CCM Mode Efficiency
Figure 20. DEM Mode Efficiency Figure 21. CCM Load Regulation at +25°C
Figure 22. CCM Line Regulation at 10A Load +25°C Figure 23. Boost Mode Waveforms, VIN = 6V, IOUT = 8A,
CCM Mode
Oscilloscope plots are taken using the ISL81401EVAL1Z evaluation board, VIN = 9V to 40V, VOUT = 12V, IOUT = 10A, unless otherwise
noted. (Continued)
0.6
0.7
0.8
0.9
1.0
1.1
1.2
1.3
02468
10
V_IMON_OUT (V)
I
OUT
(A)
TA = +25°C TA = +125°C
TA = -40°C
70
75
80
85
90
95
100
0.0 1.6 3.2 4.8 6.4
8.0
Efficiency (%)
IOUT (A)
VIN = 6V VIN = 9V
VIN = 12V VIN = 24V
VIN = 36V
70
75
80
85
90
95
100
01234567
8
Efficiency (%)
IOUT (A)
VIN = 6V VIN = 9V
VIN = 12V VIN = 24V
VIN = 36V
11.80
11.85
11.90
11.95
12.00
12.05
12.10
12.15
12.20
012345678
VOUT (V)
IOUT (A)
VIN = 6V VIN = 9V
VIN = 12V VIN = 24V
VIN = 36V
11.90
11.91
11.92
11.93
11.94
11.95
11.96
11.97
11.98
11.99
12.00
5 101520253035
VOUT (V)
VIN (V)
4µs/Div
PHASE1 5V/Div
PHASE2 10V/Div
VOUT 200mV/Div
IL 10A/Div
FN9310 Rev.0.00 Page 22 of 47
Sep 11, 2018
ISL81401, ISL81401A 3. Typical Performance Curves
Figure 24. Buck-Boost Mode Waveforms,
VIN = 12V, IOUT = 8A, CCM Mode
Figure 25. Buck Mode Waveforms, VIN = 40V, IOUT = 8A,
CCM Mode
Figure 26. DEM Mode Waveforms, VIN = 6V, IOUT = 0.01A Figure 27. Burst Mode Waveforms, VIN = 6V, IOUT = 0.1A
Figure 28. Burst Mode Waveforms, VIN = 12V, IOUT = 0.1A Figure 29. Burst Mode Waveforms, VIN = 40V, IOUT = 0.1A
Oscilloscope plots are taken using the ISL81401EVAL1Z evaluation board, VIN = 9V to 40V, VOUT = 12V, IOUT = 10A, unless otherwise
noted. (Continued)
4µs/Div
PHASE1 10V/Div
PHASE2 10V/Div
VOUT 200mV/Div
IL 10A/Div
4µs/Div
PHASE1 50V/Div
PHASE2 10V/Div
VOUT 200mV/Div
IL 10A/Div
4µs/Div
PHASE1 5V/Div
VOUT 50mV/Div
IL 1A/Div
PHASE2 10V/Div
2ms/Div
PHASE1 5V/Div
VOUT 200mV/Div
IL 10A/Div
PHASE2 10V/Div
2ms/Div
PHASE1 10V/Div
VOUT 200mV/Div
IL 10A/Div
PHASE2 10V/Div
PHASE1 50V/Div
VOUT 200mV/Div
IL 10A/Div
PHASE2 10V/Div
2ms/Div
FN9310 Rev.0.00 Page 23 of 47
Sep 11, 2018
ISL81401, ISL81401A 3. Typical Performance Curves
Figure 30. Load Transient, VIN = 6V, IOUT = 0A to 8A,
2.5A/µs, CCM
Figure 31. Load Transient, VIN = 12V, IOUT = 0A to 8A,
2.5A/µs, CCM
Figure 32. Load Transient, VIN = 40V IOUT = 0A to 8A,
2.5A/µs, CCM
Figure 33. Line Transient, VIN = 6V to 40V, 1V/ms,
IOUT = 0A
Figure 34. Line Transient, VIN = 40V to 6V, 0.5V/ms,
IOUT =0A
Figure 35. Start-Up Waveform, VIN = 6V IO = 8A, CCM
Oscilloscope plots are taken using the ISL81401EVAL1Z evaluation board, VIN = 9V to 40V, VOUT = 12V, IOUT = 10A, unless otherwise
noted. (Continued)
VOUT 200mV/Div
IOUT 5A/Div
2ms/Div
VOUT 200mV/Div
IOUT 5A/Div
2ms/Div
2ms/Div
VOUT 200mV/Div
IL 5A/Div
VOUT 200mV/Div
IL 10A/Div
VIN 20V/Div
10ms/Div
VOUT 200mV/Div
IL 10A/Div
VIN 20V/Div
20ms/Div
4ms/Div
PHASE1 5V/Div
PHASE2 10V/Div
VOUT 5V/Div
IL 10A/Div
FN9310 Rev.0.00 Page 24 of 47
Sep 11, 2018
ISL81401, ISL81401A 3. Typical Performance Curves
Figure 36. Start-Up Waveform, VIN = 12V IO = 8A, CCM Figure 37. Start-Up Waveform, VIN = 40V IO = 8A, CCM
Figure 38. OCP Response, Output Short-Circuited from
No Load to Ground and Released, CCM Mode, VIN = 12V
Figure 39. Constant Voltage (CV) and
Constant Current (CC) Operation
Figure 40. Bi-Directional Operation
VIN = 18V, VIN Regulation at 6V, Remove VIN DC Source with 1A Load Applied on Input Terminals
Oscilloscope plots are taken using the ISL81401EVAL1Z evaluation board, VIN = 9V to 40V, VOUT = 12V, IOUT = 10A, unless otherwise
noted. (Continued)
4ms/Div
PHASE1 10V/Div
PHASE2 10V/Div
VOUT 5V/Div
IL 10A/Div
4ms/Div
PHASE1 50V/Div
PHASE2 10V/Div
VOUT 5V/Div
IL 10A/Div
40ms/Div
PHASE1 10V/Div
PHASE2 10V/Div
VOUT 5V/Div
IL 20A/Div
0
2
4
6
8
10
12
14
024681012
VOUT (V)
IOUT (A)
VIN = 6V VIN = 12V
VIN = 40V
VIN 5V/Div
VOUT 10V/Div
4ms/Div
IL 20A/Div
FN9310 Rev.0.00 Page 25 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
4. Functional Description
4.1 General Description
The ISL81401 and ISL81401A implement a complete buck-boost switching control with a PWM controller,
internal drivers, references, protection circuitry, current and voltage control inputs, and monitor outputs. Refer to
Figure 5 on page 6.
The ISL81401 and ISL81401A are current-mode controllers. They use a proprietary control algorithm to
automatically switch between Buck and Boost modes as necessary to maintain a steady output voltage with
changing input voltages and dynamic external loads. The controllers integrate four control loops to regulate not
only VOUT, but also average IOUT and IIN for constant current control and VIN for reverse direction control.
The driver and protection circuits are also integrated to simplify the end design.
The part has an independent enable/disable control line, which provides a flexible power-up sequencing and a
simple VIN UVP implementation. The soft-start time is programmable by adjusting the soft-start capacitor
connected from SS/TRK.
4.2 Internal 5.3V Linear Regulator (VDD), External Bias Supply (EXTBIAS),
and 5V Linear Regulator (VCC5V)
The ISL81401 and ISL81401A provide two input pins, VIN and EXTBIAS, and two internal LDOs for VDD gate
driver supply. A third LDO generates VCC5V from VDD. VCC5V provides power to all internal functional
circuits other than the gate drivers. Bypass the linear regulator’s outputs (VDD) with a 10µF capacitor to the power
ground. Also bypass the third linear regulator output (VCC5V) with a 10µF capacitor to the signal ground. VCC5V
is monitored by a power-on-reset circuit, which disables all regulators when VCC5V falls below 3.5V.
Both LDOs from VIN and EXTBIAS can source over 75mA for VDD to power the buck and boost gate drivers.
When driving large FETs at high switching frequency, little or no regulator current may be available for external
loads. The LDO from VDD to VCC5V can also source over 75mA to supply the IC internal circuit. Although the
current consumed by the internal circuit is low, the current supplied by VCC5V to the external loads is limited by
VDD. For example, a single large FET with 15nC total gate charge requires 15nC x 300kHz = 4.5mA
(15nC x 600kHz = 9mA).
Also, at higher input voltages with larger FETs, the power dissipation across the internal 5.3V LDO increases.
Excessive power dissipation across this regulator must be avoided to prevent junction temperature rise. Thermal
protection may be triggered if die temperature increases above +150°C due to excessive power dissipation.
When large MOSFETs are used, an external 5V bias voltage can be applied to the EXTBIAS pin to alleviate
excessive power dissipation. When the voltage at the EXTBIAS pin is higher than typical 4.8V, the LDO from
EXTBIAS activates and the LDO from VIN is disconnected. The recommended maximum voltage at the
EXTBIAS pin is 36V. For applications with VOUT significantly lower than VIN, EXTBIAS is usually back biased
by VOUT to reduce the LDO power loss. EXTBIAS is allowed to activate only after soft-start is finished to avoid
early activation during the VOUT rising stage. An external UVLO circuit might be necessary to ensure smooth
soft-starting. Renesas recommends adding a 10µF capacitor on the EXTBIAS pin and using a diode to connect the
EXTBIAS pin to VOUT to avoid the EXTBIAS pin voltage being pulled low at the VOUT short-circuit condition.
The two VDD LDOs have an overcurrent limit for short-circuit protection. The VIN to VDD LDO current limit is
set to typical 120mA. The EXTBIAS to VDD LDO current limit is set to typical 160mA.
4.3 Enable (EN/UVLO) and Soft-Start Operation
Pulling the EN/UVLO pin high or low can enable or disable the controller. When the EN/UVLO pin voltage is
higher than 1.3V, the three LDOs are enabled. After the VCC5V reaches the POR threshold, the controller is
powered up to initialize its internal circuit. When EN/UVLO is higher than the 1.8V accurate UVLO threshold, the
ISL81401 and ISL81401A soft-start circuitry becomes active. The internal 2µA charge current begins charging up
the soft-start capacitor connected from the SS/TRK pin to GND. The voltage error amplifier reference voltage is
FN9310 Rev.0.00 Page 26 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
clamped to the voltage on the SS/TRK pin. The output voltage thus rises from 0V to regulation as SS/TRK rises
from 0V to 0.8V. Charging of the soft-start capacitor continues until the voltage on the SS/TRK pin reaches 3V.
Typical applications for the ISL81401 and ISL81401A use programmable analog soft-start or the SS/TRK pin for
tracking. The soft-start time can be set by the value of the soft-start capacitor connected from SS/TRK to GND.
Inrush current during start-up can be alleviated by adjusting the soft-starting time.
The typical soft-start time is set according to Equation 2:
When the soft-starting time set by external CSS or tracking is less than 1.5ms, an internal soft-start circuit of 1.5ms
takes over the soft-start.
PGOOD toggles to high when the output voltage is in regulation.
Pulling the EN/UVLO lower than the EN threshold of 1.3V disables the PWM output and internal LDOs to achieve
low standby current. The SS/TRK pin is also discharged to GND by an internal MOSFET with 70Ω rDS(ON). For
applications with a larger than 1µF capacitor on the SS/TRK pin, Renesas recommends adding a 100Ω to 1kΩ
resistor in series with the capacitor to share the power loss at the discharge.
With use of the accurate UVLO threshold, an accurate VIN Undervoltage Protection (UVP) feature can be
implemented by feeding the VIN into the EN/UVLO pin using a voltage divider, RUV1 and RUV2, shown in
Figure 41.
The VIN UVP rising threshold can be calculated by Equation 3.
where VUVLO_THR is the EN/UVLO pin UVLO rising threshold, typically 1.8V.
The VIN UVP falling threshold can be calculated by Equation 4.
where IUVLO_HYST is the UVLO hysteresis current, typically 4.2µA.
Figure 41. VIN Undervoltage Protection
tSS 0.8V
CSS
2A
-----------


=
(EQ. 2)
EN/UVLO
VIN
RUV1
RUV2
ISL81401/
ISL81041A
VUVRISE
VUVLO_THR RUV1 RUV2
+1.1x10 6–
––RUV1RUV2
RUV2
--------------------------------------------------------------------------------------------------------------------------------------------------
=
(EQ. 3)
VUVFALL
VUVLO_THR RUV1 RUV2
+IUVLO_HYST
–RUV1RUV2
RUV2
----------------------------------------------------------------------------------------------------------------------------------------------------------
=
(EQ. 4)
FN9310 Rev.0.00 Page 27 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
4.4 Tracking Operation
The ISL81401 and ISL81401A can track an external supply. To implement tracking, connect a resistive divider
between the external supply output and ground. Connect the center point of the divider to the SS/TRK pin of the
ISL81401 and ISL81401A. The resistive divider ratio sets the ramping ratio between the two voltage rails. To
implement coincident tracking, set the tracking resistive divider ratio exactly the same as the ISL81401 and
ISL81401A output resistive divider given by Equation 5 on page 27. Make sure that the voltage at SS/TRK is
greater than 0.8V when the master rail reaches regulation.
To minimize the impact of the 2µA soft-start current on the tracking function, Renesas recommends using resistors
less than 10kΩ for the tracking resistive divider.
When the SS/TRK pin voltage is pulled down to less than 0.3V by the external tracking source, the prebias startup
DEM function is enabled again. The output voltage may not be able to be pull down if the load current is not
highenough.
When Overcurrent Protection (OCP) is triggered, the internal minimum soft-start circuit determines the 50ms OCP
soft-start hiccup off-time.
4.5 Control Loops
The ISL81401 and ISL81401A are current-mode controllers that can provide an output voltage above, equal to, or
below the input voltage. Referring to Figure 2 on page 2 (Typical Application circuit) and Figure 5 on page 6
(Block Diagram), the Renesas proprietary control architecture uses a current sense resistor in series with the buck
upper FET to sense the inductor current in Buck or Boost mode. The inductor current is controlled by the voltage
on the COMP pin, which is the lowest output of the error amplifiers Gm1 - Gm4. As the simplest example, when
the output is regulated to a constant voltage, the FB_OUT pin receives the output feedback signal, which is
compared to the internal reference by Gm1. Lower output voltage creates higher COMP voltage, which leads to
higher PWM duty cycle to push more current to the output. Conversely, higher output voltage creates lower COMP
voltage, which leads to lower PWM duty cycle to reduce the current to the output.
The ISL81401 and ISL81401A have four error amplifiers (Gm1-4) which can control output voltage (Gm1), input
voltage (Gm2), input current (Gm3), and output current (Gm4). In a typical application, the output voltage is
regulated by Gm1, and the remaining error amplifiers are monitoring for excessive input or output current or an
input undervoltage condition. In other applications, such as a battery charger, the output current regulator (Gm4)
implements constant current charging until a predetermined voltage is reached, at which point the output voltage
regulator (Gm1) takes control.
4.5.1 Output Voltage Regulation Loop
The ISL81401 and ISL81401A provide a precision 0.8V internal reference voltage to set the output voltage.
Based on this internal reference, the output voltage can be set from 0.8V up to a level determined by the
feedback voltage divider, as shown in Figure 42 on page 28.
A resistive divider from the output to ground sets the output voltage. Connect the center point of the divider to
the FB_OUT pin. The output voltage value is determined by Equation 5.
where RFBO1 is the top resistor of the feedback divider network and RFBO2 is the bottom resistor connected
from FB_OUT to ground, shown in Figure 42.
VOUT 0.8V
RFBO1 RFBO2
+
RFBO2
---------------------------------------------



=
(EQ. 5)
FN9310 Rev.0.00 Page 28 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
As shown in Figure 42, the RCOMP
, CCOMP1, and CCOMP2 network connected on the Gm1 regulator output COMP
pin is needed to compensate the loop for stable operation. The loop stability can be affected by many different
factors such as VIN, VOUT, load current, switching frequency, inductor value, output capacitance, and the
compensation network on COMP pin. For most applications, 22nF is a good value for CCOMP1. A larger CCOMP1
makes the loop more stable by giving a larger phase margin, but the loop bandwidth is lower. CCOMP2 is typically
1/10th to 1/30th of CCOMP1 to filter high frequency noise. A good starting value for RCOMP is 10k. Lower RCOMP
improves stability but slows the loop response. Optimize the final compensation network with a bench test.
4.5.2 Input Voltage Regulation Loop
As shown in Figure 43, the input voltage VIN can be sensed by the FB_IN pin using a resistor divider
RFBIN1/RFBIN2 and regulated by Gm2. When the FB_IN pin voltage falls below the 0.8V reference voltage, the
COMP pin voltage is pulled low to reduce the PWM duty cycle and the input current. For applications with a
high input source impedance, such as a solar panel, the input voltage regulation loop can prevent the input
voltage from being pulled too low in high output load conditions. For applications with a low input source
impedance, such as batteries, the VIN feedback loop can prevent the battery from being over-discharged. For
applications with loads on the VIN supply, such as a DC back up system, the input voltage regulation loop can
reduce the input current to negative area to reverse power conversion direction to discharge the backup battery
or supper capacitor to supply a regulated VIN for the loads. The regulated input voltage value is determined by
Equation 6.
FB_IN is a dual-function pin. It also sets the phase angle of the clock output signal on the CLKOUT/DITHER
pin, shown in Table 2 on page 35. The VIN feedback loop is disabled when the FB_IN pin voltage is below 0.3V
or above 4.7V. The VIN feedback loop is also disabled in DEM mode and during soft-start.
Figure 42. Output Voltage Regulator
Figure 43. VIN Feedback Loop
COMP
FB_OUT
+
_
+
_0.8V
REF
GM1
VOUT
RFBO1
RFBO2 RCOMP
CCOMP1
CCOMP2
VIN 0.8V
RFBIN1 RFBIN2
+
RFBIN2
------------------------------------------------



=
(EQ. 6)
COMP
FB_IN
+
_
+
_0.8V
REF
GM2
VIN
RFBIN1
RFBIN2
FN9310 Rev.0.00 Page 29 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
4.5.3 Input and Output Average Current Monitoring and Regulation Loops
As shown in Figure 44, the ISL81401 and ISL81401A have two current sense operational amplifiers (op amps),
A1 and A2, which monitor both input and output current. The voltage signals on the input and output current
sense resistor RS_IN and RS_OUT are sent to the differential inputs of CS+/CS- and ISEN+/ISEN-, respectively,
after the RC filters RS_IN1/CS_IN1, RS_IN2/CS_IN2, RS_OUT1/CS_OUT1, and RS_OUT2/CS_OUT2. Renesas
recommends using a 1Ω value for RS_IN1, RS_IN2, RS_OUT1, and RS_OUT2, and a 10nF value for CS_IN1, CS_IN2,
CS_OUT1, and CS_OUT2 to effectively damp the switching noise without delaying the current signal too much
introducing too much error by the op amp bias current. The Gm op amps A1 and A2 then transfer the current
sense voltage signals to current signals ICS and IISEN.
where
•I
IN is the input current in Q1 drain
•V
CSOFFSET is the A1 input offset voltage
•Gm
CS is the gain of A1
•V
CSOFFSET GmCS = ICSOFFSET.
The typical value of ICSOFFSET is 20µA
where
•I
OUT is the output current in Q4 drain
•V
ISENOFFSET is the A2 input offset voltage
•Gm
ISEN is the gain of A2
•V
ISENOFFSET GmISEN = IISENOFFSET.
The typical value of IISENOFFSET is 20µA
Figure 44. Input and Output Average Current Monitoring and Regulation Loops
ICS IIN
RS_IN VCSOFFSET
+GmCS
=
(EQ. 7)
IISEN IOUT
RS_OUT VISENOFFSET
+GmISEN
=
(EQ. 8)
CS+
IMON_OUT
CS-
A1
+
_
COMP
ISEN+ ISEN-
+
_
1.2V
Gm3 Gm4
IMON_IN
Q1
Q3
VIN VOUT
Q2
Q4
RS_IN RS_OUT
L
UG1
LG1
UG2
LG2
PH1 PH2
A2
1.2V
RIM_IN RIM_OUT
RIM_IN1
CIM_IN1
CIM_IN2 CIM_OUT2
CIM_OUT1
RIM_OUT1
RS_IN1 RS_IN2
CS_IN1 CS_IN2 CS_OUT1 CS_OUT2
RS_OUT1 RS_OUT2
VCS_OFFSET VISEN_OFFSET
Ics IISEN
FN9310 Rev.0.00 Page 30 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
By connecting resistor RIM_IN and RIM_OUT on the IMON_IN and IMON_OUT pins, the ICS and IISEN current
signals are transferred to voltage signals. The RC networks on the IMON_IN and IMON_OUT pins
RIM_IN1/CIM_IN1/CIM_IN2 and RIM_OUT1/CIM_OUT1/CIM_OUT2 are needed to remove the AC content in the ICS
and IISEN signals and ensure stable loop operation. The average voltages at the IMON_IN and IMON_OUT
pins are regulated to 1.2V by Gm3 and Gm4 for constant input and output current control.
The input constant current loop set point IINCC is calculated by Equation 9.
where RIMIN is the resistance of RIM_IN.
The output constant current loop set point IOUTCC is calculated by Equation 10.
where RIMOUT is the resistance of RIM_OUT.
Similar to the voltage control loops, the loop stability can be affected by many different factors such as VIN,
VOUT, switching frequency, inductor value, output and input capacitance, and the RC network on the IMON_IN
or IMON_OUT pin. Due to the high AC content in ICS and IISEN, large CIM_IN1 and CIM_OUT1 are needed.
Larger CIM_IN1 and CIM_OUT1 can also make the loop more stable by giving a larger phase margin, but the loop
bandwidth is lower. For most applications, 47nF is a good value for CIM_IN1 and CIM_OUT1. CIM_IN2 and
CIM_OUT2 are typically 1/10th to 1/30th of CIM_IN1 and CIM_OUT1 to filter high frequency noise. RIM_IN1 and
RIM_OUT1 are needed to boost the phase margin. A good starting value for RIM_IN1 and RIM_OUT1 is 5k.
Optimize the final compensation network with iSim simulation and bench testing.
4.6 Buck-Boost Conversion Topology and Control Algorithm
The ISL81401 and ISL81401A use the Renesas proprietary buck-boost control algorithm to achieve optimized
power conversion performance. The buck-boost topology is shown in Figure 45. The ISL81401 and ISL81401A
control the four power switches Q1, Q2, Q3, and Q4 to work in either Buck or Boost mode. When VIN is far lower
than VOUT, the converter works in Boost mode. When VIN is far higher than VOUT, the converter works in Buck
mode. When VIN is equal or close to VOUT, the converter alternates between Buck and Boost mode as necessary to
provide a regulated output voltage, which is called Buck-Boost mode. Figure 46 shows the relationship between
the operation modes and VOUT - VIN.
RS_IN is a current sense resistor to sense the inductor current during Q1 on-time. As shown in the “Block Diagram”
on page 6, the sensed signal is fed into the CS+ and CS- pins and used for peak or valley current-mode control,
DEM control, input average current monitor, constant current control, and protections.
RS_OUT is a current sense resistor to sense the inductor current during Q4 on-time. As shown in the Block Diagram,
the sensed signal is fed into the ISEN+ and ISEN- pins and used for negative peak inductor current limit, output
average current monitor, constant current control, and protections.
Figure 45. Buck-Boost Topology Figure 46. Operation Modes vs VOUT - VIN
IINCC
1.2 ICSOFFSET
–xRIMIN
RIMINxRS_INxGmCS
-------------------------------------------------------------------
=
(EQ. 9)
IOUTCC
1.2 IISENOFFSET
–xRIMOUT
RIMOUTxRS_OUTxGmISEN
-------------------------------------------------------------------------------
=
(EQ. 10)
Q1
Q3
VIN VOUT
Q2
Q4
RS_IN RS_OUT
L
UG1
LG1
UG2
LG2
PH1 PH2
0
Boost Mode
Buck/Boost Mode
Buck Mode
VOUT - VIN
Q1 On, Q2 Off
Q3, Q4 PWM Switching
Q4 On, Q3 Off
Q1, Q2 PWM Switching
4-Switch PWM
Q3 Max Duty
Q3 Min Duty
Q2 Min Duty
Q2 Max Duty
FN9310 Rev.0.00 Page 31 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
4.6.1 Buck Mode Operation (VIN >> VOUT)
In Buck mode, Q4 is always on and Q3 is always off unless boot refresh or inductor negative peak current limit
is tripped. Q1 and Q2 runs in a normal peak current controlled sync buck operation mode. Q1 turns on by the
clock. During Q1 on-time, op amp A1 senses the inductor current by the voltage on RS_IN. Q1 turns off when
the sensed signal combined with the slope compensation ramp is higher than the COMP pin voltage, which is
the error signal from the upper voltage or current regulator. The equivalent circuit and operation waveforms are
shown in Figure 47.
In Buck mode, the Q1 duty cycle is given by:
DQ1 = VOUT / VIN x 100%
As VIN decreases close to VOUT, DQ1 increases close to its maximum value decided by its minimum off-time.
When DQ1 reaches its maximum value, the converter moves to Buck-Boost mode.
When VIN is much higher than VOUT, DQ1 decreases to close to its minimum duty cycle decided by its
minimum on-time. To allow stable loop operation and avoid duty cycle jitter, Renesas recommends keeping the
Q1 on-time always two to three times higher than the minimum on-time.
4.6.2 Boost Mode Operation (VIN << VOUT)
In Boost mode, the converter Q1 is always on and Q2 is always off unless boot refresh or inductor negative
peak current limit is tripped. Q3 and Q4 run in a normal valley current controlled sync boost operation mode.
Q3 turns off by the clock. During Q3 off-time, op amp A1 senses the inductor current by the voltage on RS_IN.
Q3 turns on when the sensed signal combined with the slope compensation ramp is lower than the COMP pin
voltage which is the error signal from the upper voltage or current regulator. The equivalent circuit and
operation waveforms are shown in Figure 48 on page 32.
Figure 47. Buck Mode Equivalent Circuit and Operation Waveforms
CLOCK
Q1
UG1
Q2
LG1
Q3
LG2
Q4
UG2
IL
0V
5V
Q1
VIN VOUT
Q2
RS_IN RS_OUT
L
UG1
LG1
PH1 PH2
FN9310 Rev.0.00 Page 32 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
In Boost mode, the Q3 duty cycle is given by:
DQ3 = (1 - VIN / VOUT) x 100%
As VIN increases close to VOUT, DQ3 decreases close to its minimum value decided by its minimum on-time.
When DQ3 reaches its minimum value, the converter moves to Buck-Boost mode.
When VIN is much lower than VOUT, DQ3 increases close to its maximum duty cycle decided by its minimum
off-time. To allow stable loop operation and avoid duty cycle jitter, Renesas recommends keeping the Q3
off-time always two to three times higher than the minimum off-time.
4.6.3 Buck-Boost Mode Operation (VIN >=< VOUT)
In Buck-Boost mode, the converter runs in one cycle of Buck mode followed by one cycle of Boost mode
operation mode. It takes two clock cycles to finish a full buck-boost period.
When VIN is higher than VOUT, Q3 runs in minimum duty in the Boost mode cycle. Q1 duty cycle DQ1 is
modulated in the buck cycle to keep VOUT in regulation. As VIN increases, DQ1 decreases. When DQ1 decreases
to less than 66.7% of the clock period, the converter moves to Buck mode.
When VIN is lower than VOUT, Q1 runs in maximum duty in the Buck mode cycle. Q3 duty cycle DQ3 is
modulated in the Boost mode cycle to keep VOUT in regulation. As VIN decreases, DQ3 increases. When DQ3
increases to more than 33.3% of the clock period, the converter moves to Boost mode.
Figure 48. Boost Mode Equivalent Circuit and Operation Waveforms
Q3
VIN VOUT
Q4
RS_IN RS_OUT
L
UG2
LG2
PH1 PH2
CLOCK
Q1
UG1
Q2
LG1
Q3
LG2
Q4
UG2
IL
0V
5V
FN9310 Rev.0.00 Page 33 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
4.7 Light-Load Efficiency Enhancement
The ISL81401 and ISL81401A can be set to DEM and Burst mode to improve light-load efficiency by connecting
the MODE pin to VCC5V.
When DEM mode is set, the buck sync FET driven by LG1 and the boost sync FET driven by UG2 are all running
in DEM mode. The inductor current is not allowed to reverse (discontinuous operation) depending on the zero
cross detection reference level VCROSS1 for buck sync FET and VCROSS2 for boost sync FET. At light load
condition, the converter goes into diode emulation. When the load current is less than the level set by
VIMONOUTBSTEN typical 0.84V on the IMON_OUT pin, the part enters Burst mode. Equation 11 sets the Burst
mode operation enter condition.
where (refer to Figure 44 on page 29):
RIMOUT is the resistance of RIM_OUT
ISENOFFSET is the output current sense op amp internal offset current, typical 20µA
GmISEN is the output current sense op amp Gm, typical 195µS.
The part exits Burst mode when the output current increases to higher than the level set by VIMONOUTBSTEX
typical 0.88V on the IMON_OUT pin. Equation 12 sets the Burst mode operation exit condition.
When the part enters Burst mode, the BSTEN pin goes low. To fully avoid any enter/exit chattering, a 4-10MΩ
resistor can be added between BETEN and IMON_OUT pins to further expand the hysteresis.
In Burst mode, an internal window comparator takes control of the output voltage. The comparator monitors the
FB_OUT pin voltage. When the FB_OUT pin voltage is higher than 0.82V, the controller enters Low Power Off
mode. Some of the unnecessary internal circuitries are powered off. When the FB_OUT pin voltage drops to 0.8V,
the controller wakes up and runs in a fixed level peak current controlled D/(1-D) Buck-Boost mode when
VIN -V
OUT < 2V and Buck mode when VIN -VOUT > 2V. In the D/(1-D) Buck-Boost mode, Q1 and Q3 conduct in
D*T period, where D is the duty cycle and T is the switching period. Q2 and Q4 complimentarily conduct in
(1-D)*T period. Q1 and Q3 are turned on by the clock signal and turned off when inductor current rises to the level
that the input current sense op amp input voltage reaches VBST-CS, typical 27mV. After Q1 and Q3 are turned off,
Q2 and Q4 are turned on to pass the energy stored in the inductor to the output until next cycle begins. The output
Figure 49. Buck-Boost Mode Equivalent Circuit and Operation Waveforms
CLOCK
Q1
UG1
Q2
LG1
Q3
LG2
Q4
UG2
IL
Q1
Q3
VIN VOUT
Q2
Q4
RS_IN RS_OUT
L
UG1
LG1
UG2
LG2
PH1 PH2
RIMOUTxI
SENOFFSET IOUTxRS_OUTxGmISEN
+0.84V
(EQ. 11)
RIMOUTxI
SENOFFSET IOUTxRS_OUTxGmISEN
+0.88V
(EQ. 12)
FN9310 Rev.0.00 Page 34 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
voltage increases in the wake up period. When the output reaches 0.82V again, the controller enters into Low
Power Off mode again. When the load current increases, the Low Power Off mode period decreases. When the off
mode period disappears and the load current further increases but still does not meet the Equation 12 exit condition,
the output voltage drops. When the FB_OUT pin voltage drops to 0.78V, the controller exits Burst mode and runs
in normal DEM PWM mode. The voltage error amplifier takes control of the output voltage regulation.
In Low Power Off mode, the CLKEN pin goes low. By connecting the BSTEN and CLKEN pins together in a
multiple chip parallel system, the Burst mode enter/exit and burst on/off controls are all synchronized.
Because the VOUT is controlled by a window comparator in Burst mode, higher than normal low frequency voltage
ripples appear on the VOUT, which can generate audible noise if the inductor and output capacitors are not chosen
properly. Also, the efficiency in D/(1-D) Buck-Boost mode is low. To avoid these drawbacks, the Burst mode can be
disabled by choosing a bigger RIMOUT to set the IMON_OUT pin voltage higher than 0.88V at no load condition,
shown in Equation 13. The part runs in DEM mode only. Pulse Skipping mode can also be implemented to lower the
light load power loss with much lower output voltage ripple as the VOUT is always controlled by the regulator Gm1.
4.8 Prebiased Power-Up
The ISL81401 and ISL81401A have the ability to soft-start with a prebiased output by running in forced DEM
mode during soft-start. The output voltage is not pulled down during prebiased start-up. PWM mode is not active
until the soft-start ramp reaches 90% of the output voltage times the resistive divider ratio. Forced DEM mode is
set again when the SS/TRK pin voltage is pulled to less than 0.3V by either internal or external circuit.
The overvoltage protection function is still alive during soft-start of the DEM operation.
4.9 Frequency Selection
Switching frequency selection is a trade-off between efficiency and component size. Low switching frequency
improves efficiency by reducing MOSFET switching loss. To meet the output ripple and load transient
requirements, operation at a low switching frequency would require larger inductance and output capacitance. The
switching frequency of the ISL81401 and ISL81401A is set by a resistor connected from the RT/SYNC pin to GND
according to Equation 1 on page 10.
The frequency setting curve shown in Figure 50 assists in selecting the correct value for RT.
4.10 Phase Lock Loop (PLL)
The ISL81401 and ISL81401A integrate a high performance PLL. The PLL ensures the wide range of accurate
clock frequency and phase setting. It also easily synchronizes the internal clock to an external clock with the
frequency either lower or higher than the internal setting.
Figure 50. RT vs Switching Frequency fSW
RIMOUTxISENOFFSET 0.88V
(EQ. 13)
0
500
1,000
1,500
2,000
2,500
3,000
050100150200
250
f
SW
(kHz)
R
T
(k:)
FN9310 Rev.0.00 Page 35 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
As shown in Figure 51, an external compensation network of RPLL, CPLL1, and CPLL2 is needed to connect to the
PLL_COMP pin to ensure PLL stable operation. Renesas recommends choosing 2.7kΩ for RPLL, 10nF for CPLL1,
and 820pF for CPLL2. With the recommended compensation network, the PLL stability is ensured in the full clock
frequency range of 100kHz to 600kHz.
4.11 Frequency Synchronization and Dithering
The RT/SYNC pin can synchronize the ISL81401 and ISL81401A to an external clock or the CLKOUT/DITHER
pin of another ISL81401. When the RT/SYNC pin is connected to the CLKOUT/DITHER pin of another
ISL81401, the two controllers operate in cascade synchronization with phase interleaving.
When the RT/SYNC pin is connected to an external clock, the ISL81401 and ISL81401A synchronizes to this
external clock frequency. The frequency set by the RT resistor can be either lower or higher than, or equal to the
external clock frequency.
The CLKOUT/DITHER pin outputs a clock signal with approximately 300ns pulse width. The signal frequency is
the same as the frequency set by the resistor from the RT pin to ground or the external sync clock. The signal rising
edge phase angle to the rising edge of the internal clock or the external clock to the RT/SYNC pin can be set by the
voltage applied to the FB_IN and IMON_IN pins. The phase interleaving can be implemented by the cascade
connecting of the upper chip CLKOUT/DITHER pin to the lower chip RT/SYNC pin in a parallel system. Table 2
shows the CLKOUT/DITHER phase settings with different FB_IN and IMON_IN pin voltages.
When FB_IN is connected to 4.5V, the VIN feedback control loop is disabled. When IMON_IN is connected to 4.5V,
the average input current control loop and input current hiccup OCP are disabled.
In multi-chip cascade parallel operation, the CLKOUT pin of the upstream chip is connected to the RT/SYNC pin of
the downstream chip. Renesas recommends leaving the RT/SYNC pin open for all the slave chips. The FB_IN,
SS/TRK, COMP, FB_OUT, IMON_OUT, EN/UVLO, IMON_IN, and MODE pins of all the paralleled chips should
be tied together. Refer to ISL81601 datasheet for the current sharing approach in parallel operation.
The CLKOUT/DITHER pin provides a dual function option. When a capacitor CDITHER is connected on the
CLKOUT/DITHER pin, the internal circuit disables the CLKOUT function and enables the DITHER function.
When the CLKOUT/DITHER pin voltage is lower than 1.05V, a typical 8µA current source IDITHERSO charges the
capacitor on the pin. When the capacitor voltage is charged to more than 2.2V, a typical 10µA current source
IDITHERSI discharges the capacitor on the pin. A sawtooth voltage waveform shown in Figure 52 is generated on
the CLKOUT/DITHER pin. The internal clock frequency is modulated by the sawtooth voltage on the
CLKOUT/DITHER pin. The clock frequency dither range is set to typically ±15% of the frequency set by the
resistor on the RT/SYNC pin. The dither function is lost when the chip is synchronized to an external clock.
Figure 51. PLL Compensation Network
Table 2. CLKOUT Phase Shift vs FB_IN and IMON_IN Voltage
CLKOUT Phase Shift 120° 90° 60° 180°
FB_IN Voltage Active 1 1 Active
IMON_IN Voltage 1 Active 1 Active
Note: “1” means logic high 4.7V to 5V. “Active” means logic low 0V to 4.3V.
PLL_COMP
CPLL1
RPLL
CPLL2
ISL81401/ISL81401A
FN9310 Rev.0.00 Page 36 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
The dither frequency FDITHER can be calculated by Equation 14. Renesas recommends setting CDITHER between
10nF and 1µF. With a too low CDITHER the part may not be able to set to Dither mode. With a higher CDITHER, the
discharge power loss at disable or power off is higher, leading to a higher thermal stress to the internal discharge
circuit.
4.12 Gate Drivers
The ISL81401 and ISL81401A integrate two almost identical high voltage driver pairs to drive both buck and boost
MOSFET pairs. Each driver pair consists of a gate control logic circuit, a low side driver, a level shifter, and a high
side driver.
The ISL81401 and ISL81401A incorporate an adaptive dead time algorithm that optimizes operation with varying
MOSFET conditions. This algorithm provides approximately 16ns dead time between the switching of the upper and
lower MOSFETs. This dead time is adaptive and allows operation with different MOSFETs without having to
externally adjust the dead time using a resistor or capacitor. During turn-off of the lower MOSFET, the LGATE
voltage is monitored until it reaches a threshold of 1V, at which time the UGATE is released to rise. Adaptive dead
time circuitry monitors the upper MOSFET gate voltage during UGATE turn-off. When the upper MOSFET
gate-to-source voltage drops below a threshold of 1V, the LGATE is allowed to rise. Renesas recommends not using a
resistor between the driver outputs and the respective MOSFET gates, because it can interfere with the dead time
circuitry.
The low-side gate driver is supplied from VDD and provides a 3A peak sink and 2A peak source current. The
high-side gate driver can also deliver the same currents as the low-side gate driver. Gate-drive voltage for the upper
N-channel MOSFET is generated by a flying capacitor boot circuit. A boot capacitor connected from the BOOT
pin to the PHASE node provides power to the high-side MOSFET driver. As shown in Figure 53 on page 37, the
boot capacitor is charged up to VDD by an external Schottky diode during low-side MOSFET on-time (phase node
low). To limit the peak current in the Schottky diode, an external resistor can be placed between the BOOT pin and
the boot capacitor. This small series resistor also damps any oscillations caused by the resonant tank of the parasitic
inductances in the traces of the board and the FET’s input capacitance.
Figure 52. Frequency Dithering Operation
CLKOUT/
DITHER
CDITHER
ISL81401
FDITHER
1
2.2V
1.05V
a. Frequency Dithering Operation
b. CLKOUT/DITHER Pin Voltage Waveform in Dither Operation
FDITHER
3.865x10e 6–
CDITHER
-----------------------------------------
=
(EQ. 14)
FN9310 Rev.0.00 Page 37 of 47
Sep 11, 2018
ISL81401, ISL81401A 4. Functional Description
At start-up, the low-side MOSFET turns on first and forces PHASE to ground to charge the BOOT capacitor to
5.3V if the diode voltage drop is ignored. After the low-side MOSFET turns off, the high-side MOSFET is turned
on by closing an internal switch between BOOT and UGATE. This provides the necessary gate-to-source voltage to
turn on the upper MOSFET, an action that boosts the 5.3V gate drive signal above VIN. The current required to
drive the upper MOSFET is drawn from the internal 5.3V regulator supplied from either VIN or EXTBIAS pin.
The BOOT to PHASE voltage is monitored internally. When the voltage drops to 3.9V at no switching condition, a
minimum off-time pulse is issued to turn off the upper MOSFET and turn on the low-side MOSFET to refresh the
bootstrap capacitor and maintain the upper driver bias voltage.
To optimize EMI performance or reduce phase node ringing, a small resistor can be placed between the BOOT pin
to the positive terminal of the bootstrap capacitor.
4.13 Power-Good Indicator
The power-good pin can monitor the status of the output voltage. PGOOD is true (open drain) 1.1ms after the
FB_OUT pin is within ±10% of the reference voltage.
There is no extra delay when the PGOOD pin is pulled low.
Figure 53. Upper Gate Driver Circuit
BOOT
UGATE
PHASE
VDD VIN
ISL81401/ISL81401A
CB
RBOOT
External
Schottky
FN9310 Rev.0.00 Page 38 of 47
Sep 11, 2018
ISL81401, ISL81401A 5. Protection Circuits
5. Protection Circuits
The converter output and input are monitored and protected against overload, overvoltage, and undervoltage conditions.
5.1 Input Undervoltage Lockout
The ISL81401 and ISL81401A include input UVLO protection, which keeps the devices in a reset condition until a
proper operating voltage is applied. UVLO protection shuts down the ISL81401 and ISL81401A if the input
voltage drops below 3.2V. The controller is disabled when UVLO is asserted. When UVLO is asserted, PGOOD is
valid and is deasserted. If the input voltage rises above 4V, UVLO is deasserted to allow the start-up operation.
5.2 VCC5V Power-On Reset (POR)
The ISL81401 and ISL81401A set their VCC5V POR rising threshold at 4V and falling threshold at 3.5V when
supplied by VIN. EXTBIAS can activate only after VCC5V reaches its POR rising threshold.
5.3 Overcurrent Protection (OCP)
5.3.1 Input and Output Average Overcurrent Protection
As described in “Input and Output Average Current Monitoring and Regulation Loops” on page 29, the
ISL81401 and ISL81401A can regulate both input and output currents with close loop control. This provides a
constant current type of overcurrent protection for both input and output average current. It can be set to a
hiccup type of protection by selecting a different value of the resistor connected between LG2/OC_MODE and
GND.
The input and output constant or hiccup average OCP set points IINCC and IOUTCC can be calculated by
Equations 9 and 10 in Input and Output Average Current Monitoring and Regulation Loops.
The average OCP mode is set by a resistor connected from the LG2/OC_MODE pin to ground during the
initiation stage before soft-start. During the initiation stage, the LG2/OC_MODE pin sources out a typical
10µA current IMODELG2 to set the voltage on the pin. If the pin voltage is less than 0.3V, the OCP is set to
Constant Current-mode. Otherwise, the OCP is set to hiccup mode.
In hiccup OCP mode, after the average current is higher than the set point for 32 consecutive switching cycles
the converter turns off for 50ms before a restart-up is issued.
5.3.2 First Level Pulse-by-Pulse Peak Current Limit
As shown in Figure 44 on page 29 in Input and Output Average Current Monitoring and Regulation Loops, the
inductor peak current is sensed by the shunt resistor RS_IN and op amp A1. When the voltage drop on RS_IN
reaches the set point VOCSET-CS typical 83mV, Q1 is turned off in Buck mode or Q3 is turned off in Boost
mode. The first level peak current limit set point IOCPP1 can be calculated by Equation 15.
IOCPP1
VOCSET CS–
RS_IN
----------------------------------
=
(EQ. 15)
FN9310 Rev.0.00 Page 39 of 47
Sep 11, 2018
ISL81401, ISL81401A 5. Protection Circuits
5.3.3 Second Level Hiccup Peak Current Protection
To avoid any false trip in peak current-mode operation, a minimum on or blanking time is set to the PWM
signal. The first level pulse-by-pulse current limit circuit cannot further reduce the PWM duty cycle in the
minimum on-time. In output dead short conditions, especially at high VIN, the inductor current runs away with
the minimum on PWM duty. The ISL81401 and ISL81401A integrate a second level hiccup type of peak
current protection. When the voltage drop on RS_IN reaches the set point VOCSET-CS-HIC (typical 100mV), the
converter turns off by turning off all four switches Q1, Q2, Q3, and Q4 for 50ms before a restart is issued. The
second level peak current protection set point IOCPP2 can be calculated by Equation 16.
5.3.4 Pulse-by-Pulse Negative Peak Current Limit
In cases of reverse direction operation and OVP protection, the inductor current goes to negative. The negative
current is sensed by the shunt resistor RS_OUT and op amp A2 shown in Figure 44. When the voltage drop on
RS_IN reaches the set point VOCSET-ISEN (typical -59mV), Q2 and Q4 are turned off and Q1 and Q3 are turned
on. The negative peak current limit set point IOCPPN can be calculated by Equation 17.
The device can be damaged in negative peak current limit conditions. In these conditions, the energy flows from
output to input. If the impedance of the input source or devices is not low enough, the VIN voltage increases.
When VIN increases to higher than its maximum limit, the IC can be damaged.
5.4 Overvoltage Protection
The overvoltage set point is set at 114% of the nominal output voltage set by the feedback resistors. In the case of
an overvoltage event, the IC attempts to bring the output voltage back into regulation by keeping Q1 and Q3 turned
off and Q2 and Q4 turned on. If the OV condition continues, the inductor current goes negative to trip the negative
peak current limit. The converter reverses direction to transfer energy from the output end to the input end. Input
voltage is pushed high if the input source impedance is not low enough. The IC may be damaged if the input
voltage goes to higher than its maximum limit. If the overvoltage condition is corrected and the output voltage
drops to the nominal voltage, the controller resumes work in normal PWM switching.
5.5 Over-Temperature Protection
The ISL81401 and ISL81401A incorporate an over-temperature protection circuit that shuts the IC down when a
die temperature of +160°C is reached. Normal operation resumes when the die temperature drops below +145°C
through the initiation of a full soft-start cycle. During OTP shutdown, the IC consumes only 100µA current. When
the controller is disabled, thermal protection is inactive. This helps achieve a very low shutdown current of 5µA.
IOCPP2
VOCSET-CS-HIC
RS_IN
--------------------------------------------
=
(EQ. 16)
IOCPPN
VOCSET-ISEN
RISEN
--------------------------------------
=
(EQ. 17)
FN9310 Rev.0.00 Page 40 of 47
Sep 11, 2018
ISL81401, ISL81401A 6. Layout Guidelines
6. Layout Guidelines
Careful attention to layout requirements is necessary for successful implementation of ISL81401 and ISL81401A
based DC/DC converters. The ISL81401 and ISL81401A switch at a very high frequency, so the switching times are
very short. At these switching frequencies, even the shortest trace has significant impedance. Also, the peak gate drive
current rises significantly in an extremely short time. Transition speed of the current from one device to another causes
voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade
efficiency, generate EMI, and increase device voltage stress and ringing. Careful component selection and proper
Printed Circuit Board (PCB) layout minimize the magnitude of these voltage spikes.
The three sets of critical components in a DC/DC converter using the ISL81401 and ISL81401A are the following:
• the controller
• the switching power components
• the small signal components
The switching power components are the most critical from a layout point of view because they switch a large amount
of energy, which tends to generate a large amount of noise. The critical small signal components are those connected to
sensitive nodes or those supplying critical bias currents. A multilayer PCB is recommended.
6.1 Layout Considerations
(1) Place the input capacitors, buck FETs, inductor, boost FETs, and output capacitor first. Isolate these power
components on dedicated areas of the board with their ground terminals adjacent to one another. Place the
input and output high frequency decoupling ceramic capacitors very close to the MOSFETs.
(2) If signal components and the IC are placed in a separate area to the power train, use full ground planes in the
internal layers with shared SGND and PGND to simplify the layout design. Otherwise, use separate ground
planes for the power ground and the small signal ground. Connect the SGND and PGND together close to the
IC. DO NOT connect them together anywhere else.
(3) Keep the loop formed by the input capacitor, the buck top FET, and the buck bottom FET as small as possible.
Keep the loop formed by the output capacitor, the boost top FET, and the boost bottom FET as small as
possible.
(4) Ensure the current paths from the input capacitor to the buck FETs, the power inductor, the boost FETs, and
the output capacitor are as short as possible with maximum allowable trace widths.
(5) Place the PWM controller IC close to the lower FETs. The low side FETs gate drive connections should be
short and wide. Place the IC over a quiet ground area. Avoid switching ground loop currents in this area.
(6) Place the VDD bypass capacitor very close to the VDD pin of the IC and connect its ground end to the PGND
pin. Connect the PGND pin to the ground plane by a via. Do not directly connect the PGND pin to the SGND
EPAD.
(7) Place the gate drive components (BOOT diodes and BOOT capacitors) together near the controller IC.
(8) Place the output capacitors as close to the load as possible. Use short, wide copper regions to connect output
capacitors to load to avoid inductance and resistances.
(9) Use copper filled polygons or wide short traces to connect the junction of the buck or boost upper FET, buck
or boost lower FET, and output inductor. Also keep the buck and boost PHASE nodes connection to the IC
short. DO NOT oversize the copper islands for the PHASE nodes. Because the phase nodes are subjected to
very high dv/dt voltages, the stray capacitor formed between these islands and the surrounding circuitry tends
to couple switching noise.
(10) Route all high speed switching nodes away from the control circuitry.
(11) Create a separate small analog ground plane near the IC. Connect the SGND pin to this plane. All small signal
grounding paths including feedback resistors, current monitoring resistors and capacitors, soft-starting
capacitors, loop compensation capacitors and resistors, and EN pull-down resistors should be connected to
this SGND plane.
FN9310 Rev.0.00 Page 41 of 47
Sep 11, 2018
ISL81401, ISL81401A 6. Layout Guidelines
(12) Use a pair of traces with minimum loop for the input or output current sensing connection.
(13) Ensure the feedback connection to the output capacitor is short and direct.
6.2 General EPAD Design Considerations
Figure 54 illustrates how to use vias to remove heat from the IC.
Fill the thermal pad area with vias. A typical via array fills the thermal pad footprint so that their centers are three
times the radius apart from each other. Keep the vias small but not so small that their inside diameter prevents
solder wicking through during reflow.
Connect all vias to the ground plane. The vias must have a low thermal resistance for efficient heat transfer. Ensure
a complete connection of the plated through hole to each plane.
Figure 54. PCB Via Pattern
FN9310 Rev.0.00 Page 42 of 47
Sep 11, 2018
ISL81401, ISL81401A 7. Component Selection Guideline
7. Component Selection Guideline
7.1 MOSFET Considerations
The MOSFETs are chosen for optimum efficiency given the potentially wide input voltage range and output power
requirement. Select these MOSFETs based upon rDS(ON), gate supply requirements, and thermal management
considerations.
The buck MOSFETs’ maximum operation voltage is decided by the maximum VIN voltage, and the boost
MOSFETs’ maximum operation voltage is decided by the maximum VOUT voltage. Choose the buck or boost
MOSFETs based on their maximum operation voltage with sufficient margin for safe operation.
The MOSFETs’ power dissipation is based on conduction loss and switching loss. In Buck mode, the power loss
of the buck upper and lower MOSFETs are calculated by Equations 18 and 19. The conduction losses are the main
source of power dissipation for the lower MOSFET. Only the upper MOSFET has significant switching losses,
because the lower device turns on and off into near zero voltage. The equations assume linear voltage current
transitions and do not model power loss due to the reverse recovery of the lower MOSFET’s body diode.
In Boost mode, there is only conduction loss on the buck upper MOSFET calculated by Equation 20.
In Boost mode, the boost upper and lower MOSFETs power loss are calculated by Equations 21 and 22. The
conduction losses are the main component of power dissipation for the upper MOSFET. Only the lower MOSFET
has significant switching losses, because the upper device turns on and off into near zero voltage. The equations
assume linear voltage current transitions and do not model power loss due to the reverse recovery of the upper
MOSFET’s body diode.
In Buck mode, the conduction loss exists on the boost upper MOSFET calculated by Equation 23.
A large gate-charge increases the switching time, tSW, which increases the switching losses of the buck upper and
boost lower MOSFETs. Ensure that all four MOSFETs are within their maximum junction temperature at high
ambient temperature by calculating the temperature rise according to package thermal resistance specifications.
PUPPERBUCK
IOUT
2
rDS ON
VOUT

VIN
-----------------------------------------------------------------------IOUT
VIN
tSW
fSW

2
-----------------------------------------------------------------
+=
(EQ. 18)
PLOWERBUCK
IOUT
2
rDS ON
VIN VOUT
–
VIN
---------------------------------------------------------------------------------------
=
(EQ. 19)
PUPPERBUCK
IOUT
2
VOUT
2

VIN
2

----------------------------------------------rDS ON
=
(EQ. 20)
PLOWERBOOST
IOUT
2
VOUT
2

VIN
2

----------------------------------------------VOUT VIN
–rDS ON

VOUT
----------------------------------------------------------------- IOUT
VOUT
2
tSW
fSW

2V
IN

---------------------------------------------------------------------------
+=
(EQ. 21)
PUPPERBOOST
IOUT
2
rDS ON
VOUT

VIN
-----------------------------------------------------------------------
=
(EQ. 22)
PUPPERBOOST IOUT
2
rDS ON
=
(EQ. 23)
FN9310 Rev.0.00 Page 43 of 47
Sep 11, 2018
ISL81401, ISL81401A 7. Component Selection Guideline
7.2 Inductor Selection
The inductor is selected to meet the output voltage ripple requirements. The inductor value determines the
converter’s ripple current, and the ripple voltage is a function of the ripple current and the output capacitor(s) ESR.
The ripple voltage expression is given in the capacitor selection section and the ripple current is approximated by
Equation 24 for Buck mode and Equation 25 for Boost mode.
The ripple current ratio is usually 30% to 70% of the inductor average current at the full output load condition.
7.3 Output Capacitor Selection
In general, select the output capacitors to meet the dynamic regulation requirements including ripple voltage and
load transients. Selection of output capacitors is also dependent on the inductor, so some inductor analysis is
required to select the output capacitors.
One of the parameters limiting the converter’s response to a load transient is the time required for the inductor
current to slew to its new level. The ISL81401 and ISL81401A provide either 0% or maximum duty cycle in
response to a load transient.
The response time is the time interval required to slew the inductor current from an initial current value to the load
current level. During this interval, the difference between the inductor current and the transient current level must
be supplied by the output capacitor(s). Minimizing the response time can minimize the output capacitance required.
Also, if the load transient rise time is slower than the inductor response time, as in a hard drive or CD drive, it
reduces the requirement on the output capacitor.
The maximum capacitor value required to provide the full, rising step, transient load current during the response
time of the inductor is shown in Equation 26 for Buck mode and Equation 27 for Boost mode:
where COUT is the output capacitor(s) required, L is the inductor, ITRAN is the transient load current step, VIN is the
input voltage, VOUT is output voltage, and DVOUT is the drop in output voltage allowed during the load transient.
High frequency capacitors initially supply the transient current and slow the load rate of change seen by the bulk
capacitors. The bulk filter capacitor values are generally determined by the Equivalent Series Resistance (ESR) and
voltage rating requirements as well as actual capacitance requirements.
In Buck mode, the output voltage ripple is due to the inductor ripple current and the ESR of the output capacitors as
defined by Equation 28:
where ILBuck is calculated in Equation 24.
ILBuck
VIN VOUT
–VOUT

fSW
LVIN

----------------------------------------------------------
=
(EQ. 24)
ILBoost
VOUT VIN
–VIN

fSW
LVOUT

----------------------------------------------------
=
(EQ. 25)
COUTBuck
LITRAN

2
2V
IN VOUT
–DVOUT

------------------------------------------------------------------
=
(EQ. 26)
COUTBoost
LVOUT
ITRAN

2
2V
IN
2
DVOUT

-------------------------------------------------------
=
(EQ. 27)
VRIPPLE ILBuck ESR=
(EQ. 28)
FN9310 Rev.0.00 Page 44 of 47
Sep 11, 2018
ISL81401, ISL81401A 7. Component Selection Guideline
In Boost mode, the current to the output capacitor is not continuous. The output voltage ripple is much higher as
defined by Equation 29:
where ILBoost is calculated in Equation 25 on page 43.
Place high frequency decoupling capacitors as close to the power pins of the load as physically possible. Be careful
not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load circuitry for specific decoupling requirements.
Use only specialized low-ESR capacitors intended for switching regulator applications for the bulk capacitors. In most
cases, multiple small case electrolytic capacitors perform better than a single large case capacitor.
The stability requirement on the selection of the output capacitor is that the ESR zero (fZ) is between 2kHz and
60kHz. The ESR zero can help increase phase margin of the control loop.
This requirement is shown in Equation 30:
In conclusion, the output capacitors must meet the following criteria:
• They must have sufficient bulk capacitance to sustain the output voltage during a load transient while the output
inductor current is slewing to the value of the load transient.
• The ESR must be sufficiently low to meet the desired output voltage ripple due to the supplied ripple current.
• The ESR zero should be placed in a large range to provide additional phase margin.
7.4 Input Capacitor Selection
The important parameters for the input capacitor(s) are the voltage rating and the RMS current rating. For reliable
operation, select input capacitors with voltage and current ratings above the maximum input voltage and largest
RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the
maximum input voltage and 1.5 times is a conservative guideline. In Buck mode the AC RMS input current varies
with the load giving in Equation 31:
where DC is duty cycle.
The maximum RMS current supplied by the input capacitance occurs at VIN = 2 X VOUT, DC = 50% as shown in
Equation 32:
In Boost mode, the input current is continuous. The RMS current supplied by the input capacitance is much
smaller.
Use a mix of input bypass capacitors to control the voltage ripple across the MOSFETs. Use ceramic capacitors for
the high frequency decoupling and bulk capacitors to supply the RMS current. Small ceramic capacitors can be
placed very close to the MOSFETs to suppress the voltage induced in the parasitic circuit impedances.
Solid tantalum capacitors can be used, but use caution with regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge current at power-up.
VRIPPLE
IOUT
VOUT

VIN
----------------------------------------ILBoost
2
------------------------
+



ESR=
(EQ. 29)
COUT
1
2ESRfZ

------------------------------------
=
(EQ. 30)
IRMS DC DC2
–IOUT
=
(EQ. 31)
IRMS
1
2
---IOUT
=
(EQ. 32)
FN9310 Rev.0.00 Page 45 of 47
Sep 11, 2018
ISL81401, ISL81401A 8. Revision History
8. Revision History
Rev. Date Description
0.00 Sep 11, 2018 Initial release
FN9310 Rev.0.00 Page 46 of 47
Sep 11, 2018
ISL81401, ISL81401A 9. Package Outline Drawing
9. Package Outline Drawing
L32.5x5B
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 3, 5/10
located within the zone indicated. The pin #1 identifier may be
Unless otherwise specified, tolerance : Decimal ± 0.05
Tiebar shown (if present) is a non-functional feature.
The configuration of the pin #1 identifier is optional, but must be
between 0.15mm and 0.30mm from the terminal tip.
Dimension applies to the metallized terminal and is measured
Dimensions in ( ) for Reference Only.
Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
6.
either a mold or mark feature.
3.
5.
4.
2.
Dimensions are in millimeters.1.
NOTES:
BOTTOM VIEW
DETAIL "X"
SIDE VIEW
TYPICAL RECOMMENDED LAND PATTERN
TOP VIEW
5.00 A
5.00
B
INDEX AREA
PIN 1
6
(4X) 0.15
32X 0.40 ± 0.10 4
A
32X 0.23
M0.10 C B
16 9
4X
0.50
28X
3.5
6
PIN #1 INDEX AREA
3 .30 ± 0 . 15
0 . 90 ± 0.1
BASE PLANE
SEE DETAIL "X"
SEATING PLANE
0.10 C
C
0.08 C
0 . 2 REF
C
0 . 05 MAX.
0 . 00 MIN.
5
( 3. 30 )
( 4. 80 TYP ) ( 28X 0 . 5 )
(32X 0 . 23 )
( 32X 0 . 60)
+ 0.07
- 0.05
17
25
24
8
1
32
For the most recent package outline drawing, see L32.5x5B.
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Refer to "http://www.renesas.com/" for the latest and detailed information.
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