LM4917SDBD - National Semiconductor

LM4917

Ground-Referenced, 95mW Stereo Headphone Amplifier

General Description

The LM4917 is a stereo, output capacitor-less headphone

amplifier capable of delivering 95mW of continuous average

power into a 16Ωload with less than 1% THD+N from a

single 3V power supply.

The LM4917 provides high quality audio reproduction with

minimal external components. A ground referenced output

eliminates the output coupling capacitors typically required

by single-ended loads, reducing component count, cost and

board space consumption. This makes the LM4917 ideal for

mobile phones and other portable equipment where board

space is at a premium. Eliminating the output coupling ca-

pacitors also improves low frequency response.

The LM4917 operates from a single 1.4V to 3.6V power

supply, features low 0.02% THD+N and 70dB PSRR. Inde-

pendent right/left channel low-power shutdown controls pro-

vide power saving flexibility for mono/stereo applications.

Superior click and pop suppression eliminates audible tran-

sients during start up and shutdown. Short circuit and ther-

mal overload protection protects the device during fault con-

ditions.

Key Specifications

jImproved PSRR at 1kHz 70dB (typ)

jPower Output at V

= 3V,

=16Ω, THD %1% 95mW (typ)

jShutdown Current 0.01µA (typ)

Features

nGround referenced outputs

nHigh PSRR

nAvailable in space-saving TSSOP package

nUltra low current shutdown mode

nImproved pop & click circuitry eliminates noises during

turn-on and turn-off transitions

n1.4 – 3.6V operation

nNo output coupling capacitors, snubber networks,

bootstrap capacitors

nShutdown either channel independently

Applications

nNotebook PCs

nDesktop PCs

nMobile Phone

nPDAs

nPortable electronic devices

Block Diagram

Boomer®is a registered trademark of National Semiconductor Corporation.

200893B8

FIGURE 1. Circuit Block Diagram

November 2004

LM4917 Ground-Referenced, 95mW Stereo Headphone Amplifier

Typical Application

200893B2

FIGURE 2. Typical Audio Amplifier Application Circuit

LM4917

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Connection Diagrams

TSSOP Package TSSOP Marking

200893A4

Top View

Order Number LM4917MT

See NS Package Number MTC14

200893B7

Z - Assembly Plant Code

XY - Date Code

TT - Traceability

LLP Package LLP Marking

200893C0

Top View

Order Number LM4917SD

See NS Package Number SDA14A

200893C4

Z - Assembly Plant Code

XY - Date Code

TT - Traceability

Pin Descriptions

Pin Name Function

1 SD_LC Active_Low Shutdown, Left Channel

2CP

VDD

Charge Pump Power Supply

CP+

Positive Terminal-Charge Pump Flying Capacitor

4 PGND Power Ground

CP-

Negative Terminal- Charge Pump Flying Capacitor

CP_OUT

Charge Pump Output

7-AV

Negative Power Supply-Amplifier

8 L_OUT Left Channel Output

9AV

Positive Power Supply-Amplifier

10 L_IN Left Channel Input

11 R_OUT Right Channel Output

12 SD_RC Active_Low Shutdown, Right Channel

13 R_IN Right Channel Input

14 SGND Signal Ground

LM4917

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Absolute Maximum Ratings (Note 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage 4.0V

Storage Temperature −65˚C to +150˚C

Input Voltage -0.3V to V

+ 0.3V

Power Dissipation (Note 3) Internally Limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150˚C

Thermal Resistance

(TSSOP) 40˚C/W

(TSSOP) 109˚C/W

Operating Ratings

Temperature Range

MIN

≤T

MAX

−40˚C ≤T

≤85˚C

Supply Voltage (V

) 1.4V ≤V

≤3.6V

Electrical Characteristics V

=3V(Notes 1, 2)

The following specifications apply for V

= 3V, A

= 1, and 16Ωload unless otherwise specified. Limits apply to T

= 25˚C.

Symbol Parameter Conditions LM4917 Units

(Limits)

Typ

(Note 6)

Limit

(Notes 7,

Quiescent Power Supply Current V

= 0V, I

= 0A, both channels

enabled

11 20 mA (max)

= 0V, I

= 0A, one channel

enabled

9mA

Shutdown Current V

SD_LC

SD_RC

= GND 0.01 1 µA (max)

Output Offset Voltage R

=32Ω1 10 mV (max)

Output Power THD+N = 1% (max); f = 1kHz, R

16Ω

95 50 mW (min)

THD+N = 1% (max); f = 1kHz, R

32Ω

82 mW

THD+N Total Harmonic Distortion + Noise P

= 50mW, f = 1kHz, R

=32Ω

(A-weighted) single channel

0.02 %

PSRR Power Supply Rejection Ratio

RIPPLE

= 200mV sine p-p,

f = 1kHz

f = 20kHz

SNR Signal-to-Noise Ratio R

=32Ω,P

OUT

= 20mW, f = 1kHz 100 dB

Shutdown Input Voltage High V

0.7*CPV

V (min)

Shutdown Input Voltage Low V

0.3*CPV

V (max)

Wake Up Time From Shutdown 339 µs (max)

TALK

Crosstalk R

=16Ω,P

= 1.6mW, f = 1kHz 70 dB

Input Leakage Current ±0.1 nA

Note 1: All voltages are measured with respect to the GND pin unless otherwise specified.

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions that

guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit

is given; however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX,θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX =(T

JMAX -T

A)/θJA or the number given in Absolute Maximum Ratings, whichever is lower. For the LM4917, see power

de-rating currents for more information.

Note 4: Human body model, 100pF discharged through a 1.5kΩresistor.

Note 5: Machine Model, 220pF-240pF discharged through all pins.

Note 6: Typicals are measured at 25˚C and represent the parametric norm.

Note 7: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: If the product is in shutdown mode and VDD exceeds 3.6V (to a max of 4V VDD) then most of the excess current will flow through the ESD protection circuits.

If the source impedance limits the current to a max of 10mA, then the part will be protected. If the part is enabled when VDD is above 4V circuit performance will

be curtailed or the part may be permanently damaged.

Note 10: Human body model, 100pF discharged through a 1.5kΩresistor.

LM4917

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External Components Description (Figure 1)

Components Functional Description

1. R

Inverting input resistance which sets the closed-loop gain in conjunction with R

. This resistor also forms a

high-pass filter with C

at f

=1/(2πR

2. C

Input coupling capacitor which blocks the DC voltage at the amplifier’s input terminals. Also creates a

high-pass filter with R

at f

=1/(2πR

). Refer to the section Proper Selection of External Components,

for an explanation of how to determine the value of C

3. R

Feedback resistance which sets the closed-loop gain in conjunction with R

4. C

Flying capacitor. Low ESR ceramic capacitor (≤100mΩ)

5. C

Output capacitor. Low ESR ceramic capacitor (≤100mΩ)

6. C

Tantalum capacitor. Supply bypass capacitor which provides power supply filtering. Refer to the Power Supply

Bypassing section for information concerning proper placement and selection of the supply bypass capacitor.

7. C

Ceramic capacitor. Supply bypass capacitor which provides power supply filtering. Refer to the Power Supply

Bypassing section for information concerning proper placement and selection of the supply bypass capacitor.

LM4917

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Typical Performance Characteristics

THD+N vs Frequency

= 1.4V, R

=32Ω,P

= 1mW

THD+N vs Frequency

= 1.8V, R

=16Ω,P

= 5mW

20089341 20089339

THD+N vs Frequency

= 1.8V, R

=32Ω,P

= 5mW

THD+N vs Frequency

= 1.8V, R

=32Ω,P

= 10mW

20089338 20089348

THD+N vs Frequency

= 3.0V, R

=16Ω,P

= 10mW

THD+N vs Frequency

= 3.0V, R

=16Ω,P

= 25mW

20089336 20089334

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Typical Performance Characteristics (Continued)

THD+N vs Frequency

= 3.0V, R

=16Ω,P

= 50mW

THD+N vs Frequency

= 3.0V, R

=32Ω,P

= 5mW

20089333 20089332

THD+N vs Frequency

= 3.0V, R

=32Ω,P

= 10mW

THD+N vs Frequency

= 3.0V, R

=32Ω,P

= 25mW

20089331 20089328

Gain Flatness vs Frequency

= 20kΩ,C

= 0.39µF

Output Power vs Supply Voltage

=16Ω

20089354 20089347

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Typical Performance Characteristics (Continued)

Output Power vs Supply Voltage

=32Ω

PSRR vs Frequency

= 1.8V, R

=16Ω

200893C6 20089345

PSRR vs Frequency

= 1.8V, R

=32Ω

PSRR vs Frequency

= 3.0V, R

=16Ω

20089344 20089343

PSRR vs Frequency

= 3.0V, R

=32Ω

THD+N vs Output Power

= 1.4V, R

=32Ω, f = 1kHz

20089342 20089327

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Typical Performance Characteristics (Continued)

THD+N vs Output Power

= 1.8V, R

=16Ω, f = 1kHz

THD+N vs Output Power

= 1.8V, R

=32Ω, f = 1kHz

20089326 20089325

THD+N vs Output Power

= 3.0V, R

=16Ω, f = 1kHz

THD+N vs Output Power

= 3.0V, R

=32Ω, f = 1kHz

20089324 20089322

Power Dissipation vs Output Power

= 1.8V, R

=16Ω

Power Dissipation vs Output Power

= 1.8V, R

=32Ω

20089349 20089350

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Typical Performance Characteristics (Continued)

Power Dissipation vs Output Power

= 3V, R

=16Ω

Power Dissipation vs Output Power

= 3V, R

=32Ω

20089351 20089352

Supply Current vs Supply Voltage

20089353

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Application Information

ELIMINATING THE OUTPUT COUPLING CAPACITOR

The LM4917 features a low noise inverting charge pump that

generates an internal negative supply voltage. This allows

the outputs of the LM4917 to be biased about GND instead

of a nominal DC voltage, like traditional headphone amplifi-

ers. Because there is no DC component, the large DC

blocking capacitors (typically 220µF) are not necessary. The

coupling capacitors are replaced by two, small ceramic

charge pump capacitors, saving board space and cost.

Eliminating the output coupling capacitors also improves low

frequency response. The headphone impedance and the

output capacitor form a high pass filter that not only blocks

the DC component of the output, but also attenuates low

frequencies, impacting the bass response. Because the

LM4917 does not require the output coupling capacitors, the

low frequency response of the device is not degraded by

external components.

In addition to eliminating the output coupling capacitors, the

ground referenced output nearly doubles the available dy-

namic range of the LM4917 when compared to a traditional

headphone amplifier operating from the same supply volt-

age.

OUTPUT TRANSIENT (’CLICK AND POPS’)

ELIMINATED

The LM4917 contains advanced circuitry that virtually elimi-

nates output transients (’clicks and pops’). This circuitry

prevents all traces of transients when the supply voltage is

first applied or when the part resumes operation after coming

out of shutdown mode.

To ensure optimal click and pop performance under low gain

configurations (less than 0dB), it is critical to minimize the

RC combination of the feedback resistor R

and stray input

capacitance at the amplifier inputs. A more reliable way to

lower gain or reduce power delivered to the load is to place

a current limiting resistor in series with the load as explained

in the Minimizing Output Noise / Reducing Output Power

section.

AMPLIFIER CONFIGURATION EXPLANATION

As shown in Figure 2, the LM4917 has two operational

amplifiers internally. The two amplifiers have externally con-

figurable gain, and the closed loop gain is set by selecting

the ratio of R

to R

. Consequently, the gain for each channel

of the IC is

= -(R

)

Since this an output ground-referenced amplifier, by driving

the headphone through R

OUT

(Pin 11) and L

OUT

(Pin 8), the

LM4917 does not require output coupling capacitors. The

typical single-ended amplifier configuration where one side

of the load is connected to ground requires large, expensive

output capacitors.

POWER DISSIPATION

Power dissipation is a major concern when using any power

amplifier and must be thoroughly understood to ensure a

successful design. Equation 1 states the maximum power

dissipation point for a single-ended amplifier operating at a

given supply voltage and driving a specified output load.

DMAX

=(V

)

/(2π

) (1)

Since the LM4917 has two operational amplifiers in one

package, the maximum internal power dissipation point is

twice that of the number which results from Equation 1. Even

with the large internal power dissipation, the LM4917 does

not require heat sinking over a large range of ambient tem-

perature. From Equation 1, assuming a 3V power supply and

a16Ωload, the maximum power dissipation point is 28mW

per amplifier. Thus the maximum package dissipation point

is 56mW. The maximum power dissipation point obtained

must not be greater than the power dissipation that results

from Equation 2:

DMAX

=(T

JMAX

-T

)/(θ

) (2)

For package TSSOP, θ

= 109˚C/W. T

JMAX

= 150˚C for the

LM4917. Depending on the ambient temperature, T

,ofthe

system surroundings, Equation 2 can be used to find the

maximum internal power dissipation supported by the IC

packaging. If the result of Equation 1 is greater than that of

Equation 2, then either the supply voltage must be de-

creased, the load impedance increased or T

reduced. For

the typical application of a 3V power supply, with a 16Ωload,

the maximum ambient temperature possible without violating

the maximum junction temperature is approximately 119.9˚C

provided that device operation is around the maximum

power dissipation point. Power dissipation is a function of

output power and thus, if typical operation is not around the

maximum power dissipation point, the ambient temperature

may be increased accordingly. Refer to the Typical Perfor-

mance Characteristics curves for power dissipation infor-

mation for lower output powers.

POWER SUPPLY BYPASSING

As with any power amplifier, proper supply bypassing is

critical for low noise performance and high power supply

rejection. Applications that employ a 3V power supply typi-

cally use a 4.7µF in parallel with a 0.1µF ceramic filter

capacitors to stabilize the power supply’s output, reduce

noise on the supply line, and improve the supply’s transient

response. However, their presence does not eliminate the

need for a local 0.1µF supply bypass capacitor, C

, con-

nected between the LM4917’s supply pins and ground. Keep

the length of leads and traces that connect capacitors be-

tween the LM4917’s power supply pin and ground as short

as possible.

MICRO POWER SHUTDOWN

The voltage applied to the SD_LC (shutdown left channel)

pin and the SD_RC (shutdown right channel) pin controls the

LM4917’s shutdown function. When active, the LM4917’s

micropower shutdown feature turns off the amplifiers’ bias

circuitry, reducing the supply current. The trigger point is

0.3*CPV

for a logic-low level, and 0.7*CPV

for logic-

high level. The low 0.01µA(typ) shutdown current is achieved

by appling a voltage that is as near as ground a possible to

the SD_LC/SD_RC pins. A voltage that is higher than ground

may increase the shutdown current.

There are a few ways to control the micro-power shutdown.

These include using a single-pole, single-throw switch, a

microprocessor, or a microcontroller. When using a switch,

connect an external 100kΩpull-up resistor between the

SD_LC/SD_RC pins and V

. Connect the switch between

the SD_LC/SD_RC pins and ground. Select normal amplifier

LM4917

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Application Information (Continued)

operation by opening the switch. Closing the switch con-

nects the SD_LC/SD_RC pins to ground, activating micro-

power shutdown. The switch and resistor guarantee that the

SD_LC/SD_RC pins will not float. This prevents unwanted

state changes. In a system with a microprocessor or micro-

controller, use a digital output to apply the control voltage to

the SD_LC/SD_RC pins. Driving the SD_LC/SD_RC pins

with active circuitry eliminates the pull-up resistor.

SELECTING PROPER EXTERNAL COMPONENTS

Optimizing the LM4917’s performance requires properly se-

lecting external components. Though the LM4917 operates

well when using external components with wide tolerances,

best performance is achieved by optimizing component val-

ues.

The LM4917 is unity-gain stable, giving a designer maximum

design flexibility. The gain should be set to no more than a

given application requires. This allows the amplifier to

achieve minimum THD+N and maximum signal-to-noise ra-

tio. These parameters are compromised as the closed-loop

gain increases. However, low gain demands input signals

with greater voltage swings to achieve maximum output

power. Fortunately, many signal sources such as audio

CODECs have outputs of 1V

RMS

(2.83V

P-P

). Please refer to

the Audio Power Amplifier Design section for more infor-

mation on selecting the proper gain.

Charge Pump Capacitor Selection

Choose low ESR (<100mΩ) ceramic capacitors for optimum

performance. Low ESR capacitors keep the charge pump

output impedance to a minimum, extending the headroom

on the negative supply. Choose capacitors with an X7R

dielectric for best performance over temperature.

Charge pump load regulation and output resistance is af-

fected by the value of the flying capacitor (C1). A larger

valued C1 improves load regulation and minimizes charge

pump output resistance. The switch on-resistance and ca-

pacitor ESR dominates the output resistance for capacitor

values above 2.2µF.

The output ripple is affected by the value and ESR of the

output capacitor (C2). Larger valued capacitors reduce out-

put ripple on the negative power supply. Lower ESR capaci-

tors minimizes the output ripple and reduces the output

resistance of the charge pump.

Input Capacitor Value Selection

Amplifying the lowest audio frequencies requires high value

input coupling capacitor (C

in Figure 2). A high value capaci-

tor can be expensive and may compromise space efficiency

in portable designs. In many cases, however, the speakers

used in portable systems, whether internal or external, have

little ability to reproduce signals below 150Hz. Applications

using speakers with this limited frequency response reap

little improvement by using high value input and output ca-

pacitors.

Besides affecting system cost and size, C

has an effect on

the LM4917’s click and pop performance. The magnitude of

the pop is directly proportional to the input capacitor’s size.

Thus, pops can be minimized by selecting an input capacitor

value that is no higher than necessary to meet the desired

−3dB frequency.

As shown in Figure 2, the input resistor, R

and the input

capacitor, C

, produce a -3dB high pass filter cutoff frequency

that is found using Equation (3).

i-3dB

=1/2πR

(3)

Also, careful consideration must be taken in selecting a

certain type of capacitor to be used in the system. Different

types of capacitors (tantalum, electrolytic, ceramic) have

unique performance characteristics and may affect overall

system performance.

AUDIO POWER AMPLIFIER DESIGN

Design a Dual 90mW/16ΩAudio Amplifier

Given:

Power Output 90mW

Load Impedance 16Ω

Input Level 1Vrms (max)

Input Impedance 20kΩ

Bandwidth 100Hz–20kHz ±0.50dB

The design begins by specifying the minimum supply voltage

necessary to obtain the specified output power. One way to

find the minimum supply voltage is to use the Output Power

vs Supply Voltage curve in the Typical Performance Char-

acteristics section. Another way, using Equation (5), is to

calculate the peak output voltage necessary to achieve the

desired output power for a given load impedance. For a

single-ended application, the result is Equation (5).

(4)

≥[2V

OPEAK

+(V

DOTOP

DOBOT

)] (5)

The Output Power vs Supply Voltage graph for a 16Ωload

indicates a minimum supply voltage of 3.1V. This is easily

met by the commonly used 3.3V supply voltage. The addi-

tional voltage creates the benefit of headroom, allowing the

LM4917 to produce peak output power in excess of 90mW

without clipping or other audible distortion. The choice of

supply voltage must also not create a situation that violates

maximum power dissipation as explained above in the

Power Dissipation section. Remember that the maximum

power dissipation point from Equation (1) must be multiplied

by two since there are two independent amplifiers inside the

package. Once the power dissipation equations have been

addressed, the required gain can be determined from Equa-

tion (6).

(6)

Thus, a minimum gain of 1.2 allows the LM4917 to reach full

output swing and maintain low noise and THD+N perfro-

mance. For this example, let A

= 1.5.

The amplifiers overall gain is set using the input (R

) and

feedback (R

) resistors. With the desired input impedance

set at 20kΩ, the feedback resistor is found using Equation

(7).

(7)

LM4917

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Application Information (Continued)

The value of R

is 30kΩ.

The last step in this design is setting the amplifier’s −3db

frequency bandwidth. To achieve the desired ±0.25dB pass

band magnitude variation limit, the low frequency response

must extend to at lease one−fifth the lower bandwidth limit

and the high frequency response must extend to at least five

times the upper bandwidth limit. The gain variation for both

response limits is 0.17dB, well within the ±0.25dB desired

limit. The results are

= 100Hz/5=20Hz (8)

and

= 20kHzx5=100kHz (9)

As stated in the External Components section, both R

conjunction with C

, and R

, create first order highpass fil-

ters. Thus to obtain the desired low frequency response of

100Hz within ±0.5dB, both poles must be taken into consid-

eration. The combination of two single order filters at the

same frequency forms a second order response. This results

in a signal which is down 0.34dB at five times away from the

single order filter −3dB point. Thus, a frequency of 20Hz is

used in the following equations to ensure that the response

is better than 0.5dB down at 100Hz.

≥1/(2π*20kΩ*20Hz) = 0.397µF; use 0.39µF (10)

The high frequency pole is determined by the product of the

desired high frequency pole, f

, and the closed-loop gain,

. With a closed-loop gain of 1.5 and f

= 100kHz, the

resulting GBWP = 150kHz which is much smaller than the

LM4917’s GBWP of 3MHz. This figure displays that if a

designer has a need to design an amplifier with a higher

gain, the LM4917 can still be used without running into

bandwidth limitations.

LM4917

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Application Information (Continued)

LM4917 SO DEMO BOARD ARTWORK

Top Overlay Top Layer

20089304 20089305

Bottom Layer

20089321

LM4917

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Application Information (Continued)

LM4917 LLP DEMO BOARD ARTWORK

Top Overlay Top Layer

200893C3

200893C2

Bottom Layer

200893C1

LM4917

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Application Information (Continued)

LM4917 REFERENCE DESIGN BOARDS

BILL OF MATERIALS

Part Description Qty Ref Designator

LM4917 Mono Reference Design Board 1

LM4917 Audio AMP 1 U1

Tantalum Cap 1µF 16V 10 1 Cs

Ceramic Cap 0.39µF 50V Z50 20 2 Ci

Resistor 20kΩ1/10W 5 4 Ri, Rf

Resistor 100kΩ1/10W 5 1 Rpu

Jumper Header Vertical Mount 2X1, 0.100 1 J1

PCB LAYOUT GUIDELINES

This section provides practical guidelines for mixed signal

PCB layout that involves various digital/analog power and

ground traces. Designers should note that these are only

"rule-of-thumb" recommendations and the actual results will

depend heavily on the final layout.

Minimization of THD

PCB trace impedance on the power, ground, and all output

traces should be minimized to achieve optimal THD perfor-

mance. Therefore, use PCB traces that are as wide as

possible for these connections. As the gain of the amplifier is

increased, the trace impedance will have an ever increasing

adverse affect on THD performance. At unity-gain (0dB) the

parasitic trace impedance effect on THD performance is

reduced but still a negative factor in the THD performance of

the LM4917 in a given application.

GENERAL MIXED SIGNAL LAYOUT

RECOMMENDATION

Power and Ground Circuits

For two layer mixed signal design, it is important to isolate

the digital power and ground trace paths from the analog

power and ground trace paths. Star trace routing techniques

(bringing individual traces back to a central point rather than

daisy chaining traces together in a serial manner) can

greatly enhance low level signal performance. Star trace

routing refers to using individual traces to feed power and

ground to each circuit or even device. This technique will

require a greater amount of design time, but will not increase

the final price of the board. The only extra parts required may

be some jumpers.

Single-Point Power / Ground Connections

The analog power traces should be connected to the digital

traces through a single point (link). A "PI-filter" can be helpful

in minimizing high frequency noise coupling between the

analog and digital sections. Further, place digital and analog

power traces over the corresponding digital and analog

ground traces to minimize noise coupling.

Placement of Digital and Analog Components

All digital components and high-speed digital signal traces

should be located as far away as possible from analog

components and circuit traces.

Avoiding Typical Design / Layout Problems

Avoid ground loops or running digital and analog traces

parallel to each other (side-by-side) on the same PCB layer.

When traces must cross over each other do it at 90 degrees.

Running digital and analog traces at 90 degrees to each

other from the top to the bottom side as much as possible will

minimize capacitive noise coupling and cross talk.

LM4917

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Physical Dimensions inches (millimeters) unless otherwise noted

TSSOP

Order Number LM4917MT

NS Package Number MTC14

LLP

Order Number LM4917SD

NS Package Number SDA14A

LM4917

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Notes

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves

the right at any time without notice to change said circuitry and specifications.

For the most current product information visit us at www.national.com.

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LM4917 Ground-Referenced, 95mW Stereo Headphone Amplifier