HCPL-7800#500 - Agilent / Hewlett-Packard

Isolation Amplifier

Technical Data

Features

• 15 kV/µs Common-Mode

Rejection at VCM = 1000 V

• Compact, Auto-Insertable

Standard 8-pin DIP Package

• 0.00025 V/V/°C Gain Drift

vs. Temperature

• 0.3 mV Input Offset Voltage

• 100 kHz Bandwidth

• 0.004% Nonlinearity

• Worldwide Safety Approval:

UL 1577 (3750 Vrms/1 min.)

and CSA, VDE 0884 (pending)

• Advanced Sigma-Delta (Σ−∆)

A/D Converter Technology

• Fully Differential Circuit

Topology

Applications

• Motor Phase and Rail

Current Sensing

• Inverter Current Sensing

• Switched Mode Power

Supply Signal Isolation

• General Purpose Current

Sensing and Monitoring

• General Purpose Analog

Signal Isolation

Description

The HCPL-7800(A) isolation

amplifier family was designed

for current sensing in electronic

motor drives. In a typical

implementation, motor currents

flow through an external

resistor and the resulting analog

voltage drop is sensed by the

HCPL-7800(A). A differential

output voltage is created on the

other side of the HCPL-7800(A)

optical isolation barrier. This

differential output voltage is

proportional to the motor current

and can be converted to a single-

ended signal by using an op-amp

as shown in the recommended

application circuit. Since

common-mode voltage swings of

several hundred volts in tens of

nanoseconds are common in

modern switching inverter motor

drives, the HCPL-7800(A)

was designed to ignore very high

common-mode transient slew

rates (of at least 10 kV/µs).

The high CMR capability of the

HCPL-7800(A) isolation amplifier

provides the precision and

stability needed to accurately

monitor motor current in high

noise motor control environ-

ments, providing for smoother

control (less “torque ripple”) in

various types of motor control

applications.

The product can also be used for

general analog signal isolation

applications requiring high

accuracy, stability, and linearity

under similarly severe noise con-

ditions. For general applications,

we recommend the HCPL-7800

(gain tolerance of ±3%). For

precision applications Agilent

offers the HCPL-7800A with part-

to-part gain tolerance of ±1%.

The HCPL-7800(A) utilizes sigma

delta (∑-∆) analog-to-digital

converter technology, chopper

stabilized amplifiers, and a fully

differential circuit topology.

HCPL-7800A

HCPL-7800

CAUTION: It is advised that normal static precautions be taken in handling and assembly of this component to

prevent damage and/or degradation which may be induced by ESD.

DD1

IN+

IN–

GND1

DD2

OUT+

OUT–

GND2

–

SHIELD

Functional Diagram

Note: Initial or continued variation in the color of the HCPL-7800(A)’s white mold compound is normal and does not affect device

performance or reliability.

Ordering Information

Specify Part Number followed by Option Number (if desired).

HCPL-7800A = ±1% Gain Tol.; Mean Gain = 8.00

HCPL-7800 = ±3% Gain Tol.; Mean Gain = 8.00

Option: #YYY

No Option = Standard DIP package, 50 per tube

300 = Surface Mount Option

500 = Tape/Reel Packaging Option, 1k min. per reel

Package Outline Drawings

Standard DIP Package

Together, these features deliver

unequaled isolation-mode noise

rejection, as well as excellent offset

and gain accuracy and stability

over time and temperature. This

performance is delivered in a

compact, auto-insertable, industry

standard 8-pin DIP package that

meets worldwide regulatory safety

standards. (A gull-wing surface

mount option #300 is also

available).

9.65 ± 0.25

(0.380 ± 0.010)

1.78 (0.070) MAX.

1.19 (0.047) MAX.

A 7800

YYWW

DATE CODE

1.080 ± 0.320

(0.043 ± 0.013) 2.54 ± 0.25

(0.100 ± 0.010)

0.51 (0.020) MIN.

0.65 (0.025) MAX.

4.70 (0.185) MAX.

2.92 (0.115) MIN.

DIMENSIONS IN MILLIMETERS AND (INCHES).

5678

4321

5° TYP. 0.20 (0.008)

0.33 (0.013)

7.62 ± 0.25

(0.300 ± 0.010)

6.35 ± 0.25

(0.250 ± 0.010)

Gull Wing Surface Mount Option 300

Maximum Solder Reflow Thermal Profile

240

∆T = 115°C, 0.3°C/SEC

∆T = 100°C, 1.5°C/SEC

∆T = 145°C, 1°C/SEC

TIME – MINUTES

TEMPERATURE – °C

220

200

180

160

140

120

100

260

123 456789101112

(NOTE: USE OF NON-CHLORINE ACTIVATED FLUXES IS RECOMMENDED.)

0.635 ± 0.25

(0.025 ± 0.010) 12° NOM.

0.20 (0.008)

0.33 (0.013)

9.65 ± 0.25

(0.380 ± 0.010)

0.635 ± 0.130

(0.025 ± 0.005)

7.62 ± 0.25

(0.300 ± 0.010)

9.65 ± 0.25

(0.380 ± 0.010)

6.350 ± 0.25

(0.250 ± 0.010)

1.016 (0.040)

1.194 (0.047)

1.194 (0.047)

1.778 (0.070)

9.398 (0.370)

9.960 (0.390)

4.826

(0.190)



TYP.

0.381 (0.015)

0.635 (0.025)

PAD LOCATION (FOR REFERENCE ONLY)

1.080 ± 0.320

(0.043 ± 0.013)

4.19

(0.165)MAX.

1.780

(0.070)

MAX.

1.19

(0.047)

MAX.

2.54

(0.100)

BSC

DIMENSIONS IN MILLIMETERS (INCHES).

TOLERANCES (UNLESS OTHERWISE SPECIFIED):



xx.xx = 0.01

xx.xxx = 0.005

A 7800



YYWW

LEAD COPLANARITY 

MAXIMUM: 0.102 (0.004)

Regulatory Information

The HCPL-7800(A) are pending approval by the following organizations:

VDE 0884 Insulation Characteristics*

Description Symbol Characteristic Unit

Installation classification per DIN VDE 0110/1.89, Table 1

for rated mains voltage ≤300 Vrms I-IV

for rated mains voltage ≤450 Vrms I-III

for rated mains voltage ≤600 Vrms I-II

Climatic Classification 55/100/21

Pollution Degree (DIN VDE 0110/1.89) 2

Maximum Working Insulation Voltage VIORM 891 VPEAK

Input to Output Test Voltage, Method b**

VIORM x 1.875 = VPR, 100% Production Test with VPR 1670 VPEAK

tm = 1 sec, Partial discharge < 5 pC

Input to Output Test Voltage, Method a**

VIORM x 1.5 = VPR, Type and Sample Test, VPR 1336 VPEAK

tm = 60 sec, Partial discharge < 5 pC

Highest Allowable Overvoltage VIOTM 6000 VPEAK

(Transient Overvoltage tini = 10 sec)

Safety-limiting values—maximum values

allowed in the event of a failure.

Case Temperature TS175 °C

Input Current*** IS,INPUT 400 mA

Output Power*** PS,OUTPUT 600 mW

Insulation Resistance at TS, VIO = 500 V RS>109Ω

*Insulation characteristics are guaranteed only within the safety maximum ratings which must be ensured by protective circuits within

the application. Surface Mount Classification is Class A in accordance with CECC00802.

**Refer to the optocoupler section of the Isolation and Control Components Designer’s Catalog, under Product Safety Regulations

section, (VDE 0884) for a detailed description of Method a and Method b partial discharge test profiles.

***Refer to the following figure for dependence of PS and IS on ambient temperature.

VDE

Approval under VDE 0884/06.92

with VIORM = 891 VPEAK expected

prior to product release.

Approval under UL 1577, com-

ponent recognition program up

to VISO = 3750 Vrms expected

prior to product release.

CSA

Approved under CSA Component

Acceptance Notice #5, File CA

88324 expected prior to product

release.

OUTPUT POWER – P

, INPUT CURRENT – I

– CASE TEMPERATURE – °C

20050

400

12525 75 100 150

600

800

200

100

300

500

700

175

(mW)

(mA)

Insulation and Safety Related Specifications

Parameter Symbol Value Unit Conditions

Minimum External Air Gap L(101) 7.4 mm Measured from input terminals to output

(Clearance) terminals, shortest distance through air.

Minimum External Tracking L(102) 8.0 mm Measured from input terminals to output

(Creepage) terminals, shortest distance path along body.

Minimum Internal Plastic Gap 0.5 mm Through insulation distance conductor to

(Internal Clearance) conductor, usually the straight line distance

thickness between the emitter and detector.

Tracking Resistance CTI >175 Volts DIN IEC 112/VDE 0303 Part 1

(Comparative Tracking Index)

Isolation Group III a Material Group (DIN VDE 0110, 1/89,

Table 1)

Absolute Maximum Ratings

Parameter Symbol Min. Max. Unit Note

Storage Temperature TS-55 125 °C

Operating Temperature TA- 40 100

Supply Voltage VDD1, VDD2 0 5.5 V

Steady-State Input Voltage VIN+, VIN- -2.0 VDD1 +0.5

2 Second Transient Input Voltage -6.0

Output Voltage VOUT -0.5 VDD2 +0.5

Solder Reflow Temperature Profile See Package Outline Drawings Section

Recommended Operating Conditions

Parameter Symbol Min. Max. Unit Note

Ambient Operating Temperature TA-40 85 °C

Supply Voltage VDD1, VDD2 4.5 5.5 V

Input Voltage (accurate and linear) VIN+, VIN- -200 200 mV 1

Input Voltage (functional) VIN+, VIN- -2 2 V

DC Electrical Specifications

Unless otherwise noted, all typicals and figures are at the nominal operating conditions of VIN+ = 0,

VIN- = 0 V, VDD1 = VDD2 = 5 V and TA = 25°C; all Min./Max. specifications are within the Recommended

Operating Conditions.

Parameter Symbol Min. Typ. Max. Unit Test Conditions Fig. Note

-2.0 0.3 2.0 TA = 25°C

Input Offset Voltage VOS -3.0 3.0 mV -40°C < TA < +85°C 1,2

-4.5 V < (VDD1, VDD2) < 5.5 V

Magnitude of Input

Offset Change vs. |∆VOS/∆TA| 3.0 10.0 µV/°C32

Temperature

Gain (HCPL-7800A) G17.92 8.00 8.08 -200 mV < VIN+ < 200 mV, 3

Gain (HCPL-7800) G37.76 8.00 8.24 V/V TA = 25°C, -200 mV < VIN+

< 200 mV

Magnitude of VOUT

4,5,6

Gain Change vs. |∆G/∆TA| 0.00025 V/V/°C4

Temperature

VOUT 200 mV Nonlinearity NL200 0.0037 0.35 % -200 mV < VIN+ < 200 mV 5

Magnitude of VOUT 200 mV

Nonlinearity Change |dNL200/dT| 0.0002 % / °C 7,8

vs. Temperature

VOUT 100 mV Nonlinearity NL100 0.0027 0.2 % -100 mV < VIN+ < 100 mV 6

Maximum Input

Voltage before |VIN+|MAX 308.0 mV 9

VOUT Clipping

Input Supply Current IDD1 10.86 16.0 mA VIN+ = 400 mV 10 7

Output Supply Current IDD2 11.56 16.0 VIN+ = -400 mV 8

Input Current IIN+ -0.5 5.0 µA9

Magnitude of Input

Bias Current vs. |dIIN/dT| +0.45 nA/°C11

Temperature

Coefficient

Output Low Voltage VOL 1.29 V 10

Output High Voltage VOH 3.80 V

Output Common-Mode

Voltage VOCM 2.2 2.545 2.8 V

Output Short-Circuit

Current |IOSC| 18.6 mA 11

Equivalent Input

Impedance RIN 500 kΩ

VOUT Output

Resistance ROUT 15 Ω

Input DC Common-

Mode Rejection Ratio CMRRIN 76 dB 12

AC Electrical Specifications

Unless otherwise noted, all typicals and figures are at the nominal operating conditions of VIN+ = 0,

VIN- = 0 V, VDD1 = VDD2 = 5 V and TA = 25°C; all Min./Max. specifications are within the Recommended

Operating Conditions.

Parameter Symbol Min. Typ. Max. Unit Test Conditions Fig. Note

VOUT Bandwidth (-3 dB) BW 50 100 kHz VIN+ = 200 mVpk-pk 12,13

sine wave.

VOUT Noise NOUT 31.5 mVrms VIN+ = 0.0 V 13

VIN to VOUT Signal Delay tPD10 2.03 3.3 mVrms Measured at output of 14,15

(50 – 10%) MC34081 on Figure 15.

VIN to VOUT Signal Delay tPD50 3.47 5.6 µsV

IN+ = 0 mV to 150 mV step.

(50 – 50%)

VIN to VOUT Signal Delay tPD90 4.99 9.9

(50 – 90%)

VOUT Rise/Fall Time tR/F 2.96 6.6

(10 – 90%)

Common Mode Transient CMTI 10.0 15.0 kV/µsV

CM = 1 kV, TA = 25°C1614

Immunity

Power Supply Rejection PSR 170 mVrms With recommended 15

application circuit.

Package Characteristics

Parameter Symbol Min. Typ. Max. Unit Test Conditions Fig. Note

Input-Output Momentary VISO 3750 Vrms RH < 50%, t = 1 min., 16,17

Withstand Voltage TA = 25°C

Resistance (Input-Output) RI-O >109ΩVI-O = 500 VDC 18

Capacitance (Input- CI-O 1.2 pF ƒ = 1 MHz 18

Output)

Notes:

General Note: Typical values represent

the mean value of all characterization

units at the nominal operating conditions.

Typical drift specifications are determined

by calculating the rate of change of the

specified parameter versus the drift pa-

rameter (at nominal operating conditions)

for each characterization unit, and then

averaging the individual unit rates. The

corresponding drift figures are normalized

to the nominal operating conditions and

show how much drift occurs as the par-

ticular drift parameter is varied from its

nominal value, with all other parameters

held at their nominal operating values.

Note that the typical drift specifications

in the tables below may differ from the

slopes of the mean curves shown in the

corresponding figures.

1. Agilent recommends operation with

VIN- = 0 V (tied to GND1). Limiting

VIN+ to 100 mV will improve DC

nonlinearity and nonlinearity drift. If

VIN- is brought above VDD1 – 2 V, an

internal test mode may be activated.

This test mode is for testing LED

coupling and is not intended for

customer use.

2. This is the Absolute Value of Input

Offset Change vs. Temperature.

3. Gain is defined as the slope of the

best-fit line of differential output

voltage (VOUT+–VOUT- ) vs. differential

input voltage (VIN+–VIN-) over the

specified input range.

4. This is the Absolute Value of Gain

Change vs. Temperature.

5. Nonlinearity is defined as half of the

peak-to-peak output deviation from

the best-fit gain line, expressed as a

percentage of the full-scale differential

output voltage.

6. NL100 is the nonlinearity specified over

an input voltage range of ±100 mV.

7. The input supply current decreases

as the differential input voltage

(VIN+–VIN-) decreases.

8. The maximum specified output supply

current occurs when the differential

input voltage (VIN+–VIN-) = -200 mV,

the maximum recommended operat-

ing input voltage. However, the out-

put supply current will continue to

rise for differential input voltages up

to approximately -300 mV, beyond

which the output supply current

remains constant.

9. Because of the switched-capacitor

nature of the input sigma-delta con-

verter, time-averaged values are shown.

10. When the differential input signal

exceeds approximately 308 mV, the

outputs will limit at the typical values

shown.

11. Short circuit current is the amount of

output current generated when either

output is shorted to VDD2 or ground.

12. CMRR is defined as the ratio of the

differential signal gain (signal applied

differentially between pins 2 and 3)

to the common-mode gain (input pins

tied together and the signal applied

to both inputs at the same time),

expressed in dB.

13. Output noise comes from two primary

sources: chopper noise and sigma-

delta quantization noise. Chopper

noise results from chopper stabilization

of the output op-amps. It occurs at a

specific frequency (typically 400 kHz

at room temperature), and is not

attenuated by the internal output filter.

A filter circuit can be easily added to

the external post-amplifier to reduce

the total rms output noise. The

internal output filter does eliminate

most, but not all, of the sigma-delta

quantization noise. The magnitude of

the output quantization noise is very

small at lower frequencies (below

10 kHz) and increases with increasing

frequency.

14. CMTI (Common Mode Transient

Immunity or CMR, Common Mode

Rejection) is tested by applying an

exponentially rising/falling voltage

step on pin 4 (GND1) with respect to

pin 5 (GND2). The rise time of the

test waveform is set to approximately

50 ns. The amplitude of the step is

adjusted until the differential output

(VOUT+–VOUT-) exhibits more than a

200 mV deviation from the average

output voltage for more than 1µs.

The HCPL-7800(A) will continue to

func-tion if more than 10 kV/µs

common mode slopes are applied, as

long as the breakdown voltage

limitations are observed.

15. Data sheet value is the differential

amplitude of the transient at the

output of the HCPL-7800(A) when a

1 Vpk-pk, 1 MHz square wave with

40 ns rise and fall times is applied to

both VDD1 and VDD2.

16. In accordance with UL 1577, each

optocoupler is proof tested by

applying an insulation test voltage

≥4200 Vrms for 1 second (leakage

detection current limit, II-O ≤ 5 µA).

This test is performed before the

100% production test for partial

discharge (method b) shown in

VDE 0884 Insulation Characteristic

Table.

17. The Input-Output Momentary With-

stand Voltage is a dielectric voltage

rating that should not be interpreted

as an input-output continuous voltage

rating. For the continuous voltage

rating refer to the VDE 0884 insula-

tion characteristics table and your

equipment level safety specification.

18. This is a two-terminal measurement:

pins 1–4 are shorted together and

pins 5–8 are shorted together.

Figure 3. Input Offset vs. Supply. Figure 4. Gain vs. Temperature.

Figure 1. Input Offset Voltage Test Circuit.

Figure 2. Input Offset Voltage vs.

Temperature.

Figure 5. Gain and Nonlinearity Test Circuit.

– TEMPERATURE – °C

0.6

0.5

0.3

-25

0.8

35 95

0.2

0.7

-55 125

0.4

565

– INPUT OFFSET VOLTAGE – mV

– SUPPLY VOLTAGE – V

0.37

0.36

0.39

4.75 5.0

0.33

vs. V

DD1

4.5 5.55.25

vs. V

DD2

0.34

0.38

0.35

– INPUT OFFSET VOLTAGE – mV

G – GAIN – V/V

TA – TEMPERATURE – °C

8.025

8.02

8.015

-35

8.035

25 85

8.01

8.03

-55 1255 45 105

-15 65

0.1 µF

DD2

OUT

3HCPL-7800

0.1 µF

10 K

DD1

+15 V

0.1 µF

-15 V

–AD624CD

GAIN = 100

0.47

µF 0.47

µF

0.1 µF

DD2

3HCPL-7800

0.01 µF

10 K

+15 V

0.1 µF

-15 V

–AD624CD

GAIN = 4

0.47

µF 0.47

µF

DD1

13.2

404

OUT

+15 V

0.1 µF

-15 V

–AD624CD

GAIN = 10

10 K

0.47

µF

0.1 µF

Figure 12. Gain vs. Frequency. Figure 13. Phase vs. Frequency. Figure 14. Propagation Delay vs.

Temperature.

Figure 6. Gain vs. Supply. Figure 7. Nonlinearity vs. Temperature. Figure 8. Nonlinearity vs. Supply.

Figure 9. Output Voltage vs. Input

Voltage. Figure 10. Supply Current vs. Input

Voltage. Figure 11. Input Current vs. Input

Voltage.

– OUTPUT VOLTAGE – V

– INPUT VOLTAGE – V

2.6

1.8

-0.3

4.2

-0.1 0.1 0.3



1.0

3.4

-0.5 0.5

G – GAIN – V/V

– SUPPLY VOLTAGE – V

8.028

8.032

4.75 5.0

8.024

4.5 5.55.25

8.03

8.026 vs. V

DD1

vs. V

DD2

NL – NONLINEARITY – %

– TEMPERATURE – °C

0.02

0.015

0.005

-25

0.03

35 95

0.025

-55 125

0.01

565

NL – NONLINEARITY – %

– SUPPLY VOLTAGE – V

0.005

4.75 5.0

0.002

4.5 5.55.25

0.004

0.003

vs. V

DD1

vs. V

DD2

IDD – SUPPLY CURRENT – mA

VIN – INPUT VOLTAGE – V

-0.3

-0.1 0.1 0.3

-0.5 0.5

IDD1

IDD2

IN –

INPUT CURRENT – µA

– INPUT VOLTAGE – V

-3

-0.4

-0.2 0.2 0.4

-5

-1

-0.6 0.6

-2

-4

D –

PROPAGATION DELAY – µS

– TEMPERATURE – °C

3.1

-25

5.5

56595

1.5

4.7

-55 125

3.9

2.3

Tpd 10

Tpd 50

Tpd 90

Trise



GAIN – dB

FREQUENCY (Hz)

-2

-4

10 100000

-1

-3

1000

100 10000

PHASE – DEGREES

FREQUENCY (Hz)

-100

-300

10 100000

-50

-150

1000

-200

-250

100 10000

Figure 16. CMTI Test Circuits.

Figure 15. Propagation Delay Test Circuits.

0.1 µF

DD2

OUT

3HCPL-7800

2 K

+15 V

0.1 µF

-15 V

–

+MC34081

0.1 µF

10 K

0.01 µF

DD1

IMPEDANCE LESS THAN 10 Ω.

0.1 µF

DD2

OUT

3HCPL-7800

2 K

78L05 +15 V

0.1 µF

-15 V

–

+MC34081

150

IN OUT

0.1

µF 0.1

µF

9 V

PULSE GEN.

+–

10 K

150 pF

Application Information

Power Supplies and Bypassing

The recommended supply con-

nections are shown in Figure 17.

A floating power supply (which in

many applications could be the

same supply that is used to drive

the high-side power transistor) is

regulated to 5 V using a simple

zener diode (D1); the value of

resistor R4 should be chosen to

supply sufficient current from

the existing floating supply. The

voltage from the current sensing

resistor (Rsense) is applied to the

input of the HCPL-7800(A)

through an RC anti-aliasing filter

(R2 and C2). Although the

application circuit is relatively

simple, a few recommendations

should be followed to ensure

optimal performance.

The power supply for the

HCPL -7800(A) is most often

obtained from the same supply

used to power the power

transistor gate drive circuit. If a

dedicated supply is required, in

many cases it is possible to add

an additional winding on an exist-

ing transformer. Otherwise, some

sort of simple isolated supply can

be used, such as a line powered

transformer or a high-frequency

DC-DC converter.

An inexpensive 78L05 three-

terminal regulator can also be

used to reduce the floating supply

voltage to 5 V. To help attenuate

high-frequency power supply

noise or ripple, a resistor or

inductor can be used in series

with the input of the regulator

to form a low-pass filter with

the regulator’s input bypass

capacitor.

Figure 17. Recommended Supply and Sense Resistor Connections.

HCPL-7800

C1

0.1 µF



R2



39 Ω

GATE DRIVE

CIRCUIT

FLOATING

POWER

SUPPLY

• • •

HV+

• • •

HV–

• • • –+

SENSE

MOTOR

C2

0.01 µF



D1

5.1 V

–

As shown in Figure 18, 0.1 µF

bypass capacitors (C1, C2)

should be located as close as

possible to the pins of the

HCPL-7800(A). The bypass

capacitors are required because

of the high-speed digital nature

of the signals inside the

HCPL-7800(A). A 0.01 µF

bypass capacitor (C2) is also

recommended at the input due to

the switched-capacitor nature of

the input circuit. The input

bypass capacitor also forms part

of the anti-aliasing filter, which is

recommended to prevent high-

frequency noise from aliasing

down to lower frequencies and

interfering with the input signal.

The input filter also performs an

important reliability function—it

reduces transient spikes from

ESD events flowing through the

current sensing resistor.

Figure 18: Recommended Application Circuit.

PC Board Layout

The design of the printed circuit

board (PCB) should follow good

layout practices, such as keeping

bypass capacitors close to the

supply pins, keeping output

signals away from input signals,

the use of ground and power

planes, etc. In addition, the

layout of the PCB can also affect

the isolation transient immunity

(CMTI) of the HCPL-7800(A),

due primarily to stray capacitive

coupling between the input and

the output circuits. To obtain

optimal CMTI performance, the

layout of the PC board should

minimize any stray coupling by

maintaining the maximum

possible distance between the

input and output sides of the

circuit and ensuring that any

ground or power plane on the

Figure 19. Example Printed Circuit Board Layout.

C2 C4

TO R

SENSE+

TO R

SENSE–

TO V

DD1

TO V

DD2

OUT+

OUT–

PC board does not pass directly

below or extend much wider than

the body of the HCPL-7800(A).

0.1 µF

+5 V

OUT

3U2

R1



2.00 K

+15 V C8

0.1 µF



0.1 µF



-15 V

–

+MC34081

R3



10.0 K

HCPL-7800

R4

10.0 K

C6

150 pF

U1

78L05

IN OUT

C1

C2



0.01

µF

R5



GATE DRIVE

CIRCUIT

POSITIVE

FLOATING

SUPPLY

• • •

HV+

• • •

HV–

• • • –+

SENSE

MOTOR

C5

150 pF

0.1

µF 0.1

µF

C3



C7



R2



2.00 K

can also be easily calculated by

multiplying the sense resistance

times the square of the maximum

RMS current, which is about 1 W

in the previous example. If the

power dissipation in the sense

resistor is too high, the resistance

can be decreased below the

maximum value to decrease

power dissipation. The minimum

value of the sense resistor is

limited by precision and accuracy

requirements of the design. As

the resistance value is reduced,

the output voltage across the

resistor is also reduced, which

means that the offset and noise,

which are fixed, become a larger

percentage of the signal amp-

litude. The selected value of the

sense resistor will fall somewhere

between the minimum and

maximum values, depending on

the particular requirements of

a specific design.

When sensing currents large

enough to cause significant

heating of the sense resistor, the

temperature coefficient (tempco)

of the resistor can introduce

nonlinearity due to the signal

dependent temperature rise of the

resistor. The effect increases as

the resistor-to-ambient thermal

Figure 20. Motor Output Horsepower

vs. Motor Phase Current and Supply

Voltage.

MOTOR PHASE CURRENT – A (rms)

10 25 30

035

440 V

380 V

220 V

120 V

MOTOR OUTPUT POWER – HORSEPOWER

resistance increases. This effect

can be minimized by reducing the

thermal resistance of the current

sensing resistor or by using a

resistor with a lower tempco.

Lowering the thermal resistance

can be accomplished by

repositioning the current sensing

resistor on the PC board, by

using larger PC board traces to

carry away more heat, or by

using a heat sink.

For a two-terminal current

sensing resistor, as the value

of resistance decreases, the re-

sistance of the leads become a

significant percentage of the total

resistance. This has two primary

effects on resistor accuracy.

First, the effective resistance of

the sense resistor can become

dependent on factors such as

how long the leads are, how

they are bent, how far they are

inserted into the board, and how

far solder wicks up the leads

during assembly (these issues

will be discussed in more detail

shortly). Second, the leads are

typically made from a material,

such as copper, which has a

much higher tempco than the

material from which the resistive

element itself is made, resulting

in a higher tempco overall.

Both of these effects are

eliminated when a four-terminal

current sensing resistor is used. A

four-terminal resistor has two

additional terminals that are

Kelvin-connected directly across

the resistive element itself; these

two terminals are used to monitor

the voltage across the resistive

element while the other two

terminals are used to carry the

load current. Because of the

Kelvin connection, any voltage

drops across the leads carrying

the load current should have no

impact on the measured voltage.

Current Sensing Resistors

The current sensing resistor

should have low resistance (to

minimize power dissipation), low

inductance (to minimize di/dt

induced voltage spikes which

could adversely affect operation),

and reasonable tolerance (to

maintain overall circuit accuracy).

Choosing a particular value for

the resistor is usually a compro-

mise between minimizing power

dissipation and maximizing accu-

racy. Smaller sense resistance

decreases power dissipation,

while larger sense resistance

can improve circuit accuracy by

utilizing the full input range of

the HCPL -7800(A).

The first step in selecting a sense

resistor is determining how much

current the resistor will be sensing.

The graph in Figure 20 shows the

RMS current in each phase of a

three-phase induction motor as a

function of average motor output

power (in horsepower, hp) and

motor drive supply voltage. The

maximum value of the sense re-

sistor is determined by the current

being measured and the maxi-

mum recommended input voltage

of the isolation amplifier. The

maximum sense resistance can be

calculated by taking the maxi-

mum recommended input voltage

and dividing by the peak current

that the sense resistor should see

during normal operation. For

example, if a motor will have a

maximum RMS current of 10 A

and can experience up to 50%

overloads during normal op-

eration, then the peak current is

21.1 A (=10 x 1.414 x 1.5).

Assuming a maximum input

voltage of 200 mV, the maximum

value of sense resistance in this

case would be about 10 mΩ.

The maximum average power

dissipation in the sense resistor

When laying out a PC board for

the current sensing resistors, a

couple of points should be kept

in mind. The Kelvin connections

to the resistor should be brought

together under the body of the

resistor and then run very close

to each other to the input of the

HCPL-7800(A); this minimizes

the loop area of the connection

and reduces the possibility of

stray magnetic fields from

interfering with the measured

signal. If the sense resistor is not

located on the same PC board as

the HCPL-7800(A) circuit, a

tightly twisted pair of wires can

accomplish the same thing.

Also, multiple layers of the PC

board can be used to increase

current carrying capacity.

Numerous plated-through vias

should surround each non-Kelvin

terminal of the sense resistor

to help distribute the current

between the layers of the PC

board. The PC board should use

2 or 4 oz. copper for the layers,

resulting in a current carrying

capacity in excess of 20 A.

Making the current carrying

traces on the PC board fairly

large can also improve the sense

resistor’s power dissipation

capability by acting as a heat

sink. Liberal use of vias where

the load current enters and exits

the PC board is also

recommended.

Sense Resistor Connections

The recommended method for

connecting the HCPL-7800(A) to

the current sensing resistor is

shown in Figure 18. VIN+ (pin 2

of the HPCL-7800(A)) is

connected to the positive

terminal of the sense resistor,

while VIN- (pin 3) is shorted to

GND1 (pin 4), with the power-

supply return path functioning as

the sense line to the negative

terminal of the current sense

resistor. This allows a single pair

of wires or PC board traces to

connect the HCPL-7800(A)

circuit to the sense resistor. By

referencing the input circuit to

the negative side of the sense

resistor, any load current induced

noise transients on the resistor

are seen as a common-mode

signal and will not interfere with

the current-sense signal. This is

important because the large load

currents flowing through the

motor drive, along with the

parasitic inductances inherent in

the wiring of the circuit, can

generate both noise spikes and

offsets that are relatively large

compared to the small voltages

that are being measured across

the current sensing resistor.

If the same power supply is used

both for the gate drive circuit

and for the current sensing

circuit, it is very important that

the connection from GND1 of

the HCPL-7800(A) to the sense

resistor be the only return path

for supply current to the gate

drive power supply in order to

eliminate potential ground loop

problems. The only direct

connection between the

HCPL-7800(A) circuit and the

gate drive circuit should be the

positive power supply line.

Output Side

The op-amp used in the external

post-amplifier circuit should be of

sufficiently high precision

so that it does not contribute

a significant amount of offset

or offset drift relative to the

contribution from the isolation

amplifier. Generally, op-amps

with bipolar input stages exhibit

better offset performance than

op-amps with JFET or MOSFET

input stages.

In addition, the op-amp should

also have enough bandwidth

and slew rate so that it does not

adversely affect the response

speed of the overall circuit. The

post-amplifier circuit includes a

pair of capacitors (C5 and C6)

that form a single-pole low-pass

filter; these capacitors allow the

bandwidth of the post-amp to

be adjusted independently of the

gain and are useful for reducing

the output noise from the isola-

tion amplifier. Many different op-

amps could be used in the circuit,

including: MC34082A (Motorola),

TLO32A, TLO52A, and TLC277

(Texas Instruments), LF412A

(National Semiconductor).

The gain-setting resistors in the

post-amp should have a tolerance

of 1% or better to ensure adequate

CMRR and adequate gain toler-

ance for the overall circuit.

Resistor networks can be used

that have much better ratio

tolerances than can be achieved

using discrete resistors. A

resistor network also reduces

the total number of components

for the circuit as well as the

required board space.

Please refer to Agilent

Applications Note 1078 for

additional information on using

Isolation Amplifiers.

FREQUENTLY ASKED QUESTIONS ABOUT THE HCPL-7800(A)

1. THE BASICS

1.1: Why should I use the HCPL-7800(A) for sensing current when Hall-effect sensors are

available which don’t need an isolated supply voltage?

Available in an auto-insertable, 8-pin DIP package, the HCPL-7800(A) is smaller than and has better

linearity, offset vs. temperature and Common Mode Rejection (CMR) performance than most Hall-effect

sensors. Additionally, often the required input-side power supply can be derived from the same supply that

powers the gate-drive optocoupler.

2. SENSE RESISTOR AND INPUT FILTER

2.1: Where do I get 10 mΩ resistors? I have never seen one that low.

Although less common than values above 10 Ω, there are quite a few manufacturers of resistors suitable for

measuring currents up to 50 A when combined with the HCPL-7800(A).

Example product information may be found at Dale’s web site (http://www.vishay.com/vishay/dale) and

Isotek’s web site (http://www.isotekcorp.com).

2.2: Should I connect both inputs across the sense resistor instead of grounding VIN- directly

to pin 4?

This is not necessary, but it will work. If you do, be sure to use an RC filter on both pin 2 (VIN+) and pin 3

(VIN-) to limit the input voltage at both pads.

2.3: Do I really need an RC filter on the input? What is it for? Are other values of R and C okay?

The input anti-aliasing filter (R=39 Ω, C=0.01 µF) shown in the typical application circuit is recommended

for filtering fast switching voltage transients from the input signal. (This helps to attenuate higher signal

frequencies which could otherwise alias with the input sampling rate and cause higher input offset voltage.)

Some issues to keep in mind using different filter resistors or capacitors are:

1. (Filter resistor:) Input bias current for pins 2 and 3: This is on the order of 500 nA. If you are using a

single filter resistor in series with pin 2 but not pin 3 the IxR drop across this resistor will add to the offset

error of the device. As long as this IR drop is small compared to the input offset voltage there should not be

a problem. If larger-valued resistors are used in series, it is better to put half of the resistance in series with

pin 2 and half the resistance in series with pin 3. In this case, the offset voltage is due mainly to resistor

mismatch (typically less than 1% of the resistance design value) multiplied by the input bias.

2. (Filter resistor:) The equivalent input resistance for HCPL-7800(A) is around 500 kΩ. It is therefore

best to ensure that the filter resistance is not a significant percentage of this value; otherwise the offset

voltage will be increased through the resistor divider effect. [As an example, if Rfilt = 5.5 kΩ, then VOS =

(Vin * 1%) = 2 mV for a maximum 200 mV input and VOS will vary with respect to Vin.]

3. The input bandwidth is changed as a result of this different R-C filter configuration. In fact this is one of

the main reasons for changing the input-filter R-C time constant.

4. (Filter capacitance:) The input capacitance of the HCPL-7800(A) is approximately 1.5 pF. For proper

operation the switching input-side sampling capacitors must be charged from a relatively fixed (low

impedance) voltage source. Therefore, if a filter capacitor is used it is best for this capacitor to be a few

orders of magnitude greater than the CINPUT (A value of at least 100 pF works well.)

2.4: How do I ensure that the HCPL-7800(A) is not destroyed as a result of short circuit

conditions which cause voltage drops across the sense resistor that exceed the ratings of the

HCPL-7800(A)’s inputs?

Select the sense resistor so that it will have less than 5 V drop when short circuits occur. The only other

requirement is to shut down the drive before the sense resistor is damaged or its solder joints melt. This

ensures that the input of the HCPL-7800(A) can not be damaged by sense resistors going open-circuit.

3. ISOLATION AND INSULATION

3.1: How many volts will the HCPL-7800(A) withstand?

The momentary (1 minute) withstand voltage is 3750 V rms per UL 1577 and CSA Component Acceptance

Notice #5.

4. ACCURACY

4.1: Can the signal to noise ratio be improved?

Yes. Some noise energy exists beyond the 100 kHz bandwidth of the HCPL-7800(A). Additional filtering

using different filter R,C values in the post-amplifier application circuit can be used to improve the signal to

noise ratio. For example, by using values of R3 = R4 = 10 kΩ, C5 = C6 = 470 pF in the application circuit

the rms output noise will be cut roughly by a factor of 2. In applications needing only a few kHz bandwidth

even better noise performance can be obtained. The noise spectral density is roughly 500 nV/√ Hz below

20 kHz (input referred).

4.2: Does the gain change if the internal LED light output degrades with time?

No. The LED is used only to transmit a digital pattern. Agilent has accounted for LED degradation in the

design of the product to ensure long life.

5. POWER SUPPLIES AND START-UP

5.1: What are the output voltages before the input side power supply is turned on?

VO+ is close to 1.29 V and VO- is close to 3.80 V. This is equivalent to the output response at the condition

that LED is completely off.

5.2: How long does the HCPL-7800(A) take to begin working properly after power-up?

Within 1 ms after VDD1 and VDD2 powered the device starts to work. But it takes longer time for output to

settle down completely. In case of the offset measurement while both inputs are tied to ground there is

initially VOS adjustment (about 60 ms). The output completely settles down in 100 ms after device powering

up.

6. MISCELLANEOUS

6.1: How does the HCPL-7800(A) measure negative signals with only a +5 V supply?

The inputs have a series resistor for protection against large negative inputs. Normal signals are no more

than 200 mV in amplitude. Such signals do not forward bias any junctions sufficiently to interfere with

accurate operation of the switched capacitor input circuit.

www.agilent.com/semiconductors

For product information and a complete list of

distributors, please go to our web site.

For technical assistance call:

Americas/Canada: +1 (800) 235-0312 or

(408) 654-8675

Europe: +49 (0) 6441 92460

China: 10800 650 0017

Hong Kong: (+65) 271 2451

India, Australia, New Zealand: (+65) 271 2394

Japan: (+81 3) 3335-8152(Domestic/Interna-

tional), or 0120-61-1280(Domestic Only)

Korea: (+65) 271 2194

Malaysia, Singapore: (+65) 271 2054

Taiwan: (+65) 271 2654

Data subject to change.

Obsoletes 5967-5627E

March 6, 2002

5988-5801EN