FEATURES
●LOW-NOISE PREAMP:
– Low Input Noise: 0.95nV/
√Hz
– Active Termination Noise Reduction
– Switchable Termination Value
– 80MHz Bandwidth
– 5dB to 25dB Gain
– Differential In and Out
●LOW-NOISE VARIABLE GAIN AMPLIFIER:
– Low-Noise VCA
– Up to 40dB Gain Range
– 40MHz Bandwidth
– Differential In and Out
●LOW CROSSTALK:
66dB at Max Gain, 5MHz
●HIGH-SPEED VARIABLE GAIN ADJUST
●SWITCHABLE EXTERNAL PROCESSING
APPLICATIONS
●ULTRASOUND SYSTEMS
●WIRELESS RECEIVERS
●TEST EQUIPMENT
DESCRIPTION
The VCA2616 and VCA2611 are dual, Low-Noise Preamplifiers
(LNP), plus low-noise Variable Gain Amplifiers (VGA). The
VCA2611 is an upgraded version of the VCA2616. The only
difference between the VCA2616 and the VCA2611 is the input
structure to the LNP. The VCA2616 is limited to –0.3V negative-
going input spikes; the VCA2611 is limited to –2.0V negative-
going input spikes. This change allows the user to use slower
and less expensive input clamping diodes prior to the LNP input.
In some designs, input clamping may not be required.
The combination of Active Termination (AT) and Maximum
Gain Select (MGS) allow for the best noise performance. The
VCA2616 and VCA2611 also feature low crosstalk and out-
standing distortion performance.
The LNP has differential input and output capability and is
strappable for gains of 5dB, 17dB, 22dB, or 25dB. Low input
impedance is achieved by AT, resulting in as much as a 4.6dB
improvement in noise figure over conventional shunt termina-
tion. The termination value can also be switched to accommo-
date different sources. The output of the LNP is available for
external signal processing.
The variable gain is controlled by an analog voltage whose
gain varies from 0dB to the gain set by the MGS. The ability
to program the variable gain also allows the user to optimize
dynamic range. The VCA input can be switched from the LNP
to external circuits for different applications. The output can be
used in either a single-ended or differential mode to drive high-
performance Analog-to-Digital (A/D) converters, and is cleanly
limited for optimum overdrive recovery.
The combination of low noise, gain, and gain range program-
mability makes the VCA2616 and VCA2611 versatile building
blocks in a number of applications where noise performance
is critical. The VCA2616 and VCA2611 are available in a
TQFP-48 package.
Dual, Variable-Gain Amplifier
with Low-Noise Preamp
Low Noise
Preamp
5dB to 25dB
Programmable
Gain Amplifier
24 to 45dB
Voltage
Controlled
Attenuator
Analog
Control Maximum Gain
Select
RF2
RF1FB
FBSW
LNPINP
LNPINN
LNPGS1
LNPGS2
LNPGS3
LNP
Gain
Set
Input
LNPOUTPSEL
VCAINP
LNPOUTN VCAINN VCACNTL
FBCNTL
VCAOUTP
VCAOUTN
MGS1MGS2MGS3
Maximum Gain Select
VCA2616
(1 of 2 Channels)
VCA2616
VCA2611
SBOS234E – MARCH 2002 – REVISED NOVEMBER 2004
VCA2616
www.ti.com
Copyright © 2002-2004, Texas Instruments Incorporated
All trademarks are the property of their respective owners.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
VCA2616, VCA2611
2SBOS234E
www.ti.com
ABSOLUTE MAXIMUM RATINGS(1)
SPECIFIED
PACKAGE TEMPERATURE PACKAGE ORDERING TRANSPORT
PRODUCT PACKAGE-LEAD DESIGNATOR RANGE MARKING NUMBER MEDIA, QUANTITY
VCA2616 TQFP-48 PFB –40°C to +85°C VCA2616 VCA2616YT Tape and Reel, 250
"" " " "VCA2616YR Tape and Reel, 2000
VCA2611 TQFP-48 PFB –40°C to +85°C VCA2611 VCA2611Y/250 Tape and Reel, 250
"" " " "VCA2611Y/2K Tape and Reel, 2000
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet.
PACKAGE/ORDERING INFORMATION(1)
ELECTRICAL CHARACTERISTICS
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted.
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas Instru-
ments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degrada-
tion to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
Power Supply (+VS) ............................................................................. +6V
VCA2616 Analog Input ............................................ –0.3V to (+VS + 0.3V)
VCA2611 Analog Input ............................................ –2.0V to (+VS + 0.3V)
Logic Input ............................................................... –0.3V to (+VS + 0.3V)
Case Temperature ......................................................................... +100°C
Junction Temperature .................................................................... +150°C
Storage Temperature...................................................... –40°C to +150°C
NOTE: (1) Stresses above those listed under “Absolute Maximum Ratings” may
cause permanent damage to the device. Exposure to absolute maximum
conditions for extended periods may affect device reliability.
NOTES: (1) For preamp driving VGA.
(2) Referenced to best fit dB-linear curve.
VCA2616Y, VCA2611Y
PARAMETER CONDITIONS MIN TYP MAX UNITS
PREAMPLIFIER
Input Resistance 600 kΩ
Input Capacitance 15 pF
Input Bias Current 1nA
CMRR f = 1MHz, VCACNTL = 0.2V 50 dB
Maximum Input Voltage Preamp Gain = +5dB 1 VPP
Preamp Gain = +25dB 112 mVPP
Input Voltage Noise(1) Preamp Gain = +5dB 4.2 nV/
√Hz
Preamp Gain = +25dB 0.95 nV/
√Hz
Input Current Noise Independent of Gain 0.35 pA/
√Hz
Noise Figure, RS = 75Ω, RIN = 75Ω(1) RF = 550Ω, Preamp Gain = 22dB, 6.2 dB
PGA Gain = 39dB
Bandwidth Gain = 22dB 80 MHz
PROGRAMMABLE VARIABLE GAIN AMPLIFIER
Peak Input Voltage Differential 2 VPP
–3dB Bandwidth 40 MHz
Slew Rate 300 V/µs
Output Signal Range RL ≥ 500Ω Each Side to Ground 2 VPP
Output Impedance f = 5MHz 1 Ω
Output Short-Circuit Current ±40 mA
3rd-Harmonic Distortion f = 5MHz, VOUT = 1VPP, VCACNTL = 3.0V –45 –71 dBc
2nd-Harmonic Distortion f = 5MHz, VOUT = 1VPP, VCACNTL = 3.0V –45 –63 dBc
IMD, 2-Tone VOUT = 2VPP, f = 1MHz –75 dBc
VOUT = 2VPP, f = 10MHz –75 dBc
Crosstalk
VCACNTL = 0.2V
–66 dB
Group Delay Variation 1MHz < f < 10MHz, Full Gain Range ±2ns
DC Output Level, VIN = 0 2.5 V
ACCURACY
Gain Slope 10.9 dB/V
Gain Error ±1(2) dB
Output Offset Voltage ±50 mV
Total Gain VCACNTL = 0.2V 18 21 24 dB
VCACNTL = 3.0V 47 50 53 dB
GAIN CONTROL INTERFACE
Input Voltage (VCACNTL) Range 0.2 to 3.0 V
Input Resistance 1MΩ
Response Time 40dB Gain Change, MGS = 111 0.2 µs
POWER SUPPLY
Operating Temperature Range –40 +85 °C
Specified Operating Range 4.75 5.0 5.25 V
Power Dissipation Operating, Both Channels 410 495 mW
VCA2616, VCA2611 3
SBOS234E www.ti.com
PIN CONFIGURATION
36
35
34
33
32
31
30
29
28
27
26
25
V
DD
B
NC
NC
VCA
IN
NB
VCA
IN
PB
LNP
OUT
NB
LNP
OUT
PB
SWFBB
FBB
COMP1B
COMP2B
LNP
IN
NB
GNDA
VCA
OUT
NA
VCA
OUT
PA
FBSW
CNTL
VCA
IN
SEL
VCA
CNTL
MGS
1
MGS
2
MGS
3
VCA
OUT
PB
VCA
OUT
NB
GNDB
LNP
GS3
A
LNP
GS2
A
LNP
GS1
A
LNP
IN
PA
V
DD
R
V
BIAS
V
CM
GNDR
LNP
IN
PB
LNP
GS1
B
LNP
GS2
B
LNP
GS3
B
1
2
3
4
5
6
7
8
9
10
11
12
V
DD
A
NC
NC
VCA
IN
NA
VCA
IN
PA
LNP
OUT
NA
LNP
OUT
PA
SWFBA
FBA
COMP1A
COMP2A
LNP
IN
NA
48 47 46 45 44 43 42 41 40 39 38
13 14 15 16 17 18 19 20 21 22 23
37
24
VCA2616
VCA2611
1V
DDA Channel A +Supply
2 NC Do Not Connect
3 NC Do Not Connect
4 VCAINNA Channel A VCA Negative Input
5 VCAINPA Channel A VCA Positive Input
6LNP
OUTNA Channel A LNP Negative Output
7LNP
OUTPA Channel A LNP Positive Output
8 SWFBA Channel A Switched Feedback Output
9 FBA Channel A Feedback Output
10 COMP1A Channel A Frequency Compensation 1
11 COMP2A Channel A Frequency Compensation 2
12 LNPINNA Channel A LNP Inverting Input
13 LNPGS3A Channel A LNP Gain Strap 3
14 LNPGS2A Channel A LNP Gain Strap 2
15 LNPGS1A Channel A LNP Gain Strap 1
16 LNPINPA Channel A LNP Noninverting Input
17 VDDR +Supply for Internal Reference
18 VBIAS 0.01µF Bypass to Ground
19 VCM 0.01µF Bypass to Ground
20 GNDR Ground for Internal Reference
21 LNPINPB Channel B LNP Noninverting Input
22 LNPGS1B Channel B LNP Gain Strap 1
23 LNPGS2B Channel B LNP Gain Strap 2
24 LNPGS3B Channel B LNP Gain Strap 3
25 LNPINNB Channel B LNP Inverting Input
26 COMP2B Channel B Frequency Compensation 2
27 COMP1B Channel B Frequency Compensation 1
28 FBB Channel B Feedback Output
29 SWFBB Channel B Switched Feedback Output
30 LNPOUTPB Channel B LNP Positive Output
31 LNPOUTNB Channel B LNP Negative Output
32 VCAINPB Channel B VCA Positive Input
33 VCAINNB Channel B VCA Negative Input
34 NC Do Not Connect
35 NC Do Not Connect
36 VDDB Channel B +Analog Supply
37 GNDB Channel B Analog Ground
38 VCAOUTNB Channel B VCA Negative Output
39 VCAOUTPB Channel B VCA Positive Output
40 MGS3Maximum Gain Select 3 (LSB)
41 MGS2Maximum Gain Select 2
42 MGS1Maximum Gain Select 1 (MSB)
43 VCACNTL VCA Control Voltage
44 VCAINSEL VCA Input Select, HI = External
45 FBSWCNTL Feedback Switch Control: HI = ON
46 VCAOUTPA Channel A VCA Positive Output
47 VCAOUTNA Channel A VCA Negative Output
48 GNDA Channel A Analog Ground
PIN
DESIGNATOR
DESCRIPTION PIN
DESIGNATOR
DESCRIPTION
PIN DESCRIPTIONS
Top View TQFP
VCA2616, VCA2611
4SBOS234E
www.ti.com
TYPICAL CHARACTERISTICS
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal
amplitude compared to differential operation.
GAIN vs VCA
CNTL
VCA
CNTL
(V)
0.2 1.21.00.4 0.6 0.8 1.8 2.0 2.21.61.4 2.4 2.6 2.8 3.0
Gain (dB)
65
60
55
50
45
40
35
30
25
20
15
MGS = 110
MGS = 111
MGS = 101 MGS = 100
MGS = 011
MGS = 010
MGS = 001
MGS = 000
GAIN ERROR vs TEMPERATURE
VCA
CNTL
(V)
0.2 1.0 1.20.80.4 0.6 2.0 2.21.4 1.6 1.8 2.4 2.6 2.8 3.0
Gain Error (dB)
2.0
1.5
1.0
0.5
0.0
–0.5
–1.0
–1.5
–2.0
+85°C
–40°C
+25°C
GAIN ERROR vs VCACNTL
VCACNTL (V)
0.2 1.0 1.20.80.4 0.6 2.0 2.21.4 1.6 1.8 2.4 2.6 2.8 3.0
Gain Error (dB)
2.0
1.5
1.0
0.5
0.0
–0.5
–1.0
–1.5
–2.0
10MHz
1MHz
5MHz
GAIN ERROR vs VCACNTL
VCACNTL (V)
0.2 1.0 1.20.80.4 0.6 2.0 2.21.4 1.6 1.8 2.4 2.6 2.8 3.0
Gain Error (dB)
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
MGS = 011
MGS = 000
MGS = 111
GAIN MATCH
0.2V CHA to CHB
Delta Gain (dB)
Units
50
45
40
35
30
25
20
15
10
5
0
–0.55
–0.48
–0.42
–0.35
–0.29
–0.22
–0.16
–0.09
–0.03
0.04
0.16
0.10
0.23
0.36
0.29
Delta Gain (dB)
–0.26
–0.23
–0.20
–0.17
–0.14
–0.10
0.07
0.04
0.01
0.02
0.09
0.05
0.12
0.18
0.15
Units
60
50
40
30
20
10
0
GAIN MATCH
3.0V CHA to CHB
VCA2616, VCA2611 5
SBOS234E www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500 Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal
amplitude compared to differential operation.
GAIN vs FREQUENCY
(Pre-Amp)
Frequency (Hz)
Gain (dB)
30
25
20
15
10
5
0
LNA = 25dB LNA = 22dB
LNA = 17dB
LNA = 5dB
100k 1M 10M 100M
GAIN vs FREQUENCY
VCA
(VCACNTL = 0.2V)
Frequency (Hz)
Gain (dB)
5.0
4.0
3.0
2.0
1.0
0.0
–1.0
–2.0
–3.0
–4.0
–5.0
MGS = 111
MGS = 100
MGS = 011
MGS = 000
100k 1M 10M 100M
GAIN vs FREQUENCY
VCA
(VCA
CNTL
= 3.0V)
Frequency (Hz)
Gain (dB)
45
40
35
30
25
20
15
10
5
0
MGS = 111
MGS = 100
MGS = 011
MGS = 000
100k 1M 10M 100M
GAIN vs FREQUENCY
LNA and VCA
(VCA
CNTL
= 3.0V)
Frequency (Hz)
100k 1M 10M 100M
Gain (dB)
60
50
40
30
20
10
0
LNP = 25dB
LNP = 22dB
LNP = 5dB
LNP = 17dB
GAIN vs FREQUENCY
LNA and VCA
(LNP = 22dB)
Frequency (Hz)
Gain (dB)
60
50
40
30
20
10
0
VCNTL = 3.0V
VCNTL = 1.6V
VCNTL = 0.2V
100k 1M 10M 100M
OUTPUT-REFERRED NOISE vs VCACNTL
(LNP = 25dB)
VCACNTL (V)
0.2 1.0 1.20.4 0.6 0.8 1.8 2.01.4 1.6 2.2 2.4 2.6 2.8 3.0
Noise (nV/√Hz)
2000
1800
1600
1400
1200
1000
800
600
400
200
0
RS= 50Ω
MGS = 111
MGS = 011
VCA2616, VCA2611
6SBOS234E
www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended unless otherwise noted. This results in a 6dB reduction in signal
amplitude compared to differential operation.
INPUT-REFERRED NOISE vs VCACNTL
(LNP = 25dB)
VCACNTL (V)
0.2 1.0 1.20.4 0.6 0.8 1.8 2.01.4 1.6 2.2 2.4 2.6 2.8 3.0
Noise (nV/√Hz)
24
22
20
18
16
14
12
10
8
6
4
2
0MGS = 011
MGS = 111
RS = 50Ω
INPUT-REFERRED NOISE vs R
S
(LNP = 25dB)
R
S
(Ω)
1 10 100 1k
Noise (nV√Hz)
10.0
1.0
0.1
MGS = 111
NOISE FIGURE vs RS
(LNP = 25dB)
RS (Ω)
10 100 1k
Noise Figure (dB)
9
8
7
6
5
4
3
2
1
0
MGS = 111
30
28
26
24
22
20
18
16
14
12
10
8
6
4
2
0
Noise Figure (dB)
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
VCA
CNTL
(V)
NOISE FIGURE vs VCA
CNTL
(LNP = 25dB)
MGS = 111
DISTORTION vs FREQUENCY
MGS = 000
2V
PP
DIFFERENTIAL
Frequency (Hz)
100k 1M 10M
Harmonic (dBc)
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
V
C
= 0.2, H2
V
C
= 0.2, H3 V
C
= 3.0, H2
V
C
= 3.0, H3
DISTORTION vs FREQUENCY
MGS = 011
2V
PP
DIFFERENTIAL
Frequency (Hz)
100k 1M 10M
Distortion (dBc)
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
V
C
= 0.2, H2
V
C
= 3.0, H2 V
C
= 3.0, H3
V
C
= 0.2, H3
VCA2616, VCA2611 7
SBOS234E www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500 Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal
amplitude compared to differential operation.
DISTORTION vs FREQUENCY
MGS = 111
2V
PP
DIFFERENTIAL
Frequency (Hz)
100k 1M 10M
Distortion (dBc)
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
VC = 0.2, H2
VC = 0.2, H3
VC = 3.0, H3
VC = 3.0, H2
DISTORTION vs FREQUENCY
MGS = 000
1VPP SINGLE-ENDED
Frequency (Hz)
100k 1M 10M
Distortion (dBc)
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
V
C
= 0.2, H3 V
C
= 0.2, H2
V
C
= 3.0, H2 V
C
= 3.0, H3
DISTORTION vs FREQUENCY
MGS = 011
1VPP SINGLE-ENDED
Frequency (Hz)
100k 1M 10M
Distortion (dBc)
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
V
C
= 0.2, H3 V
C
= 0.2, H2
V
C
= 3.0, H2 V
C
= 3.0, H3
DISTORTION vs FREQUENCY
MGS = 111
1VPP SINGLE-ENDED
Frequency (Hz)
100k 1M 10M
Distortion (dBc)
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
V
C
= 0.2, H2 V
C
= 0.2, H3
V
C
= 3.0, H2 V
C
= 3.0, H3
DISTORTION vs VCA
CNTL
2V
PP
DIFFERENTIAL
VCA
CNTL
(V)
Distortion (dBc)
–45
–50
–55
–60
–65
–70
–75
–80
MGS = 011, H2 MGS = 000, H2
MGS = 111, H3 MGS = 011, H3
MGS = 000, H3
MGS = 111, H2
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
DISTORTION vs VCA
CNTL
1V
PP
SINGLE-ENDED
VCA
CNTL
(V)
Distortion (dBc)
–45
–50
–55
–60
–65
–70
–75
–80
MGS = 011, H2 MGS = 000, H2
MGS = 000, H3
MGS = 111, H3
MGS = 011, H3
MGS = 111, H2
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
VCA2616, VCA2611
8SBOS234E
www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal
amplitude compared to differential operation.
–5
–15
–25
–35
–45
–55
–65
–75
–85
Crosstalk (dB)
CROSSTALK vs FREQUENCY
1VPP SINGLE-ENDED
MGS = 011
Frequency (Hz)
1M 10M 100M
VCACNTRL 0V
VCACNTRL 1.5V
VCACNTRL 3.0V
ICC vs TEMPERATURE
Temperature (°C)
–20–30–40 –100 102030405060708090
ICC (dBFS)
80.50
80.00
79.50
79.00
78.50
78.00
77.50
77.00
76.50
76.00
16
14
12
10
8
6
4
2
0
Group Delay (nS)
1M 10M 100M
Frequency (Hz)
GROUP DELAY vs FREQUENCY
V
C
= 3.0
V
C
= 0.2
VCA2616, VCA2611 9
SBOS234E www.ti.com
VCA—OVERVIEW
The magnitude of the differential VCA input signal (from the
LNP or an external source) is reduced by a programmable
attenuation factor, set by the analog VCA Control Voltage
(VCACNTL) at pin 43. The maximum attenuation factor is
further programmable by using the three MGS bits
(pins 40-42). Figure 3 illustrates this dual-adjustable charac-
teristic. Internally, the signal is attenuated by having the
analog VCACNTL vary the channel resistance of a set of
shunt-connected FET transistors. The MGS bits effectively
adjust the overall size of the shunt FET by switching parallel
components in or out under logic control. At any given
maximum gain setting, the analog variable gain characteris-
tic is linear in dB as a function of the control voltage, and is
created as a piecewise approximation of an ideal dB-linear
transfer function. The VCA gain control circuitry is common
to both channels of the VCA2616 and VCA2611.
FIGURE 1. Simplified Block Diagram of the VCA2616.
FIGURE 2. Recommended Circuit for Coupling an External
Signal into the VCA Inputs.
FIGURE 3. Swept Attenuator Characteristic.
THEORY OF OPERATION
The VCA2616 and VCA2611 are dual-channel systems con-
sisting of three primary blocks: an LNP, a VCA, and a
Programmable Gain Amplifier (PGA). For greater system
flexibility, an onboard multiplexer is provided for the VCA
inputs, selecting either the LNP outputs or external signal
inputs. Figure 1 shows a simplified block diagram of the dual-
channel system.
LNP—OVERVIEW
The LNP input may be connected to provide active-feedback
signal termination, achieving lower system noise perfor-
mance than conventional passive shunt termination. Further
lower noise performance is obtained if signal termination is
not required. The unterminated LNP input impedance is
600kΩ. The LNP can process fully differential or single-
ended signals in each channel. Differential signal processing
results in significantly reduced 2nd-harmonic distortion and
improved rejection of common-mode and power-supply noise.
The first gain stage of the LNP is AC-coupled into its output
buffer with a 4.8µs time constant (33kHz high-pass charac-
teristic). The buffered LNP outputs are designed to drive the
succeeding VCA directly or, if desired, external loads as low
as 135Ω with minimal impact on signal distortion. The LNP
employs very low impedance local feedback to achieve
stable gain with the lowest possible noise and distortion.
Four pin-programmable gain settings are available: 5dB,
17dB, 22dB, and 25dB. Additional intermediate gains can be
programmed by adding trim resistors between the Gain Strap
programming pins.
The common-mode DC level at the LNP output is nominally
2.5V, matching the input common-mode requirement of the
VCA for simple direct coupling. When external signals are
fed to the VCA, they should also be set up with a 2.5VDC
common-mode level. Figure 2 shows a circuit that demon-
strates the recommended coupling method using an external
op amp. The VCM node shown in Figure 2 is the VCM output
(pin 19). Typical R and C values are shown, yielding a high-
pass time constant similar to that of the LNP. If a different
common-mode referencing method is used, it is important
that the common-mode level be within 10mV of the VCM
output for proper operation.
V
CM
(+2.5V)
1kΩ
1kΩ
47nF To VCA
IN
Input
Signal
VCALNP
Channel A
Input
VCA
Control
PGA Channel A
Output
External
InA
Maximum
Gain
Select MGS
Analog
Control
VCALNP
Channel B
Input PGA Channel B
Output
External
InB
0
–24
VCA Attenuation (dB)
–45
Control Voltage (V)
0
Maximum Attenuation
Minimum Attenuation
3.0
VCA2616, VCA2611
10 SBOS234E
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the circuit. This reduces the susceptibility to power-supply
variation, ripple, and noise. In addition, separate power
supply and ground connections are provided for each chan-
nel and for the reference circuitry, further reducing interchannel
crosstalk.
Further details regarding the design, operation, and use of
each circuit block are provided in the following sections.
LOW-NOISE PREAMPLIFIER (LNP)—DETAIL
The LNP is designed to achieve a low-noise figure, espe-
cially when employing active termination. Figure 4 is a
simplified schematic of the LNP, illustrating the differential
input and output capability. The input stage employs low
resistance local feedback to achieve stable low-noise, low-
distortion performance with very high input impedance. Nor-
mally, low noise circuits exhibit high power consumption as
a result of the large bias currents required in both input and
output stages. The LNP uses a patented technique that
combines the input and output stages such that they share
the same bias current. Transistors Q4 and Q5 amplify the
signal at the gate-source input of Q4, the +IN side of the LNP.
The signal is further amplified by the Q1 and Q2 stage, and
then by the final Q3 and RL gain stage, which uses the same
bias current as the input devices Q4 and Q5. Devices Q6
through Q10 play the same role for signals on the –IN side.
The differential gain of the LNP is given in Equation 1:
Gain R
R
L
S
=×
2
FIGURE 4. Schematic of the Low-Noise Preamplifier (LNP).
(1)
PGA OVERVIEW AND OVERALL DEVICE
CHARACTERISTICS
The differential output of the VCA attenuator is then amplified
by the PGA circuit block. This post-amplifier is programmed
by the same MGS bits that control the VCA attenuator,
yielding an overall swept-gain amplifier characteristic in which
the VCA × PGA gain varies from 0dB (unity) to a program-
mable peak gain of 24-, 27-, 30-, 33-, 36-, 39-, 42-, or 45dB.
The Gain vs VCACNTL curve in the Typical Characteristics
shows the composite gain control characteristic of the entire
VCA2616. Setting VCACNTL to 3.0V causes the digital MGS
gain control to step in 3dB increments. Setting VCACNTL to 0V
causes all the MGS-controlled gain curves to converge at
one point. The gain at the convergence point is the LNP gain
less 6dB, because the measurement setup looks at only one
side of the differential PGA output, resulting in 6dB lower
signal amplitude.
ADDITIONAL FEATURES—OVERVIEW
Overload protection stages are placed between the attenua-
tor and the PGA, providing a symmetrically clipped output
whenever the input becomes large enough to overload the
PGA. A comparator senses the overload signal amplitude
and substitutes a fixed DC level to prevent undesirable
overload recovery effects. As with the previous stages, the
VCA is AC-coupled into the PGA. In this case, the coupling
time constant varies from 5µs at the highest gain (45dB) to
59µs at the lowest gain (25dB).
The VCA2616 includes a built-in reference, common to both
channels, to supply a regulated voltage for critical areas of
RL
93Ω
RS1
105Ω
Q3
Q4
Q5
Q2
Q1
RS2
34Ω
RS3
17Ω
LNPGS2
LNPINPLNPINN
To Bias
Circuitry
LNPGS1
LNPGS3
RL
93Ω
COMP2A VDD COMP1A
LNPOUTNLNP
OUTP
Buffer Buffer
Q8
Q7
To Bias
Circuitry
Q6
Q9
Q10
RWRW
CCOMP
4.7pF
(External
Capacitor)
VCA2616, VCA2611 11
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LNP GAIN (dB) Input-Referred Output-Referred
25 1.35 2260
22 1.41 1650
17 1.63 1060
5 4.28 597
The LNP is capable of generating a 2VPP differential signal.
The maximum signal at the LNP input is therefore 2VPP
divided by the LNP gain. An input signal greater than this
would exceed the linear range of the LNP, an especially
important consideration at low LNP gain settings.
The VCA2611 is an upgraded version of the VCA2616. The
only difference between the VCA2616 and the VCA2611 is the
input structure to the LNP. The VCA2616 is limited to –0.3V
negative-going input spikes; the VCA2611 is limited to –2.0V
negative-going input spikes. This change allows the user to
use slower and less expensive input clamping diodes prior to
the LNA input. In some designs, input clamping may not be
required.
ACTIVE FEEDBACK WITH THE LNP
One of the key features of the LNP architecture is the ability
to employ active-feedback termination to achieve superior
noise performance. Active-feedback termination achieves a
lower noise figure than conventional shunt termination, es-
sentially because no signal current is wasted in the termina-
tion resistor itself. Another way to understand this is to
consider first that the input source, at the far end of the signal
cable, has a cable-matching source resistance of RS. Using
conventional shunt termination at the LNP input, a second
terminating resistor of value RS is connected to ground.
Therefore, the signal loss is 6dB due to the voltage divider
action of the series and shunt RS resistors. The effective
source resistance has been reduced by the same factor of 2,
but the noise contribution has been reduced by only the
√2
,
only a 3dB reduction. Therefore, the net theoretical SNR
degradation is 3dB, assuming a noise-free amplifier input. (In
practice, the amplifier noise contribution will degrade both
the unterminated and the terminated noise figures, some-
what reducing the distinction between them.)
See Figure 5 for an amplifier using active feedback. This
diagram appears very similar to a traditional inverting ampli-
fier. However, the analysis is somewhat different because
the gain A in this case is not a very large open-loop op amp
gain; rather, it is the relatively low and controlled gain of the
LNP itself. Thus, the impedance at the inverting amplifier
terminal will be reduced by a finite amount, as given in the
familiar relationship of Equation 3:
RRA
IN F
=+
(
)
1
(3)
where RF is the feedback resistor (supplied externally be-
tween the LNPINP and FB terminals for each channel), A is
It is also possible to create other gain settings by connecting
an external resistor between LNPGS1 on one side, and
LNPGS2 and/or LNPGS3 on the other. In that case, the
internal resistor values (see Figure 4) should be combined
with the external resistor to calculate the effective value of RS
for use in Equation 1. The resulting expression for external
resistor value is given in Equation 2:
RR R R R Gain R R
Gain R R
EXT SL FIX L SFIX
SL
=+×
×
22 2
11
1
––
where REXT is the externally selected resistor value needed
to achieve the desired gain setting, RS1 is the fixed parallel
resistor in Figure 4, and RFIX is the effective fixed value of the
remaining internal resistors: RS2, RS3, or (RS2 || RS3), de-
pending on the pin connections.
Note that the best process and temperature stability will be
achieved by using the pre-programmed fixed-gain options of
Table I, since the gain is then set entirely by internal resistor
ratios, which are typically accurate to ±0.5%, and track quite
well over process and temperature. When combining exter-
nal resistors with the internal values to create an effective RS
value, note that the internal resistors have a typical tempera-
ture coefficient of +700ppm/°C and an absolute value toler-
ance of approximately ±5%, yielding somewhat less predict-
able and stable gain settings. With or without external resis-
tors, the board layout should use short Gain Strap connec-
tions to minimize parasitic resistance and inductance effects.
The overall noise performance of the VCA2616 and VCA2611
will vary as a function of gain. Table II shows the typical input-
and-output-referred noise densities of the entire VCA2616 and
VCA2611 for maximum VCA and PGA gain; that is, VCACNTL
set to 3.0V and all MGS bits set to 1. Note that the input-
referred noise values include the contribution of a 50Ω fixed
source impedance, and are therefore somewhat larger than
the intrinsic input noise. As the LNP gain is reduced, the noise
contribution from the VCA/PGA portion becomes more signifi-
cant, resulting in higher input-referred noise. However, the
output-referred noise, which is indicative of the overall SNR at
that gain setting, is reduced.
To preserve the low-noise performance of the LNP, the user
should take care to minimize resistance in the input lead. A
parasitic resistance of only 10Ω will contribute 0.4nV/
√Hz
.
NOISE (nV/√Hz)
TABLE II. Equivalent Noise Performance for MGS = 111 and
VCACNTL = 3.0V with 50Ω source impedance.
LNP PIN STRAPPING LNP GAIN (dB)
LNPGS1, LNPGS2, LNPGS3 Connected Together 25
LNPGS1 Connected to LNPGS3 22
LNPGS1 Connected to LNPGS2 17
All Pins Open 5
TABLE I. Pin Strappings of the LNP for Various Gains.
(2)
where RL is the load resistor in the drains of Q3 and Q8, and
RS is the resistor connected between the sources of the input
transistors Q4 and Q7. The connections for various RS com-
binations are brought out to device pins LNPGS1, LNPGS2,
and LNPGS3 (pins 13-15 for channel A, 22-24 for channel B).
These Gain Strap pins allow the user to establish one of four
fixed LNP gain options as shown in Table I.
VCA2616, VCA2611
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FIGURE 5. Configurations for Active Feedback and Conven-
tional Cable Termination.
FIGURE 6. Noise Figure for Active Termination.
FIGURE 7. Noise Figure for Conventional Termination.
FIGURE 8. Low-Frequency LNP Time Constants.
the user-selected gain of the LNP, and RIN is the resulting
amplifier input impedance with active feedback. In this case,
unlike the conventional termination above, both the signal
voltage and the RS noise are attenuated by the same factor
of 2 (6dB) before being re-amplified by the A gain setting.
This avoids the extra 3dB degradation due to the square-root
effect described earlier, the key advantage of the active
termination technique.
This previous explanation ignored the input noise contribu-
tion of the LNP itself. Also, the noise contribution of the
feedback resistor must be included for a completely correct
analysis. The curves given in Figures 6 and 7 allow the
VCA2616 and VCA2611 user to compare the achievable
noise figure for active and conventional termination methods.
The left-most set of data points in each graph give the results
for typical 50Ω cable termination, showing the worst noise
figure but also the greatest advantage of the active feedback
method.
A switch, controlled by the FBSWCNTL signal on pin 45,
enables the user to reduce the feedback resistance by
adding an additional parallel component, connected between
the LNPINP and SWFB terminals. The two different values of
feedback resistance will result in two different values of
active-feedback input resistance. Thus, the active-feedback
impedance can be optimized at two different LNP gain
settings. The switch is connected at the buffered output of
the LNP and has an ON resistance of approximately 1Ω.
When employing active feedback, the user should be careful
to avoid low-frequency instability or overload problems. Fig-
ure 8 illustrates the various low-frequency time constants.
RF
A
RIN
RIN =
RS
RS
RS
= RS
LNPIN
RF
1 + A
Active Feedback
A
Conventional Cable Termination
VCA NOISE = 3.8nV√Hz, LNP GAIN = 20dB
Source Impedance (Ω)
0 300100 200 500400 600 700 800 900 1000
Noise Figure (dB)
9
8
7
6
5
4
3
2
1
0
6.0E-10
8.0E-10
1.0E-09
1.2E-09
1.4E-09
1.6E-09
1.8E-09
2.0E-09
LNP Noise
nV/√Hz
Source Impedance (Ω)
0 300100 200 500400 600 700 900 1000800
Noise Figure (dB)
14
12
10
8
6
4
2
0
VCA NOISE = 3.8nV√Hz, LNP GAIN = 20dB
LNP Noise
nV/√Hz
6.0E-10
8.0E-10
1.0E-09
1.2E-09
1.4E-09
1.6E-09
1.8E-09
2.0E-09
RS
200kΩ
CC
CF
0.001µF
VCM
RF
44pF
Buffer
200kΩ
VCM
LNPOUTN
LNPOUTP
44pF
Gain
Stage
(VCA) LNP
Buffer
VCA2616, VCA2611 13
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Referring again to the input resistance calculation of Equa-
tion (3), and considering that the gain term “A” falls off below
21kHz, it is evident that the effective LNP input impedance
will rise below 3.6kHz, with a DC limit of approximately RF. To
avoid interaction with the feedback pole/zero at low frequen-
cies, and to avoid the higher signal levels resulting from the
rising impedance characteristic, it is recommended that the
external RFCC time constant be set to about 5µs.
Achieving the best active-feedback architecture is difficult
with conventional op amp circuit structures. The overall gain
A must be negative in order to close the feedback loop, the
input impedance must be high to maintain low current noise
and good gain accuracy, but the gain ratio must be set with
very low value resistors to maintain good voltage noise.
Using a two-amplifier configuration (noninverting for high
impedance plus inverting for negative feedback reasons)
results in excessive phase lag and stability problems when
the loop is closed. The VCA2616 and VCA2611 use a
patented architecture that achieves these requirements, with
the additional benefits of low power dissipation and differen-
tial signal handling at both input and output.
For greatest flexibility and lowest noise, the user may wish to
shape the frequency response of the LNP. The COMP1 and
COMP2 pins for each channel (pins 10 and 11 for channel A,
pins 26 and 27 for channel B) correspond to the drains of Q3
and Q8, see Figure 4. A capacitor placed between these pins
will create a single-pole low-pass response, in which the
effective R of the RC time constant is approximately 186Ω.
COMPENSATION WHEN USING ACTIVE FEEDBACK
The typical open-loop gain versus frequency characteristic for
the LNP is shown in Figure 9. The –3dB bandwidth is approxi-
mately 180MHz and the phase response is such that when
feedback is applied, the LNP will exhibit a peaked response or
might even oscillate. One method of compensating for this
undesirable behavior is to place a compensation capacitor at
the input to the LNP, as shown in Figure 10. This method is
effective when the desired –3dB bandwidth is much less than
the open-loop bandwidth of the LNP. This compensation
technique also allows the total compensation capacitor to
include any stray or cable capacitance that is associated with
the input connection. Equation 4 relates the bandwidth to the
various impedances that are connected to the LNP.
BW A1R R
2 C(R )(R )
IF
IF
=+
(
)
+
π
(4)
AVOIDING UNSTABLE PERFORMANCE
The VCA2612 and the VCA2616 are very similar in perfor-
mance in all respects, except in the area of noise performance.
See Figure 4 for a schematic of the LNP. This brings the input
noise of the VCA2616 and VCA2611 down to 1.0nV/
√Hz
compared to the input on the VCA2612 1.25nV/
√Hz
imped-
ance at the gate of either Q4 or Q7, as can be approximated
by the network shown in Figure 11. The resistive component
shown in Figure 11 is negative, which gives rise to unstable
behavior when the signal source resistance has both inductive
and capacitive elements. It should be noted that this negative
resistance is not a physical resistor, but an equivalent resis-
tance that is a function of the devices shown in Figure 4.
Normally, when an inductor and capacitor are placed in series
or parallel, there is a positive resistance in the loop that
prevents unstable behavior.
FIGURE 9. Open-Loop Gain Characteristic of LNP.
FIGURE 10. LNP with Compensation Capacitor.
25dB
Gain
–3dB Bandwidth
180MHz
Output
Input
R
F
R
I
CA
FIGURE 11. VCA2616 and VCA2611 Input Impedance.
24pF
57pF–93Ω
For the VCA2616 and VCA2611, the situation can be rem-
edied by placing an external resistor with a value of approxi-
mately 15Ω or higher in series with the input lead. The net
series resistance will be positive, and there will be no
observed instability.
Although this technique will prevent oscillations, it is not
recommended, as it will also increase the input noise. A
4.7pF external capacitor must be placed between pins
COMP2A (pin 11) and LNPINPA (pin 16), and between pins
COMP2B (pin 26) and LNPINPB (pin 21). This has the result
of making the input impedance always capacitive due to the
feedback effect of the compensation capacitor and the gain
of the LNP. Using capacitive feedback, the LNP becomes
unconditionally stable, as there is no longer a negative
component to the input impedance. The compensation
capacitor mentioned above will be reflected to the input by
the formula:
CIN = (A + 1)CCOMP (5)
VCA2616, VCA2611
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The capacitance that is determined in Equation 5 should be
added to the capacitance of Equation 4 to determine the
overall bandwidth of the LNP. The LNPINNA (pin 12) and the
LNPINNB (pin 25) should be bypassed to ground by the
shortest means possible to avoid any inductance in the lead.
LNP OUTPUT BUFFER
The differential LNP output is buffered by wideband class AB
voltage followers which are designed to drive low impedance
loads. This is necessary to maintain LNP gain accuracy,
since the VCA input exhibits gain-dependent input imped-
ance. The buffers are also useful when the LNP output is
brought out to drive external filters or other signal processing
circuitry. Good distortion performance is maintained with
buffer loads as low as 135Ω. As mentioned previously, the
buffer inputs are AC-coupled to the LNP outputs with a
3.6kHz high-pass characteristic, and the DC common-mode
level is maintained at the correct VCM for compatibility with
the VCA input.
VOLTAGE-CONTROLLED ATTENUATOR (VCA)—DETAIL
The VCA is designed to have a dB-linear attenuation charac-
teristic; that is, the gain loss in dB is constant for each equal
increment of the VCACNTL control voltage. See Figure 1 for a
block diagram of the VCA. The attenuator is essentially a
variable voltage divider consisting of one series input resis-
tor, RS, and ten identical shunt FETs, placed in parallel and
controlled by sequentially activated clipping amplifiers. Each
clipping amplifier can be thought of as a specialized voltage
comparator with a soft transfer characteristic and well-con-
trolled output limit voltages. The reference voltages V1 through
V10 are equally spaced over the 0V to 3.0V control voltage
range. As the control voltage rises through the input range of
each clipping amplifier, the amplifier output will rise from 0V
(FET completely ON) to VCM – VT (FET nearly OFF), where
VCM is the common source voltage and VT is the threshold
voltage of the FET. As each FET approaches its OFF state
and the control voltage continues to rise, the next clipping
amplifier /FET combination takes over for the next portion of
the piecewise-linear attenuation characteristic. Thus, low
control voltages have most of the FETs turned ON, while
high control voltages have most turned OFF. Each FET acts
to decrease the shunt resistance of the voltage divider
formed by RS and the parallel FET network.
The attenuator is comprised of two sections, with five parallel
clipping amplifier/FET combinations in each. Special refer-
ence circuitry is provided so that the (VCM – VT) limit voltage
will track temperature and IC process variations, minimizing
the effects on the attenuator control characteristic.
In addition to the analog VCACNTL gain setting input, the
attenuator architecture provides digitally programmable ad-
justment in eight steps, via the three MGS bits. These adjust
the maximum achievable gain (corresponding to minimum
attenuation in the VCA, with VCACNTL = 3.0V) in 3dB incre-
ments. This function is accomplished by providing multiple
FET sub-elements for each of the Q1 to Q10 FET shunt
elements (see Figure 12). In the simplified diagram of
Figure 13, each shunt FET is shown as two sub-elements,
QNA and QNB. Selector switches, driven by the MGS bits,
activate either or both of the sub-element FETs to adjust the
maximum RON and thus achieve the stepped attenuation
options.
The VCA can be used to process either differential or single-
ended signals. Fully differential operation will reduce 2nd-
harmonic distortion by about 10dB for full-scale signals.
Input impedance of the VCA will vary with gain setting, due
to the changing resistances of the programmable voltage
divider structure. At large attenuation factors (that is, low gain
settings), the impedance will approach the series resistor
value of approximately 135Ω.
As with the LNP stage, the VCA output is AC-coupled into the
PGA. This means that the attenuation-dependent DC com-
mon-mode voltage will not propagate into the PGA, and so
the PGA’s DC output level will remain constant.
Finally, note that the VCACNTL input consists of FET gate
inputs. This provides very high impedance and ensures that
multiple VCA2616 and VCA2611 devices may be connected
in parallel with no significant loading effects. The nominal
voltage range for the VCACNTL input spans from 0V to 3V.
Overdriving this input (≤ 5V) does not affect the performance.
INPUT OVERLOAD RECOVERY
One of the most important applications for the VCA2616 and
VCA2611 is processing signals in an ultrasound system. The
ultrasound signal flow begins when a large signal is applied to
a transducer, which converts electrical energy to acoustic
energy. It is not uncommon for the amplitude of the electrical
signal that is applied to the transducer to be ±50V or greater.
FIGURE 13. Programmable Attenuator Section.
R
S
Q
1A
A1
B1
VCM
INPUT OUTPUT
Programmable Attenuator Section
B2
Q
1B
Q
2A
A2
Q
2B
Q
3A
A3
Q
3B
Q
4A
A4
Q
4B
Q
5A
A5
Q
5B
VCA2616, VCA2611 15
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FIGURE 12. Piecewise Approximation to Logarithmic Control Characteristics.
R
S
Attenuator
Input Attenuator
Output
A1-A10 Attenuator Stages
Control
Input
Q
1
V
CM
0dB
–4.5dB
Q
2
Q
3
C
1
V1
Q
4
Q
5
Q
S
C
1
-C
10
Clipping Amplifiers
Attenuation Characteristic of Individual FETs
Q
6
Q
7
Q
8
Q
9
Q
10
C
2
V2
V
CM
-V
T
0V1 V2 V3 V4 V5 V6 V7 V8 V9 V10
Characteristic of Attenuator Control Stage Output
Overall Control Characteristics of Attenuator
–45dB
0dB
0.3V 3V
Control Signal
C
3
V3
C
4
V4
C
5
V5
C
6
V6
C
7
V7
C
8
V8
C
9
V9
C
10
V10
A1 A2 A3 A4 A5 A6 A7 A8 A9 A10
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FIGURE 15. Simplified Block Diagram of the PGA Section Within the VCA2616 and VCA2611.
R
S1
R
L
R
S2
VCA
OUT
P
+In
Q
11
Q
3
Q
4
Q
5
Q
1
V
CM
Q
2
VCA
OUT
N
Q
9
Q
8
Q
13
Q
14
Q
7
Q
6
Q
12
V
DD
V
CM
R
L
Q
10
–In
To Bias
Circuitry
To Bias
Circuitry
RF
CF
VDD
LNPINP
ESD Diode
Protection
Network
LNPOUTN
LNP
FIGURE 14. VCA2616 and VCA2611 Diode Bridge Protection
Circuit.
To prevent damage, it is necessary to place a protection circuit
between the transducer and the VCA2616 and VCA2611 (see
Figure 14). Care must be taken to prevent any signal from
turning the ESD diodes on. Turning on the ESD diodes inside
the VCA2616 and VCA2611 could cause the input coupling
capacitor (CC) to charge to the wrong value.
PGA POST-AMPLIFIER—DETAIL
Figure 15 shows a simplified circuit diagram of the PGA block.
As described previously, the PGA gain is programmed with
the same MGS bits which control the VCA maximum attenu-
ation factor. Specifically, the PGA gain at each MGS setting is
the inverse (reciprocal) of the maximum VCA attenuation at
that setting. Therefore, the VCA + PGA overall gain will always
be 0dB (unity) when the analog VCACNTL input is set to 0V
MGS ATTENUATOR GAIN DIFFERENTIAL ATTENUATOR +
SETTING VCACNTL = 0V to 3V PGA GAIN DIFF. PGA GAIN
000 –24dB to 0dB 24dB 0dB to 24dB
001 –27dB to 0dB 27dB 0dB to 27dB
010 –30dB to 0dB 30dB 0dB to 30dB
011 –33dB to 0dB 33dB 0dB to 33dB
100 –36dB to 0dB 36dB 0dB to 36dB
101 –39dB to 0dB 39dB 0dB to 39dB
110 –42dB to 0dB 42dB 0dB to 42dB
111 –45dB to 0dB 45dB 0dB to 45dB
TABLE III. MGS Settings.
(= maximum attenuation). For VCACNTL = 3V (no attenuation),
the VCA + PGA gain will be controlled by the programmed
PGA gain (24 to 45 dB in 3dB steps). For clarity, the gain and
attenuation factors are detailed in Table III.
The PGA architecture consists of a differential, program-
mable-gain voltage to current converter stage followed by
transimpedance amplifiers to create and buffer each side of
the differential output. The circuitry associated with the volt-
age to current converter is similar to that previously de-
scribed for the LNP, with the addition of eight selectable PGA
gain-setting resistor combinations (controlled by the MGS
bits) in place of the fixed resistor network used in the LNP.
Low input noise is also a requirement of the PGA design due
to the large amount of signal attenuation which can be
inserted between the LNP and the PGA. At minimum VCA
attenuation (used for small input signals) the LNP noise
dominates; at maximum VCA attenuation (large input sig-
nals) the PGA noise dominates. Note that if the PGA output
is used single-ended, the apparent gain will be 6dB lower.
PACKAGE OPTION ADDENDUM
www.ti.com 1-Aug-2011
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status (1) Package Type Package
Drawing Pins Package Qty Eco Plan (2) Lead/
Ball Finish MSL Peak Temp (3) Samples
(Requires Login)
VCA2611Y/250 ACTIVE TQFP PFB 48 250 Green (RoHS
& no Sb/Br) CU NIPDAU Level-3-260C-168 HR
VCA2611Y/2K ACTIVE TQFP PFB 48 2000 TBD Call TI Call TI
VCA2616YR ACTIVE TQFP PFB 48 2000 Green (RoHS
& no Sb/Br) CU NIPDAU Level-3-260C-168 HR
VCA2616YT ACTIVE TQFP PFB 48 250 Green (RoHS
& no Sb/Br) CU NIPDAU Level-3-260C-168 HR
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
VCA2611Y/250 TQFP PFB 48 250 177.8 16.4 9.6 9.6 1.5 12.0 16.0 Q2
VCA2616YR TQFP PFB 48 2000 330.0 16.8 9.6 9.6 1.5 12.0 16.0 Q2
VCA2616YT TQFP PFB 48 250 177.8 16.4 9.6 9.6 1.5 12.0 16.0 Q2
PACKAGE MATERIALS INFORMATION
www.ti.com 14-Jul-2012
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
VCA2611Y/250 TQFP PFB 48 250 210.0 185.0 35.0
VCA2616YR TQFP PFB 48 2000 367.0 367.0 38.0
VCA2616YT TQFP PFB 48 250 210.0 185.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 14-Jul-2012
Pack Materials-Page 2
MECHANICAL DATA
MTQF019A – JANUARY 1995 – REVISED JANUARY 1998
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
PFB (S-PQFP-G48) PLASTIC QUAD FLATPACK
4073176/B 10/96
Gage Plane
0,13 NOM
0,25
0,45
0,75
Seating Plane
0,05 MIN
0,17
0,27
24
25
13
12
SQ
36
37
7,20
6,80
48
1
5,50 TYP
SQ
8,80
9,20
1,05
0,95
1,20 MAX 0,08
0,50 M
0,08
0°–7°
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Falls within JEDEC MS-026
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