October 2005 1 MIC2198
MIC2198 Micrel, Inc.
MIC2198
500kHz 4mm × 4mm Synchronous Buck Controller
General Description
The MIC2198 is a high-power 500kHz synchronous buck
DC-to-DC controller housed in a small 4mm × 4mm MLF™
12-pin package. The MIC2198 operates from a wide 4.5V
to 32V input and can be programmed for output voltages
from 0.8V to 6V. The wide input voltage capability makes
the MIC2198 an ideal solution for point-of-load DC-to-DC
conversion in 5V, 12V and higher systems.
The 500kHz switching frequency allows the use of a small
inductor and small output capacitors. The current mode PWM
control along with the external COMP pin allows for ease of
stability compensation and fast transient response across a
wide range of applications.
An all N-Channel synchronous architecture and powerful
output drivers allow up to 20A of output current capability.
For higher efficiency at higher input voltages, refer to the
300kHz MIC2199.
The MIC2198 is available in a 12-pin 4mm × 4mm MLF™
package with a junction temperature range from –40°C to
125°C.
Features
• 4.5V to 32V input range
• 4mm × 4mm MLF™ package
• 500kHz PWM operation
• >90% efficiency
• Output voltage adjustable down to 0.8V
• 20A output current capability
• Drives all N-Channel MOSFETs
• Logic controlled micropower shutdown
• Cycle-by-cycle current limiting
• Adjustable undervoltage lockout
• Frequency foldback overcurrent protection
Applications
• Point-of-load DC-to-DC conversion from 5V, 12V
and higher supplies
• Telecom equipment
• Wireless modems
• Servers
• Base stations
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
Typical Application
IRF7821
IRF7821
1µH 0.01Ω 3.3V/7A
10kΩ
6.04kΩ
1nF
5.5V to 24V
VIN
EN/UVLO
COMP
CSH
LSD
VSW
VDD
BST
HSD
GND
VOUT
FB
2.2nF
2kΩ
U1 MIC2198BML
SD103BWS
0.1µF
22
µ
F
4.7µF
220µF
VIN
VOUT
5.5V–24V to 3.3V/7A Converter
MLF and MicroLeadFrame are trademarks of Amkor Technology, Inc.
50
55
60
65
70
75
80
85
90
95
100
02468
ILOAD
(A)
VIN = 5.5V
VOUT = 3.3V
Efficiency
vs. Load Current
MIC2198 Micrel, Inc.
MIC2198 2 October 2005
Pin Configuration
1
2
3
4
COMP
EN/UVLO
FB
CSH
HSD
VSW
BST
GND
VOUT
VIN
5
6
12
11
10
9
8
7
LSD
VDD
12-Pin 4×4 MLF (ML)
Pin Description
Pin Number Pin Name Pin Function
1 COMP Compensation (Output): Internal error amplifier output. Connect to capacitor
or series RC network to compensate the regulator control loop.
2 EN/UVLO Enable/Undervoltage Lockout (Input): Low-level signal powers down the
controller. Input below the 2.5V threshold disables switching and functions as
an accurate undervoltage lockout (UVLO). Input below the threshold forces
complete micropower (<0.1µA) shutdown.
3 FB Feedback (Input): Regulates FB pin to 0.8V. See “Applications Information”
for resistor divider calculations.
4 CSH Current-Sense High (Input): Current limit comparator non-inverting input. A
built-in offset of 75mV between CSH and VOUT pins in conjunction with the
current-sense resistor set the current limit threshold level. This is also the
non-inverting input to the current sense amplifier.
5 VOUT Current-Sense Low (Input): Output voltage feedback input and inverting
input for the current limit comparator and the current sense amplifier.
6 VIN Unregulated Input (Input): +4.5V to +32V supply input.
7 VDD 5V Internal Linear-Regulator (Output): VDD is the external MOSFET gate
drive supply voltage and internal supply bus for the IC. Bypass to GND with
4.7µF.
8 LSD Low-Side Drive (Output): High-current driver output for low-side N-Channel
MOSFET. Voltage swing is between ground and VDD.
9 GND Ground (Return).
10 BST Boost (Input): Provides drive voltage for the high-side MOSFET driver. The
drive voltage is higher than the input voltage by VDD minus a diode drop.
11 VSW Switch (Return): High-side MOSFET driver return.
12 HSD High-Side Drive (Output): High-current driver output for high-side MOSFET.
This node voltage swing is between ground and VIN +5V minus a diode drop.
Ordering Information
Part Number Voltage Temperature Range Package
Standard Pb-Free
MIC2198BML MIC2198YML Adj –40ºC to +125ºC 12-Pin 4x4 MLF™
October 2005 3 MIC2198
MIC2198 Micrel, Inc.
Absolute Maximum Ratings (Note 1)
Analog Supply Voltage (VIN) ........................................+34V
Digital Supply Voltage (VDD) ..........................................+7V
Driver Supply Voltage (BST) ...................................VIN + 7V
Sense Voltage (VOUT, CSH) ...............................7V to –0.3V
Enable Pin Voltage (VEN/UVLO) ....................................... VIN
Power Dissipation (PD)
4×4 MLF™ ..................................665mW @ TA = 85°C
Ambient Storage Temperature (TS) .......... –65°C to +150°C
ESD, Note 3 ................................................................ 2kV
Operating Ratings (Note 2)
Analog Supply Voltage (VIN) ......................... +4.5V to +32V
Output Voltage Range (VOUT) ....................... +0.8V to +6V
Junction Temperature (TJ) ........................ –40°C to +125°C
Package Thermal Resistance
4×4 MLF-12L (θJA) ............................................... 60°C/W
Electrical Characteristics (Note 4)
VIN = VEN = 12V; TJ = 25°C, unless noted, bold values indicate –40°C ≤ TJ ≤ +125°C
Parameter Condition Min Typ Max Units
Feedback Voltage Reference (±1%) 0.792 0.8 0.808 V
Feedback Voltage Reference (±2%) 0.784 0.816 V
Feedback Voltage Reference 4.5V < VIN < 32V, 0 < (VCSH – VOUT) < 60mV (±3%) 0.776 0.824 V
Feedback Bias Current 10 nA
Output Voltage Range 0.8 6 V
Output Voltage Line Regulation VIN = 4.5V to 32V, VCSH – VOUT = 60mV 0.03 %/V
Output Voltage Load Regulation 25mV < (VCSH – VOUT) < 60mV 0.5 %
Input and VDD Supply
Quiescent Current excluding external MOSFET gate drive current 3.5 5 mA
Shutdown Quiescent Current VEN/UVLO = 0V 0.1 5 µA
Digital Supply Voltage (VDD) IL = 0mA to 5mA 4.7 5.0 5.3 V
Undervoltage Lockout VIN upper threshold (turn-on threshold) 4.25 4.4 V
VIN lower threshold (turn-off threshold) 3.95 4.1 V
Enable/UVLO
Enable Input Threshold (rising) 0.6 1.1 1.6 V
UVLO Threshold (rising) 2.2 2.5 2.8 V
Enable Input Current VEN/UVLO = 5V 0.1 5 µA
Current Limit
Current Limit Threshold Voltage (VCSH – VOUT) 55 75 95 mV
Error Amplifier
Transconductance Error Amplifier GM
0.2 mS
Oscillator Section
Oscillator Frequency 450 500 550 kHz
Maximum Duty Cycle 70 76 %
Minimum On-Time 150 200 ns
Frequency Foldback Threshold measured at VOUT pin 0.25 0.40 0.55 V
Foldback Frequency 125 kHz
Gate Drivers
Rise/Fall Time CL = 3000pF 60 ns
Output Driver Impedance source 5 8.5 Ω
sink 3.5 6 Ω
Driver Non-Overlap Time 80 ns
Note 1. Exceeding the absolute maximum rating may damage the device.
Note 2. The device is not guaranteed to function outside its operating rating.
Note 3. Devices are ESD protected; however, handling precautions are recommended. Human body model, 1.5k in series with 100pF.
Note 4. Specification for packaged product only.
MIC2198 Micrel, Inc.
MIC2198 4 October 2005
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
-40 -20 02 04 06 08 0 100120140
TEMPERATURE (°C)
Quiescent Current
vs. Temperature
PWM
UVLO
Shutdown
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4 14 19 24 29 34
SUPPLY VOLTAGE (V)
Quiescent Current
vs. Supply Voltage
PWM
UVLO
Shutdown
9
0.790
0.795
0.800
0.805
0 10 15 20 25 30 35
SUPPLY VOLTAGE (V)
VFB
Line Regulation
5
0.795
0.800
0.805
0.810
0.815
0.820
-40 -20 02 04 06 08 0 100120140
TEMPERATURE (°C)
VFB
vs. Temperature
4.94
4.95
4.96
4.97
4.98
4.99
5.00
5.01
5.02
0 10 15 20 25 30
LOAD CURRENT (mA)
VDD
Load Regulation
5
4.80
4.85
4.90
4.95
5.00
5.05
5.10
5.15
5.20
-40 -20 02 04 06 08 0 100120140
TEMPERATURE (°C)
VDD
vs. Temperature
-10
-8
-6
-4
-2
0
2
4
6
8
10
-40 -20 02 04 06 08 0 100120140
TEMPERATURE (°C)
Oscillator Frequency
vs. Temperature
-0.15
-0.10
-0.05
0.00
0.05
0.10
0.15
0.20
0.25
0 10 15 20 25 30 35
SUPPLY VOLTAGE (V)
Oscillator Frequency
vs. Supply Voltage
5
Typical Characteristics
(VIN = 12V, unless otherwise noted)
64
66
68
70
72
74
76
78
-40 -20 02 04 06 08 0 100120140
TEMPERATURE (°C)
Overcurrent Threshold
vs. Temperature
VIN = 5V
VOUT = 3.3V
RCS = 20mV
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
0 1 2 3 5
OUTPUT CURRENT (A)
Current Limit
Foldback
4
0
1.0
2.0
3.0
4.0
5.0
6.0
0 8 12 16 20 24 28
SUPPLY VOLTAGE (V)
VDD
Line Regulation
324
October 2005 5 MIC2198
MIC2198 Micrel, Inc.
Block Diagram
Gm = 0.2×10-3
Current
Limit
Error
VIN
VDD
CORRECTIVE
RAMP
RESET
Reference
Oscillator
EN/UVLO
27
10
6
12
11
8
9
4
5
3
1
COMP
100k
MIC2198
FB
R2
R1
VOUT
CSH
VDD
BST
CIN
VIN
HSD
Q2
Q1 D1
D2
CBST
COUT
VIN
VOUT
L1
VSW
LSD
PGND
CCOMP
RCOMP
4.7µF
AV = 2
V 0.8V R1
R2
OUT = +1
V 6 V
OUT(max) = .0
Current
RCS
Control
PWM
Logic
Amp
Sense
Amp
Figure 1. Internal Block Diagram
Functional Description
The MIC2198 is a BiCMOS, switched-mode, synchronous
step-down (buck) converter controller. Current-mode control
is used to achieve superior transient line and load regulation.
An internal corrective ramp provides slope compensation
for stable operation above a 50% duty cycle. The controller
is optimized for high-efficiency, high-performance DC-DC
converter applications.
The MIC2198 block diagram is shown above.
The MIC2198 controller is divided into 5 functions.
• Control loop
• Current limit
• Reference, enable and UVLO
• MOSFET gate drive
• Oscillator
Control Loop
The MIC2198 operates in PWM (pulse-width-modulation)
mode. In PWM mode, the synchronous buck converter
forces continuous current to flow in the inductor which also
improves cross regulation of transformer coupled, multiple
output configurations.
PWM Control Loop
The MIC2198 uses current-mode control to regulate the output
voltage. This method senses the output voltage (outer loop)
and the inductor current (inner loop). It uses inductor current
and output voltage to determine the duty cycle of the buck
converter. Sampling the inductor current removes the inductor
from the control loop, which simplifies compensation.
A block diagram of the MIC2198 PWM current-mode control
loop is shown in Figure 2 and the PWM mode voltage and
current waveform is shown in Figure 3. The inductor current
is sensed by measuring the voltage across the resistor, RCS.
MIC2198 Micrel, Inc.
MIC2198 6 October 2005
A ramp is added to the amplified current-sense signal to
provide slope compensation, which is required to prevent
unstable operation at duty cycles greater than 50%.
A transconductance amplifier is used for the error amplifier,
which compares an attenuated sample of the output voltage
with a reference voltage. The output of the error amplifier is
Current
Error
VIN
VDD
CORRECTIVE
PWM
RESET
Oscillator
7
10
6
12
11
8
9
4
5
3
1
COMP
100k
MIC2198
FB
R2
R1
VOUT
CSH
VDD
BST
CIN
VIN
HSD
Q2
Q1
D1
D2
CBST
COUT
VIN
VOUT
L1
VSW
LSD
PGND
CCOMP
RCOMP
4.7µF
SR
Q
CONTROL LOGIC AND
RCS
Gm = 0.2×10-3
AV = 2
V 0.8V R1
R2
OUT = +1
Reference
PULSE-WIDTH MODULATOR
COMPARATOR
RAMP
Amp
Sense
Amp
Figure 2. PWM Operation
VDD
0V
0V
0V
0V
0A
VDD
VIN + VDD
VIN
VSW
Reset
VHSD
VLSD
ILOAD
IL1
Pulse
Figure 3. PWM-Mode Timing
the COMP (compensation) pin, which is compared to the
current-sense waveform in the PWM block. When the current
signal becomes greater than the error signal, the comparator
turns off the high-side drive. The COMP pin (pin 1) provides
access to the output of the error amplifier and allows the use
of external components to stabilize the voltage loop.
October 2005 7 MIC2198
MIC2198 Micrel, Inc.
Current Limit
The MIC2198 output current is detected by the voltage drop
across the external current-sense resistor (RCS in Figure
2.). The current limit threshold is 75mV±25mV. The current-
sense resistor must be sized using the minimum current
limit threshold. The external components must be designed
to withstand the maximum current limit. The current-sense
resistor value is calculated by the equation below:
R55mV
I
CS
OUT(max)
=
The maximum output current is:
I95mV
R
OUT(max)
CS
=
The current-sense pins CSH (pin 4) and VOUT (pin 5) are
noise sensitive due to the low signal level and high input im-
pedance. The PCB traces should be short and routed close
to each other. A small (1nF to 0.1µF) capacitor across the
pins will attenuate high frequency switching noise.
When the peak inductor current exceeds the current limit
threshold, the current limit comparator, in Figure 2, turns off
the high-side MOSFET for the remainder of the cycle. The
output voltage drops as additional load current is pulled from
the converter. When the output voltage reaches approximately
0.4V, the circuit enters frequency-foldback mode and the
oscillator frequency will drop to 125kHz while maintaining the
peak inductor current equal to the nominal 75mV across the
external current-sense resistor. This limits the maximum output
power delivered to the load under a short circuit condition.
Reference, Enable and UVLO Circuits
The output drivers are enabled when the following conditions
are satisfied:
• The VDD voltage (pin 7) is greater than its under-
voltage threshold (typically 4.25V).
• The voltage on the enable pin is greater than the
enable UVLO threshold (typically 2.5V).
The internal bias circuit generates a 0.8V bandgap reference
voltage for the voltage error amplifier and a 5V VDD voltage
for the gate drive circuit. The MIC2198 uses FB (pin 3) for
output voltage sensing.
The enable pin (pin 2) has two threshold levels, allowing
the MIC2198 to shut down in a low current mode, or turn off
output switching in UVLO mode. An enable pin voltage lower
than the shutdown threshold turns off all the internal circuitry
and reduces the input current to typically 0.1µA.
If the enable pin voltage is between the shutdown and UVLO
thresholds, the internal bias, VDD, and reference voltages are
turned on. The output drivers are inhibited from switching and
remain in a low state. Raising the enable voltage above the
UVLO threshold of 2.5V enables the output drivers.
Either of two UVLO conditions will disable the MIC2198 from
switching.
• When the VDD drops below 4.1V
• When the enable pin drops below the 2.5V threshold
MOSFET Gate Drive
The MIC2198 high-side drive circuit is designed to switch
an N-Channel MOSFET. Referring to the block diagram in
Figure 2, a bootstrap circuit, consisting of D2 and CBST, sup-
plies energy to the high-side drive circuit. Capacitor CBST is
charged while the low-side MOSFET is on and the voltage on
the VSW pin (pin 11) is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to
turn the MOSFET on. As the MOSFET turns on, the voltage
on the VSW pin increases to approximately VIN. Diode D2
is reversed biased and CBST floats high while continuing to
keep the high-side MOSFET on. When the low-side switch
is turned back on, CBST is recharged through D2.
The drive voltage is derived from the internal 5V VDD bias
supply. The nominal low-side gate drive voltage is 5V and
the nominal high-side gate drive voltage is approximately
4.5V due the voltage drop across D2. A fixed 80ns delay
between the high- and low-side driver transitions is used
to prevent current from simultaneously flowing unimpeded
through both MOSFETs.
Oscillator
The internal oscillator is free running and requires no external
components. The nominal oscillator frequency is 500kHz. If
the output voltage is below approximately 0.4V, the oscillator
operates in a frequency-foldback mode and the switching
frequency is reduced to 125kHz.
VSS VOUT
TIME
VIN = 7V
fS = 125kHz
VOUT = 0.4V
fS = 500kHz
VOUT = 3.3V
Figure 4. Startup Waveform
Above 0.4V, the switching frequency increases to 500kHz
causing the output voltage to rise a greater rate. The rise
time of the output is dependent on the output capacitance,
output voltage, and load current. The oscilloscope photo in
Figure 4 show the output voltage at startup.
MIC2198 Micrel, Inc.
MIC2198 8 October 2005
Minimum Pulsewidth
The MIC2198 has a specified minimum pulsewidth. This
minimum pulsewidth places a lower limit on the minimum
duty cycle of the buck converter.
Figure 5 shows the minimum output voltage versus input
supply voltage for the MIC2198. For example, for VIN = 15V,
VOUT = 1.65V would be the lowest achievable voltage that
conforms to the minimum-on-time.
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5 9.5 14.5 19.5 24.5 29.5
INPUT VOLTAGE (V)
Figure 5. Minimum Output Voltage
vs. Input Supply Voltage
October 2005 9 MIC2198
MIC2198 Micrel, Inc.
Applications Information
The following applications information includes component
selection and design guidelines.
Inductor Selection
Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak to peak induc-
tor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak to peak ripple
currents will increase the power dissipation in the inductor
and MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple cur-
rent. Smaller peak to peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current.
The inductance value is calculated by the equation below.
LVOUT × (VIN(max) − VOUT)
VIN(max) × fS × 0.2 × IOUT(max)
=
where:
fS = switching frequency
0.2 = ratio of AC ripple current to DC output current
VIN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
IVOUT × (VIN(max) −VOUT)
VIN(max) × fS × L
PP =
The peak inductor current is equal to the average output current
plus one half of the peak to peak inductor ripple current.
I I 0.5 I
PK OUT(max) PP
= + ×
The RMS inductor current is used to calculate the I2×R losses
in the inductor.
I I 1 1
3
I
I
INDUCTOR(rms) OUT(max) P
OUT(max)
2
= × +
Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2198 requires the use of fer-
rite materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the efficiency of the power supply.
This is especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor.
The power dissipated in the inductor is equal to the sum
of the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored. At
lower output currents, the core losses can be a significant
contributor. Core loss information is usually available from
the magnetics vendor.
Copper loss in the inductor is calculated by the equation
below:
PINDUCTORCu = IINDUCTOR(rms)2 × RWINDING
The resistance of the copper wire, RWINDING, increases with
temperature. The value of the winding resistance used should
be at the operating temperature.
RWINDING(hot) = RWINDING(20°C) × 1 + 0.0042 × (T HOT − T20°C)
where:
THOT = temperature of the wire under operating load
T20°C = ambient temperature
RWINDING(20°C) is room temperature winding
resistance (usually specified by the manufacturer)
Current-Sense Resistor Selection
Low inductance power resistors, such as metal film resistors
should be used. Most resistor manufacturers make low induc-
tance resistors with low temperature coefficients, designed
specifically for current-sense applications. Both resistance
and power dissipation must be calculated before the resis-
tor is selected. The value of RSENSE is chosen based on the
maximum output current and the maximum threshold level.
The power dissipated is based on the maximum peak output
current at the minimum overcurrent threshold limit.
R55mV
I
SENSE
OUT(max)
=
The maximum overcurrent threshold is:
I95mV
R
OVERCURRENT(max)
CS
=
The maximum power dissipated in the sense resistor is:
PD(RSENCE) = IOVERCURRENT(max)2 × RCS
MOSFET Selection
External N-Channel logic-level power MOSFETs must be
used for the high- and low-side switches. The MOSFET
gate-to-source drive voltage of the MIC2198 is regulated by
an internal 5V VDD regulator. Logic-level MOSFETs, whose
operation is specified at VGS = 4.5V must be used.
It is important to note the on-resistance of a MOSFET in-
creases with increasing temperature. A 75°C rise in junction
temperature will increase the channel resistance of the MOS-
FET by 50% to 75% of the resistance specified at 25°C. This
change in resistance must be accounted for when calculating
MOSFET power dissipation.
Total gate charge is the charge required to turn the MOSFET
on and off under specified operating conditions (VDS and
VGS). The gate charge is supplied by the MIC2198 gate drive
circuit. At 500kHz switching frequency, the gate charge can
be a significant source of power dissipation in the MIC2198.
At low output load this power dissipation is noticeable as a
reduction in efficiency. The average current required to drive
the high-side MOSFET is:
I Q f
G[high-side](avg) G S
= ×
MIC2198 Micrel, Inc.
MIC2198 10 October 2005
where:
IG[high-side](avg) =
average high-side MOSFET gate current.
QG = total gate charge for the high-side MOSFET
taken from manufacturer’s data sheet
with VGS = 5V.
fs = 500kHz
The low-side MOSFET is turned on and off at VDS = 0 because
the freewheeling diode is conducting during this time. The
switching losses for the low-side MOSFET is usually negli-
gible. Also, the gate drive current for the low-side MOSFET
is more accurately calculated using CISS at VDS = 0 instead
of gate charge.
For the low-side MOSFET:
I C V f
G[low-side](avg) ISS GS S
= × ×
Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2198 due to gate
drive is:
P V I I
GATEDRIVE IN G[high-side](avg) G[low-side](avg)
= +
( )
A convenient figure of merit for switching MOSFETs is the
on-resistance times the total gate charge (RDS(on) × QG).
Lower numbers translate into higher efficiency. Low gate-
charge logic-level MOSFETs are a good choice for use with
the MIC2198. Power dissipation in the MIC2198 package
limits the maximum gate drive current.
Parameters that are important to MOSFET switch selection
are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage rating of the MOSFETs are essentially equal to
the input voltage. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage spikes
due to circuit parasitics.
The power dissipated in the switching transistor is the sum
of the conduction losses during the on-time (PCONDUCTION)
and the switching losses that occur during the period of time
when the MOSFETs turn on and off (PAC).
P P P
SW CONDUCTION AC
= +
where:
P I R
CONDUCTION SW(rms) SW
2
= ×
P P P
AC AC(off) AC(on)
= +
RSW = on-resistance of the MOSFET switch.
Making the assumption the turn-on and turnoff transition times
are equal, the transition time can be approximated by:
tC V C V
I
TISS GS OSS IN
G
=× + ×
where:
CISS and COSS are measured at VDS = 0.
IG = gate drive current (1A for the MIC2198)
The total high-side MOSFET switching loss is:
PAC = (VIN + VD) × IPK × tT ×fS
where:
tT = switching transition time (typically 20ns to 50ns).
VD = freewheeling diode drop, typically 0.5V.
fS it the switching frequency, nominally 500kHz
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
RMS Current and MOSFET Power Dissipation
Calculation
Under normal operation, the high-side MOSFETs RMS cur-
rent is greatest when VIN is low (maximum duty cycle). The
low-side MOSFETs RMS current is greatest when VIN is high
(minimum duty cycle). However, the maximum stress the
MOSFETs see occurs during short circuit conditions, where
the output current is equal to IOVERCURRENT(max). (See the
“Sense Resistor” section). The calculations below are for
normal operation. To calculate the stress under short circuit
conditions, substitute IOVERCURRENT(max) for IOUT(max). Use
the formula below to calculate D under short circuit condi-
tions.
D 0.063 1.8 10 V
SHORTCIRCUIT 3IN
= − × ×
−
The RMS value of the high-side switch current is:
I D I I
12
SW(high side)(rms) OUT(max)
2PP2
−= × +
I 1 D I I
12
SW(low side)(rms) OUT(max)
2PP2
−= −
( )
+
where:
D = duty cycle of the converter
DV
V
OUT
IN
=×η
η = efficiency of the converter.
Converter efficiency depends on component parameters,
which have not yet been selected. For design purposes, an
efficiency of 90% can be used for VIN less than 10V and 85%
can be used for VIN greater than 10V. The efficiency can be
more accurately calculated once the design is complete. If the
assumed efficiency is grossly inaccurate, a second iteration
through the design procedure can be made.
October 2005 11 MIC2198
MIC2198 Micrel, Inc.
For the high-side switch, the maximum DC power dissipa-
tion is:
P R I
SWITCH1(dc) DS(on)1 SW1(rms)2
= ×
For the low-side switch (N-Channel MOSFET), the DC power
dissipation is:
P R I
SWITCH2(dc) DS(on)2 SW 2(rms)2
= ×
Since the AC switching losses for the low-side MOSFET is
near zero, the total power dissipation is:
P P
low-side MOSFET(max) SWITCH2(dc)
=
The total power dissipation for the high-side MOSFET is:
P P P
high sideMOSFET(max) SWITCH 1(dc) AC− = +
External Schottky Diode
An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 80ns The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must
be able to handle the peak current.
I I 2 80ns f
D(avg) OUT S
= × × ×
The reverse voltage requirement of the diode is:
V V
DIODE(rrm) IN
=
The power dissipated by the Schottky diode is:
P I V
DIODE D(avg) F
= ×
where:
VF = forward voltage at the peak diode current
The external Schottky diode, D2, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode
is used, it must be rated to handle the peak and average cur-
rent. The body diode has a relatively slow reverse recovery
time and a relatively high forward voltage drop. The power
lost in the diode is proportional to the forward voltage drop
of the diode. As the high-side MOSFET starts to turn on, the
body diode becomes a short circuit for the reverse recovery
period, dissipating additional power. The diode recovery and
the circuit inductance will cause ringing during the high-side
MOSFET turn-on.
An external Schottky diode conducts at a lower forward voltage
preventing the body diode in the MOSFET from turning on.
The lower forward voltage drop dissipates less power than
the body diode. The lack of a reverse recovery mechanism
in a Schottky diode causes less ringing and less power loss.
Depending on the circuit components and operating condi-
tions, an external Schottky diode will give a 1/2% to 1%
improvement in efficiency.
Output Capacitor Selection
The output capacitor values are usually determined by the
capacitors ESR (equivalent series resistance). Voltage rating
and RMS current capability are two other important factors in
selecting the output capacitor. Recommended capacitors are
tantalum, low-ESR aluminum electrolytics, and OS-CON.
The output capacitor’s ESR is usually the main cause of output
ripple. The maximum value of ESR is calculated by:
RV
I
ESR OUT
PP
≤∆
where:
VOUT = peak-to-peak output voltage ripple
IPP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ESR and the
output capacitance. The total ripple is calculated below:
∆V I (1 D)
C f I R
OUT PP
OUT S
2
PP ESR
2
=× −
×
+ ×
( )
where:
D = duty cycle
COUT = output capacitance value
fS = switching frequency
The voltage rating of capacitor should be twice the output
voltage for a tantalum and 20% greater for an aluminum
electrolytic or OS-CON.
The output capacitor RMS current is calculated below:
II
12
CPP
OUT(rms) =
The power dissipated in the output capacitor is:
PDISS(COUT) = ICOUT(rms)2 × RESR(COUT)
Input Capacitor Selection
The input capacitor should be selected for ripple current rating
and voltage rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the input
supply on. Tantalum input capacitor voltage rating should
be at least 2 times the maximum input voltage to maximize
reliability. Aluminum electrolytic, OS-CON, and multilayer
polymer film capacitors can handle the higher inrush currents
without voltage derating.
The input voltage ripple will primarily depend on the input
capacitors ESR. The peak input current is equal to the peak
inductor current, so:
∆V I R
IN INDUCTOR(peak) ESR(C )
IN
= ×
The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at
the maximum output current. Assuming the peak-to-peak
inductor ripple current is low:
I I D (1 D)
C (rms) OUT(max)
IN ≈ × × −
MIC2198 Micrel, Inc.
MIC2198 12 October 2005
The power dissipated in the input capacitor is:
P I R
DISS(C ) C (rms) ESR(C )
IN IN 2IN
= ×
Voltage Setting Components
The MIC2198 requires two resistors to set the output voltage
as shown in Figure 6.
Error
3
MIC2198
FB
VREF
0.8V
R2
R1
Amp
Figure 6. Voltage-Divider Configuration
The output voltage is determined by the equation:
V V 1 R1
R2
O REF
= × +
Where: VREF for the MIC2198 is typically 0.8V.
A typical value of R1 can be between 3k and 10k. If R1 is
too large it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small in value it will decrease
the efficiency of the power supply, especially at low output
loads.
Once R1 is selected, R2 can be calculated using:
R2 V R1
V V
REF
O REF
=×
−
Voltage Divider Power Dissipation
The reference voltage and R2 set the current through the
voltage divider.
IV
R2
DIVIDER REF
=
The power dissipated by the divider resistors is:
P (R1 R2) I
DIVIDER DIVIDER
2
= + ×
Efficiency Calculation and Considerations
Efficiency is the ratio of output power to input power. The
difference is dissipated as heat in the buck converter. Under
light output load, the significant contributors are:
• Supply current to the MIC2198
• MOSFET gate-charge power (included in the IC
supply current)
• Core losses in the output inductor
To maximize efficiency at light loads:
• Use a low gate-charge MOSFET or use the small-
est MOSFET, which is still adequate for maximum
output current.
• Use a ferrite material for the inductor core, which
has less core loss than an MPP or iron power
core.
Under heavy output loads the significant contributors to power
loss are (in approximate order of magnitude):
• Resistive on-time losses in the MOSFETs
• Switching transition losses in the MOSFETs
• Inductor resistive losses
• Current-sense resistor losses
• Input capacitor resistive losses (due to the capaci-
tors ESR)
To minimize power loss under heavy loads:
• Use logic-level, low on-resistance MOSFETs.
Multiplying the gate charge by the on-resistance
gives a figure of merit, providing a good balance
between low and high load efficiency.
• Slow transition times and oscillations on the voltage
and current waveforms dissipate more power during
turn-on and turnoff of the MOSFETs. A clean layout
will minimize parasitic inductance and capacitance
in the gate drive and high current paths. This will
allow the fastest transition times and waveforms
without oscillations. Low gate-charge MOSFETs
will transition faster than those with higher gate-
charge requirements.
• For the same size inductor, a lower value will
have fewer turns and therefore, lower winding re-
sistance. However, using too small of a value will
require more output capacitors to filter the output
ripple, which will force a smaller bandwidth, slower
transient response and possible instability under
certain conditions.
• Lowering the current-sense resistor value will
decrease the power dissipated in the resistor.
However, it will also increase the overcurrent
limit and will require larger MOSFETs and inductor
components.
• Use low-ESR input capacitors to minimize the
power dissipated in the capacitors ESR.
Decoupling Capacitor Selection
The 4.7µF decoupling capacitor is used to minimize noise on
the VDD pin. The placement of this capacitor is critical to the
proper operation of the IC. It must be placed right next to the
pins and routed with a wide trace. The capacitor should be a
good quality tantalum. An additional 1µF ceramic capacitor
may be necessary when driving large MOSFETs with high
gate capacitance. Incorrect placement of the VDD decoupling
capacitor will cause jitter or oscillations in the switching wave-
form and large variations in the overcurrent limit.
A 0.1µF ceramic capacitor is required to decouple the VIN.
The capacitor should be placed near the IC and connected
directly to between pin 6 (VIN) and pin 9 (GND).
October 2005 13 MIC2198
MIC2198 Micrel, Inc.
PCB Layout and Checklist
PCB layout is critical to achieve reliable, stable and efficient
performance. A ground plane is required to control EMI and
minimize the inductance in power, signal and return paths.
The following guidelines should be followed to insure proper
operation of the circuit.
• Signal and power grounds should be kept separate
and connected at only one location. Large currents
or high di/dt signals that occur when the MOSFETs
turn on and off must be kept away from the small
signal connections.
• The connection between the current-sense resis-
tor and the MIC2198 current-sense inputs (pin 4
and 5) should have separate traces, through a
10Ω resistor on each pin. The traces should be
routed as closely as possible to each other and
their length should be minimized. Avoid running
the traces under the inductor and other switching
components. The 10Ω resistor should be placed
close as possible to pins 4 and 5 on the MIC2198
and a 1nF to 0.1µF capacitor placed between pins
4 and 5 will help attenuate switching noise on the
current sense traces. This capacitor should be
placed close to pins 4 and 5.
• When the high-side MOSFET is switched on, the
critical flow of current is from the input capacitor
through the MOSFET, inductor, sense resistor,
output capacitor, and back to the input capacitor.
These paths must be made with short, wide pieces
of trace. It is good practice to locate the ground
terminals of the input and output capacitors close
to each.
• When the low-side MOSFET is switched on, cur-
rent flows through the inductor, sense resistor,
output capacitor, and MOSFET. The source of the
low-side MOSFET should be located close to the
output capacitor.
• The freewheeling diode, D1 in Figure 2, conducts
current during the dead time, when both MOSFETs
are off. The anode of the diode should be located
close to the output capacitor ground terminal and
the cathode should be located close to the input
side of the inductor.
• The 4.7µF capacitor, which connects to the VDD
terminal (pin 7) must be located right at the IC. The
VDD terminal is very noise sensitive and placement
of this capacitor is very critical. Connections must
be made with wide trace. The capacitor may be
located on the bottom layer of the board and con-
nected to the IC with multiple vias.
• The VIN bypass capacitor should be located close
to the IC and connected between pins 6 and 9.
Connections should be made with a ground and
power plane or with short, wide trace.
MIC2198 Micrel, Inc.
MIC2198 14 October 2005
Package Information
12-Pin 4×4 MLF™ (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2003 Micrel, Inc.
