LM5072MHX-80 - National Semiconductor

LM5072

LM5072 Integrated 100V Power Over Ethernet PD Interface and PWM Controller

with Aux Support

Literature Number: SNVS437C

October 30, 2007

LM5072

Integrated 100V Power Over Ethernet PD Interface and

PWM Controller with Aux Support

General Description

The LM5072 Powered Device (PD) interface and Pulse-

Width-Modulation (PWM) controller provides a complete pow-

er solution, fully compliant to IEEE 802.3af, for the PD

connecting into Power over Ethernet (PoE) networks. This

controller integrates all functions necessary to implement

both a PD powered interface and DC-DC converter with a

minimum number of external components. The LM5072 pro-

vides the flexibility for the PD to also accept power from

auxiliary sources such as AC adapters in a variety of config-

urations. The low RDS(ON) PD interface hot swap MOSFET

and programmable DC current limit extend the range of

LM5072 applications up to twice the power level of 802.3af

compliant PD devices. The 100V maximum voltage rating

simplifies selection of the transient voltage suppressor that

protects the PD from network transients. The LM5072 in-

cludes an easy-to-use PWM controller that facilitates the

various single-ended power supply topologies including the

flyback, forward and buck. The PWM control scheme is based

on peak current mode control, which provides inherent ad-

vantages including line feed-forward, cycle-by-cycle current

limit, and simplified feedback loop compensation. Two ver-

sions of the LM5072 provide either an 80% maximum duty

cycle (-80 suffix), or a 50% maximum duty cycle (-50 suffix).

Features

PD Interface

■Fully Compliant IEEE 802.3af PD Interface

■Versatile Auxiliary Power Options

■9V Minimum Auxiliary Power Operating Range

■100V Maximum Input Voltage Rating

■Programmable DC Current Limit Up To 800mA

■100V, 0.7Ω Hot Swap MOSFET

■Integrated PD Signature Resistor

■Integrated PoE Input UVLO

■Programmable Inrush Current Limit

■PD Classification Capability

■Power Good Indicator

■Thermal Shutdown Protection

PWM Controller

■Current Mode PWM Controller

■100V Start-up Regulator

■Error Amplifier with 2% Voltage Reference

■Supports Isolated and Non-Isolated Applications

■Programmable Oscillator Frequency

■Programmable Soft-Start

■800 mA Peak Gate Driver

■80% Maximum Duty Cycle with Built-in Slope

Compensation (-80 device)

■50% Maximum Duty Cycle, No Slope Compensation (-50

device)

Applications

■IEEE 802.3af Compliant PoE Powered Devices

■Non-Compliant, Application Specific Devices

■Higher Power Ethernet Powered Devices

Packages

■TSSOP-16 EP (Exposed Pad)

Simplified Application Diagram

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LM5072 Integrated 100V Power Over Ethernet PD Interface and PWM Controller with Aux Support

Connection Diagram

20184602

16 Lead TSSOP-EP

Ordering Information

Order Number Description NSC Package Type / Drawing Supplied As

LM5072MH-50 50% Duty Cycle Limit TSSOP-16EP/MXA16A 92 units per rail

LM5072MHX-50 50% Duty Cycle Limit TSSOP-16EP/MXA16A 2500 units on tape and reel

LM5072MH-80 80% Duty Cycle Limit TSSOP-16EP/MXA16A 92 units per rail

LM5072MHX-80 80% Duty Cycle Limit TSSOP-16EP/MXA16A 2500 units on tape and reel

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LM5072

Pin Descriptions

Pin Number Name Description

1 RT PWM controller oscillator frequency programming pin.

2 SS Soft-start programming pin.

3 VIN Positive supply pin for the PD interface and the internal PWM controller start-up regulator.

4 RCLASS PD classification programming pin.

5 ICL_FAUX Inrush current limit programming pin; also the front auxiliary power enable pin.

6 DCCL PD interface DC current limit programming pin.

7 VEE Negative supply pin for PD interface; connected to PoE and/or front auxiliary power return path.

8 RTN PWM controller power return; connected to the drain of the internal PD interface hot swap

MOSFET; should be externally connected to the reference ground of the PWM controller.

9 OUT PWM controller gate driver output pin.

10 VCC PWM controller start-up regulator output pin.

11 nPGOOD PD interface Power Good indicator and delay timer pin; active low state indicates PoE interface

is in normal operation.

12 CS PWM controller current sense input pin.

13 RAUX Rear auxiliary power enable pin; can be programmed for auxiliary power dominance over PoE

power.

14 FB PWM controller voltage feedback pin and inverting input of the internal error amplifier; connect

to ARTN to disable the error amplifier in isolated dc-dc converter applications.

15 COMP Output of the internal error amplifier and control input to the PWM comparator. In isolated

applications, COMP is controlled by the secondary side error amplifier via an opto-coupler.

16 ARTN PWM controller reference ground pin; should be shorted externally to the RTN pin as a single

point ground connection to improve noise immunity.

EP Exposed metal pad on the underside of the device. It is recommended to connect this pad to a

PC Board plane connected to the VEE pin to improve heat dissipation.

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LM5072

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN , RTN to VEE (Note 7) -0.3V to 100V

RAUX to ARTN -0.3V to 100V

ICL_FAUX to VEE -0.3V to 100V

DCCL, RCLASS to VEE -0.3V to 7V

nPGOOD to ARTN -0.3V to 16V

ARTN to RTN -0.3V to 0.3V

VCC, OUT to ARTN -0.3V to 16V

CS, FB, RT to ARTN -0.3V to 7V

COMP, SS to ARTN -0.3V to 5.5V

ESD Rating

Human Body Model (Note 2) 2000V

Lead Soldering Temp. (Note 3)

Wave (4 seconds)

Infrared (10 seconds)

Vapor Phase (75 seconds)

260°C

240°C

219°C

Storage Temperature -55°C to 150°C

Junction Temperature 150°C

Operating Ratings

VIN voltage 9V to 70V

External voltage applied to VCC 8V to 15V

Operating Junction Temperature -40°C to 125°C

Electrical Characteristics (Note 4) Specifications in standard type face are for TJ = +25°C and those in

boldface type apply over the full operating junction temperature range. Unless otherwise specified: VIN = 48V, FOSC = 250kHz.

Symbol Parameter Conditions Min Typ Max Units

Detection and Classification

VIN Signature Startup Voltage 1.5 V

Signature Resistance 23.25 24.5 26 kΩ

Signature Resistor Disengage/

Classification Engage

VIN Rising 11.0 12.0 12.6 V

Hysteresis 1.9 V

Classification Current Turn Off VIN Rising 22 23.5 25 V

Classification Voltage 1.213 1.25 1.287 V

Supply Current During Classification VIN = 17V 0.7 1.1 mA

Line Under Voltage Lock-Out

UVLO Release VIN Rising 36 38.5 40 V

UVLO Lock out VIN Falling 29.5 31.0 32.5 V

UVLO Hysteresis 6 V

UVLO Filter 300 µs

Power Good

VDS Required for Power Good Status 1.3 1.5 1.7 V

VDS Hysteresis of Power Good Status 0.8 1.0 1.2 V

VGS Required for Power Good Status 4.5 5.5 6.5 V

Default Delay Time of Loss-of Power

Good Status

30 µs

nPGOOD current Source 45 55 65 µA

nPGOOD Pull Down Resistance 130 250 Ω

nPGOOD Threshold 2.3 2.5 2.7 V

Hot Swap

RDS(ON) Hot Swap MOSFET Resistance 0.7 1.2 Ω

Hot Swap MOSFET Leakage 110 µA

Default Inrush Current Limit VDS = 4.0V 120 150 180 mA

Default DC Current Limit VDS = 4.0V 380 440 510 mA

Front Auxiliary DC Current Limit VDS = 4.0V 470 540 610 mA

Inrush Current Limit Programming

Accuracy

VDS = 4.0V -15 15 %

DC Current Limit Programming Accuracy VDS = 4.0V -12 12 %

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LM5072

Symbol Parameter Conditions Min Typ Max Units

Auxiliary Power Option

ICL_FAUX Threshold ICL_FAUX Pin Rising 8.1 8.7 9.3 V

ICL_FAUX Pull Down Current 50 µA

RAUX Lower Threshold (I = 22 µA) RAUX Pin Rising 2.3 2.5 3.0 V

RAUX Lower Threshold Hysteresis 0.8 V

RAUX Upper Threshold (I = 250 µA) RAUX Pin Rising 5.4 6.0 6.9 V

RAUX Lower Threshold Current 16 22 28 µA

RAUX Upper Threshold Current 187 250 313 µA

VCC Regulator

VccReg VCC Regulation (VccReg) 7.4 7.7 8V

VCC Current Limit 15 mA

VCC UVLO (Rising) VccReg

– 210 mV

VccReg

– 100 mV

VCC UVLO (Falling) 5.9 6.2 6.5 V

VIN Supply Current VCC = 10V 2.0 mA

Supply Current (Icc) VCC = 10V 3mA

VCC Regulator Dropout VIN – VCC (Note 6) 6.5 V

Error Amplifier

Gain Bandwidth 3 MHz

DC Gain 67 dB

Input Voltage 1.225 1.275 V

COMP Sink Capability 510 mA

Current Limit

ILIM Delay to Output 30 ns

Cycle by Cycle Current Limit Threshold

Voltage

0.45 0.5 0.55 V

Leading Edge Blanking Time 65 ns

CS Sink Impedance (clocked) 35 55 Ω

Soft-start

Soft-start Current Source 810 12 µA

Oscillator

Frequency1 (RT = 26.1 kΩ) 175 200 225 KHz

Frequency2 (RT = 8.7 kΩ) 515 580 645 KHz

Sync threshold 2.6 3.2 3.8 V

PWM Comparator

Delay to Output 25 ns

Min Duty Cycle 0%

Max Duty Cycle (-80 Device) 75 80 85 %

Max Duty Cycle (-50 Device) 47 50 53 %

COMP to PWM Comparator Gain 0.33

COMP Open Circuit Voltage 4.3 5.2 6.1 V

COMP Short Circuit Current 0.6 1.0 1.4 mA

Slope Compensation (LM5072-80 Device Only)

Slope Comp Amplitude 70 90 110 mV

Output Section

Output High Saturation 0.25 0.75 V

Output Low Saturation 0.25 0.75 V

trRise time CLOAD = 1nF 18 ns

tfFall time CLOAD = 1nF 15 ns

PDI Thermal Shutdown (Note 5)

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LM5072

Symbol Parameter Conditions Min Typ Max Units

Thermal Shutdown Temp. 165 °C

Thermal Shutdown Hysteresis 25 °C

Thermal Resistance

θJA Junction to Ambient MXA package 40 °C/W

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the

device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.

Note 3: For detailed information on soldering the plastic TSSOP package, refer to the Packaging Databook available from National Semiconductor.

Note 4: Minimum and Maximum limits are guaranteed through test, design, or statistical correlation using Statistical Quality Control (SQC) methods. Typical

values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purpose only. Limits are used to calculate National’s Average

Outgoing Quality Level (AOQL).

Note 5: Device thermal limitations may limit usable range.

Note 6: The VCC regulator is intended for use solely as a bias supply for the LM5072, dropout assumes 3mA of external VCC current.

Note 7: During rear auxiliary operation, the RTN pin can be approximately -0.4V with respect to VEE. This is caused by normal internal bias currents, and will

not harm the device. Application of external voltage or current must not cause the absolute maximum rating to be exceeded.

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LM5072

Typical Performance Characteristics

UVLO Threshold vs Temperature

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Default Current Limit vs Temperature

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Inrush Current Limit vs ICL_FAUX Resistor

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DC Current Limit vs. DCCL Resistor

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Programmed DC Current Limit vs Temperature

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Oscillator Frequency vs Temperature

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LM5072

Oscillator Frequency vs RT Resistance

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Error Amplifier Reference Voltage vs Temperature

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VCC vs ICC

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Input Current vs Input Voltage

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Maximum Duty Cycle vs Temperature

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Soft-start Current vs Temperature

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LM5072

Specialized Block Diagrams

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FIGURE 1. Top Level Block Diagram

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LM5072

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FIGURE 2. PWM Controller Block Diagram

Description of Operation and

Applications Information

The LM5072 integrates a fully IEEE 802.3af compliant PD in-

terface and PWM controller in a single integrated circuit,

providing a complete and low cost power solution for devices

that connect to PoE systems. The implementation requires a

minimal number of external components.

The LM5072’s Hot Swap PD interface provides four major

advantages:

1. An input voltage rating up to 100V that allows greater

flexibility when selecting a transient surge suppressor to

protect the PD from voltage transients encountered in

PoE applications.

2. The integration of the PD signature resistor and other

functions including programmable inrush current limit,

input voltage under-voltage lock-out (UVLO), PD

classification, and thermal shutdown simplifies PD

implementation.

3. The PD interface and PWM controller accept power from

auxiliary sources including AC adapters and solar cells

in various configurations and over a wide range of input

voltages. Auxiliary power input can be programmed to be

dominant over PoE power.

4. DC current limit is programmable and adjustable to

support PoE applications requiring input currents up to

700 mA.

The LM5072 includes an easy to use PWM controller based

on the peak current mode control technique. Current mode

control provides inherent advantages such as line voltage

feed-forward, cycle-by-cycle current limit, and simplified

closed-loop compensation. The controller’s PWM gate driver

is capable of sourcing and sinking peak currents of 800 mA

to directly drive the power MOSFET switch of the DC-DC

converter. The PWM controller also contains a high gain, high

bandwidth error amplifier, a high voltage startup bias regula-

tor, a programmable oscillator for a switching frequency be-

tween 50 kHz to 500 kHz, a bias supply (VCC) under-voltage

lock-out circuit, and a programmable soft-start circuit. These

features greatly simplify the design and implementation of

single ended topologies like the flyback, forward and buck.

The LM5072 is available in two versions, the LM5072-50 and

LM5072-80. As indicated in the suffix of the part number, the

maximum duty cycle of each device is limited to 50% and

80%, respectively. Internal PWM controller slope compensa-

tion is provided in the LM5072-80 version.

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LM5072

Modes of Operation

Per the IEEE 802.3af specification, when a PD is connected

to a PoE system it transitions through several operating

modes in sequence including detection, classification (op-

tional), turn on, normal operation, and power removal. Each

operating mode corresponds to a specific PoE voltage range

fed through the Ethernet cable. Figure 3 shows the IEEE

802.3af specified sequence of operating modes and the cor-

responding PD input voltages.

Current steering diode-bridges are required for the PD inter-

face to accept all allowable connections and polarities of PoE

voltage from the RJ-45 connector (see the example applica-

tion circuits in Figures 18, 19, 20 and 21). The bridge will

cause some reduction of the input voltage sensed by the

LM5072. To guarantee full compliance to the specification in

all operating modes, the LM5072 takes into account the volt-

age drop across the bridge diodes and responds appropri-

ately to the voltage received from the PoE cable. Table 1

presents the response in each operating mode to voltages

across the VIN and VEE pins.

20184617

FIGURE 3. Sequence of PoE Operating Modes

TABLE 1. Operating Modes With Respect To Input Voltage

Mode of Operation Voltage from PoE Cable per

IEEE 802.3af

LM5072 Input Voltage

(VIN pin to VEE pin)

Detection (Signature) 2.7V to 10.0V 1.5V to 10.0V

Classification 14.5V to 20.5V 12V to 23.5V

Startup 42V max 38V (UVLO Release, VIN Rising)

Normal Operation 57V to 36V 70V to 32V (UVLO, VIN Falling)

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LM5072

Detection Signature

During detection mode, a PD must present a signature resis-

tance between 23.75 kΩ and 26.25 kΩ to the PoE power

sourcing equipment. This signature impedance distinguishes

the PD from non-PoE capable equipment to protect the latter

from being accidentally damaged by inadvertent application

of PoE voltage levels. To simplify the circuit implementation,

the LM5072 integrates the 24.5 kΩ signature resistor, as

shown in Figure 4.

20184618

FIGURE 4. Detection Circuit With Integrated PD Signature

Resistor

During detection mode, the voltage across the VIN and VEE

pins is less than 10V. Once signature mode is complete, the

LM5072 will disengage the signature resistor to reduce power

loss in all other modes.

Classification

Classification is an optional feature of the IEEE802.3af spec-

ification. It is primarily used to identify the power requirements

of a particular PD device. This feature will allow the PSE to

allocate the appropriate available power to each device on the

network. Classification is performed by measuring the current

flowing into the PD during this mode. IEEE 802.3af specifies

five power classes, each corresponding to a unique range of

classification current, as presented in Table 2. The LM5072

simplifies the classification implementation by requiring a sin-

gle external resistor connected between the RCLASS and

VEE pins to program the classification current. The resistor

value required for each class is also given in Table 2.

TABLE 2. Classification Levels and Required External Resistor Value

Class PD Max Power Level ICLASS Range LM5072

RCLASS Value

From To From To

0 (Default) 0.44W 12.95W 0 mA 4 mA Open

1 0.44W 3.84W 9 mA 12 mA 130Ω

2 3.84W 6.49W 17 mA 20 mA 71.5Ω

3 6.49W 12.95W 26 mA 30 mA 46.4Ω

4 Reserved Reserved 36 mA 44 mA 31.6Ω

20184619

FIGURE 5. PD Classification – Fulfilled With a Single External Resistor

Figure 5 shows the LM5072’s implementation of PD classifi-

cation using an external resistor connected to the RCLASS

pin. During classification, the voltage across the VIN and VEE

pins is between 13V and 23.5V. In this voltage range, the

class resistor RCLASS is engaged by enabling the 1.25V

buffer amplifier and MOSFET. After classification is complete,

the voltage from the PSE will increase to the normal operating

voltage of the PoE system (48V nominal). When VIN rises

above 23.5V, the LM5072 will disengage the RCLASS resis-

tor to reduce on-chip power dissipation.

The classification feature is disabled when either the front or

rear auxiliary power options are selected, as the classification

function is not required when power is supplied from an aux-

iliary source. The classification function is also disabled when

the LM5072 reaches the thermal shutdown temperature

threshold (nominally 165°C). This may occur if the LM5072 is

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LM5072

operated at elevated ambient temperatures and the classifi-

cation time exceeds the IEEE802.3af limit of 75 ms.

When the classification option is not required, simply leave

the RCLASS pin open to set the PD to the default Class 0

state. Class 0 requires that the PSE allocate the maximum

IEEE802.3af specified power of 15.4 W (12.95 W at the PD

input terminals) to the PD.

Undervoltage Lockout (UVLO)

The LM5072’s internal preset UVLO circuit continuously mon-

itors the PoE input voltage between the VIN and VEE pins.

When the VIN voltage rises above 38V nominal, the UVLO

circuit will release the hot swap MOSFET and initiate the

startup inrush sequence. When the VIN voltage falls below

31V nominal during normal operating mode, the LM5072 dis-

ables the PD by shutting off the hot swap MOSFET.

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FIGURE 6. Preset Input UVLO Function

Figure 6 illustrates the block diagram of the LM5072 UVLO

circuit. This function requires no external components. The

UVLO signal can be over-ridden by the front auxiliary power

option (see details in the FAUX section) to allow the hot swap

MOSFET of the LM5072 to pass power from front auxiliary

power sources at voltage levels below the PoE operating volt-

age. In the rear auxiliary power application (see RAUX sec-

tion), the auxiliary power source bypasses the hot swap

MOSFET and is applied directly to the input of the DC-DC

converter. The UVLO function does not need to be over-rid-

den in this configuration.

The PD can draw a maximum current of 400 mA during stan-

dard 802.3af PoE operation. This current will cause a voltage

drop of up to 8V over a 100m long Ethernet cable. The PD

front-end current steering diode bridges may introduce an ad-

ditional 2V drop. In order to guarantee successful startup at

the minimum PoE voltage of 42V, and to continue operation

at the minimum requirement of 36V as specified by IEEE

802.3af, these voltage drops must be taken into account.

Therefore, the LM5072 UVLO thresholds have been set to

38V on the rising edge of VIN, and 31V on the falling edge of

VIN. The 7V nominal hysteresis of the UVLO function, in ad-

dition to the inrush current limit (discussed in the next section),

prevents false starts and chattering during startup.

Inrush Current Limit Programming

According to IEEE 802.3af, the input capacitance of the PD

power supply must be at least 5 µF (between the VIN and RTN

pins). Considering the capacitor tolerance and the effects of

voltage and temperature, a nominal capacitor value of at least

10 µF is recommended to ensure 5 µF minimum under all

conditions. A greater amount of capacitance may be needed

to filter the input ripple of the DC-DC converter. The input ca-

pacitors remain discharged during detection and classifica-

tion modes of the PD interface. The hot swap MOSFET is

turned on after the VIN minus VEE voltage difference rises

above the UVLO release threshold of 38V nominal. When

enabled, the hot swap MOSFET delivers a regulated inrush

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LM5072

current to charge the input capacitors of the DC-DC converter.

To prevent excessive inrush current, the LM5072 will turn on

the hot swap MOSFET in a constant current mode. The de-

fault, pre-programmed inrush current of 150 mA can be se-

lected by simply leaving the ICL_FAUX pin open.

To adjust the capacitor charging time for a particular applica-

tion requirement, the inrush limit can be programmed to any

value between 150 and 400 mA with an external resistor

(RICL) between the ICL_FAUX and VEE pins, as shown in

Figure 7. The relationship between the RICL value and the

desired inrush current limit IINRUSH satisfies the following

equation:

20184622

FIGURE 7. Input Inrush Limit Programming via RICL

The inrush current causes a voltage drop along the PoE Eth-

ernet cable (20Ω maximum) that reduces the input voltage

sensed by the LM5072. To avoid erratic turn-on (hiccups),

IINRUSH should be programmed such that the input voltage

drop due to cable resistance does not exceed the VIN-UVLO

hysteresis (6V minimum).

DC Current Limit Programming

The LM5072 provides a default DC current limit of 440 mA

nominal. This default limit can be selected by leaving the DC-

CL pin open.

The LM5072 allows the DC current limit to be programmed

within the range from 150 mA to 800 mA. Figure 8 shows the

method to program the DC current limit with an external re-

sistor, RDCCL. The relationship between the RDCCL value and

the desired DC current limit IDC satisfies the following equa-

tion:

20184624

FIGURE 8. Input DC Current Limit Programming via RDCCL

The maximum recommended DC current limit is 800 mA.

While thermal analysis should be a standard part of the mod-

ule development process, it may warrant additional attention

if the DC current limit is programmed to values in excess of

400 mA. This analysis should include evaluations of the dis-

sipation capability of LM5072 package, heat sinking proper-

ties of the PC Board, ambient temperature, and other heat

dissipation factors of the operating environment.

Power Good and Regulator Startup

The Power Good status indicates that the circuit is ready for

PWM controller startup to occur. It is established when the

input capacitors are fully charged through the hot swap MOS-

FET. Since the hot swap MOSFET is in series with the input

capacitors of the DC-DC converter, its drain-to-source volt-

age decreases as the charging occurs. Power Good is indi-

cated when the following two conditions are met: the

MOSFET drain-to-source voltage drops below 1.5V (with 1V

hysteresis), and the gate-to-source voltage is greater than 5V.

Circuitry internal to the LM5072 monitors both the drain and

gate voltages (see Figure 1), and issues the Power Good sta-

tus flag by pulling down the nPGOOD pin to a logic low level

relative to the ARTN pin.

The nPGOOD circuitry consists of a 2.5V comparator, a

130Ω pull down MOSFET, and a 50 µA pull up current source,

as shown in Figure 9. Once the Power Good status is estab-

lished, the nPGOOD pin voltage will be pulled down quickly

by the MOSFET, and the PWM controller will start as soon as

the nPGOOD pin voltage drops below the 2.5V threshold.

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LM5072

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FIGURE 9. "Powered-from-PoE" Indictor and Power Good Delay Timer

The nPGOOD pin can be configured to perform multiple func-

tions. As shown in Figure 9, it can be used to implement a

“Powered from PoE” indicator using an LED with a series cur-

rent limiting resistor connected to the VCC pin. This may be

useful when the auxiliary power source is directly connected

to the DC-DC converter stage, a situation known as

“RAUX” (see Auxiliary Power Options below). In such a con-

figuration, the nPGOOD pin will be active when the PD is

operating from PoE power but not when it is powered from the

auxiliary source. However, the “Powered from PoE” indicator

is not applicable in systems implementing the front auxiliary

power configuration “FAUX” (see Auxiliary Power Options be-

low) because both PoE and auxiliary supply current pass

through the hot swap MOSFET. In this configuration, the nP-

GOOD pin is active when either PoE power or auxiliary power

is applied. The designer should ensure that the current drawn

by the LED is not more than a few milliamps, as the VCC

regulator’s output current is limited to 15 mA and must also

supply the LM5072’s bias current and external MOSFET’s

gate charging current. Supplying an external VCC that is high-

er than the regulated level with a bench supply is an easy way

to measure VCC load during normal operation. It should also

be noted that an external load on the VCC line will increase

the dropout voltage of the VCC regulator. This may be a con-

cern when operating from a low voltage rear auxiliary supply.

The nPGOOD pin can also be used to implement a delay timer

by adding a capacitor from the nPGOOD pin to the ARTN pin.

This delay timer will prevent the interruption of the PWM

controller’s operation in the event of an intermittent loss of

Power Good status. This can be caused by PoE line voltage

transients that may occur when switching between normal

PoE power and a backup supply system (e.g. a battery or

UPS). This condition will create a new “hot swap” event if

there is a voltage difference between the backup supply and

PoE supply. Since the hot swap MOSFET will likely limit cur-

rent during such a sudden input voltage change, the nP-

GOOD pin will momentarily switch to the ”pull up” state. A

capacitor on this pin will delay the transition of the nPGOOD

pin state in order to provide continuous operation of the PWM

controller during such transients. The Power Good filter delay

time and capacitor value can be selected with the following

equation:

For example, selecting 1000 nF for CPGOOD, the delay time

will be 50 ms if no LED is used and about 0.83 ms when an

LED, drawing 3 mA, is used. The delay required for continued

operation will depend on the amplitude of the transient, the

DC current limit, the load, and the total amount of input ca-

pacitance. Note that this delay does not guarantee continued

operation. If the hot swap MOSFET is in current limit for an

extended period, it may cause a thermal limit condition. This

will result in a complete shutdown of the switching regulator,

though no elements in the system will be permanently dam-

aged and normal operation will resume momentarily.

The Power Good status will also affect the default DC current

limit. Should the sensed drain to source voltage of the hot

swap MOSFET (from ARTN to VEE) exceed 2.5V, the LM5072

will increase the DC current limit from the default 440 mA to

540 mA, thus allowing the PD to continue operation through

the transient event. This higher current limit will remain in ef-

fect until one of the following events occur: (i) the duration of

loss of Power Good status exceeds tPG_Delay, at which time

the PWM controller will be disabled, (ii) the increased power

dissipation in the hot swap MOSFET causes a thermal limit

condition as previously discussed, or (iii) the MOSFET drain

to source voltage falls below 1.5V to re-establish Power Good

status. Under this condition, the LM5072 will revert back to

the default 440 mA DC current limit once Power Good status

is restored. Note that if the DC current limit has been pro-

grammed externally with RDCCL (see the DC current limit

section), the DC current limit will remain at the programmed

level even when the Power Good status is lost.

Auxiliary Power Options

The LM5072 based PD can receive power from auxiliary

sources like AC adapters and solar cells in addition to the PoE

enabled network. This is a desirable feature when the total

system power requirements exceed the PSE’s load capacity.

Furthermore, with the auxiliary power option the PD can be

used in a standard Ethernet (non-PoE) system.

For maximum versatility, the LM5072 accepts two different

auxiliary power configurations. The first one, shown in Figure

15 www.national.com

LM5072

10, is the front auxiliary (FAUX) configuration in which the

auxiliary source is “diode OR’d” with the PoE potential re-

ceived from the Ethernet connector. The second configura-

tion, shown in Figure 11, is the rear auxiliary (RAUX) option

in which the auxiliary power bypasses the PoE interface and

is connected directly to the input of the DC-DC converter

through a diode. The FAUX option is desirable if the auxiliary

power voltage is similar to the PoE input voltage. However,

when the auxiliary supply voltage is much lower than the PoE

input voltage, the RAUX option is more favorable because the

current from the auxiliary supply is not limited by the hot swap

MOSFET DC current limit. A comparison of the FAUX and

RAUX options is presented in Table 3. Note the ICL_FAUX

and RAUX pins are not reverse voltage protected. If complete

reverse protection is desired, series blocking diodes are nec-

essary.

20184627

FIGURE 10. The FAUX Configuration

20184628

FIGURE 11. The RAUX Configuration

www.national.com 16

LM5072

TABLE 3. Comparison Between FAUX and RAUX Operation

Tradeoff FAUX Operation RAUX Operation

Hot Swap Protection / Current Limit

Protection

Automatically provided by the hot swap

MOSFET.

Requires a series resistor to limit the inrush

current during hot swap.

Minimum Auxiliary Voltage

(at the IC pins)

Limited to 18V by the signature

detection mode, or by the power

requirement (current limit).

Only limited by 9V minimum input requirement.

Auxiliary Dominance Over PoE Cannot be forced without external

components.

Can be forced with appropriate RAUX pin

configuration.

Use of nPGOOD Pin as “Powered from

PoE” Indicator

Not applicable as power is delivered

through the hot swap interface in both

PoE and FAUX modes.

Supported.

Transient Protection Excellent due to active MOSFET

current limit.

Fair due to passive resistor current limit.

The term “Auxiliary Dominance” mentioned in Table 3 means

that when the auxiliary power source is connected, it will al-

ways power the PD regardless of the state of PoE power. “Aux

dominance” is achievable only with the RAUX option, as not-

ed in the table.

If the PD is not designed for aux dominance, either the FAUX

or RAUX power sources will deliver power to the PD only un-

der the following two conditions: (i) If auxiliary power is applied

before PoE power, it will prevent the PD’s detection by the

PSE and will supply power indefinitely. This occurs because

the PoE input bridge rectifiers will be reverse biased, so no

detection signature will be observed. Under this condition,

when the auxiliary supply is removed, power will not be main-

tained because it will take some time for the PSE to perform

signature detection and classification before it will supply

power. (ii) If auxiliary power is applied after PoE power is al-

ready present but has a higher voltage than PoE, it may

assume power delivery responsibility. Under the second

case, if the supplied voltages are comparable, the load cur-

rent may be shared inversely proportional to the respective

output impedances of each supply. (The output impedance of

the PSE supply is increased by the cable series resistance).

If PoE power is applied first and has a higher voltage than the

non-dominant aux power source, it will continue powering the

PD even when the aux power source becomes available. In

this case, should PoE power be removed, the auxiliary source

will assume power delivery and supply the DC-DC loads with-

out interruption.

If either FAUX or RAUX power is supplied prior to PoE power,

it will prevent the recognition of the PD by the PSE. Conse-

quently, continuity of power delivery cannot be guaranteed

because the PoE supply will not be present when auxiliary

power is removed.

FAUX Option

With the FAUX option, the LM5072 hot swap MOSFET pro-

vides inrush and DC current limit protection for the auxiliary

power source. To select the FAUX configuration for an auxil-

iary voltage lower than nominal PoE voltages, the ICL_FAUX

pin must be forced above its high threshold to override the

VIN UVLO function. Note that when the ICL_FAUX pin is

pulled high to override VIN UVLO, it also overrides the inrush

current limit programmed by RICL, if present. In this case, the

inrush current will revert back to the default 150 mA limit.

Pulling up the ICL_FAUX pin will increase the default DC cur-

rent limit to 540 mA. This increase in DC current limit is

necessary because higher current is required to support the

PD output power at the lower input potentials observed with

auxiliary sources. In cases where the auxiliary supply voltage

is comparable to the PoE voltage, there is no need to pull-up

the ICL_FAUX pin to override VIN UVLO, and the default DC

current limit remains at 440 mA. However, if the DC current

limit is externally programmed with RDCCL, the condition of the

ICL_FAUX pin will not affect the programmed DC current limit.

In other words, programmed DC current limit can be consid-

ered a “hard limit” that will not vary in any configuration.

RAUX Option

The RAUX option is desirable when the auxiliary supply volt-

age is significantly lower than the PoE voltage or when aux

dominance is desired. The inrush and DC current limits of the

LM5072 do not protect or limit the RAUX power source, and

an additional resistor in the RAUX input path will be needed

to provide transient protection.

To select the RAUX option without aux dominance, simply pull

up the RAUX pin to the auxiliary power supply voltage through

a high value resistor. Depending on the auxiliary supply volt-

age, the resistor value should be selected such that the

current flowing into the RAUX pin is approximately 100 µA

when the pin is mid-way between the lower and upper RAUX

thresholds (approximately 4V). For example, with an 18V

non-dominant rear auxiliary supply, the pull up resistor should

be:

If the PSE load capacity is limited and insufficient, aux domi-

nance will be a desired feature to off load PoE power for other

PDs that do not have auxiliary power available. Aux domi-

nance is achieved by pulling the RAUX pin up to the auxiliary

supply voltage through a lower value (~5 kΩ) resistor that de-

livers at least 250 µA into the RAUX pin. When this higher

RAUX current level is detected, the LM5072 shuts down the

PD interface. In aux dominant mode, the auxiliary power

source will supply the PD system as soon as it is applied. PD

operation will not be interrupted when the aux power source

is connected. The PoE source may or may not actually be

removed by the PSE, although the DC current from the net-

work cable is effectively reduced to zero (< 150 µA). IEEE

802.3af requires the AC input impedance to be greater than

2 MΩ to ensure PoE power removal. This condition is not sat-

isfied when the auxiliary power source is applied. The PSE

17 www.national.com

LM5072

may remove power from a port based on the reduction in DC

current. This is commonly known as DC Maintain Power Sig-

nature (DC MPS), a common feature in many PSE systems.

The high voltage startup regulator of the PWM controller does

not have low dropout capability and will not be able to provide

VCC when the potential from VIN to RTN is less than 14.5V

(no external VCC load). In this case, the auxiliary voltage

should supply VCC directly via diode OR-ing to ensure suc-

cessful startup.

When using the RAUX configuration, the positive potential

connection of the 0.1 µF signature capacitor should be moved

from VIN to RTN/ARTN as shown in Figure 11. This provides

a high frequency, low impedance path for the IC's substrate

during rear auxiliary operation. Placing the capacitor here will

not affect signature mode.

It should be noted that rear auxiliary non-dominance does not

imply PoE dominance. PoE dominance is difficult to achieve

in any PoE system if continuity of power is desired. When the

PoE voltage appears, the PSE and PD interface must con-

tinue delivering load current in addition to charging the input

capacitor bank from the auxiliary voltage to the PoE voltage.

The situation is further complicated by the fact that for a given

delivered power level, the load current is much higher at the

lower input voltages typically used in auxiliary supplies. As is

the case during any inrush sequence, very high power is dis-

sipated in the hot swap MOSFET. Consequently, attempting

to achieve inrush completion while delivering load current is

highly ill advised. Lastly, current delivered to the system may

be limited by the PSE, the PD, or both.

A Note About FAUX and RAUX Pin

False Input State Detection

The ICL_FAUX and RAUX pins are used to sense the pres-

ence of auxiliary power sources. The input voltage of each pin

must remain low when the auxiliary power sources are ab-

sent. However, the Or-ing diodes feeding the auxiliary power

are not ideal and leak reverse current that can flow from the

PoE input to both the ICL_FAUX and RAUX pins. When PoE

power is applied, these leakage currents may elevate the po-

tentials of the ICL_FAUX and RAUX pins to false logic states.

One of two failure modes may be observed when the power

diode feeding the front auxiliary input leaks excessively. First,

the current may corrupt the inrush current limit programming,

if that feature has been implemented. Second, the leakage

current may elevate the voltage on the pin to the ICL_FAUX

input threshold, which will force UVLO release. This would

certainly interrupt any attempt by the LM5072 PD interface to

perform the signature or classification functions.

When the power diode that feeds the rear auxiliary input

leaks, the false signal could imply a rear auxiliary supply is

present. In this case, the internal hot swap MOSFET will be

turned off. This would of course block PoE power flow and

cause the circuit to prevent startup.

This leakage problem at the control input pins can be easily

solved. As shown in Figure 12, an additional pull-down resis-

tor (Rpd) across each auxiliary power control input provides

a path for the diode leakage current so that it will not create

false states on the ICL_FAUX or RAUX pins.

20184629

FIGURE 12. Bypassing Resistor – Prevents False

ICL_FAUX and RAUX Pin Signaling

High Voltage Startup Regulator

The LM5072 contains a startup bias regulator that allows the

VIN pin to be connected directly to PoE network voltages as

high as 100V. The regulator output is connected to the VCC

pin, providing an initial DC bias voltage of 7.7V nominal to

start the PWM controller. The regulator is internally current

limited to no less than 15 mA to prevent excessive power dis-

sipation. For VCC voltage stability and noise immunity, a ca-

pacitor ranging between 0.1 µF to 10 µF is required between

the VCC and ARTN pins. Though the current capability of the

regulator exceeds the requirements of the IC, no external DC

load drawing more than 3 mA should be applied to the output.

A small amount of current for a “Powered from PoE” indicator

LED (see Power Good section) is acceptable. After the DC-

DC converter reaches steady state operation, the VCC voltage

is typically elevated by an auxiliary winding of the power

transformer. The sustained VCC voltage should be greater

than 8.1V to guarantee the current supplied by the startup

regulator is reduced to zero. Increasing the VCC pin voltage

above the regulation level of the startup regulator automati-

cally disables the regulator, thus reducing the power dissipa-

tion inside the LM5072. The power savings can be significant

as many high voltage MOSFETs require a relatively large

amount of gate charge and the gate drive current adds directly

to the VCC current draw.

A VCC under-voltage lock-out circuit monitors the VCC voltage

to prevent the PWM controller from operating as the VCC volt-

age rises during startup or falls during shutdown. The PWM

controller is enabled when the VCC voltage rising edge ex-

ceeds 7.6V and disabled when the VCC voltage falling edge

drops below 6.25V.

Error Amplifier

The LM5072 contains a wide-bandwidth, high-gain error am-

plifier to regulate the output voltage in non-isolated applica-

tions. The amplifier’s non-inverting input is set to a fixed

reference voltage of 1.25V, while the inverting input is con-

nected to the FB pin. The open-drain output of the amplifier

is connected to the COMP pin, which is pulled up internally

through a 5 kΩ resistor to an internal 5V bias voltage. Feed-

back loop compensation can be easily implemented by plac-

ing the compensation network, represented by “Zcomp”,

between the FB and COMP pins as shown in Figure 13.

www.national.com 18

LM5072

20184630

FIGURE 13. Internal Error Amplifier – Used for Non-

isolated Output Applications

For isolated applications, the error amplifier function is locat-

ed on the isolated secondary side. The LM5072’s error am-

plifier can be disabled by connecting the FB pin to the ARTN

pin. As shown in Figure 14, an opto-coupler is normally used

to send the feedback signal across the isolation boundary to

the COMP pin. The internal pull-up resistor on the COMP pin

now serves as the pull-up bias for the opto-coupler transistor.

20184631

FIGURE 14. The Internal Error Amplifier – Bypassed in

Isolated Output Applications

Current Sense and Limit

The LM5072 CS pin senses the transformer primary current

signal for current mode control and current limiting of the sup-

ply. As shown in Figure 15, the current sense function can be

fulfilled by a simple sense resistor RSENSE inserted between

the RTN and the source of the primary MOSFET switch.

The RSENSE resistor should be non-inductive, and a low pass

filter should be used to reject the switching noise on the

sensed signal. A simple RC filter using 100Ω and 1 nF is typ-

ically sufficient. The filter capacitor must be located close to

the CS and ARTN pins. In order to prevent noise propagation

and to improve the noise immunity of the current sense, it is

very important to minimize the return path of the current sense

signal. This is accomplished with direct connection to the

ARTN pin and a single point connection to the RTN pin on the

PC Board layout.

20184632

FIGURE 15. Current Sense Schemes

20184633

FIGURE 16. Typical Current Sense Waveform Having a

Leading Edge Spike

The current sense signal is also used for cycle-by-cycle over-

current protection. When the CS pin signal exceeds 0.5V, the

PWM pulse of that cycle will be immediately terminated. The

desired cycle-by-cycle over-current protection level is

achieved by selecting the proper value of current sense re-

sistor that produces 0.5V at the CS pin. For the LM5072-80,

the slope compensation reduces the current limit threshold by

about 20% maximum at the 80% maximum duty cycle.

The typical current sense waveform as shown in Figure 16

has a spike at the leading edge. This spike is mainly caused

by the large gate drive current that flows through the current

sense resistor at turn-on (up to 0.8A). The reverse recovery

of the rectifier diode on the secondary side and the cross

conduction of the primary MOSFET and sync MOSFET (if

used) may also contribute to this leading edge spike. With a

relatively small external RC filter, this spike can still cause a

false over-current condition that terminates the PWM output

pulse. To avoid this problem, an internal blanking circuit is

provided within the LM5072 as shown in Figure 15. An internal

MOSFET is turned on to short the CS pin to ARTN at the end

of each cycle. This MOSFET switch remains on for an addi-

tional 65ns after the beginning of the next PWM cycle, thus

blanking out the leading edge spike on the current sense sig-

nal.

Soft-Start

The LM5072 incorporates a soft-start feature which forces the

PWM duty cycle to grow progressively during startup such

that the output voltage increases gradually to the steady state

level. The soft-start process reduces or prevents both the

surge of inrush current and the associated overshoot of the

output voltage during startup. The LM5072 achieves soft-start

using an internal 10 µA current source to charge an external

capacitor connected to the SS pin. The capacitor voltage lim-

its the voltage at the COMP pin which directly controls the

PWM duty cycle. The rate of the soft-start ramp can be ad-

justed by varying the value of the external capacitor. Note that

the slope of the supply’s output voltage is influenced by the

load condition and the total output capacitance of the supply,

as well as the soft-start programming. The supply should be

19 www.national.com

LM5072

started slowly enough such that the input current is limited

below the hot swap MOSFET DC current limit.

Gate Driver and Maximum Duty

Cycle Limit

The LM5072’s gate drive (OUT) pin can source and sink a

peak current of 800 mA directly to the gate of the DC-DC

converter’s power MOSFET switch. To serve a variety of ap-

plications, the LM5072 is available with two options for max-

imum PWM duty cycle. The LM5072-80 operates at duty

cycles up to 80% while the LM5072-50 limits the PWM duty

cycle to 50%.

Oscillator, Shutdown and Sync

Capability

The LM5072 requires a single external resistor connected

between the RT and ARTN pins to set the oscillator frequency

(FOSC). The RT timing resistor should be located very close to

the IC and connected directly to the RT and ARTN pins. The

following equation describes the relationship between FOSC

and the RT resistor value:

The LM5072 can also be synchronized to an external clock

signal with a frequency higher than the programmed oscillator

frequency determined by the RT resistor. The clock signal

should be coupled into the RT pin through a 100 pF capacitor,

as shown in Figure 17. Successful synchronization requires

the peak voltage of the sync pulse signal to be greater than

3.7V at the RT pin, and pulse width between 15 and 150 ns

(set by external components). The RT resistor is always re-

quired, whether the oscillator is operated in “free-running”

mode or with external synchronization.

20184635

FIGURE 17. Oscillator Synchronization Implementation

Special attention should be paid to the relationship between

the oscillator frequency and the PWM switching frequency.

For the LM5072-50 version, the programmed oscillator fre-

quency is internally divided by two in order to facilitate the

50% duty cycle limit. The PWM output switching frequency is

therefore one half of the programmed oscillator frequency.

The frequency divider is not used in the LM5072-80 and

therefore the PWM output frequency is the same as the os-

cillator frequency. These relationships also apply to external

synchronization frequency versus PWM output frequency.

PWM Comparator / Slope

Compensation

The PWM comparator produces the PWM duty cycle by com-

paring the current sense ramp signal with an error voltage

derived from the error amplifier output. The error amplifier

output voltage at the COMP pin is offset by 1.4V and then

further attenuated by a 3:1 resistor divider before it is pre-

sented to the PWM comparator input.

The PWM duty cycle increases with the voltage at the COMP

pin. The controller output duty cycle reduces to zero when the

COMP pin voltage drops below approximately 1.4V.

For duty cycles greater than 50%, current mode control loops

are subject to sub-harmonic oscillation. This instability can be

eliminated by adding an additional fixed slope voltage ramp

signal to the current sense signal. This technique is commonly

known as “slope compensation”. For the LM5072-80 version

with its maximum duty cycle of 80%, slope compensation is

integrated by injecting a 45 μA current ramp from the oscillator

into the current sense signal path (see Figure 2). The 45 μA

peak ramping current flows through an internal 2 kΩ resistor

to produce a fixed voltage ramp at the PWM comparator input.

Additional slope compensation may be added by increasing

the source impedance of the current sense signal with an ex-

ternal resistor between the CS pin and the source of the

current sense signal. The feature is disabled for the

LM5072-50 version because the duty cycle is limited to 50%

and slope compensation is not required.

Thermal Protection

The LM5072 includes internal thermal shutdown circuitry to

protect the IC in the event the maximum junction temperature

is exceeded. This circuit prevents catastrophic overheating

due to accidental overload of the hot swap MOSFET or other

circuitry. Typically, thermal shutdown is activated at 165°C,

causing the hot swap MOSFET and classification regulator to

be disabled. The PWM controller is disabled after the PGOOD

timer has expired. Thermal limit is not enabled unless the

module is being powered through the front end and the hot

swap MOSFET is enhanced. VCC current limit provides an

adequate level of protection for this 15 mA regulator. The

thermal protection is non-latching, therefore after the temper-

ature drops by the 25°C nominal hysteresis, the hot swap

MOSFET is re-activated and a soft-start is initiated to restore

the LM5072 to normal operation. If the cause of overheating

has not been eliminated, the circuit will hiccup in and out of

the thermal shutdown mode.

PCB Layout Guidelines

Before processing the Printed Circuit Board (PCB) layout, the

engineer should make all necessary adjustments to the

schematic to suite the application. The reader may notice that

the LM5072 evaluation board is designed with dual outputs,

both FAUX and RAUX power options, and some re-configu-

ration flexibility features (refer to Figure 19). However, many

devices can be removed for a particular application. Recom-

mendations on simplifying Figure 19 to suit a given application

are as follows:

1. When selecting the FAUX power option only, delete C3,

D1, D2, J3, P3, P4, R1, R2, R13, and R29.

2. When selecting the RAUX power option only, delete R30,

D3, D7, J2, P1, P2 and R6.

3. When neither FAUX nor RAUX power options are

selected, delete all the parts mentioned in (1) and (2)

above.

4. When only a single output is required, delete C11 through

C14, C17, D8, J6, J7, L2, R10 and Z4. Modify T1 design

to delete the unwanted second output winding and

increase the copper used for the single output winding.

www.national.com 20

LM5072

This re-configuration should make use of the spare pins

of the transformer.

5. R24 should be deleted from the schematic completely,

being replaced by a short connection for an isolated

application, or by an open for a non-isolated application.

6. Jumpers P5 and P6 (Figure 20) should be deleted from

the schematic completely, being replaced by a short

connection for an isolated application, or by an open for

a non-isolated application.

7. When the output is non-isolated, delete C20, C22, C25,

R7, R11, R16, R17, R24, U2, and U3. Replace C28 with

a short connection, and replace P5 and P6 with short

connections.

8. One may also modify the number of input and output

capacitors to achieve a more optimized design.

Consider the following when starting the PCB design:

1. Try to use both sides of the PCB for part placement to

facilitate both layout and routing.

2. Place the power components in a pattern that minimizes

the lengths of the high current paths on the PCB.

3. Place the LM5072 and its critical peripheral parts closely.

Bypass capacitors and transient protection elements

should be near the LM5072.

4. Route the critical traces first, including both power and

signal traces. Make the length of the trace as short as

possible, and avoid excessive use of via holes.

5. Pay attention to grounding issues. Each reference

ground should be a copper plane or island. Use via holes

if necessary for direct connections of devices to their

appropriate return ground plane or island. Identify the

following ground returns:

Primary power return COM: C4, C5, C6, R14, R15,

R29, C3, P4, J3-pins 2 and 3, U1-pin 8, C28, and C29

are all returned to the COM ground plane.

Primary control signal return, a ground return island:

C19, T1-pin 2, C23, U2-pin 3, R24, C26, C21, and U1-

pin 16 are all returned to this island, and the island should

be single point connected to the COM ground plane.

Secondary power return IGND: T1-pins 6 and 7, C7

through C10, C12 through C17, C28, Z4, J5, and J7 are

all returned to the IGND ground plane.

Secondary control signal return, a ground return

island: R18, U3 and C20 are all returned to this island,

and the island should be single point connected to the

IGND ground plane.

Also consider the following during PCB layout and routing.

1. Place the following power components in each group as

close as possible:

C4, C5 (if used), the primary winding of T1, Q1, and

R14/R15. The high frequency switching current (pulse

current) flows through these parts in a loop. The physical

area enclosed by the loop should be as small as possible.

D5, C7 through C10, and the secondary winding of T1

for the main output. The high frequency switching current

for the main output rail flows through these parts in a loop.

The physical area enclosed by the loop should be as

small as possible.

D8, C12 and C13, and the secondary winding of T1 for

the second output, if used. The high frequency switching

current for the second output rail flows through these

parts in a loop. The physical area enclosed by the loop

should be as small as possible.

L3, C15, C16, J4 and J5 (if posts are used). L3 should

also be as close as possible to the capacitor bank

consisting of C7 through C10 in order to minimize the

conduction losses on the PCB. Ceramic capacitor C15

should be placed directly at the output port.

L2, C14, C17, Z4, J6 and J7 (if posts are used) for the

second output rail. L2 should also be as close as possible

to C12 and C13 in order to minimize the conduction

losses on the PCB. Ceramic capacitor C14 should be

directly placed at the output port

2. U1 (LM5072) should be placed close to Q1 in the

orientation such that the gate drive output pin (OUT, Pin

9) is close to Q1’s gate.

3. (iii) Z2 and C27 must be placed directly across the VIN

and VEE pins for best protection against input transients.

In a rear auxiliary application, C27 should be removed

and C29 should be installed very close to the RTN and

VEE pins.

4. C19 should be placed directly across the VCC and ARTN

pins.

5. C23 should be placed directly across the CS and ARTN

pins.

6. R21 should be placed directly across the RT and ARTN

pins.

7. C26 should be placed directly across the SS and ARTN

pins.

8. C21 should be placed directly across the nPGOOD and

ARTN pins.

9. R25 should be directly routed from the output port.

10. R9 should be directly routed from R14/R15.

11. D6 and Z1 should be placed to achieve the shortest

connection from C4 or C5 to the drain pad(s) of Q1 for

better snubbing.

12. C2 and R4 should be placed to achieve the shortest

connection across D5.

13. Q1, D5, D8, and U1 (LM5072) should be installed on

thermal pads having adequate thermal vias down

through all PCB Layers and an exposed thermal pad on

the other side of the PCB.

14. Avoid spiral trace pattern.

15. Avoid placing switching traces near any traces in the

regulator feedback loop.

16. Pay attention to trace width. Try to make the power traces

as wide as possible. Conversely, do not make signal

traces wider than needed.

After the first placement and routing is completed, make nec-

essary modifications to optimize the design.

21 www.national.com

LM5072

Application Example #1

Figure 18 shows an application example of a single isolated output solution for the PD. Both front auxiliary (FAUX) and rear auxiliary

(RAUX) power options are given, although only one option may be needed in practice. Note that for the RAUX option, D2 is only

installed when the supply voltage of the auxiliary power source would cause the VIN voltage to be below 14.5V.

20184636

FIGURE 18. PD with Isolated, Single Output Solution

Application Example #2

Figure 19 shows an example of an isolated, dual-output solution for the PD. The 3.3V output is tightly regulated while the 5V output

is cross-regulated. Both front auxiliary (FAUX) and rear auxiliary (RAUX) power options are given, although only one option may

be needed in practice. Note that for the RAUX option, D2 is only installed when the supply voltage of the auxiliary power source

is lower than 14.5V.

20184637

FIGURE 19. PD with Isolated, Dual Output Solution

www.national.com 22

LM5072

Application Example #3:

Figure 20 shows an application example of the non-isolated version of Figure 18 . This non-isolated version saves many parts

used in the isolated feedback example shown in Figure 18. Similar simplification also applies to the non-isolated version of Figure

19.

20184638

FIGURE 20. PD Solution with Non-Isolated Flyback Topology

Application Example #4

Figure 21 shows an application example of a PD solution using the buck topology. Q2, a dual PNP transistor, is employed in the

output voltage sensing to achieve temperature compensation for the regulated output.

20184639

FIGURE 21. PD Solution with Buck Topology

23 www.national.com

LM5072

Physical Dimensions inches (millimeters) unless otherwise noted

Package Number MXA16A

www.national.com 24

LM5072

Notes

25 www.national.com

LM5072

Notes

LM5072 Integrated 100V Power Over Ethernet PD Interface and PWM Controller with Aux Support

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