Figure 1. Typical Application – Not a Simplified Circuit (a) and
Output Characteristic Tolerance Envelopes (b).
Product Highlights
Cost Effective Linear/RCC Replacement
Lowest cost and component count, constant voltage (CV)
or constant voltage/constant current (CV/CC) solution
Extremely simple circuit configuration
Up to 75% lighter power supply reduces shipping cost
Primary based CV/CC solution eliminates 10 to 20 secondary
components for low system cost
Combined primary clamp, feedback, IC supply, and loop
compensation functions–minimizes external components
Fully integrated auto-restart for short circuit and open loop
fault protection–saves external component costs
42 kHz operation simplifies EMI filter design
Much Higher Performance Over Linear/RCC
Universal input range allows worldwide operation
Up to 70% reduction in power dissipation–reduces enclosure
size significantly
CV/CC output characteristic without secondary feedback
System level thermal and current limit protection
Meets all single point failure requirements with only one
additional clamp capacitor
Controlled current in CC region provides inherent soft-start
Optional opto feedback improves output voltage accuracy
EcoSmart - Extremely Energy Efficient
Consumes <300 mW at 265 VAC input with no load
Meets Blue Angel, Energy Star, and EC requirements
No current sense resistors–maximizes efficiency
Applications
Linear transformer replacement in all ≤3 W applications
Chargers for cell phones, cordless phones, PDAs, digital
cameras, MP3/portable audio devices, shavers, etc.
Home appliances, white goods and consumer electronics
Constant output current LED lighting applications
TV standby and other auxiliary supplies
Description
LinkSwitch is specifically designed to replace low power linear
transformer/RCC chargers and adapters at equal or lower system
cost with much higher performance and energy efficiency.
LNK500 is a lower cost version of the LNK501 with a wider
tolerance output CC characteristic. LinkSwitch introduces a
revolutionary patented topology for the design of low power
switching power supplies that rivals the simplicity and low
cost of linear adapters, and enables a much smaller, lighter, and
Table 1. Notes: 1. Output power for designs in an enclosed adapter
measured at 50 °C ambient. 2. See Figure 1 (b) for Min (CV only
designs) and Typ (CV/CC charger designs) power points identified
on output characteristic. 3. Uses higher reflected voltage transformer
designs for increased power capability see Key Application
Considerations section. 4. For lead-free package options, see Part
Ordering Information.
attractive package when compared with the traditional “brick.
With efficiency of up to 75% and <300 mW no-load consumption,
a LinkSwitch solution can save the end user enough energy
over a linear design to completely pay for the full power
supply cost in less than one year. LinkSwitch integrates a
700 V power MOSFET, PWM control, high voltage start-up, current
limit, and thermal shutdown circuitry, onto a monolithic IC.
LNK500
LinkSwitch® Family
Energy Efficient, CV or CV/CC Switcher for
Very Low Cost Adapters and Chargers
®
December 2004
PI-3415-021103
Wide Range
HV DC Input
DC
Output
(VO)
Min
(CV only)
Example Characteristic
Typ
(CV/CC)
(a)
(b)
For Circuit
Shown Above With Optional
Secondary Feedback**
*Estimated tolerance achievable in high volume production
including transformer and other component tolerances.
**See Optional Secondary Feedback section.
LinkSwitch
±10%
±25%*
±5%
±25%*
D S
C
IOIO
VOVO
OUTPUT POWER TABLE1
PRODUCT4230 VAC ±15% 85-265 VAC No-Load
Input
Power
Min2Typ2Min2Typ2
LNK500
P or G
3.2 W 4 W 2.4 W 3 W <300 mW
4.3 W 5.5 W 2.9 W 3.5 W <500 mW3
2
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Pin Functional Description
DRAIN (D) Pin:
Power MOSFET drain connection. Provides internal operating
current for start-up. Internal current limit sense point for drain
current.
CONTROL (C) Pin:
Error amplifier and feedback current input pin for duty cycle
and current limit control. Internal shunt regulator connection
to provide internal bias current during normal operation. It is
also used as the connection point for the supply bypass and
auto-restart/compensation capacitor.
SOURCE (S) Pin:
Output MOSFET source connection for high voltage power
return. Primary side control circuit common and reference
point.
Figure 3. Pin Configuration.
PI-3417-111802
SD
S
S
S
C
5
7
8
S
4
2
3
1
P Package (DIP-8B)
G Package (SMD-8B)
LNK500
Figure 2. Block Diagram.
PI-3416-032603
SHUTDOWN/
AUTO-RESTART
PWM
COMPARATOR
CLOCK
SAW
OSCILLATOR
INTERNAL
SUPPLY
5.6 V
4.7
V
SOURCE
S
R
Q
DMAX
-
+
CONTROL
-
+
5.6
V
IFB
ZC
VC
+
-
EDGE
0
1
HYSTERETIC
THERMAL
SHUTDOWN
LEADING
EDGE
BLANKING
CURRENT
LIMIT
ADJUST
LOW
FREQUENCY
OPERATION
SHUNT REGULATOR/
ERROR AMPLIFIER
+
-
DRAIN
IDCS
CURRENT LIMIT
COMPARATOR
RE
÷ 8
3
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C
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LinkSwitch Functional Description
The duty cycle, current limit and operating frequency
relationships with CONTROL pin current are shown in
Figure 4. Figure 5 shows a typical power supply outline
schematic which is used below to describe the LinkSwitch
operation.
Power Up
During power up, as VIN is first applied (Figure 5), the CONTROL
pin capacitor C1 is charged through a switched high voltage
current source connected internally between the DRAIN and
CONTROL pins (see Figure 2). When the CONTROL pin
voltage reaches approximately 5.6 V relative to the SOURCE
pin, the high voltage current source is turned off, the internal
control circuitry is activated and the high voltage internal
MOSFET starts to switch. At this point, the charge stored on
C1 is used to supply the internal consumption of the chip.
Constant Current (CC) Operation
As the output voltage, and therefore the reflected voltage
across the primary transformer winding ramp up, the feedback
CONTROL current IC increases. As shown in Figure 4, the
internal current limit increases with IC and reaches ILIM when IC
is equal to IDCT
. The internal current limit vs. IC characteristic
is designed to provide an approximately constant power supply
output current as the power supply output voltage rises.
Constant Voltage (CV) Operation
When IC exceeds IDCS, typically 2 mA (Figure 4), the maximum
duty cycle is reduced. At a value of IC that depends on power
supply input voltage, the duty cycle control limits LinkSwitch
peak current below the internal current limit value. At this point
the power supply transitions from CC to CV operation. With
minimum input voltage in a typical universal input design, this
transition occurs at approximately 30% duty cycle. Resistor R1
(Figure 5) is therefore initially selected to conduct a value of IC
approximately equal to IDCT when VOUT is at the desired value
at the minimum power supply input voltage. The final choice
of R1 is made when the rest of the circuit design is complete.
When the duty cycle drops below approximately 4%, the
frequency is reduced, which reduces energy consumption under
light load conditions.
Auto-Restart Operation
When a fault condition, such as an output short circuit or open
loop, prevents flow of an external current into the CONTROL
pin, the capacitor C1 discharges towards 4.7 V. At 4.7 V,
auto-restart is activated, which turns the MOSFET off and puts
the control circuitry in a low current fault protection mode. In
auto-restart, LinkSwitch periodically restarts the power supply
so that normal power supply operation can be restored when
the fault is removed.
Figure 4. CONTROL Characteristics.
Figure 5. Power Supply Outline Schematic.
VOUT
VIN
D2
C4
D
S
C
C2
R1
R2
C1
D1
Link
witc
PI-2715-112102
PI-2799-112102
Internal Current Limit
CONTROL Current IC
IDCS
CONTROL Current IC
ICD1
CONTROL Current IC
Duty Cycle
Frequency
ILIM
IDCT
77%
30%
3.8%
fOSC
fOSC(low)
Auto-restart
Auto-restart
Auto-restart
4
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The characteristics described above provide an approximate
CV/CC power supply output without the need for secondary-
side voltage or current feedback. The output voltage regulation
is influenced by how well the voltage across C2 tracks the
reflected output voltage. This tracking is influenced by the
value of the transformer leakage inductance which introduces
an error. Resistor R2 and capacitor C2 partially filter the
leakage inductance voltage spike reducing this error. This
circuitry, used with standard transformer construction
techniques provides much better output load regulation than a
linear transformer, making this an ideal power supply solution
in many low power applications. If tighter load regulation is
required, an optocoupler configuration can be used while still
employing the constant output current characteristics provided
by LinkSwitch.
Optional Secondary Feedback
Figure 6 shows a typical power supply outline schematic using
LinkSwitch with optocoupler feedback to improve output
voltage regulation. On the primary side, the schematic differs
from Figure 5 by the addition of R3, C3 and optocoupler U1.
R3 forms a potential divider with R1 to limit the U1 collector
emitter voltage.
On the secondary side, the addition of voltage sense circuit
components R4, VR1 and U1 LED provide the voltage feedback
signal. In the example shown, a simple Zener (VR1) reference
is used though a precision TL431 reference is typically needed
to provide ±5% output voltage tolerancing and cable drop
compensation, if required. Resistor R4 provides biasing for VR1.
The regulated output voltage is equal to the sum of the VR1
Zener voltage plus the forward voltage drop of the U1 LED.
Resistor R5 is an optional low value resistor to limit U1 LED
peak current due to output ripple. Manufacturerʼs specifications
for U1 current and VR1 slope resistance should be consulted
to determine whether R5 is required.
U1 is arranged with collector connected to primary ground and
emitter to the anode of D1. This connection keeps the opto in
an electrically “quiet” position in the circuit. If the opto was
Figure 6. Power Supply Outline Schematic with Optocoupler Feedback.
Figure 7. Influence of the Optocoupler on the Power Supply Output Characteristic.
LNK500
LinkSwitch
85-265
VAC
VOUT
RTN
SD
CC1
C3
C2
U1
R4
R5
VR1
R1
R2
D1
R3
PI-3418-071304
U1
Output Voltage
Tolerance envelope
without optocoupler
Inherent
CC to CV
transition
point
Load variation
during battery
charging
Voltage
feedback
threshold
Characteristic with
optocoupler
Typical inherent
characteristic without
optocoupler
PI-2788-092101
Output Current
5
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Figure 8. Output Characteristic with Optocoupler Regulation (Reduced Voltage Feedback Threshold).
Output Voltage
Output Current
VO(MAX)
Tolerance envelope
without optocoupler
Characteristic with
optocoupler
Power supply peak
output power curve
Typical inherent
characteristic without
optocoupler
PI-2790-112102
Inherent
CC to CV
transition
point
Load variation
during battery
charging Characteristic observed with
load variation often applied during
laboratory bench testing
Voltage
feedback
threshold
instead placed on the cathode side of D1, it would become a
switching node, generating additional common mode EMI
currents through its internal parasitic capacitance.
The feedback configuration in Figure 6 is simply a resistive
divider made up of R1 and R3 with D1, R2, C1 and C2 rectifying,
filtering and smoothing the primary winding voltage signal. The
optocoupler therefore effectively adjusts the resistor divider ratio
to control the DC voltage across R1 and therefore, the feedback
current received by the LinkSwitch CONTROL pin.
When the power supply operates in the constant current (CC)
region, for example when charging a battery, the output voltage
is below the voltage feedback threshold defined by U1 and
VR1 and the optocoupler is fully off. In this region, the circuit
behaves exactly as previously described with reference to
Figure 5 where the reflected voltage increases with increasing
output voltage and the LinkSwitch internal current limit is
adjusted to provide an approximate CC output characteristic.
Note that for similar output characteristics in the CC region,
the value of R1 in Figure 5 will be equal to the value of R1+R3
in Figure 6.
When the output reaches the voltage feedback threshold set by
U1 and VR1, the optocoupler turns on. Any further increase
in the power supply output voltage results in the U1 transistor
current increasing, which increases the percentage of the
reflected voltage appearing across R1. The resulting increase
in the LinkSwitch CONTROL current reduces the duty cycle
according to Figure 4 and therefore, maintains the output
voltage regulation.
Normally, R1 and R3 are chosen to be equal in value. However,
increasing R3 (while reducing R1 to keep R1 + R3 constant)
increases loop gain in the CV region, improving load regulation.
The extent to which R3 can be increased is limited by opto
transistor voltage and dissipation ratings and should be fully
tested before finalizing a design. The values of C2 and C3 are
less important other than to make sure they are large enough
to have very little influence on the impedance of the voltage
division circuit set up by R1, R3 and U1 at the switching
frequency. Normally, the values of C2 and C3 in Figure 6 are
chosen equal to the value of C2 in Figure 5, though the voltage
rating may be reduced depending on the relative values of R1
and R2 discussed above. See Applications section for typical
values of components.
Figure 7 shows the influence of optocoupler feedback on the
output characteristic. The envelope defined by the dashed lines
represent the worst case power supply DC output voltage and
current tolerances (unit-to-unit and over the input voltage
range) if an optocoupler is not used. A typical example of an
inherent (without optocoupler) output characteristic is shown
dotted. This is the characteristic that would result if U1, R4 and
VR1 were removed. The optocoupler feedback results in the
characteristic shown by the solid line. The load variation arrow in
Figure 7 represents the locus of the output characteristic normally
seen during a battery charging cycle. The two characteristics
are identical as the output voltage rises but then separate as
shown when the voltage feedback threshold is reached. This
is the characteristic seen if the voltage feedback threshold is
above the output voltage at the inherent CC to CV transition
point also indicated in Figure 7.
Figure 8 shows a case where the voltage feedback threshold is
set below the voltage at the inherent CC to CV transition point.
In this case, as the output voltage rises, the secondary feedback
circuit takes control before the inherent CC to CV transition
occurs. In an actual battery charging application, this simply
limits the output voltage to a lower value.
6
LNK500
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However, in laboratory bench tests, it is often more convenient
to test the power supply output characteristic starting from a
low output current and gradually increasing the load. In this
case, the optocoupler feedback regulates the output voltage until
the peak output power curve is reached as shown in Figure 8.
Under these conditions, the output current will continue to rise
until the peak power point is reached and the optocoupler turns
off. Once the optocoupler is off, the CONTROL pin feedback
current is determined only by R1 and R3 and the output current
therefore folds back to the inherent CC characteristic as shown.
Since this type of load transition does not normally occur in a
battery charger, the output current never overshoots the inherent
constant current value in the actual application.
In some applications it may be necessary to avoid any output
current overshoot, independent of the direction of load variation.
To achieve this goal, the minimum voltage feedback threshold
should be set at VO(MAX). This will ensure that the voltage at the
CC to CV transition point of the inherent characteristic will
always occur below the voltage feedback threshold. However, the
output voltage tolerance is then increased, since the inherent CV
characteristic tolerance below VO(MAX) is added to the tolerance
of the optocoupler feedback circuit.
Applications Example
The circuit shown in Figure 9 shows a typical implementation
of an approximate constant voltage / constant current (CV/CC)
charger using LinkSwitch. This design delivers 2.75 W with
a nominal peak power point voltage of 5.5 V and a current of
500 mA. Efficiency is greater than 70% over an input range
of 85 VAC to 265 VAC.
The bridge rectifier, BR1, rectifies the AC input. Resistor, RF1
is a fusible type providing protection from primary-side short
circuits. The rectified AC is smoothed by C1 and C2 with
inductor L1 forming a pi-filter in conjunction with C1 and C2
to filter conducted EMI. The switching frequency of 42 kHz
allows such a simple EMI filter to be used without the need for
a Y capacitor while still meeting international EMI standards.
When power is applied, high voltage DC appears at the DRAIN
pin of LinkSwitch (U1). The CONTROL pin capacitor C3 is then
charged through a switched high voltage current source connected
internally between the DRAIN and CONTROL pins. When
the CONTROL pin reaches approximately 5.6 V relative to the
SOURCE pin, the internal current source is turned off. The internal
control circuitry is activated and the high voltage MOSFET starts
to switch, using the energy in C3 to power the IC.
When the MOSFET is on, the high voltage DC bus is connected
to one end of the transformer primary, the other end being
connected to primary return. As the current ramps in the
primary of flyback transformer T1, energy is stored. This
energy is delivered to the output when the MOSFET turns off
each switching cycle.
The secondary of the transformer is rectified and filtered by D6
and C5 to provide the DC output to the load.
LinkSwitch dramatically simplifies the secondary side by
controlling both the constant voltage and constant current regions
entirely from the primary side. This is achieved by monitoring
the primary-side VOR (voltage output reflected).
Diode D5 and capacitor C4 form the primary clamp network.
This both limits the peak drain voltage due to leakage inductance
and provides a voltage across C4, which is equal to the VOR plus
an error due to the parasitic leakage inductance. Resistor R2
filters the leakage inductance spike and reduces the error in the
value of the VOR. Resistor R1 converts this voltage into a current
that is fed into the CONTROL pin to regulate the output.
During CV operation the output is regulated through control of
the duty cycle. As the current into the CONTROL pin exceeds
approximately 2 mA, the duty cycle begins to reduce, reaching
30% at a CONTROL pin current of 2.3 mA.
Under light or no-load conditions, when the duty cycle reaches
approximately 4%, the switching frequency is reduced to lower
energy consumption.
If the output load is increased beyond the peak power point
(defined by 0.5·LP·ILIM
2·f), the output voltage and VOR falls.
The reduced CONTROL pin current will lower the internal
LinkSwitch current limit (current limit control) providing an
approximately constant current output characteristic. If the
load is increased and the CONTROL pin current falls below
approximately 1 mA, the CONTROL pin capacitor C3 will
discharge and the supply enters auto-restart.
Current limit control removes the need for any secondary side
current sensing components (sense resistor, transistor, opto
coupler and associated components). Removing the secondary
sense circuit dramatically improves efficiency, giving the
associated benefit of reduced enclosure size.
Key Application Considerations
Design Output Power
Table 1 (front page) provides guidance for the maximum
continuous output power from a given device under the
conditions specified.
The output of chargers (CV/CC) are normally specified at
the typical output peak power point. Conversely, non-charger
applications (CV only, which applies to many converters such
as adapters, standby/auxiliary supplies and other embedded
AC-DC converters) where CC operation is not required, are
normally specified at the minimum output power they will
supply under worst case conditions.
7
LNK500
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Figure 9. 2.75 W Constant Voltage/Constant Current (CV/CC) Charger using LinkSwitch.
Figure 10. Measured Output Characteristic of the Circuit in Figure 9.
1000 200 300 400 500 600 700
Output Current (mA)
Output Voltage (V)
0
1
2
3
4
5
6
7
8
9
10
PI-3420-111802
VIN = 85 VAC
VIN = 115 VAC
VIN = 185 VAC
VIN = 265 VAC
C1
4.7 µF
400 V
C2
4.7 µF
400 V
RF1
10 1 W
Fusible
L1
1 mH
R1
20.5 k
1%
R2
100
D5
1N4937
C4
0.1 µF
100 V
116 T
#34 AWG
EE13
LP = 2.55 mH
15 T
#30 AWG
TIW
3
4
15
T1
6
D6
11DQ06
C5
470 µF
10 V
85-265
VAC
U1
LNK500
LinkSwitch
5.5 V,
500 mA
RTN
BR1
1 A, 600 V
PI-3419-071304
PERFORMANCE SUMMARY
Output Power: 2.75 W
Efficiency: 72%
No Load
Consumption: 260 mW, 230 VAC
200 mW, 115 VAC
C3
0.22 µF
50 V
DS
C
8
LNK500
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To aid the designer, the power table reflects these differences. For
CV/CC designs the typical power column and for CV designs
the minimum power column should be used, respectively.
Additionally, figures are based on the following conditions:
1. The minimum DC input bus voltage is 90 V or higher. This
corresponds to a filter capacitor of 3
µF/W for universal input
and 1 µF/W for 230 VAC or 115 VAC input with doubler
input stage.
2. Design is a discontinuous mode flyback converter.
Continuous mode designs can result in loop instability and
are therefore not recommended. For typical output power
figures, nominal values for primary inductance and I2f are
assumed. For minimum output power figures, primary
inductance minus 10% and the minimum I2f value are
assumed. For no-load consumption <300 mW, a VOR in the
range 40 V to 60 V is assumed. For no-load consumption
<500 mW and higher output power capability, a VOR in the
range 60 V to 100 V is assumed.
3. A secondary output of 5 V with a Schottky rectifier diode.
4. Assumed efficiency of 70%.
5. The part is board mounted with SOURCE pins soldered to
sufficient area of copper to keep the die temperature at or
below 100 °C.
6. An output cable with a total resistance of 0.2 .
In addition to the thermal environment (sealed enclosure,
ventilated, open frame, etc), the maximum power capability
of LinkSwitch in a given application depends on transformer
core size, efficiency, primary inductance tolerance, minimum
specified input voltage, input storage capacitance, output voltage,
output diode forward drop, etc., and can be different from the
values shown in Table 1.
Transformer Design
To provide an approximately CV/CC output, the transformer
should be designed to be discontinuous; all the energy stored
in the transformer is transferred to the secondary during the
MOSFET off time. Energy transfer in discontinuous mode is
independent of line voltage.
The peak power point prior to entering constant current operation
is defined by the maximum power transferred by the transformer.
The power transferred is given by the expression P = 0.5·LP·I2·f,
where LP is the primary inductance, I2 is the primary peak current
squared and f is the switching frequency.
To simplify analysis, the data sheet parameter table specifies an
I2f coefficient. This is the product of current limit squared and
switching frequency normalized to the feedback parameter IDCT
.
This provides a single term that specifies the variation of the
peak power point in the power supply due to LinkSwitch.
As primary inductance tolerance is part of the expression
that determines the peak output power point (start of the CC
characteristic) this parameter should be well controlled. For
an estimated overall constant current tolerance of ±25% the
primary inductance tolerance should be ±10% or better. This
is achievable using standard low cost, center leg gapping
techniques where the gap size is typically 0.08 mm or larger.
Smaller gap sizes are possible but require non-standard, tighter
ferrite AL tolerances.
Other gapping techniques such as film gapping allow tighter
tolerances (±7% or better) with associated improvements in
the tolerance of the peak power point. Please consult your
transformer vendor for guidance.
Core gaps should be uniform. Uneven core gapping, especially
with small gap sizes, may cause variation in the primary
inductance with flux density (partial saturation) and make the
constant current region non-linear. To verify uniform gapping
it is recommended that the primary current wave-shape be
examined while feeding the supply from a DC source. The
gradient is defined as di/dt = V/L and should remain constant
throughout the MOSFET on time. Any change in gradient of
the current ramp is an indication of uneven gapping.
Measurements made using an LCR bridge should not be solely
relied upon; typically these instruments only measure at currents
of a few milliamps. This is insufficient to generate high enough
flux densities in the core to show uneven gapping.
For a typical EE13 core using center leg gapping, a 0.08 mm
gap (ALG of 190 nH/t2) allows a primary inductance tolerance of
±10% to be maintained in standard high volume production.
This allows the EE13 to be used in designs up to 2.75 W with
less than 300 mW no-load consumption. If film gapping is used
then this increases to 3 W. Moving to a larger core, EE16 for
example, allows a 3 W output with center leg gapping.
The transformer turns ratio should be selected to give a VOR
(output voltage reflected through secondary to primary turns
ratio) of 40 V to 60 V. In designs not required to meet 300 mW
no-load consumption targets, the transformer can be designed
with higher VOR as long as discontinuous mode operation is
maintained. This increases the output power capability. For
example, a 230 VAC input design using an EE19 transformer
core with VOR >70 V, is capable of delivering up to 5.5 W typical
output power. Note: the linearity of the CC region of the power
supply output characteristic is influenced by VOR. If this is an
important aspect of the application, the output characteristic
should be checked before finalizing the design.
Output Characteristic Variation
Both the device tolerance and external circuit govern the overall
tolerance of the LinkSwitch output characteristic. Estimated
peak power point tolerances for a 3 W design are ±10% for
9
LNK500
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voltage and ±25% for current limit for overall variation in high
volume manufacturing. This includes device and transformer
tolerances and line variation. Lower power designs may have
poorer constant current linearity.
As the output load reduces from the peak power point, the
output voltage will tend to rise due to tracking errors compared
to the load terminals. Sources of these errors include the
output cable drop, output diode forward voltage and leakage
inductance, which is the dominant cause. As the load reduces,
the primary operating peak current reduces, together with the
leakage inductance energy, which reduces the peak charging
of the clamp capacitor. With a primary leakage inductance of
50 µH, the output voltage typically rises 30% over a 100% to
5% load change.
At very light or no-load, typically less than 2 mA of output current,
the output voltage rises due to leakage inductance peak charging
of the secondary. This voltage rise can be reduced with a small
preload with little change to no-load power consumption.
The output voltage load variation can be improved across the
whole load range by adding an optocoupler and secondary
reference (Figure 6). The secondary reference is designed to only
provide feedback above the normal peak power point voltage
to maintain the correct constant current characteristic.
Component Selection
The schematic shown in Figure 5 outlines the key components
needed for a LinkSwitch supply.
Clamp diode – D1
Diode D1 should be either a fast (trr <250 ns) or ultra-fast
type (trr <50 ns), with a voltage rating of 600 V or higher. Fast
recovery types are preferred, being typically lower cost. Slow
diodes are not recommended; they can allow excessive DRAIN
ringing and the LinkSwitch to be reverse biased.
Clamp Capacitor – C2
Capacitor C2 should be a 0.1 µF, 100 V capacitor. Low cost
metallized plastic film types are recommended. The tolerance
of this part has a very minor effect on the output characteristic
so any of the standard ±5%, ±10% or ±20% tolerances are
acceptable. Ceramic capacitors are not recommended. The
common dielectrics used such as Y5U or Z5U are not stable
with voltage or temperature and may cause output instability.
Ceramic capacitors with high stability dielectrics may be used
but are expensive compared to metallized film types.
CONTROL Pin Capacitor – C1
Capacitor C1 is used during start-up to power LinkSwitch and
sets the auto-restart frequency. For designs that have a battery
load, this component should have a value of 0.22 µF and for
resistive loads, a value of 1 µF. This ensures there is sufficient
time during start-up for the output voltage to reach regulation.
Any capacitor type is acceptable with a voltage rating of
10 V or above.
Feedback Resistor – R1
The value of R1 is selected to give a feedback current into the
CONTROL pin of approximately 2.3 mA at the peak output
power point of the supply. The actual value depends on the VOR
selected during design. Any 0.25 W resistor is suitable.
Output Diode – D2
Either PN fast, PN ultra-fast or Schottky diodes can be used,
depending on the efficiency target for the supply, Schottky
diodes giving higher efficiency then PN diodes. The diode
voltage rating should be sufficient to withstand the output
voltage plus the input voltage transformed through the turns
ratio (a typical VOR of 50 V requires a diode PIV of 50 V).
Slow recovery diodes are not recommended (1N400X types).
Output Capacitor – C4
Capacitor C4 should be selected such that its voltage and ripple
current specifications are not exceeded.
LinkSwitch Layout considerations
Primary Side Connections
Since the SOURCE pins in a LinkSwitch supply are switching
nodes, the copper area connected to SOURCE together with C1,
C2 and R1 (Figure 5) should be minimized, within the thermal
contraints of the design, to reduce EMI coupling.
The CONTROL pin capacitor C1 should be located as close as
possible to the SOURCE and CONTROL pins.
To minimize EMI coupling from the switching nodes on the
primary to both the secondary and AC input, the LinkSwitch
should be positioned away from the secondary of the transformer
and AC input.
Routing the primary return trace from the transformer primary
around LinkSwitch and associated components further reduces
coupling.
Y capacitor
If a Y capacitor is required, it should be connected close to the
transformer secondary output return pin(s) and the primary bulk
capacitor negative return. Such placement will maximize the
EMI benefit of the Y capacitor and avoid problems in common-
mode surge testing.
10
LNK500
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Figure 11. Recommended Circuit Board Layout for LinkSwitch using P Package.
Quick Design Checklist
As with any power supply design, all LinkSwitch designs
should be verified on the bench to make sure that component
specifications are not exceeded under worst case conditions.
Note: In a LinkSwitch circuit, the SOURCE is a switching
node. This should be taken into consideration during testing.
Oscilloscope measurements should be made with probe grounded
to DC voltages such as primary return or DC rail but not to
SOURCE. Power supply input voltage should always be supplied
using an isolation transformer. The following minimum set of
tests is strongly recommended:
1. Maximum drain voltage – Verify that VDS does not exceed
675 V at highest input voltage and peak output power.
2. Maximum drain current At maximum ambient temperature,
maximum input voltage and peak output power, verify drain
current waveforms at start-up for any signs of transformer
saturation and excessive leading edge current spikes.
LinkSwitch has a minimum leading edge blanking time of
200 ns to prevent premature termination of the on-cycle.
Verify that the leading edge current spike event is below
current limit at the end of the 200 ns blanking period.
3. Thermal check At peak output power, minimum input
voltage and maximum ambient temperature, verify that the
temperature specifications are not exceeded for LinkSwitch,
transformer, output diode and output capacitors. Enough
thermal margin should be allowed for part-to-part variation of
the RDS(ON) of LinkSwitch as specified in the data sheet. Under
low line, peak power, a maximum LinkSwitch SOURCE pin
temperature of 100 °C is recommended to allow for these
variations.
4. Centered output characteristic – Using a transformer with
nominal primary inductance and at an input voltage midway
between low and high line, verify that the peak power point
occurs at the desired nominal output current, with the correct
output voltage. If this does not occur then the design should
be refined to ensure the overall tolerance limits are met.
Design Tools
Up to date information on design tools can be found at the
Power Integrations website: www.powerint.com.
+
-
HV DC
Input
PI-2900-070202
Transformer
+
-
DC Out
D
S
S S SS
C
LinkSwitch
Y1-
Capacitor
Input Filter
Capacitor
Output
Capacitor
11
LNK500
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12/04
ABSOLUTE MAXIMUM RATINGS(1,4)
DRAIN Voltage .................................. ................ -0.3 V to 700 V
DRAIN Peak Current......................................400 mA
CONTROL Voltage ................................................ -0.3 V to 9 V
CONTROL Current (not to exceed 9 V)............100 mA
Storage Temperature .......................................... -65 °C to 150 °C
Operating Junction Temperature(3) ..................... -40 °C to 150 °C
Lead Temperature(4) ........................................................260 °C
Notes:
1. All voltages referenced to SOURCE, TA = 25 °C.
2. Normally limited by internal circuitry.
3. 1/16 in. from case for 5 seconds.
4. Maximum ratings specified may be applied, one at a time,
without causing permanent damage to the product.
Exposure to Absolute Maximum Rating conditions for
extended periods of time may affect product reliability.
THERMAL IMPEDANCE
Thermal Impedance: P or G Package:
(θJA) ........................... 70 °C/W(2); 55 °C/W(3)
(θJC)(1) ............................................... 11 °C/W
Notes:
1. Measured on pin 2 (SOURCE) close to plastic interface.
2. Soldered to 0.36 sq. in. (232 mm2), 2 oz. (610 g/m2) copper clad.
3. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 12
(Unless Otherwise Specified)
Min Typ Max Units
CONTROL FUNCTIONS
Switching
Frequency fOSC IC = IDCT
, TJ = 25 °C 34.5 42 49.5 kHz
Low Switching
Frequency fOSC(LOW)
Duty Cycle = DCLF
TJ = 25 °C 24 30 36 kHz
Duty Cycle at Low
Switching
Frequency
DCLF
Frequency Switching from fOSC to
fOSC(LOW), TJ = 25 °C 2.4 3.8 5.2 %
Low Frequency
Duty Cycle Range DC(RANGE) Frequency = fOSC(LOW), TJ = 25 °C 1.8 3.15 4.5 %
Maximum Duty
Cycle DCMAX IC = 1.5 mA 74 77 80 %
PWM Gain DCREG IC = IDCT, TJ = 25 °C -0.45 -0.35 -0.25 %/µA
CONTROL Pin
Current at 30%
Duty Cycle
IDCT
TJ = 25 °C
See Figure 4 2.21 2.30 2.39 mA
CONTROL Pin
Voltage VC(IDCT) IC = IDCT 5.5 5.75 6 V
Dynamic
Impedance ZCIC = IDCT, TJ = 25 °C 60 90 120
12
LNK500
C
12/04
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 12
(Unless Otherwise Specified)
Min Typ Max Units
SHUTDOWN/AUTO-RESTART
CONTROL Pin
Charging Current IC(CH) TJ = 25 °C VC = 0 V -4.5 -3.25 -2 mA
VC = 5.15 V -2.3 -1.3 -0.3
Control/Supply/
Discharge Current
ICD1 TJ = 25 °C Output MOSFET Enabled 0.95 1.06 1.14 mA
ICD2 TJ = 25 °C Output MOSFET Disabled 0.7 0.9 1.1
Auto-Restart
Threshold Voltage VC(AR) 5.6 V
Auto-Restart
Hysteresis Voltage VC(AR)hyst 0.9 V
Auto-Restart Duty
Cycle DC(AR)
Short Circuit Applied at
Power Supply Output 8 %
Auto-Restart
Frequency f(AR)
S2 Open
C1 = 0.22 µF (See Figure 12) 300 Hz
CIRCUIT PROTECTION
Self-Protection
Current Limit ILIM
TJ = 25 °C
di/dt = 90 mA/µs
See Note C
228 254 280 mA
I2f Coefficient I2f
TJ = 25 °C
di/dt = 90 mA/µs
See Notes C, D
2412 2710 3008 A2Hz
Current Limit at
Auto-Restart ILIM(AR) IC = ICD1, TJ = 25 °C 158 mA
Power Up Reset
Threshold Voltage VC(RESET) 1.5 2.75 4.0 V
Leading Edge
Blanking Time tLEB IC = IDCT, TJ = 25 °C 200 300 ns
Current Limit Delay tIL(D) TJ = 25 °C 100 ns
Thermal Shutdown
Temperature IC = IDCT 125 135 °C
Thermal Shutdown
Hysteresis 70 °C
13
LNK500
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12/04
NOTES:
A. For specifications with negative values, a negative temperature coefficient corresponds to an increase in magnitude with
increasing temperature, and a positive temperature coefficient corresponds to a decrease in magnitude with increasing
temperature.
B. Breakdown voltage may be checked against minimum BVDSS specification by ramping the DRAIN pin voltage up to but not
exceeding minimum BVDSS.
C. IC is increased gradually to obtain maximum current limit at di/dt of 90 mA/µs. Increasing IC further would terminate the cycle
through duty cycle control.
D. This parameter is normalized to IDCT to correlate to power supply output current (it is multiplied by IDCT(nominal)/IDCT).
E. It is possible to start up and operate LinkSwitch at DRAIN voltages well below 36 V. However, the CONTROL pin charging
current is reduced, which affects start-up time, auto-restart frequency, and auto-restart duty cycle. Refer to the characteristic
graph on CONTROL pin charge current (IC) vs. DRAIN voltage (Figure 13) for low voltage operation characteristics.
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 12
(Unless Otherwise Specified)
Min Typ Max Units
OUTPUT
ON-State
Resistance RDS(ON) ID = 25 mA TJ = 25 °C 28 32
TJ = 25 °C 42 48
OFF-State Drain
Leakage Current IDSS
VC = 6.2 V
VD = 560 V, TA = 125 °C50 µA
Breakdown Voltage BVDSS
See Note B
VC = 6.2 V, TA = 25 °C700 V
DRAIN Supply
Voltage See Note E 36 50 V
14
LNK500
C
12/04
Figure 12. LinkSwitch General Test Circuit.
Figure 13. IC vs. DRAIN Voltage.
Figure 14. Duty Cycle Measurement.
120
100
80
40
60
20
0
0.0 2.0 4.0 8.06.0 10.0 12.0 14.0
CONTROL Pin Voltage (V)
PI-2895-102303
CONTROL Pin Current (mA)
Figure 15. CONTROL Pin I-V Characteristic.
90
50
60
70
80
0
2.15 2.25 2.35 2.45 2.55 2.65
CONTROL Pin Current (mA)
Duty Cycle (%)
PI-2902-051904
20
10
40
30
Figure 16. Duty Cycle vs. CONTROL Pin Current.
2
1.2
1.6
0
0 20 40 60 80 100
DRAIN Voltage (V)
CONTROL Pin
Charging Current (mA)
PI-2901-071602
0.4
0.8
VC = 5.15 V
D
C
SS
S
S
S
LinkSwitch
PI-2894-031004
C1
0.22 µF
S2
40 V 40 V
750
10 k
S1
15
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12/04
Typical Performance Characteristics
Figure 17. Breakdown Voltage vs. Temperature.
1.200
1.000
0.800
0.400
0.600
0.200
0.000
-50 0 50 100 150
Junction Temperature (°C)
PI-2896-062802
Switching Frequency
(Normalized for 25 °C)
1.200
1.000
0.800
0.400
0.600
0.200
0.000
-50 -25 0 25 50 75 100 125 150
Junction Temperature (°C)
PI-2897-062802
Current Limit
(Normalized for 25 °C)
1.2
1
0.8
0.4
0.6
0.2
0
-50 0 50 100 150
Temperature (°C)
PI-2899-062802
PWM Gain (Normalized for 25 °C)
Figure 18. Switching Frequency vs. Temperature.
Figure 19. Current Limit vs. Temperature.
1.1
1.0
0.9
-50 -25 0 25 50 75 100 125 150
Junction Temperature (°C)
Breakdown Voltage
(Normalized to 25 °C)
PI-2213-012301
Figure 22. PWM Gain vs. Temperature.
1.200
1.000
0.800
0.400
0.600
0.200
0.000
-50 0 50 100 150
Junction Temperature (°C)
PI-2898-062802
IDCT (Normalized for 25 °C)
Figure 21. IDCT vs. Temperature.
Figure 20. I2f Coefficient vs. Temperature.
1.2
0.8
1.0
0.0-50 -25 0 25 7550 100 150125
Junction Temperature (°C)
I2f Coefficient
(Normalized for 25 °C)
PI-2910-071602
0.2
0.6
0.4
16
LNK500
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Typical Performance Characteristics (cont.)
Figure 23. Output Characteristics (DRAIN Current vs.
DRAIN Voltage).
Drain Voltage (V)
Drain Current (mA)
300
250
200
100
50
150
0
0 2 4 6 8 10
TCASE=25 °C
TCASE=100 °C
PI-2222-031401
17
LNK500
C
12/04
Notes:
1. Package dimensions conform to JEDEC specification
MS-001-AB (Issue B 7/85) for standard dual-in-line (DIP)
package with .300 inch row spacing.
2. Controlling dimensions are inches. Millimeter sizes are
shown in parentheses.
3. Dimensions shown do not include mold flash or other
protrusions. Mold flash or protrusions shall not exceed
.006 (.15) on any side.
4. Pin locations start with Pin 1, and continue counter-clock-
wise to Pin 8 when viewed from the top. The notch and/or
dimple are aids in locating Pin 1. Pin 6 is omitted.
5. Minimum metal to metal spacing at the package body for
the omitted lead location is .137 inch (3.48 mm).
6. Lead width measured at package body.
7. Lead spacing measured with the leads constrained to be
perpendicular to plane T.
.008 (.20)
.015 (.38)
.300 (7.62) BSC
(NOTE 7)
.300 (7.62)
.390 (9.91)
.367 (9.32)
.387 (9.83)
.240 (6.10)
.260 (6.60)
.125 (3.18)
.145 (3.68)
.057 (1.45)
.068 (1.73)
.120 (3.05)
.140 (3.56)
.015 (.38)
MINIMUM
.048 (1.22)
.053 (1.35)
.100 (2.54) BSC
.014 (.36)
.022 (.56)
-E-
Pin 1
SEATING
PLANE
-D-
-T-
P08B
DIP-8B
PI-2551-121504
D S .004 (.10)
T E D S .010 (.25) M
(NOTE 6)
.137 (3.48)
MINIMUM
PART ORDERING INFORMATION
LinkSwitch Product Family
Series Number
Package Identifier
G Plastic Surface Mount DIP
P Plastic DIP
Lead Finish
Blank Standard (Sn Pb)
N Pure Matte Tin (Pb-Free)
Tape & Reel and Other Options
Blank Standard Configurations
TL Tape & Reel, 1 k pcs minimum, G package only
LNK 500 G N - TL
18
LNK500
C
12/04
SMD-8B
PI-2546-121504
.004 (.10)
.012 (.30)
.036 (0.91)
.044 (1.12)
.004 (.10)
0 -
° 8°
.367 (9.32)
.387 (9.83)
.048 (1.22) .009 (.23)
.053 (1.35)
.032 (.81)
.037 (.94)
.125 (3.18)
.145 (3.68)
-D-
Notes:
1. Controlling dimensions are
inches. Millimeter sizes are
shown in parentheses.
2. Dimensions shown do not
include mold flash or other
protrusions. Mold flash or
protrusions shall not exceed
.006 (.15) on any side.
3. Pin locations start with Pin 1,
and continue counter-clock-
wise to Pin 8 when viewed
from the top. Pin 6 is omitted.
4. Minimum metal to metal
spacing at the package body
for the omitted lead location
is .137 inch (3.48 mm).
5. Lead width measured at
package body.
6. D and E are referenced
datums on the package
body.
.057 (1.45)
.068 (1.73)
(NOTE 5)
E S
.100 (2.54) (BSC)
.372 (9.45)
.240 (6.10) .388 (9.86)
.137 (3.48)
MINIMUM
.260 (6.60) .010 (.25)
-E-
Pin 1
D S .004 (.10)
G08B
.420
.046 .060 .060 .046
.080
Pin 1
.086
.186
.286
Solder Pad Dimensions
19
LNK500
C
12/04
20
LNK500
C
12/04
Revision Notes Date
B 1) Released Final Data Sheet. 3/03
C 1) Added lead-free ordering information. 12/04
For the latest updates, visit our website: www.powerint.com
Power Integrations may make changes to its products at any time. Power Integrations has no liability arising from your use of any information, device or circuit
described herein nor does it convey any license under its patent rights or the rights of others. POWER INTEGRATIONS MAKES NO WARRANTIES HEREIN
AND SPECIFICALLY DISCLAIMS ALL WARRANTIES INCLUDING, WITHOUT LIMITATION, THE IMPLIED WARRANTIES OF MERCHANTABILITY,
FITNESS FOR A PARTICULAR PURPOSE, AND NON-INFRINGEMENT OF THIRD PARTY RIGHTS.
PATENT INFORMATION
The products and applications illustrated herein (including circuits external to the products and transformer construction) may be covered by one or more U.S.
and foreign patents or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A complete list of Power Integrationsʼ patents
may be found at www.powerint.com.
LIFE SUPPORT POLICY
POWER INTEGRATIONSʼ PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS
WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein:
1. Life support devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when
properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user.
2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life
support device or system, or to affect its safety or effectiveness.
The PI logo, TOPSwitch, TinySwitch, LinkSwitch, DPA-Switch and EcoSmart are registered trademarks of
Power Integrations. PI Expert and PI FACTS are trademarks of Power Integrations. ©Copyright 2004, Power Integrations
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