AN377 - STMicroelectronics

APPLICATION NOTE

TEA5101A- RGB HIGH VOLTAGE AMPLIFIER

BASIC OPERATION AND APPLICATIONS

AN377/0594

By Ch. MATHELET

SUMMARY Page

I DESCRIPTION ........................................................ 3

I.1 INPUTSTAGE . ....................................................... 3

I.2 OUTPUTSTAGE . . . . . . . . .. . . . . . .. . . . . . . . . . . . . . . . . . . ................... 3

I.3 BEAM CURRENT MONITORING. . . ....................................... 4

I.4 PROTECTION CIRCUITS. . . .. . . . . . . . . . . . . . . . . . . . . . . . . ................... 4

I.4.1. MOS Protection. . . . .. . . . . . . . . . . . ....................................... 4

I.4.2. ProtectionAgainstElectrostaticDischarges . . . . . . . . . . . . . . . ................... 4

I.4.3. FlashoverProtection. ................................................... 4

II FUNCTIONAL DESCRIPTION............................................ 4

II.1 VOLTAGEAMPLIFIER. . . . . . . . . . . . . . . . . . . . . ............................. 5

II.1.1 Bias Conditions. ....................................................... 5

II.1.2. DynamicOperation. . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ................... 5

II.1.2.1. White To Black Transition . .. . . ......................................... 5

II.1.2.2. Black To WhiteTransition . ............................................. 5

II.2 BEAM CURRENT MONITORING. . . ....................................... 5

II.2.1. StationaryState. . . . .. . . . . . . . . . . . ....................................... 5

II.2.2. TransientPhase . . . . . . . . . . . . . . . . ....................................... 5

III EXTERNAL COMPONENTS CALCULATION ................................ 6

III.1 COMPONENTS VALUECALCULATION . . . .. . . . . . . . . . . . . ................... 7

III.1.1. Feedback resistor . . . ................................................... 7

III.1.2. Input resistor. . . . . . . . . . ................................................ 7

III.1.3. Bias resistor . . . . ...................................................... 7

III.1.4. Current measurement resistor . . . . . . . . . . . . . . . ............................. 7

III.2 DISSIPATEDPOWER . . . . . . . . . ......................................... 8

III.2.1. Measurementmethod................................................... 8

III.2.2. Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . ................... 8

III.2.2.1. Staticpower. .. . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .................... 8

III.2.2.2. Measurementwith sinusoidal input . . . .. . . . . . ............................. 8

III.2.2.3. Measurementin a TVset . .. . . ......................................... 9

III.2.3. Design of externalcomponents. . . . . ....................................... 9

III.2.3.1. Heatsink . .. . . . . . ................................................... 9

III.2.3.2. Powerrating of feedbackresistor . .. . . . . . . . . . . . . . . . . . . ................... 9

IV APPLICATION HINTS .................................................. 9

IV.1 DYNAMIC PERFORMANCES . . . . . ....................................... 9

IV.2 CROSSTALK . . . . . . . . . . . . ............................................. 10

IV.3 FLASHOVERPROTECTION . . . . . . . . . . . . . . . . . . . . . . . . . . ................... 10

IV.4 OUTPUTSWING . .. . . . .. . . . . . . . . . . . . . . . . . ............................. 12

IV.5 LOW CURRENT MEASUREMENTS . . . . . . . . . . ............................. 13

V APPLICATION EXAMPLES.............................................. 14

V.1 APPLICATIONDESCRIPTION. . . . . . . . . . . . . . . . . . . . . . . . . ................... 14

V.2 PERFORMANCES EVALUATION . . . . . . . . . . . . . . . . . . . . . . ................... 14

V.2.1. Measurementsconditions. . . ............................................. 14

V.2.2. Results . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . ................... 14

V.2.2.1. Bandwidth. . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . .................... 14

V.2.2.2. Crosstalk. .. . . . . . . . . . . . . . . . . . ....................................... 14

V.2.2.3. Transition times. . . . . . . . . . . . . . . ....................................... 21

1/21

The aim of this ApplicationNote is to describe the

basic operation of the TEA5101Avideo amplifier

and to providethe user withbasichints forthe best

utilization of the device and therealisationof high

performance applications. Application examples

are also provided to assist the designer in the

maximum exploitation of the circuit.

GENERAL

The control of state-of-the-art color cathode ray

tubes requires high performance video amplifiers

which must satisfy both tube and video processor

characteristics.

When considering tube characteristics (see Fig-

ures 13 and 14),wenote that a 130V cutoffvoltage

is necessary to ensure a 5mApeak current.How-

ever 150V is a more appropriatevalue if the satu-

ration effect of the amplifier is to be taken into

account. Asthe dispersionrange ofthe threeguns

is ± 12%, the cutoff voltage should be adjustable

from 130V to 170V. The G2 voltage, from 700 to

1500V allows overall adjustment of the cutoff volt-

age for similar tube types.

A 200V supply voltage of the video amplifier is

necessaryto achieve a correctblanking operation.

In addition, the video amplifier should have an

output saturation voltage drop lower than 15V, as

a drive voltage of 130V (resp. 115V)is necessary

to obtain a beam current of 4 mAfor a gun which

has a cutoff point of 170V(resp. 130V).

Note :Forallthecalculationsdiscussedabove,the

G1 voltage is assumed to be 0V.

The video processor characteristics must also be

considered. As it generallydelivers an outputvolt-

age of 2 to 3V, the video amplifier must provide a

closedloop DC gain of approximately40.

The video amplifier dynamic performances must

also meet the requirementsofgooddefinitioneven

with RGBinputsignals(teletext,homecomputer...),

e.g. 1mmresolutionona 54cmCRTwidthscanned

in 52µs. Consequently, a slew rate better than

2000V/µs, i.e. rise and fall times lower than 50ns,

is needed.In addition,transitiontimesmust be the

sameforthethreechannelssoas toavoidcoloured

transitionswhen displaying white characters. The

bandwidth of a video amplifier satisfying all these

requirementsmust be at least 7MHz for high level

signals and 10MHz for small signals.

One major feature of a video amplifier is its capa-

bility to monitor the beam currentof the tube. This

function is necessary with modern video proces-

sors:

- forautomaticadjustmentof cutoffandalso,where

required,videogain in order to improve the long

term performances by compensation for aging

effects through the life of the CRT. This adjust-

ment can be done either sequentially(gun after

gun) or in a parallel mode.

- for limiting the average beamcurrent

Avideo amplifiermust also be flashover protected

and provide high crosstalk performances. Cros-

stalkeffectsare mainlycaused byparasiticcapaci-

tors and thus increase withthe signalfrequency.A

crosstalk level of -20dB at 5MHz is generally ac-

ceptable.

Table 1 summarizes the main features of a high

performancevideo amplifier.

Table 1 : Main Featuresof aHigh Performance

Video Amplifier

Maximum Supply Voltage 220V

Output voltage swing ”Average” 100V

Output voltage swing ”Peak” 130V

Low level saturation (refered to VG1) 15V

Closed loop gain 40

Transition time 50ns

Large signalbandwidth 7MHz

Small signalbandwidth 10MHz

Beam current monitoring

Flash over protection

Crosstalk at 5MHz -20dB

The SGS-THOMSONMicroelectronics TEA5101A

is ahighperformanceandlargebandwidth3 chan-

nel video amplifier which fulfills all the criteria dis-

cussedabove. Designed in a 250V DMOS bipolar

technology,it operates with a 200V power supply

and candeliver 100V peak-to-peakoutput signals

with rise and fall times equal to 50ns.

The 5101A features a large signal bandwidth of

8MHz, which can be extended to 10MHzfor small

signals(50 Vpp).

Each channel incorporates a PMOS transistor to

monitor the beam current. The circuit provides

internal protection against electrostaticdischarges

and high voltage CRT discharges.

The bestutilization of theTEA5101Ahigh perform-

ance features such as dynamic characteristics,

crosstalk,or flashover protection requires opti-

mizedapplication implementation. This aspect will

be discussed in the fourthpart of this document.

TEA5101A APPLICATION NOTE

2/21

40kΩ

0.8kΩ

1kΩ

(12, 9)15VDD

(11, 6)

20kΩ

2.5kΩ

6kΩ

1.5kΩ

350Ω

35Ω35Ω3pF

GND

(3, 4)

(10,7)

5101A-02.EPS

Figure1

I - DESCRIPTION

The completeschematicdiagram of one channelof the TEA5101Ais shownin Figure1.

I.1 - Input Stage

Thedifferentialinputstageconsistsofthetransistor

T1and T2and theresistors R4,R5andR6.

This stage is biased by a voltagesource T3,R1,R2

and R3.

VB(T1)=(1+R

3)xV

)≅3.8V

Each amplifier is biased by a separate voltage

source in order to reduce internal crosstalk. The

load oftheinputstage iscomposedof thetransistor

T4(cascodeconfiguration)andtheresistor R7.The

cascode configuration has been chosen so as to

reduce the Miller input capacitance. The voltage

gain ofthe inputstageisfixedby R7andtheemitter

degenerationresistorsR5,R6,andtheT1,T2internal

emitter resistances. The voltage gain is approxi-

mately 50dB.

Using a bipolar transistor T4and a polysilicon re-

sistor R7gives rise to a very low parasitic capaci-

tance at the output of this stage (about 1.5pF).

Hence the rise and fall times are about 50ns for a

100V peak-to-peak signal (between 50V and

150V).

I.2 - Output Stage

The output stage is a quasi-complementaryclass

B push-pullstage.Thisdesignensuresasymetrical

load of the first stage for both rising and falling

signals. The positive output stage is made of the

DMOS transistor T5,and the negative output stage

is madeof the transistorsPMOST6andDMOST7.

The compoundconfigurationT6-T7isequivalentto

a singlePMOS. Asingle PMOS transistorcapable

of sinking the total current would have been too

large.

By virtue of the symetrical drive properties of the

output stage the rise and fall times are equal(50ns

for 100V DC outputvoltage).

TEA5101A APPLICATIONNOTE

3/21

I.3 - Beam Current Monitoring

This function is performed by the PMOS transistor

T8insourcefollower configuration.The voltage on

the source (cathode output) follows the gate volt-

age (feedback output). The beam current is ab-

sorbed via T8. On the drain of T8, this currentwill

be monitoredby thevideoprocessor.

I.4 - ProtectionCircuits

I.4.1 - MOS protection

Four zener diodesDZ(1-4) are connectedbetween

gate and sourceof each MOS in order to prevent

the voltage from reaching the breakdown volt-

age.Hence the VGS voltage is internally limited to

±15V.

I.4.2 - Protectionagainst electrostatic dis-

charges

All the input/outputpins of the TEA5101Aare pro-

tected bythe diodesD1-D7which limit the overvol-

tage due to ESD.

I.4.3 - FlashoverProtection

A high voltage and high current diode D5is con-

nected between each output and the high voltage

power supply. During a flash, most ofthe current is

generally absorbedby thespark gap connectedto

the CRTsocket.Theremaining currentisabsorbed

by the high voltage decoupling capacitor through

the diode D5. Hence the cathode voltage is

clampedto the supply voltage and the output volt-

age does not exceedthis value.

II - FUNCTIONAL DESCRIPTION

The schematic diagram of one TEA5101Achannel with its associated external components is shown in

Figure 2.

6kΩ

1.5kΩ

T2T1

350Ω

2.5kΩ

R6R5

35Ω35Ω

Input

3pF

Dz2Dz1

40kΩT5

T6 T7

0.8kΩ

1kΩ

R10

20kΩ

Dz3

D7 6

Video Processor

Dz4 D5

9Feedback

Output

(12, 15)

RfC

VDD

(3, 4)

GND

CRT

(10, 13)

5101A-04.EPS

Figure2

TEA5101A APPLICATION NOTE

4/21

II.1 - Voltage Amplifier

II.1.1 - Biasconditions Vin =V

ref

The bias point is fixed by the feedback resistor

Rf,the bias resistor Rp, and by the internal refer-

ence voltagewhen Vin =V

ref.

IfVOis the output voltage (pin 9) :

VO=(1+Rf

Rp)xV

ref (1)

In this state T1and T2are conducting. A current

flows in R7andT4soT5is on. TheT5draincurrent

is fed to the amplifier input through the feedback

resistor. The current in R7is:

I(R7)=VDD −VO−VGS(T5)

R7≅VDD −VO

and the current in T5and Rfis :

I(T5)=VO−Vref

Rf≅VO

Thus the totalcurrentabsorbedby eachchannelof

the TEA5101Ais :

VDD

R7+VOx(1

Rf−1

R7)

The cathode(pin 7) output voltage is:

VO+VGS(T8)=VO

The beam currentis absorbed by T8and Rm. The

voltage developedacross Rmbythis currentis fed

to thevideoprocessorin order to monitor thebeam

current.

II.1.2 - Dynamicoperation

TheTEA5101Aoperatesasaclosedloopamplifier,

with its voltage gain fixed by the resistors Rfand

Re.

SincetheopenloopgainAisnotinfinite,theresistor

Rpand the input impedance Rin must be consid-

ered.Hence the voltagegain is

G=− R

ex1

1+1

A(1+R

p⁄⁄R

e⁄⁄R

in)(2)

II.1.2.1- Input voltageVin <V

ref (blackpicture)

In this case the current flowing in R7and T1de-

creases whilst the collector voltage of T4and the

output voltage bothincrease. In the extremecase,

I(T1) = I(R7) = 0 andVO=V

DD-VGS(T5)

In orderto chargethe tubecapacitorthe voltageis

fed to the cathode output in twoways:

- throughthePMOS (witha VGSdifference)forthe

low frequencypart

- through the capacitor C for the high frequency

part(output signalleading edge)

To correctly transmit the rising edge, the value of

the capacitorC must be high compared to CL.

With the current values used (C = 1nF,CL=10pF),

the attenuationis very small(0.99)

II.1.2.2 - Input voltage Vin >Vref (white picture)

In thiscase,thecurrent in R7andT1increaseswith

anaccompanyingdropofT4’scollectorvoltageuntil

T1and T4are saturated. At this point:

VO≅VC(T4)≅VCC

During a high to low transition (i.e. black-white

picture),thebeamcurrentisabsorbedintwoways:

- through the capacitor C and the compound

PMOS T6-T7for the high frequency part (falling

edge)

- throughthe PMOST8and the resistor Rmfor the

low frequencypart.

II.2 - Beam Current Monitoring

II.2.1 - Stationary state

The beam current monitoring is performed by the

PMOST8andtheresistorRm.Whenmeasuringlow

currents (leakage, quasi cutoff),the Rmvalue is

generally high. When measuring high currents

(drive, average or peak beam current),Rmis gen-

erally bypassed by a lower impedance.

It should be noted that the currentsupplied by the

three guns flows through this resistor.Hence,with

too large a value for the resistor Rm,the cathode

voltage of the tubes will become too high for the

required operating current values.This is a funda-

mental difference betweenthe TEA5101Aand dis-

crete video amps. In discrete video amps, the

currentmonitoring transistor isa high voltagePNP

bipolarwhich may saturate. In thiscase the beam

current can flow through the transistor base and it

is nolongermonitoredby thevideoprocessor.This

effect does not occur with the TEA5101A.

II.2.2 - Transientphase : low current measure-

ments

The cut-off adjustment sequence is generally as

follows:

In a first step, the cathode is set to a high voltage

(180V) in order to blank the CRT and to measure

the leakage current. In a second step, the tube is

slighly switched on to measure a very low current

(quasi cut-offcurrent). This operation isperformed

by setting the cathode voltageto about 150V and

adjustingit until thepropercurrent isobtained.The

maximum time available to do this operation is

generally about 52µs.

Figure 3 shows the simplified diagram of the

TEA5101Aoutput,thevoltagesduringthedifferent

steps,and the stationary state the system must

reach for correct adjustment.

TEA5101A APPLICATIONNOTE

5/21

VC1.5V

180V

181.5V

BLANKING CUT-OFF

151.5V

2.5V

150V

152.5V

τ2

τ2=RxC=1

τ1= R x C = 10ns

1nF

1kΩ

5101A-05.EPS

Figure3

Duringtheblankingphase,thetubeisswitchedoff,

the PMOS is switched off and its VGS voltage is

equal to the pinch-off voltage (about 1.5V). The

voltages at the different nodes are shown in figure

3 (V(9)= 180V, V(k) =181.5V). The falling edge of

the cutoff pulse is instantaneouslytransmitted by

the capacitor C. When the stationary state is

reached,the cathodevoltage will be 152.5Vif the

voltage on pin9 is 150V,as the VGSvoltage ofthe

conductingPMOSis about2.5V.

We cansee thatthe voltage onC must increase by

an amountof∆Vc = 1V. This chargeis furnishedby

the tube capacitor which is discharged by an

amount of ∆VCL= 29V with a time constant equal

to R x CL(10 ns). By considering the energy

balance, we can calculate the maximum charge

∆Vmax that CLcan furnishedto C

∆Vmax =√

Cx∆VCL ≅3V

Sincethisvoltageisgreaterthan∆VC, thecapacitor

C can be charged and the stationary state is

reached without any contribution being required

from the tube current,i.e. the whole tube current

canflowthroughthePMOSandtheadjustmentcan

be performedcorrectly.

Considering higher voltage and beam current

swings, the margin is greater because:

- the voltage swing across the tube capacitor is

greater

- the tube current is higher and the picture is not

disturbedeven if part of thebeamcurrent isused

to chargethe capacitorC.

III - EXTERNAL COMPONENTS CALCULATION

The implementation of the TEA5101Ain an appli-

cationrequiresthe determinationof externalcom-

ponent values. These components are Rf,R

and Rm(seeFigure 4).The dissipatedpowerin the

IC and in the feedback resistor Rfmust also be

calculated in order to correctly choose the power

ratings of the heatsink and resistors.

TEA5101A APPLICATION NOTE

6/21

III.1 - Components Value Calculation

From equations 1 and 2 in section II-1, both the

value oftheDC outputvoltage andthevoltagegain

depend directly on the resistor Rf. HenceRfmust

be determinedfirst before calculating the value of

Reand Rpin order to obtain the correct gain and

DC outputvoltage.

III.1.1 - Feedback resistorRf

The value ofRfmust beas low as possiblein order

to obtain the optimum dynamic performance from

the TEA5101A(seesectionIV-1).Atypicalvalueof

Rfis39 kΩ.

III.1.2 - Input resistorRe

The voltage gain is calculated from the following

formula (see section II-1):

G=−R

1+1

A(1+R

p⁄⁄R

e⁄⁄R

in)

Since the open loop gain A is high enough(50dB),

we canapproximatethe calculation:

G @−Rf

where Reis generally implemented as a variable

value for channelgain adjustment.

If the gain adjustmentrange Gmin,G

max is known:

Re min =Rf

Gmax and Re max =Rf

Gmin

With Gmin = 15 and Gmax =80:

ewillbemadeofa 2.2kΩpotentiometerand470Ω

fixed resistor.

III.1.3 - Bias resistor Rp

Rpmust be chosen in such a way that the black

level outputvoltage VOUT(BLK) isequal to the cutoff

voltage, which is a characteristic of the tube cur-

rently used, when the DC black level input voltage

VIN(BLK) isthe meanvalueof the adjustmentrange

of the video processor. This is the optimum condi-

tion to ensure a correct adjustment during the

lifetime of the tube. Rpcan be calculated by con-

sidering the TEA5101Aas an operational amplifier

and applying the usualformula :

Rp=Vref

Vout (BLK)−V

ref

Rf+Vin (BLK)−Vref

-IfV

in(BLK)=V

ref Rp=Vref

Vout (BLK)−Vref xR

For a 150Vblack level :

Rp =1kΩwith Rf=39kΩ

-IfV

in (BLK) ≠Vref:

Rp= 1.2kΩwith Vin (BLK)= 2.7V

Rf=39kΩ

Re= 1.5kΩ

Rp= 680Ωwith Vin (BLK)= 6.7V

Rf=39kΩ

Re= 1.5kΩ

fora 150Vblack level

III.1.4 - Current measurement resistor Rm

Rmmustbe determinedby takinginto accountthe

quasi cutoffcurrent Ico and theinput voltageVCof

the videoprocessor.

Rm=VC

ICO

- Withthe videoprocessorTEA5031D (VC=2V) :

Rm=120kΩwith ICO =16µA

REF 7

VIDEO

PROC.

IC.O.

VIN(BLK)

VOUT(BLK)

5101A-06.EPS

Figure4

TEA5101A APPLICATIONNOTE

7/21

IC.O

120kΩ

82kΩ

TDA3562A

Pin18

12V

5101A-07.EPS

Figure5

- With the videoprocessorTDA3562A(VC= 0.5V)

which requires a DC biased input ”Black current

stabilization” (pin 18), the schematicdiagram is

the following:

The DCbias is 12 x 82

120 +82 =5V

The quasicutoff current is

0.5 (1

120 +1

82)x1x10

−3=10µA

III.2 - DissipatedPower in External Compo-

nents

The only components dissipating power are the

TEA5101A and the feedback resistor. The dissi-

pated power has a constant static componentand

a dynamic component which increases with fre-

quency. The theoretical calculation is not suffi-

cientlyaccuratetodeterminethe correctdissipated

power. The best way consists of measuring the

power in different configurations of the circuit:

steady state (no input), sinusoidal input,and in situ

(in a TV set with a video input signal). The meas-

urementmethodwill bedescribed firstand thenthe

results and calculationswill be discussed.

III.2.1 - Measurementmethod

The dissipatedpowercanbedeterminedby meas-

uring the average supply current IDD (principally

high voltagesupplycurrentVDD)andbysubtracting

the power dissipated in the external components

from the calculatedpower delivered by this supply

voltage.

The power delivered by the high voltage power

supplyis : P= VDD xI

The power dissipated in the external components

(principallythe feedbackresistor Rf)is:

- for the static part: PSR=3xV

2OUT (AVG)

- for the dynamic part: PDR=3xV

2OUT (RMS)

When the IC is driven by a sinusoidal signal (ca-

pacitive drive),the measurement and calculation

are straightforward:

-V

OUT(AVG) = VOUT(DC)

-V

OUT(RMS) = VOUT(peakto peak)

2x√2

With VOUT (DC)= 100V and

VOUT (peak to peak) = 100V andRf= 39kΩ

PSR = 0.8W

PDR = 0.1W

Measurements are more difficult to carry out when

the IC is working in a TV set. VOUT(AVG) can be

measuredwith an oscilloscope (difference of level

betweenACandDCcoupling)andVOUT (RMS) can

be measured by connectingan RMS voltmeter to

the feedback resistor. In this case we have the

followingresults (see section 2.2.3) :

-V

OUT (AVG) =130V and PSR = 1.3W

-V

OUT (RMS) = 32V and PDR = 80mW

In each case, the term PDR can be neglected as a

reasonableapproximation.Hence,the powerdissi-

patedby the IC willbe:

Pi=V

DD xI

DD -3V2OUT (AVG)

and the power dissipated in Rfwill be :

Pr=V2OUT (AVG)

III.2.2 - Results

III.2.2.1 - Static power

Table2 shows the measured valuesof IDD andthe

calculated power for three values of Vout and for

VDD = 200V

Table 2

VOUT (V) IDD (mA) Pi(W) Pr(W)

50 16 3 0.065

100 15 2.2 0.25

150 14.6 1.2 0.6

We can see that the static power dissipated in the

IC decreases with VOUT increasing, but obviously

the power dissipated by Rfincreases as VOUT

increases.

III.2.2.2 - Measurementwith sinusoidal input

Table3 summarizestheresultsobtainedfromprac-

ticalmeasurementsas functionsof VOUT(DC) and

of the frequency (the three channels are driven

simultaneously).

We can see that when driving the IC with a HF

sinusoidal signal, care must be taken to avoid

excessive temperatureincrease.

TEA5101A APPLICATION NOTE

8/21

Table 3

VOUT

(V) IDD

1MHz

(mA)

IDD

7MHz

(mA)

VOUT (PP)

1MHz

(V)

VOUT (PP)

7MHz

(V)

1MHz

(W)

7MHz

(W) Pr

(W)

50 20.7 44.6 66 50 3.9 8.7 0.065

100 20 59.5 100 80 3 11 0.25

150 18 45 100 67 1.7 8.2 0.6

III.2.2.3 - Measurement in a TV set

Wehavedeterminedtheworst casesofdissipation

in a TV set. These trials have been carried out on

one particular TV set, and may not be repre-

sentative for all TV sets. In this particular TV set,

the worst cases of dissipation occur with noise

signal (fromHF tuner)and witha multiburstpattern

(0.8 to 4.8MHz)in RGB mode.

Table 4 summarizes the resultsin these two cases

when the brightness control is set to minand max

value (the contrastcontrol is set to max).

Table 4

VOUT

(AVG)

(V) IDD

(mA) VDD

(V) Pi

(W) Pr

(W)

Bright.max Noise

Bright.min 148

188 22.2

23.3 218

224 3.15

2.5 0.56

0.9

Bright.max Multiburst

Bright.min 131

158 23.6

22 213

221 3.7

2.9 0.44

0.64

III.2.3 - Design of heatsink and external com-

ponents

III.2.3.1 - Heatsink

As discussed above, the power dissipated in the

IC in a TV set can reach about 4W. In this case, a

12oC/W heatsink seems to be sufficient. Such a

heatsinkwill give Tj= 115oC for Troom =60

The resulting margin guaranteescorrect reliability.

III.2.3.2 - Feedbackresistors

1 Watt type feedback resistors must be used, as

they may need to dissipate0.9W when the TVset

is working and up to 1W when the TV is blanked

(VOUT = 200V), for example when the security of

the scanningprocessoris activated.

IV - APPLICATION HINTS

IV.1 - Dynamic Performances

Figure 6 shows the simplified schematic diagram

of the TEA5101Ain AC mode.

RpCIN

A(s)

5101A-08.EPS

Figure 6

Cfis the parasiticcapacitor between the input and

the output.

Cinis theparasiticcapacitorbetween the input and

ground. The voltage gainversus frequency canbe

deduced from the formula (2) in chapter II sec-

tion 1.2 :

G(s) = Rf

Re(1+RfCfs)⋅1

1+1

A(s)



1+Rf

Req

1+Req Cin s

1+RfCfs





with Req=Rp//Re//Rinand A(s) open loop gain

A(s) is a second order function such as

1+bs +as2

with a = 9 x 10-16 s2,b=60x10

-9 s,AO=400

Assuming Req xC

in =RfxC

, we find:

G(s)=− R

e(1+R

fs)x1

1+B

AO +Bbs +B

AO+Bas2

with B =1 + Rf

Req

We see that the closed loop amplifieris equivalent

to acombinationofasecondordercircuitanda first

orderone.The lattercomprisesthe feedbackresis-

tor and the parasitic capacitor between input and

output.

With the current values : Rf= 39kΩ,R

e=2kΩ,

in = 14kΩ,R

p=1.2kΩ,C

f=0.5pF,Cin = 15pF

TEA5101A APPLICATIONNOTE

9/21

we have Req xCin = 10ns, RfxC

f=20ns, B =56

The second ordercircuitcharacteristicsare :

Natural frequency :

Fn=1

2xπxaxAO+B

B=15MHz

dampingfactor :

z=b

2xaxB

AO +B=0.35

The cutoff frequencyof the first order circuit is :

fC=1

2xπR

fxC

f=8MHz

The amplifier response is thus the combination of

the responses of these two circuits. The contribu-

tion of the parasitic capacitor Cfto the frequency

response is very important. If the value of Cfis too

high, the contributionof thefirst ordercircuit willbe

of overriding importance and the resulting band-

width of the amplifier will be too small. If the value

ofCfis toolow, theresponsecurvewill havea peak

(due to the secondorder circuit). A ”ringing” effect

will be present on pulse-type signalsand an insta-

bilityandoscillationcanoccurat somefrequencies.

This capacitoris generallytoo high.It consists of:

- the self parasitic capacitor of the feedback resis-

tor

- the parasitic capacitordue to the PCB layout.

Practically,the best bandwidth performances are

achievedby:

- thesmallestinput-outputcapacitorandthesmall-

est capacitorbetween an input and ground

- using a feedback resistor with the smallest pos-

siblevalue butlarge enoughto yielda sufficiently

high gain.

- using a feedback resistor with small parasitic

capacitance (typ 0.2pF). Some resistors have

0.5 or 0.8 pFparasitic capacitor.

The parasiticcapacitors discussed aboveare usu-

allythe oneswhich need to be taken into account.

However any other parasitic capacitor or inductor

canmodify thefrequencyresponse.For instance,a

too large capacitor value between the feedback

outputand groundcan createa dominantpoleand

causea potentialrisk of oscillation.

IV.2 - Crosstalk

Figure7 showsthe differentparasiticlinksinducing

crosstalk.

The crosstalkcan be causedby:

- parasiticcoupling betweenthe inputs (Cpi)

- parasiticcoupling betweenthe outputs(Cpo)

- parasiticcoupling between an outputand a near

input of anotherchannel (Cp).

CpCpo

5101A-09.EPS

Figure 7

Parasiticcouplingmaybe capacitiveor be caused

by HF radiations.

The third type of parasitic coupling is predominant

since it involves the addition by feedback at rela-

tively high level(output) signals to relatively low

level(input)signals.Forexample,a0.1pFCppara-

sitic capacitor between an output and the input of

another channel will act asa differenciatorwith the

feedbackresistor Rf= 39kΩ.

The transfer function of this integrator will be Rfx

Cpxs( 0.2jat 8MHz)and thusthe crosstalkwill be

-14dB at 8MHz. The parasitic coupling between

inputs and outputs must be minimized to achieve

an acceptablecrosstalk(-20dBat 5MHz). Thiscan

be doneby crossingonly the inputwires and sepa-

rating the input and output leads. High voltage

components and wires must be laid out as far as

possiblefrom small signalwires,even if this results

in a larger circuit board.

HF radiations from the feedbackresistor must not

induced a voltage signal at the input of another

channel.This can be achievedby :

- spacingout the feed back resistors

- mounting these resistors in the same direction

and strictlyaligned one underanother.

- mounting these resistors 1cm above the PC

board

- using ground connections to insulate the input

wires

IV.3 - Flashover Protection

A picture tube has generally several high voltage

dischargesin its lifetime.This is dueto thefact that

the vacuum is not perfect coupled with the pres-

enceof metallicparticlesevaporatedfromtheelec-

trodes.Hence,short circuits (very brieffortunately)

can occur between two electrodes,one of which is

usually the anode (at EHT potential). An overvol-

tage can be induced on the cathodes or on the

TEA5101A APPLICATION NOTE

10/21

suppliesevenifaflashoccursonanelectrodeother

thana cathode,becauseofthepossibilityof flashes

in series or overvoltagesdue to inductive links on

the video board or on the chassis.These overvol-

tages can destroy an IC particularlythe video am-

plifierwhichisthemostvulnerablesinceitisdirectly

connectedto the tube.

The tube manufacturers have made much pro-

gress in technology in order to reduce the fre-

quency of flashes and their associated energy

(increased quality of vacuum, internal resistance

for ”softflash” tubes). Nevertheless, some protec-

tionmeasuresare suggestedbythetubemanufac-

turers :

- connect spark gaps on eachelectrode

(1 to3kV or12kV forfocus)

- connect the spark gaps to a separated ground

directlyconnectedtothechassisgroundby anon

inductivelink

- connectthecathodesor gridsbyprotectiveresis-

tors.These resistors must be able to withstand

12kV (20kV for focus)instantaneous voltages

without breakdown and without any change of

value following successive flashes. Theseresis-

tors must beof a non-capacitivetype.1/2W (1W

for focus) hot molded carbon type resistors are

well suited for this application.

- the grid and cathode connections on the PC

board must be as short as possible and spaced

well away from other connections in order to

avoid parasiticinductions.

Furthermore, the TEA5101A has been provided

with an additionaleffective feature to improve the

flashoverprotection.Asdescribed in section I-4, a

protection device has been included comprising a

high voltage high current diodewhich is connected

between each output and the high voltage power

supply.Theequivalentdiagramof thisprotectionis

shownin Figure 8.

CHASSIS

VDD

SPARK GAP

2KV

TEA5101A

5101A-10.EPS

Figure8

The flash current is diverted to the ground through

the diode and the decouplingcapacitor C.

Two kinds of flashescan occur:

1) low resistance flashes during which the spark

gaps are activated since the cathode voltage ex-

ceedsthebreakdownvalueof thespark gap.In this

case the equivalent diagram is the following :

Ctube

1nF SPARK GAP

2KV

5101A-11.EPS

Ifflash current (≅1000A)

Lfinductance of the connection (≅10µH)

Figure 9

Ctube previously charged to 28kV is instantane-

ously dischargedduring

∆t=Ctube x∆V

If=30ns

Sincethe voltageacross thespark gapfallsalmost

instantaneouslyto 2000V, the peak current I flow-

ing intothe diodeis (assuming VCis held by good

decoupling):

I=Vex∆t

Lf=6A

To ensurea variation of VClessthan 10V,C must

C>Ix∆t

∆V

Ceg C >18nF

The decoupling must have good HF charac-

teristics.

2) high resistance flashesin which the sparkgaps

are not activated. In this case the equivalentdia-

gram is the following :

ID5

R1kΩ

Ctube

1nF V

5101A-12.EPS

Figure 10

TEA5101A APPLICATIONNOTE

11/21

IfV<2kV,I< 2000

R, I < 2A andRt≅ 12kΩ

The timeconstantof the flashis RtxC

tube =12µs,

the decaytime isapproximatevely30µs. Thevalue

of C must be

C>∆txI

∆V

Ceg C >6µF

in orderto ensure a VC variation less than 10V.

The total decoupling will be made up by a 10µF

electrolytic capacitor connected in parallel with a

22nF plasticfilmcapacitorwith goodHFproperties.

It must beplaced veryclose to theTEA5101Ato be

efficient.Otherwise,the equivalentdiagram will be

the following(case of low resistance flash).

220V

LP2

≈1µH

R5I

CHASSIS

SPARK

GAP

Ctube

5101A-13.EPS

Figure11

∆VC=Ix∆t

C+L

P1 xI

∆t

∆V

C= 210V with Lp1 =1 µH andLp2 =0

InthiscasetheVDD voltagecanrise toadangerous

value (+210V increase) and the protection is not

efficient.

If the connection between the socket ground and

the chassisground isinductive (Lp2 ≠0), theeffect

is the same.

However in this case, all the TV IC’s,and not only

the TEA5101A,willbeexposedto destructiveover-

voltages.

IV.4 - Output Swing

The simplified diagram of this function is shown in

Figure 12 (see Chapter II and chapterIII ).

The current delivered by a CRT is given by the

R1Rm

CRTTEA5101A

5101A-14.EPS

Figure 12

characteristiccurves (Figures 13 and 14).

The minimum value of Vk(due to all the voltage

dropsin the resistorsand in theamplifier)is given

by the equation (see Figure 12) :

Vk=(R+R

on +R1+3xRm

)xI

t=Req xI

t(1)

with Ron : on state PMOS resistance

To findthe maximum available currentItmax,we can

draw the curves of the equation (1) on the tube

characteristics. Itmax will be given by the intersec-

tion point of the curves.Since the tube charac-

teristics are: ItvsVcutoff +V

G1 -V

kthe equation(1)

must be changedto

It=VCUTOFF +VG1 −Vk

Req (2)

Assuming VG1= 0, we can draw the curves of

equation (2) for several values of Vcutoff (eg 150V

and 200V) and several values of Req

(eg 5k,10k,15k,20k)(see Figures 13 and 14). We

can see fromthese curves that Req must have the

following values to allow the tube to source 4mA

per gun :

Req ≤5kΩfor a 150Vcutoff point

or Req ≤15kΩfor a 200Vcutoff point

As Ron value is approximatively1.7kΩ, the meas-

urementresistor must be as low as possible.

Working with highercutoff point would be an alter-

native solution. But a 200V cutoff point seems to

be too high a value since in this case the supply

voltage would be greater than 200V and would

affect reliability performances.

TEA5101A APPLICATION NOTE

12/21

10 10 10

Heater Voltage - 6.3V

Anode-to-grid No 1 Voltage - 25kV

Grid No 3 -to-grid No 1 Voltage (each gun)

adjusted to provide spot cut-off

-Zero - Bias point

150V

200V

100V

5kΩ

10kΩ

15kΩ

20kΩ

E=50V

Kc1

VIDEO SIGNAL VOLTAGE PER GUN

V=V -V

CUT-OFF G1K CUT-OFF

V = 150V)(

ANODE CURRENT PERGUN (µA)

5101A-15.EPS

Figure13

10 10 10

Heater Voltage- 6.3V

Anode-to-grid No 1 Voltage - 25kV

Grid No 3 -to-grid No 1 Voltage(each gun)

adjusted to provide spot cut-off

-Zero - Bias point

100V

5kΩ

10kΩ

15kΩ

20kΩ

VIDEO SIGNAL VOLTAGE PER GUN

V=V -V

CUT-OFF G1K CUT-OFF

V= 200V)(

ANODECURRENTPERGUN ( µA)

E=50V

KcL

200V

150V

5101A-16.EPS

Figure14

Another solution consists of connecting a zener

diode as shown in Figure 15.With this device the

5101A-17.EPS

Figure 15

high current operation of the TEA5101Ais similar

to thatof a discrete amplifier (with PNP) operation.

Forlowcurrents,ifthezenervoltageis greaterthan

the VGS voltage, the zener diode is biased off and

the beam current flows through the measurement

resistor. When the cathodevoltage (pin 7) drop is

limited because of the pin 6 voltage and when the

pin 9 voltage continues to decrease,thezener di-

ode is switched on when V7-V

9=V

. In thiscase

the beamcurrentis absorbedby thevoltageampli-

fier and the tube can provide larger current val-

ues.Nevertheless, the pin 7 output voltage will

follow the pin 9 voltagewith a VZdifference.

Sincethe pin 9 voltageis internallylimited to 14V,

the output voltage will be limited to 22V with a 8V

zenerdiode.

The CRT bias voltages shown on the previous

curves are referenced to the G1voltage. The

TEA5101Aisreferencedtoground.Wecanchoose

towork with a G1voltage greater than ground and

thus the low level saturation is not taken into ac-

count. In this case, the cutoff points must be in-

creased. When choosing VG1 = 12V, the cutoff

points will be adjusted to 170V (instead of 150V).

Since the powersupply is 200V,30V are available

to ensure correct blanking operation.The DC out-

put voltage must be increased by 12V from its

previousvalue.

Note that all the phenomena described in this

section concern a static or quasi-static (15kHz)

operation (e.g. white picture or rather large white

pattern on a black background). When current

peaks occur (e.g white characters insertion or

straight luminance transition), the peaks will be

absorbedbythecouplingcapacitorandthevoltage

amplifier,andhence thetube willbe ableto source

a greatercurrent.

IV.5 - Low Current Measurements

We have seen in section II-2.2 how the beam

current monitoring works (see Figure 3). We have

TEA5101A APPLICATIONNOTE

13/21

1N4148

5101A-18.EPS

Figure16

seen that the capacitor C must charge again after

the blanking phase.

This charge is generally furnished by the tube

capacitor independently from the beam cur-

rent.However,ifduringthe blankingphase, the out-

put voltage is too low (e.g. the PMOS is reverse

biased (-20V) because of a too high leakage cur-

rent or when measuring with an oscilloscope

probe),the ∆VC required to charge C again will be

greater than the maximum charge available from

the tube capacitor.Hence the beam current will

have to charge C in a first step.

Since this current is rather low during the cutoff

adjustement phase, a long time will be spent to

chargeC. The currentabsorbed bythe PMOSand

fed to the videoprocessorwill not be equal to the

beamcurrentandthecutoffadjustementwill notbe

correct.

Hence the reverse voltage across the capacitor C

mustbe limited by a diode connectedas follows:

With thisconfiguration,the voltageacross Cwill be

-0.6V max. Since this voltage must be 2.5V in the

stationary state (see section II-2.2), the voltage

across C must be increased by 3.1V and this

charge canbe supplied by CL. We canalso slightly

decrease the value of C. However if C is too low,

the HFbehaviour will be impaired.

V - APPLICATION EXAMPLES

V.1 - Application Description

Figures 17 and 18 show two applications,one for

a 45AX tube and the videoprocessor TDA3562A

(application1), the otherdesignedforS4 typetube

and thevideoprocessorTEA5031D(application2).

In these twoapplications, the nominalgain is 28dB

and the output black level is 150V.The quasi cutoff

currents are respectively 10µAand16µAfor appli-

cations 1 and2.

These applications are implemented using the

same PCboardespeciallydesignedto allowdiffer-

ent options for tube biasing, power supply decou-

pling and connections.This PC board allows also

two differenttubesockets(jedecB8274orB10277)

to be connected. Both beam current monitoring

modes(sequentialand parallel)are possible.

The layout and the electrical diagram of the PC

boardare shownin Figures19 and 20.

V.2 - PerformanceEvaluation

As seen in chapter IV, the dynamicperformances

(bandwidth, crosstalk) of the TEA5101A is very

dependent on the PCB layout.Consequently, the

evaluationboard has been designed to obtain the

best results.

To evaluate the performance, the best way is to

work outside of the TV set by driving the amplifier

by an HF generator (or a network analyser) while

simulating the load conditions fixed by the CRT,

since AC performancesare directly determined by

the load.

V.2.1 - Measurement conditions

The schematic diagramsof the AC measurements

areshowninFigures21and 22.The conditionsare

as follows :

-BIASING : V

OUTDC = 100V by choosing

R11 =R21 =R31 = 1.5kΩand VDD = 200V

- AC GAIN = 50 by adjustingP10, P20, P30

- LOADING:

- by a 8.2pF capacitor and the probe capacitor

(2pF), the sumis equivalentto the capacitance

of a CRT with the socketand the spark gaps

- the 1MΩresistors connected between each

output and VDD allow the conduction of the

beam current monitoring PMOS transistor in

such a way that VADC =VB

DC= 100V.

- DRIVING by a 1µF capacitor, the HF generator

beingloaded by 50Ω.

- the dynamic power dissipated in the IC will in-

crease with frequency.To avoid the temperature

increasing, it is necessary todo very quick meas-

urementsor to use a low Rth (7oC/W) heatsinkin

forced convection configuration.Such conditions

are notpresentin aTV setsincethe drivingsignal

will be a video signal instead of a pure HF signal.

V.2.2 - Results

V.2.2.1 - Bandwidth

Thecurves Figures 23 and24 showthe frequency

responsesof one channel with 100Vpp and 50Vpp

output voltages.

The bandwidths are approximatively 8MHz at

100Vpp and up to 10MHzat 50Vpp.

V.2.2.2 - Crosstalk

The curves Figures 25, 26 and 27 show the cros-

stalkfor this application.Thecrosstalk isalmostthe

same for the six differentcombinationsof the three

channels. Theworst value is -24dB at 5MHz.

TEA5101A APPLICATION NOTE

14/21

825

REF

VDD

VREF

Rin

Gin

Bin

R11

820Ω

R21

820Ω

R31

820Ω

R32

39kΩ

R22

39kΩ

R12

39kΩ

R10

390Ω

R20

390Ω

R30

390Ω

P10

2.2kΩ

P20

2.2kΩ

P30

2.2kΩ

C10

470pF

C20

470pF

C30

470pF

B CATHODE

G CATHODE

R CATHODE

R13

1kΩ1/2W

R14 4.7kΩ

R23

1kΩ1/2W

R24 4.7kΩ

R33

1kΩ1/2W

R34 4.7kΩ

12V

CHASSIS

120kΩ

68kΩ

VCC

12V C2

10mF

250V

0.1µF

10µF

63V

0.1µF

250V

CHASSIS

GROUND

47ΩVDD

200V

CRT

GROUND

HEATER

1nF

630V

G2 10kΩ1/2W

I/O LOW LEVEL CONNECTOR C1

I/O HIGH LEVEL CONNECTORC2

CRT CONNECTOR

PIN CONNECTION

TEA5101A

5101A-19.EPS

Figure 17 : Application1 (45AX Tube, TDA3562A) - Electrical Diagram

TEA5101A APPLICATIONNOTE

15/21

825

REF

VDD

VREF

Rin

Gin

Bin

R11

820Ω

R21

820Ω

R31

820W

R32

39kΩ

R22

39kΩ

R12

39kΩ

R10

390Ω

R20

390Ω

R30

390Ω

P10

2.2kΩ

P20

2.2kΩ

P30

2.2kΩ

C10

1nF

C20

1nF

C30

1nF

B CATHODE

G CATHODE

R CATHODE

R13

1kΩ1/2W

R14 4.7kΩ

R23

1kΩ1/2W

R24 4.7kΩ

R33

1kΩ1/2W

R34 4.7kΩ

VCC

12V C2

10µF

250V

0.1µF

10µF

63V

0.1µF

250V

CHASSIS

GROUND

47ΩVDD

200V

CRT

GROUND

HEATER

2 x 22nF

630V

G2 10kΩ1/2W

TEA5101A

100kΩ

R15

100kΩ

R CUT-OFF

R25

100kΩ

G CUT-OFF

B CUT-OFF

R35

100kΩ

I/O LOW LEVEL CONNECTOR C1

I/O HIGH LEVEL CONNECTOR C2

CRT CONNECTOR

PIN CONNECTION

5101A-20.EPS

Figure 18 : Application2 (PIL24 Tube,TEA5031D) - ElectricalDiagram

TEA5101A APPLICATION NOTE

16/21

5101-21A.EPS/ 5101-21B.TIF

Figure 19 : TEA5101AEvaluationBoard Layout and Components View

COPPER SIDE

COMPONENT SIDE

TEA5101A APPLICATIONNOTE

17/21

825

REF

VDD

VREF

Rin

Gin

Bin

R11

1.5kΩ

R21

1.5kΩ

R31

1.5kΩ

R32

39kΩ

R22

39kΩ

R12

39kΩ

R10

390Ω

R20

390Ω

R30

390Ω

P10

2.2kΩ

P20

2.2kΩ

P30

2.2kΩ

C10

1nF

C20

1nF

C30

1nF

B CATHODE

G CATHODE

R CATHODE

R13

1kΩ1/2W

R14 4.7kΩ

R23

1kΩ1/2W

R24 4.7kΩ

R33

1kΩ1/2W

R34 4.7kΩ

VCC

12V C2

10µF

250V

0.1µF

10µF

63V

0.1µF

250V

CHASSIS

GROUND

47ΩVDD

200V

CRT

GROUND

HEATER

2 x 22nF

630V

G2 10kΩ1/2W

TEA5101A

100kΩ

R15

120kΩ

R CUT-OFF

R25

120kΩ

G CUT-OFF

B CUT-OFF

R35

120kΩ

I/O LOW LEVEL CONNECTOR C1

I/O HIGH LEVEL CONNECTOR C2

CRT CONNECTOR

PIN CONNECTION

5101A-22.EPS

Figure 20 : TEA5101AEvaluationBoard ElectricalSchematic Diagram

TEA5101A APPLICATION NOTE

18/21

50kΩ

R12

39kΩ

R11

1.5kΩ

1µFR10

390Ω

P10

2.2kΩ

VREF

VDD

C10

1nF R13

1kΩ1/2W

VDD

1MΩ

8P2

100nF

ACTIVE

PROBE

-40dB

CL2pF

PASSIVE

PROBE

-20dB

HP 3577

NETWORK

ANALYSER

AATTENUATOR

-20dB

4.7kΩ

5101A-23.EPS

Figure 21 : BandwidthMeasurement Configuration

ATTENUATOR

-20dB

50kΩ

R12

39kΩ

R11

1.5kΩ

1µFR10

390Ω

P10

2.2kΩ

VREF

VDD

C10

1nF R13

1kΩ1/2W

VDD

1MΩ

8P2

100nF

ACTIVE

PROBE

-40dB

CL 2pF

HP 3577

NETWORK

ANALYSER

4.7kΩ

ACTIVEPROBE

-40dB

R32

39kΩ

R21

1.5kΩ

1µFR20

390Ω

P20

2.2kΩ

VREF

VDD

C20

1nF R23

1kΩ1/2W

VDD

1MΩ

8P2

100nF

4.7kΩ

5101A-24.EPS

Figure 22 : Crosstalk Measurement Configuration

TEA5101A APPLICATIONNOTE

19/21

5101A-25.TIF

Figure 23 : Frequency Response of R Channel

(100VPP)

5101A-26.TIF

Figure 24 : FrequencyResponseof R Channel

(50VPP)

5101A-27.TIF

Figure 25 : Crosstalkbetween R Channel and G

and B Ones

5101A-28.TIF

Figure 26 : Crosstalk betweenGRChannel and R

and B Ones

5101A-29.TIF

Figure 27 : Crosstalk betweenB Channeland R

and G Ones

TEA5101A APPLICATION NOTE

20/21

Information furnishedis believed to be accurateand reliable. However, SGS-THOMSON Microelectronics assumes no responsibility

for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result

from its use. No licence is granted by implication or otherwise underany patent or patentrights of SGS-THOMSON Microelectronics.

Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all

information previously supplied. SGS-THOMSON Microelectronics products are not authorized for use as critical components in life

support devices or systems without express written approval of SGS-THOMSON Microelectronics.

Purchase of I2C Components of SGS-THOMSON Microelectronics, conveys a license under the Philips

I2C Patent. Rights to use these components in a I2C system, is granted provided that the system conformsto

the I2C Standard Specifications as defined by Philips.

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5101A-30.TIF

Figures 28A and 28B : TEA5101AR ChannelStep Response

5101A-31.TIF

V.2.2.3- Transitiontimes

The curves Figure 28 show respectively the R, G,

B rise and fall times of respectively49ns and 48ns

with a 100Vpp output voltage (between 50 and

150V).

The difference between rise times of the three

channelsis less than1ns.

Thedifferencebetweenfalltimesofthethreechan-

nels is less than 2ns.

The delay time at rising output is 48ns.

The delay time at falling output is 50ns.

The differencebetweenthedelaytimesis less than

2ns.

The slewrate is about 2000V/µs.

Theseresultsverifythehighperformanceavailable

from the TEA5101Avideo amplifier which make it

very suitable for high endapplications.

TEA5101A APPLICATIONNOTE

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