LM3524DM - NATIONAL SEMICONDUCTOR

LM2524D,LM3524D

LM2524D/LM3524D Regulating Pulse Width Modulator

Literature Number: SNVS766D

May 2, 2008

LM2524D/LM3524D

Regulating Pulse Width Modulator

General Description

The LM3524D family is an improved version of the industry

standard LM3524. It has improved specifications and addi-

tional features yet is pin for pin compatible with existing 3524

families. New features reduce the need for additional external

circuitry often required in the original version.

The LM3524D has a ±1% precision 5V reference. The current

carrying capability of the output drive transistors has been

raised to 200 mA while reducing VCEsat and increasing VCE

breakdown to 60V. The common mode voltage range of the

error-amp has been raised to 5.5V to eliminate the need for

a resistive divider from the 5V reference.

In the LM3524D the circuit bias line has been isolated from

the shut-down pin. This prevents the oscillator pulse ampli-

tude and frequency from being disturbed by shut-down. Also

at high frequencies (≃300 kHz) the max. duty cycle per output

has been improved to 44% compared to 35% max. duty cycle

in other 3524s.

In addition, the LM3524D can now be synchronized external-

ly, through pin 3. Also a latch has been added to insure one

pulse per period even in noisy environments. The LM3524D

includes double pulse suppression logic that insures when a

shut-down condition is removed the state of the T-flip-flop will

change only after the first clock pulse has arrived. This feature

prevents the same output from being pulsed twice in a row,

thus reducing the possibility of core saturation in push-pull

designs.

Features

■Fully interchangeable with standard LM3524 family

■±1% precision 5V reference with thermal shut-down

■Output current to 200 mA DC

■60V output capability

■Wide common mode input range for error-amp

■One pulse per period (noise suppression)

■Improved max. duty cycle at high frequencies

■Double pulse suppression

■Synchronize through pin 3

Connection Diagram

865002

Top View

Order Number LM2524DN or LM3524DN

See NS Package Number N16E

Order Number LM3524DM

See NS Package Number M16A

LM2524D/LM3524D Regulating Pulse Width Modulator

Block Diagram

865001

www.national.com 2

LM2524D/LM3524D

Absolute Maximum Ratings (Note 5)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage 40V

Collector Supply Voltage

(LM2524D) 55V

(LM3524D) 40V

Output Current DC (each) 200 mA

Oscillator Charging Current (Pin 7) 5 mA

Internal Power Dissipation 1W

Operating Junction Temperature

Range (Note 2)

LM2524D −40°C to +125°C

LM3524D 0°C to +125°C

Maximum Junction Temperature 150°

Storage Temperature Range −65°C to +150°C

Lead Temperature (Soldering 4 sec.)

M, N Pkg. 260°C

Electrical Characteristics

(Note 1)

LM2524D LM3524D

Symbol Parameter Conditions Tested Design Tested Design Units

Typ Limit Limit Typ Limit Limit

(Note 3) (Note 4) (Note 3) (Note 4)

REFERENCE SECTION

VREF Output Voltage 54.85 4.80 54.75 VMin

5.15 5.20 5.25 VMax

VRLine Line Regulation VIN = 8V to 40V 10 15 30 10 25 50 mVMax

VRLoad Load Regulation IL = 0 mA to 20 mA 10 15 25 10 25 50 mVMax

Ripple Rejection f = 120 Hz 66 66 dB

IOS Short Circuit VREF = 0 25 25 mA Min

Current 50 50

180 200 mA Max

NOOutput Noise 10 Hz ≤ f ≤ 10 kHz 40 100 40 100 μVrms Max

Long Term TA = 125°C 20 20 mV/kHr

Stability

OSCILLATOR SECTION

fOSC Max. Freq. RT = 1k, CT = 0.001 μF550 500 350 kHzMin

(Note 7)

fOSC Initial RT = 5.6k, CT = 0.01 μF 17.5 17.5 kHzMin

Accuracy (Note 7) 20 20

22.5 22.5 kHzMax

RT = 2.7k, CT = 0.01 μF 34 30 kHzMin

(Note 7) 38 38

42 46 kHzMax

ΔfOSC Freq. Change VIN = 8 to 40V 0.5 1 0.5 1.0 %Max

with VIN

ΔfOSC Freq. Change TA = −55°C to +125°C

with Temp. at 20 kHz RT = 5.6k, 5 5 %

CT = 0.01 μF

VOSC Output Amplitude RT = 5.6k, CT = 0.01 μF3 2.4 3 2.4 VMin

(Pin 3) (Note 8)

tPW Output Pulse RT = 5.6k, CT = 0.01 μF0.5 1.5 0.5 1.5 μsMax

Width (Pin 3)

Sawtooth Peak RT = 5.6k, CT = 0.01 μF3.4 3.6 3.8 3.8 VMax

Voltage

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LM2524D/LM3524D

LM2524D LM3524D

Symbol Parameter Conditions Tested Design Tested Design Units

Typ Limit Limit Typ Limit Limit

(Note 3) (Note 4) (Note 3) (Note 4)

Sawtooth Valley RT = 5.6k, CT = 0.01 μF1.1 0.8 0.6 0.6 VMin

Voltage

ERROR-AMP SECTION

VIO Input Offset VCM = 2.5V 2 8 10 2 10 mVMax

Voltage

IIB Input Bias VCM = 2.5V 1 8 10 1 10 μAMax

Current

IIO Input Offset VCM = 2.5V 0.5 1.0 10.5 1 μAMax

Current

ICOSI Compensation VIN(I) − VIN(NI) = 150 mV 65 65 μAMin

Current (Sink) 95 95

125 125 μAMax

ICOSO Compensation VIN(NI) − VIN(I) = 150 mV −125 −125 μAMin

Current (Source) −95 −95

−65 −65 μAMax

AVOL Open Loop Gain RL = ∞, VCM = 2.5 V 80 74 60 80 70 60 dBMin

VCMR Common Mode 1.5 1.4 1.5 VMin

Input Voltage

Range

5.5 5.4 5.5 VMax

CMRR Common Mode 90 80 90 80 dBMin

Rejection Ratio

GBW Unity Gain AVOL = 0 dB, VCM = 2.5V 3 2 MHz

Bandwidth

VOOutput Voltage RL = ∞ 0.5 0.5 VMin

Swing 5.5 5.5 VMax

PSRR Power Supply VIN = 8 to 40V 80 70 80 65 dbMin

Rejection Ratio

COMPARATOR SECTION

Minimum Duty Pin 9 = 0.8V, 0 0 0 0 %Max

Cycle [RT = 5.6k, CT = 0.01 μF]

Maximum Duty Pin 9 = 3.9V, 49 45 49 45 %Min

Cycle [RT = 5.6k, CT = 0.01 μF]

Maximum Duty Pin 9 = 3.9V, 44 35 44 35 %Min

Cycle [RT = 1k, CT = 0.001 μF]

VCOMPZ Input Threshold Zero Duty Cycle 1 1 V

(Pin 9)

VCOMPM Input Threshold Maximum Duty Cycle 3.5 3.5 V

(Pin 9)

IIB Input Bias −1 −1 μA

Current

CURRENT LIMIT SECTION

VSEN Sense Voltage V(Pin 2) − V(Pin 1) ≥ 180 180 mVMin

150 mV 200 200

220 220 mVMax

TC-Vsense Sense Voltage T.C. 0.2 0.2 mV/°C

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LM2524D/LM3524D

LM2524D LM3524D

Symbol Parameter Conditions Tested Design Tested Design Units

Typ Limit Limit Typ Limit Limit

(Note 3) (Note 4) (Note 3) (Note 4)

Common Mode −0.7 −0.7 VMin

Voltage Range V5 − V4 = 300 mV 1 1 VMax

SHUT DOWN SECTION

VSD High Input V(Pin 2) − V(Pin 1) ≥10.5 10.5 VMin

Voltage 150 mV 1.5 1.5 VMax

ISD High Input I(pin 10) 1 1 mA

Current

OUTPUT SECTION (EACH OUTPUT)

VCES Collector Emitter IC ≤ 100 μA 55 40 VMin

Voltage Breakdown

ICES Collector Leakage VCE = 60V

Current VCE = 55V 0.1 50 μAMax

VCE = 40V 0.1 50

VCESAT Saturation IE = 20 mA 0.2 0.5 0.2 0.7 VMax

Voltage IE = 200 mA 1.5 2.2 1.5 2.5

VEO Emitter Output IE = 50 mA 18 17 18 17 VMin

Voltage

tRRise Time VIN = 20V,

IE = −250 μA200 200 ns

RC = 2k

tFFall Time RC = 2k 100 100 ns

SUPPLY CHARACTERISTICS SECTION

VIN Input Voltage After Turn-on 8 8 VMin

Range 40 40 VMax

T Thermal Shutdown (Note 2) 160 160 °C

Temp.

IIN Stand By Current VIN = 40V (Note 6) 5 10 5 10 mA

Note 1: Unless otherwise stated, these specifications apply for TA = TJ = 25°C. Boldface numbers apply over the rated temperature range: LM2524D is −40° to

85°C and LM3524D is 0°C to 70°C. VIN = 20V and fOSC = 20 kHz.

Note 2: For operation at elevated temperatures, devices in the N package must be derated based on a thermal resistance of 86°C/W, junction to ambient. Devices

in the M package must be derated at 125°C/W, junction to ambient.

Note 3: Tested limits are guaranteed and 100% tested in production.

Note 4: Design limits are guaranteed (but not 100% production tested) over the indicated temperature and supply voltage range. These limits are not used to

calculate outgoing quality level.

Note 5: Absolute maximum ratings indicate limits beyond which damage to the device may occur. DC and AC electrical specifications do not apply when operating

the device beyond its rated operating conditions.

Note 6: Pins 1, 4, 7, 8, 11, and 14 are grounded; Pin 2 = 2V. All other inputs and outputs open.

Note 7: The value of a Ct capacitor can vary with frequency. Careful selection of this capacitor must be made for high frequency operation. Polystyrene was used

in this test. NPO ceramic or polypropylene can also be used.

Note 8: OSC amplitude is measured open circuit. Available current is limited to 1 mA so care must be exercised to limit capacitive loading of fast pulses.

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LM2524D/LM3524D

Typical Performance Characteristics

Switching Transistor

Peak Output Current

vs Temperature

865028

Maximum Average Power

Dissipation (N, M Packages)

865029

Maximum & Minimum

Duty Cycle Threshold

Voltage

865030

Output Transistor

Saturation Voltage

865031

Output Transistor Emitter

Voltage

865032

Reference Transistor

Peak Output Current

865033

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LM2524D/LM3524D

Standby Current

vs Voltage

865034

Standby Current

vs Temperature

865035

Current Limit Sense Voltage

865036

Test Circuit

865004

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LM2524D/LM3524D

Functional Description

INTERNAL VOLTAGE REGULATOR

The LM3524D has an on-chip 5V, 50 mA, short circuit pro-

tected voltage regulator. This voltage regulator provides a

supply for all internal circuitry of the device and can be used

as an external reference.

For input voltages of less than 8V the 5V output should be

shorted to pin 15, VIN, which disables the 5V regulator. With

these pins shorted the input voltage must be limited to a max-

imum of 6V. If input voltages of 6V–8V are to be used, a pre-

regulator, as shown in Figure 1, must be added.

865010

*Minimum CO of 10 μF required for stability.

FIGURE 1.

OSCILLATOR

The LM3524D provides a stable on-board oscillator. Its fre-

quency is set by an external resistor, RT and capacitor, CT. A

graph of RT, CT vs oscillator frequency is shown is Figure 2.

The oscillator's output provides the signals for triggering an

internal flip-flop, which directs the PWM information to the

outputs, and a blanking pulse to turn off both outputs during

transitions to ensure that cross conduction does not occur.

The width of the blanking pulse, or dead time, is controlled by

the value of CT, as shown in Figure 3. The recommended

values of RT are 1.8 kΩ to 100 kΩ, and for CT, 0.001 μF to 0.1

μF.

If two or more LM3524D's must be synchronized together, the

easiest method is to interconnect all pin 3 terminals, tie all pin

7's (together) to a single CT, and leave all pin 6's open except

one which is connected to a single RT. This method works well

unless the LM3524D's are more than 6≃ apart.

A second synchronization method is appropriate for any cir-

cuit layout. One LM3524D, designated as master, must have

its RTCT set for the correct period. The other slave LM3524D

(s) should each have an RTCT set for a 10% longer period. All

pin 3's must then be interconnected to allow the master to

properly reset the slave units.

The oscillator may be synchronized to an external clock

source by setting the internal free-running oscillator frequen-

cy 10% slower than the external clock and driving pin 3 with

a pulse train (approx. 3V) from the clock. Pulse width should

be greater than 50 ns to insure full synchronization.

865005

FIGURE 2.

865006

FIGURE 3.

ERROR AMPLIFIER

The error amplifier is a differential input, transconductance

amplifier. Its gain, nominally 86 dB, is set by either feedback

or output loading. This output loading can be done with either

purely resistive or a combination of resistive and reactive

components. A graph of the amplifier's gain vs output load

resistance is shown in Figure 4.

865007

FIGURE 4.

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LM2524D/LM3524D

The output of the amplifier, or input to the pulse width modu-

lator, can be overridden easily as its output impedance is very

high (ZO ≃ 5 MΩ). For this reason a DC voltage can be applied

to pin 9 which will override the error amplifier and force a par-

ticular duty cycle to the outputs. An example of this could be

a non-regulating motor speed control where a variable volt-

age was applied to pin 9 to control motor speed. A graph of

the output duty cycle vs the voltage on pin 9 is shown in Figure

The duty cycle is calculated as the percentage ratio of each

output's ON-time to the oscillator period. Paralleling the out-

puts doubles the observed duty cycle.

865008

FIGURE 5.

The amplifier's inputs have a common-mode input range of

1.5V–5.5V. The on board regulator is useful for biasing the

inputs to within this range.

CURRENT LIMITING

The function of the current limit amplifier is to override the

error amplifier's output and take control of the pulse width.

The output duty cycle drops to about 25% when a current limit

sense voltage of 200 mV is applied between the +CL and

−CLsense terminals. Increasing the sense voltage approxi-

mately 5% results in a 0% output duty cycle. Care should be

taken to ensure the −0.7V to +1.0V input common-mode

range is not exceeded.

In most applications, the current limit sense voltage is pro-

duced by a current through a sense resistor. The accuracy of

this measurement is limited by the accuracy of the sense re-

sistor, and by a small offset current, typically 100 μA, flowing

from +CL to −CL.

OUTPUT STAGES

The outputs of the LM3524D are NPN transistors, capable of

a maximum current of 200 mA. These transistors are driven

180° out of phase and have non-committed open collectors

and emitters as shown in Figure 6.

865009

FIGURE 6.

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LM2524D/LM3524D

Typical Applications

865011

FIGURE 7. Positive Regulator, Step-Up Basic Configuration (IIN(MAX) = 80 mA)

865012

FIGURE 8. Positive Regulator, Step-Up Boosted Current Configuration

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LM2524D/LM3524D

865013

FIGURE 9. Positive Regulator, Step-Down Basic Configuration (IIN(MAX) = 80 mA)

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LM2524D/LM3524D

865014

FIGURE 10. Positive Regulator, Step-Down Boosted Current Configuration

865015

FIGURE 11. Boosted Current Polarity Inverter

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LM2524D/LM3524D

BASIC SWITCHING REGULATOR THEORY

AND APPLICATIONS

The basic circuit of a step-down switching regulator circuit is

shown in Figure 12, along with a practical circuit design using

the LM3524D in Figure 15.

865016

FIGURE 12. Basic Step-Down Switching Regulator

The circuit works as follows: Q1 is used as a switch, which

has ON and OFF times controlled by the pulse width modu-

lator. When Q1 is ON, power is drawn from VIN and supplied

to the load through L1; VA is at approximately VIN, D1 is re-

verse biased, and Co is charging. When Q1 turns OFF the

inductor L1 will force VA negative to keep the current flowing

in it, D1 will start conducting and the load current will flow

through D1 and L1. The voltage at VAis smoothed by the L1,

Co filter giving a clean DC output. The current flowing through

L1 is equal to the nominal DC load current plus some ΔIL

which is due to the changing voltage across it. A good rule of

thumb is to set

ILP-P ≃ 40% × Io.

865017

FIGURE 13. Relation of Switch Timing to Inductor Current in Step-Down Regulator

Neglecting VSAT, VD, and settling ΔIL+ = ΔIL−;

where T = Total Period

The above shows the relation between VIN, Vo and duty cycle.

as Q1 only conducts during tON.

The efficiency, η, of the circuit is:

ηMAX will be further decreased due to switching losses in Q1.

For this reason Q1 should be selected to have the maximum

possible fT, which implies very fast rise and fall times.

CALCULATING INDUCTOR L1

Since ΔIL+ = ΔIL− = 0.4Io

Solving the above for L1

where: L1 is in Henrys

f is switching frequency in Hz

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LM2524D/LM3524D

Also, see LM1578 data sheet for graphical methods of induc-

tor selection.

CALCULATING OUTPUT FILTER CAPACITOR Co:

Figure 13 shows L1's current with respect to Q1's tON and

tOFF times (VA is at the collector of Q1). This curent must flow

to the load and Co. Co's current will then be the difference

between IL, and Io.

Ico = IL − Io

From Figure 13 it can be seen that current will be flowing into

Co for the second half of tON through the first half of tOFF, or a

time, tON/2 + tOFF/2. The current flowing for this time is ΔIL/4.

The resulting ΔVc or ΔVo is described by:

For best regulation, the inductor's current cannot be allowed

to fall to zero. Some minimum load current Io, and thus in-

ductor current, is required as shown below:

865019

FIGURE 14. Inductor Current Slope in Step-Down

Regulator

A complete step-down switching regulator schematic, using

the LM3524D, is illustrated in Figure 15. Transistors Q1 and

Q2 have been added to boost the output to 1A. The 5V reg-

ulator of the LM3524D has been divided in half to bias the

error amplifier's non-inverting input to within its common-

mode range. Since each output transistor is on for half the

period, actually 45%, they have been paralleled to allow

longer possible duty cycle, up to 90%. This makes a lower

possible input voltage. The output voltage is set by:

where VNI is the voltage at the error amplifier's non-inverting

input.

Resistor R3 sets the current limit to:

Figures 16, 17 and show a PC board layout and stuffing dia-

gram for the 5V, 1A regulator of Figure 15. The regulator's

performance is listed in Table 1.

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LM2524D/LM3524D

865020

*Mounted to Staver Heatsink No. V5-1.

Q1 = BD344

Q2 = 2N5023

L1 = >40 turns No. 22 wire on Ferroxcube No. K300502 Torroid core.

FIGURE 15. 5V, 1 Amp Step-Down Switching Regulator

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LM2524D/LM3524D

TABLE 1.

Parameter Conditions Typical

Characteristics

Output Voltage VIN = 10V, Io = 1A 5V

Switching Frequency VIN = 10V, Io = 1A 20 kHz

Short Circuit VIN = 10V 1.3A

Current Limit

Load Regulation VIN = 10V 3 mV

Io = 0.2 − 1A

Line Regulation ΔVIN = 10 − 20V, 6 mV

Io = 1A

Efficiency VIN = 10V, Io = 1A 80%

Output Ripple VIN = 10V, Io = 1A 10 mVp-p

865021

FIGURE 16. 5V, 1 Amp Switching Regulator, Foil Side

865022

FIGURE 17. Stuffing Diagram, Component Side

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LM2524D/LM3524D

THE STEP-UP SWITCHING REGULATOR

Figure 18 shows the basic circuit for a step-up switching reg-

ulator. In this circuit Q1 is used as a switch to alternately apply

VIN across inductor L1. During the time, tON, Q1 is ON and

energy is drawn from VIN and stored in L1; D1 is reverse bi-

ased and Io is supplied from the charge stored in Co. When

Q1 opens, tOFF, voltage V1 will rise positively to the point

where D1 turns ON. The output current is now supplied

through L1, D1 to the load and any charge lost from Co during

tON is replenished. Here also, as in the step-down regulator,

the current through L1 has a DC component plus some ΔIL.

ΔIL is again selected to be approximately 40% of IL. Figure

19 shows the inductor's current in relation to Q1's ON and

OFF times.

865023

FIGURE 18. Basic Step-Up Switching Regulator

865024

FIGURE 19. Relation of Switch Timing to Inductor Current in Step-Up Regulator

Since ΔIL+ = ΔIL−, VINtON = VotOFF − VINtOFF,

and neglecting VSAT and VD1

The above equation shows the relationship between VIN, Vo

and duty cycle.

In calculating input current IIN(DC), which equals the inductor's

DC current, assume first 100% efficiency:

for η = 100%, POUT = PIN

This equation shows that the input, or inductor, current is

larger than the output current by the factor (1 + tON/tOFF). Since

this factor is the same as the relation between Vo and VIN, IIN

(DC) can also be expressed as:

So far it is assumed η = 100%, where the actual efficiency or

ηMAX will be somewhat less due to the saturation voltage of

Q1 and forward on voltage of D1. The internal power loss due

to these voltages is the average IL current flowing, or IIN,

through either VSAT or VD1. For VSAT = VD1 = 1V this power

loss becomes IIN(DC) (1V). ηMAX is then:

This equation assumes only DC losses, however ηMAX is fur-

ther decreased because of the switching time of Q1 and D1.

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LM2524D/LM3524D

In calculating the output capacitor Co it can be seen that Co

supplies Io during tON. The voltage change on Co during this

time will be some ΔVc = ΔVo or the output ripple of the regu-

lator. Calculation of Co is:

where: Co is in farads, f is the switching frequency,

ΔVo is the p-p output ripple

Calculation of inductor L1 is as follows:

VIN is applied across L1

where: L1 is in henrys, f is the switching frequency in Hz

To apply the above theory, a complete step-up switching reg-

ulator is shown in Figure 20. Since VIN is 5V, VREF is tied to

VIN. The input voltage is divided by 2 to bias the error

amplifier's inverting input. The output voltage is:

The network D1, C1 forms a slow start circuit.

This holds the output of the error amplifier initially low thus

reducing the duty-cycle to a minimum. Without the slow start

circuit the inductor may saturate at turn-on because it has to

supply high peak currents to charge the output capacitor from

0V. It should also be noted that this circuit has no supply re-

jection. By adding a reference voltage at the non-inverting

input to the error amplifier, see Figure 21, the input voltage

variations are rejected.

The LM3524D can also be used in inductorless switching

regulators. Figure 22 shows a polarity inverter which if con-

nected to Figure 20 provides a −15V unregulated output.

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LM2524D/LM3524D

865025

L1 = > 25 turns No. 24 wire on Ferroxcube No. K300502 Toroid core.

FIGURE 20. 15V, 0.5A Step-Up Switching Regulator

865026

FIGURE 21. Replacing R3/R4 Divider in Figure 20 with Reference Circuit Improves Line Regulation

865027

FIGURE 22. Polarity Inverter Provides Auxiliary −15V Unregulated Output from Circuit of Figure 20

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LM2524D/LM3524D

Physical Dimensions inches (millimeters) unless otherwise noted

Molded Surface-Mount Package (M)

Order Number LM3524DM

NS Package Number M16A

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LM2524D/LM3524D

Molded Dual-In-Line Package (N)

Order Number LM2524DN or LM3524DN

NS Package Number N16E

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LM2524D/LM3524D

Notes

LM2524D/LM3524D Regulating Pulse Width Modulator

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