REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
Complete 14-Bit, 1.25 MSPS
Monolithic A/D Converter
AD9241
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700 World Wide Web Site: http://www.analog.com
Fax: 617/326-8703 © Analog Devices, Inc., 1997
FUNCTIONAL BLOCK DIAGRAM
VINA
CAPT
CAPB
SENSE OTR
BIT 1
(MSB)
BIT 14
(LSB)
VREF
DVSS
AVSS
AD9241
SHA
DIGITAL CORRECTION LOGIC
OUTPUT BUFFERS
VINB
1V
REFCOM
5
5
4
4
4
44
14
DVDDAVDD
CLK
MODE
SELECT
MDAC3
GAIN = 8
MDAC2
GAIN = 8
MDAC1
GAIN = 16
A/D
A/D
A/DA/D
DRVDD
DRVSS
CML
FEATURES
Monolithic 14-Bit, 1.25 MSPS A/D Converter
Low Power Dissipation: 60 mW
Single +5 V Supply
Integral Nonlinearity Error: 2.5 LSB
Differential Nonlinearity Error: 0.6 LSB
Input Referred Noise: 0.36 LSB
Complete: On-Chip Sample-and-Hold Amplifier and
Voltage Reference
Signal-to-Noise and Distortion Ratio: 78.0 dB
Spurious-Free Dynamic Range: 88.0 dB
Out-of-Range Indicator
Straight Binary Output Data
44-Pin MQFP
PRODUCT HIGHLIGHTS
The AD9241 offers a complete single-chip sampling 14-bit,
analog-to-digital conversion function in a 44-pin Metric Quad
Flatpack.
Low Power and Single Supply
The AD9241 consumes only 60 mW on a single +5 V power
supply.
Excellent DC Performance Over Temperature
The AD9241 provides no missing codes, and excellent tempera-
ture drift performance over the full operating temperature range.
Excellent AC Performance and Low Noise
The AD9241 provides nearly 13 ENOB performance and has an
input referred noise of 0.36 LSB rms.
Flexible Analog Input Range
The versatile onboard sample-and-hold (SHA) can be configured
for either single-ended or differential inputs of varying input spans.
Flexible Digital Outputs
The digital outputs can be configured to interface with +3 V and
+5 V CMOS logic families.
PRODUCT DESCRIPTION
The AD9241 is a 1.25 MSPS, single supply, 14-bit analog-to-
digital converter (ADC). It combines a low cost, high speed
CMOS process and a novel architecture to achieve the resolution
and speed of existing hybrid implementations at a fraction of the
power consumption and cost. It is a complete, monolithic ADC
with an on-chip, high performance, low noise sample-and-hold
amplifier and programmable voltage reference. An external refer-
ence can also be chosen to suit the dc accuracy and temperature
drift requirements of the application. The device uses a multistage
differential pipelined architecture with digital output error correc-
tion logic to guarantee no missing codes over the full operating
temperature range.
The input of the AD9241 is highly flexible, allowing for easy
interfacing to imaging, communications, medical, and data-
acquisition systems. A truly differential input structure allows
for both single-ended and differential input interfaces of varying
input spans. The sample-and-hold amplifier (SHA) is equally
suited for both multiplexed systems that switch full-scale voltage
levels in successive channels as well as sampling single-channel
inputs at frequencies up to and beyond the Nyquist rate. Also,
the AD9241 performs well in communication systems employ-
ing Direct-IF Down Conversion since the SHA in the differen-
tial input mode can achieve excellent dynamic performance well
beyond its specified Nyquist frequency of 0.625 MHz.
A single clock input is used to control all internal conversion
cycles. The digital output data is presented in straight binary
output format. An out-of-range (OTR) signal indicates an over-
flow condition which can be used with the most significant bit
to determine low or high overflow.
REV. 0
–2–
AD9241–SPECIFICATIONS
DC SPECIFICATIONS
Parameter AD9241 Units
RESOLUTION 14 Bits min
MAX CONVERSION RATE 1.25 MHz min
INPUT REFERRED NOISE
VREF = 1 V 0.9 LSB rms typ
VREF
= 2.5 V 0.36 LSB rms typ
ACCURACY
Integral Nonlinearity (INL) ±2.5 LSB typ
Differential Nonlinearity (DNL) ±0.6 LSB typ
±1.0 LSB max
INL
1
±2.5 LSB typ
DNL
1
±0.7 LSB typ
No Missing Codes 14 Bits Guaranteed
Zero Error (@ +25°C) 0.3 % FSR max
Gain Error (@ +25°C)
2
1.5 % FSR max
Gain Error (@ +25°C)
3
0.75 % FSR max
TEMPERATURE DRIFT
Zero Error 3.0 ppm/°C typ
Gain Error
2
20.0 ppm/°C typ
Gain Error
3
5.0 ppm/°C typ
POWER SUPPLY REJECTION 0.1 % FSR max
ANALOG INPUT
Input Span (with VREF = 1.0 V) 2 V p-p min
Input Span (with VREF = 2.5 V) 5 V p-p max
Input (VINA or VINB) Range 0 V min
AVDD V max
Input Capacitance 16 pF typ
INTERNAL VOLTAGE REFERENCE
Output Voltage (1 V Mode) 1 Volts typ
Output Voltage Tolerance (1 V Mode) ±14 mV max
Output Voltage (2.5 V Mode) 2.5 Volts typ
Output Voltage Tolerance (2.5 V Mode) ±35 mV max
Load Regulation
4
5.0 mV max
REFERENCE INPUT RESISTANCE 5 kΩ typ
POWER SUPPLIES
Supply Voltages
AVDD +5 V (±5% AVDD
Operating)
DVDD +5 V (±5% DVDD
Operating)
DRVDD +5 V (±5% DRVDD
Operating)
Supply Current
IAVDD 13.0 mA max (10 mA typ )
IDRVDD 1.0 mA max (1 mA typ )
IDVDD 3.0 mA max (2 mA typ )
POWER CONSUMPTION 65 mW typ
85 mW max
NOTES
1
VREF =1 V.
2
Including internal reference.
3
Excluding internal reference.
4
Load regulation with 1 mA load current (in addition to that required by the AD9241).
Specification subject to change without notice.
(AVDD = +5 V, DVDD = +5 V, DRVDD = +5 V, fSAMPLE = 1.25 MSPS, VREF = 2.5 V, VINB = 2.5 V,
TMIN to TMAX unless otherwise noted)
AC SPECIFICATIONS
Parameter AD9241 Units
SIGNAL-TO-NOISE AND DISTORTION RATIO (S/N+D)
f
INPUT
= 100 kHz 78.0 dB typ
f
INPUT
= 500 kHz 74.5 dB min
77.0 dB typ
EFFECTIVE NUMBER OF BITS (ENOB)
f
INPUT
= 100 kHz 12.7 Bits typ
f
INPUT
= 500 kHz 12.1 Bits min
12.5 Bits typ
SIGNAL-TO-NOISE RATIO (SNR)
f
INPUT
= 100 kHz 79.0 dB typ
f
INPUT
= 500 kHz 75.5 dB min
79.0 dB typ
TOTAL HARMONIC DISTORTION (THD)
f
INPUT
= 100 kHz –88.0 dB typ
f
INPUT
= 500 kHz –77.5 dB max
–88.0 dB typ
SPURIOUS FREE DYNAMIC RANGE
f
INPUT
= 100 kHz 88.0 dB typ
f
INPUT
= 500 kHz 86.0 dB typ
DYNAMIC PERFORMANCE
Full Power Bandwidth 25 MHz typ
Small Signal Bandwidth 25 MHz typ
Aperture Delay 1 ns typ
Aperture Jitter 4 ps rms typ
Acquisition to Full-Scale Step (0.0025%) 240 ns typ
Overvoltage Recovery Time 167 ns typ
Specifications subject to change without notice.
DIGITAL SPECIFICATIONS
Parameters Symbol AD9241 Units
LOGIC INPUTS
High Level Input Voltage V
IH
+3.5 V min
Low Level Input Voltage V
IL
+1.0 V max
High Level Input Current (V
IN
= DVDD) I
IH
±10 µA max
Low Level Input Current (V
IN
= 0 V) I
IL
±10 µA max
Input Capacitance C
IN
5 pF typ
LOGIC OUTPUTS (with DRVDD = 5 V)
High Level Output Voltage (I
OH
= 50 µA) V
OH
+4.5 V min
High Level Output Voltage (I
OH
= 0.5 mA) V
OH
+2.4 V min
Low Level Output Voltage (I
OL
= 1.6 mA) V
OL
+0.4 V max
Low Level Output Voltage (I
OL
= 50 µA) V
OL
+0.1 V max
Output Capacitance C
OUT
5 pF typ
LOGIC OUTPUTS (with DRVDD = 3 V)
High Level Output Voltage (I
OH
= 50 µA) V
OH
+2.4 V min
Low Level Output Voltage (I
OL
= 50 µA) V
OL
+0.7 V max
Specifications subject to change without notice.
AD9241
REV. 0 –3–
(AVDD = +5 V, DVDD = +5 V, DRVDD = +5 V, fSAMPLE = 1.25 MSPS, VREF = 2.5 V, AIN = –0.5 dBFS, AC Coupled/
Differential Input, TMIN to TMAX unless otherwise noted)
(AVDD = +5 V, DVDD = +5 V, TMIN to TMAX unless otherwise noted)
AD9241
REV. 0
–4–
ABSOLUTE MAXIMUM RATINGS*
With
Respect
Parameter to Min Max Units
AVDD AVSS –0.3 +6.5 V
DVDD DVSS –0.3 +6.5 V
AVSS DVSS –0.3 +0.3 V
AVDD DVDD –6.5 +6.5 V
DRVDD DRVSS –0.3 +6.5 V
DRVSS AVSS –0.3 +0.3 V
REFCOM AVSS –0.3 +0.3 V
CLK DVSS –0.3 DVDD
+ 0.3 V
Digital Outputs DRVSS –0.3 DRVDD
+ 0.3 V
VINA, VINB AVSS –0.3 AVDD
+ 0.3 V
VREF AVSS –0.3 AVDD
+ 0.3 V
SENSE AVSS –0.3 AVDD
+ 0.3 V
CAPB, CAPT AVSS –0.3 AVDD
+ 0.3 V
Junction Temperature +150 °C
Storage Temperature –65 +150 °C
Lead Temperature
(10 sec) +300 °C
*Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum ratings
for extended periods may effect device reliability.
SWITCHING SPECIFICATIONS
Parameters Symbol AD9241 Units
Clock Period
1
t
C
800 ns min
CLOCK Pulse Width High t
CH
360 ns min
CLOCK Pulse Width Low t
CL
360 ns min
Output Delay t
OD
8 ns min
13 ns typ
19 ns max
Pipeline Delay (Latency) 3 Clock Cycles
NOTES
1
The clock period may be extended to 1 ms without degradation in specified performance @ +25 °C.
Specifications subject to change without notice.
(TMIN to TMAX with AVDD = +5 V, DVDD = +5 V, DRVDD = +5 V, CL = 20 pF)
WARNING!
ESD SENSITIVE DEVICE
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD9241 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
t
CL
t
CH
t
C
t
OD
DATA 1
DATA
OUTPUT
INPUT
CLOCK
ANALOG
INPUT
S1 S2
S3 S4
Figure 1. Timing Diagram
THERMAL CHARACTERISTICS
Thermal Resistance
44-Pin MQFP
θ
JA
= 53.2°C/W
θ
JC
= 19°C/W
ORDERING GUIDE
Temperature Package Package
Model Range Description Option*
AD9241AS –40
o
C to +85
o
C 44-Pin MQFP S-44
AD9241EB Evaluation Board
*S = Metric Quad Flatpack.
PIN CONNECTION
3
4
5
6
7
1
2
10
11
8
9
40 39 3841424344 36 35 3437
29
30
31
32
33
27
28
25
26
23
24
PIN 1
IDENTIFIER
TOP VIEW
(Not to Scale)
12 13 14 15 16 17 18 19 20 21 22
BIT 5
BIT 4
BIT 3
BIT 8
BIT 13
BIT 12
BIT 11
BIT 9
BIT 7
BIT 6
NC
NC
NC
CML
NC
CAPT
NC
REFCOM
VREF
SENSE
NC
AVSS
AVDD
NC
AD9241
DVSS
AVSS
DVDD
AVDD
DRVSS
DRVDD
CLK
NC = NO CONNECT
NC
NC
NC
(LSB) BIT 14
NC
OTR
BIT 1 (MSB)
BIT 2
BIT 10
CAPB
NC
VINB
VINA
AD9241
REV. 0 –5–
overvoltage (50% greater than full-scale range), measured from
the time the overvoltage signal reenters the converter’s range.
TEMPERATURE DRIFT
The temperature drift for zero error and gain error specifies the
maximum change from the initial (+25°C) value to the value at
T
MIN
or T
MAX
.
POWER SUPPLY REJECTION
The specification shows the maximum change in full scale,
from the value with the supply at the minimum limit to the
value with the supply at its maximum limit.
APERTURE JITTER
Aperture jitter is the variation in aperture delay for successive
samples and is manifested as noise on the input to the A/D.
APERTURE DELAY
Aperture delay is a measure of the sample-and-hold amplifier
(SHA) performance and is measured from the rising edge of the
clock input to when the input signal is held for conversion.
SIGNAL-TO-NOISE AND DISTORTION (S/N+D, SINAD)
RATIO
S/N+D is the ratio of the rms value of the measured input sig-
nal to the rms sum of all other spectral components below the
Nyquist frequency, including harmonics but excluding dc.
The value for S/N+D is expressed in decibels.
EFFECTIVE NUMBER OF BITS (ENOB)
For a sine wave, SINAD can be expressed in terms of the num-
ber of bits. Using the following formula,
N = (SINAD – 1.76)/6.02
it is possible to get a measure of performance expressed as N,
the effective number of bits.
Thus, the effective number of bits for a device for sine wave
inputs at a given input frequency can be calculated directly
from its measured SINAD.
TOTAL HARMONIC DISTORTION (THD)
THD is the ratio of the rms sum of the first six harmonic
components to the rms value of the measured input signal; this
is expressed as a percentage or in decibels.
SIGNAL-TO-NOISE RATIO (SNR)
SNR is the ratio of the rms value of the measured input signal
to the rms sum of all other spectral components below the
Nyquist frequency, excluding the first six harmonics and dc.
The value for SNR is expressed in decibels.
SPURIOUS FREE DYNAMIC RANGE (SFDR)
SFDR is the difference in dB between the rms amplitude of the
input signal and the peak spurious signal.
TWO-TONE SFDR
The ratio of the rms value of either input tone to the rms value
of the peak spurious component. The peak spurious component
may or may not be an IMD product. It may be reported in dBc
(i.e., degrades as signal level is lowered) or in dBFS (always
related back to converter full scale).
PIN FUNCTION DESCRIPTIONS
Pin
Number Name Description
1 DVSS Digital Ground
2, 29 AVSS Analog Ground
3 DVDD +5 V Digital Supply
4, 28 AVDD +5 V Analog Supply
5 DRVSS Digital Output Driver Ground
6 DRVDD Digital Output Driver Supply
7 CLK Clock Input Pin
8–10 NC No Connect
11 BIT 14 Least Significant Data Bit (LSB)
12–23 BIT 13–BIT 2 Data Output Bits
24 BIT 1 Most Significant Data Bit (MSB)
25 OTR Out of Range
26, 27, 30 NC No Connect
31 SENSE Reference Select
32 VREF Reference I/O
33 REFCOM Reference Common
34, 35, 38 NC No Connect
40, 43, 44
36 CAPB Noise Reduction Pin
37 CAPT Noise Reduction Pin
39 CML Common-Mode Level (Midsupply)
41 VINA Analog Input Pin (+)
42 VINB Analog Input Pin (–)
DEFINITIONS OF SPECIFICATION
INTEGRAL NONLINEARITY (INL)
INL refers to the deviation of each individual code from a line
drawn from “negative full scale” through “positive full scale.”
The point used as “negative full scale” occurs 1/2 LSB before
the first code transition. “Positive full scale” is defined as a level
1 1/2 LSB beyond the last code transition. The deviation is
measured from the middle of each particular code to the true
straight line.
DIFFERENTIAL NONLINEARITY (DNL, NO MISSING
CODES)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed
no missing codes to 14-bit resolution indicates that all 16384
codes, respectively, must be present over all operating ranges.
ZERO ERROR
The major carry transition should occur for an analog value
1/2 LSB below VINA = VINB. Zero error is defined as the
deviation of the actual transition from that point.
GAIN ERROR
The first code transition should occur at an analog value
1/2 LSB above negative full scale. The last transition should
occur at an analog value 1 1/2 LSB below the nominal full
scale. Gain error is the deviation of the actual difference
between first and last code transitions, and the ideal differ-
ence between first and last code transitions.
OVERVOLTAGE RECOVERY TIME
Overvoltage recovery time is defined as that amount of time
required for the ADC to achieve a specified accuracy after an
AD9241
REV. 0
–6–
Typical Differential AC Characterization Curves/Plots
(AVDD = +5 V, DVDD = +5 V, DRVDD = +5 V, fSAMPLE =
1.25 MSPS, TA = +258C, Differential Input)
INPUT FREQUENCY – MHz
SINAD – dB
80
75
40
0.01 0.1 10.0
1.0
70
65
45
60
55
50
–0.5dBFS
–6.0dBFS
–20dBFS
Figure 2. SINAD vs. Input Frequency
(Input Span = 5 V, V
CM
= 2.5 V)
INPUT FREQUENCY – MHz
SINAD – dB
80
75
40
0.01 0.1 10.01.0
70
65
45
60
55
50
–0.5dBFS
–6.0dBFS
–20.0dBFS
Figure 5. SINAD vs. Input Frequency
(Input Span = 2 V, V
CM
= 2.5 V)
SAMPLE RATE – MSPS
–40
–1000.1 1.0 10.0
–50
–80
–60
–70
THD – dB
–90
5V SPAN
2V SPAN
Figure 8. THD vs. Sample Rate
(f
IN
= 0.3 MHz, A
IN
= –0.5 dBFS,
V
CM
= 2.5 V)
–0.5dBFS
–6.0dBFS
–20.0dBFS
INPUT FREQUENCY – MHz
THD – dB
–40
–50
–100
0.01 0.1 10.0
1.0
–70
–80
–90
–60
Figure 3. THD vs. Input Frequency
(Input Span = 5 V, V
CM
= 2.5 V)
–0.5dBFS
–6.0dBFS
–20.0dBFS
INPUT FREQUENCY – MHz
THD – dB
–40
–50
–100
0.01 0.1 10.01.0
–70
–80
–90
–60
Figure 6. THD vs. Input Frequency
(Input Span = 2 V, V
CM
= 2.5 V)
AIN – dBFS
SFDR – dBc AND dBFS
110
40
–45 –39 –33 –27 –21 –15
100
90
80
60
50
70
dBc - 5V
dBFS - 5V
dBc - 2V
dBFS - 5V
–9 –3 0
Figure 9. Single Tone SFDR
(f
IN
= 0.6 MHz, V
CM
= 2.5 V)
8
FUND
5326794
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
0FREQUENCY – kHz
AMPLITUDE – dB
–110
–120
–130
–140
–150
–160
–170 100 200 300 400 500 600
Figure 4. Typical FFT, f
IN
>
500 kHz
(Input Span = 5 V, V
CM
= 2.5 V)
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
0FREQUENCY – kHz
AMPLITUDE – dB
–110
–120
–130
–140
–150
–160
–170 100 200 300 400 500 600
FUND
6
7
32
589
4
Figure 7. Typical FFT, f
IN
>
500 kHz
(Input Span = 2 V, V
CM
= 2.5 V)
INPUT POWER LEVEL (f1 = f2) – dBFS
WORST CASE SPURIOUS – dBc AND dBFS
110
60
–40 –35 0
–30 –25 –20 –15 –10 –5
105
90
85
75
65
100
95
80
70
5V SPAN - dBFS
5V SPAN - dBc
2V SPAN - dBFS
2V SPAN - dBc
Figure 10. Dual Tone SFDR
(f
1
= 0.5 MHz, f
2
= 0.6 MHz,
V
CM
= 2.5 V)
AD9241
REV. 0 –7–
Other Characterization Curves/Plots
(AVDD = +5 V, DVDD = +5 V, DRVDD = +5 V, fSAMPLE = 1.25 MSPS, TA = +258C,
Single-Ended Input)
CODE
3.0
–1.5
0
INL – LSB
2.5
0.5
0.0
–0.5
–1.0
2.0
1.5
–2.0
–2.5
–3.0 16383
1.0
Figure 11. Typical INL
(Input Span = 5 V)
INPUT FREQUENCY – MHz
SINAD – dB
90
85
50
0.01 0.1 10.0
80
75
55
70
65
60
–0.5dBFS
–6.0dBFS
–20.0dBFS
1.0
Figure 14. SINAD vs. Input Frequency
(Input Span = 2 V, V
CM
= 2.5 V)
INPUT FREQUENCY – MHz
SINAD – dB
90
85
50
0.01 0.1 10.0
80
75
55
70
65
60
–0.5dBFS
–6dBFS
–20dBFS
45
40 1.0
Figure 17. SINAD vs. Input Frequency
(Input Span = 5 V, V
CM
= 2.5 V)
CODE
1.0
0
0.8
0.2
0.0
0.6
0.4
–0.2
–0.4
–0.6
–0.8
–1.0
DNL – LSB
16383
100%
Figure 12. Typical DNL
(Input Span = 5 V)
INPUT FREQUENCY – MHz
THD – dB
–40
–50
–100
0.01 0.1 10.0
–70
–80
–90
–60
–0.5dBFS
–6dBFS
–20dBFS
1.0
Figure 15. THD vs. Input Frequency
(Input Span = 2 V, V
CM
= 2.5 V)
INPUT FREQUENCY – MHz
THD – dB
–40
–50
–100
0.01 0.1 1.0
–70
–80
–90
–60
–0.5dBFS
–20dBFS
10.0
–6dBFS
Figure 18. THD vs. Input Frequency
(Input Span = 5 V, V
CM
= 2.5 V)
N–1
12,901,627
1,137,700 1,146,291
N N+1
HITS
CODE
Figure 13. “Grounded-Input”
Histogram (Input Span = 5 V)
FREQUENCY – MHz
CMR – dB
90
0.01 0.1 10.0
1.0
40
50
60
70
80
100
Figure 16. CMR vs. Input Frequency
(Input Span = 2 V, V
CM
= 2.5 V)
TEMPERATURE – 8C
V
REF
ERROR – V
0.01
–0.004
–0.01
–60 –40 140
–20 0 20 40 60 80 100 120
0.008
–0.002
–0.006
–0.008
0.002
0
0.006
0.004
Figure 19. Typical Voltage Reference
Error vs. Temperature
AD9241
REV. 0
–8–
converter. Specifically, the input to the A/D core is the difference
of the voltages applied at the VINA and VINB input pins.
Therefore, the equation,
VCORE = VINA – VINB (1)
defines the output of the differential input stage and provides the
input to the A/D core.
The voltage, V
CORE
, must satisfy the condition,
–VREF
≤
V
CORE
≤
VREF (2)
where VREF is the voltage at the VREF pin.
While an infinite combination of VINA and VINB inputs exist
to satisfy Equation 2, an additional limitation is placed on the
inputs by the power supply voltages of the AD9241. The power
supplies bound the valid operating range for VINA and VINB.
The condition,
AVSS – 0.3 V < VINA < AVDD + 0.3 V (3)
AVSS – 0.3 V < VINB < AVDD + 0.3 V
where AVSS is nominally 0 V and AVDD is nominally +5 V,
defines this requirement. Thus, the range of valid inputs for
VINA and VINB is any combination that satisfies both Equa-
tions 2 and 3.
For additional information showing the relationship between
VINA, VINB, VREF and the digital output of the AD9241, see
Table IV.
Refer to Table I and Table II for a summary of the various
analog input and reference configurations.
ANALOG INPUT OPERATION
Figure 21 shows the equivalent analog input of the AD9241,
which consists of a differential sample-and-hold amplifier
(SHA). The differential input structure of the SHA is highly
flexible, allowing the devices to be easily configured for either a
differential or single-ended input. The dc offset, or common-
mode voltage, of the input(s) can be set to accommodate either
single-supply or dual supply systems. Also, note that the analog
inputs, VINA and VINB, are interchangeable, with the exception
that reversing the inputs to the VINA and VINB pins results in a
polarity inversion.
C
S
Q
S1
Q
H1
VINA
VINB
C
S
Q
S1
C
PIN
–
C
PAR
C
PIN
+
C
PAR
Q
S2
C
H
Q
S2
C
H
Figure 21. Simplified Input Circuit
INTRODUCTION
The AD9241 uses a four-stage pipeline architecture with a
wideband input sample-and-hold amplifier (SHA) implemented
on a cost-effective CMOS process. Each stage of the pipeline,
excluding the last, consists of a low resolution flash A/D con-
nected to a switched capacitor DAC and interstage residue
amplifier (MDAC). The residue amplifier amplifies the differ-
ence between the reconstructed DAC output and the flash input
for the next stage in the pipeline. One bit of redundancy is used
in each of the stages to facilitate digital correction of flash er-
rors. The last stage simply consists of a flash A/D.
The pipeline architecture allows a greater throughput rate at the
expense of pipeline delay or latency. This means that while the
converter is capable of capturing a new input sample every clock
cycle, it actually takes three clock cycles for the conversion to be
fully processed and appear at the output. This latency is not a
concern in most applications. The digital output, together with
the out-of-range indicator (OTR), is latched into an output
buffer to drive the output pins. The output drivers can be con-
figured to interface with +5 V or +3.3 V logic families.
The AD9241 uses both edges of the clock in its internal timing
circuitry (see Figure 1 and specification page for exact timing
requirements). The A/D samples the analog input on the rising
edge of the clock input. During the clock low time (between the
falling edge and rising edge of the clock), the input SHA is in
the sample mode; during the clock high time it is in the hold
mode. System disturbances just prior to the rising edge of the
clock and/or excessive clock jitter may cause the input SHA to
acquire the wrong value and should be minimized.
ANALOG INPUT AND REFERENCE OVERVIEW
Figure 20, a simplified model of the AD9241, highlights the rela-
tionship between the analog inputs, VINA, VINB, and the
reference voltage, VREF. Like the voltage applied to the top of
the resistor ladder in a flash A/D converter, the value VREF defines
the maximum input voltage to the A/D core. The minimum input
voltage to the A/D core is automatically defined to be –VREF.
V
CORE
VINA
VINB
+VREF
–VREF
A/D
CORE 14
AD9241
Figure 20. Equivalent Functional Input Circuit
The addition of a differential input structure gives the user an
additional level of flexibility that is not possible with traditional
flash converters. The input stage allows the user to easily config-
ure the inputs for either single-ended operation or differential
operation. The A/D’s input structure allows the dc offset of the
input signal to be varied independently of the input span of the
AD9241
REV. 0 –9–
The input SHA of the AD9241 is optimized to meet the perfor-
mance requirements for some of the most demanding commu-
nication, imaging and data acquisition applications, while
maintaining low power dissipation. Figure 22 is a graph of the
full-power bandwidth of the AD9241, typically 40 MHz. Note
that the small signal bandwidth is the same as the full-power
bandwidth. The settling time response to a full-scale stepped
input is shown in Figure 23 and is typically less than 80 ns to
0.0025%. The low input referred noise of 0.36 LSB’s rms is
displayed via a grounded histogram and is shown in Figure 13.
FREQUENCY – MHz
AMPLITUDE – dB
2
0
–12
0.01 1.0 10.0 100
–2
–4
–6
–8
–10
0.1
Figure 22. Full-Power Bandwidth
SETTLING TIME – ns
CODE
16000
12000
006010 20 30 40 50
8000
4000
70 80
Figure 23. Settling Time
The SHA’s optimum distortion performance for a differential or
single-ended input is achieved under the following two condi-
tions: (1) the common-mode voltage is centered around mid-
supply (i.e., AVDD/2 or approximately 2.5 V) and (2) the input
signal voltage span of the SHA is set at its lowest (i.e., 2 V input
span). This is due to the sampling switches, Q
S1
, being CMOS
switches whose R
ON
resistance is very low but has some signal
dependency causing frequency-dependent ac distortion while
the SHA is in the track mode. The R
ON
resistance of a CMOS
switch is typically lowest at its midsupply, but increases sym-
metrically as the input signal approaches either AVDD or
AVSS. A lower input signal voltage span centered at midsupply
reduces the degree of R
ON
modulation.
Figure 24 compares the AD9241’s THD vs. frequency perfor-
mance for a 2 V input span with a common-mode voltage of
1 V and 2.5 V. Note the difference in the amount of degrada-
tion in THD performance as the input frequency increases.
Similarly, note how the THD performance at lower frequencies
becomes less sensitive to the common-mode voltage. As the
input frequency approaches dc, the distortion will be domi-
nated by static nonlinearities such as INL and DNL. It is
important to note that these dc static nonlinearities are inde-
pendent of any R
ON
modulation.
FREQUENCY – MHz
THD – dB
–40
–45
–85
–50
–75
–55
–60
–65
–70
–80
0.01 0.1 10.0
VCM = 1V
VCM = 2.5V
1.0
Figure 24. THD vs. Frequency for V
CM
= 2.5 V and 1.0 V
(A
IN
= –0.5 dB, Input Span = 2.0 V p-p)
Due to the high degree of symmetry within the SHA topology, a
significant improvement in distortion performance for differen-
tial input signals with frequencies up to and beyond Nyquist can
be realized. This inherent symmetry provides excellent cancella-
tion of both common-mode distortion and noise. In addition,
the required input signal voltage span is reduced by a factor of
two, which further reduces the degree of R
ON
modulation and
its effects on distortion.
The optimum noise and dc linearity performance for either
differential or single-ended inputs is achieved with the largest
input signal voltage span (i.e., 5 V input span) and matched
input impedance for VINA and VINB. Note that only a slight
degradation in dc linearity performance exists between the 2 V and
5 V input span as specified in AD9241 DC SPECIFICATIONS.
Referring to Figure 21, the differential SHA is implemented
using a switched-capacitor topology. Hence, its input imped-
ance and its subsequent effects on the input drive source should
be understood to maximize the converter’s performance. The
combination of the pin capacitance, C
PIN
, parasitic capacitance
C
PAR,
and the sampling capacitance, C
S
, is typically less than
16 pF. When the SHA goes into track mode, the input source
must charge or discharge the voltage stored on C
S
to the new
input voltage. This action of charging and discharging C
S
,
which
is approximately 4 pF, averaged over a period of time and for a
given sampling frequency, F
S
, makes the input impedance ap-
pear to have a benign resistive component (i.e., 83 kΩ at F
S
=
1.25 MSPS). However, if this action is analyzed within a sam-
pling period (i.e., T = <1/F
S
), the input impedance is dynamic
due to the instantaneous requirement of charging and discharg-
ing C
S
. A series resistor inserted between the input drive source
and the SHA input, as shown in Figure 25, provides effective
isolation.
AD9241
REV. 0
–10–
10µF
VINA
VINB
SENSE
AD9241
0.1µF
R
S
*
V
CC
V
EE
R
S
*
VREF
REFCOM
*OPTIONAL SERIES RESISTOR
Figure 25. Series Resistor Isolates Switched-Capacitor
SHA Input from Op Amp. Matching Resistors Improve
SNR Performance
The optimum size of this resistor is dependent on several fac-
tors, including the AD9241 sampling rate, the selected op amp
and the particular application. In most applications, a 30 Ω to
50 Ω resistor is sufficient. Some applications may require a
larger resistor value to reduce the noise bandwidth or possibly
limit the fault current in an overvoltage condition. Other appli-
cations may require a larger resistor value as part of an antialiasing
filter. In any case, since the THD performance is dependent on
the series resistance and the above mentioned factors, optimiz-
ing this resistor value for a given application is encouraged.
A slight improvement in SNR performance and dc offset
performance is achieved by matching the input resistance con-
nected to VINA and VINB. The degree of improvement is depen-
dent on the resistor value and the sampling rate. For series
resistor values greater than 100 Ω, the use of a matching
resistor is encouraged.
The noise or small-signal bandwidth of the AD9241 is the same
as its full-power bandwidth. For noise sensitive applications, the
excessive bandwidth may be detrimental and the addition of a
series resistor and/or shunt capacitor can help limit the wide-
band noise at the A/D’s input by forming a low-pass filter. Note,
however, that the combination of this series resistance with the
equivalent input capacitance of the AD9241 should be evalu-
ated for those time-domain applications that are sensitive to the
input signal’s absolute settling time. In applications where har-
monic distortion is not a primary concern, the series resistance
may be selected in combination with the SHA’s nominal 16 pF
of input capacitance to set the filter’s 3 dB cutoff frequency.
A better method of reducing the noise bandwidth, while possi-
bly establishing a real pole for an antialiasing filter, is to add
some additional shunt capacitance between the input (i.e.,
VINA and/or VINB) and analog ground. Since this additional
shunt capacitance combines with the equivalent input capaci-
tance of the AD9241, a lower series resistance can be selected
to establish the filter’s cutoff frequency while not degrading the
distortion performance of the device. The shunt capacitance
also acts as a charge reservoir, sinking or sourcing the additional
charge required by the hold capacitor, C
H
, further reducing
current transients seen at the op amp’s output.
The effect of this increased capacitive load on the op amp driv-
ing the AD9241 should be evaluated. To optimize performance
when noise is the primary consideration, increase the shunt
capacitance as much as the transient response of the input signal
will allow. Increasing the capacitance too much may adversely
affect the op amp’s settling time, frequency response and distor-
tion performance.
Table I. Analog Input Configuration Summary
Input Input Input Range (V) Figure
Connection Coupling Span (V) VINA
1
VINB
1
# Comments
Single-Ended DC 2 0 to 2 1 32, 33 Best for stepped input response applications, suboptimum
THD and noise performance, requires ±5 V op amp.
2 × VREF 0 to VREF 32, 33 Same as above but with improved noise performance due to
2 × VREF increase in dynamic range. Headroom/settling time require-
ments of ±5 V op amp should be evaluated.
5 0 to 5 2.5 32, 33 Optimum noise performance, excellent THD performance. Requires
op amp with VCC > +5 V due to insufficient headroom @ 5 V.
2 × VREF 2.5 – VREF 2.5 39 Optimum THD performance with VREF = 1, noise performance
to improves while THD performance degrades as VREF increases
2.5 + VREF to 2.5 V. Single supply operation (i.e., +5 V) for many op amps.
Single-Ended AC 2 or 0 to 1 or 1 or VREF 34 Suboptimum ac performance due to input common-mode
2 × VREF 0 to 2 × VREF level not biased at optimum midsupply level (i.e., 2.5 V).
5 0 to 5 2.5 34 Optimum noise performance, excellent THD performance.
2 × VREF 2.5 – VREF 2.5 35 Flexible input range, Optimum THD performance with
to VREF = 1. Noise performance improves while THD performance
2.5 + VREF degrades as VREF increases to 2.5 V.
Differential AC or 2 2 to 3 3 to 2 29–31 Optimum full-scale THD and SFDR performance well beyond
DC the A/Ds Nyquist frequency.
2 × VREF 2.5 – VREF/2 2.5 + VREF/2 29–31 Same as 2 V to 3 V input range with the exception that full-scale
to to THD and SFDR performance can be traded off for better noise
2.5 + VREF/2 2.5 – VREF/2 performance.
5 1.25 to 3.75 3.75 to 1.25 29–31 Widest dynamic range (i.e., ENOBs) due to Optimum Noise
performance.
1
VINA and VINB can be interchanged if signal inversion is required.
AD9241
REV. 0 –11–
REFERENCE OPERATION
The AD9241 contains an onboard bandgap reference that pro-
vides a pin-strappable option to generate either a 1 V or 2.5 V
output. With the addition of two external resistors, the user can
generate reference voltages other than 1 V and 2.5 V. Another
alternative is to use an external reference for designs requiring
enhanced accuracy and/or drift performance. See Table II for a
summary of the pin-strapping options for the AD9241 reference
configurations.
Figure 26 shows a simplified model of the internal voltage refer-
ence of the AD9241. A pin-strappable reference amplifier buff-
ers a 1 V fixed reference. The output from the reference amplifier,
A1, appears on the VREF pin. The voltage on the VREF pin
determines the full-scale input span of the A/D. This input span
equals,
Full-Scale Input Span = 2
×
VREF
A2
5kΩ
5kΩ
5kΩ
5kΩ
LOGIC
DISABLE
A2
7.5kΩ
LOGIC
A1
5kΩ
DISABLE
A1
1V
TO
A/D
AD9241
CAPT
CAPB
VREF
SENSE
REFCOM
Figure 26. Equivalent Reference Circuit
The voltage appearing at the VREF pin, and the state of the
internal reference amplifier, A1, are determined by the voltage
appearing at the SENSE pin. The logic circuitry contains two
comparators that monitor the voltage at the SENSE pin. The
comparator with the lowest set point (approximately 0.3 V)
controls the position of the switch within the feedback path of
A1. If the SENSE pin is tied to REFCOM, the switch is
connected to the internal resistor network thus providing a
VREF of 2.5 V. If the SENSE pin is tied to the VREF pin via a
short or resistor, the switch is connected to the SENSE pin. A
short will provide a VREF of 1.0 V while an external resistor
network will provide an alternative VREF between 1.0 V and
2.5 V. The second comparator controls internal circuitry that
will disable the reference amplifier if the SENSE pin is tied to
AVDD. Disabling the reference amplifier allows the VREF pin
to be driven by an external voltage reference.
The actual reference voltages used by the internal circuitry of
the AD9241 appear on the CAPT and CAPB pins. For proper
operation when using the internal or an external reference, it is
necessary to add a capacitor network to decouple these pins.
Figure 27 shows the recommended decoupling network. This
capacitive network performs the following three functions: (1) in
conjunction with the reference amplifier, A2, it provides a low
source impedance over a large frequency range to drive the A/D
internal circuitry, (2) it provides the necessary compensation for
A2, and (3) it bandlimits the noise contribution from the refer-
ence. The turn-on time of the reference voltage appearing be-
tween CAPT and CAPB is approximately 15 ms and should be
evaluated in any power-down mode of operation.
0.1µF 10µF
0.1µF
0.1µF
CAPT
CAPB
AD9241
Figure 27. Recommended CAPT/CAPB Decoupling Network
The A/D’s input span may be varied dynamically by changing
the differential reference voltage appearing across CAPT and
CAPB symmetrically around 2.5 V (i.e., midsupply). To change
the reference at speeds beyond the capabilities of A2, it will be
necessary to drive CAPT and CAPB with two high speed, low
noise amplifiers. In this case, both internal amplifiers (i.e., A1
and A2) must be disabled by connecting SENSE to AVDD and
VREF to REFCOM, and the capacitive decoupling network
removed. The external voltages applied to CAPT and CAPB
must be 2.5 V + Input Span/4 and 2.5 V – Input Span/4, respec-
tively where the input span can be varied between 2 V and 5 V.
Note that those samples within the pipeline A/D during any
reference transition will be corrupted and should be discarded.
Table II. Reference Configuration Summary
Reference Input Span (VINA–VINB)
Operating Mode (V p-p) Required VREF (V) Connect To
INTERNAL 2 1 SENSE VREF
INTERNAL 5 2.5 SENSE REFCOM
INTERNAL 2 ≤ SPAN ≤ 5 AND 1 ≤ VREF ≤ 2.5 AND R1 VREF AND SENSE
SPAN = 2 × VREF VREF = (1 + R1/R2) R2 SENSE AND REFCOM
EXTERNAL 2 ≤ SPAN ≤ 51 ≤ VREF ≤ 2.5 SENSE AVDD
(NONDYNAMIC) VREF EXT. REF.
EXTERNAL 2 ≤ SPAN ≤ 5 CAPT and CAPB SENSE AVDD
(DYNAMIC) Externally Driven VREF REFCOM
EXT. REF. 1 CAPT
EXT. REF. 2 CAPB
AD9241
REV. 0
–12–
DRIVING THE ANALOG INPUTS
INTRODUCTION
The AD9241 has a highly flexible input structure allowing it to
interface with single-ended or differential input interface cir-
cuitry. The applications shown in sections Driving the Analog
Inputs and Reference Configurations, along with the informa-
tion presented in Input and Reference Overview of this data
sheet, give examples of both single-ended and differential opera-
tion. Refer to Tables I and II for a list of the different possible
input and reference configurations and their associated figures
in the data sheet.
The optimum mode of operation, analog input range and asso-
ciated interface circuitry, will be determined by the particular
applications performance requirements as well as power supply
options. For example, a dc coupled single-ended input may be
appropriate for many data acquisition and imaging applications.
Also, many communication applications requiring a dc coupled
input for proper demodulation can take advantage of the excel-
lent single-ended distortion performance of the AD9241. The
input span should be configured so the system’s performance
objectives and the headroom requirements of the driving op amp
are simultaneously met.
Alternatively, the differential mode of operation provides the
best THD and SFDR performance over a wide frequency range.
A transformer coupled differential input should be considered
for the most demanding spectral-based applications that allow
ac coupling (e.g., Direct IF to Digital Conversion). The dc
coupled differential mode of operation also provides an enhance-
ment in distortion and noise performance at higher input spans.
Furthermore, it allows the AD9241 to be configured for a 5 V
span using op amps specified for +5 V or ±5 V operation.
Single-ended operation requires that VINA be ac or dc coupled
to the input signal source, while VINB of the AD9241 be biased
to the appropriate voltage corresponding to a midscale code
transition. Note that signal inversion may be easily accom-
plished by transposing VINA and VINB.
Differential operation requires that VINA and VINB be simulta-
neously driven with two equal signals that are in and out of
phase versions of the input signal. Differential operation of the
AD9241 offers the following benefits: (1) Signal swings are
smaller and therefore linearity requirements placed on the input
signal source may be easier to achieve, (2) Signal swings are
smaller and therefore may allow the use of op amps that may
otherwise have been constrained by headroom limitations,
(3) Differential operation minimizes even-order harmonic prod-
ucts and (4) Differential operation offers noise immunity based
on the device’s common-mode rejection as shown in Figure 16.
As is typical of most CMOS devices, exceeding the supply limits
will turn on internal parasitic diodes resulting in transient cur-
rents within the device. Figure 28 shows a simple means of clamp-
ing a dc coupled input with the addition of two series resistors and
two diodes. Note that a larger series resistor could be used to limit
the fault current through D1 and D2, but should be evaluated
since it can cause a degradation in overall performance.
AVDD
RS1
30Ω
VCC
VEE
D2
1N4148
D1
1N4148
RS2
20ΩAD9243
Figure 28. Simple Clamping Circuit
DIFFERENTIAL MODE OF OPERATION
Since not all applications have a signal preconditioned for differ-
ential operation, there is often a need to perform a single-ended-
to-differential conversion. A single-ended-to-differential conversion
can be realized with an RF transformer or a dual op amp differ-
ential driver. The optimum method depends on whether the
application requires the input signal to be ac or dc coupled to
AD9241.
AC Coupling via an RF Transformer
In applications that do not need to be dc coupled, an RF trans-
former with a center tap is the best method of generating differ-
ential inputs for the AD9241. It provides all the benefits of
operating the A/D in the differential mode without contributing
additional noise or distortion. An RF transformer has the added
benefit of providing electrical isolation between the signal source
and the A/D.
Figure 29 shows the schematic of the suggested transformer
circuit. The circuit uses a Mini-Circuits RF transformer, model
#T4-6T, which has an impedance ratio of four (turns ratio of
2). The schematic assumes that the signal source has a 50 Ω
source impedance. The 1:4 impedance ratio requires the 200 Ω
secondary termination for optimum power transfer and VSWR.
The centertap of the transformer provides a convenient means
of level-shifting the input signal to a desired common-mode
voltage. Optimum performance can be realized when the centertap
is tied to CML of the AD9241 which is the common-mode bias
level of the internal SHA.
VINA
CML
VINB
AD9241
0.1µF
200Ω
MINI-CIRCUITS
T4-6T
50Ω
Figure 29. Transformer Coupled Input
Transformers with other turns ratios may also be selected to
optimize the performance of a given application. For example, a
given input signal source or amplifier may realize an improve-
ment in distortion performance at reduced output power levels
and signal swings. Hence, selecting a transformer with a higher
impedance ratio (i.e., Mini-Circuits T16-6T with a 1:16 imped-
ance ratio) effectively “steps up” the signal level, further reduc-
ing the driving requirements of the signal source.
AD9241
REV. 0 –13–
DC Coupling with Op Amps
Applications that require dc coupling can also benefit by driv-
ing the AD9241 differentially. Since the signal swing require-
ments of each input is reduced by a factor of two in the differential
mode, the AD9241 can be configured for a 5 V input span in a
+5 V or ±5 V system. This allows various high performance op
amps specified for +5 V and ±5 V operation to be configured in
various differential driver topologies. The optimum op amp
driver topology depends on whether the common-mode voltage
of the single-ended-input signal requires level-shifting.
Figure 30 shows a cross-coupled differential driver circuit best
suited for systems in which the common-mode signal of the
input is already biased to approximately midsupply (i.e., 2.5 V).
The common-mode voltage of the differential output is set by
the voltage applied to the “+” input of A2. The closed loop
gain of this symmetrical driver can easily be set by R
IN
and R
F
.
For more insight into the operation of this cross-coupled driver,
please refer to the AD8042 data sheet.
VINA
VINB
CML
AD9241
1kΩ
0.1µF
1kΩ
1kΩ
1kΩ
1kΩ
1kΩ
V
IN
V
CML
–VIN
AVDD/2
V
CML
+VIN
AD8042
AD8042
33Ω
33Ω
C
F
*
*OPTIONAL NOISE/BAND LIMITING CAPACITOR
R
F
R
IN
A1
A2
Figure 30. Cross-Coupled Differential Driver
The driver circuit shown in Figure 31 is best suited for systems
in which the bipolar input signal is referenced to AGND and
requires proper level shifting. This driver circuit provides the
ability to level-shift the input signal to within the common-
mode range of the AD9241. The two op amps are configured as
matched difference amplifiers, with the input signal applied to
opposing inputs to provide the differential output. The common-
mode offset voltage is applied to the noninverting resistor net-
work that provides the proper level-shifting. The circuit also
employs optional diodes and pull-up resistors that may help
improve the op amps’ distortion performance by reducing their
headroom requirements. Rail-to-rail output amplifiers such as
the AD8042 have sufficient headroom and do not require these
optional components.
VINA
VINB
CML
AD9241
390Ω
390Ω
V
IN
V
CML
–VIN
V
CML
+VIN
AVDD
390Ω
390Ω220.2Ω
390Ω
AVDD
390Ω
220.2Ω
390Ω
AD8047
AD8047
2.5kΩ
33Ω
100Ω
0.1µF 1µF
0.1µF
OP113
33Ω
390Ω
0.1µF
0.1µF
Figure 31. Differential Driver with Level-Shifting
SINGLE-ENDED MODE OF OPERATION
The AD9241 can be configured for single-ended operation
using dc or ac coupling. In either case, the input of the A/D
must be driven from an operational amplifier that will not de-
grade the A/D’s performance. Because the A/D operates from a
single supply, it will be necessary to level-shift ground-based
bipolar signals to comply with its input requirements. Both dc
and ac coupling provide this necessary function, but each
method results in different interface issues that may influence
the system design and performance.
DC COUPLING AND INTERFACE ISSUES
Many applications require the analog input signal to be dc
coupled to the AD9241. An operational amplifier can be con-
figured to rescale and level-shift the input signal to make it
compatible with the selected input range of the A/D. The input
range to the A/D should be selected on the basis of system
performance objectives as well as the analog power supply
availability since this will place certain constraints on the op
amp selection.
Many of the new high performance op amps are specified for
only ±5 V operation and have limited input/output swing capa-
bilities. Hence, the selected input range of the AD9241 should
be sensitive to the headroom requirements of the particular op
amp to prevent clipping of the signal. Also, since the output of
a dual supply amplifier can swing below –0.3 V, clamping its
output should be considered in some applications.
In some applications, it may be advantageous to use an op amp
specified for single supply +5 V operation since it will inher-
ently limit its output swing to within the power supply rails.
Rail-to-rail output amplifiers such as the AD8041 allow the
AD9241 to be configured with larger input spans, which im-
proves the noise performance.
AD9241
REV. 0
–14–
If the application requires the largest single-ended input range
(i.e., 0 V to 5 V) of the AD9241, the op amp will require larger
supplies to drive it. Various high speed amplifiers in the Op
Amp Selection Guide of this data sheet can be selected to
accommodate a wide range of supply options. Once again, clamp-
ing the output of the amplifier should be considered for these
applications. Alternatively, a single-ended-to-differential op amp
driver circuit using the AD8042 could be used to achieve the
5 V input span while operating from a single +5 V supply, as
discussed in the previous section.
Two dc coupled op amp circuits using a noninverting and inverting
topology are discussed below. Although not shown, the nonin-
verting and inverting topologies can easily be configured as part
of an antialiasing filter by using a Sallen-Key or Multiple-Feed-
back topology, respectively. An additional R-C network can be
inserted between the op amp’s output and the AD9241 input to
provide a real pole.
Simple Op Amp Buffer
In the simplest case, the input signal to the AD9241 will already
be biased at levels in accordance with the selected input range.
It is merely a matter of providing an adequately low source imped-
ance for the VINA and VINB analog input pins of the A/D. Figure
32 shows the recommended configuration for a single-ended drive
using an op amp. In this case, the op amp is shown in a nonin-
verting unity gain configuration driving the VINA pin. The
internal reference drives the VINB pin. Note that the addition of
a small series resistor of 30 Ω to 50 Ω connected to VINA and
VINB will be beneficial in nearly all cases. Refer to section
Analog Input Operation for a discussion on resistor selection.
Figure 32 shows the proper connection for a 0 V to 5 V input
range. Alternative single ended input ranges of 0 V to 2 × VREF
can also be realized with the proper configuration of VREF
(refer to the section Using the Internal Reference).
10µF
VINA
VINB
SENSE
AD9241
0.1µF
R
S
+V
–V
R
S
VREF
5V
0V U1
2.5V
Figure 32. Single-Ended AD9241 Op Amp Drive Circuit
Op Amp with DC Level Shifting
Figure 33 shows a dc-coupled level shifting circuit employing an
op amp, A1, to sum the input signal with the desired dc offset.
Configuring the op amp in the inverting mode with the given
resistor values results in an ac signal gain of –1. If the signal
inversion is undesirable, interchange the VINA and VINB con-
nections to reestablish the original signal polarity. The dc volt-
age at VREF sets the common-mode voltage of the AD9241. For
example, when VREF = 2.5 V, the output level from the op amp
will also be centered around 2.5 V. The use of ratio matched,
thin-film resistor networks will minimize gain and offset errors.
Also, an optional pull-up resistor, R
P
, may be used to reduce the
output load on VREF to ±1 mA.
0V
DC
+VREF
–VREF VINA
VINB
AD9241
0.1µF
500Ω*
0.1µF
500Ω*
71
2
345
A1
6
NC
NC
+V
CC
500Ω*
R
S
VREF
500Ω*
R
S
R
P
**
AVDD
*OPTIONAL RESISTOR NETWORK-OHMTEK ORNA500D
**OPTIONAL PULL-UP RESISTOR WHEN USING INTERNAL REFERENCE
Figure 33. Single-Ended Input With DC-Coupled Level Shift
AC COUPLING AND INTERFACE ISSUES
For applications where ac coupling is appropriate, the op amp’s
output can easily be level-shifted to the common-mode voltage,
V
CM
, of the AD9241 via a coupling capacitor. This has the
advantage of allowing the op amps common-mode level to be
symmetrically biased to its midsupply level (i.e., (V
CC
+ V
EE
)/2).
Op amps that operate symmetrically with respect to their power
supplies typically provide the best ac performance as well as the
greatest input/output span. Hence, various high speed/performance
amplifiers that are restricted to +5 V/–5 V operation and/or
specified for +5 V single-supply operation can easily be config-
ured for the 5 V or 2 V input span of the AD9241, respectively.
The best ac distortion performance is achieved when the A/D is
configured for a 2 V input span and common-mode voltage of
2.5 V. Note that differential transformer coupling, another form of
ac coupling, should be considered for optimum ac performance.
Simple AC Interface
Figure 34 shows a typical example of an ac-coupled, single-
ended configuration. The bias voltage shifts the bipolar, ground-
referenced input signal to approximately VREF. The value for
C1 and C2
will depend on the size of the resistor, R. The ca-
pacitors, C1 and C2, are typically a 0.1 µF ceramic and 10 µF
tantalum capacitor in parallel to achieve a low cutoff frequency
while maintaining a low impedance over a wide frequency range.
The combination of the capacitor and the resistor form a high-
pass filter with a high-pass –3 dB frequency determined by the
equation,
f
–3 dB
= 1/(2 × π × R × (C1 + C2))
The low impedance VREF voltage source both biases the VINB
input and provides the bias voltage for the VINA input. Figure
34 shows the VREF configured for 2.5 V. Thus the input range
C2
VINA
VINB
SENSE
AD9241
C1
R
+5V
–5V R
S
VREF
+VREF
0V
–VREF
V
IN
C2
C1
R
S
Figure 34. AC-Coupled Input
AD9241
REV. 0 –15–
of the A/D is 0 V to 5 V. Other input ranges could be selected
by changing VREF, but the A/D’s distortion performance will
degrade slightly as the input common-mode voltage deviates
from its optimum level of 2.5 V.
Alternative AC Interface
Figure 35 shows a flexible ac coupled circuit that can be config-
ured for different input spans. Since the common-mode voltage
of VINA and VINB are biased to midsupply independent of
VREF, VREF can be pin-strapped or reconfigured to achieve
input spans between 2 V and 5 V p-p. The AD9241’s CMRR,
along with the symmetrical coupling R-C networks, will reject
both power supply variations and noise. The resistors, R, estab-
lish the common-mode voltage. They may have a high value (e.g.,
5 kΩ) to minimize power consumption and establish a low cutoff
frequency. The capacitors, C1
and C2, are typically a 0.1 µF
ceramic and 10 µF tantalum capacitor in parallel to achieve a
low cutoff frequency while maintaining a low impedance over a
wide frequency range. R
S
isolates the buffer amplifier from the
A/D input. The optimum performance is achieved when VINA
and VINB are driven via symmetrical networks. The high-pass
f
–3 dB
point can be approximated by the equation,
f
–3 dB
= 1/(2 × π × R/2 × (C1 + C2))
C2
VINA
VINB
AD9241
C1
R
+5V
–5V
R
S
V
IN
C1
C2
RR
S
+5V
R
R
+5V
Figure 35. AC-Coupled Input-Flexible Input Span, V
CM
= 2.5 V
OP AMP SELECTION GUIDE
Op amp selection for the AD9241 is highly dependent on a
particular application. In general, the performance requirements
of any given application can be characterized by either time
domain or frequency domain parameters. In either case, one
should carefully select an op amp that preserves the performance
of the A/D. This task becomes challenging when one considers
the high performance capabilities of the AD9241, coupled with
other external system level requirements such as power con-
sumption and cost.
The ability to select the optimal op amp may be further compli-
cated by limited power supply availability and/or limited accept-
able supplies for a desired op amp. Newer, high performance op
amps typically have input and output range limitations in accor-
dance with their lower supply voltages. As a result, some op
amps will be more appropriate in systems where ac-coupling is
allowable. When dc-coupling is required, op amps without
headroom constraints, such as rail-to-rail op amps or those
where larger supplies can be used, should be considered. The
following section describes some op amps currently available
from Analog Devices. The system designer is always encouraged
to contact the factory or local sales office to be updated on
Analog Devices’ latest amplifier product offerings. Highlights of
the areas where the op amps excel, and where they may limit the
performance of the AD9241, are also included.
AD812: Dual, 145 MHz Unity GBW, Single-Supply Cur-
rent Feedback, +5 V to ±15 V Supplies
Best Applications: Differential and/or Low Imped-
ance Input Drivers
Limits: THD above 1 MHz
AD8011: f
–3 dB
= 300 MHz, +5 V or ±5 V Supplies, Current
Feedback
Best Applications: Single-Supply, AC/DC-Coupled,
Good AC Specs, Low Noise, Low Power (5 mW)
Limits: THD above 5 MHz, Usable Input/Output
Range
AD8013: Triple, f
–3 dB
= 230 MHz, +5 V or ±5 V supplies,
Current Feedback, Disable Function
Best Applications: 3:1 Multiplexer, Good AC Specs
Limits: THD above 5 MHz, Input Range
AD9631: 220 MHz Unity GBW, 16 ns Settling to 0.01%,
±5 V Supplies
Best Applications: Best AC Specs, Low Noise,
AC-Coupled
Limits: Usable Input/Output Range, Power
Consumption
AD8047: 130 MHz Unity GBW, 30 ns Settling to 0.01%,
±5 V Supplies
Best Applications: Good AC Specs, Low Noise,
AC-Coupled
Limits: THD > 5 MHz, Usable Input Range
AD8041: Rail-to-Rail, 160 MHz Unity GBW, 55 ns Settling
to 0.01%, +5 V Supply, 26 mW
Best Applications: Low Power, Single-Supply Sys-
tems, DC-Coupled, Large Input Range
Limits: Noise with 2 V Input Range
AD8042: Dual AD8041
Best Applications: Differential and/or Low Imped-
ance Input Drivers
Limits: Noise with 2 V Input Range
REFERENCE CONFIGURATIONS
The figures associated with this section on internal and external
reference operation do not show recommended matching series
resistors for VINA and VINB for the purpose of simplicity.
Please refer to section Driving the Analog Inputs, Introduction, for
a discussion of this topic. Also, the figures do not show the
decoupling network associated with the CAPT and CAPB pins.
Please refer to the Reference Operation section for a discussion of
the internal reference circuitry and the recommended decoupling
network shown in Figure 27.
USING THE INTERNAL REFERENCE
Single-Ended Input with 0 to 2 3 VREF Range
Figure 36 shows how to connect the AD9241 for a 0 V to 2 V or
0 V to 5 V input range via pin strapping the SENSE pin. An inter-
mediate input range of 0 to 2 × VREF can be established using
the resistor programmable configuration in Figure 38 and con-
necting VREF to VINB.
In either case, both the common-mode voltage and input span
are directly dependent on the value of VREF. More specifically,
the common-mode voltage is equal to VREF while the input
span is equal to 2 × VREF. Thus, the valid input range extends
from 0 to 2 × VREF. When VINA is ≤ 0 V, the digital output
will be 0000 Hex; when VINA is ≥ 2 × VREF, the digital output
will be 3FFF Hex.
AD9241
REV. 0
–16–
Shorting the VREF pin directly to the SENSE pin places the
internal reference amplifier in unity-gain mode and the resultant
VREF output is 1 V. Therefore, the valid input range is 0 V to
2 V. However, shorting the SENSE pin directly to the REFCOM
pin configures the internal reference amplifier for a gain of 2.5
and the resultant VREF output is 2.5 V. Thus, the valid input
range becomes 0 V to 5 V. The VREF pin should be bypassed
to the REFCOM pin with a 10 µF tantalum capacitor in parallel
with a low-inductance 0.1 µF ceramic capacitor.
10µF
VINA
VREF
AD9241
0.1µF VINB
2xVREF
0V
SHORT FOR 0 TO 2V
INPUT SPAN
SENSE
SHORT FOR 0 TO 5V
INPUT SPAN
REFCOM
Figure 36. Internal Reference (2 V p-p Input Span,
V
CM
= 1 V, or 5 V p-p Input Span, V
CM
= 2.5 V)
Single-Ended or Differential Input, V
CM
= 2.5 V
Figure 37 shows the single-ended configuration that gives the
best SINAD performance. To optimize dynamic specifications,
center the common-mode voltage of the analog input at
approximately 2.5 V by connecting VINB to VREF, a low-
impedance 2.5 V source. As described above, shorting the
SENSE pin directly to the REFCOM pin results in a 2.5 V
reference voltage and a 5 V p-p input span. The valid range
for input signals is 0 V to 5 V. The VREF pin should be by-
passed to the REFCOM pin with a 10 µF tantalum capacitor in
parallel with a low inductance 0.1 µF ceramic capacitor.
This reference configuration could also be used for a differential
input wherein VINA and VINB are driven via a transformer as
shown in Figure 29. In this case, the common-mode voltage,
V
CM
, is set at midsupply by connecting the transformers center
tap to CML of the AD9241. VREF can be configured for 1 V or
2.5 V by connecting SENSE to either VREF or REFCOM
respectively. Note that the valid input range for each of the
differential inputs is one half of the single-ended input and thus
becomes V
CM
– VREF/2 to V
CM
+ VREF/2.
0.1µF
10µF
VINA
VINB
VREF
SENSE
REFCOM
AD9241
5V
0V
2.5V
Figure 37. Internal Reference—5 V p-p Input Span,
V
CM
= 2.5 V
Resistor Programmable Reference
Figure 38 shows an example of how to generate a reference
voltage other than 1 V or 2.5 V with the addition of two ex-
ternal resistors and a bypass capacitor. Use the equation,
VREF = 1 V × (1 + R1/R2),
to determine appropriate values for R1 and R2. These resistors
should be in the 2 kΩ to 100 kΩ range. For the example shown,
R1 equals 2.5 kΩ and R2 equals 5 kΩ. From the equation above,
the resultant reference voltage on the VREF pin is 1.5 V. This
sets the input span to be 3 V p-p. To assure stability, place a
0.1 µF ceramic capacitor in parallel with R1.
The common-mode voltage can be set to VREF by connecting
VINB to VREF to provide an input span of 0 to 2 × VREF.
Alternatively, the common-mode voltage can be set to 2.5 V
by connecting VINB to a low impedance 2.5 V source. For
the example shown, the valid input signal range for VINA is
1 V to 4 V since VINB is set to an external, low impedance
2.5 V source. The VREF pin should be bypassed to the REFCOM
pin with a 10 µF tantalum capacitor in parallel with a low induc-
tance 0.1 µF ceramic capacitor.
1.5V
C1
0.1µF
10µF
VINA
VINB
VREF
SENSE
REFCOM
AD9241
4V
1V
2.5V
R1
2.5kΩ
R2
5kΩ
0.1µF
Figure 38. Resistor Programmable Reference (3 V p-p
Input Span, V
CM
= 2.5 V)
USING AN EXTERNAL REFERENCE
Using an external reference may enhance the dc performance
of the AD9241 by improving drift and accuracy. Figures 39
through 41 show examples of how to use an external reference
with the A/D. Table III is a list of suitable voltage references
from Analog Devices. To use an external reference, the user
must disable the internal reference amplifier and drive the VREF
pin. Connecting the SENSE pin to AVDD disables the inter-
nal reference amplifier.
Table III. Suitable Voltage References
Initial Operating
Output Drift Accuracy Current
Voltage (ppm/8C) % (max) (mA)
Internal 1.00 26 1.4 N/A
AD589 1.235 10–100 1.2–2.8 50
AD1580 1.225 50–100 0.08–0.8 50
REF191 2.048 5–25 0.1–0.5 45
Internal 2.50 26 1.4 N/A
REF192 2.50 5–25 0.08–0.4 45
REF43 2.50 10–25 0.06–0.1 600
AD780 2.50 3–7 0.04–0.2 1000
The AD9241 contains an internal reference buffer, A2 (see
Figure 26), that simplifies the drive requirements of an external
reference. The external reference must be able to drive a ≈5 kΩ
(±20%) load. Note that the bandwidth of the reference buffer is
AD9241
REV. 0 –17–
deliberately left small to minimize the reference noise contribu-
tion. As a result, it is not possible to change the reference volt-
age rapidly in this mode without removing the CAPT/CAPB
Decoupling Network and driving these pins directly.
Variable Input Span with V
CM
= 2.5 V
Figure 39 shows an example of the AD9241 configured for an
input span of 2 × VREF centered at 2.5 V. An external 2.5 V
reference drives the VINB pin thus setting the common-mode
voltage at 2.5 V. The input span can be independently set by a
voltage divider consisting of R1 and R2, which generates the
VREF signal. A1 buffers this resistor network and drives VREF.
Choose this op amp based on accuracy requirements. It is
essential that a minimum of a 10 µF capacitor in parallel with a
0.1 µF low inductance ceramic capacitor decouple the reference
output to ground.
2.5V+VREF
2.5V–VREF
2.5V
+5V
0.1µF
22µF
VINA
VINB
VREF
SENSE
AD9241
+5V
R2
0.1µF
A1
R1
0.1µF
2.5V
REF
Figure 39. External Reference, V
CM
= 2.5 V (2.5 V on VINB,
Resistor Divider to Make VREF)
Single-Ended Input with 0 to 2 3 VREF Range
Figure 40 shows an example of an external reference driving
both VINB and VREF. In this case, both the common mode
voltage and input span are directly dependent on the value of
VREF. More specifically, the common-mode voltage is equal to
VREF while the input span is equal to 2 × VREF. Thus, the
valid input range extends from 0 to 2 × VREF. For example, if
the REF191, a 2.048 external reference, was selected, the valid
input range extends from 0 V to 4.096 V. In this case, 1 LSB of
the AD9241 corresponds to 0.250 mV. It is essential that a
minimum of a 10 µF capacitor in parallel with a 0.1 µF low induc-
tance ceramic capacitor decouple the reference output to ground.
2xREF
0V
+5V
10µF
VINA
VINB
VREF
SENSE
AD9241
+5V
0.1µF
VREF
0.1µF
0.1µF
Figure 40. Input Range = 0 V to 2
×
VREF
Low Cost/Power Reference
The external reference circuit shown in Figure 41 uses a low
cost 1.225 V external reference (e.g., AD580 or AD1580) along
with an op amp and transistor. The 2N2222 transistor acts in
conjunction with 1/2 of an OP282 to provide a very low imped-
ance drive for VINB. The selected op amp need not be a high
speed op amp and may be selected based on cost, power and
accuracy.
3.75V
1.25V
+5V
10µF
VINA
VINB
VREF
SENSE
AD9241
+5V
0.1µF
316Ω
1kΩ
0.1µF
1/2
OP282
10µF 0.1µF
7.5kΩ
AD1580
1kΩ
1kΩ820Ω
+5V
2N2222
1.225V
Figure 41. External Reference Using the AD1580 and Low
Impedance Buffer
DIGITAL INPUTS AND OUTPUTS
Digital Outputs
The AD9241 output data is presented in positive true straight
binary for all input ranges. Table IV indicates the output data
formats for various input ranges, regardless of the selected input
range. A twos-complement output data format can be created
by inverting the MSB.
Table IV. Output Data Format
Input (V) Condition (V) Digital Output OTR
VINA –VINB < – VREF 00 0000 0000 0000 1
VINA –VINB = – VREF 00 0000 0000 0000 0
VINA –VINB = 0 10 0000 0000 0000 0
VINA –VINB = + VREF – 1 LSB 11 1111 1111 1111 0
VINA –VINB ≥ + VREF 11 1111 1111 1111 1
111111 1111 1111
111111 1111 1111
111111 1111 1110
OTR
–FS +FS
–FS+1/2 LSB
+FS –1/2 LSB–FS –1/2 LSB
+FS –1 1/2 LSB
000000 0000 0001
000000 0000 0000
000000 0000 0000
1
0
0
0
0
1
OTR DATA OUTPUTS
Figure 42. Output Data Format
Out Of Range (OTR)
An out-of-range condition exists when the analog input voltage
is beyond the input range of the converter. OTR is a digital
output that is updated along with the data output corresponding
to the particular sampled analog input voltage. Hence, OTR has
the same pipeline delay (latency) as the digital data. It is LOW
when the analog input voltage is within the analog input range.
AD9241
REV. 0
–18–
It is HIGH when the analog input voltage exceeds the input
range as shown in Figure 42. OTR will remain HIGH until the
analog input returns within the input range and another conver-
sion is completed. By logical ANDing OTR with the MSB and
its complement, overrange high or underrange low conditions
can be detected. Table V is a truth table for the over/underrange
circuit in Figure 43, which uses NAND gates. Systems requiring
programmable gain conditioning of the AD9241 input signal
can immediately detect an out-of-range condition, thus elimi-
nating gain selection iterations. Also, OTR can be used for
digital offset and gain calibration.
Table V. Out-of-Range Truth Table
OTR MSB Analog Input Is
0 0 In Range
0 1 In Range
1 0 Underrange
1 1 Overrange
OVER = “1”
UNDER = “1”
MSB
OTR
MSB
Figure 43. Overrange or Underrange Logic
Digital Output Driver Considerations (DRVDD)
The AD9241 output drivers can be configured to interface with
+5 V or 3.3 V logic families by setting DRVDD to +5 V or 3.3 V
respectively. The AD9241 output drivers are sized to provide
sufficient output current to drive a wide variety of logic families.
However, large drive currents tend to cause glitches on the
supplies and may affect SINAD performance. Applications
requiring the AD9241 to drive large capacitive loads or large
fanout may require additional decoupling capacitors on DRVDD.
In extreme cases, external buffers or latches may be required.
Clock Input and Considerations
The AD9241 internal timing uses the two edges of the clock
input to generate a variety of internal timing signals. The clock
input must meet or exceed the minimum specified pulse width
high and low (t
CH
and t
CL
) specifications for the given A/D, as
defined in the Switching Specifications section at the beginning
of the data sheet, to meet the rated performance specifications.
For example, the clock input to the AD9241 operating at 1.25
MSPS may have a duty cycle between 45% to 55% to meet this
timing requirement since the minimum specified t
CH
and t
CL
is
360 ns. For clock rates below 1.25 MSPS, the duty cycle may
deviate from this range to the extent that both t
CH
and t
CL
are
satisfied.
All high speed, high resolution A/Ds are sensitive to the quality
of the clock input. The degradation in SNR at a given full-scale
input frequency (f
IN
) due only to aperture jitter (t
A
) can be
calculated with the following equation:
SNR = 20 log
10
[1/(2 π f
IN
t
A
)]
In the equation, the rms aperture jitter, t
A
, represents the root-
sum square of all the jitter sources including the clock input,
analog input signal and A/D aperture jitter specification. For
example, if a 1.0 MHz full-scale sine wave is sampled by an A/D
with a total rms jitter of 15 ps, the SNR performance of the A/D
will be limited to 80.5 dB. Undersampling applications are
particularly sensitive to jitter.
The clock input should be treated as an analog signal in cases
where aperture jitter may affect the dynamic range of the
AD9241. As such, supplies for clock drivers should be separated
from the A/D output driver supplies to avoid modulating the
clock signal with digital noise. Low jitter crystal controlled oscil-
lators make the best clock sources. If the clock is generated from
another type of source (by gating, dividing or other method), it
should be retimed by the original clock at the last step.
Most of the power dissipated by the AD9241 is from the analog
power supply. However, lower clock speeds will slightly reduce
digital current. Figure 44 shows the relationship between power
and clock rate.
CLOCK RATE – MHz
150
140
110
6
POWER – mW
5
130
120
5V p-p
2V p-p
100
90
80
43210
70
60 78910
Figure 44. AD9241 Power Consumption vs. Clock
Frequency
GROUNDING AND DECOUPLING
Analog and Digital Grounding
Proper grounding is essential in any high speed, high resolution
system. Multilayer printed circuit boards (PCBs) are recom-
mended to provide optimal grounding and power schemes. The
use of ground and power planes offers distinct advantages:
1. The minimization of the loop area encompassed by a signal
and its return path.
2. The minimization of the impedance associated with ground
and power paths.
3. The inherent distributed capacitor formed by the power
plane, PCB insulation and ground plane.
These characteristics result in both a reduction of electro-
magnetic interference (EMI) and an overall improvement in
performance.
It is important to design a layout that prevents noise from coupling
onto the input signal. Digital signals should not be run in paral-
lel with input signal traces, and should be routed away from the
input circuitry. While the AD9241 features separate analog and
digital ground pins, it should be treated as an analog compo-
nent. The AVSS, DVSS and DRVSS pins must be joined to-
gether directly under the AD9241. A solid ground plane under
the A/D is acceptable if the power and ground return currents
are carefully managed. Alternatively, the ground plane under
AD9241
REV. 0 –19–
the A/D may contain serrations to steer currents in predictable
directions where cross-coupling between analog and digital
would otherwise be unavoidable. The AD9241/EB ground lay-
out shown in Figure 52 depicts the serrated type of arrange-
ment. The analog and digital grounds are connected by a jumper
below the A/D.
Analog and Digital Supply Decoupling
The AD9241 features separate analog and digital supply and
ground pins, helping to minimize digital corruption of sensitive
analog signals.
FREQUENCY – kHz
120
PSRR – dBFS
100
1000
80
60
40 100101
DVDD
AVDD
Figure 45. PSRR vs. Frequency
Figure 45 shows the power supply rejection ratio vs. frequency
for a 200 mV p-p ripple applied to both AVDD and DVDD.
In general, AVDD, the analog supply, should be decoupled to
AVSS, the analog common, as close to the chip as physically
possible. Figure 46 shows the recommended decoupling for the
analog supplies; 0.1 µF ceramic chip capacitors should provide
adequately low impedance over a wide frequency range. Note
that the AVDD and AVSS pins are co-located on the AD9241
to simplify the layout of the decoupling capacitors and provide
the shortest possible PCB trace lengths. The AD9241/EB power
plane layout shown in Figure 53 depicts a typical arrangement
using a multilayer PCB.
0.1µF
AVDD
AVSS
AD9241
0.1µF
AVDD
AVSS
Figure 46. Analog Supply Decoupling
The CML is an internal analog bias point used internally by the
AD9241. This pin must be decoupled with at least a 0.1 µF
capacitor as shown in Figure 47. The dc level of CML is ap-
proximately AVDD/2. This voltage should be buffered if it is to
be used for any external biasing.
0.1µF CML
AD9241
Figure 47. CML Decoupling
The digital activity on the AD9241 chip falls into two general
categories: correction logic and output drivers. The internal
correction logic draws relatively small surges of current, prima-
rily during the clock transitions. The output drivers draw large
current impulses while the output bits are changing. The size
and duration of these currents are a function of the load on the
output bits: large capacitive loads are to be avoided. Note that
the internal correction logic of the AD9241 is referenced DVDD
while the output drivers are referenced to DRVDD.
The decoupling shown in Figure 48 (a 0.1 µF ceramic chip
capacitor) is appropriate for a reasonable capacitive load on the
digital outputs (typically 20 pF on each pin). Applications in-
volving greater digital loads should consider increasing the digi-
tal decoupling proportionally and/or using external buffers/
latches.
0.1µF DVDD
DVSS
AD9241
DRVDD
DRVSS
0.1µF
Figure 48. Digital Supply Decoupling
A complete decoupling scheme will also include large tantalum
or electrolytic capacitors on the PCB to reduce low-frequency
ripple to negligible levels. Refer to the AD9241/EB schematic
and layouts in Figures 49–53 for more information regarding the
placement of decoupling capacitors.
AD9241
REV. 0
–20–
Figure 49. Evaluation Board Schematic
2
3
AD817
V
EE
U3
AR7
1kΩ
C16
0.1µF
R8
316Ω
A
Q1
2N2222
A
C17
10µF
16V
A
C18
0.1µF
R6
820Ω
+5VA
TP25
R4
50Ω
JP10 R3
15kΩ
A
C12
0.1µF
A
R5
10kΩC13
10µF
16V
A
V
IN
V
OUT
GND
6
2
4
REF43
A
EXTERNAL REFERENCE DRIVE
U2
VCC
C14
0.1µF
6
7
4
V
CC
A
C15
0.1µF
C19
0.1µF
6
7
2
34
V
CC
AD845
A
C21
0.1µF
V
EE
U4
A
R11
500Ω
C20
0.1µF
A
R14
10kΩ
A
CW
R13
10kΩ
BUFFER
JP23
R10
500Ω
1
2
3
A
B
JP24
DIRECT COUPLE OPTION
C38
?
AC COUPLE OPTION
JP14
JP13
A
R9
50Ω
A
J1
VIN
R12
33Ω
R15
33Ω
+5VA
A
D2
1N5711
D1
1N5711
AC COUPLE OPTION
+5VA
A
D4
1N5711
D3
1N5711
R39
?
2J8
4J8
6J8
8J8
10 J8
12 J8
14 J8
16 J8
18 J8
20 J8
22 J8
24 J8
26 J8
39 J8
28 J8
29 J8
30 J8
31 J8
32 J8
34 J8
35 J8
36 J8
37 J8
38 J8
NC
NC
NC
+5VA
C26
0.1µF
U8
DECOUPLING
A
11 10
U8
98
U8
34
U8
12
U8
13 12
U8
A
+5VD
C22
0.1µF
U5
DECOUPLING
56
U5
12
U5
34
U5
SPARE GATES
JP18
JP17
R20
22.1Ω13 J8
TP10
R21
22.1Ω11 J8
TP11
R22
22.1Ω9J8
TP12
R23
22.1Ω7J8
TP13
R24
22.1Ω5J8
TP14
R25
22.1Ω3J8
TP15
R26
22.1Ω1J8
TP16
R27
22.1Ω33 J8
TP3
R28
22.1Ω27 J8
TP4
R29
22.1Ω25 J8
TP5
R30
22.1Ω23 J8
TP17
R31
22.1Ω21 J8
TP6
R32
22.1Ω19 J8
TP7
R33
22.1Ω17 J8
TP8
R34
22.1Ω15 J8
TP9
11
12
13
14
15
16
17
18
C24
0.1µF
+DRVDD
74HC541N
20
Y7
Y6
Y5
Y4
Y3
Y2
Y1
Y0
+5VD
U6
G1
G2
A7
A6
A5
A4
A3
A2
A1
A0
GND
1
19
9
8
7
6
5
4
3
2
10
D7
D8
D9
D10
D11
D12
D13
11
12
13
14
15
16
17
18
C25
0.1µF
+DRVDD
74HC541N
20
Y7
Y6
Y5
Y4
Y3
Y2
Y1
Y0
+5VD
U7
G1
G2
A7
A6
A5
A4
A3
A2
A1
A0
GND
1
19
9
8
7
6
5
4
3
2
10
CLK
D0
D1
D2
D3
D4
D5
D6
A
J9
CLKIN
U5
13 12
U5
98
JP15
CLKB
JP16
CLK
U5
11 10
U8
65
C23
0.1µF
TP2
A
R19
50Ω
R40
?
R41
?
A
R16
5kΩ
+5VA
R18
5kΩ
R17
1kΩCW
BIT1
BIT2
BIT3
BIT4
BIT5
BIT6
BIT7
BIT8
BIT9
BIT10
BIT11
BIT12
BIT13
BIT14
OTRVREF
SENSE
REFCOM
CAPT
CAPB
CML
VINA
VINB
U1
AD9241MQFP
AVDD2
AVDD1
AVSS2
AVSS1
32
31
33
37
36
39
41
42
28 4229
TP24
C8
0.1µF A
A
JP7 +5VA
C9
0.1µF
A
25
24
23
22
21
20
19
18
17
16
15
14
13
12
11
D13
D12
D11
D10
D9
D8
D7
D6
D5
D4
D3
D2
D1
D0
5
7
ADC_CLK
C43
0.1µF C10
0.1µF C11
0.1µF
+DRVDD
C2
0.1µF
+C1
10µF
16V
A
JP6
JP3
+5VA
JP4
JP5
R1
10kΩ
R2
10kΩ
A
C41
0.1µF
JP2
TPC
TPD
A
C6
0.1µF
+C5
10µF
16V
C3
0.1µF
C4
0.1µF
A
CML
C7
0.1µF
A
VINA2
VINA1 JP11 BA
321
VINB2
VINB1 JP12 BA
321
JP8 +5VD
D13
ADC_CLK
A
DRVSS
DVSS
DRVDD
DVDD
T1
6
5
4
1
2
3
PRI SEC
R36
200Ω
A
C36
15pF
R37
33Ω
R38
33Ω
VINA1
VINB1
A
C37
15pF
JP21 TPC
JP22 TPD
JP1 CML
C42
0.1µF
R35
50Ω
A
A
J10
AIN
ATP26
SJ6
40 J8
AA
+C28
22µF
25V
C32
0.1µF
L1
TP18
J2+5A
AA
+C29
22µF
25V
C33
0.1µF
L2
TP19
J3+5D
AA
+C30
22µF
25V
C34
0.1µF
L3
TP20
J4+VCC
AA
+C31
22µF
25V
C35
0.1µF
L4
TP21
J5–VEE
+C39
22µF
25V
C40
0.1µF
L5
TP27
J11
+5_OR _+3 +DRVDD
VEE
VCC
+5VD
+5VA
J6
TP23
J7
A
JG1-WIRE ETCH
CKT SIDE
5 SETS OF
PADS TO
CONNECT
GROUNDS
JG1
SJ1
SJ2
SJ3
SJ4
SJ5
AGND
DGND
TP22
TP1
163
CLK
VINA2
VINB2
TPD
TPD
TPC
74HC14
74HC04
AD9241
REV. 0 –21–
Figure 50. Evaluation Board Component Side Layout (Not to Scale)
Figure 51. Evaluation Board Solder Side Layout (Not to Scale)
AD9241
REV. 0
–22–
Figure 52. Evaluation Board Ground Plane Layout (Not to Scale)
Figure 53. Evaluation Board Power Plane Layout (Not to Scale)
AD9241
REV. 0 –23–
OUTLINE DIMENSIONS
Dimensions shown in mm and (inches).
44-Pin Metric Quad Flatpack (MQFP)
(S-44)
TOP VIEW
(PINS DOWN)
12
44
1
11
22
23
34
33
0.45 (0.018)
0.3 (0.012)
13.45 (0.529)
12.95 (0.510)
8.45 (0.333)
8.3 (0.327)
10.1 (0.398)
9.90 (0.390)
0.8 (0.031)
BSC
2.1 (0.083)
1.95 (0.077)
0.23 (0.009)
0.13 (0.005)
0.25 (0.01)
MIN
SEATING
PLANE
0
°
MIN
2.45 (0.096)
MAX
1.03 (0.041)
0.73 (0.029)
C2961–10–4/97PRINTED IN U.S.A.
–24–
