LT8303
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For more information www.linear.com/LT8303
Typical applicaTion
FeaTures DescripTion
100VIN Micropower Isolated
Flyback Converter with
150V/450mA Switch
The LT
®
8303 is a micropower high voltage isolated flyback
converter. By sampling the isolated output voltage directly
from the primary-side flyback waveform, the part requires
no third winding or opto-isolator for regulation. The output
voltage is programmed with a single external resistor.
Internal compensation and soft-start further reduce external
component count. Boundary mode operation provides a
small magnetic solution with excellent load regulation.
Low ripple Burst Mode operation maintains high efficiency
at light load while minimizing the output voltage ripple.
A 450mA, 150V DMOS power switch is integrated along
with all high voltage circuitry and control logic into a 5-lead
ThinSOT™ package.
The LT8303 operates from an input voltages range of 5.5V
to 100V and can deliver up to 5W of isolated output power.
The high level of integration and the use of boundary mode
and low ripple Burst Mode operations result in a simple to
use, low component count, and high efficiency application
solution for isolated power delivery.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497,
and 7471522.
6V to 80VIN, 5VOUT Isolated Flyback Converter
applicaTions
n 5.5V to 100V Input Voltage Range
n 450mA, 150V Internal DMOS Power Switch
n Up to 5W of Output Power
n Low Quiescent Current:
70µA in Sleep Mode
280µA in Active Mode
n Boundary Mode Operation at Heavy Load
n Low Ripple Burst Mode
®
Operation at Light Load
n Minimum Load <0.5% (Typ) of Full Output
n VOUT Set with a Single External Resistor
n No Transformer Third Winding or Opto-Isolator
Required for Regulation
n Accurate EN/UVLO Threshold and Hysteresis
n Internal Compensation and Soft-Start
n 5-Lead TSOT-23 Package
n Isolated Telecom, Datacom, Automotive, Industrial,
and Medical Power Supplies
n Isolated Auxiliary/Housekeeping Power Supplies
Efficiency vs Load Current
V
IN
= 12V
V
IN
= 24V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
100
200
300
400
500
600
700
800
900
40
50
60
70
80
90
100
EFFICIENCY (%)
LT8303
T1
6:1
RFB
SW
150µH 4.2µH
EN/UVLO
4.7µF VIN
VIN
6V TO 80V
VOUT
+
5V
VOUT
–
GND
316k 2.5mA TO 0.33A (VIN = 12V)
2.5mA TO 0.52A (VIN = 24V)
2.5mA TO 0.73A (VIN = 48V)
2.5mA TO 0.84A (VIN = 72V)
•
•
100µF
8303 TA01a
LT8303
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For more information www.linear.com/LT8303
pin conFiguraTionabsoluTe MaxiMuM raTings
SW (Note 2) ........................................................... 150V
VIN ......................................................................... 100V
EN/UVLO ................................................................... VIN
RFB ...................................................... VIN – 0.5V to VIN
Current into RFB ................................................... 200µA
Operating Junction Temperature Range (Notes 3, 4)
LT8303E, LT8303I ............................. –40°C to 125°C
LT8303H ............................................ –40°C to 150°C
Storage Temperature Range .............. –65°C to 150°C
(Note 1)
EN/UVLO 1
GND 2
TOP VIEW
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
RFB 3
5 V
IN
4 SW
θJA = 215°C/W
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT8303ES5#PBF LT8303ES5#TRPBF LTGXH 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8303IS5#PBF LT8303IS5#TRPBF LTGXH 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8303HS5#PBF LT8303HS5#TRPBF LTGXH 5-Lead Plastic TSOT-23 –40°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
http://www.linear.com/product/LT8303#orderinfo
LT8303
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elecTrical characTerisTics
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 150V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 150V as shown
in Figure 5.
Note 3: The LT8303E is guaranteed to meet performance specifications
from 0°C to 125°C operating junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT
VIN Input Voltage Range 5.5 100 V
VIN UVLO Threshold Rising
Falling
5.3
3.2
5.5 V
V
IQVIN Quiescent Current VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)
1.5
200
70
280
2.5 µA
µA
µA
µA
EN/UVLO Shutdown Threshold For Lowest Off IQl0.3 0.75 V
EN/UVLO Enable Threshold Falling
Hysteresis
l1.186 1.223
0.016
1.284 V
V
IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
VEN/UVLO = 1.3V
–0.1
2.1
–0.1
0
2.5
0
0.1
2.9
0.1
µA
µA
µA
fMAX Maximum Switching Frequency 320 350 380 kHz
fMIN Minimum Switching Frequency 5 7 9 kHz
tON(MIN) Minimum Switch-On Time 160 ns
tOFF(MIN) Minimum Switch-Off Time VIN = VEN/UVLO = 12V 350 ns
tOFF(MAX) Maximum Switch-Off Time Backup Timer 200 µs
ISW(MAX) Maximum SW Current Limit 450 535 620 mA
ISW(MIN) Minimum SW Current Limit 70 105 140 mA
SW Over Current Limit To Initiate Soft-Start 1 A
RDS(ON) Switch On-Resistance ISW = 100mA 3.2 Ω
ILKG Switch Leakage Current VIN = 100V, VSW = 150V 0.1 0.5 µA
IRFB RFB Regulation Current l97.5 100 102.5 µA
RFB Regulation Current Line Regulation 5.5V ≤ VIN ≤ 100V 0.001 0.01 %/V
design, characterization and correlation with statistical process controls.
The LT8303I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8303H is guaranteed over the full –40°C to
150°C operating junction temperature range. High junction temperatures
degrade operating lifetimes. Operating lifetime is derated at junction
temperatures greater than 125°C.
Note 4: The LT8303 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
LT8303
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Typical perForMance characTerisTics
Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms
VIN Shutdown Current
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
Output Load and Line Regulation Output Temperature Variation
Switching Frequency
vs Load Current
TA = 25°C, unless otherwise noted.
FRONT PAGE APPLICATION
V
IN
= 12V
V
IN
= 24V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
100
200
300
400
500
600
700
800
900
4.7
4.8
4.9
5.0
5.1
5.2
5.3
OUTPUT VOLTAGE (V)
8303 G01
FRONT PAGE APPLICATION
V
IN
= 48V
I
OUT
= 3mA
I
OUT
= 200mA
I
OUT
= 700mA
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
4.7
4.8
4.9
5.0
5.1
5.2
5.3
OUTPUT VOLTAGE (V)
8303 G02
FRONT PAGE APPLICATION
V
IN
= 12V
V
IN
= 24V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
100
200
300
400
500
600
700
800
900
0
50
100
150
200
250
300
350
400
FREQUENCY (kHz)
8303 G03
FRONT PAGE APPLICATION
V
IN
= 48V, I
OUT
= 700mA
2µs/DIV
V
SW
50V/DIV
V
OUT
50mV/DIV
8303 G04
FRONT PAGE APPLICATION
V
IN
= 48V, I
OUT
= 200mA
2µs/DIV
V
SW
50V/DIV
V
OUT
50mV/DIV
8303 G05
FRONT PAGE APPLICATION
V
IN
= 48V, I
OUT
= 3mA
20µs/DIV
V
SW
50V/DIV
V
OUT
50mV/DIV
8303 G06
T
J
= –50°C
T
J
= 25°C
T
J
= 150°C
V
IN
(V)
0
20
40
60
80
100
0
2
4
6
8
10
I
Q
(µA)
8303 G07
T
J
= 150°C
T
J
= 25°C
T
J
= –50°C
V
IN
(V)
0
20
40
60
80
100
40
50
60
70
80
90
100
I
Q
(µA)
8303 G08
T
J
= 150°C
T
J
= 25°C
T
J
= –50°C
V
IN
(V)
0
20
40
60
80
100
240
260
280
300
320
340
I
Q
(µA)
8303 G09
LT8303
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For more information www.linear.com/LT8303
Typical perForMance characTerisTics
RDS(ON) Switch Current Limit Maximum Switching Frequency
Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time
EN/UVLO Enable Threshold EN/UVLO Hysteresis Current RFB Regulation Current
TA = 25°C, unless otherwise noted.
RISING
FALLING
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
1.20
1.21
1.22
1.23
1.24
1.25
1.26
1.27
1.28
V
EN/UVLO
(V)
8303 G10
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
1
2
3
4
5
I
HYST
(µA)
8303 G11
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
95
96
97
98
99
100
101
102
103
104
105
I
RFB
(µA)
8303 G12
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
2
4
6
8
10
RESISTANCE (Ω)
8303 G13
1SW = 100mA
MAXIMUM CURRENT LIMIT
MINIMUM CURRENT LIMIT
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
500
600
700
I
SW
(mA)
8303 G14
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
500
FREQUENCY (kHz)
8303 G15
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
TIME (ns)
8303 G17
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
TIME (ns)
8303 G18
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
4
8
12
16
20
FREQUENCY (kHz)
8303 G16
LT8303
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pin FuncTions
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8303. Pull the pin
below 0.3V to shut down the LT8303. This pin has an ac-
curate 1.223V threshold and can be used to program a VIN
undervoltage lockout (UVLO) threshold using a resistor
divider from VIN to ground. A 2.5µA current hysteresis
allows the programming of VIN UVLO hysteresis. If neither
function is used, tie this pin directly to VIN.
GND (Pin 2): Ground. Tie this pin directly to local ground
plane.
RFB (Pin 3): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer pri-
mary SW pin. The ratio of the RFB resistor to the internal
trimmed 12.23k resistor, times the internal bandgap
reference, determines the output voltage (plus the effect
of any non-unity transformer turns ratio). Minimize trace
area at this pin.
SW (Pin 4): Drain of the 150V Internal DMOS Power
Switch. Minimize trace area at this pin to reduce EMI and
voltage spikes.
VIN (Pin 5): Input Supply. The VIN pin supplies current
to internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the RFB pin. Locally
bypass this pin to ground with a capacitor.
block DiagraM
8303 BD
–
+
–
+
OSCILLATOR
1:4
S
R Q
1.223V
25µA
M2M3
BOUNDARY
DETECTOR
DRIVER
–
+
A2
A3
RSENSE
M1
gm
RREF
12.23kΩ
RFB
2.5µA
R2
EN/UVLO
M4
3 45
–
+
1.223V
A1
1
REFERENCE
REGULATORS
VIN
2
GND
RFB SWVIN
V
IN
T1
NPS:1
D
OUT
LSEC
LPRI
VOUT
+
VOUT
–
•
•
COUT
CIN
R1
LT8303
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For more information www.linear.com/LT8303
operaTion
The LT8303 is a current mode switching regulator IC
designed specially for the isolated flyback topology. The
key problem in isolated topologies is how to communicate
the output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicate this information across the isolation bound-
ary. Opto-isolator circuits waste output power, and the
extra components increase the cost and physical size of
the power supply. Opto-isolators can also cause system
issues due to limited dynamic response, nonlinearity, unit-
to-unit variation and aging over lifetime. Circuits employing
extra transformer windings also exhibit deficiencies, as
using an extra winding adds to the transformer’s physical
size and cost, and dynamic response is often mediocre.
The LT8303 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this manner,
neither opto-isolator nor extra transformer winding is re-
quired for regulation. Since the LT8303 operates in either
boundary conduction mode or discontinuous conduction
mode, the output voltage is always sampled on the SW
pin when the secondary current is zero. This method im-
proves load regulation without the need of external load
compensation components.
The LT8303 is a simple to use micropower isolated flyback
converter housed in a 5-lead TSOT-23 package. The output
voltage is programmed with a single external resistor. By
integrating the loop compensation and soft-start inside, the
part further reduces the number of external components.
As shown in the Block Diagram, many of the blocks are
similar to those found in traditional switching regulators
including reference, regulators, oscillator, logic, current
amplifier, current comparator, driver, and power switch.
The novel sections include a flyback pulse sense circuit,
a sample-and-hold error amplifier, and a boundary mode
detector, as well as the additional logic for boundary
conduction mode, discontinuous conduction mode, and
low ripple Burst Mode operation.
Boundary Conduction Mode Operation
The LT8303 features boundary conduction mode operation
at heavy load, where the chip turns on the primary power
switch when the secondary current is zero. Boundary
conduction mode is a variable frequency, variable peak-
current switching scheme. The power switch turns on
and the transformer primary current increases until an
internally controlled peak current limit. After the power
switch turns off, the voltage on the SW pin rises to the
output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on.
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc-
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit sub-harmonic oscillation.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode in-
creases the switching frequency and decreases the switch
peak current at the same ratio. Running at a higher switching
frequency up to several MHz increases switching and gate
charge losses. To avoid this scenario, the LT8303 has an
additional internal oscillator, which clamps the maximum
switching frequency to be less than 350kHz (typical). Once
the switching frequency hits the internal frequency clamp,
the part starts to delay the switch turn-on and operates in
discontinuous conduction mode.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8303 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling
of the output voltage. The inherent minimum switch cur-
rent limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.
As the load gets very light, the LT8303 starts to fold back
the switching frequency while keeping the minimum switch
current limit. So the load current is able to decrease while
still allowing minimum switch-off time for the sample-and-
hold error amplifier. Meanwhile, the part switches between
sleep mode and active mode, thereby reducing the effec-
LT8303
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operaTion
tive quiescent current to improve light load efficiency. In
this condition, the LT8303 operates in low ripple Burst
Mode. The typical 7kHz minimum switching frequency
Output Voltage
The RFB resistor as depicted in the Block Diagram is the
only external resistor used to program the output voltage.
The LT8303 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the VIN supply. The
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = Output diode forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current IRFB by
the flyback pulse sense circuit (M2 and M3). This cur-
rent IRFB also flows through the internal trimmed 12.23k
RREF resistor to generate a ground-referred voltage. The
resulting voltage feeds to the inverting input of the sample-
and-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (ISEC • ESR) term in the VFLBK equation can be
assumed to be zero.
The bandgap reference voltage VBG, 1.223V, feeds to the
non-inverting input of the sample-and-hold error ampli-
fier. The relatively high gain in the overall loop causes
the voltage across RREF resistor to be nearly equal to the
applicaTions inForMaTion
determines how often the output voltage is sampled and
also the minimum load requirement.
bandgap reference voltage VBG. The resulting relationship
between VFLBK and VBG can be expressed as:
VFLBK
RFB
⎛
⎝
⎜
⎞
⎠
⎟•RREF =VBG
or
VFLBK =VBG
RREF
⎛
⎝
⎜
⎞
⎠
⎟•RFB =IRFB •RFB
VBG = Bandgap reference voltage
IRFB = RFB regulation current = 100µA
Combination with the previous VFLBK equation yields an
equation for VOUT, in terms of the RFB resistor, transformer
turns ratio, and diode forward voltage:
VOUT =100µA •RFB
NPS
⎛
⎝
⎜
⎞
⎠
⎟−VF
Output Temperature Coefficient
The first term in the VOUT equation does not have tempera-
ture dependence, but the output diode forward voltage VF
has a significant negative temperature coefficient (–1mV/°C
to –2mV/°C). Such a negative temperature coefficient pro-
duces approximately 200mV to 300mV voltage variation
on the output voltage across temperature.
For higher voltage outputs, such as 12V and 24V, the output
diode temperature coefficient has a negligible effect on the
output voltage regulation. For lower voltage outputs, such
as 3.3V and 5V, however, the output diode temperature
coefficient does count for an extra 2% to 5% output voltage
regulation. For customers requiring tight output voltage
regulation across temperature, please refer to other LTC
parts with integrated temperature compensation features.
LT8303
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applicaTions inForMaTion
Selecting Actual RFB Resistor Value
The LT8303 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
which will affect the output voltage and force a re-evaluation
of the RFB resistor value. Therefore, a simple two-step
process is required to choose feedback resistor RFB.
Rearrangement of the expression for VOUT in the Output
Voltage section yields the starting value for RFB:
RFB =NPS •VOUT +VF
( )
100µA
VOUT = Output voltage
VF = Output diode forward voltage = ~0.3V
NPS = Transformer effective primary-to-secondary
turns ratio
Power up the application with the starting RFB value and
other components connected, and measure the regulated
output voltage, VOUT(MEAS). The final RFB value can be
adjusted to:
RFB(FINAL) =
V
OUT
VOUT(MEAS)
•RFB
Once the final RFB value is selected, the regulation accuracy
from board to board for a given application will be very
consistent, typically under ±5% when including device
variation of all the components in the system (assuming
resistor tolerances and transformer windings matching
within ±1%). However, if the transformer or the output
diode is changed, or the layout is dramatically altered,
there may be some change in VOUT.
Output Power
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and output
currents which make it similar to a non-isolated buck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In ad-
dition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 120V dur-
ing the switch-off time. 30V of margin is left for leakage
inductance voltage spike. To achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 120V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 30V and a maximum input
voltage of 80V. A six-to-one winding ratio fits this design
example perfectly and outputs equal to 4.35W at 80V but
lowers to 2.95W at 30V.
The following equations calculate output power:
P
OUT = η
•V
IN
•D•I
SW(MAX)
•0.5
η = Efficiency =85%
D=DutyCycle =VOUT +VF
( )
•NPS
VOUT +VF
( )
•NPS +V
IN
ISW(MAX) = Maximum switch current limit = 450mA
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applicaTions inForMaTion
Figure 4. Output Power for 24V Output
Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output
Figure 3. Output Power for 12V Output
Primary Inductance Requirement
The LT8303 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected output
voltage. In order to ensure proper sampling, the second-
ary winding needs to conduct current for a minimum of
350ns. The following equation gives the minimum value
for primary-side magnetizing inductance:
LPRI ≥tOFF(MIN) •NPS •VOUT +VF
( )
ISW(MIN)
tOFF(MIN) = Minimum switch-off time = 350ns
ISW(MIN) = Minimum switch current limit = 105mA
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8303 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blank-
ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
LPRI ≥
t
ON(MIN)
•V
IN(MAX)
ISW(MIN)
tON(MIN) = Minimum Switch-On Time = 160ns
N = 12:1
N = 8:1
N = 6:1
N = 4:1
ASSUME 80% EFFICIENCY
MAXIMUM
OUTPUT
POWER
INPUT VOLTAGE (V)
0
20
40
60
80
100
0
1
2
3
4
5
6
OUTPUT POWER (W)
8303 F01
N = 8:1
N = 6:1
N = 4:1
N = 2:1
ASSUME 85% EFFICIENCY
MAXIMUM
OUTPUT
POWER
INPUT VOLTAGE (V)
0
20
40
60
80
100
0
1
2
3
4
5
6
OUTPUT POWER (W)
8303 F02
N = 2:1
N = 3:2
N = 1:1
N = 1:2
ASSUME 85% EFFICIENCY
MAXIMUM
OUTPUT
POWER
INPUT VOLTAGE (V)
0
20
40
60
80
100
0
1
2
3
4
5
6
OUTPUT POWER (W)
8303 F04
N = 4:1
N = 3:1
N = 2:1
N = 1:1
ASSUME 85% EFFICIENCY
MAXIMUM
OUTPUT
POWER
INPUT VOLTAGE (V)
0
20
40
60
80
100
0
1
2
3
4
5
6
OUTPUT POWER (W)
8303 F03
LT8303
11
8303fa
For more information www.linear.com/LT8303
applicaTions inForMaTion
In general, choose a transformer with its primary mag-
netizing inductance about 40% to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Selecting a Transformer
T
ransformer specification and design is perhaps the most
critical part of successfully applying the LT8303. In addition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Linear Technology has worked with several leading mag-
netic component manufacturers to produce pre-designed
flyback transformers for use with the LT8303. Table 1
shows the details of these transformers.
Turns Ratio
Note that when choosing the RFB resistor to set output
voltage, the user has relative freedom in selecting a trans-
former turns ratio to suit a given application. In contrast,
the use of simple ratios of small integers, e.g., 4:1, 2:1,
1:1, provides more freedom in settling total turns and
mutual inductance.
Table 1. Predesigned Transformers – Typical Specifications
TRANSFORMER
PART NUMBER
DIMENSION
(W × L × H) (mm)
LPRI, µH
TYP
LLKG, µH
TYP (MAX) NP: NSVENDOR
TARGET APPLICATION
VIN (V) VOUT (V) IOUT (A)
750315825 13.36 × 10.16 × 8.64 150 3 (6) 8:1 Wurth Elektronik 36 to 75 3.3 0.9
750315826 13.36 × 10.16 × 8.64 150 2 (4) 6:1 Wurth Elektronik 36 to 75 5 0.65
750315827 13.36 × 10.16 × 8.65 150 1.8 (3.6) 4:1 Wurth Elektronik 36 to 75 5 0.5
750315828 13.36 × 10.16 × 8.66 150 1.6 (3.2) 2:1 Wurth Elektronik 36 to 75 12 0.25
750315829 13.36 × 10.16 × 8.67 150 1.5 (3) 1:1 Wurth Elektronik 36 to 75 24 0.12
750315830 13.36 × 10.16 × 8.68 150 1.9 (3.8) 1:2 Wurth Elektronik 36 to 75 48 0.06
750315833 13.36 × 10.16 × 8.71 150 1.5 (3) 2:1:1 Wurth Elektronik 36 to 75 12/12 0.12/0.12
750315834 13.36 × 10.16 × 8.72 150 2.6 (5.2) 6:1:1 Wurth Elektronik 36 to 75 5/5 0.32/0.32
PS15-108 14 × 10 × 9.2 150 (5) 8:1 Sumida 36 to 75 3.3 0.9
PS15-109 14 × 10 × 9.2 150 (5) 6:1 Sumida 36 to 75 5 0.65
PS15-110 14 × 10 × 9.2 150 (5) 4:1 Sumida 36 to 75 5 0.5
PS15-111 14 × 10 × 9.2 150 (5) 2:1 Sumida 36 to 75 12 0.25
PS15-112 14 × 10 × 9.2 150 (5) 1:1 Sumida 36 to 75 24 0.12
PS15-113 14 × 10 × 9.2 150 (5) 1:2 Sumida 36 to 75 48 0.06
LT8303
12
8303fa
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Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V or
5V), a larger N:1 turns ratio can be used with multiple
primary windings relative to the secondary to maximize
the transformer’s current gain (and output power).
However, remember that the SW pin sees a voltage that
is equal to the maximum input supply voltage plus the
output voltage multiplied by the turns ratio. In addition,
leakage inductance will cause a voltage spike (VLEAKAGE)
on top of this reflected voltage. This total quantity needs
to remain below the 150V absolute maximum rating of
the SW pin to prevent breakdown of the internal power
switch. Together these conditions place an upper limit
on the turns ratio, NPS, for a given application. Choose a
turns ratio low enough to ensure:
NPS <
150V
−
V
IN(MAX) −
V
LEAKAGE
VOUT +VF
For lower output power levels, choose a smaller N:1 turns
ratio to alleviate the SW pin voltage stress. Although a
1:N turns ratio makes it possible to have very high output
voltages without exceeding the breakdown voltage of the
internal power switch, the multiplied parasitic capacitance
through turns ratio coupled with the relatively resistive
150V internal power switch may cause the switch turn-on
current spike ringing beyond 160ns leading-edge blanking,
thereby producing light load instability in certain applica-
tions. So any 1:N turns ratio should be fully evaluated
before its use with the LT8303.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci-
fies turns ratio accuracy within ±1%.
applicaTions inForMaTion
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core is
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8303, the saturation
current should always be specified by the transformer
manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding re-
sistance due to the boundary/discontinuous conduction
mode operation of the LT8303.
Leakage Inductance and Snubbers
Transformer leakage inductance on either the primary or
secondary causes a voltage spike to appear on the primary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored en-
ergy must be dissipated. It is very important to minimize
transformer leakage inductance.
When designing an application, adequate margin should be
kept for the worst-case leakage voltage spikes even under
overload conditions. In most cases shown in Figure 5, the
reflected output voltage on the primary plus VIN should
be kept below 120V. This leaves at least 30V margin for
the leakage spike across line and load conditions. A larger
voltage margin will be required for poorly wound trans-
formers or for excessive leakage inductance.
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig-
ger boundary mode detector, the LT8303 internally blanks
the boundary mode detector for approximately 250ns. Any
remaining voltage ringing after 250ns may turn the power
switch back on again before the secondary current falls
to zero. So the leakage inductance spike ringing should
be limited to less than 250ns.
LT8303
13
8303fa
For more information www.linear.com/LT8303
applicaTions inForMaTion
A snubber circuit is recommended for most applications.
Two types of snubber circuits shown in Figure 6 that can
protect the internal power switch include the DZ (diode-
Zener) snubber and the RC (resistor-capacitor) snubber. The
DZ snubber ensures well defined and consistent clamping
voltage and has slightly higher power efficiency, while the
RC snubber quickly damps the voltage spike ringing and
provides better load regulation and EMI performance.
Figure 5 shows the flyback waveforms with the DZ and
RC snubbers.
For the DZ snubber, proper care must be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leak-
age inductance spike. Choose a diode that has a reverse-
voltage rating higher than the maximum SW pin voltage.
The Zener diode breakdown voltage should be chosen to
balance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown.
Use the following equation to make the proper choice:
VZENER(MAX) ≤ 150V – VIN(MAX)
For an application with a maximum input voltage of 80V,
choose a 62V Zener diode, the VZENER(MAX) of which is
around 65V and below the 70V maximum.
The power loss in the clamp will determine the power rat-
ing of the Zener diode. Power loss in the clamp is highest
at maximum load and minimum input voltage. The switch
current is highest at this point along with the energy stored
in the leakage inductance. A 0.5W Zener will satisfy most
applications when the highest VZENER is chosen.
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
Figure 6. Snubber Circuits
8303 F05
VSW
tOFF > 350ns
VLEAKAGE
tSP < 250ns
VSW VSW
TIME
No Snubber with DZ Snubber with RC Snubber
tOFF > 350ns
VLEAKAGE
tSP < 250ns
TIME
tOFF > 350ns
VLEAKAGE
tSP < 250ns
TIME
<150V
<120V
<150V
<120V
<150V
<120V
8303 F06
DZ Snubber RC Snubber
Lℓ
Z
D
C
R
•
•
Lℓ
•
•
LT8303
14
8303fa
For more information www.linear.com/LT8303
applicaTions inForMaTion
Tables 2 and 3 show some recommended diodes and
Zener diodes.
Table 2. Recommended Zener Diodes
PART
VZENER
(V)
POWER
(W) CASE VENDOR
MMSZ5266BT1G 68 0.5 SOD-123 On Semi
MMSZ5270BT1G 91 0.5 SOD-123
CMHZ5266B 68 0.5 SOD-123 Central
Semiconductor
CMHZ5267B 75 0.5 SOD-123
BZX84J-68 68 0.5 SOD323F NXP
BZX100A 100 0.5 SOD323F
Table 3. Recommended Diodes
PART I (A)
VREVERSE
(V) CASE VENDOR
BAV21W 0.625 200 SOD-123 Diodes Inc.
BAV20W 0.625 150 SOD-123
The recommended approach for designing an RC snubber
is to measure the period of the ringing on the SW pin when
the power switch turns off without the snubber and then
add capacitance (starting with 100pF) until the period of
the ringing is 1.5 to 2 times longer. The change in period
will determine the value of the parasitic capacitance, from
which the parasitic inductance can be determined from
the initial period, as well. Once the value of the SW node
capacitance and inductance is known, a series resistor can
be added to the snubber capacitance to dissipate power
and critically dampen the ringing. The equation for deriving
the optimal series resistance using the observed periods
( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance
(CSNUBBER) is:
CPAR =
C
SNUBBER
tPERIOD(SNUBBED)
tPERIOD
⎛
⎝
⎜
⎞
⎠
⎟
2
−
1
LPAR =tPERIOD2
CPAR •4π2
RSNUBBER =LPAR
CPAR
Figure 7. Undervoltage Lockout (UVLO)
LT8303
GND
EN/UVLO
R1
RUN/STOP
CONTROL
(OPTIONAL)
R2
V
IN
8303 F07
Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
may need to be sized for thermal dissipation.
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin imple-
ments undervoltage lockout (UVLO). The EN/UVLO pin
falling threshold is set at 1.223V with 16mV hysteresis.
In addition, the EN/UVLO pin sinks 2.5µA when the volt-
age at the pin is below 1.223V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
V
IN(UVLO+)=
1.239V •(R1
+
R2)
R2 +2.5µA •R1
V
IN(UVLO−)=1.223V •(R1+R2)
R2
Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8303 in shutdown with quiescent current less than 2.5µA.
LT8303
15
8303fa
For more information www.linear.com/LT8303
Minimum Load Requirement
The LT8303 samples the isolated output voltage from
the primary-side flyback pulse waveform. The flyback
pulse occurs once the primary switch turns off and the
secondary winding conducts current. In order to sample
the output voltage, the LT8303 has to turn on and off at
least for a minimum amount of time and with a minimum
frequency. The LT8303 delivers a minimum amount of
energy even during light load conditions to ensure ac-
curate output voltage information. The minimum energy
delivery creates a minimum load requirement, which can
be approximately estimated as:
ILOAD(MIN) =LPRI •ISW(MIN)
2•fMIN
2•V
OUT
LPRI = Transformer primary inductance
ISW(MIN) = Minimum switch current limit = 140mA (Max)
fMIN = Minimum switching frequency = 9kHz (Max)
The LT8303 typically needs less than 0.5% of its full output
power as minimum load. Alternatively, a Zener diode with its
breakdown of 20% higher than the output voltage can serve
as a minimum load if pre-loading is not acceptable. For a 5V
output, use a 6V Zener with cathode connected to the output.
Output Short Protection
When the output is heavily overloaded or shorted, the
reflected SW pin waveform rings longer than the internal
blanking time. After the 350ns minimum switch-off time,
the excessive ring falsely trigger the boundary mode
detector and turn the power switch back on again before
the secondary current falls to zero. Under this condition,
the LT8303 runs into continuous conduction mode at
350kHz maximum switching frequency. Depending on
the VIN supply voltage, the switch current may run away
and exceed 450mA maximum current limit. Once the
switch current hits 1A over current limit, a soft-start cycle
initiates and throttles back both switch current limit and
switch frequency. This output short protection prevents the
switch current from running away and limits the average
output diode current.
applicaTions inForMaTion
Design Example
Use the following design example as a guide to design
applications for the LT8303. The design example involves
designing a 12V output with a 200mA load current and an
input range from 30V to 80V.
VIN(MIN) = 30V, VIN(NOM) = 48V, VIN(MAX) = 80V,
VOUT = 12V, IOUT = 200mA
Step 1: Select the Transformer Turns Ratio.
NPS <
150V
−
V
IN(MAX) −
V
LEAKAGE
VOUT +VF
VLEAKAGE = Margin for transformer leakage spike = 30V
VF = Output diode forward voltage = ~0.3V
Example:
NPS <
150V
−
80V
−
30V
12V +0.3V
=3.3
The choice of transformer turns ratio is critical in deter-
mining output current capability of the converter. Table 4
shows the switch voltage stress and output current capa-
bility at different transformer turns ratio.
Table 4. Switch Voltage Stress and Output Current Capability
vs Turns Ratio
NPS
VSW(MAX) at
VIN(MAX) (V)
IOUT(MAX) at
VIN(MIN) (mA) DUTY CYCLE (%)
1:1 92.3 139 13 to 29
2:1 104.6 215 24 to 45
3:1 116.9 264 32 to 55
Since both NPS = 2 and NPS = 3 can meet the 200mA output
current requirement, NPS = 2 is chosen in this example
to allow more margin for transformer leakage inductance
voltage spike.
LT8303
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applicaTions inForMaTion
Step 2: Determine the Primary Inductance.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
LPRI ≥tOFF(MIN) •NPS •VOUT +VF
( )
ISW(MIN)
LPRI ≥tON(MIN) •VIN(MAX)
ISW(MIN)
tOFF(MIN) = 350ns
tON(MIN) = 160ns
ISW(MIN) = 105mA
Example:
LPRI ≥
350ns •2•(12V
+
0.3V)
105mA =82µH
LPRI ≥160ns •80V
105mA
=122µH
Most transformers specify primary inductance with a toler-
ance of ±20%. With other component tolerance considered,
choose a transformer with its primary inductance 40% to
60% larger than the minimum values calculated above.
LPRI = 150µH is then chosen in this example.
The transformer also needs to be rated for the correct
saturation current level across line and load conditions. A
saturation current rating larger than 620mA is necessary
to work with the LT8303. The PS15-111 from Sumida is
chosen as the flyback transformer.
Step 3: Choose the Output Diode.
Two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
as the average current requirement for the output diode.
A conservative metric is the maximum switch current limit
multiplied by the turns ratio,
IDIODE(MAX) = ISW(MAX) • NPS
Example:
IDIODE(MAX) = 1.07A
Next calculate reverse voltage requirement using maxi-
mum VIN:
VREVERSE =VOUT +
V
IN(MAX)
NPS
Example:
VREVERSE =12V +
72V
2
=48V
The DFLS2100 (2A, 100V diode) from Diodes Inc. is chosen.
Step 4: Choose the Output Capacitor.
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the equation below to
calculate the output capacitance:
COUT =LPRI •ISW2
2•VOUT •ΔVOUT
Example:
Design for output voltage ripple less than 1% of VOUT,
i.e., 120mV.
COUT =150µH •(0.535A)2
2•12V •0.12V
=14.9µF
Remember ceramic capacitors lose capacitance with ap-
plied voltage. The capacitance can drop to 40% of quoted
capacitance at the maximum voltage rating. So a 22µF,
25V rating X5R or X7R ceramic capacitor is chosen.
Step 5: Design Snubber Circuit.
The snubber circuit protects the power switch from leakage
inductance voltage spike. A DZ snubber is recommended
for this application because of lower leakage inductance
and larger voltage margin. The Zener and the diode need
to be selected.
LT8303
17
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The maximum Zener breakdown voltage is set according
to the maximum VIN:
VZENER(MAX) ≤ 150V – VIN(MAX)
Example:
VZENER(MAX) ≤ 150V – 80V = 70V
A 62V Zener with a maximum of 65V will provide optimal
protection and minimize power loss. So a 62V, 0.5W Zener
from Central Semiconductor (CMHZ5265B) is chosen.
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
VREVERSE > VSW(MAX)
VSW(MAX) = VIN(MAX) + VZENER(MAX)
Example:
VREVERSE > 144V
A 200V, 1A diode from Central Semiconductor
(CMMRIU-02) is chosen.
Step 6: Select the RFB Resistor.
Use the following equation to calculate the starting value
for RFB:
RFB =
N
PS
•(V
OUT +
V
F
)
100µA
Example:
RFB =
2•(12V
+
0.3V)
100µA =246k
Depending on the tolerance of standard resistor values,
the precise resistor value may not exist. For 1% standard
values, a 243k resistor in series with a 3.01k resistor
should be close enough. As discussed in the Application
Information section, the final RFB value should be adjusted
on the measured output voltage.
applicaTions inForMaTion
Step 7: Select the EN/UVLO Resistors.
Determine the amount of hysteresis required and calculate
R1 resistor value:
VIN(HYS) = 2.5µA • R1
Example:
Choose 2.5V of hysteresis,
R1 = 1M
Determine the UVLO thresholds and calculate R2 resistor
value:
V
IN(UVLO+)=
1.239V •(R1
+
R2)
R2
+2.5µA •R1
Example:
Set VIN UVLO rising threshold to 34.5V,
R2 = 49.9k
VIN(UVLO+) = 28.6V
VIN(UVLO–) = 25.7V
Step 8: Ensure minimum load.
The theoretical minimum load can be approximately
estimated as:
ILOAD(MIN) =150µH •(140mA)2•9kHz
2•12V
=1.1mA
Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the con-
verter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 1mA. In this example, a 12.1k resistor is selected
as the minimum load.
LT8303
18
8303fa
For more information www.linear.com/LT8303
Typical applicaTions
30V to 80VIN, 3.3VOUT Isolated Flyback Converter
Efficiency vs Load Current Output Load and Line Regulation
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (A)
0
0.2
0.4
0.6
0.8
1.0
1.2
40
50
60
70
80
90
100
EFFICIENCY (%)
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (A)
0
0.2
0.4
0.6
0.8
1.0
1.2
3.10
3.15
3.20
3.25
3.30
3.35
3.40
3.45
3.50
OUTPUT VOLTAGE (V)
8303 TA02c
LT8303
T1
8:1
D2
D1
Z1
RFB
SW
150µH 2.3µH
EN/UVLO
1M
4.7µF
100V
49.9k
VIN
VIN
30V TO 80V
VOUT
+
3.3V
4mA TO 0.9A (VIN
= 36V)
4mA TO 1A (VIN = 48V)
4mA TO 1.1A (VIN
= 72V)
VOUT–
GND
287k D1: CENTRAL CMMR1U-02
D2: DIODES SBR3U30P1-7
T1: SUMIDA PS15-108
Z1: CENTRAL CMHZ5265B
•
•
330µF
6.3V
8303 TA02a
LT8303
19
8303fa
For more information www.linear.com/LT8303
Typical applicaTions
30V to 80VIN, 5VOUT Isolated Flyback Converter
Efficiency vs Load Current Output Load and Line Regulation
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
100
200
300
400
500
600
700
800
900
40
50
60
70
80
90
100
EFFICIENCY (%)
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
100
200
300
400
500
600
700
800
900
4.7
4.8
4.9
5.0
5.1
5.2
5.3
OUTPUT VOLTAGE (V)
8303 TA03c
LT8303
T1
6:1
D2
RFB
SW
Z1
D1
150µH 4.2µH
EN/UVLO
1M
4.7µF
100V
49.9k
VIN
VIN
30V TO 80V
VOUT
+
5V
2.5mA TO 0.65A (VIN
= 36V)
2.5mA TO 0.73A (VIN
= 48V)
2.5mA TO 0.84A (VIN
= 72V)
VOUT–
GND
316k D1: CENTRAL CMMR1U-02
D2: DIODES SBR3U30P1-7
T1: SUMIDA PS15-109
Z1: CENTRAL CMHZ5265B
•
•
100µF
10V
8303 TA03a
LT8303
20
8303fa
For more information www.linear.com/LT8303
Efficiency vs Load Current Output Load and Line Regulation
Typical applicaTions
30V to 80VIN, 12VOUT Isolated Flyback Converter
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
40
80
120
160
200
240
280
320
11.4
11.6
11.8
12.0
12.2
12.4
12.6
OUTPUT VOLTAGE (V)
8303 TA04c
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
40
80
120
160
200
240
280
320
40
50
60
70
80
90
100
EFFICIENCY (%)
LT8303
T1
2:1
D2
Z1
D1
RFB
SW
150µH 37.5µH
EN/UVLO
1M
4.7µF
100V
49.9k
VIN
VIN
30V TO 80V
VOUT
+
12V
1mA TO 250mA (VIN
= 36V)
1mA TO 270mA (VIN
= 48V)
1mA TO 310mA (VIN
= 72V)
VOUT–
GND
249k D1: CENTRAL CMMR1U-02
D2: DIODES DFLS2100-7
T1: SUMIDA PS15-111
Z1: CENTRAL CMHZ5265B
•
•
22µF
25V
8303 TA04a
LT8303
21
8303fa
For more information www.linear.com/LT8303
Typical applicaTions
LT8303
T1
1:1
D2
Z1
D1
RFB
SW
150µH 150µH
EN/UVLO
1M
4.7µF
100V
49.9k
VIN
VIN
30V TO 80V
VOUT
+
24V
0.6mA TO 120mA (VIN
= 36V)
0.6mA TO 140mA (VIN
= 48V)
0.6mA TO 150mA (VIN
= 72V)
VOUT–
GND
249k D1: CENTRAL CMMR1U-02
D2: DIODES DFLS1200-7
T1: SUMIDA PS15-112
Z1: CENTRAL CMHZ5265B
•
•
22µF
50V
8303 TA05a
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
20
40
60
80
100
120
140
160
40
50
60
70
80
90
100
EFFICIENCY (%)
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
20
40
60
80
100
120
140
160
22.8
23.2
23.6
24.0
24.4
24.8
25.2
OUTPUT VOLTAGE (V)
8303 TA05c
Efficiency vs Load Current Output Load and Line Regulation
30V to 80VIN, 24VOUT Isolated Flyback Converter
LT8303
22
8303fa
For more information www.linear.com/LT8303
package DescripTion
Please refer to http://www.linear.com/product/LT8303#packaging for the most recent package drawings.
1.50 – 1.75
(NOTE 4)
2.80 BSC
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
DATUM ‘A’
0.09 – 0.20
(NOTE 3) S5 TSOT-23 0302
PIN ONE
2.90 BSC
(NOTE 4)
0.95 BSC
1.90 BSC
0.80 – 0.90
1.00 MAX 0.01 – 0.10
0.20 BSC
0.30 – 0.50 REF
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3.85 MAX
0.62
MAX
0.95
REF
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
1.4 MIN
2.62 REF
1.22 REF
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635)
LT8303
23
8303fa
For more information www.linear.com/LT8303
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision hisTory
REV DATE DESCRIPTION PAGE NUMBER
A 1/17 Added H-grade version 2, 3
LT8303
24
8303fa
For more information www.linear.com/LT8303
LINEAR TECHNOLOGY CORPORATION 2016
LT REV A 0117 • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT8303
relaTeD parTs
Typical applicaTion
30V to 80VIN, 48VOUT Isolated Flyback Converter
PART NUMBER DESCRIPTION COMMENTS
LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23
LT8304 100VIN Micropower Isolated Flyback Converter with 150V/2A Switch Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E
LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23
LT8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6mA Switch Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E
LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead
TSOT-23
LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12)
LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components
LT3757/LT3759/
LT3758
40V/100V Flyback/Boost Controller Universal Controllers with Small Package and Powerful
Gate Drive
LT3957/LT3958 40V/80V Boost/Flyback Converter Monolithic with Integrated 5A/3.3A Switch
LT C3803/LT C3803-3/
LT C3803-5
200kHz/300kHz Flyback Controller in SOT-23 VIN and VOUT Limited Only by External Components
LT C3805/LT C3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited Only by External Components
LT8303
T1
1:2
D2
Z1
D1
RFB
SW
150µH 600µH
EN/UVLO
1M
4.7µF
100V
49.9k
VIN
VIN
30V TO 80V
VOUT+
48V
0.3mA TO 60mA (VIN
= 36V)
0.3mA TO 70mA (VIN
= 48V)
0.3mA TO 75mA (VIN
= 72V)
VOUT–
GND
243k D1: CENTRAL CMMR1U-02
D2: DIODES SBR1U400P1-7
T1: SUMIDA PS15-113
Z1: CENTRAL CMHZ5265B
•
•
4.7µF
100V
8303 TA06a
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
10
20
30
40
50
60
70
80
40
50
60
70
80
90
100
EFFICIENCY (%)
V
IN
= 36V
V
IN
= 48V
V
IN
= 72V
LOAD CURRENT (mA)
0
10
20
30
40
50
60
70
80
45.6
46.4
47.2
48.0
48.8
49.6
50.4
OUTPUT VOLTAGE (V)
8303 TA06c
Efficiency vs Load Current Output Load and Line Regulation