LM2574N-12G - ON Semiconductor L.L.C.

August, 2006 − Rev. 8 1Publication Order Number:

LM2574/D

LM2574, NCV2574

0.5 A, Adjustable Output

Voltage, Step−Down

Switching Regulator

The LM2574 series of regulators are monolithic integrated circuits

ideally suited for easy and convenient design of a step−down

switching regulator (buck converter). All circuits of this series are

capable of driving a 0.5 A load with excellent line and load regulation.

These devices are available in fixed output voltages of 3.3 V, 5.0 V,

12 V, 15 V, and an adjustable output version.

These regulators were designed to minimize the number of external

components to simplify the power supply design. Standard series of

inductors optimized for use with the LM2574 are offered by several

different inductor manufacturers.

Since the LM2574 converter is a switch−mode power supply, its

efficiency is significantly higher in comparison with popular

three−terminal linear regulators, especially with higher input voltages.

In most cases, the power dissipated by the LM2574 regulator is so low,

that the copper traces on the printed circuit board are normally the only

heatsink needed and no additional heatsinking is required.

The LM2574 features include a guaranteed ±4% tolerance on output

voltage within specified input voltages and output load conditions, and

±10% on the oscillator frequency (±2% over 0°C to +125°C). External

shutdown is included, featuring 60 mA (typical) standby current. The

output switch includes cycle−by−cycle current limiting, as well as

thermal shutdown for full protection under fault conditions.

Features

•3.3 V, 5.0 V, 12 V, 15 V, and Adjustable Output Versions

•Adjustable Version Output Voltage Range, 1.23 to 37 V ±4% max

over Line and Load Conditions

•Guaranteed 0.5 A Output Current

•Wide Input Voltage Range: 4.75 to 40 V

•Requires Only 4 External Components

•52 kHz Fixed Frequency Internal Oscillator

•TTL Shutdown Capability, Low Power Standby Mode

•High Efficiency

•Uses Readily Available Standard Inductors

•Thermal Shutdown and Current Limit Protection

•NCV Prefix for Automotive and Other Applications Requiring Site

and Control Changes

•Pb−Free Packages are Available*

Applications

•Simple and High−Efficiency Step−Down (Buck) Regulators

•Efficient Pre−regulator for Linear Regulators

•On−Card Switching Regulators

•Positive to Negative Converters (Buck−Boost)

•Negative Step−Up Converters

•Power Supply for Battery Chargers

*For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting

Techniques Reference Manual, SOLDERRM/D.

16 SO−16 WB

DW SUFFIX

CASE 751G

PIN CONNECTIONS

(Top View)

Pwr Gnd

ON/OFF

Sig Gnd

Output

Vin

* No internal connection, but should be soldered to

* PC board for best heat transfer.

(Top View)

Sig Gnd

ON/OFF

Pwr Gnd

Output

Vin

See detailed ordering and shipping information in the package

dimensions section on page 24 of this data sheet.

ORDERING INFORMATION

See general marking information in the device marking

section on page 24 of this data sheet.

DEVICE MARKING INFORMATION

PDIP−8

N SUFFIX

CASE 626

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Figure 1. Block Diagram and Typical Application

7.0 − 40 V

Unregulated

DC Input

330 mH

Pwr

Gnd

+Vin

Cin

22 mF4ON

/OFF3

Output

Feedback

1N5819 Cout

220 mF

Typical Application (Fixed Output Voltage Versions)

Representative Block Diagram and Typical Application

Unregulated

DC Input

+Vin

Cout

Feedback

Cin

1.0 k Output

Pwr Gnd

ON/OFF

Reset

Latch

Thermal

Shutdown

52 kHz

Oscillator

1.235 V

Band−Gap

Reference

Freq

Shift

18 kHz

Comparator

Fixed Gain

Error Amplifier

Current

Limit

Driver

1.0 Amp

Switch

ON/OFF

3.1 V Internal

Regulator

Vout

Load

Output

Voltage Versions

3.3 V

5.0 V

12 V

15 V

(W)

1.7 k

3.1 k

8.84 k

11.3 k

For adjustable version

R1 = open, R2 = 0 W

LM2574

5.0 V Regulated

Output 0.5 A Load

Sig

Gnd

Sig Gnd

(12) (14)

(3)

(4) (6) (5)

(5)

(12)

(3)

(4)

(14)

(6)

NOTE: Pin numbers in ( ) are for the SO−16W package.

ABSOLUTE MAXIMUM RATINGS (Absolute Maximum Ratings indicate limits beyond which damage to the device may occur).

Rating Symbol Value Unit

Maximum Supply Voltage Vin 45 V

ON/OFF Pin Input Voltage −−0.3 V ≤ V ≤ +Vin V

Output Voltage to Ground (Steady State) − −1.0 V

DW Suffix, Plastic Package Case 751G

Max Power Dissipation PDInternally Limited W

Thermal Resistance, Junction−to−Air RqJA 145 °C/W

N Suffix, Plastic Package Case 626

Max Power Dissipation PDInternally Limited W

Thermal Resistance, Junction−to−Ambient RqJA 100 °C/W

Thermal Resistance, Junction−to−Case RqJC 5.0 °C/W

Storage Temperature Range Tstg −65°C to +150°C°C

Minimum ESD Rating − 2.0 kV

(Human Body Model: C = 100 pF, R = 1.5 kW)

Lead Temperature (Soldering, 10 seconds) − 260 °C

Maximum Junction Temperature TJ150 °C

Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit

values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied,

damage may occur and reliability may be affected.

NOTE: ESD data available upon request.

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OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee

specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics).

Rating Symbol Value Unit

Operating Junction Temperature Range TJ−40 to +125 °C

Supply Voltage Vin 40 V

SYSTEM PARAMETERS ([Note 1] Test Circuit Figure 16)

ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable

version, Vin = 25 V for the 12 V version, Vin = 30 V for the 15 V version. ILoad = 100 mA. For typical values TJ = 25°C, for min/max values

TJ is the operating junction temperature range that applies [Note 2], unless otherwise noted).

Characteristic Symbol Min Typ Max Unit

LM2574−3.3 ([Note 1] Test Circuit Figure 16)

Output Voltage (Vin = 12 V, ILoad = 100 mA, TJ = 25°C) Vout 3.234 3.3 3.366 V

Output Voltage (4.75 V ≤ Vin ≤ 40 V, 0.1 A ≤ ILoad ≤ 0.5 A) Vout V

TJ = 25°C 3.168 3.3 3.432

TJ = −40 to +125°C 3.135 − 3.465

Efficiency (Vin = 12 V, ILoad = 0.5 A) η− 72 − %

LM2574−5 ([Note 1] Test Circuit Figure 16)

Output Voltage (Vin = 12 V, ILoad = 100 mA, TJ = 25°C) Vout 4.9 5.0 5.1 V

Output Voltage (7.0 V ≤ Vin ≤ 40 V, 0.1 A ≤ ILoad ≤ 0.5 A) Vout V

TJ = 25°C4.8 5.0 5.2

TJ = −40 to +125°C4.75 5.25

Efficiency (Vin = 12 V, ILoad = 0.5 A) η− 77 − %

LM2574−12 ([Note 1] Test Circuit Figure 16)

Output Voltage (Vin = 25 V, ILoad = 100 mA, TJ = 25°C) Vout 11.76 10 12.24 V

Output Voltage (15 V ≤ Vin ≤ 40 V, 0.1 A ≤ ILoad ≤ 0.5 A) Vout V

TJ = 25°C11.52 12 12.48

TJ = −40 to +125°C11.4 − 12.6

Efficiency (Vin = 15 V, ILoad = 0.5 A) η− 88 − %

LM2574−15 ([Note 1] Test Circuit Figure 16)

Output Voltage (Vin = 30 V, ILoad = 100 mA, TJ = 25°C) Vout 14.7 15 15.3 V

Output Voltage (18 V < Vin < 40 V, 0.1 A < ILoad < 0.5 A) Vout V

TJ = 25°C 14.4 15 15.6

TJ = −40 to +125°C 14.25 15.75

Efficiency (Vin = 18 V, ILoad = 0.5 A) η− 88 − %

LM2574 ADJUSTABLE VERSION ([Note 1] Test Circuit Figure 16)

Feedback Voltage Vin = 12 V, ILoad = 100 mA, Vout = 5.0 V, TJ = 25°C VFB 1.217 1.23 1.243 V

Feedback Voltage 7.0 V ≤ Vin ≤ 40 V, 0.1 A ≤ ILoad ≤ 0.5 A, Vout = 5.0

VVFBT V

TJ = 25°C 1.193 1.23 1.267

TJ = −40 to +125°C 1.18 1.28

Efficiency (Vin = 12 V, ILoad = 0.5 A, Vout = 5.0 V) η− 77 − %

1. External components such as the catch diode, inductor, input and output capacitors can affect the switching regulator system performance.

When the LM2574 is used as shown in the Figure 16 test circuit, the system performance will be as shown in the system parameters section

of the Electrical Characteristics.

2. Tested junction temperature range for the LM2574, NCV2574: Tlow = −40°C Thigh = +125°C.

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SYSTEM PARAMETERS ([Note 3] Test Circuit Figure 16)

ELECTRICAL CHARACTERISTICS (continued) (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and

Adjustable version, Vin = 25 V for the 12 V version, Vin = 30 V for the 15 V version. ILoad = 100 mA. For typical values TJ = 25°C, for

min/max values TJ is the operating junction temperature range that applies [Note 4], unless otherwise noted).

Characteristic Symbol Min Typ Max Unit

ALL OUTPUT VOLTAGE VERSIONS

Feedback Bias Current Vout = 5.0 V (Adjustable Version Only) IbnA

TJ = 25°C − 25 100

TJ = −40 to +125°C − − 200

Oscillator Frequency (Note 5) fOkHz

TJ = 25°C − 52 −

TJ = 0 to +125°C 47 52 58

TJ = −40 to +125°C 42 − 63

Saturation Voltage (Iout = 0.5 A, [Note 6]) Vsat V

TJ = 25°C − 1.0 1.2

TJ = −40 to +125°C − − 1.4

Max Duty Cycle (“on”) (Note 7) DC 93 98 − %

Current Limit Peak Current (Notes 5 and 6) ICL A

TJ = 25°C 0.7 1.0 1.6

TJ = −40 to +125°C 0.65 − 1.8

Output Leakage Current (Notes 8 and 9), TJ = 25°C ILmA

Output = 0 V − 0.6 2.0

Output = − 1.0 V − 10 30

Quiescent Current (Note 8) IQmA

TJ = 25°C − 5.0 9.0

TJ = −40 to +125°C − − 11

Standby Quiescent Current (ON/OFF Pin = 5.0 V (“off”)) Istby mA

TJ = 25°C − 60 200

TJ = −40 to +125°C − − 400

ON/OFF Pin Logic Input Level V

Vout = 0 V VIH

TJ = 25°C 2.2 1.4 −

TJ = −40 to +125°C 2.4 − −

Nominal Output Voltage VIL

TJ = 25°C − 1.2 1.0

TJ = −40 to +125°C − − 0.8

ON/OFF Pin Input Current mA

ON/OFF Pin = 5.0 V (“off”), TJ = 25°C IIH − 15 30

ON/OFF Pin = 0 V (“on”), TJ = 25°C IIL − 0 5.0

3. External components such as the catch diode, inductor, input and output capacitors can affect the switching regulator system performance.

When the LM2574 is used as shown in the Figure 16 test circuit, the system performance will be as shown in the system parameters section

of the Electrical Characteristics.

4. Tested junction temperature range for the LM2574, NCV2574: Tlow = −40°C Thigh = +125°C.

5. The oscillator frequency reduces to approximately 18 kHz in the event of an output short or an overload which causes the regulated output

voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average power dissipation of th e

IC by lowering the minimum duty cycle from 5% down to approximately 2%.

6. Output (Pin 2) sourcing current. No diode, inductor or capacitor connected to the output pin.

7. Feedback (Pin 4) removed from output and connected to 0 V.

8. Feedback (Pin 4) removed from output and connected to 12 V for the Adjustable, 3.3 V, and 5.0 V versions, and 25 V for the 12 V and 15 V

versions, to force the output transistor OFF.

9. Vin = 40 V.

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Istby, STANDBY QUIESCENT CURRENT ( A)μ

IQ, QUIESCENT CURRENT (mA)

out, OUTPUT VOLTAGE CHANGE (%)

TJ, JUNCTION TEMPERATURE (°C)

IO, OUTPUT CURRENT (A)

TJ, JUNCTION TEMPERATURE (°C)

Vin, INPUT VOLTAGE (V)

INPUT − OUTPUT DIFFERENTIAL (V)

TJ, JUNCTION TEMPERATURE (°C)

out, OUTPUT VOLTAGE CHANGE (%)

Figure 2. Normalized Output Voltage

TJ, JUNCTION TEMPERATURE (°C)

Figure 3. Line Regulation

Vin = 20 V

ILoad = 100 mA

Normalized at TJ = 25°C

Figure 4. Dropout Voltage Figure 5. Current Limit

Figure 6. Quiescent Current Figure 7. Standby Quiescent Current

ILoad = 100 mA

TJ = 25°C

3.3 V, 5.0 V and ADJ

12 V and 15 V

Vin = 25 V

ILoad = 100 mA

ILoad = 500 A

Vin = 12 V

Vin = 40 V

L = 300 mH

ILoad = 500 mA

ILoad = 100 mA

Vout = 5.0 V

Measured at

Ground Pin

TJ = 25°C

VON/OFF = 5.0 V

TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16)

1.0

0.8

0.6

0.4

0.2

−0.2

−0.4

−0.6

−0.8

−1.0

1.4

1.2

1.0

0.8

0.6

0.4

0.2

−0.2

−0.4

−0.6

2.0

1.5

1.0

0.5

1.4

1.3

1.2

1.1

1.0

0.9

0.8

0.7

8.0

6.0

4.0

200

180

160

140

120

100

1251007560250−25−50 403530252015105.00

1251007560250−25−50 1251007560250−25−50

403530252015105.00 1251007560250−25−50

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, INPUT VOLTAGE (V)Vin

Vsat, SATURATION VOLTAGE (V)

IFB, FEEDBACK PIN CURRENT (nA)

5 ms/DIV

TJ, JUNCTION TEMPERATURE (°C)

SWITCH CURRENT (A)

5 ms/DIV

TJ, JUNCTION TEMPERATURE (°C)

NORMALIZED FREQUENCY (%)

Figure 8. Oscillator Frequency

TJ, JUNCTION TEMPERATURE (°C)

Figure 9. Switch Saturation Voltage

Figure 10. Minimum Operating Voltage Figure 11. Feedback Pin Current

Figure 12. Continuous Mode Switching Waveforms

out

= 5.0 V, 500 mA Load Current, L = 330 mHFigure 13. Discontinuous Mode Switching Waveform

out

= 5.0 V, 100 mA Load Current, L = 100 mH

Vin = 1.23 V

ILoad = 100 mA

Adjustable Version Only

Vin = 12 V

Normalized at 25°C

Adjustable Version Only

A: Output Pin Voltage, 10 V/DIV.

B: Inductor Current, 0.2 A/DIV.

C: Output Ripple Voltage, 20 mV/DIV, AC−Coupled

A: Output Pin Voltage, 10 V/DIV.

B: Inductor Current, 0.2 A/DIV.

C: Output Ripple Voltage, 20 mV/DIV, AC−Coupled

TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) (continued)

8.0

6.0

4.0

2.0

−2.0

−4.0

−6.0

−8.0

1.3

1.2

1.1

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

5.0

4.5

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

100

−20

−40

−60

−80

−100

1251007550250−25−50 0 0.1 0.2 0.3 0.4 0.5

1251007550250−25−50 1251007550250−25−50

20 V

10 V

0.6 A

0.4 A

0.2 A

20 mV

20 V

10 V

0.4 A

0.2 A

20 mV

−40°C

25°C

125°C

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200 ms/DIV200 ms/DIV

Figure 14. 500 mA Load Transient Response for

Continuous Mode Operation, L = 330 mH, C

out

= 300 mFFigure 15. 250 mA Load Transient Response for

Discontinuous Mode Operation, L = 68 mH, C

out

= 470 m

A: Output Voltage, 50 mV/DIV, AC Coupled

B: 100 mA to 500 mA Load Pulse

A: Output Voltage, 50 mV/DIV, AC Coupled

B: 50 mA to 250 mA Load Pulse

TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) (continued)

50 mV

500 mA

50 mV

200 mA

100 mA

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Figure 16. Test Circuit and Layout Guidelines

1N5819

330 mH

Output

Feedback

Cout

220 mF

Cin

22 mF

LM2574

Fixed Output

34ON

/OFFPwr

Gnd

Vin

Load

Vout

1N5819

330 mH

Output

Feedback

Cout

220 mF

Cin

22 mF

LM2574

Adjustable

Vin

Load

Vout

5.0 V

Fixed Output Voltage Versions

Adjustable Output Voltage Versions

Vout +Vrefǒ1.0 )R2

R1Ǔ

R2 +R1ǒVout

Vref

–1.0Ǔ

Where Vref = 1.23 V, R1

between 1.0 kW and 5.0 kW

6.12 k

2.0 k

7.0 − 40 V

Unregulated

DC Input

2 Sig

Gnd

34ON/OFFPwr

Gnd

2 Sig

Gnd

7.0 V − 40 V

Unregulated

DC Input

Cin −22 mF, 60 V, Aluminium Electrolytic

Cout − 220 mF, 25 V, Aluminium Electrolytic

D1 − Schottky, 1N5819

L1 − 330 mH, (For 5.0 Vin, 3.3 Vout, use 100 mH)

R1 − 2.0 k, 0.1%

R2 − 6.12 k, 0.1%

NOTE: Pin numbers in ( ) are for the SO−16W package.

(12)

(3)

(6) (4)

(5)

(14)

(12)

(3)

(14)

(6) (4)

(5)

PCB LAYOUT GUIDELINES

As in any switching regulator, the layout of the printed

circuit board is very important. Rapidly switching currents

associated with wiring inductance, stray capacitance and

parasitic inductance of the printed circuit board traces can

generate voltage transients which can generate

electromagnetic interferences (EMI) and affect the desired

operation. As indicated in the Figure 16, to minimize

inductance and ground loops, the length of the leads

indicated by heavy lines should be kept as short as possible.

For best results, single−point grounding (as indicated) or

ground plane construction should be used.

On the other hand, the PCB area connected to the Pin 7

(emitter of the internal switch) of the LM2574 should be

kept to a minimum in order to minimize coupling to sensitive

circuitry.

Another sensitive part of the circuit is the feedback. It is

important to keep the sensitive feedback wiring short. To

assure this, physically locate the programming resistors near

to the regulator, when using the adjustable version of the

LM2574 regulator.

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PIN FUNCTION DESCRIPTION

Symbol Description (Refer to Figure 1)

SO−16W PDIP−8

12 5 Vin This pin is the positive input supply for the LM2574 step−down switching regulator. In order to

minimize voltage transients and to supply the switching currents needed by the regulator, a

suitable input bypass capacitor must be present (Cin in Figure 1).

14 7 Output This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is

typically 1.0 V. It should be kept in mind that the PCB area connected to this pin should be kept

to a minimum in order to minimize coupling to sensitive circuitry.

4 2 Sig Gnd Circuit signal ground pin. See the information about the printed circuit board layout.

6 4 Pwr Gnd Circuit power ground pin. See the information about the printed circuit board layout.

3 1 Feedback This pin senses regulated output voltage to complete the feedback loop. The signal is divided by

the internal resistor divider network R2, R1 and applied to the non−inverting input of the internal

error amplifier. In the Adjustable version of the LM2574 switching regulator , this pin is the direct

input of the error amplifier and the resistor network R2, R1 is connected externally to allow

programming of the output voltage.

5 3 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the

total input supply current to approximately 80 mA. The input threshold voltage is typically 1.5 V.

Applying a voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this

pin is lower than 1.5 V or if this pin is left open, the regulator will be in the “on” condition.

DESIGN PROCEDURE

Buck Converter Basics

The LM2574 is a “Buck” or Step−Down Converter which

is the most elementary forward−mode converter. Its basic

schematic can be seen in Figure 17.

The operation of this regulator topology has two distinct

time periods. The first one occurs when the series switch is

on, the input voltage is connected to the input of the inductor.

The output of the inductor is the output voltage, and the

rectifier (or catch diode) is reverse biased. During this

period, since there is a constant voltage source connected

across the inductor, the inductor current begins to linearly

ramp upwards, as described by the following equation:

IL(on) +ǒVin –V

outǓton

During this “on” period, energy is stored within the core

material in the form of magnetic flux. If the inductor is

properly designed, there is sufficient energy stored to carry

the requirements of the load during the “off” period.

Figure 17. Basic Buck Converter

DVin RLoad

Cout

Power

Switch

The next period is the “off” period of the power switch.

When the power switch turns off, the voltage across the

inductor reverses its polarity and is clamped at one diode

voltage drop below ground by the catch diode. Current now

flows through the catch diode thus maintaining the load

current loop. This removes the stored energy from the

inductor. The inductor current during this time is:

IL(off) +ǒVout –V

DǓtoff

This period ends when the power switch is once again

turned on. Regulation of the converter is accomplished by

varying the duty cycle of the power switch. It is possible to

describe the duty cycle as follows:

d+ton

T, where T is the period of switching.

For the buck converter with ideal components, the duty

cycle can also be described as:

d+Vout

Vin

Figure 18 shows the buck converter idealized waveforms

of the catch diode voltage and the inductor current.

Figure 18. Buck Converter Idealized Waveforms

Power

Switch

Power

Switch

Off

Power

Switch

Off

Power

Switch

Power

Switch

Von(SW)

VD(FWD)

Time

ILoad(AV)

Imin

Ipk

Diode Diode

Power

Switch

Diode VoltageInductor Current

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Procedure (Fixed Output Voltage Version) In order to simplify the switching regulator design, a step−by−step design

procedure and example is provided.

Procedure Example

Given Parameters:

Vout = Regulated Output Voltage (3.3 V, 5.0 V, 12 V or 15 V)

Vin(max) = Maximum Input Voltage

ILoad(max) = Maximum Load Current

Given Parameters:

Vout = 5.0 V

Vin(max) = 15 V

ILoad(max) = 0.4 A

1. Controller IC Selection

According to the required input voltage, output voltage and

current, select the appropriate type of the controller IC output

voltage version.

1. Controller IC Selection

According to the required input voltage, output voltage,

current polarity and current value, use the LM2574−5

controller IC.

2. Input Capacitor Selection (Cin)

To prevent large voltage transients from appearing at the input

and for stable operation of the converter, an aluminium or

tantalum electrolytic bypass capacitor is needed between the

input pin +Vin and ground pin Gnd. This capacitor should be

located close to the IC using short leads. This capacitor should

have a low ESR (Equivalent Series Resistance) value.

2. Input Capacitor Selection (Cin)

A 22 mF, 25 V aluminium electrolytic capacitor located near

to the input and ground pins provides sufficient bypassing.

3. Catch Diode Selection (D1)

A.Since the diode maximum peak current exceeds the

regulator maximum load current, the catch diode current

rating must be at least 1.2 times greater than the maximum

load current. For a robust design the diode should have a

current rating equal to the maximum current limit of the

LM2574 to be able to withstand a continuous output short.

B.The reverse voltage rating of the diode should be at least

1.25 times the maximum input voltage.

3. Catch Diode Selection (D1)

A.For this example the current rating of the diode is 1.0 A.

B.Use a 20 V 1N5817 Schottky diode, or any of the

suggested fast recovery diodes shown in Table 1.

4. Inductor Selection (L1)

A.According to the required working conditions, select the

correct inductor value using the selection guide from

Figures 19 to 23.

B.From the appropriate inductor selection guide, identify the

inductance region intersected by the Maximum Input Voltage

line and the Maximum Load Current line. Each region is

identified by an inductance value and an inductor code.

C.Select an appropriate inductor from the several different

manufacturers part numbers listed in Table 2. The designer

must realize that the inductor current rating must be higher

than the maximum peak current flowing through the inductor.

This maximum peak current can be calculated as follows:

where ton is the “on” time of the power switch and

For additional information about the inductor, see the inductor

section in the “EXTERNAL COMPONENTS” section of this

data sheet.

Ip(max)+ILoad(max))ǒVin *VoutǓton

ton +Vout

Vin x1.0

fosc

4. Inductor Selection (L1)

A.Use the inductor selection guide shown in Figure 20.

B.From the selection guide, the inductance area

intersected by the 15 V line and 0.4 A line is 330.

C.Inductor value required is 330 mH. From Table 2, choose

an inductor from any of the listed manufacturers.

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Procedure (Fixed Output Voltage Version) (continued) In order to simplify the switching regulator design, a step−by−step

design procedure and example is provided.

Procedure Example

5. Output Capacitor Selection (Cout)

A.Since the LM2574 is a forward−mode switching regulator

with voltage mode control, its open loop 2−pole−1−zero

frequency characteristic has the dominant pole−pair

determined by the output capacitor and inductor values. For

stable operation and an acceptable ripple voltage,

(approximately 1% of the output voltage) a value between

100 mF and 470 mF is recommended.

B.Due to the fact that the higher voltage electrolytic capacitors

generally have lower ESR (Equivalent Series Resistance)

numbers, the output capacitor’s voltage rating should be at

least 1.5 times greater than the output voltage. For a 5.0 V

regulator, a rating at least 8.0 V is appropriate, and a 10 V

or 16 V rating is recommended.

5. Output Capacitor Selection (Cout)

A.Cout = 100 mF to 470 mF standard aluminium electrolytic.

B.Capacitor voltage rating = 20 V.

Procedure (Adjustable Output Version: LM2574−ADJ)

Procedure Example

Given Parameters:

Vout = Regulated Output Voltage

Vin(max) = Maximum DC Input Voltage

ILoad(max) = Maximum Load Current

Given Parameters:

Vout = 24 V

Vin(max) = 40 V

ILoad(max) = 0.4 A

1. Programming Output Voltage

To select the right programming resistor R1 and R2 value (see

Figure 2) use the following formula:

where Vref = 1.23 V

Resistor R1 can be between 1.0 kW and 5.0 kW. (For best

temperature coefficient and stability with time, use 1% metal

film resistors).

Vout +Vrefǒ1.0 )R2

R1Ǔ

R2 +R1ǒVout

Vref *1.0Ǔ

1. Programming Output Voltage (selecting R1 and R2)

Select R1 and R2 :

Vout = 1.23 Select R1 = 1.0 kW

R2 = 18.51 kW, choose a 18.7 kW metal film resistor.

ǒ1.0 )R2

R1Ǔ

R2 +R1ǒVout

Vref *1.0Ǔ+1.0 kǒ10 V

1.23 V *1.0Ǔ

2. Input Capacitor Selection (Cin)

To prevent large voltage transients from appearing at the input

and for stable operation of the converter, an aluminium or

tantalum electrolytic bypass capacitor is needed between the

input pin +Vin and ground pin Gnd. This capacitor should be

located close to the IC using short leads. This capacitor should

have a low ESR (Equivalent Series Resistance) value.

For additional information see input capacitor section in the

“EXTERNAL COMPONENTS” section of this data sheet.

2. Input Capacitor Selection (Cin)

A 22 mF aluminium electrolytic capacitor located near the

input and ground pin provides sufficient bypassing.

3. Catch Diode Selection (D1)

A.Since the diode maximum peak current exceeds the

regulator maximum load current the catch diode current

rating must be at least 1.2 times greater than the maximum

load current. For a robust design, the diode should have a

current rating equal to the maximum current limit of the

LM2574 to be able to withstand a continuous output short.

B.The reverse voltage rating of the diode should be at least

1.25 times the maximum input voltage.

3. Catch Diode Selection (D1)

A. For this example, a 1.0 A current rating is adequate.

B.Use a 50 V MBR150 Schottky diode or any suggested

fast recovery diodes in Table 1.

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Procedure (Adjustable Output Version: LM2574−ADJ)

Procedure Example

4. Inductor Selection (L1)

A.Use the following formula to calculate the inductor Volt x

microsecond [V x ms] constant:

B.Match the calculated E x T value with the corresponding

number on the vertical axis of the Inductor Value Selection

Guide shown in Figure 23. This E x T constant is a measure

of the energy handling capability of an inductor and is

dependent upon the type of core, the core area, the number

of turns, and the duty cycle.

C.Next step is to identify the inductance region intersected by

the E x T value and the maximum load current value on the

horizontal axis shown in Figure 27.

D.From the inductor code, identify the inductor value. Then

select an appropriate inductor from Table 2. The inductor

chosen must be rated for a switching frequency of 52 kHz

and for a current rating of 1.15 x ILoad. The inductor current

rating can also be determined by calculating the inductor

peak current:

where ton is the “on” time of the power switch and

For additional information about the inductor, see the inductor

section in the “External Components” section of this data

sheet.

ton +Vout

Vin x1.0

fosc

Ip(max)+ILoad(max))ǒVin *VoutǓton

ExT+(Vin *Vout)Vout

Vin x106

F[Hz]ƪVxmsƫ

4. Inductor Selection (L1)

C.ILoad(max) = 0.4 A

Inductance Region = 1000

D.Proper inductor value = 1000 mH

Choose the inductor from Table 2.

ExT+(40 *24) x 24

40 x1000

52 +105ƪVxmsƫ

ExT+185ƪVxmsƫ

Calculate E x T ƪVxmsƫconstant :

5. Output Capacitor Selection (Cout)

A.Since the LM2574 is a forward−mode switching regulator with

voltage mode control, its open loop 2−pole−1−zero frequency

characteristic has the dominant pole−pair determined by the

output capacitor and inductor values.

For stable operation, the capacitor must satisfy the following

requirement:

B.Capacitor values between 10 mF and 2000 mF will satisfy the

loop requirements for stable operation. To achieve an

acceptable output ripple voltage and transient response, the

output capacitor may need to be several times larger than the

above formula yields.

C.Due to the fact that the higher voltage electrolytic capacitors

generally have lower ESR (Equivalent Series Resistance)

numbers, the output capacitor’s voltage rating should be at

least 1.5 times greater than the output voltage. For a 5.0 V

regulator, a rating of at least 8.0 V is appropriate, and a 10 V

or 16V rating is recommended.

Cout w13,300 Vin(max)

Vout xL

ƪmHƫƪmFƫ

5. Output Capacitor Selection (Cout)

To achieve an acceptable ripple voltage, select

Cout = 100 mF electrolytic capacitor.

Cout w13,300 x 40

24 x 1000 +22.2 mF

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ET, VOLTAGE TIME (V s)μ

Vin , MAXIMUM INPUT VOLTAGE (V) Vin , MAXIMUM INPUT VOLTAGE (V)

Vin , MAXIMUM INPUT VOLTAGE (V)Vin , MAXIMUM INPUT VOLTAGE (V)

IL, MAXIMUM LOAD CURRENT (A)

Figure 19. LM2574−3.3

IL, MAXIMUM LOAD CURRENT (A)

Figure 20. LM2574−5

680

Figure 21. LM2574−12 Figure 22. LM2574−15

Figure 23. LM2574−ADJ

150

470

220

100

330

1000

330

680

470

150

220

2200

470

1500

1000

330

680

220

2200

470

1500

1000

680

2200

470

1500

1000

330

680

220

150

100

LM2574 Series Buck Regulator Design Procedures (continued)

Indicator Value Selection Guide (For Continuous Mode Operation)

9.0

8.0

7.0

6.0

5.0

9.0

8.0

7.0

250

200

150

100

0.50.40.30.20.150.1 0.50.40.30.20.150.1

0.50.40.30.20.150.1

330

220

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Table 1. Diode Selection Guide gives an overview about through−hole diodes for

an effective design. Device listed in bold are available from ON Semiconductor

1.0 Amp Diodes

Schottky Fast Recovery

20 V 1N5817

MBR120P

MUR110

(rated to 100 V)

30 V 1N5818

MBR130P

40 V 1N5819

MBR140P

50 V MBR150

60 V MBR160

Table 2. Inductor Selection Guide

Inductor

Value Pulse Engineering Tech 39 Renco NPI

68 mH*55 258 SN RL−1284−68 NP5915

100 mH*55 308 SN RL−1284−100 NP5916

150 mH52625 55 356 SN RL−1284−150 NP5917

220 mH52626 55 406 SN RL−1284−220 NP5918/5919

330 mH52627 55 454 SN RL−1284−330 NP5920/5921

470 mH52628 * RL−1284−470 NP5922

680 mH52629 55 504 SN RL−1284−680 NP5923

1000 mH52631 55 554 SN RL−1284−1000 *

1500 mH* * RL−1284−1500 *

2200 mH* * RL−1284−2200 *

* : Contact Manufacturer

Table 3. Example of Several Inductor Manufacturers Phone/Fax Numbers

Pulse Engineering Inc. Phone

Fax + 1−619−674−8100

+ 1−619−674−8262

Pulse Engineering Inc. Europe Phone

Fax + 353−9324−107

+ 353−9324−459

Renco Electronics Inc. Phone

Fax + 1−516−645−5828

+ 1−516−586−5562

Tech 39 Phone

Fax + 33−1−4115−1681

+ 33−1−4709−5051

NPI/APC Phone

Fax + 44−634−290−588

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EXTERNAL COMPONENTS

Input Capacitor (Cin)

The Input Capacitor Should Have a Low ESR

For stable operation of the switch mode converter a low

ESR (Equivalent Series Resistance) aluminium or solid

tantalum bypass capacitor is needed between the input pin

and the ground pin, to prevent large voltage transients from

appearing at the input. It must be located near the regulator

and use short leads. With most electrolytic capacitors, the

capacitance value decreases and the ESR increases with

lower temperatures. For reliable operation in temperatures

below −25°C larger values of the input capacitor may be

needed. Also paralleling a ceramic or solid tantalum

capacitor will increase the regulator stability at cold

temperatures.

RMS Current Rating of Cin

The important parameter of the input capacitor is the RMS

current rating. Capacitors that are physically large and have

large surface area will typically have higher RMS current

ratings. For a given capacitor value, a higher voltage

electrolytic capacitor will be physically lar ger than a lower

voltage capacitor, and thus be able to dissipate more heat to

the surrounding air, and therefore will have a higher RMS

current rating. The consequences of operating an

electrolytic capacitor beyond the RMS current rating is a

shortened operating life. In order to assure maximum

capacitor operating lifetime, the capacitor’s RMS ripple

current rating should be:

Irms u1.2 x d x ILoad

where d is the duty cycle, for a continuous mode buck

regulator

d+ton

T+Vout

Vin

and d+ton

T+|Vout|

|Vout|)Vin

for a buck−boost regulator.

Output Capacitor (Cout)

For low output ripple voltage and good stability, low ESR

output capacitors are recommended. An output capacitor

has two main functions: it filters the output and provides

regulator loop stability. The ESR of the output capacitor a nd

the peak−to−peak value of the inductor ripple current are the

main factors contributing to the output ripple voltage value.

Standard aluminium electrolytics could be adequate for

some applications but for quality design, low ESR types are

recommended.

An aluminium electrolytic capacitor’s ESR value is

related to many factors, such as the capacitance value, the

voltage rating, the physical size and the type of construction.

In most cases, the higher voltage electrolytic capacitors have

lower ESR value. Often capacitors with much higher

voltage ratings may be needed to provide low ESR values,

that are required for low output ripple voltage.

The Output Capacitor Requires an ESR Value that has

an Upper and Lower Limit

As mentioned above, a low ESR value is needed for low

output ripple voltage, typically 1% to 2% of the output

voltage. But if the selected capacitor’s ESR is extremely low

(below 0.03 W), there is a possibility of an unstable feedback

loop, resulting in oscillation at the output. This situation can

occur when a tantalum capacitor, that can have a very low

ESR, is used as the only output capacitor.

At Low Temperatures, Put in Parallel Aluminium

Electrolytic Capacitors with Tantalum Capacitors

Electrolytic capacitors are not recommended for

temperatures below −25°C. The ESR rises dramatically at

cold temperatures and typically rises 3 times at −25°C and

as much as 10 times at −40°C. Solid tantalum capacitors

have much better ESR spec at cold temperatures and are

recommended for temperatures below −25°C. They can be

also used in parallel with aluminium electrolytics. The value

of the tantalum capacitor should be about 10% or 20% of the

total capacitance. The output capacitor should have at least

50% higher RMS ripple current rating at 52 kHz than the

peak−to−peak inductor ripple current.

Catch Diode

Locate the Catch Diode Close to the LM2574

The LM2574 is a step−down buck converter, it requires a

fast diode to provide a return path for the inductor current

when the switch turns off. This diode must be located close

to the LM2574 using short leads and short printed circuit

traces to avoid EMI problems.

Use a Schottky or a Soft Switching

Ultra−Fast Recovery Diode

Since the rectifier diodes are very significant source of

losses within switching power supplies, choosing the

rectifier that best fits into the converter design is an

important process. Schottky diodes provide the best

performance because of their fast switching speed and low

forward voltage drop.

They provide the best efficiency especially in low output

voltage applications (5.0 V and lower). Another choice

could be Fast−Recovery, or Ultra−Fast Recovery diodes. It

has to be noted, that some types of these diodes with an

abrupt turnoff characteristic may cause instability or EMI

troubles.

A fast−recovery diode with soft recovery characteristics

can better fulfill some quality, low noise design

requirements. Table 1 provides a list of suitable diodes for

the LM2574 regulator. Standard 50/60 Hz rectifier diodes,

such as the 1N4001 series or 1N5400 series are NOT

suitable.

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Inductor

The magnetic components are the cornerstone of all

switching power supply designs. The style of the core and

the winding technique used in the magnetic component’s

design have a great influence on the reliability of the overall

power supply.

Using an improper or poorly designed inductor can cause

high voltage spikes generated by the rate of transitions in

current within the switching power supply, and the

possibility of core saturation can arise during an abnormal

operational mode. Voltage spikes can cause the

semiconductors to enter avalanche breakdown and the part

can instantly fail if enough energy is applied. It can also

cause significant RFI (Radio Frequency Interference) and

EMI (Electro−Magnetic Interference) problems.

Continuous and Discontinuous Mode of Operation

The LM2574 step−down converter can operate in both the

continuous and the discontinuous modes of operation. The

regulator works in the continuous mode when loads are

relatively heavy, the current flows through the inductor

continuously and never falls to zero. Under light load

conditions, the circuit will be forced to the discontinuous

mode when inductor current falls to zero for certain period

of time (see Figure 24 and Figure 25). Each mode has

distinctively different operating characteristics, which can

affect the regulator performance and requirements. In many

cases the preferred mode of operation is the continuous

mode. It o ffers greater output power, lower peak currents in

the switch, inductor and diode, and can have a lower output

ripple voltage. On the other hand it does require larger

inductor values to keep the inductor current flowing

continuously, especially at low output load currents and/or

high input voltages.

To simplify the inductor selection process, an inductor

selection guide for the LM2574 regulator was added to this

data sheet (Figures 19 through 23). This guide assumes that

the regulator is operating in the continuous mode, and

selects an inductor that will allow a peak−to−peak inductor

ripple current to be a certain percentage of the maximum

design load current. This percentage is allowed to change as

different design load currents are selected. For light loads

(less than approximately 0.2 A) it may be desirable to

operate the regulator in the discontinuous mode, because t he

inductor value and size can be kept relatively low.

Consequently, the percentage of inductor peak−to−peak

current increases. This discontinuous mode of operation is

perfectly acceptable for this type of switching converter.

Any buck regulator will be forced to enter discontinuous

mode if the load current is light enough.

Selecting the Right Inductor Style

Some important considerations when selecting a core type

are core material, cost, the output power of the power supply,

the physical volume the inductor must fit within, and the

amount of EMI (Electro−Magnetic Interference) shielding

that the core must provide. There are many different styles

of inductors available, such as pot core, E−core, toroid and

bobbin core, as well as different core materials such as

ferrites and powdered iron from different manufacturers.

For high quality design regulators the toroid core seems to

be the best choice. Since the magnetic flux is contained

within the core, it generates less EMI, reducing noise

problems in sensitive circuits. The least expensive is the

bobbin core type, which consists of wire wound on a ferrite

rod core. This type of inductor generates more EMI due to

the fact that its core is open, and the magnetic flux is not

contained within the core.

When multiple switching regulators are located on the

same printed circuit board, open core magnetics can cause

interference between two or more of the regulator circuits,

especially a t high currents due to mutual coupling. A toroid,

pot core or E−core (closed magnetic structure) should be

used in such applications.

Do Not Operate an Inductor Beyond its Maximum

Rated Current

Exceeding an inductor’s maximum current rating may

cause the inductor to overheat because of the copper wire

losses, o r the core may saturate. Core saturation occurs when

the flux density is too high and consequently the cross

sectional area of the core can no longer support additional

lines of magnetic flux.

This causes the permeability of the core to drop, the

inductance value decreases rapidly and the inductor begins

to look mainly resistive. It has only the dc resistance of the

winding. This can cause the switch current to rise very

rapidly and force the LM2574 internal switch into

cycle−by−cycle current limit, thus reducing the dc output

load current. This can also result in overheating of the

inductor and/or the LM2574. Different inductor types have

different saturation characteristics, and this should be kept

in mind when selecting an inductor.

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HORIZONTAL TIME BASE: 5.0 ms/DIV

VERTRICAL RESOLUTION 200 mADV

Figure 24. Continuous Mode Switching

Current Waveforms

0.5 A

0 A

0.5 A

0 A

Power

Switch

Current

Waveform

Inductor

Current

Waveform

VERTICAL RESOLUTION 100 mADV

HORIZONTAL TIME BASE: 5.0 ms/DIV

Figure 25. Discontinuous Mode Switching

Current Waveforms

0.1 A

0 A

0.1 A

0 A

Power

Switch

Current

Waveform

Inductor

Current

Waveform

GENERAL RECOMMENDATIONS

Output Voltage Ripple and Transients

Source of the Output Ripple

Since the LM2574 is a switch mode power supply

regulator, its output voltage, if left unfiltered, will contain a

sawtooth ripple voltage at the switching frequency. The

output ripple voltage value ranges from 0.5% to 3% of the

output voltage. It is caused mainly by the inductor sawtooth

ripple current multiplied by the ESR of the output capacitor.

Short Voltage Spikes and How to Reduce Them

The regulator output voltage may also contain short

voltage spikes at the peaks of the sawtooth waveform (see

Figure 26). These voltage spikes are present because of the

fast switching action of the output switch, and the parasitic

inductance of the output filter capacitor. There are some

other important factors such as wiring inductance, stray

capacitance, a s well as the scope probe used to evaluate these

transients, all these contribute to the amplitude of these

spikes. To minimize these voltage spikes, low inductance

capacitors should be used, and their lead lengths must be

kept short. The importance of quality printed circuit board

layout design should also be highlighted.

HORIZONTAL TIME BASE: 5.0 ms/DIV

VERTRICAL RESOLUTION 20 mV/DIV

Voltage spikes caused by switching action of the output

switch and the parasitic inductance of the output capacitor

Figure 26. Output Ripple Voltage Waveforms

Unfiltered

Output

Voltage

Filtered

Output

Voltage

Minimizing the Output Ripple

In order t o m inimize t he o utput ripple v oltage i t i s p ossible

to enlarge the inductance value of the inductor L1 and/or to

use a l ar ger v alue o utput c apacitor. There i s a lso a nother w ay

to smooth the output by means of an additional LC filter

(20 mH, 100 mF), that can be added to the output (see

Figure 35) to further reduce the amount of output ripple and

transients. With such a filter it is possible to reduce the

output ripple voltage transients 10 times or more. Figure 26

shows the dif ference between filtered and unfiltered output

waveforms of the regulator shown in Figure 34.

The upper waveform is from the normal unfiltered output

of the converter , while the lower waveform shows the output

ripple voltage filtered by an additional LC filter.

Heatsinking and Thermal Considerations

The LM2574 is available in both 8−pin DIP and SO−16L

packages. W hen us ed in t he t ypical a pplication t he c opper l ead

frame conducts the majority of the heat from the die, through

the leads, to the printed circuit copper. The copper and the

board are the heatsink for this package and the other heat

producing components, such as the catch diode and inductor.

For the best thermal performance, wide copper traces

should be used and all ground and unused pins should be

soldered to generous amounts of printed circuit board

copper, such as a ground plane. Large areas of copper

provide the best transfer of heat to the surrounding air. One

exception t o this is the output (switch) pin, which should not

have large areas of copper in order to minimize coupling to

sensitive circuitry.

Additional improvement in heat dissipation can be

achieved even by using of double sided or multilayer boards

which can provide even better heat path to the ambient.

Using a socket for the 8−pin DIP package is not

recommended because socket represents an additional

thermal resistance, and as a result the junction temperature

will be higher.

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Since the current rating of the LM2574 is only 0.5 A, the

total package power dissipation for this switcher is quite

low, ranging from approximately 0.1 W up to 0.75 W under

varying conditions. In a carefully engineered printed circuit

board, the through−hole DIP package can easily dissipate u p

to 0.75 W, even at ambient temperatures of 60°C, and still

keep the maximum junction temperature below 125°C.

Thermal Analysis and Design

The following procedure must be performed to determine

the operating junction temperature. First determine:

1. PD(max) − maximum regulator power dissipation in

the application.

2. TA(max) − maximum ambient temperature in the

application.

3. TJ(max) − maximum allowed junction temperature

(125°C for the LM2574). For a conservative

design, the maximum junction temperature

should not exceed 110°C to assure safe

operation. For every additional +10°C

temperature rise that the junction must

withstand, the estimated operating lifetime

of the component is halved.

4. RqJC − package thermal resistance junction−case.

5. RqJA − package thermal resistance junction−ambient.

(Refer to Absolute Maximum Ratings on page 2 of this data

sheet or RqJC and RqJA values).

The following formula is to calculate the approximate

total power dissipated by the LM2574:

PD = (Vin x IQ) + d x ILoad x Vsat

where d is the duty cycle and for buck converter

d+ton

T+VO

Vin,

IQ (quiescent current) and Vsat can be found in the

LM2574 data sheet,

Vin is minimum input voltage applied,

VO is the regulator output voltage,

ILoad is the load current.

MBR150

68 mH

Output

Feedback

8.0 to 25 V

Unregulated

DC Input

Cin

22 mF

34ON

/OFFPwr

Gnd

+Vin

−12 V @ 100 mA

Regulated

Output

Cout

680 mF

LM2574−12

2 Sig

Gnd

Figure 27. Inverting Buck−Boost Develops −12 V

(6)

(12)

(4)

(5)

(3)

(14)

The dynamic switching losses during turn−on and

turn−off can be neglected if a proper type catch diode is used.

The junction temperature can be determined by the

following expression:

TJ = (RqJA)(PD) + TA

where (RqJA)(PD) represents the junction temperature rise

caused by the dissipated power and TA is the maximum

ambient temperature.

Some Aspects That can Influence Thermal Design

It should be noted that the package thermal resistance and

the junction temperature rise numbers are all approximate,

and there are many factors that will affect these numbers,

such as PC board size, shape, thickness, physical position,

location, board temperature, as well as whether the

surrounding air is moving or still. At higher power levels the

thermal resistance decreases due to the increased air current

activity.

Other factors are trace width, total printed circuit copper

area, copper thickness, single− or double−sided, multilayer

board, the amount of solder on the board or even color of the

traces.

The size, quantity and spacing of other components on the

board can also influence its effectiveness to dissipate the

heat. Some of them, like the catch diode or the inductor will

generate some additional heat.

ADDITIONAL APPLICATIONS

Inverting Regulator

An inverting buck−boost regulator using the LM2574−12

is shown in Figure 27. This circuit converts a positive input

voltage to a negative output voltage with a common ground

by bootstrapping the regulators ground to the negative

output voltage. By grounding the feedback pin, the regulator

senses the inverted output voltage and regulates it.

In t his e xample t he L M2574−12 i s u sed t o g enerate a − 12 V

output. The maximum input voltage in this case cannot

exceed 2 8 V because the maximum voltage appearing across

the regulator is the absolute sum of the input and output

voltages and this must be limited to a maximum of 40 V.

This circuit configuration is able to deliver approximately

0.1 A to the output when the input voltage is 8.0 V or higher.

At lighter loads the minimum input voltage required drops

to approximately 4.7 V, because the buck−boost regulator

topology can produce an output voltage that, in its absolute

value, is either greater or less than the input voltage.

Since the switch currents in this buck−boost configuration

are higher than in the standard buck converter topology, the

available output current is lower.

This type of buck−boost inverting regulator can also

require a larger amount of startup input current, even for

light loads. This may overload an input power source with

a current limit less than 0.6 A.

Because o f the relatively high startup currents required by

this inverting regulator topology, the use of a delayed startup

or an undervoltage lockout circuit is recommended.

While using a delayed startup arrangement, the input

capacitor can charge up to a higher voltage before the

switch−mode regulator begins to operate.

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The high input current needed for startup is now partially

supplied by the input capacitor Cin.

Design Recommendations:

The inverting regulator operates in a different manner

than the buck converter and so a different design procedure

has to be used to select the inductor L1 or the output

capacitor Cout.

The output capacitor values must be larger than what is

normally required for buck converter designs. Low input

voltages or high output currents require a lar ge value output

capacitor (in the range of thousands of mF).

The recommended range of inductor values for the

inverting converter design is between 68 mH and 220 mH. To

select an inductor with an appropriate current rating, the

inductor peak current has to be calculated.

MBR150

68 mH

Output

Feedback

12 to 25 V

Unregulated

DC Input

Cin

22 mF

/50 V

43ON/OFF Pwr

Gnd

+Vin

−12 V @ 100 mA

Regulated

Output

Cout

680 mF

/16 V

LM2574−12

0.1 mF

47 k R2

47 k

2 Sig

Gnd

Figure 28. Inverting Buck−Boost Regulator with

Delayed Startup

(3)

(12) (14)

(5)

(6) (4)

The following formula is used to obtain the peak inductor

current:

Ipeak [

ILoadǒVin )|VO|Ǔ

Vin )Vin xt

2L1

where ton +|VO|

Vin )|VO|x1.0

fosc

, and fosc = 52 kHz.

Under normal continuous inductor current operating

conditions, the worst case occurs when Vin is minimal.

It has been already mentioned above, that in some

situations, the delayed startup or the undervoltage lockout

features could be very useful. A delayed startup circuit

applied to a buck−boost converter is shown in Figure 28.

Figure 34 in the “Undervoltage Lockout” section describes

an undervoltage lockout feature for the same converter

topology.

With the inverting configuration, the use of the ON/OFF

pin requires some level shifting techniques. This is caused

by the fact, that the ground pin of the converter IC is no

longer at ground. Now, the ON/OFF pin threshold voltage

(1.3 V approximately) has to be related to the negative

output voltage level. There are many different possible

shutdown methods, two of them are shown in Figures 29

and 30.

LM2574−XX

and

3 Gnds

Pins

ON/OFF

+Vin

47 k

Cin

22 mF

NOTE: This picture does not show the complete circuit.

47 k

470

Shutdown

Input

MOC8101

−Vout

Off

5.0 V

+Vin

Figure 29. Inverting Buck−Boost Regulator Shutdow

Circuit Using an Optocoupler

(5)

(12)

(4)

and

(6)

NOTE: This picture does not show the complete circuit.

5.6 k

2N3906

LM2574−XX

and

3 Gnds

Pins

ON/OFF

12 k −Vout

+Vin

Shutdown

Input

Off

+Vin

Cin

22 mF

Figure 30. Inverting Buck−Boost Regulator Shutdow

Circuit Using a PNP Transistor

(5)

(12)

(4)

and

(6)

Negative Boost Regulator

This example is a variation of the buck−boost topology

and it is called negative boost regulator. This regulator

experiences relatively high switch current, especially at low

input voltages. The internal switch current limiting results in

lower output load current capability.

LM2574, NCV2574

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The circuit in Figure 31 shows the negative boost

configuration. The input voltage in this application ranges

from −5.0 to −12 V and provides a regulated −12 V output.

If the input voltage is greater than −12 V, the output will rise

above −12 V accordingly, but will not damage the regulator.

1N5817

330 mH

Output

Feedback

Vout = −12 V

Load Current

60 mA for Vin = −5.2 V

120 mA for Vin = −7.0 V

Vin

Cout

1000 mF

Cin

22 mF

LM2574−12

34 ON/OFFPwr

Gnd

+Vin

2Sig

Gnd

−5.0 to −12 V

Figure 31. Negative Boost Regulator

(5)

(12)

(3)

(14)

(4)(6)

Design Recommendations:

The same design rules as for the previous inverting

buck−boost converter can be applied. The output capacitor

Cout must be chosen lar ger than what would be required for

a standard buck converter. Low input voltages or high output

currents require a large value output capacitor (in the range

of thousands of mF). The recommended range of inductor

values for the negative boost regulator is the same as for

inverting converter design.

Another important point is that these negative boost

converters cannot provide any current limiting load

protection in the event of a short in the output so some other

means, such as a fuse, may be necessary to provide the load

protection.

Delayed Startup

There are some applications, like the inverting regulator

already mentioned above, which require a higher amount of

startup current. In such cases, if the input power source is

limited, this delayed startup feature becomes very useful.

To provide a time delay between the time when the input

voltage is applied and the time when the output voltage

comes up, the circuit in Figure 32 can be used. As the input

voltage is applied, the capacitor C1 charges up, and the

voltage across the resistor R2 falls down. When the voltage

on the ON/OFF pin falls below the threshold value 1.3 V, the

regulator starts up. Resistor R1 is included to limit the

maximum voltage applied to the ON/OFF pin. It reduces t h e

power supply noise sensitivity, and also limits the capacitor

C1 discharge current, but its use is not mandatory.

When a high 50 Hz or 60 Hz (100 Hz or 120 Hz

respectively) ripple voltage exists, a long delay time can

cause some problems by coupling the ripple into the

ON/OFF pin, the regulator could be switched periodically

on and off with the line (or double) frequency.

47 k

LM2574−XX

and

3 Gnds

Pins

ON/OFF

47 k

+Vin

0.1 mF

Cin

22 mF

NOTE: This picture does not show the complete circuit.

Figure 32. Delayed Startup Circuitry

(5)

(12)

(4)

and

(6)

Undervoltage Lockout

Some applications require the regulator to remain of f until

the input voltage reaches a certain threshold level. Figure 33

shows an undervoltage lockout circuit applied to a buck

regulator. A version of this circuit for buck−boost converter

is shown in Figure 34. Resistor R3 pulls the ON/OFF pin

high and keeps the regulator off until the input voltage

reaches a predetermined threshold level, which is

determined by the following expression:

Vth [VZ1 )ǒ1.0 )R2

R1ǓVBE(Q1)

10 k

1N5242B

10 k

2N3904

47 k

Cin

22 mF

LM2574−XX

and

3 Gnds

Pins

ON/OFF

+Vin

NOTE: This picture does not show the complete circuit.

Figure 33. Undervoltage Lockout Circuit for

Buck Converter

(5)

(12)

(4)

and

(6)

LM2574, NCV2574

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15 k

1N5242

15 k

2N3904

68 k

Cin

22 mF

LM2574−XX

and

3 Gnds

Pins

ON/OFF

+Vin

−Vout

NOTE: This picture does not show the complete circuit (see Figure 27).

Figure 34. Undervoltage Lockout Circuit for

Buck−Boost Converter

(5)

(12)

(4)

and

(6)

Adjustable Output, Low−Ripple Power Supply

A 0.5 A output current capability power supply that

features an adjustable output voltage is shown in Figure 35.

This regulator delivers 0.5 A into 1.2 to 35 V output. The

input voltage ranges from roughly 3.0 to 40 V. In order to

achieve a 10 or more times reduction of output ripple, an

additional L−C filter is included in this circuit.

1N5819

150 mH

Output

Feedback

50 k

1.1 k

20 mHOutput

Voltage

1.2 to 35 V @ 0.5 A

Optional Output

Ripple Filter

40 V Max

Unregulated

DC Input

Cout

1000 mFC1

100 mF

Cin

22 mF

LM2574−ADJ

34ON

/OFFPwr

Gnd

+Vin

2 Sig

Gnd

Figure 35. 1.2 to 35 V Adjustable 500 mA Power Supply with Low Output Ripple

(12)

(5)

(3)

(14)

(6) (4)

LM2574, NCV2574

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The LM2574−5 Step−Down Voltage Regulator with 5.0 V @ 0.5 A Output Power Capability.

Typical Application With Through−Hole PC Board Layout

1N5819

330 mH

Output

Feedback

Unregulated

DC Input

+Vin = 7.0 to 40 V

220 mF

22 mF

LM2574−5

34ON

/OFFPwr

Gnd

+Vin

Regulated Output

+Vout = 5.0 V @ 0.5 A

Gnd Gnd

C1 − 22 mF, 63 V, Aluminium Electrolytic

C2 − 220 mF, 16 V, Aluminium Electrolytic

D1 − 1.0 A, 40 V, Schottky Rectifier, 1N5819

L1 − 330 mH, RL−1284−330, Renco Electronics

2 Sig

Gnd

Figure 36. Schematic Diagram of the LM2574−5 Step−Down Converter

(12)

(5)

(3)

(14)

(6) (4)

LM2574−5.0

C1 C2

D1 Vout

Gnd

+Vin

Figure 37. PC Board Layout Component Side

NOTE: Not to scale.

Figure 38. PC Board Layout Copper Side

NOTE: Not to scale.

LM2574, NCV2574

http://onsemi.com

The LM2574−ADJ Step−Down Voltage Regulator with 5.0 V @ 0.5 A Output Power Capability Typical

Application With Through−Hole PC Board Layout

1N5819

330 mH

Output

Feedback

6.12 kW

2.0 kW

22 mHRegulated

Output Filtered

Vout = 5.0 V @ 0.5 A

Output

Ripple Filter

Unregulated

DC Input

220 mF

100 mF

22 mF

LM2574−ADJ

34ON

/OFFPwr

Gnd

+Vin

2 Sig

Gnd

+Vin = 7.0 to 40 V

Gnd Gnd

C1 − 22 mF, 63 V, Aluminium Electrolytic

C2 − 220 mF, 16 V, Aluminium Electrolytic

C3 − 100 mF, 16 V Aluminium Electrolytic

D1 − 1.0 A, 40 V, Schottky Rectifier, 1N5819

L1 − 330 mH, RL−1284−330, Renco Electronics

L2 − 25 mH, SFT52501, TDK

R1 − 2.0 kW, 0.1%, 0.25 W

R2 − 6.12 kW, 0.1%, 0.25 W

Figure 39. Schematic Diagram of the 5.0 V @ 0.5 A Step−Down Converter Using the LM2574−ADJ

(An additional LC filter is included to achieve low output ripple voltage)

(12)

(5)

(3)

(14)

(6) (4)

LM2574

C1 C2

Vout

Gnd

+Vin C3

+Gnd

R1 R2

Figure 40. PC Board Layout Component Side

NOTE: Not to scale.

Figure 41. PC Board Layout Copper Side

NOTE: Not to scale.

References

•Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990

•Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995

LM2574, NCV2574

http://onsemi.com

ORDERING INFORMATION

Device Nominal Output

Voltage Operating Junction

Temperature Range Package Shipping

LM2574DW−ADJ

1.23 V to 37 V TJ = −40° to +125°C

SO−16 WB 47 Units/Rail

LM2574DW−ADJR2 SO−16 WB 1000 Units/Tape & Reel

LM2574DW−ADJR2G SO−16 WB

(Pb−Free)

LM2574N−ADJ PDIP−8 50 Units/Rail

LM2574N−ADJG PDIP−8

(Pb−Free)

NCV2574DW−ADJR2 SO−16 WB 1000 Units/Tape & Reel

NCV2574DW−ADJR2G SO−16 WB

(Pb−Free)

LM2574N−3.3 3.3 V TJ = −40° to +125°CPDIP−8

50 Units/Rail

LM2574N−3.3G PDIP−8

(Pb−Free)

LM2574N−5 5.0 V TJ = −40° to +125°CPDIP−8

LM2574N−5G PDIP−8

(Pb−Free)

LM2574N−12 12 V TJ = −40° to +125°CPDIP−8

LM2574N−12G PDIP−8

(Pb−Free)

LM2574N−15 15 V TJ = −40° to +125°CPDIP−8

LM2574N−15G PDIP−8

(Pb−Free)

†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging

Specifications Brochure, BRD8011/D.

*NCV devices: Tlow = −40°C, Thigh = +125°C. Guaranteed by Design. NCV prefix is for automotive and other applications requiring site and

change control.

SO−16 WB

DW SUFFIX

CASE 751G

MARKING DIAGRAMS

LM2574DW−A

AWLYYWWG

PDIP−8

N SUFFIX

CASE 626

2574−xxx

AWL

YYWWG

xxx = 3.3, 5.0, 12, 15, or ADJ

A = Assembly Location

WL = Wafer Lot

Y = Year

WW = Work Week

G = Pb−Free Package

2574N−xxx

AWL

YYWWG

CV2574DW−A

AWLYYWWG

*NCV part

LM2574, NCV2574

http://onsemi.com

PACKAGE DIMENSIONS

SO−16 WB

DW SUFFIX

CASE 751G−03

ISSUE C

14X

B16X

SEATING

PLANE

0.25 B S

16 9

hX 45_

0.25

H8X

NOTES:

1. DIMENSIONS ARE IN MILLIMETERS.

2. INTERPRET DIMENSIONS AND TOLERANCES

PER ASME Y14.5M, 1994.

3. DIMENSIONS D AND E DO NOT INLCUDE

MOLD PROTRUSION.

4. MAXIMUM MOLD PROTRUSION 0.15 PER SIDE.

5. DIMENSION B DOES NOT INCLUDE DAMBAR

PROTRUSION. ALLOWABLE DAMBAR

PROTRUSION SHALL BE 0.13 TOTAL IN

EXCESS OF THE B DIMENSION AT MAXIMUM

MATERIAL CONDITION.

DIM MIN MAX

MILLIMETERS

A2.35 2.65

A1 0.10 0.25

B0.35 0.49

C0.23 0.32

D10.15 10.45

E7.40 7.60

e1.27 BSC

H10.05 10.55

h0.25 0.75

L0.50 0.90

q0 7

PDIP−8

N SUFFIX

CASE 626−05

ISSUE L

NOTES:

1. DIMENSION L TO CENTER OF LEAD WHEN

FORMED PARALLEL.

2. PACKAGE CONTOUR OPTIONAL (ROUND OR

SQUARE CORNERS).

3. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

NOTE 2 −A−

−B−

−T−

SEATING

PLANE

GDK

0.13 (0.005) B M

DIM MIN MAX MIN MAX

INCHESMILLIMETERS

A9.40 10.16 0.370 0.400

B6.10 6.60 0.240 0.260

C3.94 4.45 0.155 0.175

D0.38 0.51 0.015 0.020

F1.02 1.78 0.040 0.070

G2.54 BSC 0.100 BSC

H0.76 1.27 0.030 0.050

J0.20 0.30 0.008 0.012

K2.92 3.43 0.115 0.135

L7.62 BSC 0.300 BSC

M−−− 10 −−− 10

N0.76 1.01 0.030 0.040

LM2574, NCV2574

http://onsemi.com

ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice

to any products herein. SCILLC makes no warranty , representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability

arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.

“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All

operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights

nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications

intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should

Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, af filiates,

and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death

associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal

Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.

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LM2574/D

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