© Semiconductor Components Industries, LLC, 2006
August, 2006 − Rev. 8 1Publication Order Number:
LM2574/D
LM2574, NCV2574
0.5 A, Adjustable Output
Voltage, Step−Down
Switching Regulator
The LM2574 series of regulators are monolithic integrated circuits
ideally suited for easy and convenient design of a step−down
switching regulator (buck converter). All circuits of this series are
capable of driving a 0.5 A load with excellent line and load regulation.
These devices are available in fixed output voltages of 3.3 V, 5.0 V,
12 V, 15 V, and an adjustable output version.
These regulators were designed to minimize the number of external
components to simplify the power supply design. Standard series of
inductors optimized for use with the LM2574 are offered by several
different inductor manufacturers.
Since the LM2574 converter is a switch−mode power supply, its
efficiency is significantly higher in comparison with popular
three−terminal linear regulators, especially with higher input voltages.
In most cases, the power dissipated by the LM2574 regulator is so low,
that the copper traces on the printed circuit board are normally the only
heatsink needed and no additional heatsinking is required.
The LM2574 features include a guaranteed ±4% tolerance on output
voltage within specified input voltages and output load conditions, and
±10% on the oscillator frequency (±2% over 0°C to +125°C). External
shutdown is included, featuring 60 mA (typical) standby current. The
output switch includes cycle−by−cycle current limiting, as well as
thermal shutdown for full protection under fault conditions.
Features
3.3 V, 5.0 V, 12 V, 15 V, and Adjustable Output Versions
Adjustable Version Output Voltage Range, 1.23 to 37 V ±4% max
over Line and Load Conditions
Guaranteed 0.5 A Output Current
Wide Input Voltage Range: 4.75 to 40 V
Requires Only 4 External Components
52 kHz Fixed Frequency Internal Oscillator
TTL Shutdown Capability, Low Power Standby Mode
High Efficiency
Uses Readily Available Standard Inductors
Thermal Shutdown and Current Limit Protection
NCV Prefix for Automotive and Other Applications Requiring Site
and Control Changes
Pb−Free Packages are Available*
Applications
Simple and High−Efficiency Step−Down (Buck) Regulators
Efficient Pre−regulator for Linear Regulators
On−Card Switching Regulators
Positive to Negative Converters (Buck−Boost)
Negative Step−Up Converters
Power Supply for Battery Chargers
*For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting
Techniques Reference Manual, SOLDERRM/D.
1
16 SO−16 WB
DW SUFFIX
CASE 751G
PIN CONNECTIONS
*
(Top View)
12
Pwr Gnd
ON/OFF
*
*
*
*
11
10
9
5
6
7
8
*
16
*
*
FB
Sig Gnd
Output
*
Vin
15
14
13
1
2
3
4
*
* No internal connection, but should be soldered to
* PC board for best heat transfer.
*
(Top View)
8
FB
Sig Gnd
ON/OFF
Pwr Gnd
Output
*
Vin
7
6
5
1
2
3
4
See detailed ordering and shipping information in the package
dimensions section on page 24 of this data sheet.
ORDERING INFORMATION
See general marking information in the device marking
section on page 24 of this data sheet.
DEVICE MARKING INFORMATION
PDIP−8
N SUFFIX
CASE 626
1
8
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Figure 1. Block Diagram and Typical Application
7.0 − 40 V
Unregulated
DC Input
L1
330 mH
Pwr
Gnd
+Vin
5
Cin
22 mF4ON
/OFF3
Output
7
Feedback
1
D1
1N5819 Cout
220 mF
Typical Application (Fixed Output Voltage Versions)
Representative Block Diagram and Typical Application
Unregulated
DC Input
+Vin
5
Cout
Feedback
1
Cin
L1
D1
R2
R1
1.0 k Output
7
Pwr Gnd
4
ON/OFF
3
Reset
Latch
Thermal
Shutdown
52 kHz
Oscillator
1.235 V
Band−Gap
Reference
Freq
Shift
18 kHz
Comparator
Fixed Gain
Error Amplifier
Current
Limit
Driver
1.0 Amp
Switch
ON/OFF
3.1 V Internal
Regulator
Vout
Load
Output
Voltage Versions
3.3 V
5.0 V
12 V
15 V
R2
(W)
1.7 k
3.1 k
8.84 k
11.3 k
For adjustable version
R1 = open, R2 = 0 W
LM2574
5.0 V Regulated
Output 0.5 A Load
Sig
Gnd
2
Sig Gnd
2
(12) (14)
(3)
(4) (6) (5)
(5)
(12)
(3)
(4)
(14)
(6)
NOTE: Pin numbers in ( ) are for the SO−16W package.
ABSOLUTE MAXIMUM RATINGS (Absolute Maximum Ratings indicate limits beyond which damage to the device may occur).
Rating Symbol Value Unit
Maximum Supply Voltage Vin 45 V
ON/OFF Pin Input Voltage −0.3 V V +Vin V
Output Voltage to Ground (Steady State) −1.0 V
DW Suffix, Plastic Package Case 751G
Max Power Dissipation PDInternally Limited W
Thermal Resistance, Junction−to−Air RqJA 145 °C/W
N Suffix, Plastic Package Case 626
Max Power Dissipation PDInternally Limited W
Thermal Resistance, Junction−to−Ambient RqJA 100 °C/W
Thermal Resistance, Junction−to−Case RqJC 5.0 °C/W
Storage Temperature Range Tstg −65°C to +150°C°C
Minimum ESD Rating 2.0 kV
(Human Body Model: C = 100 pF, R = 1.5 kW)
Lead Temperature (Soldering, 10 seconds) 260 °C
Maximum Junction Temperature TJ150 °C
Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit
values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied,
damage may occur and reliability may be affected.
NOTE: ESD data available upon request.
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OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee
specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics).
Rating Symbol Value Unit
Operating Junction Temperature Range TJ−40 to +125 °C
Supply Voltage Vin 40 V
SYSTEM PARAMETERS ([Note 1] Test Circuit Figure 16)
ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable
version, Vin = 25 V for the 12 V version, Vin = 30 V for the 15 V version. ILoad = 100 mA. For typical values TJ = 25°C, for min/max values
TJ is the operating junction temperature range that applies [Note 2], unless otherwise noted).
Characteristic Symbol Min Typ Max Unit
LM2574−3.3 ([Note 1] Test Circuit Figure 16)
Output Voltage (Vin = 12 V, ILoad = 100 mA, TJ = 25°C) Vout 3.234 3.3 3.366 V
Output Voltage (4.75 V Vin 40 V, 0.1 A ILoad 0.5 A) Vout V
TJ = 25°C 3.168 3.3 3.432
TJ = −40 to +125°C 3.135 3.465
Efficiency (Vin = 12 V, ILoad = 0.5 A) η 72 %
LM2574−5 ([Note 1] Test Circuit Figure 16)
Output Voltage (Vin = 12 V, ILoad = 100 mA, TJ = 25°C) Vout 4.9 5.0 5.1 V
Output Voltage (7.0 V Vin 40 V, 0.1 A ILoad 0.5 A) Vout V
TJ = 25°C4.8 5.0 5.2
TJ = −40 to +125°C4.75 5.25
Efficiency (Vin = 12 V, ILoad = 0.5 A) η 77 %
LM2574−12 ([Note 1] Test Circuit Figure 16)
Output Voltage (Vin = 25 V, ILoad = 100 mA, TJ = 25°C) Vout 11.76 10 12.24 V
Output Voltage (15 V Vin 40 V, 0.1 A ILoad 0.5 A) Vout V
TJ = 25°C11.52 12 12.48
TJ = −40 to +125°C11.4 12.6
Efficiency (Vin = 15 V, ILoad = 0.5 A) η 88 %
LM2574−15 ([Note 1] Test Circuit Figure 16)
Output Voltage (Vin = 30 V, ILoad = 100 mA, TJ = 25°C) Vout 14.7 15 15.3 V
Output Voltage (18 V < Vin < 40 V, 0.1 A < ILoad < 0.5 A) Vout V
TJ = 25°C 14.4 15 15.6
TJ = −40 to +125°C 14.25 15.75
Efficiency (Vin = 18 V, ILoad = 0.5 A) η 88 %
LM2574 ADJUSTABLE VERSION ([Note 1] Test Circuit Figure 16)
Feedback Voltage Vin = 12 V, ILoad = 100 mA, Vout = 5.0 V, TJ = 25°C VFB 1.217 1.23 1.243 V
Feedback Voltage 7.0 V Vin 40 V, 0.1 A ILoad 0.5 A, Vout = 5.0
VVFBT V
TJ = 25°C 1.193 1.23 1.267
TJ = −40 to +125°C 1.18 1.28
Efficiency (Vin = 12 V, ILoad = 0.5 A, Vout = 5.0 V) η 77 %
1. External components such as the catch diode, inductor, input and output capacitors can affect the switching regulator system performance.
When the LM2574 is used as shown in the Figure 16 test circuit, the system performance will be as shown in the system parameters section
of the Electrical Characteristics.
2. Tested junction temperature range for the LM2574, NCV2574: Tlow = −40°C Thigh = +125°C.
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SYSTEM PARAMETERS ([Note 3] Test Circuit Figure 16)
ELECTRICAL CHARACTERISTICS (continued) (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and
Adjustable version, Vin = 25 V for the 12 V version, Vin = 30 V for the 15 V version. ILoad = 100 mA. For typical values TJ = 25°C, for
min/max values TJ is the operating junction temperature range that applies [Note 4], unless otherwise noted).
Characteristic Symbol Min Typ Max Unit
ALL OUTPUT VOLTAGE VERSIONS
Feedback Bias Current Vout = 5.0 V (Adjustable Version Only) IbnA
TJ = 25°C 25 100
TJ = −40 to +125°C 200
Oscillator Frequency (Note 5) fOkHz
TJ = 25°C 52
TJ = 0 to +125°C 47 52 58
TJ = −40 to +125°C 42 63
Saturation Voltage (Iout = 0.5 A, [Note 6]) Vsat V
TJ = 25°C 1.0 1.2
TJ = −40 to +125°C 1.4
Max Duty Cycle (“on”) (Note 7) DC 93 98 %
Current Limit Peak Current (Notes 5 and 6) ICL A
TJ = 25°C 0.7 1.0 1.6
TJ = −40 to +125°C 0.65 1.8
Output Leakage Current (Notes 8 and 9), TJ = 25°C ILmA
Output = 0 V 0.6 2.0
Output = − 1.0 V 10 30
Quiescent Current (Note 8) IQmA
TJ = 25°C 5.0 9.0
TJ = −40 to +125°C 11
Standby Quiescent Current (ON/OFF Pin = 5.0 V (“off”)) Istby mA
TJ = 25°C 60 200
TJ = −40 to +125°C 400
ON/OFF Pin Logic Input Level V
Vout = 0 V VIH
TJ = 25°C 2.2 1.4
TJ = −40 to +125°C 2.4
Nominal Output Voltage VIL
TJ = 25°C 1.2 1.0
TJ = −40 to +125°C 0.8
ON/OFF Pin Input Current mA
ON/OFF Pin = 5.0 V (“off”), TJ = 25°C IIH 15 30
ON/OFF Pin = 0 V (“on”), TJ = 25°C IIL 0 5.0
3. External components such as the catch diode, inductor, input and output capacitors can affect the switching regulator system performance.
When the LM2574 is used as shown in the Figure 16 test circuit, the system performance will be as shown in the system parameters section
of the Electrical Characteristics.
4. Tested junction temperature range for the LM2574, NCV2574: Tlow = −40°C Thigh = +125°C.
5. The oscillator frequency reduces to approximately 18 kHz in the event of an output short or an overload which causes the regulated output
voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average power dissipation of th e
IC by lowering the minimum duty cycle from 5% down to approximately 2%.
6. Output (Pin 2) sourcing current. No diode, inductor or capacitor connected to the output pin.
7. Feedback (Pin 4) removed from output and connected to 0 V.
8. Feedback (Pin 4) removed from output and connected to 12 V for the Adjustable, 3.3 V, and 5.0 V versions, and 25 V for the 12 V and 15 V
versions, to force the output transistor OFF.
9. Vin = 40 V.
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Istby, STANDBY QUIESCENT CURRENT ( A)μ
IQ, QUIESCENT CURRENT (mA)
V
out, OUTPUT VOLTAGE CHANGE (%)
TJ, JUNCTION TEMPERATURE (°C)
IO, OUTPUT CURRENT (A)
TJ, JUNCTION TEMPERATURE (°C)
Vin, INPUT VOLTAGE (V)
Vin, INPUT VOLTAGE (V)
INPUT − OUTPUT DIFFERENTIAL (V)
TJ, JUNCTION TEMPERATURE (°C)
V
out, OUTPUT VOLTAGE CHANGE (%)
Figure 2. Normalized Output Voltage
TJ, JUNCTION TEMPERATURE (°C)
Figure 3. Line Regulation
Vin = 20 V
ILoad = 100 mA
Normalized at TJ = 25°C
Figure 4. Dropout Voltage Figure 5. Current Limit
Figure 6. Quiescent Current Figure 7. Standby Quiescent Current
ILoad = 100 mA
TJ = 25°C
3.3 V, 5.0 V and ADJ
12 V and 15 V
Vin = 25 V
ILoad = 100 mA
ILoad = 500 A
Vin = 12 V
Vin = 40 V
L = 300 mH
ILoad = 500 mA
ILoad = 100 mA
Vout = 5.0 V
Measured at
Ground Pin
TJ = 25°C
VON/OFF = 5.0 V
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16)
1.0
0.8
0.6
0.4
0.2
0
−0.2
−0.4
−0.6
−0.8
−1.0
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
−0.2
−0.4
−0.6
2.0
1.5
1.0
0.5
0
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
20
18
16
14
12
10
8.0
6.0
4.0
200
180
160
140
120
100
80
60
40
20
0
1251007560250−25−50 403530252015105.00
1251007560250−25−50 1251007560250−25−50
403530252015105.00 1251007560250−25−50
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, INPUT VOLTAGE (V)Vin
Vsat, SATURATION VOLTAGE (V)
IFB, FEEDBACK PIN CURRENT (nA)
A
B
C
5 ms/DIV
TJ, JUNCTION TEMPERATURE (°C)
SWITCH CURRENT (A)
5 ms/DIV
TJ, JUNCTION TEMPERATURE (°C)
NORMALIZED FREQUENCY (%)
Figure 8. Oscillator Frequency
TJ, JUNCTION TEMPERATURE (°C)
Figure 9. Switch Saturation Voltage
Figure 10. Minimum Operating Voltage Figure 11. Feedback Pin Current
Figure 12. Continuous Mode Switching Waveforms
V
out
= 5.0 V, 500 mA Load Current, L = 330 mHFigure 13. Discontinuous Mode Switching Waveform
s
V
out
= 5.0 V, 100 mA Load Current, L = 100 mH
Vin = 1.23 V
ILoad = 100 mA
Adjustable Version Only
Vin = 12 V
Normalized at 25°C
Adjustable Version Only
A
B
C
A: Output Pin Voltage, 10 V/DIV.
B: Inductor Current, 0.2 A/DIV.
C: Output Ripple Voltage, 20 mV/DIV, AC−Coupled
A: Output Pin Voltage, 10 V/DIV.
B: Inductor Current, 0.2 A/DIV.
C: Output Ripple Voltage, 20 mV/DIV, AC−Coupled
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) (continued)
8.0
6.0
4.0
2.0
0
−2.0
−4.0
−6.0
−8.0
10
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
100
80
60
40
20
0
−20
−40
−60
−80
−100
1251007550250−25−50 0 0.1 0.2 0.3 0.4 0.5
1251007550250−25−50 1251007550250−25−50
20 V
10 V
0
0.6 A
0.4 A
0.2 A
0
20 mV
AC
20 V
10 V
0
0.4 A
0.2 A
0
20 mV
AC
−40°C
25°C
125°C
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A
B
200 ms/DIV200 ms/DIV
Figure 14. 500 mA Load Transient Response for
Continuous Mode Operation, L = 330 mH, C
out
= 300 mFFigure 15. 250 mA Load Transient Response for
Discontinuous Mode Operation, L = 68 mH, C
out
= 470 m
F
A: Output Voltage, 50 mV/DIV, AC Coupled
B: 100 mA to 500 mA Load Pulse
A
B
A: Output Voltage, 50 mV/DIV, AC Coupled
B: 50 mA to 250 mA Load Pulse
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) (continued)
50 mV
AC
500 mA
0
50 mV
AC
200 mA
100 mA
0
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Figure 16. Test Circuit and Layout Guidelines
D1
1N5819
L1
330 mH
Output
7
1
Feedback
Cout
220 mF
Cin
22 mF
LM2574
Fixed Output
1
34ON
/OFFPwr
Gnd
Vin
Load
Vout
D1
1N5819
L1
330 mH
Output
7
1
Feedback
Cout
220 mF
Cin
22 mF
LM2574
Adjustable
1
Vin
Load
Vout
5.0 V
Fixed Output Voltage Versions
Adjustable Output Voltage Versions
Vout +Vrefǒ1.0 )R2
R1Ǔ
R2 +R1ǒVout
Vref
–1.0Ǔ
Where Vref = 1.23 V, R1
between 1.0 kW and 5.0 kW
R2
6.12 k
R1
2.0 k
7.0 − 40 V
Unregulated
DC Input
2 Sig
Gnd
34ON/OFFPwr
Gnd
2 Sig
Gnd
7.0 V − 40 V
Unregulated
DC Input
Cin −22 mF, 60 V, Aluminium Electrolytic
Cout 220 mF, 25 V, Aluminium Electrolytic
D1 Schottky, 1N5819
L1 330 mH, (For 5.0 Vin, 3.3 Vout, use 100 mH)
R1 2.0 k, 0.1%
R2 6.12 k, 0.1%
NOTE: Pin numbers in ( ) are for the SO−16W package.
(12)
(3)
(6) (4)
(5)
(14)
(12)
(3)
(14)
(6) (4)
(5)
PCB LAYOUT GUIDELINES
As in any switching regulator, the layout of the printed
circuit board is very important. Rapidly switching currents
associated with wiring inductance, stray capacitance and
parasitic inductance of the printed circuit board traces can
generate voltage transients which can generate
electromagnetic interferences (EMI) and affect the desired
operation. As indicated in the Figure 16, to minimize
inductance and ground loops, the length of the leads
indicated by heavy lines should be kept as short as possible.
For best results, single−point grounding (as indicated) or
ground plane construction should be used.
On the other hand, the PCB area connected to the Pin 7
(emitter of the internal switch) of the LM2574 should be
kept to a minimum in order to minimize coupling to sensitive
circuitry.
Another sensitive part of the circuit is the feedback. It is
important to keep the sensitive feedback wiring short. To
assure this, physically locate the programming resistors near
to the regulator, when using the adjustable version of the
LM2574 regulator.
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PIN FUNCTION DESCRIPTION
Pin
Symbol Description (Refer to Figure 1)
SO−16W PDIP−8
12 5 Vin This pin is the positive input supply for the LM2574 step−down switching regulator. In order to
minimize voltage transients and to supply the switching currents needed by the regulator, a
suitable input bypass capacitor must be present (Cin in Figure 1).
14 7 Output This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is
typically 1.0 V. It should be kept in mind that the PCB area connected to this pin should be kept
to a minimum in order to minimize coupling to sensitive circuitry.
4 2 Sig Gnd Circuit signal ground pin. See the information about the printed circuit board layout.
6 4 Pwr Gnd Circuit power ground pin. See the information about the printed circuit board layout.
3 1 Feedback This pin senses regulated output voltage to complete the feedback loop. The signal is divided by
the internal resistor divider network R2, R1 and applied to the non−inverting input of the internal
error amplifier. In the Adjustable version of the LM2574 switching regulator , this pin is the direct
input of the error amplifier and the resistor network R2, R1 is connected externally to allow
programming of the output voltage.
5 3 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the
total input supply current to approximately 80 mA. The input threshold voltage is typically 1.5 V.
Applying a voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this
pin is lower than 1.5 V or if this pin is left open, the regulator will be in the “on” condition.
DESIGN PROCEDURE
Buck Converter Basics
The LM2574 is a “Buck” or Step−Down Converter which
is the most elementary forward−mode converter. Its basic
schematic can be seen in Figure 17.
The operation of this regulator topology has two distinct
time periods. The first one occurs when the series switch is
on, the input voltage is connected to the input of the inductor.
The output of the inductor is the output voltage, and the
rectifier (or catch diode) is reverse biased. During this
period, since there is a constant voltage source connected
across the inductor, the inductor current begins to linearly
ramp upwards, as described by the following equation:
IL(on) +ǒVin –V
outǓton
L
During this “on” period, energy is stored within the core
material in the form of magnetic flux. If the inductor is
properly designed, there is sufficient energy stored to carry
the requirements of the load during the “off” period.
Figure 17. Basic Buck Converter
DVin RLoad
L
Cout
Power
Switch
The next period is the “off” period of the power switch.
When the power switch turns off, the voltage across the
inductor reverses its polarity and is clamped at one diode
voltage drop below ground by the catch diode. Current now
flows through the catch diode thus maintaining the load
current loop. This removes the stored energy from the
inductor. The inductor current during this time is:
IL(off) +ǒVout –V
DǓtoff
L
This period ends when the power switch is once again
turned on. Regulation of the converter is accomplished by
varying the duty cycle of the power switch. It is possible to
describe the duty cycle as follows:
d+ton
T, where T is the period of switching.
For the buck converter with ideal components, the duty
cycle can also be described as:
d+Vout
Vin
Figure 18 shows the buck converter idealized waveforms
of the catch diode voltage and the inductor current.
Figure 18. Buck Converter Idealized Waveforms
Power
Switch
Power
Switch
Off
Power
Switch
Off
Power
Switch
On
Power
Switch
On
Von(SW)
VD(FWD)
Time
Time
ILoad(AV)
Imin
Ipk
Diode Diode
Power
Switch
Diode VoltageInductor Current
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Procedure (Fixed Output Voltage Version) In order to simplify the switching regulator design, a step−by−step design
procedure and example is provided.
Procedure Example
Given Parameters:
Vout = Regulated Output Voltage (3.3 V, 5.0 V, 12 V or 15 V)
Vin(max) = Maximum Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 5.0 V
Vin(max) = 15 V
ILoad(max) = 0.4 A
1. Controller IC Selection
According to the required input voltage, output voltage and
current, select the appropriate type of the controller IC output
voltage version.
1. Controller IC Selection
According to the required input voltage, output voltage,
current polarity and current value, use the LM2574−5
controller IC.
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin Gnd. This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
2. Input Capacitor Selection (Cin)
A 22 mF, 25 V aluminium electrolytic capacitor located near
to the input and ground pins provides sufficient bypassing.
3. Catch Diode Selection (D1)
A.Since the diode maximum peak current exceeds the
regulator maximum load current, the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design the diode should have a
current rating equal to the maximum current limit of the
LM2574 to be able to withstand a continuous output short.
B.The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
3. Catch Diode Selection (D1)
A.For this example the current rating of the diode is 1.0 A.
B.Use a 20 V 1N5817 Schottky diode, or any of the
suggested fast recovery diodes shown in Table 1.
4. Inductor Selection (L1)
A.According to the required working conditions, select the
correct inductor value using the selection guide from
Figures 19 to 23.
B.From the appropriate inductor selection guide, identify the
inductance region intersected by the Maximum Input Voltage
line and the Maximum Load Current line. Each region is
identified by an inductance value and an inductor code.
C.Select an appropriate inductor from the several different
manufacturers part numbers listed in Table 2. The designer
must realize that the inductor current rating must be higher
than the maximum peak current flowing through the inductor.
This maximum peak current can be calculated as follows:
where ton is the “on” time of the power switch and
For additional information about the inductor, see the inductor
section in the “EXTERNAL COMPONENTS” section of this
data sheet.
Ip(max)+ILoad(max))ǒVin *VoutǓton
2L
ton +Vout
Vin x1.0
fosc
4. Inductor Selection (L1)
A.Use the inductor selection guide shown in Figure 20.
B.From the selection guide, the inductance area
intersected by the 15 V line and 0.4 A line is 330.
C.Inductor value required is 330 mH. From Table 2, choose
an inductor from any of the listed manufacturers.
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Procedure (Fixed Output Voltage Version) (continued) In order to simplify the switching regulator design, a step−by−step
design procedure and example is provided.
Procedure Example
5. Output Capacitor Selection (Cout)
A.Since the LM2574 is a forward−mode switching regulator
with voltage mode control, its open loop 2−pole−1−zero
frequency characteristic has the dominant pole−pair
determined by the output capacitor and inductor values. For
stable operation and an acceptable ripple voltage,
(approximately 1% of the output voltage) a value between
100 mF and 470 mF is recommended.
B.Due to the fact that the higher voltage electrolytic capacitors
generally have lower ESR (Equivalent Series Resistance)
numbers, the output capacitor’s voltage rating should be at
least 1.5 times greater than the output voltage. For a 5.0 V
regulator, a rating at least 8.0 V is appropriate, and a 10 V
or 16 V rating is recommended.
5. Output Capacitor Selection (Cout)
A.Cout = 100 mF to 470 mF standard aluminium electrolytic.
B.Capacitor voltage rating = 20 V.
Procedure (Adjustable Output Version: LM2574−ADJ)
Procedure Example
Given Parameters:
Vout = Regulated Output Voltage
Vin(max) = Maximum DC Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 24 V
Vin(max) = 40 V
ILoad(max) = 0.4 A
1. Programming Output Voltage
To select the right programming resistor R1 and R2 value (see
Figure 2) use the following formula:
where Vref = 1.23 V
Resistor R1 can be between 1.0 kW and 5.0 kW. (For best
temperature coefficient and stability with time, use 1% metal
film resistors).
Vout +Vrefǒ1.0 )R2
R1Ǔ
R2 +R1ǒVout
Vref *1.0Ǔ
1. Programming Output Voltage (selecting R1 and R2)
Select R1 and R2 :
Vout = 1.23 Select R1 = 1.0 kW
R2 = 18.51 kW, choose a 18.7 kW metal film resistor.
ǒ1.0 )R2
R1Ǔ
R2 +R1ǒVout
Vref *1.0Ǔ+1.0 kǒ10 V
1.23 V *1.0Ǔ
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin Gnd. This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
For additional information see input capacitor section in the
“EXTERNAL COMPONENTS” section of this data sheet.
2. Input Capacitor Selection (Cin)
A 22 mF aluminium electrolytic capacitor located near the
input and ground pin provides sufficient bypassing.
3. Catch Diode Selection (D1)
A.Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design, the diode should have a
current rating equal to the maximum current limit of the
LM2574 to be able to withstand a continuous output short.
B.The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
3. Catch Diode Selection (D1)
A. For this example, a 1.0 A current rating is adequate.
B.Use a 50 V MBR150 Schottky diode or any suggested
fast recovery diodes in Table 1.
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Procedure (Adjustable Output Version: LM2574−ADJ)
Procedure Example
4. Inductor Selection (L1)
A.Use the following formula to calculate the inductor Volt x
microsecond [V x ms] constant:
B.Match the calculated E x T value with the corresponding
number on the vertical axis of the Inductor Value Selection
Guide shown in Figure 23. This E x T constant is a measure
of the energy handling capability of an inductor and is
dependent upon the type of core, the core area, the number
of turns, and the duty cycle.
C.Next step is to identify the inductance region intersected by
the E x T value and the maximum load current value on the
horizontal axis shown in Figure 27.
D.From the inductor code, identify the inductor value. Then
select an appropriate inductor from Table 2. The inductor
chosen must be rated for a switching frequency of 52 kHz
and for a current rating of 1.15 x ILoad. The inductor current
rating can also be determined by calculating the inductor
peak current:
where ton is the “on” time of the power switch and
For additional information about the inductor, see the inductor
section in the “External Components” section of this data
sheet.
ton +Vout
Vin x1.0
fosc
Ip(max)+ILoad(max))ǒVin *VoutǓton
2L
ExT+(Vin *Vout)Vout
Vin x106
F[Hz]ƪVxmsƫ
4. Inductor Selection (L1)
A.
B.
C.ILoad(max) = 0.4 A
Inductance Region = 1000
D.Proper inductor value = 1000 mH
Choose the inductor from Table 2.
ExT+(40 *24) x 24
40 x1000
52 +105ƪVxmsƫ
ExT+185ƪVxmsƫ
Calculate E x T ƪVxmsƫconstant :
5. Output Capacitor Selection (Cout)
A.Since the LM2574 is a forward−mode switching regulator with
voltage mode control, its open loop 2−pole−1−zero frequency
characteristic has the dominant pole−pair determined by the
output capacitor and inductor values.
For stable operation, the capacitor must satisfy the following
requirement:
B.Capacitor values between 10 mF and 2000 mF will satisfy the
loop requirements for stable operation. To achieve an
acceptable output ripple voltage and transient response, the
output capacitor may need to be several times larger than the
above formula yields.
C.Due to the fact that the higher voltage electrolytic capacitors
generally have lower ESR (Equivalent Series Resistance)
numbers, the output capacitor’s voltage rating should be at
least 1.5 times greater than the output voltage. For a 5.0 V
regulator, a rating of at least 8.0 V is appropriate, and a 10 V
or 16V rating is recommended.
Cout w13,300 Vin(max)
Vout xL
ƪmHƫƪmFƫ
5. Output Capacitor Selection (Cout)
A.
To achieve an acceptable ripple voltage, select
Cout = 100 mF electrolytic capacitor.
Cout w13,300 x 40
24 x 1000 +22.2 mF
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ET, VOLTAGE TIME (V s)μ
Vin , MAXIMUM INPUT VOLTAGE (V) Vin , MAXIMUM INPUT VOLTAGE (V)
Vin , MAXIMUM INPUT VOLTAGE (V)Vin , MAXIMUM INPUT VOLTAGE (V)
IL, MAXIMUM LOAD CURRENT (A)
IL, MAXIMUM LOAD CURRENT (A)
IL, MAXIMUM LOAD CURRENT (A)
IL, MAXIMUM LOAD CURRENT (A)
Figure 19. LM2574−3.3
IL, MAXIMUM LOAD CURRENT (A)
Figure 20. LM2574−5
680
Figure 21. LM2574−12 Figure 22. LM2574−15
Figure 23. LM2574−ADJ
150
470
220
100
330
1000
330
680
470
150
220
2200
470
1500
1000
330
680
220
2200
470
1500
1000
680
2200
470
1500
1000
330
680
220
150
100
68
LM2574 Series Buck Regulator Design Procedures (continued)
Indicator Value Selection Guide (For Continuous Mode Operation)
60
20
15
12
10
9.0
8.0
7.0
6.0
5.0
60
30
20
15
12
10
9.0
8.0
7.0
60
40
30
25
20
18
17
16
15
14
60
40
30
25
22
20
19
18
17
250
200
150
100
80
60
50
40
30
20
15
10
0.50.40.30.20.150.1 0.50.40.30.20.150.1
0.50.40.30.20.150.1 0.50.40.30.20.150.1
0.50.40.30.20.150.1
330
220
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Table 1. Diode Selection Guide gives an overview about through−hole diodes for
an effective design. Device listed in bold are available from ON Semiconductor
VR
1.0 Amp Diodes
Schottky Fast Recovery
20 V 1N5817
MBR120P
MUR110
(rated to 100 V)
30 V 1N5818
MBR130P
40 V 1N5819
MBR140P
50 V MBR150
60 V MBR160
Table 2. Inductor Selection Guide
Inductor
Value Pulse Engineering Tech 39 Renco NPI
68 mH*55 258 SN RL−1284−68 NP5915
100 mH*55 308 SN RL−1284−100 NP5916
150 mH52625 55 356 SN RL−1284−150 NP5917
220 mH52626 55 406 SN RL−1284−220 NP5918/5919
330 mH52627 55 454 SN RL−1284−330 NP5920/5921
470 mH52628 * RL−1284−470 NP5922
680 mH52629 55 504 SN RL−1284−680 NP5923
1000 mH52631 55 554 SN RL−1284−1000 *
1500 mH* * RL−1284−1500 *
2200 mH* * RL−1284−2200 *
* : Contact Manufacturer
Table 3. Example of Several Inductor Manufacturers Phone/Fax Numbers
Pulse Engineering Inc. Phone
Fax + 1−619−674−8100
+ 1−619−674−8262
Pulse Engineering Inc. Europe Phone
Fax + 353−9324−107
+ 353−9324−459
Renco Electronics Inc. Phone
Fax + 1−516−645−5828
+ 1−516−586−5562
Tech 39 Phone
Fax + 33−1−4115−1681
+ 33−1−4709−5051
NPI/APC Phone
Fax + 44−634−290−588
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EXTERNAL COMPONENTS
Input Capacitor (Cin)
The Input Capacitor Should Have a Low ESR
For stable operation of the switch mode converter a low
ESR (Equivalent Series Resistance) aluminium or solid
tantalum bypass capacitor is needed between the input pin
and the ground pin, to prevent large voltage transients from
appearing at the input. It must be located near the regulator
and use short leads. With most electrolytic capacitors, the
capacitance value decreases and the ESR increases with
lower temperatures. For reliable operation in temperatures
below −25°C larger values of the input capacitor may be
needed. Also paralleling a ceramic or solid tantalum
capacitor will increase the regulator stability at cold
temperatures.
RMS Current Rating of Cin
The important parameter of the input capacitor is the RMS
current rating. Capacitors that are physically large and have
large surface area will typically have higher RMS current
ratings. For a given capacitor value, a higher voltage
electrolytic capacitor will be physically lar ger than a lower
voltage capacitor, and thus be able to dissipate more heat to
the surrounding air, and therefore will have a higher RMS
current rating. The consequences of operating an
electrolytic capacitor beyond the RMS current rating is a
shortened operating life. In order to assure maximum
capacitor operating lifetime, the capacitors RMS ripple
current rating should be:
Irms u1.2 x d x ILoad
where d is the duty cycle, for a continuous mode buck
regulator
d+ton
T+Vout
Vin
and d+ton
T+|Vout|
|Vout|)Vin
for a buck−boost regulator.
Output Capacitor (Cout)
For low output ripple voltage and good stability, low ESR
output capacitors are recommended. An output capacitor
has two main functions: it filters the output and provides
regulator loop stability. The ESR of the output capacitor a nd
the peak−to−peak value of the inductor ripple current are the
main factors contributing to the output ripple voltage value.
Standard aluminium electrolytics could be adequate for
some applications but for quality design, low ESR types are
recommended.
An aluminium electrolytic capacitors ESR value is
related to many factors, such as the capacitance value, the
voltage rating, the physical size and the type of construction.
In most cases, the higher voltage electrolytic capacitors have
lower ESR value. Often capacitors with much higher
voltage ratings may be needed to provide low ESR values,
that are required for low output ripple voltage.
The Output Capacitor Requires an ESR Value that has
an Upper and Lower Limit
As mentioned above, a low ESR value is needed for low
output ripple voltage, typically 1% to 2% of the output
voltage. But if the selected capacitor’s ESR is extremely low
(below 0.03 W), there is a possibility of an unstable feedback
loop, resulting in oscillation at the output. This situation can
occur when a tantalum capacitor, that can have a very low
ESR, is used as the only output capacitor.
At Low Temperatures, Put in Parallel Aluminium
Electrolytic Capacitors with Tantalum Capacitors
Electrolytic capacitors are not recommended for
temperatures below −25°C. The ESR rises dramatically at
cold temperatures and typically rises 3 times at −25°C and
as much as 10 times at −40°C. Solid tantalum capacitors
have much better ESR spec at cold temperatures and are
recommended for temperatures below −25°C. They can be
also used in parallel with aluminium electrolytics. The value
of the tantalum capacitor should be about 10% or 20% of the
total capacitance. The output capacitor should have at least
50% higher RMS ripple current rating at 52 kHz than the
peak−to−peak inductor ripple current.
Catch Diode
Locate the Catch Diode Close to the LM2574
The LM2574 is a step−down buck converter, it requires a
fast diode to provide a return path for the inductor current
when the switch turns off. This diode must be located close
to the LM2574 using short leads and short printed circuit
traces to avoid EMI problems.
Use a Schottky or a Soft Switching
Ultra−Fast Recovery Diode
Since the rectifier diodes are very significant source of
losses within switching power supplies, choosing the
rectifier that best fits into the converter design is an
important process. Schottky diodes provide the best
performance because of their fast switching speed and low
forward voltage drop.
They provide the best efficiency especially in low output
voltage applications (5.0 V and lower). Another choice
could be Fast−Recovery, or Ultra−Fast Recovery diodes. It
has to be noted, that some types of these diodes with an
abrupt turnoff characteristic may cause instability or EMI
troubles.
A fast−recovery diode with soft recovery characteristics
can better fulfill some quality, low noise design
requirements. Table 1 provides a list of suitable diodes for
the LM2574 regulator. Standard 50/60 Hz rectifier diodes,
such as the 1N4001 series or 1N5400 series are NOT
suitable.
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Inductor
The magnetic components are the cornerstone of all
switching power supply designs. The style of the core and
the winding technique used in the magnetic component’s
design have a great influence on the reliability of the overall
power supply.
Using an improper or poorly designed inductor can cause
high voltage spikes generated by the rate of transitions in
current within the switching power supply, and the
possibility of core saturation can arise during an abnormal
operational mode. Voltage spikes can cause the
semiconductors to enter avalanche breakdown and the part
can instantly fail if enough energy is applied. It can also
cause significant RFI (Radio Frequency Interference) and
EMI (Electro−Magnetic Interference) problems.
Continuous and Discontinuous Mode of Operation
The LM2574 step−down converter can operate in both the
continuous and the discontinuous modes of operation. The
regulator works in the continuous mode when loads are
relatively heavy, the current flows through the inductor
continuously and never falls to zero. Under light load
conditions, the circuit will be forced to the discontinuous
mode when inductor current falls to zero for certain period
of time (see Figure 24 and Figure 25). Each mode has
distinctively different operating characteristics, which can
affect the regulator performance and requirements. In many
cases the preferred mode of operation is the continuous
mode. It o ffers greater output power, lower peak currents in
the switch, inductor and diode, and can have a lower output
ripple voltage. On the other hand it does require larger
inductor values to keep the inductor current flowing
continuously, especially at low output load currents and/or
high input voltages.
To simplify the inductor selection process, an inductor
selection guide for the LM2574 regulator was added to this
data sheet (Figures 19 through 23). This guide assumes that
the regulator is operating in the continuous mode, and
selects an inductor that will allow a peak−to−peak inductor
ripple current to be a certain percentage of the maximum
design load current. This percentage is allowed to change as
different design load currents are selected. For light loads
(less than approximately 0.2 A) it may be desirable to
operate the regulator in the discontinuous mode, because t he
inductor value and size can be kept relatively low.
Consequently, the percentage of inductor peak−to−peak
current increases. This discontinuous mode of operation is
perfectly acceptable for this type of switching converter.
Any buck regulator will be forced to enter discontinuous
mode if the load current is light enough.
Selecting the Right Inductor Style
Some important considerations when selecting a core type
are core material, cost, the output power of the power supply,
the physical volume the inductor must fit within, and the
amount of EMI (Electro−Magnetic Interference) shielding
that the core must provide. There are many different styles
of inductors available, such as pot core, E−core, toroid and
bobbin core, as well as different core materials such as
ferrites and powdered iron from different manufacturers.
For high quality design regulators the toroid core seems to
be the best choice. Since the magnetic flux is contained
within the core, it generates less EMI, reducing noise
problems in sensitive circuits. The least expensive is the
bobbin core type, which consists of wire wound on a ferrite
rod core. This type of inductor generates more EMI due to
the fact that its core is open, and the magnetic flux is not
contained within the core.
When multiple switching regulators are located on the
same printed circuit board, open core magnetics can cause
interference between two or more of the regulator circuits,
especially a t high currents due to mutual coupling. A toroid,
pot core or E−core (closed magnetic structure) should be
used in such applications.
Do Not Operate an Inductor Beyond its Maximum
Rated Current
Exceeding an inductors maximum current rating may
cause the inductor to overheat because of the copper wire
losses, o r the core may saturate. Core saturation occurs when
the flux density is too high and consequently the cross
sectional area of the core can no longer support additional
lines of magnetic flux.
This causes the permeability of the core to drop, the
inductance value decreases rapidly and the inductor begins
to look mainly resistive. It has only the dc resistance of the
winding. This can cause the switch current to rise very
rapidly and force the LM2574 internal switch into
cycle−by−cycle current limit, thus reducing the dc output
load current. This can also result in overheating of the
inductor and/or the LM2574. Different inductor types have
different saturation characteristics, and this should be kept
in mind when selecting an inductor.
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HORIZONTAL TIME BASE: 5.0 ms/DIV
VERTRICAL RESOLUTION 200 mADV
Figure 24. Continuous Mode Switching
Current Waveforms
0.5 A
0 A
0.5 A
0 A
Power
Switch
Current
Waveform
Inductor
Current
Waveform
VERTICAL RESOLUTION 100 mADV
HORIZONTAL TIME BASE: 5.0 ms/DIV
Figure 25. Discontinuous Mode Switching
Current Waveforms
0.1 A
0 A
0.1 A
0 A
Power
Switch
Current
Waveform
Inductor
Current
Waveform
GENERAL RECOMMENDATIONS
Output Voltage Ripple and Transients
Source of the Output Ripple
Since the LM2574 is a switch mode power supply
regulator, its output voltage, if left unfiltered, will contain a
sawtooth ripple voltage at the switching frequency. The
output ripple voltage value ranges from 0.5% to 3% of the
output voltage. It is caused mainly by the inductor sawtooth
ripple current multiplied by the ESR of the output capacitor.
Short Voltage Spikes and How to Reduce Them
The regulator output voltage may also contain short
voltage spikes at the peaks of the sawtooth waveform (see
Figure 26). These voltage spikes are present because of the
fast switching action of the output switch, and the parasitic
inductance of the output filter capacitor. There are some
other important factors such as wiring inductance, stray
capacitance, a s well as the scope probe used to evaluate these
transients, all these contribute to the amplitude of these
spikes. To minimize these voltage spikes, low inductance
capacitors should be used, and their lead lengths must be
kept short. The importance of quality printed circuit board
layout design should also be highlighted.
HORIZONTAL TIME BASE: 5.0 ms/DIV
VERTRICAL RESOLUTION 20 mV/DIV
Voltage spikes caused by switching action of the output
switch and the parasitic inductance of the output capacitor
Figure 26. Output Ripple Voltage Waveforms
Unfiltered
Output
Voltage
Filtered
Output
Voltage
Minimizing the Output Ripple
In order t o m inimize t he o utput ripple v oltage i t i s p ossible
to enlarge the inductance value of the inductor L1 and/or to
use a l ar ger v alue o utput c apacitor. There i s a lso a nother w ay
to smooth the output by means of an additional LC filter
(20 mH, 100 mF), that can be added to the output (see
Figure 35) to further reduce the amount of output ripple and
transients. With such a filter it is possible to reduce the
output ripple voltage transients 10 times or more. Figure 26
shows the dif ference between filtered and unfiltered output
waveforms of the regulator shown in Figure 34.
The upper waveform is from the normal unfiltered output
of the converter , while the lower waveform shows the output
ripple voltage filtered by an additional LC filter.
Heatsinking and Thermal Considerations
The LM2574 is available in both 8−pin DIP and SO−16L
packages. W hen us ed in t he t ypical a pplication t he c opper l ead
frame conducts the majority of the heat from the die, through
the leads, to the printed circuit copper. The copper and the
board are the heatsink for this package and the other heat
producing components, such as the catch diode and inductor.
For the best thermal performance, wide copper traces
should be used and all ground and unused pins should be
soldered to generous amounts of printed circuit board
copper, such as a ground plane. Large areas of copper
provide the best transfer of heat to the surrounding air. One
exception t o this is the output (switch) pin, which should not
have large areas of copper in order to minimize coupling to
sensitive circuitry.
Additional improvement in heat dissipation can be
achieved even by using of double sided or multilayer boards
which can provide even better heat path to the ambient.
Using a socket for the 8−pin DIP package is not
recommended because socket represents an additional
thermal resistance, and as a result the junction temperature
will be higher.
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Since the current rating of the LM2574 is only 0.5 A, the
total package power dissipation for this switcher is quite
low, ranging from approximately 0.1 W up to 0.75 W under
varying conditions. In a carefully engineered printed circuit
board, the through−hole DIP package can easily dissipate u p
to 0.75 W, even at ambient temperatures of 60°C, and still
keep the maximum junction temperature below 125°C.
Thermal Analysis and Design
The following procedure must be performed to determine
the operating junction temperature. First determine:
1. PD(max) maximum regulator power dissipation in
the application.
2. TA(max) maximum ambient temperature in the
application.
3. TJ(max) maximum allowed junction temperature
(125°C for the LM2574). For a conservative
design, the maximum junction temperature
should not exceed 110°C to assure safe
operation. For every additional +10°C
temperature rise that the junction must
withstand, the estimated operating lifetime
of the component is halved.
4. RqJC package thermal resistance junction−case.
5. RqJA package thermal resistance junction−ambient.
(Refer to Absolute Maximum Ratings on page 2 of this data
sheet or RqJC and RqJA values).
The following formula is to calculate the approximate
total power dissipated by the LM2574:
PD = (Vin x IQ) + d x ILoad x Vsat
where d is the duty cycle and for buck converter
d+ton
T+VO
Vin,
IQ (quiescent current) and Vsat can be found in the
LM2574 data sheet,
Vin is minimum input voltage applied,
VO is the regulator output voltage,
ILoad is the load current.
D1
MBR150
L1
68 mH
Output
7
1
Feedback
8.0 to 25 V
Unregulated
DC Input
Cin
22 mF
5
34ON
/OFFPwr
Gnd
+Vin
−12 V @ 100 mA
Regulated
Output
Cout
680 mF
LM2574−12
2 Sig
Gnd
Figure 27. Inverting Buck−Boost Develops −12 V
(6)
(12)
(4)
(5)
(3)
(14)
The dynamic switching losses during turn−on and
turn−off can be neglected if a proper type catch diode is used.
The junction temperature can be determined by the
following expression:
TJ = (RqJA)(PD) + TA
where (RqJA)(PD) represents the junction temperature rise
caused by the dissipated power and TA is the maximum
ambient temperature.
Some Aspects That can Influence Thermal Design
It should be noted that the package thermal resistance and
the junction temperature rise numbers are all approximate,
and there are many factors that will affect these numbers,
such as PC board size, shape, thickness, physical position,
location, board temperature, as well as whether the
surrounding air is moving or still. At higher power levels the
thermal resistance decreases due to the increased air current
activity.
Other factors are trace width, total printed circuit copper
area, copper thickness, single− or double−sided, multilayer
board, the amount of solder on the board or even color of the
traces.
The size, quantity and spacing of other components on the
board can also influence its effectiveness to dissipate the
heat. Some of them, like the catch diode or the inductor will
generate some additional heat.
ADDITIONAL APPLICATIONS
Inverting Regulator
An inverting buck−boost regulator using the LM2574−12
is shown in Figure 27. This circuit converts a positive input
voltage to a negative output voltage with a common ground
by bootstrapping the regulators ground to the negative
output voltage. By grounding the feedback pin, the regulator
senses the inverted output voltage and regulates it.
In t his e xample t he L M2574−12 i s u sed t o g enerate a 12 V
output. The maximum input voltage in this case cannot
exceed 2 8 V because the maximum voltage appearing across
the regulator is the absolute sum of the input and output
voltages and this must be limited to a maximum of 40 V.
This circuit configuration is able to deliver approximately
0.1 A to the output when the input voltage is 8.0 V or higher.
At lighter loads the minimum input voltage required drops
to approximately 4.7 V, because the buck−boost regulator
topology can produce an output voltage that, in its absolute
value, is either greater or less than the input voltage.
Since the switch currents in this buck−boost configuration
are higher than in the standard buck converter topology, the
available output current is lower.
This type of buck−boost inverting regulator can also
require a larger amount of startup input current, even for
light loads. This may overload an input power source with
a current limit less than 0.6 A.
Because o f the relatively high startup currents required by
this inverting regulator topology, the use of a delayed startup
or an undervoltage lockout circuit is recommended.
While using a delayed startup arrangement, the input
capacitor can charge up to a higher voltage before the
switch−mode regulator begins to operate.
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The high input current needed for startup is now partially
supplied by the input capacitor Cin.
Design Recommendations:
The inverting regulator operates in a different manner
than the buck converter and so a different design procedure
has to be used to select the inductor L1 or the output
capacitor Cout.
The output capacitor values must be larger than what is
normally required for buck converter designs. Low input
voltages or high output currents require a lar ge value output
capacitor (in the range of thousands of mF).
The recommended range of inductor values for the
inverting converter design is between 68 mH and 220 mH. To
select an inductor with an appropriate current rating, the
inductor peak current has to be calculated.
D1
MBR150
L1
68 mH
Output
7
1
Feedback
12 to 25 V
Unregulated
DC Input
Cin
22 mF
/50 V
5
43ON/OFF Pwr
Gnd
+Vin
−12 V @ 100 mA
Regulated
Output
Cout
680 mF
/16 V
LM2574−12
C1
0.1 mF
R1
47 k R2
47 k
2 Sig
Gnd
Figure 28. Inverting Buck−Boost Regulator with
Delayed Startup
(3)
(12) (14)
(5)
(6) (4)
The following formula is used to obtain the peak inductor
current:
Ipeak [
ILoadǒVin )|VO|Ǔ
Vin )Vin xt
on
2L1
where ton +|VO|
Vin )|VO|x1.0
fosc
, and fosc = 52 kHz.
Under normal continuous inductor current operating
conditions, the worst case occurs when Vin is minimal.
It has been already mentioned above, that in some
situations, the delayed startup or the undervoltage lockout
features could be very useful. A delayed startup circuit
applied to a buck−boost converter is shown in Figure 28.
Figure 34 in the “Undervoltage Lockout” section describes
an undervoltage lockout feature for the same converter
topology.
With the inverting configuration, the use of the ON/OFF
pin requires some level shifting techniques. This is caused
by the fact, that the ground pin of the converter IC is no
longer at ground. Now, the ON/OFF pin threshold voltage
(1.3 V approximately) has to be related to the negative
output voltage level. There are many different possible
shutdown methods, two of them are shown in Figures 29
and 30.
LM2574−XX
5
2
and
4
3 Gnds
Pins
ON/OFF
+Vin
R2
47 k
Cin
22 mF
NOTE: This picture does not show the complete circuit.
R1
47 k
R3
470
Shutdown
Input
MOC8101
−Vout
Off
On
5.0 V
0
+Vin
Figure 29. Inverting Buck−Boost Regulator Shutdow
n
Circuit Using an Optocoupler
(5)
(12)
(4)
and
(6)
NOTE: This picture does not show the complete circuit.
R2
5.6 k
Q1
2N3906
LM2574−XX
5
2
and
4
3 Gnds
Pins
ON/OFF
R1
12 k −Vout
+Vin
Shutdown
Input
Off
On
+V
0
+Vin
Cin
22 mF
Figure 30. Inverting Buck−Boost Regulator Shutdow
n
Circuit Using a PNP Transistor
(5)
(12)
(4)
and
(6)
Negative Boost Regulator
This example is a variation of the buck−boost topology
and it is called negative boost regulator. This regulator
experiences relatively high switch current, especially at low
input voltages. The internal switch current limiting results in
lower output load current capability.
LM2574, NCV2574
http://onsemi.com
20
The circuit in Figure 31 shows the negative boost
configuration. The input voltage in this application ranges
from −5.0 to −12 V and provides a regulated −12 V output.
If the input voltage is greater than −12 V, the output will rise
above −12 V accordingly, but will not damage the regulator.
1N5817
330 mH
Output
7
1
Feedback
Vout = −12 V
Load Current
60 mA for Vin = −5.2 V
120 mA for Vin = −7.0 V
Vin
L1
D1
Cout
1000 mF
Cin
22 mF
LM2574−12
5
34 ON/OFFPwr
Gnd
+Vin
2Sig
Gnd
−5.0 to −12 V
Figure 31. Negative Boost Regulator
(5)
(12)
(3)
(14)
(4)(6)
Design Recommendations:
The same design rules as for the previous inverting
buck−boost converter can be applied. The output capacitor
Cout must be chosen lar ger than what would be required for
a standard buck converter. Low input voltages or high output
currents require a large value output capacitor (in the range
of thousands of mF). The recommended range of inductor
values for the negative boost regulator is the same as for
inverting converter design.
Another important point is that these negative boost
converters cannot provide any current limiting load
protection in the event of a short in the output so some other
means, such as a fuse, may be necessary to provide the load
protection.
Delayed Startup
There are some applications, like the inverting regulator
already mentioned above, which require a higher amount of
startup current. In such cases, if the input power source is
limited, this delayed startup feature becomes very useful.
To provide a time delay between the time when the input
voltage is applied and the time when the output voltage
comes up, the circuit in Figure 32 can be used. As the input
voltage is applied, the capacitor C1 charges up, and the
voltage across the resistor R2 falls down. When the voltage
on the ON/OFF pin falls below the threshold value 1.3 V, the
regulator starts up. Resistor R1 is included to limit the
maximum voltage applied to the ON/OFF pin. It reduces t h e
power supply noise sensitivity, and also limits the capacitor
C1 discharge current, but its use is not mandatory.
When a high 50 Hz or 60 Hz (100 Hz or 120 Hz
respectively) ripple voltage exists, a long delay time can
cause some problems by coupling the ripple into the
ON/OFF pin, the regulator could be switched periodically
on and off with the line (or double) frequency.
R1
47 k
LM2574−XX
5
2
and
4
3 Gnds
Pins
ON/OFF
R2
47 k
+Vin
+Vin
C1
0.1 mF
Cin
22 mF
NOTE: This picture does not show the complete circuit.
Figure 32. Delayed Startup Circuitry
(5)
(12)
(4)
and
(6)
Undervoltage Lockout
Some applications require the regulator to remain of f until
the input voltage reaches a certain threshold level. Figure 33
shows an undervoltage lockout circuit applied to a buck
regulator. A version of this circuit for buck−boost converter
is shown in Figure 34. Resistor R3 pulls the ON/OFF pin
high and keeps the regulator off until the input voltage
reaches a predetermined threshold level, which is
determined by the following expression:
Vth [VZ1 )ǒ1.0 )R2
R1ǓVBE(Q1)
R1
10 k
Z1
1N5242B
R2
10 k
Q1
2N3904
R3
47 k
Cin
22 mF
LM2574−XX
5
2
and
4
3 Gnds
Pins
ON/OFF
+Vin
+Vin
NOTE: This picture does not show the complete circuit.
Figure 33. Undervoltage Lockout Circuit for
Buck Converter
(5)
(12)
(4)
and
(6)