Figure 1. Typical Forward Converter Application.
DPA423-426
DPA-Switch® Family
Highly Integrated DC-DC Converter ICs for
Power over Ethernet & Telecom DC-DC
May 2006
Product Highlights
Highly Integrated Solution
• Eliminates 20-50 external components–saves space, cost
• Integrates 220 V high frequency MOSFET, PWM control
• Lower cost plastic DIP surface mount (G package) and
through-hole (P package) options for designs ≤35 W
• Thermally efficient MO-169-7C (S-PAK) and
TO-263-7C (R package) options for high power applications
Superior Performance and Flexibility
• Eliminates all external current sensing circuitry
• Built-in auto-restart for output overload/open loop protection
• Pin selectable 300/400 kHz fixed frequency
• Wide input (line) voltage range: 16-75 VDC
• Externally programmable current limit
• Source connected tab reduces EMI
• Line under-voltage (UV) detection: meets ETSI standards
• Line overvoltage (OV) shutdown protection
• UV/OV limits gate drive voltage for synchronous rectification
• Fully integrated soft-start for minimum stress/overshoot
• Supports forward or flyback topology
• Cycle skipping: regulation to zero load without pre-load
• Hysteretic thermal shutdown for automatic fault recovery
• RoHS compliant P, G and S package options
EcoSmart®– Energy Efficient
• Extremely low consumption at no load
• Cycle skipping at light load for high standby efficiency
Applications
• PoE applications, VoIP phones, WLAN, security cameras
• Telco central office equipment: xDSL, ISDN, PABX
• Distributed power architectures (24 V/48 V bus)
• Industrial controls
Description
The DPA-Switch IC family is a highly integrated solution for
DC-DC conversion applications for 16-75 VDC input.
DPA-Switch uses the same proven topology as TOPSwitch,
cost effectively integrating a power MOSFET, PWM
control, fault protection and other control circuitry onto
a single CMOS chip. High performance features are
enabled with three user configurable pins. Hysteretic
thermal shutdown is also provided. In addition, all critical
Table 1. Notes: 1. Maximum output power is limited by device internal
current limit. 2. See Applications Considerations section for complete
description of assumptions and for output powers with other input voltage
ranges. 3. For device dissipation of 1.5 W or below, use P or G packages.
Device dissipation above 1.5 W is possible with S and R packages.
4. Packages: P: DIP-8, G: SMD-8, R: TO-263-7C, S: MO-169-7C. For lead-free
package options, see Part Ordering Information. 5. Available in S and R
package only. 6. Due to higher switching losses, the DPA425 may not deliver
additional power compared to a smaller device.
PI-2770-032002
D
S
C
DPA-Switch
VIN
VO
F
X
L
CONTROL
SENSE
CIRCUIT
RESET/
CLAMP
CIRCUIT
OUTPUT POWER TABLE
36-75 VDC INPUT RANGE (FORWARD)2
Total Device
Dissipation3
PRODUCT4
0.5 W 1 W 2.5 W 4 W 6 W
Max
Power
Output1
DPA423 12 W 16 W - - - 18 W
DPA424 16 W 23 W 35 W - - 35 W
DPA425 23 W 32 W 50 W 62 W 70 W
DPA426525 W 35 W 55 W 70 W 83 W 100 W
36-75 VDC INPUT RANGE (FLYBACK)2
Total Device
Dissipation3
PRODUCT4
0.5 W 0.75 W 1 W 1.5 W
Max
Power
Output1
DPA423 9 W 13 W - - 13 W
DPA424 10 W 14.5 W 18 W 24 W 26 W
DPA425 -6-6-625.5 W 52 W
®
parameters (i.e. current limit, frequency, PWM gain) have
tight temperature and absolute tolerance, to simplify design
and reduce system cost.
DPA423-426
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2
Section List
Functional Block Diagram ....................................................................................................................................... 3
Pin Functional Description ...................................................................................................................................... 3
DPA-Switch Family Functional Description ............................................................................................................ 4
CONTROL (C) Pin Operation .................................................................................................................................. 4
Oscillator and Switching Frequency ....................................................................................................................... 5
Pulse Width Modulator & Maximum Duty Cycle ..................................................................................................... 6
Minimum Duty Cycle and Cycle Skipping .............................................................................................................. 6
Error Amplifier ......................................................................................................................................................... 6
On-chip Current Limit with External Programmability ............................................................................................. 6
Line Under-Voltage Detection (UV) ......................................................................................................................... 6
Line Overvoltage Shutdown (OV) ........................................................................................................................... 7
Line Feed-Forward with DCMAX Reduction .............................................................................................................. 7
Remote ON/OFF ..................................................................................................................................................... 7
Synchronization ...................................................................................................................................................... 8
Soft-Start ................................................................................................................................................................. 8
Shutdown/Auto-Restart ........................................................................................................................................... 8
Hysteretic Over-Temperature Protection ................................................................................................................. 8
Bandgap Reference ............................................................................................................................................... 8
High-Voltage Bias Current Source .......................................................................................................................... 8
Using Feature Pins ..................................................................................................................................................... 9
FREQUENCY (F) Pin Operation .............................................................................................................................. 9
LINE-SENSE (L) Pin Operation ............................................................................................................................... 9
EXTERNAL CURRENT LIMIT (X) Pin Operation ...................................................................................................... 9
Typical Uses of FREQUENCY (F) Pin ...................................................................................................................... 12
Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins ...................................................... 12
Application Examples .............................................................................................................................................. 15
Key Application Considerations ............................................................................................................................. 18
DPA-Switch Design Considerations ...................................................................................................................... 18
DPA-Switch Layout Considerations ...................................................................................................................... 19
Quick Design Checklist ........................................................................................................................................ 20
Design Tools ......................................................................................................................................................... 20
Product Specifications and Test Conditions ......................................................................................................... 22
Typical Performance Characteristics .................................................................................................................. 28
Part Ordering Information ....................................................................................................................................... 31
Package Outlines .................................................................................................................................................... 32
DPA423-426
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Figure 2. Functional Block Diagram.
PI-2760-070501
SHUTDOWN/
AUTO-RESTART
PWM
COMPARATOR
CLOCK
SAW
300/400 kHz
CONTROLLED
TURN-ON
GATE DRIVER
CURRENT LIMIT
COMPARATOR
INTERNAL UV
COMPARATOR
INTERNAL
SUPPLY
5.8 V
4.8 V
SOURCE (S)
S
R
Q
DMAX
STOP SOFT-
START
-
+
CONTROL (C)
LINE-SENSE (L)
EXTERNAL
CURRENT LIMIT (X)
FREQUENCY (F)
-
+
5.8 V
1 V
IFB
RE
ZC
VC
+
-
LEADING
EDGE
BLANKING
÷ 8
1
HYSTERETIC
THERMAL
SHUTDOWN
SHUNT REGULATOR/
ERROR AMPLIFIER
+
-
DRAIN (D)
ON/OFF
SOFT
START
DCMAX
VBG
DCMAX
VBG + VT
0
OV/UV
VI (LIMIT)
CURRENT
LIMIT
ADJUST
LINE
SENSE
SOFT START
CYCLE
SKIPPING
STOP LOGIC
OSCILLATOR
Figure 3. Pin Configuration (top view).
Pin Functional Description
DRAIN (D) Pin:
High voltage power MOSFET drain output. The internal startup
bias current is drawn from this pin through a switched high-
voltage current source. Internal current limit sense point for
drain current.
CONTROL (C) Pin:
Error amplifier and feedback current input pin for duty cycle
control. Internal shunt regulator connection to provide
internal bias current during normal operation. It is also used
as the connection point for the supply bypass and auto-restart/
compensation capacitor.
LINE-SENSE (L) Pin:
Input pin for overvoltage (OV), under-voltage (UV) lock out,
line feed-forward with the maximum duty cycle (DCMAX)
reduction, remote ON/OFF and synchronization. A connection
to SOURCE pin disables all functions on this pin.
EXTERNAL CURRENT LIMIT (X) Pin:
Input pin for external current limit adjustment and remote
ON/OFF. A connection to SOURCE pin disables all functions
on this pin.
FREQUENCY (F) Pin:
Input pin for selecting switching frequency: 400 kHz if
connected to SOURCE pin and 300 kHz if connected to
CONTROL pin.
SOURCE (S) Pin:
Output MOSFET source connection for the power return.
Primary side control circuit common and reference point.
Tab Internally Connected
to SOURCE Pin
(See layout considerations)
R Package
(TO-263-7C)
S-PAK
(MO-169-7C)
1 2 3 4 5 7
C L X S F D
1 2 3 4 5 7
C L X S F D
FS
L
X
S
D
S
C
4
2
3
1
P Package (DIP-8)
G Package (SMD-8)
5
7
8
6
PI-4030-071305
DPA423-426
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DPA-Switch Family Functional
Description
DPA-Switch is an integrated switched mode power supply
chip that converts a current at the control input to a duty cycle
at the open drain output of a high voltage power MOSFET.
During normal operation the duty cycle of the power MOSFET
decreases linearly with increasing CONTROL pin current as
shown in Figure 4. A patented high-voltage CMOS technology
allows both the high-voltage power MOSFET and all the low
voltage control circuitry to be cost effectively integrated onto
a single monolithic chip.
In addition to the standard TOPSwitch features, such as the
high-voltage start-up, the cycle-by-cycle current limiting, loop
compensation circuitry, auto-restart and thermal shutdown,
DPA-Switch also offers many advanced features that reduce
system cost and increase power supply performance and design
flexibility. Following is a summary of the advanced features:
1. A fully integrated 5 ms soft-start limits peak currents and
voltages during start-up and reduces or eliminates output
overshoot in most applications.
2. A 75% maximum duty cycle (DCMAX) together with the
line feed-forward with DCMAX reduction feature makes
DPA-Switch well suited for both flyback and forward
topologies.
3. High switching frequency (400 kHz/300 kHz, pin selectable)
allows the use of smaller size transformers and offers high
bandwidth for power supply control loop.
4. Cycle skipping operation at light load minimizes standby
power consumption (typically <10 mA input current).
5. Line under-voltage ensures glitch free operations at both
power-up and power-down and is tightly toleranced over
process and temperature to meet system level requirements
common in DC to DC converters (e.g. ETSI).
6. Line overvoltage protects DPA-Switch against excessive
input voltage and line surge.
7. External current limit adjustment allows the setting of the
current limit externally to a lower level near the operating
peak current and, if desired, further adjusts the level
gradually as line voltage rises. This makes possible an ideal
implementation of overload power limiting.
8. Synchronization function allows the synchronization of
DPA-Switch operation to an external lower frequency.
9. Remote ON/OFF feature permits DPA-Switch based power
supplies to be easily switched on/off using logic signals.
Maximum input current consumption is 2 mA in remote
OFF.
10. Hysteretic over-temperature shutdown provides automatic
recovery from thermal fault.
11. Tight absolute tolerances and small temperature variations
on switching frequency, current limit, and under-voltage
lock out threshold (UV).
Three pins, LINE-SENSE (L), EXTERNAL CURRENT LIMIT
(X) and FREQUENCY (F), are used to implement all the pin
controllable features. A resistor from the LINE-SENSE pin to DC
input bus implements line UV, line OV and line feed-forward with
DCMAX reduction. A resistor from the EXTERNAL CURRENT
LIMIT pin to the SOURCE pin sets current limit externally. In
addition, remote ON/OFF may be implemented through either
the LINE-SENSE pin or the EXTERNAL CURRENT LIMIT
pin depending on the polarity of the logic signal available as
well as other system specific considerations. Shorting both the
LINE-SENSE and the EXTERNAL CURRENT LIMIT pins to
the SOURCE pin disables line OV, line UV, line feed-forward
with DCMAX reduction, external current limit, remote ON/OFF
and synchronization. The FREQUENCY pin sets the switching
frequency to 400 kHz if connected to the SOURCE pin, or
300 kHz if connected to the CONTROL pin. This pin should
not be left open. Please refer to “Using Feature Pins” section for
detailed information regarding the proper use of those pins.
CONTROL (C) Pin Operation
The CONTROL pin is a low impedance node that is capable
of receiving a combined supply and feedback current. During
normal operation, a shunt regulator is used to separate the
feedback signal from the supply current. CONTROL pin voltage
VC is the supply voltage for the control circuitry including the
MOSFET gate driver. An external bypass capacitor closely
connected between the CONTROL and SOURCE pins is required
to supply the instantaneous gate drive current. The total amount
of capacitance connected to this pin also sets the auto-restart
timing as well as control loop compensation.
When the DC input voltage is applied to the DRAIN pin during
start-up, the MOSFET is initially off, and the CONTROL
pin capacitor is charged through the switched high voltage
current source connected internally between the DRAIN and
CONTROL pins. When the CONTROL pin voltage VC reaches
Figure 4. Relationship of Duty Cycle to CONTROL Pin Current.
PI-2761-112102
Duty Cycle (%)
IC (mA)
IL = 115 µA
IL < IL(DC) IC (SKIP)
Slope = PWM Gain
ICD1
IB
Auto-restart
75
4
42
DPA423-426
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approximately 5.8 V, the control circuitry is activated and the
soft-start begins. The soft-start circuit gradually increases
the duty cycle of the MOSFET from zero to the maximum
value over approximately 5 ms. The high voltage current
source is turned off at the end of the soft-start. If no external
feedback/supply current is fed into the CONTROL pin by the
end of the soft-start, the CONTROL pin will start discharging
in response to the supply current drawn by the control circuitry
and the gate current of the switching MOSFET driver. If the
power supply is designed properly, and no fault condition such
as open loop or overloaded output exists, the feedback loop
will close, providing external CONTROL pin current, before
the CONTROL pin voltage has had a chance to discharge to
the lower threshold voltage of approximately 4.8 V (internal
supply under-voltage lockout threshold). When the externally
fed current charges the CONTROL pin to the shunt regulator
voltage of 5.8 V, current in excess of the consumption of the
chip is shunted to SOURCE through resistor RE as shown in
Figure 2. This current flowing through RE controls the duty cycle
of the power MOSFET to provide closed loop regulation. The
shunt regulator has a finite low output impedance ZC that sets
the gain of the error amplifier when used in a primary feedback
configuration. The dynamic impedance ZC of the CONTROL
pin together with the external CONTROL pin capacitance sets
the dominant pole for the control loop.
When a fault condition such as an open loop or overloaded output
prevents the flow of an external current into the CONTROL
pin, the capacitor on the CONTROL pin discharges towards
4.8 V. At 4.8 V auto-restart is activated which turns the output
MOSFET off and puts the control circuitry in a low current
standby mode. The high-voltage current source turns on and
charges the external capacitance again. A hysteretic internal
supply under-voltage comparator keeps VC within a window
of typically 4.8 V to 5.8 V by turning the high-voltage current
source on and off as shown in Figure 5. The auto-restart circuit
has a divide-by-8 counter that prevents the output MOSFET
from turning on again until eight discharge/charge cycles have
elapsed. This is accomplished by enabling the output MOSFET
only when the divide-by-8 counter reaches full count (S7).
The counter effectively limits DPA-Switch power dissipation
as well as the maximum power delivered to the power supply
output by reducing the auto-restart duty cycle to typically 4%.
Auto-restart mode continues until output voltage regulation is
again achieved through closure of the feedback loop.
Oscillator and Switching Frequency
The internal oscillator linearly charges and discharges an internal
capacitance between two voltage levels to create a sawtooth
waveform for the pulse width modulator. The oscillator sets
both the pulse width modulator latch and the current limit latch
at the beginning of each cycle.
The nominal switching frequency of 400 kHz was chosen to
minimize the transformer size and to allow faster power supply
loop response. The FREQUENCY pin, when shorted to the
CONTROL pin, lowers the switching frequency to 300 kHz,
which may be preferable in some applications such as those
employing secondary synchronous rectification. Otherwise, the
FREQUENCY pin should be connected to the SOURCE pin
for the default 400 kHz.
Figure 5. Typical Waveforms for (1) Power Up (2) Normal Operation (3) Auto-restart (4) Power Down.
PI-3867-050602
S1 S2 S6 S7 S1 S2 S6 S7S0 S1 S7
S0 S0 5.8 V
4.8 V
S7
0 V
0 V
0 V
VLINE
VC
VDRAIN
VOUT
Note: S0 through S7 are the output states of the auto-restart counter
2
1234
0 V
~
~
~
~
~
~~
~
S6 S7
~
~
~
~
~
~
VUV
~
~
~
~
~
~
~
~
S2
~
~
DPA423-426
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6
Pulse Width Modulator and Maximum Duty Cycle
The pulse width modulator implements voltage mode control
by driving the output MOSFET with a duty cycle inversely
proportional to the current into the CONTROL pin that
is in excess of the internal supply current of the chip (see
Figure 4). The excess current is the feedback error signal that
appears across RE (see Figure 2). This signal is filtered by an RC
network with a typical corner frequency of 30 kHz to reduce the
effect of switching noise in the chip supply current generated by
the MOSFET gate driver. The filtered error signal is compared
with the internal oscillator sawtooth waveform to generate the
duty cycle waveform. As the control current increases, the duty
cycle decreases. A clock signal from the oscillator sets a latch
that turns on the output MOSFET. The pulse width modulator
resets the latch, turning off the output MOSFET. Note that a
minimum current must be driven into the CONTROL pin before
the duty cycle begins to change.
The maximum duty cycle, DCMAX is set at a default maximum
value of 75% (typical). However, by connecting the
LINE-SENSE to the DC input bus through a resistor
with appropriate value, the maximum duty cycle can be
made to decrease from 75% to 33% (typical) as shown in
Figure 7 when input line voltage increases (see Line Feed-
Forward with DCMAX Reduction).
Minimum Duty Cycle and Cycle Skipping
To maintain power supply output regulation, the pulse width
modulator reduces duty cycle as the load at the power supply
output decreases. This reduction in duty cycle is proportional to
the current flowing into the CONTROL pin. As the CONTROL
pin current increases, the duty cycle reduces linearly towards a
minimum value specified as minimum duty cycle, DCMIN. After
reaching DCMIN, if CONTROL pin current is increased further
by approximately 2 mA, the pulse width modulator will force
the duty cycle from DCMIN to zero in a discrete step (refer to
Figure 4). This feature allows a power supply to operate in a
cycle skipping mode when the load consumes less power than
the DPA-Switch delivers at minimum duty cycle, DCMIN. No
additional control is needed for the transition between normal
operation and cycle skipping. As the load increases or decreases,
the power supply automatically switches between normal and
cycle skipping mode as necessary.
Cycle skipping may be avoided, if so desired, by connecting
a minimum load at the power supply output such that the duty
cycle remains at a level higher than DCMIN at all times.
Error Amplifier
The shunt regulator can also perform the function of an error
amplifier in primary side feedback applications. The shunt
regulator voltage is accurately derived from a temperature-
compensated bandgap reference. The gain of the error amplifier is
set by the CONTROL pin dynamic impedance. The CONTROL
pin clamps external circuit signals to the VC voltage level.
The CONTROL pin current in excess of the supply current
is separated by the shunt regulator and flows through RE as a
voltage error signal.
On-chip Current Limit with External Programmability
The cycle-by-cycle peak drain current limit circuit uses the
output MOSFET ON-resistance as a sense resistor. A current
limit comparator compares the output MOSFET on-state
drain to source voltage, VDS(ON) with a threshold voltage. At
the current limit, VDS(ON) exceeds the threshold voltage and the
MOSFET is turned off until the start of the next clock cycle.
The current limit comparator threshold voltage is temperature
compensated to minimize the variation of the current limit due
to temperature related changes in RDS(ON) of the output MOSFET.
The default current limit of DPA-Switch is preset internally.
However, with a resistor connected between EXTERNAL
CURRENT LIMIT pin and SOURCE pin, the current limit can
be programmed externally to a lower level between 25% and
100% of the default current limit. Please refer to the graphs
in the Typical Performance Characteristics section for the
selection of the resistor value. By setting current limit low,
a larger DPA-Switch than necessary for the power required
can be used to take advantage of the lower RDS(ON) for higher
efficiency/smaller heat sinking requirements. With a second
resistor connected between the EXTERNAL CURRENT
LIMIT pin and the DC input bus, the current limit is reduced
with increasing line voltage, allowing a true power limiting
operation against line variation to be implemented in a flyback
configuration.
The leading edge blanking circuit inhibits the current limit
comparator for a short time after the output MOSFET is turned
on. The leading edge blanking time has been set so that, if a
power supply is designed properly, current spikes caused by
primary-side capacitance and secondary-side rectifier reverse
recovery time should not cause premature termination of the
switching pulse.
The current limit after the leading edge blanking time is as shown
in Figure 31. To avoid triggering the current limit in normal
operation, the drain current waveform should stay within the
envelope shown.
Line Under-Voltage Detection (UV)
At power up, UV keeps DPA-Switch off until the input line
voltage reaches the under voltage upper threshold. At power
down, UV holds DPA-Switch on until the input voltage falls
below the under voltage lower threshold. A single resistor
connected from the LINE-SENSE pin to the DC input bus sets
UV upper and lower thresholds. To avoid false triggering by
noise, a hysteresis is implemented which sets the UV lower
threshold typically at 94% of the UV upper threshold. If the UV
lower threshold is reached during operation without the power
supply losing regulation and the condition stays longer than
10 µs (typical), the device will turn off and stay off until the
DPA423-426
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Figure 6. Synchronization Timing Diagram.
PI-2762-070501
Oscillator
(SAW)
DMAX
2 V
0 V
V
L
tON
fSYNC ≥ 128 kHz; tOFF ≤ 7.7 µs; 120 ns ≤ tON ≤ 2250 ns for fOSC = 400 kHz
≤ 3080 ns for fOSC = 300 kHz
tOFF
ON OFF SYNC
Time
UV upper threshold has been reached again. Then, a soft-start
will be initiated the next time CONTROL pin voltage reaches
5.8.V. If the power supply loses regulation before reaching the
UV lower threshold, the device will enter auto-restart. At the end
of each auto-restart cycle (S7), the UV comparator is enabled. If
the UV upper threshold is not exceeded, the MOSFET will be
disabled during the next cycle (see Figure 5). The UV feature
can be disabled independent of OV feature.
Line Overvoltage Shutdown (OV)
The same resistor used for UV also sets an overvoltage
threshold which, once exceeded, will force the DPA-Switch
output into the off-state within one switching cycle. The ratio
of OV and UV thresholds is preset at 2.7 as can be seen in
Figure 7. When the MOSFET is off, the input voltage surge
capability is increased to the voltage rating of the MOSFET
(220 V), due to the absence of the reflected voltage and leakage
spikes on the drain. A small amount of hysteresis is provided on
the OV threshold to prevent noise triggering. The OV feature
can be disabled independent of the UV feature as shown in
Figure 13.
Line Feed-Forward with DCMAX Reduction
The same resistor used for UV and OV also implements line
voltage feed-forward that minimizes output line ripple and
reduces power supply output sensitivity to line transients.
This feed-forward operation is illustrated in Figure 4 by the
different values of IL. Note that for the same CONTROL pin
current, higher line voltage results in smaller operating duty
cycle. As an added feature, the maximum duty cycle DCMAX
is also reduced from 75% (typical) at a voltage slightly higher
than the UV threshold to 33% (typical) at the OV threshold
(see Figures 4, 7). Limiting DCMAX at higher line voltages
helps prevent transformer saturation due to large load transients
in forward converter applications. DCMAX of 33% at the OV
threshold was chosen to ensure that the power capability of
the DPA-Switch is not restricted by this feature under normal
operation.
Remote ON/OFF
Remote ON/OFF control describes operation where the IC is
turned on or off for long periods as opposed to the cycle-by-
cycle on/off control, which is described in the Synchronization
section below.
DPA-Switch can be turned on or off by controlling the current into
the LINE-SENSE pin or out from the EXTERNAL CURRENT
LIMIT pin (see Figure 7). This allows easy implementation of
remote ON/OFF control of DPA-Switch in several different
ways. A transistor or an optocoupler output connected between
the EXTERNAL CURRENT LIMIT pin and the SOURCE pin
implements this function with “active-on” (Figures 17, 19 and 21)
while a transistor or an optocoupler output connected between
the LINE-SENSE pin and the CONTROL pin implements the
function with “active-off” (Figures 18, 20 and 22).
When a signal is received at the LINE-SENSE pin or the
EXTERNAL CURRENT LIMIT pin to disable the output
through any of the pin functions such as OV, UV and remote
ON/OFF, DPA-Switch always completes its current switching
cycle before the output is forced off. The internal oscillator is
stopped at the end of the current cycle and stays there as long
as the disable signal exists. When the signal at the above pins
changes state from disable to enable, the internal oscillator
starts the next switching cycle.
The remote ON/OFF feature can be used as a standby or power
switch to turn off the DPA-Switch and keep it in a very low
power consumption state for indefinitely long periods. If the
DPA-Switch is held in remote-off state for longer than 10 µs
DPA423-426
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8
(typical), the CONTROL pin goes into the hysteretic mode
of operation. In this mode, the CONTROL pin goes through
alternate charge and discharge cycles between 4.8 V and
5.8 V (see CONTROL Pin Operation section above) and the IC
runs entirely off the high voltage DC input, but with very low
power consumption (30 mW typical at 48 V with LINE-SENSE
and EXTERNAL CURRENT LIMIT pins open). When the
DPA-Switch is remotely turned on after entering this mode,
it will initiate a normal start-up sequence with soft-start the
next time the CONTROL pin reaches 5.8 V. In the worst case,
the delay from remote on to start-up can be equal to the full
discharge/charge cycle time of the CONTROL pin, which is
approximately 36 ms for a 22 µF CONTROL pin capacitor. This
reduced-consumption remote-off mode can eliminate expensive
and unreliable in-line mechanical switches. It also allows for
microprocessor-controlled turn-on and turn-off sequences that
may be required in certain applications.
Synchronization
In addition to sensing incoming current for OV, UV and
remote ON/OFF, the LINE-SENSE pin also monitors its pin
voltage through a 1 V threshold comparator. A pin voltage
below 1 V turns on DPA-Switch. When the voltage at LINE-
SENSE pin rises beyond 1 V to disable the output, DPA-Switch
completes its current switching cycle before the output is
forced off (similar to remote ON/OFF operation). The internal
oscillator is stopped at the end of the current cycle awaiting the
LINE-SENSE pin voltage to go low to start the next cycle. This
allows the use of the 1 V threshold to synchronize DPA-Switch
to an external signal with a frequency lower than its internal
switching frequency. A transistor or an optocoupler output
connected between the LINE-SENSE pin and the SOURCE
pin implements this function (see Figure 24). Please refer to
Figure 6 for the timing waveforms of synchronization
operation.
In order to be recognized as a synchronization pulse, the
LINE-SENSE pin needs to stay low (on-time) for at least
120 ns but no more than 2250 ns for 400 kHz (or 3080 ns for
300 kHz) internal switching frequency. In addition, the off-
time must be kept below 7.7 µs, which is a limitation set by
the lowest synchronization frequency of 128 kHz allowed by
the chip. The effective DCMAX for synchronization operation
can be calculated as 0.75 • fSYNC/fOSC. An off-time longer than
7.7 µs may force the CONTROL pin to go into the hysteretic
mode and initiate a soft-start cycle at next turn-on.
Soft-Start
Two on-chip soft-start functions are activated at start-up with
a duration of 5 ms (typical). Maximum duty cycle starts from
0% and linearly increases to the default maximum of 75% at
the end of the 5 ms duration and the current limit starts from
about 85% and linearly increases to 100% at the end of the
5 ms duration. In addition to start-up, soft-start is also activated
at each restart attempt during auto-restart and when restarting
after being in hysteretic regulation of CONTROL pin voltage
(VC), due to remote off or thermal shutdown conditions. This
effectively minimizes current and voltage stresses on the output
MOSFET, the clamp circuit and the output rectifier during
start-up. This feature also helps minimize output overshoot and
prevents saturation of the transformer during start-up.
Shutdown/Auto-Restart
To minimize DPA-Switch power dissipation under fault
conditions, the shutdown/auto-restart circuit turns the power
supply on and off at an auto-restart duty cycle of typically 4%
if an out of regulation condition persists. Loss of regulation
interrupts the external current into the CONTROL pin. VC
regulation changes from shunt mode to the hysteretic auto-
restart mode as described in CONTROL pin operation section.
When the fault condition is removed, the power supply output
becomes regulated, VC regulation returns to shunt mode, and
normal operation of the power supply resumes.
Hysteretic Over-Temperature Protection
Over temperature protection is provided by a precision analog
circuit that turns the output MOSFET off when the junction
temperature exceeds the thermal shutdown temperature
(137 °C typical). When the junction temperature cools to below
the hysteretic temperature (110 °C typical), normal operation
resumes providing automatic recovery. VC is regulated in
hysteretic mode and a 4.8 V to 5.8 V (typical) sawtooth waveform
is present on the CONTROL pin while in thermal shutdown.
Bandgap Reference
All critical DPA-Switch internal voltages are derived from a
temperature-compensated bandgap reference. This reference
is also used to generate a temperature-compensated current
reference that is trimmed to accurately set the switching
frequency, current limit, and the line OV/UV thresholds.
DPA-Switch has improved circuitry to maintain all of the above
critical parameters within very tight absolute and temperature
tolerances.
High-Voltage Bias Current Source
This current source biases DPA-Switch from the DRAIN pin
and charges the CONTROL pin external capacitance during
start-up or hysteretic operation. Hysteretic operation occurs
during auto-restart, remote off and over-temperature shutdown.
In this mode of operation, the current source is switched on and
off with an effective duty cycle of approximately 20%. This
duty cycle is determined by the ratio of CONTROL pin charge
(IC(CH)) and discharge currents (ICD1 and ICD2). This current source
is turned off during normal operation when the output MOSFET
is switching. The effect of the current source switching may be
seen on the DRAIN voltage waveform as small disturbances,
which is normal.
DPA423-426
Q
5/06 9
Using Feature Pins
FREQUENCY (F) Pin Operation
The FREQUENCY pin is a digital input pin. Shorting the
FREQUENCY pin to SOURCE pin selects the nominal
switching frequency of 400 kHz (Figure 9) which is suited for
most applications. For other applications that may benefit from
lower switching frequency, a 300 kHz switching frequency can
be selected by shorting the FREQUENCY pin to the CONTROL
pin (Figure 10). This pin should not be left open.
LINE-SENSE (L) Pin Operation
When current is fed into the LINE-SENSE pin, it works as a
voltage source of approximately 2.6 V up to a maximum current
of +240 µA (typical). At +240 µA, this pin turns into a constant
current sink. Refer to Figure 8. In addition, a comparator with
a threshold of 1 V is connected at the pin and is used to detect
when the pin is shorted to the SOURCE pin.
There are a total of five functions available through the use of
the LINE-SENSE pin: OV, UV, line feed-forward with DCMAX
reduction, remote ON/OFF and synchronization. Shorting the
LINE-SENSE pin to the SOURCE pin disables all five functions.
The LINE-SENSE pin is typically used for line sensing by
connecting a resistor from this pin to the positive input DC
voltage bus to implement OV, UV and line feed-forward with
DCMAX reduction over line voltage. In this mode, the value of
the resistor determines the line OV/UV thresholds, and the
DCMAX is reduced linearly with input DC high voltage starting
from just above the UV threshold. This pin can also be used
as the input pin for remote ON/OFF and synchronization.
An external transistor placed between the LINE-SENSE pin
and the CONTROL pin realizes remote ON/OFF via UV or
OV threshold. Synchronization is available by connecting
an open drain external MOSFET between the LINE-SENSE
pin and the SOURCE pin to generate synchronization pulse.
Each time the MOSFET turns on, the falling edge of the
LINE-SENSE pin voltage initiates a new switching cycle. The
lowest synchronization frequency guaranteed by DPA-Switch is
128 kHz. Refer to Table 2 for possible combinations of the
functions with example circuits shown in Figure 11 through
Figure 24. A description of specific functions in terms of the
LINE-SENSE pin I/V characteristic is shown in Figure 7 (right
hand side). The horizontal axis represents LINE-SENSE pin
current with positive polarity indicating currents flowing into
the pin. The meaning of the vertical axes varies with functions.
For those that control the on/off states of the output such as
UV, OV and remote ON/OFF, the vertical axis represents
the enable/disable states of the output. UV triggers at IUV
(+50 µA typical with 4 µA hysteresis) and OV triggers at IOV
(+135 µA typical with 4 µA hysteresis). Between the UV and
OV thresholds, the output is enabled. For line feed-forward with
DCMAX reduction, the vertical axis represents the magnitude
of the DCMAX Line feed-forward with DCMAX reduction lowers
maximum duty cycle from 75% at IL(DC) (+55 µA typical) to
33% at IOV (+135 µA).
EXTERNAL CURRENT LIMIT (X) Pin Operation
When current is drawn out of the EXTERNAL CURRENT
LIMIT pin, it works as a voltage source of approximately
1.3 V up to a maximum current of -230 µA (typical). At
-230 µA, it turns into a constant current source (refer to Figure 8).
There are two functions available through the use of the
EXTERNAL CURRENT LIMIT pin: external current limit
and remote ON/OFF. Shorting the EXTERNAL CURRENT
LIMIT pin and SOURCE pin disables both functions. In high
efficiency applications, this pin can be used to reduce the current
limit externally to a value close to the operating peak current,
by connecting the pin to the SOURCE pin through a resistor.
LINE-SENSE AND EXTERNAL CURRENT LIMIT PIN TABLE*
Figure Number 11 12 13 14 15 16 17 18 19 20 21 22 23 24
Three Terminal Operation ✓
Under-Voltage ✓ ✓ ✓ ✓ ✓
Overvoltage ✓ ✓ ✓ ✓ ✓
Line Feed-Forward (DCMAX)✓ ✓ ✓ ✓
Overload Power Limiting ✓
External Current Limit ✓ ✓ ✓ ✓ ✓ ✓
Remote ON/OFF ✓ ✓ ✓ ✓ ✓ ✓
Synchronization ✓
*This table is only a partial list of many LINE-SENSE and EXTERNAL CURRENT LIMIT pin configurations that are possible.
Table 2. Typical LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Configurations.
▼
DPA423-426
Q
5/06
10
Figure 7. LINE-SENSE and EXTERNAL CURRENT LIMIT Pin Characteristics.
-250 -200 -150 -100 -50 50 100 150 200 2500 0
PI-2778-080801
Output
MOSFET
Switching
(Enabled)
(Disabled)
(Enabled)
(Disabled)
ILIMIT (Default)
Current
Limit
VBG
-21.5 µA
-25.5 µA
VBG + VTP
IUV(U)
IREM(U) IOV(U)
X Pin Voltage
Output
MOSFET
Switching
Maximum
Duty Cycle
L Pin Voltage
DCMAX (75%)
Note: These figures provide idealized functional characteristics with typical performance values. Please refer to the
parametric table and typical performance characteristics sections of the data sheet for measured data.
X Pin Current (µA) L Pin Current (µA)
131 µA
135 µA
47 µA
The pin can also be used as a remote ON/OFF control input.
Table 2 shows several different ways of using this pin. See
Figure 7 for a description of the functions where the horizontal
axis (left hand side) represents the EXTERNAL CURRENT
LIMIT pin current. The meaning of the vertical axes varies
with function. For those that control the on/off states of the
output such as remote ON/OFF, the vertical axis represents the
enable/disable states of the output. For external current limit,
the vertical axis represents the magnitude of the ILIMIT
. Please
see graphs in the Typical Performance Characteristics section
for the current limit programming range and the selection of
the appropriate resistor value.
DPA423-426
Q
5/06 11
Figure 8. LINE-SENSE (L), and EXTERNAL CURRENT LIMIT (X) Pin Input Simplified Schematic.
VBG + VT
1 V
VBG
230 µA
240 µA
CONTROL (C)
(Voltage Sense)
(Positive Current Sense - Under-Voltage,
Overvoltage, ON/OFF Maximum Duty
Cycle Reduction)
(Negative Current Sense - ON/OFF,
Current Limit Adjustment)
PI-2765-061704
DPA-Switch
LINE-SENSE (L)
EXTERNAL CURRENT LIMIT (X)
DPA423-426
Q
5/06
12
Figure 9. 400 kHz Frequency Operation. Figure 10. 300 kHz Frequency Operation.
Typical Uses of FREQUENCY (F) Pin
PI-2654-071700
DC
Input
Voltage
+
-
D
S
C
CONTROL
F
PI-2655-071700
DC
Input
Voltage
+
-
D
S
C
CONTROL
F
Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins
Figure 11. Three Terminal Operation (LINE-SENSE and
EXTERNAL CURRENT LIMIT Features Disabled.
FREQUENCY Pin can be tied to SOURCE or
CONTROL Pin).
Figure 12. Line-Sensing for Under-Voltage, Overvoltage and
Line Feed-forward.
Figure 13. Line-Sensing for Under-Voltage Only (Overvoltage
Disabled).
Figure 14. Line-Sensing for Overvoltage Only (Under-Voltage
Disabled). Maximum Duty Cycle will be reduced at
Low Line.
PI-2767-091302
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
RLS
619 kΩ
1%
VUV = IUV x RLS + VL (IL = IUV)
VOV = IOV x RLS + VL (IL = IOV)
For RLS = 619 kΩ
VUV = 33.3 V
VOV = 86.0 V
PI-2852-121504
DC
Input
Voltage
+
-
D L
S
C
VUV = RLS x IUV +
VL (IL = IUV)
For Values Shown
VUV = 33.1 V
RLS
15 V
464 kΩ
1%
150 kΩ
1%
CONTROL
PI-2853-091302
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
RLS
1N4148
VOV = IOV x RLS +
VL (IL = IOV)
For Values Shown
VOV = 86.2 V
590 kΩ
1%
30 kΩ
1%
DPA423-426
Q
5/06 13
Figure 15. Externally Set Current Limit. Figure 16. Current Limit Reduction with Line Voltage.
X
PI-2836-011904
DC
Input
Voltage
+
-
D
S
C
RIL
For RIL = 12 kΩ
ILIMIT = 64%
See Figure 34 for other
resistor values (RIL)
For RIL = 25 kΩ
ILIMIT = 34%
CONTROL
X
PI-2854-050602
DC
Input
Voltage
+
-
D
S
C
363 kΩ
RLS
4.2 kΩ
RIL
100% @ 36 VDC
64% @ 72 VDC
ILIMIT =
ILIMIT =
CONTROL
Figure 17. Active-on (Fail Safe) Remote ON/OFF. Figure 18. Active-off Remote ON/OFF. Maximum Duty Cycle will
be reduced.
Figure 19. Active-on Remote ON/OFF with Externally Set Current
Limit.
Figure 20. Active-off Remote ON/OFF with Externally Set Current
Limit.
X
ON/OFF
47 kΩ
PI-2856-072602
DC
Input
Voltage
+
-
D
S
C
RIL QR
For RIL = 12 kΩ
ILIMIT = 64%
For RIL = 25 kΩ
ILIMIT = 34%
QR can be an optocoupler
output or can be replaced
by a MOSFET or manual
switch.
CONTROL
PI-2855-050602
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
47 kΩ
QR
RMC
37.4 kΩ
QR can be an
optocoupler output or
can be replaced
by a manual switch.
ON/OFF
X
PI-2625-040501
DC
Input
Voltage
+
-
D
S
C
ON/OFF
47 KΩ
QR can be an optocoupler
output or can be replaced by
a manual switch.
QR
CONTROL
Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)
DPA423-426
Q
5/06
14
PI-2858-072602
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
47 kΩ
619 kΩ
1%
QR
RLS
ON/OFF
For RLS = 619 kΩ
VUV = 33.3 V
VOV = 86.0 V
QR can be an optocoupler
output or can be replaced
by a MOSFET or manual
switch.
X
ON/OFF
47 kΩ
PI-2859-050602
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
RIL
RLS
QR
619 kΩ
1%
DCMAX@36 V = 75%
DCMAX@72 V = 42%
For RIL = 12 kΩ
ILIMIT = 64%
QR can be an optocoupler
output or can be replaced
by a manual switch.
Figure 21. Active-on Remote ON/OFF with LINE-SENSE and
EXTERNAL CURRENT LIMIT.
Figure 22. Active-off Remote ON/OFF with LINE-SENSE.
Figure 23. Line-Sensing and Externally Set Current Limit.
PI-3868-050602
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
ON/OFF
47 kΩ
QR can be an optocoupler
output.
QR
For timing requirements,
see Figure 6.
Figure 24. Synchronization.
X
PI-2837-011904
DC
Input
Voltage
+
-
D
S
C
CONTROL
L
RIL
RLS
12 kΩ
619 KΩ
1%
For RLS = 619 kΩ
DCMAX@36 V = 75%
DCMAX@72 V = 42%
For RIL = 12 kΩ
ILIMIT = 64%
See Figure 34 for other
resistor values (RIL)
to select different ILIMIT
values
VUV = 33.3 V
VOV = 86.0 V
Typical Uses of LINE-SENSE (L) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)
DPA423-426
Q
5/06 15
Application Examples
Figure 25. A High Efficiency 30 W, 5 V, Telecom Input DC-DC Converter.
U1
DPA425R
D1
BAV
19WS
D2
C5
220 nF
VR1
SMBJ
150 C6
68 µF
10 V
C7
1 nF
1.5 kV R14
10 Ω
T1
R3
18.2 kΩ
1%
R1
619 kΩ
1%
R4
1.0 Ω
C10
100 µF
10 V
C11
100 µF
10 V
C12
1 µF
10 V
36-75 VDC
L1
1 µH
2.5 A L2
C1, C2 & C3
1 µF
100 V
D L
S X F
C
CONTROL
CONTROL
U2
PC357N1T
PI-3472-061704
DPA-Switch
5 V, 6 A
RTN
Q2
Si4888
DY
C9* R5*
*Optional components
R15
10 ΩR16
10 kΩ
D4
BAV19WS
R17
10 Ω
Q1
Si4888
DY
C4
4.7 µF
20 V
U2
D3
BAV19WS
U3
LM431AIM3
R9
220 Ω
R11
10.0 kΩ
1%
R10
10.0 kΩ
1%
R12
5.1 Ω
R7
10 kΩ
C14
1 µF
C13
10 µF
10 V
C16
100 nF
R6
150 Ω
VIN
+
VIN
C17
3300 pF
High Efficiency 30 W Forward Converter
The circuit shown in Figure 25 is a typical implementation of a
single output DC-DC converter using DPA-Switch in a forward
configuration with synchronous rectification. This design
delivers 30 W at 5 V, from a 36 VDC to 75 VDC input with a
nominal efficiency at 48 VDC of 90% using the DPA425R.
By taking advantage of many of the built-in features of the
DPA-Switch, the design is greatly simplified compared to a
discrete implementation. Resistor R1 programs the input under-
voltage and overvoltage thresholds to typically 33 V and 86 V
respectively. This resistor also linearly reduces the maximum
duty from the internal maximum of 75% at 36 V to 42% at
72 V to prevent core saturation during load transients at high input
voltages. The DPA-Switch internal thresholds are toleranced
and characterized so the designer can guarantee the converter
will begin operation at 36 V, necessary to meet ETSI standards,
without the cost of an external reference IC.
The current limit is externally set by resistor R3 to just above the
drain current level needed for maximum load regulation to limit
the maximum overload power of the converter. The externally
programmable current limit feature also allows a larger
DPA-Switch family member to be selected. Using the X pin, the
current limit can be adjusted to the same level. A large device
reduces conduction losses and improves efficiency without
requiring any other circuit changes. This has been used here
to replace the DPA424R with a DPA425R.
The selectable 300/400 kHz switching frequency is set to 300 kHz
by connecting the FREQUENCY (F) pin to CONTROL (C).
DRAIN voltage clamping is provided by VR1, which keeps
the peak DRAIN voltage within acceptable limits. Transformer
core reset is provided by the gate capacitance of Q1 with R17
in series. Optional reset capacitance C9 with R5 can be added
if necessary to supplement the gate capacitance of Q1.
The output of the transformer is rectified using MOSFETs
to provide synchronous rectification. The UV/OV function,
together with the turns ratio of the transformer, defines the
maximum MOSFET gate voltage, allowing the very simple
gate drive arrangement, without the need for drive windings
or a drive IC. During primary on-time, capacitor C17 couples
charge through resistor R15 to drive the gate of the forward
MOSFET, Q2. Capacitor C17 provides a DC isolated drive for
Q2, preventing gate overstress on Q1 during power down. The
time constant formed by R16 and C17 is selected to be much
longer than one switching cycle. Diode D4 resets the voltage
on capacitor C17 before the next switching cycle. During the
primary off-time, the diode D2 provides a conduction path for
the energy in inductor L2 while Q1 is still off. The transformer
DPA423-426
Q
5/06
16
reset voltage on the secondary winding directly drives a positive
voltage on the gate of catch MOSFET, Q1. MOSFET Q1
provides a low loss conduction path for a substantial portion
of the primary off-time. An isolated auxiliary winding on L2,
rectified and filtered by D1 and C4, provides the bias supply
for the optocoupler transistor. Output regulation is achieved
by using secondary side voltage reference, U3. The resistor
divider formed by R10 and R11, together with the reference
voltage, determines the output voltage. Diode D3 and C13 form
a soft-finish network that, together with the internal duty cycle
and current limit soft-start of the DPA-Switch, prevent output
overshoot at start-up. Resistor R7 ensures that the soft-finish
capacitor is discharged quickly when the output falls out of
regulation. Control loop response is shaped by R6, C16, R12,
C14, R9, R4 and C5, providing a wide bandwidth and good
phase margin at gain crossover. Since the PWM control in
DPA-Switch is voltage mode, no slope compensation is required
for duty cycles above 50%.
Cost Effective 6.6 W Flyback Converter
The DPA-Switch flyback power supply provides a cost effective
solution for high density PoE and VoIP DC-DC applications.
Figure 26 shows a typical implementation of a single output
flyback converter using the DPA423G. For applications that
require input to output isolation, this simple, low component
count design delivers 6.6 W at 3.3 V from a 36 VDC to 57 VDC
input with a nominal efficiency at 48 VDC of 80%.
Resistor R2 programs the input under-voltage and overvoltage
thresholds to 33 V and 86 V respectively. Resistors R1 and R3
program the internal device current limit. The addition of line
sense resistor R1 reduces the current limit with increasing input
voltage, preventing excessive overload output current. In this
design the overload output current varies less than ±2.5% across
the entire input voltage range. Controlling the current limit also
reduces secondary component stress and leakage inductance
spikes, allowing the use of a lower VRRM (30 V rather than
40 V) Schottky output diode, D2.
The primary side Zener clamp VR1 ensures the peak drain
voltage is kept below the 220 V BVDSS rating of U1 under
input surge and overvoltage events. During normal operation,
VR1 does not conduct and C2 is sufficient to limit the peak
drain voltage.
The primary bias winding provides CONTROL pin current after
start-up. Diode D3 rectifies the bias winding, while components
R5 and C8 reduce high frequency switching noise and prevent
peak charging of the bias voltage. Capacitor C3 provides
local decoupling of U1 and should be physically close to the
CONTROL and SOURCE pins. Energy storage for start-up
and auto-restart timing is provided by C4.
The secondary is rectified by D2 and the Low ESR tantalum
output capacitors, C5-C7, minimizing switching ripple and
maximizing efficiency. A small footprint secondary output choke
L1 and ceramic output capacitor C9 are adequate to reduce high
frequency noise and ripple to below 35 mV peak-peak under
full load conditions.
The output voltage is sensed by the voltage divider formed
by resistors R8 and R9 and is fed to the low voltage 1.24 V
reference U3. Feedback compensation is provided by R6, R7
D
S
C
L
FX
CONTROL
DPA-Switch
C4
22 µF
10 V
C3
0.1 µF
50 V C10
0.33 µF
C11
0.1 µF
C8
1 µF
50 V
C9
1 µF
10 V
C7
330 µF
6 V
L1
1 µH, 2A
C6
330 µF
6 V
C5
330 µF
6 V
C1
1 µF
100 V
C2
47 pF
200 V
VR1
SMAJ
150A
R1
1 MΩ
1%
R3
8.66 kΩ
1%
R4
5.1 Ω
R9
20 kΩ
1%
R8
34 kΩ
1%
R7
1 kΩ
R5
100 Ω
R6
51 Ω
R2
619 kΩ
1%
9, 101
2
3
6, 7
4
T1
5
U1
DPA423G U2
PC357
U3
CAT431L,
SOT23
D3
BAV19, SOD323
D2
SL43
+VIN
36 - 57 VDC
J1-1
3.3 V, 2 A
J2-1
-VIN
J1-2
RTN
J2-2
PI-3806-061704
Figure 26. A Cost Effective 6.6 W, 3.3 V Flyback DC-DC Converter.
DPA423-426
Q
5/06 17
and C11 together with C4 and R4. Capacitor C10 provides a
soft-finish characteristic, preventing output overshoot during
start-up.
Low Cost PoE VoIP Phone Converter
The basic circuitry to support IEEE standard 802.3af Power
over Ethernet (PoE) is straightforward. Class 0 signature and
classification circuits can be implemented with a single resistor
and the required under-voltage lockout function is a voltage
controlled pass-switch. By adding this circuitry to the front
end of a DPA converter, a low cost and low component count
PoE powered device (PD) power supply can be implemented.
Figure 27 shows a typical PD solution.
The PoE specification requires the PD to provide three
fundamental functions: discovery, classification, and pass-
switch connection.
When input voltage is applied to the PD, it must present the
correct discovery signature impedance in the voltage range of
2.5 VDC to 10 VDC. This impedance is provided by R51 in
Figure 27.
The second “classification” phase occurs at input voltages
15 VDC to 20 VDC. The PD must draw a specified current to
identify the device class (Class 0 specifies 0.5 mA to 4 mA).
This is again accomplished by resistor R51.
In the third phase, the bipolar pass-switch (Q51 in Figure 27)
connects the input voltage to the power supply at voltages
above approximately 30 VDC (28 V+VR52). Zener diode VR51
conducts, driving the current through resistor R52 to the base
of Q51. Resistor R53 prevents turn-on under other conditions.
Once the Power supply has started, components D51, D52, C51
and R54 enhance the base-current drive by coupling power from
the power supply bias winding.
Once the three start up phases have been successfully completed,
the DPA-Switch is allowed to function as a forward converter
(described in Figure 25 and accompanying text).
Figure 27. PoE Interface Circuit Using a Bipolar Transistor Pass-Switch and DPA424P.
U1
DPA424P
D6
BAV
19WS
Q22
Si4804
D21
SL13
15 V
Q21
Si4804
C2
1 µF
100 V
VR1
SMAJ
150 C5
47 µF
10 V
D41
BAV19WS
D31
20CJQ060
R21
10 Ω
R22
10 Ω
R23
10 kΩ
VR21
C21
2.2 nF
U2
T1
R1
649 kΩ
1%
R3
1.0 Ω
C22-C24
100 µF 5 V
VR41
6.8 V
D42
IN4148
C31
100 µF
10 V
C41
4.7 µF, 35 V
C25
1 µF
10 V
L1
1 µH 2.5 A
PoE Interface
L2
16 µH 4 A
C6
4.7 µF
20 V
D11
BAV19WS
U3
LM431AIM3
C11
2.2 µF
10 V
R14
1 kΩ
R15
10 kΩ
1%
R16
10 kΩ
1%
R4
160 Ω
R12
150 Ω
R13
11 Ω
C1
1 µF
100 V
U2
PC357
N1T R11
10 kΩ
PI-3824-040706
DPA-Switch
5 V, 2.4 A
RTN
7.5 V, 0.4 A
20 V, 10 mA
C13
68 nF
4
5
3
6
7
1 8
7 2
7
8
6
5
4 3
Ethernet
(RJ-45)
Connector
R2
13.3 kΩ
1%
R52
20 kΩ
D101
DL4002
DL4002
D102
D103
DL4002
DL4002
D104
D105
DL4002
DL4002
D106
D107
DL4002
DL4002
D108
R22
10 kΩ
R23
174 k
1%
R21
10 k
R51
24.9 kΩ
1% 1/4 W
D51
BAV19
D52
BAV19
C51
1 nF
50 V
R54
20 Ω
VR51
28 V
(1,2)
(4,5)
(3,6)
(7,8)
D
S
C
L
FX
CONTROL
C12
100 nF
R53
20 kΩ
Q51
TIP29C (100 V/1 A)
or MMBTA06
C4
220 nF
Q20
MMBTS3906
DPA423-426
Q
5/06
18
Key Application Considerations
DPA-Switch Design Considerations
Power Table
This section provides a description of the assumptions used
to generate the power tables (Tables 1 and 3 through 6) and
explains how to use the information provided by them.
All Power tables: Tables 1 and 3 through 6
• Maximum output power is limited by the device internal
current limit. This is the peak output power which could
become the continuous output power, provided adequate
heat sinking is used.
• Data assumes adequate heat sinking to keep the junction
temperature at or below 100 °C and worst case RDS(ON) at
TJ = 100 °C.
• The use of P and G packages are recommended for device
dissipation equal to or less than 1.5 W only due to package
thermal limitation. For device dissipation above 1.5 W, use
S and R packages.
Forward power tables: Tables 1 (upper half), 3 and 4
• Output power figures are based on forward topology using
Schottky diode rectification. Up to 5% higher output power
is possible using synchronous rectification.
• Dissipation data assumes a diode loss representing 6%
of the total output power and combined loss in magnetic
components representing 6% of the total output power.
DPA-Switch losses are based on a ratio between conduction
and switching losses of approximately 3:1. These
assumptions are typical for a single 5 V output forward
converter design using Schottky rectification and adequately
designed magnetic components.
Flyback power tables: Tables 1 (lower half), 5 and 6
• Output power and dissipation figures are based on a 5 V
output using Schottky diode rectification with an efficiency
of 85%. Values are generated by calculation based on
I2 • RDS(ON) losses and characterization of switching losses,
correlated to bench measurement of each DPA-Switch
device.
• Device dissipations above 1.5 W are possible using the
S and R packages. However the forward converter topology
is recommended for such higher power designs.
The power tables provide two types of information. The first is
the expected device dissipation for a given output power. The
second is the maximum power output. Each table specifies
the input voltage range and assumes a single 5 V output using
Schottky diode rectification.
For example, referring to Table 1, for 36 VDC to 75 VDC input
range, DPA424 would typically dissipate 1 W in a 23 W forward
converter and has a maximum power capacity of 35 W. In the
Table 4. Output Power Table for 24-48 VDC Input Voltage
(See Table 3 for Notes).
Table 3. Output Power Table for 16-32 VDC Input Voltage.
Notes: 1. Limited by device internal current limit. 2. See text in this
section for a complete description of assumptions. 3. See Part Ordering
Information.
Table 5. Flyback Output Power Table for 16-32 VDC Input
Voltage (See Table 3 for Notes).
OUTPUT POWER TABLE
16-32 VDC RANGE (FORWARD)2
Total Device
Dissipation
PRODUCT3
0.5 W 1 W 2.5 W 4 W 6 W
Max
Power
Output1
DPA423 5 W 7 W - - - 7.5 W
DPA424 7 W 10 W 15 W - - 15.5 W
DPA425 10 W 14 W 22 W 27 W 31 W
DPA426 12 W 16.5 W 25 W 31 W 37 W 43 W
OUTPUT POWER TABLE
24-48 VDC RANGE (FORWARD)2
Total Device
Dissipation
PRODUCT3
0.5 W 1 W 2.5 W 4 W 6 W
Max
Power
Output1
DPA423 8 W 11 W - - - 11.5 W
DPA424 11 W 16 W 23.5 W - - 25 W
DPA425 16 W 22 W 35 W 43 W 47 W
DPA426 18 W 25 W 39 W 48 W 58 W 65 W
OUTPUT POWER TABLE
16-32 VDC RANGE (FLYBACK)2
Total Device
Dissipation
PRODUCT3
0.5 W 0.75 W 1 W 1.5 W
Max
Power
Output1
DPA423 5 W - - - 6 W
DPA424 6.5 W 8.5 W 10 W - 11 W
DPA425 7 W 10 W 12 W 15 W 22 W
OUTPUT POWER TABLE
24-48 VDC RANGE (FLYBACK)2
Total Device
Dissipation
PRODUCT3,4
0.5 W 0.75 W 1 W 1.5 W
Max
Power
Output1
DPA423 7 W - - - 8.5 W
DPA424 8.5 W 11.5 W 14 W - 17 W
Table 6. Flyback Output Power Table for 24-48 VDC Input Voltage.
Notes: 1. Maximum output power is limited by device internal
current limit. 2. See text in this section for a complete description of
assumptions. 3. See Part Ordering Information. 4. Higher switching losses
may prevent DPA425 from delivering more power than a smaller device.
DPA423-426
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Soft-Start
Generally a power supply experiences maximum stress at
start-up before the feedback loop achieves regulation. For a
period of 5 ms the on-chip soft-start linearly increases the duty
cycle from zero to the default DCMAX at turn-on. In addition,
the primary current limit increases from 85% to 100% over the
same period. This causes the output voltage to rise in an orderly
manner allowing time for the feedback loop to take control of
the duty cycle. This integrated soft-start reduces the stress on
the DPA-Switch MOSFET, clamp circuit and output diode(s),
and helps prevent transformer saturation during start-up. Also,
soft-start limits the amount of output voltage overshoot, and in
many applications eliminates the need for a soft-finish capacitor.
If necessary, to remove output overshoot, a soft-finish capacitor
may be added to the secondary reference.
Switching Frequency
The FREQUENCY pin of DPA-Switch offers a switching
frequency option of 400 kHz or 300 kHz. Operating at 300 kHz
will increase the amount of magnetization energy stored in the
transformer. This is ideal for applications using synchronous
rectification driven directly from the transformer secondary
where this energy can be used to drive the catch MOSFET
gate.
Transformer Design
It is recommended that the forward converter transformer be
designed for maximum operating flux swing of 1500 Gauss
and a peak flux density of 3500 Gauss. When operating at the
maximum current limit of the selected DPA-Switch (during
overload conditions), neither magnetic component (transformer
and output inductor) should be allowed to saturate. When a larger
device than necessary has been selected, reducing the internal
current limit close to the operating peak current limits overload
power and minimizes the size of the secondary components.
No-load and Standby Consumption
Cycle skipping operation at light or no load can significantly
reduce power loss. In addition this operating mode ensures
that the output maintains regulation even without an external
minimum load. However, if cycle skipping is undesirable in
a particular application, it can be avoided by adding sufficient
pre-load.
DPA-Switch Layout Considerations
The DPA-Switch can operate with large DRAIN current, the
following guidelines should be carefully followed.
Primary Side Connections
The tab of DPA-Switch R package and S-PAK is the intended
return path for the high switching currents. Therefore, the tab
should be connected by wide, low impedance traces back to
same converter, DPA425 would dissipate 0.5 W. Selecting
DPA425 with associated reduced dissipation would increase
overall converter efficiency by approximately 2%.
Issues Affecting Dissipation:
1) Using synchronous rectification will tend to reduce device
dissipation.
2) Designs with lower output voltages and higher currents
will tend to increase the device dissipation listed in the
power table.
3) Reduced input voltage decreases the available output
power for the same device dissipation. Tables 3 to 6 are
the power tables for 16 VDC and 24 VDC input voltages.
Input voltages below 16 V are possible, but since the internal
start-up current source is not specified at voltages below
16 V, an external chip supply current should be fed into the
CONTROL pin approximately equal to but less than ICD1.
DPA-Switch Selection
Use Tables 1 and 3 through 6 to select the DPA-Switch based
on device dissipation. Selecting the optimum DPA-Switch
depends upon required maximum output power, efficiency, heat
sinking constraints and cost goals. With the option to externally
reduce current limit, a larger DPA-Switch may be used for
lower power applications where higher efficiency is needed
or minimal heat sinking is available. Generally, selecting the
next larger device, than is required for power delivery will give
the highest efficiency. Selecting even larger devices may give
little or no improvement in efficiency due to the improvement
in conduction losses being negated by larger device switching
losses. Figure 50 provides information on switching losses.
This together with conduction loss calculations give an estimate
of device dissipation.
Primary Clamp
A primary clamp network is recommended to keep the peak
DRAIN voltage due to primary leakage inductance to below
the BVDSS specification. A Zener diode combined with a small
value capacitor connected across the primary winding is a low
cost and low part count implementation.
Output Rectification
Rectification of the secondary is typically performed using
Schottky diodes or synchronous rectification. Schottky diodes
are selected for peak inverse voltage, output current, forward drop
and thermal conditions. Synchronous rectification requires the
additional complication of providing gate drive. The specified
line under-voltage and line overvoltage thresholds of DPA-Switch
simplifies deriving gate drive directly from the transformer
secondary winding for many applications. The turns ratio of
the transformer together with the under/overvoltage thresholds
defines the minimum and maximum gate voltages, removing
the need for Zeners to clamp the gate voltage.
DPA423-426
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20
the input decoupling capacitor. The SOURCE pin should not
be used to return the power currents; incorrect operation of the
device may result. The SOURCE is only intended as a signal
ground. The device tab (SOURCE) is the correct connection
for high current with the R package and S-PAK.
The CONTROL pin bypass capacitor should be located as
close as possible to the SOURCE and CONTROL pins and its
SOURCE connection trace should not be shared by the main
MOSFET switching currents. All SOURCE pin referenced
components connected to the LINE-SENSE or EXTERNAL
CURRENT LIMIT pins should also be located closely between
their respective pin and SOURCE. Once again, the SOURCE
connection trace of these components should not be shared
by the main MOSFET switching currents. It is critical that
the tab (SOURCE) power switching currents are returned to the
input capacitor through a separate trace that is not
shared by the components connected to CONTROL,
LINE-SENSE or EXTERNAL CURRENT LIMIT pins.
Any traces to the L or X pins should be kept as short as possible
and away from the DRAIN trace to prevent noise coupling.
LINE-SENSE resistor (R1 in Figure 25) should be located close
to the L pin to minimize the trace length on the L pin side.
In addition to the CONTROL pin capacitor (C6 in Figure 25),
a high frequency bypass capacitor in parallel is recommended
as close as possible between SOURCE and CONTROL pins for
better noise immunity. The feedback optocoupler output should
also be located close to the CONTROL and SOURCE pins of
DPA-Switch.
Heat Sinking
To maximize heat sinking of the DPA-Switch S, R or G package
and the other power components, special thermally conductive
PC board material (aluminum clad PC board) is recommended.
This has an aluminum sheet bonded to the PC board during
the manufacturing process to provide heat sinking directly
and allow the attachment of an external heat sink. If normal
PC board material is used (such as FR4), providing copper
areas on both sides of the board and using thicker copper will
improve heat sinking.
If an aluminum clad board is used then shielding of switching
nodes is recommended. This consists of an area of copper placed
directly underneath switching nodes such as the drain node,
and output diode to provide an electrostatic shield to prevent
coupling to the aluminum substrate. These areas are connected
to input negative in the case of the primary and output return
for secondary. This reduces the amount of capacitive coupling
into the insulated aluminum substrate that can then appear on
the output as ripple and high frequency noise.
Quick Design Checklist
As with any power supply design, all DPA-Switch designs
should be verified on the bench to make sure that component
specifications limits are not exceeded under worst case
conditions. The following minimum set of tests for DPA-Switch
forward converters is strongly recommended:
1. Maximum drain voltage – Verify that peak VDS does not
exceed minimum BVDSS at highest input voltage and
maximum overload output power. It is normal, however, to
have additional margin of approximately 25 V below BVDSS
to allow for other power supply component unit-to-unit
variations. Maximum overload output power occurs when
the output is loaded to a level just before the power supply
goes into auto-restart (loss of regulation).
2. Transformer reset margin – Drain voltage should also be
checked at highest input voltage with a severe load step
(50-100%) to verify adequate transformer reset margin. This
test shows the duty cycle at high input voltage, placing the
most demand on the transformer reset circuit.
3. Maximum drain current – At maximum ambient temperature,
maximum input voltage and maximum output load, verify
drain current waveforms at start-up for any signs of
transformer or output inductor saturation and excessive
leading edge current spikes. DPA-Switch has a leading edge
blanking time of 100 ns to prevent premature termination
of the on cycle. Verify that the leading edge current spike
does not extend beyond the blanking period.
4. Thermal check – At maximum output power, minimum
input voltage and maximum ambient temperature, verify
that temperature specifications are not exceeded for the
transformer, output diodes, output choke(s) and output
capacitors. The DPA-Switch is fully protected against over-
temperature conditions by its thermal shutdown feature. It
is recommended that sufficient heat sinking is provided
to keep the tab temperature at or below 115 °C (S and R
packages), SOURCE pins at or below 100 °C (P/G packages)
under worst case continuous load conditions (at low input
voltage, maximum ambient and full load). This provides
adequate margin to minimum thermal shutdown temperature
(130 °C) to account for part-to-part RDS(ON) variation. When
monitoring device temperatures, note that the junction-
to-case thermal resistance should be accounted for when
estimating die temperature.
Design Tools
Up-to-date information on design tools is available at the Power
Integrations website: www.powerint.com.
DPA423-426
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Figure 28. Layout Considerations for DPA-Switch Using S-PAK or R Package.
V
V
PI-2883-060602
Solder Side
Component Side
TOP VIEW
S
L
C
X
Transformer
-
+
V
+
-
DC
Out
Maximize hatched copper area for optimum heat sinking
V
Via between board layers
V
Opto-
coupler
Inductor
(Coupled)
Output
Diode
D
DPA-Switch
DC
In
Figure 29. Layout Considerations for DPA-Switch Using G Package.
PI-3805-012904
T
r
a
n
s
f
o
r
m
e
r
Opto-
coupler
+
DC
Out
DC
In
+
-
V
V
V
V
V
V
V
V
V
V
DPA-Switch Heatsink
Bottom Diode
Heatsink
-
V
V
C L X F
S D S
DPA-Switch
S
Top Side PCB
Bottom Side PCB
(Two sided printed circuit board)
TOP VIEW
Maximize hatched copper area for optimum heat sinking
Via between board layers
V
-
DPA423-426
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22
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 33
(Unless Otherwise Specified)
Min Typ Max Units
CONTROL FUNCTIONS
Switching
Frequency fOSC TJ = 25 °C
FREQUENCY Pin
Connected to SOURCE 375 400 425
kHz
FREQUENCY Pin
Connected to CONTROL 282 300 317
Duty Cycle (Prior
to Cycle Skipping) DCMIN 4 6 %
Maximum Duty
Cycle DCMAX IC = ICD1
VL = 0 V 71 75 79
%
IL = 80 µA52 62 71
IL = 115 µA32 42 57
Control Current
at Start of Cycle
Skipping
IC(skip) TJ = 25 °C; fOSC = 400 kHz
DPA423 7.2 9.0
mA
DPA424 8.2 10.0
DPA425 10.0 12.0
DPA426 11.5 14.0
External Bias
Current IBTJ = 25 °C; fOSC = 400 kHz
DPA423 2 2.8 3.5
mA
DPA424 2.5 3.5 4.4
DPA425 3.6 4.8 6.0
DPA426 4.4 5.7 7.1
ABSOLUTE MAXIMUM RATINGS(1,4)
DRAIN Voltage .................................. ................ -0.3 V to 220 V
DRAIN Peak Current: DPA423......................................1.75 A
DPA424....................................... 3.5 A
DPA425........................................... 7A
DPA426....................................... 9.6 A
CONTROL Voltage ................................................ -0.3 V to 9 V
CONTROL Current .................................................... 100 mA
LINE SENSE Pin Voltage ...................................... -0.3 V to 9 V
EXTERNAL CURRENT LIMIT Pin Voltage ........ -0.3 V to 9 V
FREQUENCY Pin Voltage .................................... -0.3 V to 9 V
Storage Temperature .......................................... -65 °C to 150 °C
Operating Junction Temperature(2) ..................... -40 °C to 150 °C
Lead Temperature(3) ...................................................... 260 °C
Notes:
1. All voltages referenced to SOURCE, TA = 25 °C.
2. Normally limited by internal circuitry.
3. 1/16” from case for 5 seconds.
4. Maximum ratings specified may be applied, one at a time,
without causing permanent damage to the product.
Exposure to Absolute Maximum Rating conditions for
extended periods of time may affect product reliability.
THERMAL IMPEDANCE
Thermal Impedance: P or G Package:
(θJA) ........................... 70 °C/W(1); 60 °C/W(2)
(θJC)(3) ............................................11 °C/W
R Package:
(θJA) ...............................................40 °C/W(4)
(θJA) ...............................................30 °C/W(5)
(θJC)(6) ...............................................2 °C/W
S-PAK:
(θJA) ...............................................49 °C/W(4)
(θJA) ...............................................39 °C/W(5)
(θJC)(6) ...............................................2 °C/W
Notes:
1. Soldered to 0.36 sq. in. (232 mm2), 2 oz. (610 g/m2) copper clad.
2. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.
3. Measured on pin 7 (SOURCE) close to plastic interface.
4. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.
5. Soldered to 3 sq. in. (1935 mm2), 2 oz. (610 g/m2) copper clad.
6. Measured at the back surface of tab.
DPA423-426
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5/06 23
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 33
(Unless Otherwise Specified)
Min Typ Max Units
CONTROL FUNCTIONS (cont.)
Softstart Time tSOFT TJ = 25 °C; DCMIN to DCMAX 5 7.2 ms
PWM Gain DCreg TJ = 25 °C; IC = (IC(skip) + IB)/2 -28 -22 -18 %/mA
PWM Gain
Temperature Drift See Note A -0.01 %/mA/°C
Dynamic
Impedance ZCTJ = 25 °C; IC = (IC(skip) + IB)/2 10 15 22 Ω
Dynamic
Impedance
Temperature Drift
0.18 %/°C
CONTROL Pin
Internal Filter Pole 30 kHz
SHUTDOWN/AUTO-RESTART
CONTROL Pin
Charging Current IC(CH)
During Startup and Auto-Restart:
VC = 5.0 V; VD = 16 V & 40 V; TJ = 25 °C -5.2 -4 -3
mA
Average Current at the Beginning of
Softstart: VC = 5.0 V;
VD = 16 V & 40 V; TJ = 25 °C
-19
Charging Current
Temperature Drift See Note A -0.6 %/°C
Auto-Restart
Upper Threshold
Voltage
VC(AR)U 5.8 V
Auto-Restart
Lower Threshold
Voltage
VC(AR)L 4.5 4.8 5.1 V
Auto-Restart
Hysteresis Voltage VC(AR)Hyst 0.8 1 V
Auto-Restart Duty
Cycle DC(AR)
CCONTROL = 22 µF; fOSC = 400 kHz;
VX = 0 V 10 %
Auto-Restart
Frequency f(AR)
CCONTROL = 22 µF; fOSC = 400 kHz;
VX = 0 V 3.8 Hz
LINE-SENSE (L) AND EXTERNAL CURRENT LIMIT (X) INPUTS
Line Under-
Voltage Threshold
Current and
Hysteresis (L Pin)
IUV TJ = 25 °C
Threshold from Off to On 48 50 52
µAThreshold from On to Off 44.5 47 49.5
Hysteresis 2 3
DPA423-426
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24
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 33
(Unless Otherwise Specified)
Min Typ Max Units
LINE-SENSE (L) AND EXTERNAL CURRENT LIMIT (X) INPUTS (cont.)
Line Overvoltage
or Remote ON/
OFF Threshold
Current and
Hysteresis (L Pin)
IOV TJ = 25 °C
Threshold from On to Off 135 149
µAThreshold from Off to On 117 131
Hysteresis 4
Remote ON/OFF
Negative Thresh-
old Current and
Hysteresis
(X Pin)
IREM TJ = 25 °C
Threshold from On to Off -27 -21.5 -16
µAThreshold from Off to On -25.5
Hysteresis 4.5
L Pin Short Circuit
Current IL(SC)
VL = VC175 240 380 µA
VL = 0 V -230 -170
X Pin Short Circuit
Current IX(SC) VX = 0 V Normal Mode -270 -230 -185 µA
Remote OFF using L Pin -105 -85 -65
Line Pin Voltage
(Positive Current) VL
IL = IUV 2.05 2.35 2.6
V
IL = IOV 2.1 2.5 2.9
X Pin Voltage
(Negative Current) VX
IX = -50 µA 1.35
IX = -150 µA 1.25
Maximum Duty
Cycle Reduction
Onset Threshold
Current
IL(DC) TJ = 25 °C 55 µA
Remote OFF
DRAIN Supply
Current
ID(RMT)
VD = 40 V
VX = 0 V
VL = Floating 0.6 1.1
mA
VL = VC0.9 1.5
L Pin Voltage
Turn-On Thresh-
old in Synchro-
nous Mode
VL(TH) 0.6 1 1.4 V
On-Time Pulse
Width for
Synchronization
ton(sync)
fOSC = 400 kHz 120 2250
ns
fOSC = 300 kHz 120 3080
Off-Time Pulse
Width for
Synchronization
toff(sync) 0.25 7.7 µs
Synchronous
Turn-On Delay tdelay(sync)
From Synchronous On to Drain
Turn-On 250 ns
DPA423-426
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5/06 25
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 33
(Unless Otherwise Specified)
Min Typ Max Units
FREQUENCY (F) INPUT
FREQUENCY Pin
Threshold Voltage VF1.1 4 V
FREQUENCY Pin
Input Current IF
VF = 0 V -0.38 µA
VF = VC17 120
FREQUENCY Pin
Delay Time tdelay(VF) 2µs
CIRCUIT PROTECTION
Self Protection
Current Limit (See
Note B)
ILIMIT TJ = 25 °C
DPA423 di/dt = 300 mA/µs 1.16 1.25 1.34
A
DPA424 di/dt = 600 mA/µs 2.32 2.50 2.68
DPA425 di/dt = 1.25 A/µs 4.65 5.00 5.35
DPA426 di/dt = 1.75 A/µs 6.50 7.00 7.50
Initial Current
Limit IINIT VD = 35 V 0.9 x
ILIMIT(min)
A
Leading Edge
Blanking Time tLEB TJ = 25 °C 100 ns
Current Limit
Delay tIL(D) IC = (IC(skip) + IB)/2 100 ns
Thermal Shut-
down Temperature TJ(SD) 130 137 145 °C
Thermal Shut-
down Hysteresis TJ(SD)hyst 27 °C
Power-Up Reset
Threshold Voltage VC(RESET) 1.5 2.75 4 V
OUTPUT
ON-State
Resistance RDS(ON)
DPA423
ID = 300 mA
TJ = 25 °C 1.30 1.50
Ω
TJ = 100 °C 2.00 2.30
DPA424
ID = 600 mA
TJ = 25 °C 0.65 0.75
TJ = 100 °C 1.00 1.15
DPA425
ID = 1.25 A
TJ = 25 °C 0.33 0.38
TJ = 100 °C 0.50 0.58
DPA426
ID = 1.75 A
TJ = 25 °C 0.24 0.28
TJ = 100 °C 0.37 0.43
OFF-State Drain
Leakage Current IDSS
VX, VL = Floating;
VD = 150 V;
TJ = 125 °C;
IC = (IC(skip) + IB)/2
DPA423 65
µA
DPA424 130
DPA425 260
DPA426 360
DPA423-426
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26
Parameter Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 33
(Unless Otherwise Specified)
Min Typ Max Units
OUTPUT (cont.)
Breakdown
Voltage BVDSS
VX, VL = Floating; TJ = 25 °C;
IC = (IC(skip) + IB)/2; See Note C 220 V
Rise Time tRMeasured in a Typical Application 10 ns
Fall Time tFMeasured in a Typical Application 10 ns
SUPPLY VOLTAGE CHARACTERISTICS
DRAIN Supply
Voltage See Note D 16 V
Shunt Regulator
Voltage VC(SHUNT) IC = (IC(skip) + IB)/2: TJ = 25 °C 5.6 5.85 6.0 V
Shunt Regulator
Temperature Drift IC = (IC(skip) + IB)/2 ±50 PPM/°C
CONTROL
Supply/Discharge
Current
ICD1
Output
MOSFET Enabled
VL = 0 V;
fOSC = 400 kHz
DPA423 1.9 2.3 2.7
mA
DPA424 2.6 3.0 3.4
DPA425 3.7 4.3 4.8
DPA426 4.8 5.4 6
Output MOSFET Disabled
VL = 0 V; fOSC = 400 kHz 0.4 0.73 1.2
ICD2
NOTES:
A. For specifications with negative values, a negative temperature coefficient corresponds to an increase in
magnitude with increasing temperature, and a positive temperature coefficient corresponds to a decrease in
magnitude with increasing temperature.
B. For externally adjusted current limit values, please refer to Figure 35 (Current Limit vs. External Current Limit
Resistance) in the Typical Performance Characteristics section.
C. Breakdown voltage may be checked against minimum BVDSS specification by ramping the DRAIN pin voltage up
to but not exceeding minimum BVDSS.
D. It is possible to start up and operate
DPA-Switch
at DRAIN voltages well below 16 V. However, the CONTROL
pin charging current is reduced, which affects start-up time, auto-restart frequency, and auto-restart duty cycle.
Refer to Figure 45, the characteristic graph on CONTROL pin charge current (IC) vs. DRAIN voltage for low
voltage operation characteristics.