DC053A-A - Linear Technology

LT1372/LT1377

500kHz and 1MHz

High Efficiency

1.5A Switching Regulators

■

Boost Regulators

■

CCFL Backlight Driver

■

Laptop Computer Supplies

■

Multiple Output Flyback Supplies

■

Inverting Supplies

The LT

1372/LT1377 are monolithic high frequency

switching regulators. They can be operated in all standard

switching configurations including boost, buck, flyback,

forward, inverting and “Cuk.” A 1.5A high efficiency switch

is included on the die, along with all oscillator, control and

protection circuitry. All functions of the LT1372/LT1377

are integrated into 8-pin SO/PDIP packages.

The LT1372/LT1377 typically consumes only 4mA quies-

cent current and has higher efficiency than previous parts.

High frequency switching allows for very small inductors

to be used. All surface mount components consume less

than 0.5 square inch of board space.

New design techniques increase flexibility and maintain

ease of use. Switching is easily synchronized to an exter-

nal logic level source. A logic low on the shutdown pin

reduces supply current to 12µA. Unique error amplifier

circuitry can regulate positive or negative output voltage

while maintaining simple frequency compensation tech-

niques. Nonlinear error amplifier transconductance re-

duces output overshoot on start-up or overload recovery.

Oscillator frequency shifting protects external compo-

nents during overload conditions.

■

Faster Switching with Increased Efficiency

■

Uses Small Inductors: 4.7µH

■

All Surface Mount Components

■

Only 0.5 Square Inch of Board Space

■

Low Minimum Supply Voltage: 2.7V

■

Quiescent Current: 4mA Typ

■

Current Limited Power Switch: 1.5A

■

Regulates Positive or Negative Outputs

■

Shutdown Supply Current: 12µA Typ

■

Easy External Synchronization

■

8-Pin SO or PDIP Packages

, LTC and LT are registered trademarks of Linear Technology Corporation.

12V Output Efficiency

OUTPUT CURRENT (A)

0.01

EFFICIENCY (%)

0.1 1

LT1372 • TA02

100 V

= 5V

5V-to-12V Boost Converter

LT1372/LT1377

6, 7

GND

LT1372 • TA01

S/S

L1*

4.7µH

C1**

22µFC4**

22µF

0.047µFC3

0.0047µF

6.19k

53.6k

OUT

†

12V

MBRS120T3

OFF *FOR LT1372 USE 10µH

COILCRAFT DO1608-472 (4.7µH) OR

COILCRAFT DT3316-103 (10µH) OR

SUMIDA CD43-4R7 (4.7µH) OR

SUMIDA CD73-100KC (10µH) OR

**AVX TPSD226M025R0200

4.7µH

10µH

OUT

(LT1377)

0.25A

0.35A

OUT

(LT1372)

0.29A

†

MAX I

OUT

+ +

FEATURES

DESCRIPTIO

APPLICATIO S

TYPICAL APPLICATIO

LT1372/LT1377

(Note 1)

Supply Voltage ....................................................... 30V

Switch Voltage

LT1372/LT1377 .................................................. 35V

LT1372HV .......................................................... 42V

S/S Pin Voltage....................................................... 30V

Feedback Pin Voltage (Transient, 10ms) .............. ±10V

Feedback Pin Current........................................... 10mA

Negative Feedback Pin Voltage

(Transient, 10ms)............................................. ±10V

Operating Junction Temperature Range

Commercial ........................................ 0°C to 125°C*

Industrial ......................................... –40°C to 125°C

Short Circuit ......................................... 0°C to 150°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

ORDER PART NUMBER

LT1372CN8

LT1372HVCN8

LT1372CS8

LT1372HVCS8

LT1372IN8

LT1372HVIN8

LT1372IS8

LT1372HVIS8

LT1377CS8

LT1377IS8

TOP VIEW

NFB

S/S

GND

GND S

N8 PACKAGE

8-LEAD PDIP S8 PACKAGE

8-LEAD PLASTIC SO

JMAX

= 125°C, θ

= 100°C/ W (N8)

JMAX

= 125°C, θ

= 120°C/ W (S8)

S8 PART MARKING

1372

1372I 1377

1377I

1372H

1372HI

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

REF

Reference Voltage Measured at Feedback Pin 1.230 1.245 1.260 V

= 0.8V ●1.225 1.245 1.265 V

Feedback Input Current V

= V

REF

250 550 nA

●900 nA

Reference Voltage Line Regulation 2.7V ≤ V

≤ 25V, V

= 0.8V ●0.01 0.03 %/V

NFB

Negative Feedback Reference Voltage Measured at Negative Feedback Pin –2.540 –2.490 –2.440 V

Feedback Pin Open, V

= 0.8V ●–2.570 –2.490 –2.410 V

NFB

Negative Feedback Input Current V

NFB

= V

NFR

●–45 –30 –15 µA

Negative Feedback Reference Voltage 2.7V ≤ V

≤ 25V, V

= 0.8V ●0.01 0.05 %/V

Line Regulation

Error Amplifier Transconductance ∆I

= ±25µA 1100 1500 1900 µmho

●700 2300 µmho

Error Amplifier Source Current V

= V

REF

– 150mV, V

= 1.5V ●120 200 350 µA

Error Amplifier Sink Current V

= V

REF

+ 150mV, V

= 1.5V ●1400 2400 µA

Error Amplifier Clamp Voltage High Clamp, V

= 1V 1.70 1.95 2.30 V

Low Clamp, V

= 1.5V 0.25 0.40 0.52 V

Error Amplifier Voltage Gain 500 V/V

Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V

f Switching Frequency 2.7V ≤ V

≤ 25V

LT1372 450 500 550 kHz

0°C ≤ T

≤ 125°C●430 500 580 kHz

–40°C ≤ T

< 0°C (I Grade) 400 580 kHz

LT1377 0.90 1 1.10 MHz

0°C ≤ T

≤ 125°C●0.86 1 1.16 MHz

–40°C ≤ T

< 0°C (I Grade) 0.80 1.16 MHz

The ● denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at TA = 25°C.

VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.

Consult factory for parts specified with wider operating temperature ranges.

*Units shipped prior to Date Code 9552 are rated at 100°C maximum

operating temperature.

ABSOLUTE AXI U RATI GS

WWWU

PACKAGE/ORDER I FOR ATIO

ELECTRICAL CHARACTERISTICS

LT1372/LT1377

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Maximum Switch Duty Cycle ●85 95 %

Switch Current Limit Blanking Time 130 260 ns

BV Output Switch Breakdown Voltage LT1372/LT1377 ●35 47 V

LT1372HV

0°C ≤ T

≤ 125°C●42 47 V

–40°C ≤ T

< 0°C (I Grade) 40 V

SAT

Output Switch “On” Resistance I

= 1A ●0.5 0.8 Ω

LIM

Switch Current Limit Duty Cycle = 50% ●1.5 1.9 2.7 A

Duty Cycle = 80% (Note 2) ●1.3 1.7 2.5 A

∆I

Supply Current Increase During Switch On-Time 15 25 mA/A

∆I

Control Voltage to Switch Current 2A/V

Transconductance

Minimum Input Voltage ●2.4 2.7 V

Supply Current 2.7V ≤ V

≤ 25V ●4 5.5 mA

Shutdown Supply Current 2.7V ≤ V

≤ 25V, V

S/S

≤ 0.6V

0°C ≤ T

≤ 125°C●12 30 µA

–40°C ≤ T

< 0°C (I Grade) 50 µA

Shutdown Threshold 2.7V ≤ V

≤ 25V ●0.6 1.3 2 V

Shutdown Delay ●51225µs

S/S Pin Input Current 0V ≤ V

S/S

≤ 5V ●–10 15 µA

Synchronization Frequency Range LT1372 ●600 800 kHz

LT1377 ●1.2 1.6 MHz

Switch Saturation Voltage

vs Switch Current

TEMPERATURE (°C)

–50

1.8

INPUT VOLTAGE (V)

2.0

2.2

2.4

2.6

050

100 150

LT1372 • G03

2.8

3.0

–25 25 75 125

Minimum Input Voltage

vs Temperature

DUTY CYCLE (%)

SWITCH CURRENT LIMIT (A)

1.0

2.0

3.0

0.5

1.5

2.5

20 40 60 80

LT1372 • G02

10010

030 50 70 90

25°C AND

125°C

–55°C

Switch Current Limit

vs Duty Cycle

TYPICAL PERFORMANCE CHARACTERISTICS

SWITCH CURRENT (A)

SWITCH SATURATION VOLTAGE (V)

0.6

0.8

1.0

1.6

LT1372 • G01

0.4

0.2

0.5

0.7

0.9

0.3

0.1

00.4 0.8 1.2 2.0

1.4

0.2 0.6 1.0 1.8

100°C

150°C

25°C

–55°C

Note 1: Absolute Maximum Ratings are those values beyond which the life

of the device may be impaired.

The ● denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at TA = 25°C.

VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.

ELECTRICAL CHARACTERISTICS

Note 2: For duty cycles (DC) between 50% and 90%, minimum

guaranteed switch current is given by I

LIM

= 0.667 (2.75 – DC).

LT1372/LT1377

Error Amplifier Output Current

vs Feedback Pin Voltage

Shutdown Delay and Threshold

vs Temperature

TEMPERATURE (°C)

–50

SHUTDOWN DELAY (µs)

SHUTDOWN THRESHOLD (V)

050 75

LT1372 • G04

0.2

0.6

0.8

1.0

2.0

1.4

0.4

1.6

1.8

1.2

–25 25 100 125 150

SHUTDOWN THRESHOLD

SHUTDOWN DELAY

S/S Pin Input Current

vs Voltage Error Amplifier Transconductance

vs Temperature

Switching Frequency

vs Feedback Pin Voltage

VC Pin Threshold and High

Clamp Voltage vs Temperature

FEEDBACK PIN VOLTAGE (V)

400

ERROR AMPLIFIER OUTPUT CURRENT (µA)

–300

–200

–100

300

100

–0.1 0.1

200

–0.3 –0.2 V

REF

–55°C

125°C

25°C

LT1372 • G06

Minimum Synchronization

Voltage vs Temperature

TEMPERATURE (°C)

–50

MINIMUM SYNCHRONIZATION VOLTAGE (V

P-P

)

0.5

1.0

1.5

2.0

050

100 150

LT1372 • G05

2.5

3.0

–25 25 75 125

SYNC

= 700kHz (LT1372)

SYNC

= 1.4MHz (LT1377)

LT1377

LT1372

S/S PIN VOLTAGE (V)

–1

S/S PIN INPUT CURRENT (µA)

LT1372 • G07

–1

–3

–2

–4

–5 135

2469

= 5V

FEEDBACK PIN VOLTAGE (V)

SWITCHING FREQUENCY (% OF TYPICAL)

110

0.8

LT1372 • G08

100

10 0.2 0.4 0.6

0.1 0.9

0.3 0.5 0.7 1.0

TEMPERATURE (°C)

–50

TRANSCONDUCTANCE (µmho)

200

600

800

1000

2000

1400

050 75

LT1372 • G09

400

1600

1800

1200

–25 25 100 125 150

= ∆I (V

)

∆V (FB)

TEMPERATURE (°C)

–50

0.4

PIN VOLTAGE (V)

0.6

1.0

1.2

1.4

2.4

1.8

050 75

LT1372 • G10

0.8

2.0

2.2

1.6

–25 25 100 125 150

HIGH CLAMP

THRESHOLD

TEMPERATURE (°C)

–50

FEEDBACK INPUT CURRENT (nA)

400

500

600

150

LT1372 • G11

300

200

0050 100

100

800

700

–25 25 75 125

REF

Feedback Input Current

vs Temperature

TEMPERATURE (°C)

–50

NEGATIVE FEEDBACK INPUT CURRENT (µA)

–30

050 75

LT1372 • G12

–40

–10

–20

–25 25 100 125 150

NFB

NFR

Negative Feedback Input Current

vs Temperature

TYPICAL PERFOR A CE CHARACTERISTICS

LT1372/LT1377

(Pin 1): The compensation pin is used for frequency

compensation, current limiting and soft start. It is the

output of the error amplifier and the input of the current

comparator. Loop frequency compensation can be per-

formed with an RC network connected from the V

pin to

ground.

FB (Pin 2): T

he feedback pin is used for positive output

voltage sensing and oscillator frequency shifting. It is the

inverting input to the error amplifier. The noninverting

input of this amplifier is internally tied to a 1.245V

reference. Load on the FB pin should not exceed 250µA

when the NFB pin is used. See Applications Information.

NFB (Pin 3): The negative feedback pin is used for negative

output voltage sensing. It is connected to the inverting

input of the negative feedback amplifier through a 100k

source resistor.

S/S (Pin 4): Shutdown and Synchronization Pin. The S/S

pin is logic level compatible. Shutdown is active low and

the shutdown threshold is typically 1.3V. For normal

operation, pull the S/S pin high, tie it to V

or leave it

floating. To synchronize switching, drive the S/S pin be-

tween 600kHz and 800kHz (LT1372) or 1.2MHz to 1.6MHz

(LT1377).

(Pin 5): Bypass input supply pin with 10µF or more. The

part goes into undervoltage lockout when V

drops below

2.5V. Undervoltage lockout stops switching and pulls the

pin low.

GND S (Pin 6): The ground sense pin is a “clean” ground.

The internal reference, error amplifier and negative feed-

back amplifier are referred to the ground sense pin. Con-

nect it to ground. Keep the ground path connection to the

output resistor divider and the V

compensation network

free of large ground currents.

GND (Pin 7): The ground pin is the emitter connection of

the power switch and has large currents flowing through it.

It should be connected directly to a good quality ground

plane.

(Pin 8): The switch pin is the collector of the power

switch and has large currents flowing through it. Keep the

traces to the switching components as short as possible to

minimize radiation and voltage spikes.

–

NFBA

NFB

S/S

100k

50k

0.08Ω

–

VIN

GND LT1372 • BD

GND SENSE

1.245V

REF

5:1 FREQUENCY

SHIFT

OSCSYNC

SHUTDOWN

DELAY AND RESET LOW DROPOUT

2.3V REG ANTI-SAT

LOGIC DRIVER

SWITCH

–

AV ≈ 6

COMP

PI FU CTIO S

BLOCK DIAGRA

LT1372/LT1377

The LT1372/LT1377 are current mode switchers. This

means that switch duty cycle is directly controlled by

switch current rather than by output voltage. Referring to

the block diagram, the switch is turned “On” at the start of

each oscillator cycle. It is turned “Off” when switch current

reaches a predetermined level. Control of output voltage is

obtained by using the output of a voltage sensing error

amplifier to set current trip level. This technique has

several advantages. First, it has immediate response to

input voltage variations, unlike voltage mode switchers

which have notoriously poor line transient response.

Second, it reduces the 90° phase shift at mid-frequencies

in the energy storage inductor. This greatly simplifies

closed-loop frequency compensation under widely vary-

ing input voltage or output load conditions. Finally, it

allows simple pulse-by-pulse current limiting to provide

maximum switch protection under output overload or

short conditions. A low dropout internal regulator pro-

vides a 2.3V supply for all internal circuitry. This low

dropout design allows input voltage to vary from 2.7V to

25V with virtually no change in device performance. A

500kHz (LT1372) or 1MHz (LT1377) oscillator is the basic

clock for all internal timing. It turns “On” the output switch

via the logic and driver circuitry. Special adaptive anti-sat

circuitry detects onset of saturation in the power switch

and adjusts driver current instantaneously to limit switch

saturation. This minimizes driver dissipation and provides

very rapid turn-off of the switch.

A 1.245V bandgap reference biases the positive input of

the error amplifier. The negative input of the amplifier is

brought out for positive output voltage sensing. The error

amplifier has nonlinear transconductance to reduce out-

put overshoot on start-up or overload recovery. When

the feedback voltage exceeds the reference by 40mV,

error amplifier transconductance increases ten times,

which reduces output overshoot. The feedback input also

invokes oscillator frequency shifting, which helps pro-

tect components during overload conditions. When the

feedback voltage drops below 0.6V, the oscillator fre-

quency is reduced 5:1. Lower switching frequency allows

full control of switch current limit by reducing minimum

switch duty cycle.

Unique error amplifier circuitry allows the LT1372/LT1377

to directly regulate negative output voltages. The negative

feedback amplifier’s 100k source resistor is brought out

for negative output voltage sensing. The NFB pin regulates

at –2.49V while the amplifier output internally drives the

FB pin to 1.245V. This architecture, which uses the same

main error amplifier, prevents duplicating functions and

maintains ease of use. Consult Linear Technology Market-

ing for units that can regulate down to –1.25V.

The error signal developed at the amplifier output is

brought out externally. This pin (V

) has three different

functions. It is used for frequency compensation, current

limit adjustment and soft starting. During normal regula-

tor operation this pin sits at a voltage between 1V (low

output current) and 1.9V (high output current). The error

amplifier is a current output (g

) type, so this voltage can

be externally clamped for lowering current limit. Like-

wise, a capacitor coupled external clamp will provide soft

start. Switch duty cycle goes to zero if the V

pin is pulled

below the control pin threshold, placing the LT1372/

LT1377 in an idle mode.

Positive Output Voltage Setting

The LT1372/LT1377 develops a 1.245V reference (V

REF

)

from the FB pin to ground. Output voltage is set by

connecting the FB pin to an output resistor divider

(Figure 1). The FB pin bias current represents a small

error and can usually be ignored for values of R2 up to 7k.

The suggested value for R2 is 6.19k. The NFB pin is

normally left open for positive output applications.

Figure 1. Positive Output Resistor Divider

R1 V

OUT

= V

REF

1 +

REF

OUT

()

R1 = R2 – 1

()

OUT

1.245

LT1372 • F01

OPERATIO

APPLICATIO S I FOR ATIO

WUUU

LT1372/LT1377

Positive fixed voltage versions are available (consult

Linear Technology marketing).

Negative Output Voltage Setting

The LT1372/LT1377 develops a –2.49V reference (V

NFR

)

from the NFB pin to ground. Output voltage is set by

connecting the NFB pin to an output resistor divider

(Figure 2). The –30µA NFB pin bias current (I

NFB

) can

cause output voltage errors and should not be ignored.

This has been accounted for in the formula in Figure 2. The

suggested value for R2 is 2.49k. The FB pin is normally left

open for negative output application. See Dual Polarity

Output Voltage Sensing for limitatins on FB pin loading

when using the NFB pin.

Shutdown and Synchronization

The dual function S/S pin provides easy shutdown and

synchronization. It is logic level compatible and can be

pulled high, tied to V

or left floating for normal operation.

A logic low on the S/S pin activates shutdown, reducing

the part’s supply current to 12µA. Typical synchronization

range is from 1.05 to 1.8 times the part’s natural switching

frequency, but is only guaranteed between 600kHz and

800kHz (LT1372) or 1.2MHz and 1.6MHz (LT1377). At

start-up, the synchronization signal should not be applied

until the feedback pin is above the frequency shift voltage

of 0.7V. If the NFB pin is used, synchronization should not

be applied until the NFB pin is more negative than –1.4V.

A 12µs resetable shutdown delay network guarantees the

part will not go into shutdown while receiving a synchro-

nization signal.

Caution should be used when synchronizing above 700kHz

(LT1372) or 1.4MHz (LT1377) because at higher sync

frequencies the amplitude of the internal slope compensa-

tion used to prevent subharmonic switching is reduced.

This type of subharmonic switching only occurs when the

duty cycle of the switch is above 50%. Higher inductor

values will tend to eliminate problems.

Thermal Considerations

Care should be taken to ensure that the worst-case input

voltage and load current conditions do not cause exces-

sive die temperatures. The packages are rated at 120°C/W

for SO (S8) and 130°C/W for PDIP (N8).

Average supply current (including driver current) is:

= 4mA + DC (I

/60 + I

× 0.004)

= switch current

DC = switch duty cycle

Switch power dissipation is given by:

= (I

)

× R

× DC

= output switch “On” resistance

Total power dissipation of the die is the sum of supply

current times supply voltage plus switch power:

D(TOTAL)

= (I

× V

) + P

Dual Polarity Output Voltage Sensing

Certain applications benefit from sensing both positive

and negative output voltages. One example is the “Dual

Output Flyback Converter with Overvoltage Protection”

circuit shown in the Typical Applications section. Each

output voltage resistor divider is individually set as de-

scribed above. When both the FB and NFB pins are used,

the LT1372/LT1377 acts to prevent either output from

going beyond its set output voltage. For example in this

application, if the positive output were more heavily loaded

than the negative, the negative output would be greater

and would regulate at the desired set-point voltage. The

positive output would sag slightly below its set-point

voltage. This technique prevents either output from going

unregulated high at no load. Please note that the load on

the FB pin should not exceed 250µA when the NFB pin is

used. This situation occurs when the resistor dividers are

used at

both

FB and NFB. True load on FB is not the full

divider current unless the positive output is shorted to

ground. See Dual Output Flyback Converter application.

R1 –V

OUT

= V

NFB

+ I

NFB

(R1)1 +

LT1372 • F02

NFB

NFR

NFB

–V

OUT

()

R1 =

+ 30 × 10

–6

V

OUT

– 2.49

( ) ( )

2.49

Figure 2. Negative Output Resistor Divider

APPLICATIO S I FOR ATIO

WUUU

LT1372/LT1377

Choosing the Inductor

For most applications the inductor will fall in the range of

2.2µH to 22µH. Lower values are chosen to reduce physi-

cal size of the inductor. Higher values allow more output

current because they reduce peak current seen by the

power switch, which has a 1.5A limit. Higher values also

reduce input ripple voltage and reduce core loss.

When choosing an inductor you might have to consider

maximum load current, core and copper losses, allowable

component height, output voltage ripple, EMI, fault

current in the inductor, saturation, and of course, cost.

The following procedure is suggested as a way of handling

these somewhat complicated and conflicting requirements.

1. Assume that the average inductor current for a boost

converter is equal to load current times V

OUT

and

decide whether or not the inductor must withstand

continuous overload conditions. If average inductor

current at maximum load current is 0.5A, for instance,

a 0.5A inductor may not survive a continuous 1.5A

overload condition. Also be aware that boost convert-

ers are not short circuit protected, and that under

output short conditions, inductor current is limited only

by the available current of the input supply.

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current

can be significantly higher than output current, espe-

cially with smaller inductors and lighter loads, so don’t

omit this step. Powdered iron cores are forgiving be-

cause they saturate softly, whereas ferrite cores satu-

rate abruptly. Other core materials fall in between

somewhere. The following formula assumes continu-

ous mode operation but it errors only slightly on the

high side for discontinuous mode, so it can be used for

all conditions.

IPEAK = IOUT ×

VIN = Minimum Input Voltage

f = 500kHz Switching Frequency (LT1372) or

1MHz Switching Frequency (LT1377)

VOUT

VIN

VIN(VOUT – VIN)

2(f)(L)(VOUT)

3. Decide if the design can tolerate an “open” core geom-

etry like a rod or barrel, which have high magnetic field

radiation, or whether it needs a closed core like a toroid

to prevent EMI problems. One would not want an open

core next to a magnetic storage media for instance!

This is a tough decision because the rods or barrels are

temptingly cheap and small, and there are no helpful

guidelines to calculate when the magnetic field radia-

tion will be a problem.

4. Start shopping for an inductor which meets the re-

quirements of core shape, peak current (to avoid

saturation), average current (to limit heating) and fault

current. If the inductor gets too hot, wire insulation will

melt and cause turn-to-turn shorts. Keep in mind that

all good things like high efficiency, low profile and high

temperature operation will increase cost, sometimes

dramatically.

5. After making an initial choice, consider the secondary

things like output voltage ripple, second sourcing, etc.

Use the experts in the Linear Technology application

department if you feel uncertain about the final choice.

They have experience with a wide range of inductor

types and can tell you about the latest developments in

low profile, surface mounting, etc.

Output Capacitor

The output capacitor is normally chosen by its effective

series resistance, (ESR), because this is what determines

output ripple voltage. At 500kHz, any polarized capacitor

is essentially resistive. To get low ESR takes

volume

, so

physically smaller capacitors have high ESR. The ESR

range for typical LT1372 and LT1377 applications is

0.05Ω to 0.5Ω. A typical output capacitor is an AVX type

TPS, 22µF at 25V, with a guaranteed ESR less than 0.2Ω.

This is a “D” size surface mount solid tantalum capacitor.

TPS capacitors are specially constructed and tested for

low ESR, so they give the lowest ESR for a given volume.

To further reduce ESR, multiple output capacitors can be

used in parallel. The value in microfarads is not particu-

larly critical, and values from 22µF to greater than 500µF

work well, but you cannot cheat mother nature on ESR.

If you find a tiny 22µF solid tantalum capacitor, it will have

high ESR, and output ripple voltage will be terrible. Table

1 shows some typical solid tantalum surface mount

capacitors.

APPLICATIO S I FOR ATIO

WUUU

LT1372/LT1377

RIPPLE

f = 500kHz Switching frequency (LT1372) or,

1MHz Switching frequency (LT1377)

0.3(V

)(V

OUT

– V

)

(f)(L)(V

OUT

)

The input capacitor can see a very high surge current when

a battery or high capacitance source is connected “live”

and solid tantalum capacitors can fail under this condition.

Several manufacturers have developed a line of solid

tantalum capacitors specially tested for surge capability

(AVX TPS series, for instance), but even these units may

fail if the input voltage approaches the maximum voltage

rating of the capacitor. AVX recommends derating capaci-

tor voltage by 2:1 for high surge applications. Ceramic and

aluminum electrolytic capacitors may also be used and

have a high tolerance to turn-on surges.

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now

becoming available in smaller case sizes. These are tempt-

ing for switching regulator use because of their very low

ESR. Unfortunately, the ESR is so low that it can cause

loop stability problems. Solid tantalum capacitor ESR

generates a loop “zero” at 5kHz to 50kHz that is instrumen-

tal in giving acceptable loop phase margin. Ceramic ca-

pacitors remain capacitive to beyond 300kHz and usually

resonate with their ESL before ESR becomes effective.

They are appropriate for input bypassing because of their

high ripple current ratings and tolerance of turn-on surges.

Linear Technology plans to issue a Design Note on the use

of ceramic capacitors in the near future.

Output Diode

The suggested output diode (D1) is a 1N5818 Schottky or

its Motorola equivalent, MBR130. It is rated at 1A average

forward current and 30V reverse voltage. Typical forward

voltage is 0.42V at 1A. The diode conducts current only

during switch off time. Peak reverse voltage for boost

converters is equal to regulator output voltage. Average

forward current in normal operation is equal to output

current.

Table 1. Surface Mount Solid Tantalum Capacitor

ESR and Ripple Current

E CASE SIZE ESR (MAX Ω) RIPPLE CURRENT (A)

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

AVX TAJ 0.7 to 0.9 0.4

D CASE SIZE

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

AVX TAJ 0.9 to 2.0 0.36 to 0.24

C CASE SIZE

AVX TPS 0.2 (Typ) 0.5 (Typ)

AVX TAJ 1.8 to 3.0 0.22 to 0.17

B CASE SIZE

AVX TAJ 2.5 to 10 0.16 to 0.08

Many engineers have heard that solid tantalum capacitors

are prone to failure if they undergo high surge currents.

This is historically true and type TPS capacitors are

specially tested for surge capability, but surge ruggedness

is not a critical issue with the

output

capacitor. Solid

tantalum capacitors fail during very high

turn-on

surges,

which do not occur at the output of regulators. High

discharge

surges, such as when the regulator output is

dead shorted, do not harm the capacitors.

Single inductor boost regulators have large RMS ripple

current in the output capacitor, which must be rated to

handle the current. The formula to calculate this is:

Output Capacitor Ripple Current (RMS)

IRIPPLE (RMS) = IOUT

= IOUT

VOUT – VIN

VIN

1 – DC

Input Capacitors

The input capacitor of a boost converter is less critical due

to the fact that the input current waveform is triangular and

does not contain large squarewave currents as is found in

the output capacitor. Capacitors in the range of 10µF to

100µF with an ESR of 0.3Ω or less work well up to full 1.5A

switch current. Higher ESR capacitors may be acceptable

at low switch currents. Input capacitor ripple current for

boost converter is :

APPLICATIO S I FOR ATIO

WUUU

LT1372/LT1377

(magnetic) radiation is minimized by keeping output di-

ode, switch pin, and output bypass capacitor leads as

short as possible. E field radiation is kept low by minimiz-

ing the length and area of all traces connected to the switch

pin. A ground plane should always be used under the

switcher circuitry to prevent interplane coupling.

The high speed switching current path is shown schemati-

cally in Figure 3. Minimum lead length in this path is

essential to ensure clean switching and low EMI. The path

including the switch, output diode, and output capacitor is

the only one containing nanosecond rise and fall times.

Keep this path as short as possible.

LOAD

OUT

L1 SWITCH

NODE

LT1372 • F03

HIGH

FREQUENCY

CIRCULATING

PATH

Figure 3

More Help

For more detailed information on switching regulator

circuits, please see Application Note 19. Linear Technol-

ogy also offers a computer software program, SwitcherCAD,

to assist in designing switching converters. In addition,

our applications department is always ready to lend a

helping hand.

Frequency Compensation

Loop frequency compensation is performed on the output

of the error amplifier (V

pin) with a series RC network.

The main pole is formed by the series capacitor and the

output impedance (≈500kΩ) of the error amplifier. The

pole falls in the range of 2Hz to 20Hz. The series resistor

creates a “zero” at 1kHz to 5kHz, which improves loop

stability and transient response. A second capacitor,

typically one-tenth the size of the main compensation

capacitor, is sometimes used to reduce the switching

frequency ripple on the V

pin. V

pin ripple is caused by

output voltage ripple attenuated by the output divider and

multiplied by the error amplifier. Without the second

capacitor, V

pin ripple is:

Pin Ripple =

RIPPLE

= Output ripple (V

P–P

)

= Error amplifier transconductance

(≈1500µmho)

= Series resistor on V

OUT

= DC output voltage

1.245(V

RIPPLE

)(g

)(R

)

OUT

)

To prevent irregular switching, V

pin ripple should be

kept below 50mV

P–P

Worst-case V

pin ripple occurs at

maximum output load current and will also be increased

if poor quality (high ESR) output capacitors are used. The

addition of a 0.0047µF capacitor on the V

pin reduces

switching frequency ripple to only a few millivolts. A low

value for R

will also reduce V

pin ripple, but loop phase

margin may be inadequate.

Switch Node Considerations

For maximum efficiency, switch rise and fall time are

made as short as possible. To prevent radiation and high

frequency resonance problems, proper layout of the com-

ponents connected to the switch node is essential. B field

APPLICATIO S I FOR ATIO

WUUU

LT1372/LT1377

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection of circuits as described herein will not infringe on existing patent rights.

TYPICAL APPLICATIONS N

Low Ripple 5V to –3V “Cuk”

†

Converter

LT1372/LT1377

S/S

GND

GND S

NFB

2k R2

4.99k

0.047µF

0.1µF

OUT

–3V

250mA

LT1372 • TA05

47µF

16V

22µF

10V

47µF

16V

0.0047µF

•• 41

L1*

D1**

SUMIDA CLS62-100L

MOTOROLA MBR0520LT3

PATENTS MAY APPLY

†

Dual Output Flyback Converter with Overvoltage ProtectionPositive-to-Negative Converter with Direct Feedback

LT1372/LT1377

2.7V TO 13V

6, 7

*DALE LPE-4841-100MB (605) 665-9301

GND

NFB

LT1372 • TA04

S/S

P6KE-20A

1N4148

MBRS140T3

22µF

1.21k

13k

0.047µF

0.0047µFR3

2.49k

12.1k

–V

OUT

–15V

OUT

15V

47µF

OFF

2, 3

6, 7

T1*

•

LT1372/LT1377

2.7V TO 16V

6, 7

*COILTRONICS CTX10-2 (407) 241-7876

GND

NFB

LT1372 • TA03

S/S

P6KE-15A

1N4148

MBRS130LT3

22µF

0.047µF

0.0047µFR1

2.49k

–V

OUT†

–5V

47µF

OFF

OUT

0.3A

0.5A

0.75A

T1*

•

†

MAX I

OUT

1N4148

1N5818

1N4148

562Ω*

20k

DIMMING

10k

330Ω

12345

10µFC1

0.1µF

4.5V

TO 30V

GND

S/S

16, 7

LT1372/LT1377

2µF

0.1µF

33µH

LT1372 • TA06

C1 = WIMA MKP-20

L1 = COILCRAFT DT3316-333

T1 = COILTRONICS CTX 110609

* = 1% FILM RESISTOR

DO NOT SUBSTITUTE COMPONENTS

Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001

LAMP

27pF

5mA MAX

2.2µF

2.7V TO

5.5V

22k

1N4148

OPTIONAL REMOTE

DIMMING

COILTRONICS (407) 241-7876

COILCRAFT (708) 639-6400

OFF

CCFL BACKLIGHT APPLICATION CIRCUITS

CONTAINED IN THIS DATA SHEET ARE

COVERED BY U.S. PATENT NUMBER 5408162

AND OTHER PATENTS PENDING

90% Efficient CCFL Supply

LT1372/LT1377

TYPICAL APPLICATIONS N

2 Li-Ion Cell to 5V SEPIC Converter

LT1372/LT1377

GND

4V TO 9V

6, 7

LT1372 • TA07

S/S

33µF

20V

0.047µFC5

0.0047µF

6.19k

18.7k

OUT†

100µF

10V

OFF

OUT

0.45A

0.55A

0.65A

0.72A

L1A*

10µH

•

L1B*

10µH

1µF

C1 = AVX TPSD 336M020R0200

C2 = TOKIN 1E105ZY5U-C103-F

C3 = AVX TPSD107M010R0100

*SINGLE INDUCTOR WITH TWO WINDINGS

COILTRONICS CTX10-1

†

MAX I

OUT

MBRS130LT3

RELATED PARTS

PACKAGE DESCRIPTION

Dimensions in inches (millimeters) unless otherwise noted.

PART NUMBER DESCRIPTION COMMENTS

LT1370 High Efficiency DC/DC Converter 42V, 6A, 500kHz Switch

LT1767 1.5A, 1.25MHz Step-Down Switching Regulator 3V to 25V Input, V

REF

= 1.2V, Synchronizable Up to 2MHz, MSOP Package

LT1374 High Efficiency Step-Down Switching Regulator 25V, 4.5A, 500kHz Switch

LTC1735-1 High Efficiency Step-Down Controller with Power Good Output Fault Protection, 16-Pin SSOP and SO-8

LTC®3402 Single Cell, High Current (2A), Micropower, Synchronous V

= 0.7V to 5V, Up to 95% Efficiency Synchronizable Oscillator

3MHz Step-Up DC/DC Converter from 100kHz to 3MHz

 LINEAR TECHNOLOGY CORPORATION 1995

sn13727 13727fbs LT/TP 0401 2K REV B • PRINTED IN THE USA

N8 Package

8-Lead PDIP (Narrow 0.300)

(LTC DWG # 05-08-1510)

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

●

FAX: (408) 434-0507

●

www.linear-tech.com

N8 1098

0.009 – 0.015

(0.229 – 0.381)

0.300 – 0.325

(7.620 – 8.255)

0.325 +0.035

–0.015

+0.889

–0.381

8.255

()

0.100

(2.54)

BSC

0.065

(1.651)

TYP

0.045 – 0.065

(1.143 – 1.651)

0.130 ± 0.005

(3.302 ± 0.127)

0.020

(0.508)

MIN

0.018 ± 0.003

(0.457 ± 0.076)

0.125

(3.175)

MIN

12 34

8765

0.255 ± 0.015*

(6.477 ± 0.381)

0.400*

(10.160)

MAX

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)

0.016 – 0.050

(0.406 – 1.270)

0.010 – 0.020

(0.254 – 0.508)× 45°

0°– 8° TYP

0.008 – 0.010

(0.203 – 0.254)

SO8 1298

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

TYP

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

BSC

1234

0.150 – 0.157**

(3.810 – 3.988)

8765

0.189 – 0.197*

(4.801 – 5.004)

0.228 – 0.244

(5.791 – 6.197)

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE