10 MHz, 20 V/s, G = 1, 2, 4, 8 iCMOS Programmable Gain Instrumentation Amplifier AD8251 FEATURES FUNCTIONAL BLOCK DIAGRAM DGND WR Small package: 10-lead MSOP Programmable gains: 1, 2, 4, 8 Digital or pin-programmable gain setting Wide supply: 5 V to 15 V Excellent dc performance High CMRR: 98 dB (minimum), G = 8 Low gain drift: 10 ppm/C (maximum) Low offset drift: 1.8 V/C (maximum), G = 8 Excellent ac performance Fast settling time: 785 ns to 0.001% (maximum) High slew rate: 20 V/s (minimum) Low distortion: -110 dB THD at 1 kHz, 10 V swing High CMRR over frequency: 80 dB to 50 kHz (minimum) Low noise: 18 nV/Hz, G = 8 (maximum) Low power: 4.1 mA 2 4 OUT 7 +IN 10 8 3 9 +VS -VS REF 06287-001 AD8251 Figure 1. 25 20 Data acquisition Biomedical analysis Test and measurement G=8 15 GAIN (dB) G=4 GENERAL DESCRIPTION The AD8251 user interface consists of a parallel port that allows users to set the gain in one of two ways (see Figure 1). A 2-bit word sent via a bus can be latched using the WR input. An alternative is to use the transparent gain mode where the state of the logic levels at the gain port determines the gain. A0 5 LOGIC -IN 1 APPLICATIONS 10 G=2 5 G=1 0 -5 06287-002 The AD8251 is an instrumentation amplifier with digitally programmable gains that has G input impedance, low output noise, and low distortion, making it suitable for interfacing with sensors and driving high sample rate analog-to-digital converters (ADCs). It has a high bandwidth of 10 MHz, low THD of -110 dB, and fast settling time of 785 ns (maximum) to 0.001%. Offset drift and gain drift are guaranteed to 1.8 V/C and 10 ppm/C, respectively, for G = 8. In addition to its wide input common voltage range, it boasts a high common-mode rejection of 80 dB at G = 1 from dc to 50 kHz. The combination of precision dc performance coupled with high speed capabilities makes the AD8251 an excellent candidate for data acquisition. Furthermore, this monolithic solution simplifies design and manufacturing and boosts performance of instrumentation by maintaining a tight match of internal resistors and amplifiers. A1 6 -10 1k 10k 100k 1M 100M 10M FREQUENCY (Hz) Figure 2. Gain vs. Frequency Table 1. Instrumentation Amplifiers by Category General Purpose AD82201 AD8221 AD8222 AD82241 AD8228 1 Zero Drift AD82311 AD85531 AD85551 AD85561 AD85571 Mil Grade AD620 AD621 AD524 AD526 AD624 Low Power AD6271 AD6231 AD82231 High Speed PGA AD8250 AD8251 AD8253 Rail-to-rail output. The AD8251 is available in a 10-lead MSOP package and is specified over the -40C to +85C temperature range, making it an excellent solution for applications where size and packing density are important considerations. Rev. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2007-2010 Analog Devices, Inc. All rights reserved. AD8251 TABLE OF CONTENTS Features .............................................................................................. 1 Power Supply Regulation and Bypassing ................................ 18 Applications....................................................................................... 1 Input Bias Current Return Path ............................................... 18 General Description ......................................................................... 1 Input Protection ......................................................................... 18 Functional Block Diagram .............................................................. 1 Reference Terminal .................................................................... 19 Revision History ............................................................................... 2 Common-Mode Input Voltage Range ..................................... 19 Specifications..................................................................................... 3 Layout .......................................................................................... 19 Timing Diagram ........................................................................... 5 RF Interference ........................................................................... 20 Absolute Maximum Ratings............................................................ 6 Driving an ADC ......................................................................... 20 Maximum Power Dissipation ..................................................... 6 Applications..................................................................................... 21 ESD Caution.................................................................................. 6 Differential Output .................................................................... 21 Pin Configuration and Function Descriptions............................. 7 Setting Gains with a Microcontroller ...................................... 21 Typical Performance Characteristics ............................................. 8 Data Acquisition......................................................................... 22 Theory of Operation ...................................................................... 16 Outline Dimensions ....................................................................... 23 Gain Selection ............................................................................. 16 Ordering Guide .......................................................................... 23 REVISION HISTORY 11/10--Rev. A to Rev. B Changes to Voltage Offset, Offset RTI VOS, Average TC Parameter in Table 2......................................................................... 3 Updated Outline Dimensions ....................................................... 23 5/08--Rev. 0 to Rev. A Changes to Table 1............................................................................ 1 Changes to Table 2.............................................................................3 Changes to Table 3.............................................................................6 Inserted Figure 17; Renumbered Sequentially ..............................9 Inserted Figure 29........................................................................... 11 Changes to Timing for Latched Gain Mode Section ................. 17 5/07--Revision 0: Initial Version Rev. B | Page 2 of 24 AD8251 SPECIFICATIONS +VS = 15 V, -VS = -15 V, VREF = 0 V @ TA = 25C, G = 1, RL = 2 k, unless otherwise noted. Table 2. Parameter COMMON-MODE REJECTION RATIO (CMRR) CMRR to 60 Hz with 1 k Source Imbalance G=1 G=2 G=4 G=8 CMRR to 50 kHz G=1 G=2 G=4 G=8 NOISE Voltage Noise, 1 kHz, RTI G=1 G=2 G=4 G=8 0.1 Hz to 10 Hz, RTI G=1 G=2 G=4 G=8 Current Noise, 1 kHz Current Noise, 0.1 Hz to 10 Hz VOLTAGE OFFSET Offset RTI VOS Over Temperature Average TC Offset Referred to the Input vs. Supply (PSR) INPUT CURRENT Input Bias Current Over Temperature Average TC Input Offset Current Over Temperature Average TC DYNAMIC RESPONSE Small Signal -3 dB Bandwidth G=1 G=2 G=4 G=8 Settling Time 0.01% G=1 G=2 G=4 G=8 Conditions Min Typ 80 86 92 98 98 104 110 110 Max Unit +IN = -IN = -10 V to +10 V dB dB dB dB +IN = -IN = -10 V to +10 V 80 84 86 86 dB dB dB dB 40 27 22 18 nV/Hz nV/Hz nV/Hz nV/Hz 2.5 2.5 1.8 1.2 V p-p V p-p V p-p V p-p pA/Hz pA p-p (70 + 200/G) (90 + 300/G) (0.6 + 1.5/G) (2 + 7/G) (200 + 600/G) (260 + 900/G) (1.2 + 5/G) (6 + 20/G) V V V/C V/V 5 30 40 400 30 30 160 nA nA pA/C nA nA pA/C 5 60 G = 1, 2, 4, 8 T = -40C to +85C T = -40C to +85C VS = 5 V to 15 V T = -40C to +85C T = -40C to +85C 5 T = -40C to +85C T = -40C to +85C 10 10 8 2.5 MHz MHz MHz MHz OUT = 10 V step 615 460 460 625 Rev. B | Page 3 of 24 ns ns ns ns AD8251 Parameter Settling Time 0.001% G=1 G=2 G=4 G=8 Slew Rate G=1 G=2 G=4 G=8 Total Harmonic Distortion + Noise GAIN Gain Range Gain Error G=1 G = 2, 4, 8 Gain Nonlinearity G=1 G=2 G=4 G=8 Gain vs. Temperature INPUT Input Impedance Differential Common Mode Input Operating Voltage Range Over Temperature OUTPUT Output Swing Over Temperature Short-Circuit Current REFERENCE INPUT RIN IIN Voltage Range Gain to Output DIGITAL LOGIC Digital Ground Voltage, DGND Digital Input Voltage Low Digital Input Voltage High Digital Input Current Gain Switching Time 1 tSU tHD t WR -LOW t WR -HIGH Conditions OUT = 10 V step Min Typ Max Unit 785 700 700 770 ns ns ns ns 20 30 30 30 -110 f = 1 kHz, RL = 10 k, 10 V, G = 1, 10 Hz to 22 kHz bandpass filter G = 1, 2, 4, 8 OUT = 10 V V/s V/s V/s V/s dB 1 OUT = -10 V to +10 V RL = 10 k, 2 k, 600 RL = 10 k, 2 k, 600 RL = 10 k, 2 k, 600 RL = 10 k, 2 k, 600 All gains 3 8 V/V 0.03 0.04 % % 9 12 12 15 10 ppm ppm ppm ppm ppm/C G||pF G||pF V V 5.3||0.5 1.25||2 VS = 5 V to 15 V T = -40C to +85C -VS + 1.5 -VS + 1.6 +VS - 1.5 +VS - 1.7 T = -40C to +85C -13.5 -13.5 +13.5 +13.5 37 20 +IN, -IN, REF = 0 1 +VS -VS 1 0.0001 Referred to GND Referred to GND Referred to GND -VS + 4.25 DGND 2.8 0 +VS - 2.7 2.1 +VS 1 325 See Figure 3 timing diagram See Figure 3 timing diagram See Figure 3 timing diagram See Figure 3 timing diagram Rev. B | Page 4 of 24 20 10 20 40 V V mA k A V V/V V V V A ns ns ns ns ns AD8251 Parameter POWER SUPPLY Operating Range Quiescent Current, +IS Quiescent Current, -IS Over Temperature TEMPERATURE RANGE Specified Performance 1 Conditions Min Typ Max Unit 4.1 3.7 15 4.5 4.5 4.5 V mA mA mA +85 C 5 T = -40C to +85C -40 Add time for the output to slew and settle to calculate the total time for a gain change. TIMING DIAGRAM tWR-HIGH tWR-LOW WR tHD 06287-003 tSU A0, A1 Figure 3. Timing Diagram for Latched Gain Mode (See the Timing for Latched Gain Mode Section) Rev. B | Page 5 of 24 AD8251 ABSOLUTE MAXIMUM RATINGS 1 2 3 Rating 17 V See Figure 4 Indefinite1 +VS + 13 V to -VS - 13 V +VS + 13 V, -VS - 13 V2 VS -65C to +125C -40C to +85C 300C 140C 112C/W 140C Assumes the load is referenced to midsupply. Current must be kept to less than 6 mA. Temperature for specified performance is -40C to +85C. For performance to +125C, see the Typical Performance Characteristics section. Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. MAXIMUM POWER DISSIPATION The maximum safe power dissipation in the AD8251 package is limited by the associated rise in junction temperature (TJ) on the die. The plastic encapsulating the die locally reaches the junction temperature. At approximately 140C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit can change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8251. Exceeding a junction temperature of 140C for an extended period can result in changes in silicon devices, potentially causing failure. The still air thermal properties of the package and PCB (JA), the ambient temperature (TA), and the total power dissipated in the package (PD) determine the junction temperature of the die. The junction temperature is calculated as The difference between the total drive power and the load power is the drive power dissipated in the package. PD = Quiescent Power + (Total Drive Power - Load Power) V V PD = (VS x I S ) + S x OUT RL 2 VOUT 2 - R L In single-supply operation with RL referenced to -VS, the worst case is VOUT = VS/2. Airflow increases heat dissipation, effectively reducing JA. In addition, more metal directly in contact with the package leads from metal traces, through holes, ground, and power planes reduces the JA. Figure 4 shows the maximum safe power dissipation in the package vs. the ambient temperature on a four-layer JEDEC standard board. 2.00 1.75 1.50 1.25 1.00 0.75 0.50 0.25 0 -40 -20 0 20 40 60 80 100 120 AMBIENT TEMPERATURE (C) Figure 4. Maximum Power Dissipation vs. Ambient Temperature ESD CAUTION TJ = TA + (PD x JA ) Rev. B | Page 6 of 24 06287-004 Parameter Supply Voltage Power Dissipation Output Short-Circuit Current Common-Mode Input Voltage Differential Input Voltage Digital Logic Inputs Storage Temperature Range Operating Temperature Range3 Lead Temperature (Soldering, 10 sec) Junction Temperature JA (Four-Layer JEDEC Standard Board) Package Glass Transition Temperature The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the package due to the load drive for all outputs. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). Assuming the load (RL) is referenced to midsupply, the total drive power is VS/2 x IOUT, some of which is dissipated in the package and some in the load (VOUT x IOUT). MAXIMUM POWER DISSIPATION (W) Table 3. AD8251 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS -IN 1 DGND 2 10 +IN AD8251 9 REF 8 +VS TOP VIEW A0 4 (Not to Scale) 7 OUT A1 5 6 WR 06287-005 -VS 3 Figure 5. Pin Configuration Table 4. Pin Function Descriptions Pin No. 1 2 3 4 5 6 7 8 9 10 Mnemonic -IN DGND -VS A0 A1 WR OUT +VS REF +IN Description Inverting Input Terminal. True differential input. Digital Ground. Negative Supply Terminal. Gain Setting Pin (LSB). Gain Setting Pin (MSB). Write Enable. Output Terminal. Positive Supply Terminal. Reference Voltage Terminal. Noninverting Input Terminal. True differential input. Rev. B | Page 7 of 24 AD8251 TYPICAL PERFORMANCE CHARACTERISTICS TA = 25C, +VS = +15 V, -VS = -15 V, RL = 10 k, unless otherwise noted. 2700 800 2400 700 NUMBER OF UNITS 1800 1500 1200 900 600 500 400 300 200 600 100 06287-006 300 0 -120 -90 -60 -30 0 30 60 90 06287-009 NUMBER OF UNITS 2100 0 -30 120 -20 -10 0 10 20 30 INPUT OFFSET CURRENT (nA) CMRR (V/V) Figure 9. Typical Distribution of Input Offset Current Figure 6. Typical Distribution of CMRR, G = 1 90 500 80 NOISE (nV/Hz) 300 200 100 -200 -100 0 100 50 G=1 40 30 G=2 20 G=4 G=8 06287-007 0 60 06287-010 NUMBER OF UNITS 70 400 10 0 200 1 10 INPUT OFFSET VOLTAGE, VOSI , RTI (V) 100 1k 10k 100k FREQUENCY (Hz) Figure 10. Voltage Spectral Density Noise vs. Frequency Figure 7. Typical Distribution of Offset Voltage, VOSI 600 400 0 -30 -20 -10 0 10 20 2V/DIV 1s/DIV 30 INPUT BIAS CURRENT (nA) Figure 11. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 1 Figure 8. Typical Distribution of Input Bias Current Rev. B | Page 8 of 24 06287-011 200 06287-008 NUMBER OF UNITS 800 AD8251 150 130 G=4 PSRR (dB) 110 90 G=2 G=1 70 G=8 50 10 10 06287-016 06287-012 30 1s/DIV 1.25V/DIV 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 15. Positive PSRR vs. Frequency, RTI Figure 12. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 8 150 18 16 130 14 G=4 PSRR (dB) 10 8 G=8 90 70 G=1 6 50 4 G=2 2 0 1 10 100 1k 10k 10 10 100k 06287-017 06287-013 30 100 1k 10k 100k 1M FREQUENCY (Hz) FREQUENCY (Hz) Figure 13. Current Noise Spectral Density vs. Frequency Figure 16. Negative PSRR vs. Frequency, RTI 1s/DIV 9 8 7 6 5 4 3 2 1 0 0.01 0.1 1 WARM-UP TIME (minutes) Figure 17. Change in Offset Voltage, RTI vs. Warmup Time Figure 14. 0.1 Hz to 10 Hz Current Noise Rev. B | Page 9 of 24 10 06287-117 140pA/DIV CHANGE IN OFFSET VOLTAGE, RTI (V) 10 06287-014 NOISE (pA/Hz) 110 12 20 15 15 10 IB+ 5 CMRR (V/V) 10 5 IB- 0 -5 0 IOS -10 -60 -40 -20 0 20 40 60 80 100 120 -15 -50 140 06287-022 -10 -5 06287-019 INPUT BIAS CURRENT AND OFFSET CURRENT (nA) AD8251 -30 -10 10 TEMPERATURE (C) 30 50 70 90 130 110 TEMPERATURE (C) Figure 18. Input Bias Current and Offset Current vs. Temperature Figure 21. CMRR vs. Temperature, G = 1 140 25 G=4 VS = 15V VIN = 200mV p-p RL = 2k G=8 20 G=8 120 15 G=2 GAIN (dB) CMRR (dB) G=4 100 80 10 G=2 5 G=1 G=1 0 60 40 10 100 1k 10k 100k 06287-023 06287-020 -5 -10 1k 1M 10k 100k FREQUENCY (Hz) 1M 100M 10M FREQUENCY (Hz) Figure 19. CMRR vs. Frequency Figure 22. Gain vs. Frequency 140 40 GAIN NONLINEARITY (10ppm/DIV) 30 120 100 G=4 G=2 G=1 60 40 10 100 1k 10k 100k 10 0 -10 -20 -30 -40 -10 1M -8 -6 -4 -2 0 2 4 6 8 10 OUTPUT VOLTAGE (V) FREQUENCY (Hz) Figure 20. CMRR vs. Frequency, 1 k Source Imbalance 20 06287-024 80 06287-021 CMRR (dB) G=8 Figure 23. Gain Nonlinearity vs. Output Voltage, G = 1, RL = 10 k, 2 k, 600 Rev. B | Page 10 of 24 40 16 30 12 20 10 0 -10 -20 -8 -6 -4 -2 0 2 4 6 8 8 0V, +3.85V -4V, +2.2V 4 -4V, -2V -4 -8 -14.2V, -7.1V +14V, -7V 0V, -13.5V -12 -8 -4 30 12 COMMON-MODE VOLTAGE (V) GAIN NONLINEARITY (10ppm/DIV) 16 20 10 0 -10 -20 06287-026 -30 -4 -2 0 2 4 6 8 VS 15V -4V, +4V 0 -10 -20 06287-027 -30 2 +4V, +3.9V VS = 5V 0 -4 -4V, -3.9V 0V, -3.9V +4V, -4V -8 -13V, -13.1V -12 0V, -13.5V -8 -4 0 +13V, -13.5V 4 8 12 16 4 6 8 10 OUTPUT VOLTAGE (V) 35 IB+ IB- IOS 30 25 20 15 10 5 0 -15 06287-129 INPUT BIAS CURRENT AND OFFSET CURRENT (nA) GAIN NONLINEARITY (10ppm/DIV) 10 0 0V, +4V 4 Figure 28. Input Common-Mode Voltage Range vs. Output Voltage, G = 8 20 -2 +13V, +13V OUTPUT VOLTAGE (V) 30 -4 16 8 -16 -16 10 40 -6 12 -12 Figure 25. Gain Nonlinearity vs. Output Voltage, G = 4, RL = 10 k, 2 k, 600 -8 8 0V, +13.5V -13V, +13.5V OUTPUT VOLTAGE (V) -40 -10 4 Figure 27. Input Common-Mode Voltage Range vs. Output Voltage, G = 1 40 -6 0 OUTPUT VOLTAGE (V) Figure 24. Gain Nonlinearity vs. Output Voltage, G = 2, RL = 10 k, 2 k, 600 -8 +4V, -2V 0V, -3.9V OUTPUT VOLTAGE (V) -40 -10 +4V, +2V VS = 5V 0 -16 -16 10 +14V, +7V 0V, 15V 06287-029 -40 -10 -14.2V, +7.1V -12 06287-025 -30 0V, +13.5V 06287-028 COMMON-MODE VOLTAGE (V) GAIN NONLINEARITY (10ppm/DIV) AD8251 -10 -5 0 5 10 15 COMMON-MODE VOLTAGE (V) Figure 26. Gain Nonlinearity vs. Output Voltage, G = 8, RL = 10 k, 2 k, 600 Figure 29. Input Bias Current and Offset Current vs. Common-Mode Voltage Rev. B | Page 11 of 24 AD8251 +VS +125C -0.2 +25C OUTPUT VOLTAGE SWING (V) REFERRED TO SUPPLY VOLTAGES -1 -40C -2 +2 -40C +25C +1 +125C 4 6 +85C 8 10 12 -0.6 -0.8 -1.0 -40C -40C 1.0 0.8 0.6 -VS 4 6 8 10 12 14 16 Figure 33. Output Voltage Swing vs. Supply Voltage, G = 8, RL = 10 k 15 15 +25C +VS 10 +85C FAULT CONDITION (OVER DRIVEN INPUT) G=8 +IN 0 -IN -5 -10 -15 -16 -12 -8 -4 0 4 8 12 5 0 -5 +125C -40C -10 -VS -40C +125C +85C 06287-034 FAULT CONDITION (OVER DRIVEN INPUT) G=8 5 OUTPUT VOLTAGE SWING (V) 10 06287-031 CURRENT (mA) +85C SUPPLY VOLTAGE (VS) Figure 30. Input Voltage Limit vs. Supply Voltage, G = 1, VREF = 0 V, RL = 10 k +25C -15 100 16 10k 1k DIFFERENTIAL INPUT VOLTAGE (V) LOAD RESISTANCE () Figure 34. Output Voltage Swing vs. Load Resistance Figure 31. Fault Current Draw vs. Input Voltage, G = 8, RL = 10 k +VS +VS -0.2 +85C -0.4 +85C -0.6 OUTPUT VOLTAGE SWING (V) REFERRED TO SUPPLY VOLTAGES -0.4 +125C -0.8 -1.0 -40C +25C +25C -40C 1.0 0.8 0.6 +85C 0.4 +125C 06287-032 OUTPUT VOLTAGE SWING (V) REFERRED TO SUPPLY VOLTAGES +25C +125C 0.4 SUPPLY VOLTAGE (VS) 0.2 -VS +25C 0.2 16 14 +125C +85C -0.4 4 6 8 10 12 14 16 SUPPLY VOLTAGE (VS) +125C -0.8 -1.2 +25C -40C -1.6 -2.0 2.0 -40C 1.6 +25C 1.2 0.8 +125C 0.4 -VS 06287-035 -VS 06287-030 INPUT VOLTAGE (V) REFERRED TO SUPPLY VOLTAGES +85C 06287-033 +VS +85C 4 6 8 10 12 14 OUTPUT CURRENT (mA) Figure 32. Output Voltage Swing vs. Supply Voltage, G = 8, RL = 2 k Rev. B | Page 12 of 24 Figure 35. Output Voltage Swing vs. Output Current 16 AD8251 NO LOAD 47pF 100pF 5V/DIV 376ns TO 0.01% 640ns TO 0.001% 2s/DIV 2s/DIV 06287-036 20mV/DIV 06287-039 0.002%/DIV Figure 36. Small Signal Pulse Response for Various Capacitive Loads Figure 39. Large Signal Pulse Response and Settling Time, G = 4, RL = 10 k 5V/DIV 5V/DIV 364ns TO 0.01% 522ns TO 0.001% 585ns TO 0.01% 723ns TO 0.001% 06287-037 2s/DIV 06287-040 0.002%/DIV 0.002%/DIV 2s/DIV Figure 40. Large Signal Pulse Response and Settling Time, G = 8, RL = 10 k Figure 37. Large Signal Pulse Response and Settling Time, G = 1, RL = 10 k 5V/DIV 400ns TO 0.01% 600ns TO 0.001% 25mV/DIV 2s/DIV Figure 41. Small Signal Response, G = 1, RL = 2 k, CL = 100 pF Figure 38. Large Signal Pulse Response and Settling Time, G = 2, RL = 10 k Rev. B | Page 13 of 24 06287-041 2s/DIV 06287-038 0.002%/DIV AD8251 1200 1000 SETTLED TO 0.001% TIME (ns) 800 600 SETTLED TO 0.01% 400 06287-045 2s/DIV 06287-042 25mV/DIV 200 0 2 4 6 8 10 12 14 16 18 20 STEP SIZE (V) Figure 42. Small Signal Response, G = 2, RL = 2 k, CL = 100 pF Figure 45. Settling Time vs. Step Size, G = 1, RL = 10 k 1200 1000 TIME (ns) 800 SETTLED TO 0.001% 600 400 200 06287-046 2s/DIV 06287-043 25mV/DIV SETTLED TO 0.01% 0 2 4 6 8 10 12 14 16 18 20 STEP SIZE (V) Figure 46. Settling Time vs. Step Size, G = 2, RL = 10 k Figure 43. Small Signal Response, G = 4, RL = 2 k, CL = 100 pF 1200 1000 TIME (ns) 800 SETTLED TO 0.001% 600 400 06287-047 2s/DIV 200 06287-044 25mV/DIV SETTLED TO 0.01% 0 2 4 6 8 10 12 14 16 18 STEP SIZE (V) Figure 44. Small Signal Response, G = 8, RL = 2 k, CL = 100 pF Figure 47. Settling Time vs. Step Size, G = 4, RL = 10 k Rev. B | Page 14 of 24 20 AD8251 -50 1200 -55 -60 1000 -65 -70 THD + N (dB) TIME (ns) 800 SETTLED TO 0.001% 600 400 -75 -80 G=8 G=4 -85 G=2 -90 -95 -100 SETTLED TO 0.01% -105 06287-048 -110 0 2 4 6 8 10 12 14 16 18 -115 -120 10 20 Figure 48. Settling Time vs. Step Size, G = 8, RL = 10 k -55 -60 -65 -75 -80 G=8 G=4 -100 -105 -110 G=2 -115 -120 10 06287-049 THD + N (dB) -70 -95 G=1 100 1k 10k 1k 10k 100k Figure 50. Total Harmonic Distortion + Noise vs. Frequency, 10 Hz to 500 kHz Band-Pass Filter, RL = 2 k -50 -90 100 FREQUENCY (Hz) STEP SIZE (V) -85 G=1 06287-050 200 100k 1M FREQUENCY (Hz) Figure 49. Total Harmonic Distortion + Noise vs. Frequency, 10 Hz to 22 kHz Band-Pass Filter, RL = 2 k Rev. B | Page 15 of 24 1M AD8251 THEORY OF OPERATION +VS +VS A0 A1 2.2k +VS -VS -VS 2.2k -IN 10k A1 10k -VS +VS DIGITAL GAIN CONTROL OUT A3 -VS +VS +VS 10k 10k A2 REF +IN 2.2k +VS -VS -VS +VS 2.2k -VS 06287-061 DGND WR -VS Figure 51. Simplified Schematic Transparent Gain Mode The easiest way to set the gain is to program it directly via a logic high or logic low voltage applied to A0 and A1. Figure 52 shows an example of this gain setting method, referred to throughout the data sheet as transparent gain mode. Tie WR to the negative supply to engage transparent gain mode. In this mode, any change in voltage applied to A0 and A1 from logic low to logic high, or vice versa, immediately results in a gain change. Table 5 is the truth table for transparent gain mode, and Figure 52 shows the AD8251 configured in transparent gain mode. All internal amplifiers employ distortion cancellation circuitry and achieve high linearity and ultralow THD. Laser trimmed resistors allow for a maximum gain error of less than 0.03% for G = 1 and minimum CMRR of 98 dB for G = 8. A pinout optimized for high CMRR over frequency enables the AD8251 to offer a guaranteed minimum CMRR over frequency of 80 dB at 50 kHz (G = 1). The balanced input reduces the parasitics that, in the past, adversely affected CMRR performance. GAIN SELECTION +15V 10F 0.1F WR A1 A0 +IN +5V G=8 AD8251 REF -IN DGND 10F Logic low and logic high voltage limits are listed in the Specifications section. Typically, logic low is 0 V and logic high is 5 V; both voltages are measured with respect to DGND. See Table 2 for the permissible voltage range of DGND. The gain of the AD8251 can be set using two methods. -15V +5V DGND 0.1F -15V NOTE: 1. IN TRANSPARENT GAIN MODE, WR IS TIED TO -VS. THE VOLTAGE LEVELS ON A0 AND A1 DETERMINE THE GAIN. IN THIS EXAMPLE, BOTH A0 AND A1 ARE SET TO LOGIC HIGH, RESULTING IN A GAIN OF 8. 06287-051 The AD8251 is a monolithic instrumentation amplifier based on the classic 3-op-amp topology, as shown in Figure 51. It is fabricated on the Analog Devices, Inc., proprietary iCMOS(R) process that provides precision, linear performance, and a robust digital interface. A parallel interface allows users to digitally program gains of 1, 2, 4, and 8. Gain control is achieved by switching resistors in an internal, precision resistor array (as shown in Figure 51). Although the AD8251 has a voltage feedback topology, the gain bandwidth product increases for gains of 1, 2, and 4 because each gain has its own frequency compensation. This results in maximum bandwidth at higher gains. Figure 52. Transparent Gain Mode, A0 and A1 = High, G = 8 Rev. B | Page 16 of 24 AD8251 Table 5. Truth Table Logic Levels for Transparent Gain Mode Table 6. Truth Table Logic Levels for Latched Gain Mode WR A1 A0 Gain WR A1 A0 Gain -VS -VS -VS -VS Low Low High High Low High Low High 1 2 4 8 High to low High to low High to low High to low Low to low Low to high High to high Low Low High High X1 X1 X1 Low High Low High X1 X1 X1 Change to 1 Change to 2 Change to 4 Change to 8 No change No change No change Latched Gain Mode Some applications have multiple programmable devices such as multiplexers or other programmable gain instrumentation amplifiers on the same PCB. In such cases, devices can share a data bus. The gain of the AD8251 can be set using WR as a latch, allowing other devices to share A0 and A1. Figure 53 shows a schematic using this method, known as latched gain mode. The AD8251 is in this mode when WR is held at logic high or logic low, typically 5 V and 0 V, respectively. The voltages on A0 and A1 are read on the downward edge of the WR signal as it transitions from logic high to logic low. This latches in the logic levels on A0 and A1, resulting in a gain change. See the truth table in Table 6 for more information on these gain changes. +15V WR 10F 0.1F A1 A1 A0 +IN +5V 0V +5V 0V WR + A0 G = PREVIOUS STATE +5V 0V G=8 AD8251 REF - -IN DGND 0.1F -15V NOTE: 1. ON THE DOWNWARD EDGE OF WR, AS IT TRANSITIONS FROM LOGIC HIGH TO LOGIC LOW, THE VOLTAGES ON A0 AND A1 ARE READ AND LATCHED IN, RESULTING IN A GAIN CHANGE. IN THIS EXAMPLE, THE GAIN SWITCHES TO G = 8. X = don't care. On power-up, the AD8251 defaults to a gain of 1 when in latched gain mode. In contrast, if the AD8251 is configured in transparent gain mode, it starts at the gain indicated by the voltage levels on A0 and A1 at power-up. Timing for Latched Gain Mode In latched gain mode, logic levels at A0 and A1 must be held for a minimum setup time, tSU, before the downward edge of WR latches in the gain. Similarly, they must be held for a minimum hold time of tHD after the downward edge of WR to ensure that the gain is latched in correctly. After tHD, A0 and A1 can change logic levels, but the gain does not change (until the next downward edge of WR). The minimum duration that WR can be held high is t WR -HIGH, and the minimum duration that WR can be held low is t WR -LOW. Digital timing specifications are listed in Table 2. The time required for a gain change is dominated by the settling time of the amplifier. A timing diagram is shown in Figure 54. When sharing a data bus with other devices, logic levels applied to those devices can potentially feed through to the output of the AD8251. Feedthrough can be minimized by decreasing the edge rate of the logic signals. Furthermore, careful layout of the PCB also reduces coupling between the digital and analog portions of the board. Pull-up or pull-down resistors should be used to provide a well-defined voltage at the A0 and A1 pins. DGND 06287-052 10F 1 Figure 53. Latched Gain Mode, G = 8 tWR-HIGH tWR-LOW WR tHD 06287-053 tSU A0, A1 Figure 54. Timing Diagram for Latched Gain Mode Rev. B | Page 17 of 24 AD8251 INCORRECT POWER SUPPLY REGULATION AND BYPASSING The AD8251 has high PSRR. However, for optimal performance, a stable dc voltage should be used to power the instrumentation amplifier. Noise on the supply pins can adversely affect performance. As in all linear circuits, bypass capacitors must be used to decouple the amplifier. +VS AD8251 AD8251 REF Place a 0.1 F capacitor close to each supply pin. A 10 F tantalum capacitor can be used farther away from the part (see Figure 55) and, in most cases, it can be shared by other precision integrated circuits. REF -VS -VS TRANSFORMER TRANSFORMER +VS +VS 0.1F WR A1 +IN CORRECT +VS +VS 10F AD8251 A0 AD8251 REF OUT AD8251 10M LOAD -VS REF -IN REF -VS THERMOCOUPLE THERMOCOUPLE DGND +VS -VS +VS C Figure 55. Supply Decoupling, REF, and Output Referred to Ground C 1 fHIGH-PASS = 2RC AD8251 C REF INPUT BIAS CURRENT RETURN PATH The AD8251 input bias current must have a return path to its local analog ground. When the source, such as a thermocouple, cannot provide a return current path, one should be created (see Figure 56). R AD8251 C REF R -VS -VS CAPACITIVELY COUPLED CAPACITIVELY COUPLED 06287-055 DGND 10F 06287-054 0.1F Figure 56. Creating an IBIAS Return Path INPUT PROTECTION All terminals of the AD8251 are protected against ESD. Note that 2.2 k series resistors precede the ESD diodes as shown in Figure 51. The resistors limit current into the diodes and allow for dc overload conditions 13 V above the positive supply and 13 V below the negative supply. An external resistor should be used in series with each input to limit current for voltages greater than 13 V beyond either supply rail. In either scenario, the AD8251 safely handles a continuous 6 mA current at room temperature. For applications where the AD8251 encounters extreme overload voltages, external series resistors and low leakage diode clamps, such as BAV199Ls, FJH1100s, or SP720s, should be used. Rev. B | Page 18 of 24 AD8251 REFERENCE TERMINAL The reference terminal, REF, is at one end of a 10 k resistor (see Figure 51). The instrumentation amplifier output is referenced to the voltage on the REF terminal; this is useful when the output signal needs to be offset to voltages other than its local analog ground. For example, a voltage source can be tied to the REF pin to level shift the output so that the AD8251 can interface with a single-supply ADC. The allowable reference voltage range is a function of the gain, common-mode input, and supply voltages. The REF pin should not exceed either +VS or -VS by more than 0.5 V. For best performance, especially in cases where the output is not measured with respect to the REF terminal, source impedance to the REF terminal should be kept low because parasitic resistance can adversely affect CMRR and gain accuracy. INCORRECT CORRECT AD8251 The output voltage of the AD8251 develops with respect to the potential on the reference terminal. Take care to tie REF to the appropriate local analog ground or to connect it to a voltage that is referenced to the local analog ground. Coupling Noise To prevent coupling noise onto the AD8251, follow these guidelines: * Do not run digital lines under the device. * Run the analog ground plane under the AD8251. * Shield fast switching signals with digital ground to avoid radiating noise to other sections of the board, and never run them near analog signal paths. * Avoid crossover of digital and analog signals. * Connect digital and analog ground at one point only (typically under the ADC). * Use large traces on the power supply lines to ensure a low impedance path. Decoupling is necessary; follow the guidelines listed in the Power Supply Regulation and Bypassing section. AD8251 VREF VREF + Common-Mode Rejection - 06287-056 OP1177 Figure 57. Driving the Reference Pin COMMON-MODE INPUT VOLTAGE RANGE The 3-op-amp architecture of the AD8251 applies gain and then removes the common-mode voltage. Therefore, internal nodes in the AD8251 experience a combination of both the gained signal and the common-mode signal. This combined signal can be limited by the voltage supplies even when the individual input and output signals are not. Figure 27 and Figure 28 show the allowable common-mode input voltage ranges for various output voltages, supply voltages, and gains. LAYOUT Grounding In mixed-signal circuits, low level analog signals need to be isolated from the noisy digital environment. Designing with the AD8251 is no exception. Its supply voltages are referenced to an analog ground. Its digital circuit is referenced to a digital ground. Although it is convenient to tie both grounds to a single ground plane, the current traveling through the ground wires and PCB can cause errors. Therefore, use separate analog and digital ground planes. Analog and digital ground should meet at one point only: star ground. The AD8251 has high CMRR over frequency, giving it greater immunity to disturbances, such as line noise and its associated harmonics, in contrast to typical instrumentation amplifiers whose CMRR falls off around 200 Hz. The typical instrumentation amplifiers often need common-mode filters at their inputs to compensate for this shortcoming. The AD8251 is able to reject CMRR over a greater frequency range, reducing the need for input common-mode filtering. Careful board layout maximizes system performance. To maintain high CMRR over frequency, lay out the input traces symmetrically. Ensure that the traces maintain resistive and capacitive balance; this holds for additional PCB metal layers under the input pins and traces. Source resistance and capacitance should be placed as close to the inputs as possible. Should a trace cross the inputs (from another layer), it should be routed perpendicular to the input traces. Rev. B | Page 19 of 24 AD8251 RF INTERFERENCE DRIVING AN ADC RF rectification is often a problem when amplifiers are used in applications where there are strong RF signals. The disturbance can appear as a small dc offset voltage. High frequency signals can be filtered with a low-pass RC network placed at the input of the instrumentation amplifier, as shown in Figure 58. The filter limits the input signal bandwidth according to the following relationship: An instrumentation amplifier is often used in front of an ADC to provide CMRR. Usually, instrumentation amplifiers require a buffer to drive an ADC. However, the low output noise, low distortion, and low settle time of the AD8251 make it an excellent ADC driver. 1 FilterFreq DIFF = FilterFreq CM = 2 R( 2C D + C C ) 1 2 RC C where CD 10 CC. +15V 0.1F 10F In Figure 59, a 1 nF capacitor and a 49.9 resistor create an antialiasing filter for the AD7612. The 1 nF capacitor stores and delivers the necessary charge to the switched capacitor input of the ADC. The 49.9 series resistor reduces the burden of the 1 nF load from the amplifier and isolates it from the kickback current injected from the switched capacitor input of the AD7612. Selecting too small a resistor improves the correlation between the voltage at the output of the AD8251 and the voltage at the input of the AD7612 but may destabilize the AD8251. A tradeoff must be made between selecting a resistor small enough to maintain accuracy and large enough to maintain stability. +15V CC R +IN 10F 0.1F WR VOUT AD8251 CD R +12V A1 A0 +IN -12V 0.1F 0.1F REF -IN 49.9 AD8251 CC REF 10F 06287-057 -15V +5V -IN ADR435 DGND 10F Figure 58. RFI Suppression Values of R and CC should be chosen to minimize RFI. A mismatch between the R x CC at the positive input and the R x CC at negative input degrades the CMRR of the AD8251. By using a value of CD that is 10 times larger than the value of CC, the effect of the mismatch is reduced and performance is improved. Rev. B | Page 20 of 24 DGND 0.1F 06287-058 0.1F AD7612 1nF -15V Figure 59. Driving an ADC AD8251 APPLICATIONS DIFFERENTIAL OUTPUT SETTING GAINS WITH A MICROCONTROLLER +15V In certain applications, it is necessary to create a differential signal. High resolution ADCs often require a differential input. In other cases, transmission over a long distance can require differential signals for better immunity to interference. 10F 0.1F WR A1 A0 +IN Figure 61 shows how to configure the AD8251 to output a differential signal. An op amp, the AD817, is used in an inverting topology to create a differential voltage. VREF sets the output midpoint according to the equation shown in the figure. Errors from the op amp are common to both outputs and are thus common mode. Likewise, errors from using mismatched resistors cause a common-mode dc offset error. Such errors are rejected in differential signal processing by differential input ADCs or instrumentation amplifiers. MICROCONTROLLER + AD8251 REF - -IN DGND DGND 0.1F 06287-059 10F -15V Figure 60. Programming Gain Using a Microcontroller When using this circuit to drive a differential ADC, VREF can be set using a resistor divider from the ADC reference to make the output ratiometric with the ADC. +12V 0.1F AMPLITUDE WR +5V A1 A0 +IN -5V AMPLITUDE + VOUTA = VIN + VREF 2 AD8251 VIN G=1 - 0.1F +2.5V 0V -2.5V REF TIME 4.99k DGND - -12V -12V 4.99k + AD817 +12V VREF 0V 10pF AMPLITUDE 10F 0.1F -12V 0.1F 10F DGND VOUTB = -VIN + VREF 2 Figure 61. Differential Output with Level Shift Rev. B | Page 21 of 24 +2.5V 0V -2.5V TIME 06287-060 +12V AD8251 -70 DATA ACQUISITION -80 The AD8251 makes an excellent instrumentation amplifier for use in data acquisition systems. Its wide bandwidth, low distortion, low settling time, and low noise enable it to condition signals in front of a variety of 16-bit ADCs. -90 AMPLITUDE (dB) -100 Figure 63 shows a schematic of the AD825x data acquisition demonstration board. The quick slew rate of the AD8251 allows it to condition rapidly changing signals from the multiplexed inputs. An FPGA controls the AD7612, AD8251, and ADG1209. In addition, mechanical switches and jumpers allow users to pin strap the gains when in transparent gain mode. -110 -120 -130 -140 -150 06287-062 -160 -170 -180 This system achieved -106 dB of THD at 1 kHz and a signal-tonoise ratio of 91 dB during testing, as shown in Figure 62. 0 5 10 15 20 25 30 35 40 45 FREQUENCY (kHz) Figure 62. FFT of the AD825x DAQ Demo Board Using the AD8251 1 kHz Signal JMP 0.1F +CH2 +CH3 +CH4 -CH4 -CH3 -CH2 -CH1 806 806 806 DGND EN 4 S1A DGND JMP +5V DGND 5 S2A 2k 2 6 S3A ALTERA EPF6010ATC144-3 DGND 6 0 7 S4A 0 CC +IN DA 8 10 + 0 0 CD -IN DB 9 11 S3B CC GND 15 A0 12 S2B 806 S1B A1 VSS 16 1 5 WR A1 4 A0 AD8251 REF ADG1209 13 -VS 2k 10F 2 10 S4B 806 +5V GND VDD 806 806 -12V - +VS -VS 9 3 DGND 7 OUT +IN 0 49.9 AD7612 1nF ADR435 8 1 C4 0.1F C3 0.1F 3 +12V -12V JMP 0.1F -12V +5V 2k DGND JMP +5V R8 2k 06287-067 +CH1 + 10F 14 806 JMP +12V + +12V DGND Figure 63. Schematic of ADG1209, AD8251, and AD7612 in the AD825x DAQ Demo Board Rev. B | Page 22 of 24 50 AD8251 OUTLINE DIMENSIONS 3.10 3.00 2.90 10 3.10 3.00 2.90 1 5.15 4.90 4.65 6 5 PIN 1 IDENTIFIER 0.50 BSC 0.95 0.85 0.75 15 MAX 1.10 MAX 0.30 0.15 6 0 0.23 0.13 0.70 0.55 0.40 COMPLIANT TO JEDEC STANDARDS MO-187-BA 091709-A 0.15 0.05 COPLANARITY 0.10 Figure 64. 10-Lead Mini Small Outline Package [MSOP] (RM-10) Dimensions shown in millimeters ORDERING GUIDE Model1 AD8251ARMZ AD8251ARMZ-RL AD8251ARMZ-R7 AD8251-EVALZ 1 Temperature Range -40C to +85C -40C to +85C -40C to +85C Package Description 10-Lead Mini Small Outline Package [MSOP] 10-Lead Mini Small Outline Package [MSOP] 10-Lead Mini Small Outline Package [MSOP] Evaluation Board Z = RoHS Compliant Part. Rev. B | Page 23 of 24 Package Option RM-10 RM-10 RM-10 Branding H0T H0T H0T AD8251 NOTES (c)2007-2010 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D06287-0-11/10(B) Rev. B | Page 24 of 24