CMOS 300 MSPS Quadrature
Complete DDS
AD9854
Rev. C
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However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
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FEATURES
300 MHz internal clock rate
FSK, BPSK, PSK, CHIRP, AM operation
Dual integrated 12-bit D/A converters
Ultrahigh speed comparator, 3 ps rms jitter
Excellent dynamic performance:
80 dB SFDR @ 100 MHz (±1 MHz) AOUT
4× to 20× programmable reference clock multiplier
Dual 48-bit programmable frequency registers
Dual 14-bit programmable phase offset registers
12-bit amplitude modulation and programmable
shaped on/off keying function
Single-pin FSK and BPSK data interface
PSK capability via I/O interface
Linear or nonlinear FM chirp functions with single-pin
frequency hold function
Frequency-ramped FSK
< 25 ps rms total jitter in clock generator mode
Automatic bidirectional frequency sweeping
SIN(x)/x correction
Simplified control interfaces:
10 MHz serial, 2-wire or 3-wire SPI®-compatible
100 MHz parallel 8-bit programming
3.3 V single supply
Multiple power-down functions
Single-ended or differential input reference clock
Small 80-lead LQFP packaging
APPLICATIONS
Agile, quadrature LO frequency synthesis
Programmable clock generators
FM chirp source for radar and scanning systems
Test and measurement equipment
Commercial and amateur RF exciters
FUNCTIONAL BLOCK DIAGRAM
DIGITAL MULTIPLIERS
SYSTEM
CLOCK
DAC RSET
INV.
SINC
FILTER
FREQUENCY
ACCUMULATOR
ACC 1
I/O PORT BUFFERS
COMPARATOR
PROGRAMMING REGISTERS
4×– 20×
REF CLK
MULTIPLIER
DIFF/SINGLE
SELECT
REFERENCE
CLOCK IN
FSK/BPSK/HOLD
DATA IN
BIDIRECTIONAL
INTERNAL/EXTERNA
L
I/O UPDATE CLOCK
READ WRITE SERIAL/
PARALLEL
SELECT
6-BIT ADDRESS
OR SERIAL
PROGRAMMING
LINES
8-BIT
PARALLEL
LOAD
MASTER
RESET
+VS
GND
CLOCK
OUT
ANALOG
IN
SHAPED
ON/OFF
KEYING
ANALOG
OUT
ANALOG
OUT
INTERNAL
PROGRAMMABLE
UPDATE CLOCK
PHASE-TO-
AMPLITUDE
CONVERTER
PROGRAMMABLE
AMPLITUDE AND
RATE CONTROL
DQ
CK ÷2
INT
EXT
SYSTEM
CLOCK
REF
CLK
BUFFER
SYSTEM
CLOCK
MUX
DELTA
FREQUENCY
RATE TIMER
SYSTEM
CLOCK
DELTA
FREQUENCY
WORD
FREQUENCY
TUNING
WORD 1
FREQUENCY
TUNING
WORD 2
FIRST 14-BIT
PHASE/OFFSET
WORD
SECOND 14-BIT
PHASE/OFFSET
WORD
12-BIT DC
CONTROL
MUX
SYSTEM CLOCK
PHASE
ACCUMULATOR
ACC 2
DDS CORE 12-BIT
"I"
DAC
12-BIT
"Q" DAC OR
CONTROL
DAC
I
Q
12
MUX MUX
MUX
MUX
SYSTEM
CLOCK
SYSTEM
CLOCK
48 48 48 14 14
BUS
12
12
14
17
17
4848
48
AD9854
MODE SELECT
2
3
DEMUX
00636-B-001
MUX
MUX
12
INV.
SINC
FILTER
12
12
12
12
"I" AND "Q" 12-BIT
AM MODULATION
Figure 1.
AD9854
Rev. C | Page 2 of 52
TABLE OF CONTENTS
General Description..........................................................................3
Specifications......................................................................................4
Absolute Maximum Ratings.............................................................8
Explanation of Test Levels........................................................... 8
Pin Configuration and Function Descriptions..............................9
Typical Performance Characteristics ........................................... 12
Typical Applications ....................................................................... 16
Theory of Operation ...................................................................... 19
Modes of Operation ................................................................... 19
Using the AD9854 .......................................................................... 28
Internal and External Update Clock........................................ 28
Shaped On/Off Keying .............................................................. 28
I and Q DACs.............................................................................. 29
Control DAC ............................................................................... 29
Inverse SINC Function .............................................................. 30
REFCLK Multiplier .................................................................... 30
Programming the AD9854............................................................ 32
Parallel I/O Operation ............................................................... 33
Serial Port I/O Operation.......................................................... 33
General Operation of the Serial Interface ....................................35
Instruction Byte .......................................................................... 36
Serial Interface Port Pin Descriptions ..................................... 36
Notes on Serial Port Operation ................................................ 36
MSB/LSB Transfers..........................................................................37
Control Register Description.................................................... 37
Power Dissipation and Thermal Considerations ........................39
Thermal Impedance................................................................... 39
Junction Temperature Considerations .................................... 39
Evaluation of Operating Conditions........................................ 40
Thermally Enhanced Package Mounting Guidelines............ 41
Evaluation Board .............................................................................42
Evaluation Board Instructions.................................................. 42
General Operating Instructions ............................................... 42
Using the Provided Software .................................................... 44
Support ........................................................................................ 44
Outline Dimensions ........................................................................51
Ordering Guide .......................................................................... 51
REVISION HISTORY
9/04—Data Sheet Changed from Rev. B to Rev. C
Updated Format..................................................................Universal
Changes to Table 1............................................................................ 4
Changes to Footnote 2 ..................................................................... 7
Changes to Explanation of Test Levels Section ............................ 8
Changes to Theory of Operation Section.................................... 17
Changes to Single Tone (Mode 000) Section .............................. 17
Changes to Ramped FSK (Mode 010) Section ........................... 18
Changes to Basic FM Chirp Programming Steps Section......... 23
Changes to Figure 50...................................................................... 27
Changes to Evaluation Board Operating Instructions Section 40
Changes to Filtered IOUT1 and the Filtered IOUT2 Section... 41
Changes to Using the Provided Software Section ...................... 42
Changes to Figure 68...................................................................... 45
Changes to Figure 69...................................................................... 46
Updated Outline Dimensions ....................................................... 50
Changes to Ordering Guide .......................................................... 50
3/02—Data Sheet Changed from Rev. A to Rev. B
Updated Format..................................................................Universal
Renumbered Figures and Tables ......................................Universal
Changes to General Description Section .......................................1
Changes to Functional Block Diagram...........................................1
Changes to Specifications Section...................................................4
Changes to Absolute Maximum Ratings Section..........................7
Changes to Pin Function Descriptions...........................................8
Changes to Figure 3........................................................................ 10
Deleted two Typical Performance Characteristics Graphs ....... 11
Changes to Inverse SINC Function Section................................ 28
Changes to Differential REFCLK Enable Section ...................... 28
Changes to Figure 52...................................................................... 30
Changes to Parallel I/O Operation Section................................. 32
Changes to General Operation of the Serial Interface Section 33
Changes to Figure 57...................................................................... 34
Replaced Operating Instructions Section ................................... 40
Changes to Figure 68...................................................................... 44
Changes to Figure 69...................................................................... 45
Changes to Customer Evaluation Board Table........................... 46
AD9854
Rev. C | Page 3 of 52
GENERAL DESCRIPTION
The AD9854 digital synthesizer is a highly integrated device
that uses advanced DDS technology, coupled with two internal
high speed, high performance quadrature D/A converters to
form a digitally programmable I and Q synthesizer function.
When referenced to an accurate clock source, the AD9854
generates highly stable, frequency-phase amplitude-program-
mable sine and cosine outputs that can be used as an agile LO
in communications, radar, and many other applications. The
AD9854’s innovative high speed DDS core provides 48-bit
frequency resolution (1 MHz tuning resolution with 300 MHz
SYSCLK). Maintaining 17 bits assures excellent SFDR. The
AD9854’s circuit architecture allows the generation of simul-
taneous quadrature output signals at frequencies up to
150 MHz, which can be digitally tuned at a rate of up to
100 million new frequencies per second. The sine wave output
(externally filtered) can be converted to a square wave by the
internal comparator for agile clock generator applications.
The device provides two 14-bit phase registers and a single pin
for BPSK operation. For higher-order PSK operation, the I/O
interface may be used for phase changes. The 12-bit I and
Q DACs, coupled with the innovative DDS architecture, provide
excellent wideband and narrow-band output SFDR. The Q DAC
can also be configured as a user-programmable control DAC if
the quadrature function is not desired. When configured with
the comparator, the 12-bit control DAC facilitates static duty
cycle control in high speed clock generator applications. Two
12-bit digital multipliers permit programmable amplitude
modulation, shaped on/off keying, and precise amplitude
control of the quadrature output. Chirp functionality is also
included to facilitate wide bandwidth frequency sweeping
applications. The AD9854’s programmable 4× to 20× REFCLK
multiplier circuit generates the 300 MHz system clock internally
from a lower frequency external reference clock. This saves the
user the expense and difficulty of implementing a 300 MHz
system clock source. Direct 300 MHz clocking is also accom-
modated with either single-ended or differential inputs. Single-
pin conventional FSK and the enhanced spectral qualities of
ramped FSK are supported.The AD9854 uses advanced 0.35
micron CMOS technology to provide a high level of
functionality on a single 3.3 V supply.
The AD9854 is available in a space-saving 80-lead LQFP
surface-mount package and a thermally enhanced 80-lead
LQFP package. The AD9854 is pin-for-pin compatible with the
AD9852 single-tone synthesizer. It is specified to operate over
the extended industrial temperature range of −40°C to +85°C.
AD9854
Rev. C | Page 4 of 52
SPECIFICATIONS
VS = 3.3 V ± 5%, RSET = 3.9 kΩ, external reference clock frequency = 30 MHz with REFCLK multiplier enabled at 10× for AD9854ASQ,
external reference clock frequency = 20 MHz with REFCLK multiplier enabled at 10× for AD9854AST, unless otherwise noted.
Table 1.
AD9854ASQ AD9854AST
Parameter Temp
Tes t
Level Min Typ Max Min Typ Max Unit
REF CLOCK INPUT CHARACTERISTICS1
Internal System Clock Frequency Range
REFCLK Multiplier Enabled Full VI 20 300 20 200 MHz
REFCLK Multiplier Disabled Full VI DC 300 DC 200 MHz
External REF Clock Frequency Range
REFCLK Multiplier Enabled Full VI 5 75 5 50 MHz
REFCLK Multiplier Disabled Full VI DC 300 DC 200 MHz
Duty Cycle 25°C IV 45 50 55 45 50 55 %
Input Capacitance 25°C IV 3 3 pF
Input Impedance 25°C IV 100 100 kΩ
Differential Mode Common-Mode
Voltage Range
Minimum Signal Amplitude225°C IV 400 400 mV p-p
Common-Mode Range 25°C IV 1.6 1.75 1.9 1.6 1.75 1.9 V
VIH (Single-Ended Mode) 25°C IV 2.3 2.3 V
VIL (Single-Ended Mode) 25°C IV 1 1 V
DAC STATIC OUTPUT CHARACTERISTICS
Output Update Speed Full I 300 200 MSPS
Resolution 25°C IV 12 12 Bits
I and Q Full-Scale Output Current 25°C IV 5 10 20 5 10 20 mA
I and Q DAC DC Gain Imbalance325°C I −0.5 +0.15 +0.5 −0.5 +0.15 +0.5 dB
Gain Error 25°C I −6 +2.25 −6 +2.25 % FS
Output Offset 25°C I 2 2 µA
Differential Nonlinearity 25°C I 0.3 1.25 0.3 1.25 LSB
Integral Nonlinearity 25°C I 0.6 1.66 0.6 1.66 LSB
Output Impedance 25°C IV 100 100 kΩ
Voltage Compliance Range 25°C I −0.5 +1.0 −0.5 +1.0 V
DAC DYNAMIC OUTPUT CHARACTERISTICS
I and Q DAC Quad. Phase Error 25°C IV 0.2 1 0.2 1 Degrees
DAC Wideband SFDR
1 MHz to 20 MHz AOUT 25°C V 58 58 dBc
20 MHz to 40 MHz AOUT 25°C V 56 56 dBc
40 MHz to 60 MHz AOUT 25°C V 52 52 dBc
60 MHz to 80 MHz AOUT 25°C V 48 48 dBc
80 MHz to 100 MHz AOUT 25°C V 48 48 dBc
100 MHz to 120 MHz AOUT 25°C V 48 48 dBc
DAC Narrow-Band SFDR
10 MHz AOUT(±1 MHz) 25°C V 83 83 dBc
10 MHz AOUT (±250 kHz) 25°C V 83 83 dBc
10 MHz AOUT (±50 kHz) 25°C V 91 91 dBc
41 MHz AOUT (±1 MHz) 25°C V 82 82 dBc
41 MHz AOUT (±250 kHz) 25°C V 84 84 dBc
41 MHz AOUT (±50 kHz) 25°C V 89 89 dBc
119 MHz AOUT (±1 MHz) 25°C V 71 71 dBc
119 MHz AOUT (±250 kHz) 25°C V 77 77 dBc
119 MHz AOUT (±50 kHz) 25°C V 83 83 dBc
AD9854
Rev. C | Page 5 of 52
AD9854ASQ AD9854AST
Parameter Temp
Tes t
Level Min Typ Max Min Typ Max Unit
Residual Phase Noise
(AOUT = 5 MHz, Ext. CLK = 30 MHz,
REFCLK Multiplier Engaged at 10×)
1 kHz Offset 25°C V 140 140 dBc/Hz
10 kHz Offset 25°C V 138 138 dBc/Hz
100 kHz Offset 25°C V 142 142 dBc/Hz
(AOUT = 5 MHz, Ext. CLK = 300 MHz,
REFCLK Multiplier Bypassed)
1 kHz Offset 25°C V 142 142 dBc/Hz
10 kHz Offset 25°C V 148 148 dBc/Hz
100 kHz Offset 25°C V 152 152 dBc/Hz
Pipeline Delays4, , 5 6
DDS Core (Phase Accumulator and
Phase-to-Amp Converter)
25°C IV 33 33 SysClk
Cycles
Frequency Accumulator 25°C IV 26 26 SysClk
Cycles
Inverse Sinc Filter 25°C IV 16 16 SysClk
Cycles
Digital Multiplier 25°C IV 9 9 SysClk
Cycles
DAC 25°C IV 1 1 SysClk
Cycles
I/O Update Clock (INT MODE) 25°C IV 2 2 SysClk
Cycles
I/O Update Clock (EXT MODE) 25°C IV 3 3 SysClk
Cycles
MASTER RESET DURATION 25°C IV 10 10 SysClk
Cycles
COMPARATOR INPUT CHARACTERISTICS
Input Capacitance 25°C V 3 3 pF
Input Resistance 25°C IV 500 500 kΩ
Input Current 25°C I ±1 ±5 ±1 ±5 µA
Hysteresis 25°C IV 10 20 10 20 mV p-p
COMPARATOR OUTPUT CHARACTERISTICS
Logic 1 Voltage, High Z Load Full VI 3.1 3.1 V
Logic 0 Voltage, High Z Load Full VI 0.16 0.16 V
Output Power, 50 Ω Load,
120 MHz Toggle Rate
25°C I 9 11 9 11 dBm
Propagation Delay 25°C IV 3 3 ns
Output Duty Cycle Error725°C I −10 ±1 +10 −10 ±1 +10 %
Rise/Fall Time, 5 pF Load 25°C V 2 2 ns
Toggle Rate, High Z Load 25°C IV 300 350 300 350 MHz
Toggle Rate, 50 Ω Load 25°C IV 375 400 375 400 MHz
Output Cycle-to-Cycle Jitter8 IV 4.0 4.0 Ps rms
AD9854
Rev. C | Page 6 of 52
AD9854ASQ AD9854AST
Parameter Temp
Tes t
Level Min Typ Max Min Typ Max Unit
COMPARATOR NARROW-BAND SFDR9
10 MHz (±1 MHz) 25°C V 84 84 dBc
10 MHz (±250 MHz) 25°C V 84 84 dBc
10 MHz (±50 MHz) 25°C V 92 92 dBc
41 MHz (±1 MHz) 25°C V 76 76 dBc
41 MHz (±250 MHz) 25°C V 82 82 dBc
41 MHz (±50 MHz) 25°C V 89 89 dBc
119 MHz (±1 MHz) 25°C V 73 dBc
119 MHz (±250 MHz) 25°C V 73 dBc
119 MHz (±50 MHz) 25°C V 83 dBc
CLOCK GENERATOR OUTPUT JITTER9
5 MHz AOUT 25°C V 23 23 Ps rms
40 MHz AOUT 25°C V 12 12 Ps rms
100 MHz AOUT 25°C V 7 7 Ps rms
PARALLEL I/O TIMING CHARACTERISTICS
TASU (Address Setup Time to WR Signal
Active)
Full IV 8.0 7.5 8.0 7.5 ns
TADHW (Address Hold Time to WR Signal
Inactive)
Full IV 0 0 ns
TDSU (Data Setup Time to WR Signal Inactive) Full IV 3.0 1.6 3.0 1.6 ns
TDHD (Data Hold Time to WR Signal Inactive) Full IV 0 0 ns
TWRLOW (WR Signal Minimum Low Time) Full IV 2.5 1.8 2.5 1.8 ns
TWRHIGH (WR Signal Minimum High Time) Full IV 7 7 ns
TWR (Minimum Write Time) Full IV 10.5 10.5 ns
TADV (Address to Data Valid Time) Full V 15 15 15 15 ns
TADHR (Address Hold Time to RD Signal
Inactive)
Full IV 5 5 ns
TRDLOV (RD Low-to-Output Valid) Full IV 15 15 ns
TRDHOZ (RD High-to-Data Three-State) Full IV 10 10 ns
SERIAL I/O TIMING CHARACTERISTICS
TPRE (CS Setup Time) Full IV 30 30 ns
TSCLK (Period of Serial Data Clock) Full IV 100 100 ns
TDSU (Serial Data Setup Time) Full IV 30 30 ns
TSCLKPWH (Serial Data Clock Pulse Width High) Full IV 40 40 ns
TSCLKPWL (Serial Data Clock Pulse Width Low) Full IV 40 40 ns
TDHLD (Serial Data Hold Time) Full IV 0 0 ns
TDV (Data Valid Time) Full V 30 30 ns
CMOS LOGIC INPUTS10
Logic 1 Voltage 25°C I 2.2 2.2 V
Logic 0 Voltage 25°C I 0.8 0.8 V
Logic 1 Current 25°C IV ±5 ±12 µA
Logic 0 Current 25°C IV ±5 ±12 µA
Input Capacitance 25°C V 3 3 pF
POWER SUPPLY11
+VS Current12 25°C I 1050 1210 755 865 mA
+VS Current13 25°C I 710 816 515 585 mA
+VS Current14 25°C I 600 685 435 495 mA
PDISS12 25°C I 3.475 4.190 2.490 3.000 W
PDISS13 25°C I 2.345 2.825 1.700 2.025 W
PDISS14 25°C I 1.975 2.375 1.435 1.715 W
PDISS Power-Down Mode 25°C I 1 50 1 50 mW
AD9854
Rev. C | Page 7 of 52
1 The reference clock inputs are configured to accept a 1 V p-p (typical) dc offset square or sine wave centered at one-half the applied VDD or a 3 V TTL-level pulse input.
2 An internal 400 mV p-p differential voltage swing equates to 200 mV p-p applied to both REFCLK input pins.
3 The I and Q gain imbalance is digitally adjustable to less than 0.01 dB.
4 Pipeline delays of each individual block are fixed; however, if the eight top MSBs of a tuning word are all zeros, the delay appears longer. This is due to insufficient
phase accumulation per a system CLK period to produce enough LSB amplitude to the D/A converter.
5 If a feature such as the inverse sinc, which has 16 pipeline delays, can be bypassed, the total delay is reduced by that amount.
6 The I/O update CLK transfers data from the I/O port buffers to the programming registers. This transfer is measured in system clocks.
7 Change in duty cycle from 1 MHz to 100 MHz with 1 V p-p sine wave input and 0.5 V threshold.
8 Represents comparator’s inherent cycle-to-cycle jitter contribution. Input signal is a 1 V, 40 MHz square wave. Measurement device Wavecrest DTS – 2075.
9 Comparator input originates from analog output section via external 7-pole elliptic LPF. Single-ended input, 0.5 V p-p. Comparator output terminated in 50 Ω.
10 Avoid overdriving digital inputs. (Refer to equivalent circuits in .) Figure 3
11 Simultaneous operation at the maximum ambient temperature of 85°C and the maximum internal clock frequency of 200 MHz for the 80-lead LQFP, or 300 MHz for
the thermally enhanced 80-lead LQFP, may cause the maximum die junction temperature of 150°C to be exceeded. Refer to the
section for derating and thermal management information.
Power Dissipation and Thermal
Considerations
12 All functions engaged.
13 All functions except inverse sinc engaged.
14 All functions except inverse sinc and digital multipliers engaged.
AD9854
Rev. C | Page 8 of 52
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Rating
Maximum Junction Temperature 150°C
VS4 V
Digital Inputs −0.7 V to +VS
Digital Output Current 5 mA
Storage Temperature −65°C to +150°C
Operating Temperature −40°C to +85°C
Lead Temperature (Soldering, 10 s) 300°C
Maximum Clock Frequency (ASQ) 300 MHz
Maximum Clock Frequency (AST) 200 MHz
θJA (ASQ) 16°C/W
θJA (AST) 38°C/W
θJC (ASQ) 2°C/W
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
EXPLANATION OF TEST LEVELS
1. 100% production tested.
3. Sample tested only.
4. Parameter is guaranteed by design and characterization
testing.
5. Parameter is a typical value only.
6. Devices are 100% production tested at 25°C and
guaranteed by design and characterization testing for
industrial operating temperature range.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
AD9854
Rev. C | Page 9 of 52
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
00634-B-002
80 79 78 77 76 71 70 69 6875 74 73 72
21 22 23 24 25 26 27 28 29 30 31 32 33
1
2
3
4
5
6
7
8
9
10
11
13
12
60
59
58
57
56
55
54
53
52
51
50
49
48
NC = NO CONNECT
AD9854
TOP VIEW
(Not to Scale)
WR/SCLK
RD/CS
DVDD
DVDD
DVDD
DGND
DGND
DGND
FSK/BPSK/HOLD
SHAPED KEYING
AVDD
AVDD
AGND
DVDD
DVDD
DGND
DGND
DGND
DGND
DVDD
DVDD
DGND
MASTER RESET
S/P SELECT
REFCLK
REFCLK
D7
D6
D5
D4
D3
D2
D1
D0
DVDD
DVDD
DGND
DGND
NC
AVDD
AGND
NC
NC
DAC R
SET
DACBP
AVDD
AGND
IOUT2
IOUT2
AVDD
IOUT1
IOUT1
PIN 1
INDICATOR
14
15
16
17
18
20
19
47
46
45
44
43
42
41
A5
A4
A3
A2/IO RESET
A1/SDO
A0/SDIO
I/O UD CLK
AGND
AGND
AGND
AVDD
VINN
VINP
AGND
64 63 62 6167 66 65
34 35 36 37 38 39 40
AGND
NC
VOUT
AVDD
AVDD
AGND
AGND
AGND
AGND
AVDD
DIFF CLK ENABLE
NC
AGND
PLL FILTER
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. Mnemonic Description
1 to 8 D7 to D0 8-Bit Bidirectional Parallel Programming Data Inputs. Used only in parallel programming mode.
9, 10, 23, 24, 25,
73, 74, 79, 80
DVDD Connections for the Digital Circuitry Supply Voltage. Nominally 3.3 V more positive than AGND
and DGND.
11, 12, 26, 27, 28,
72, 75, 76, 77, 78
DGND Connections for Digital Circuitry Ground Return. Same potential as AGND.
13, 35, 57, 58, 63 NC No Internal Connection.
14 to 19 A5 toA0 Six-Bit Parallel Address Inputs for Program Registers. Used only in parallel programming mode.
Pin 17 (A2), Pin 18 (A1), and Pin 19 (A0) have a second function when the serial programming
mode is selected, as described next.
(17) A2/IO RESET
Allows an IO RESET of the serial communications bus that is unresponsive due to improper
programming protocol. Resetting the serial bus in this manner does not affect previous
programming nor does it invoke the default programming values listed in Table 7. Active high.
(18) A1/SDO Unidirectional Serial Data Output. Used in 3-wire serial communication mode.
(19) A0/SDIO Bidirectional Serial Data Input/Output. Used in 2-wire serial communication mode.
20 I/O UD CLK Bidirectional I/O Update CLK. Direction is selected in control register. If selected as an input, a
rising edge transfers the contents of the I/O port buffers to the programming registers. If I/O UD
CLK is selected as an output (default), an output pulse (low to high) of an eight-system-clock-
cycle duration indicates that an internal frequency update has occurred.
21 WR/SCLK Write Parallel Data to I/O Port Buffers. Shared function with SCLK. Serial clock signal associated
with the serial programming bus. Data is registered on the rising edge. This pin is shared with WR
when the parallel mode is selected. Mode dependent on Pin 70 (S/P select).
AD9854
Rev. C | Page 10 of 52
Pin No. Mnemonic Description
22 RD/CS Read Parallel Data from Programming Registers. Shared function with CS. Chip-select signal
associated with the serial programming bus. Active Low. This pin is shared with RD when parallel
mode is selected.
29 FSK/BPSK/HOLD
Multifunction pin according to the mode of operation selected in the programming control
register. In FSK mode, logic low selects F1, logic high selects F2. In BPSK mode, logic low selects
Phase 1, logic high selects Phase 2. In chirp mode, logic high engages the hold function, causing
the frequency accumulator to halt at its current location. To resume or commence chirp mode,
logic low is asserted.
30 SHAPED KEYING
Must first be selected in the programming control register to function. A logic high causes the I
and Q DAC outputs to ramp up from zero-scale to full-scale amplitude at a preprogrammed rate.
Logic low causes the full-scale output to ramp down to zero scale at the preprogrammed rate.
31, 32, 37, 38, 44,
50, 54, 60, 65
AVDD Connections for the Analog Circuitry Supply Voltage. Nominally 3.3 V more positive than AGND
and DGND.
33, 34, 39, 40, 41,
45, 46, 47, 53, 59,
62, 66, 67
AGND Connections for Analog Circuitry Ground Return. Same potential as DGND.
36 VOUT
Internal High Speed Comparator’s Noninverted Output Pin. Designed to drive 10 dBm to 50 Ω
load as well as standard CMOS logic levels.
42 VINP Voltage Input Positive. The internal high speed comparator’s noninverting input.
43 VINN Voltage Input Negative. The internal high speed comparator’s inverting input.
48 IOUT1 Unipolar Current Output of the I or Cosine DAC. (Refer to Figure 3.)
49 IOUT1 Complementary Unipolar Current Output of the I or Cosine DAC.
51 IOUT2 Complementary Unipolar Current Output of the Q or Sine DAC.
52 IOUT2
Unipolar Current Output of the Q or Sine DAC. This DAC can be programmed to accept external
12-bit data in lieu of internal sine data, allowing the AD9854 to emulate the AD9852 control DAC
function.
55 DACBP
Common Bypass Capacitor Connection for both I and Q DACs. A 0.01 µF chip capacitor from this
pin to AVDD improves harmonic distortion and SFDR slightly. No connect is permissible (slight
SFDR degradation).
56 DAC RSET Common Connection for both I and Q DACs to set the full-scale output current. RSET = 39.9/IOUT.
Normal RSET range is from 8 kΩ (5 mA) to 2 kΩ (20 mA).
61 PLL FILTER
Provides the connection for the external zero compensation network of the REFCLK multiplier’s
PLL loop filter. The zero compensation network consists of a 1.3 kΩ resistor in series with a 0.01 µF
capacitor. The other side of the network should be connected to AVDD as close as possible to
Pin 60. For optimum phase noise performance, the REFCLK multiplier can be bypassed by setting
the Bypass PLL bit in Control Register 1E.
64 DIFF CLK ENABLE Differential REFCLK Enable. A high level of this pin enables the differential clock inputs, REFCLK
and REFCLK (Pins 69 and 68, respectively).
68 REFCLK The Complementary Differential Clock Signal (180 Degrees Out of Phase). User should tie this pin
high or low when single-ended clock mode is selected. Same signal levels as REFCLK.
69 REFCLK
Single-Ended Reference Clock Input (CMOS Logic Levels Required) or One of Two Differential
Clock Signals. In differential ref clock mode, both inputs can be CMOS logic levels or have greater
than 400 mV p-p square or sine waves centered about 1.6 V dc.
70 S/P SELECT Selects Serial Programming Mode (Logic low) or Parallel Programming Mode (Logic High).
71 MASTER RESET
Initializes the serial/parallel programming bus to prepare for user programming; sets
programming registers to a do-nothing state defined by the default values listed in Table 7.
Active on logic high. Asserting master reset is essential for proper operation on power-up.
AD9854
Rev. C | Page 11 of 52
00636-B-003
VINP/
VINN
AVDD
I
OUT
I
OUTB
MUST TERMINATE OUTPUTS
FOR CURRENT FLOW. DO
NOT EXCEED THE OUTPUT
VOLTAGE COMPLIANCE RATING.
COMPARATOR
OUT
AVDD
DVDD
DIGITAL
IN
AVOID OVERDRIVING
DIGITAL INPUTS. FORWARD
BIASING ESD DIODES MAY
COUPLE DIGITAL NOISE
ONTO POWER PINS.
A. DAC OUTPUTS B. COMPARATOR OUTPUT C. COMPARATOR INPUT D. DIGITAL INPUTS
AVDD
Figure 3. Equivalent Input and Output Circuits
AD9854
Rev. C | Page 12 of 52
TYPICAL PERFORMANCE CHARACTERISTICS
Figure 4 to Figure 9 indicate the wideband harmonic distortion performance of the AD9854 from 19.1 MHz to 119.1 MHz fundamental
output, reference clock = 30 MHz, REFCLK multiplier = 10. Each graph plotted from 0 MHz to 150 MHz (Nyquist).
0
START 0Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 15MHz/ STOP 150MHz
00636-0-004
Figure 4. Wideband SFDR, 19.1 MHz
0
START 0Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 15MHz/ STOP 150MHz
00636-0-005
Figure 5. Wideband SFDR, 39.1 MHz
0
START 0Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 15MHz/ STOP 150MHz
00636-0-006
Figure 6. Wideband SFDR, 59.1 MHz
0
START 0Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 15MHz/ STOP 150MHz
00636-0-007
Figure 7. Wideband SFDR, 79.1 MHz
0
START 0Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 15MHz/ STOP 150MHz
00636-B-008
Figure 8. Wideband SFDR, 99.1 MHz
0
START 0Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 15MHz/ STOP 150MHz
00636-B-009
Figure 9. Wideband SFDR, 119.1 MHz
AD9854
Rev. C | Page 13 of 52
Figure 10 to Figure 13 show the trade-off in elevated noise floor, increased phase noise, and discrete spurious energy when the internal
REFCLK multiplier circuit is engaged. Plots with wide (1 MHz) and narrow (50 kHz) spans are shown. Compare the noise floor of
Figure 11 and Figure 13 to Figure 14 and Figure 15. The improvement seen in Figure 11 and Figure 13 is a direct result of sampling the
fundamental at a higher rate. Sampling at a higher rate spreads the quantization noise of the DAC over a wider bandwidth, which
effectively lowers the noise floor.
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 100kHz/ SPAN 1MHz
00636-B-010
Figure 10. Narrow-Band SFDR, 39.1 MHz, 1 MHz BW,
300 MHz REFCLK with REFCLK Multiply Bypassed
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 5kHz/ SPAN 50kHz
00636-B-011
Figure 11. Narrow-Band SFDR, 39.1 MHz, 50 kHz BW,
300 MHz REFCLK with REFCLK Multiply Bypassed
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 100kHz/ SPAN 1MHz
00636-B-012
Figure 12. Narrow-Band SFDR, 39.1 MHz, 1 MHz BW,
30 MHz REFCLK with REFCLK Multiply = 10x
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 5kHz/ SPAN 50kHz
00636-B-013
Figure 13. Narrow-Band SFDR, 39.1 MHz, 50 kHz BW,
30 MHz REFCLK with REFCLK Multiply = 10x
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 5kHz/ SPAN 50kHz
00636-B-014
Figure 14. Narrow-Band SFDR, 39.1 MHz, 50 kHz BW,
100 MHz REFCLK with REFCLK Multiply Bypassed
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 5kHz/ SPAN 50kHz
00636-B-015
Figure 15. Narrow-Band SFDR, 39.1 MHz, 50 kHz BW,
10 MHz REFCLK with REFCLK Multiply = 10x
AD9854
Rev. C | Page 14 of 52
Figure 16 and Figure 17 show the narrow-band performance of the AD9854 when operating with a 20 MHz reference clock and the
REFCLK multiplier enabled at 10× vs. a 200 MHz reference clock with REFCLK multiplier bypassed.
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100 5kHz/ SPAN 50kHz
00636-B-016
Figure 16. Narrow-Band SFDR, 39.1 MHz, 50 kHz BW,
200 MHz REFCLK with REFCLK Multiply Bypassed
0
CENTER 39.1MHz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–
100 5kHz/ SPAN 50kHz
00636-B-017
Figure 17. Narrow-Band SFDR, 39.1 MHz, 50 kHz BW,
20 MHz REFCLK with REFCLK Multiply = 10x
–100
–110
–150
–120
–130
–140
–160
–170
PHASE NOISE (dBc/Hz)
A
OUT
= 80MHz
A
OUT
= 5MHz
00636-B-018
FREQUENCY (Hz)
10 1M100 100k10k1k
Figure 18. Residual Phase Noise, 300 MHz REFCLK with
REFCLK Multiplier Bypassed
FREQUENCY (Hz)
–90
–100
–140
–110
–120
–130
–150
–16010 1M100 100k10k1k
PHASE NOISE (dBc/Hz)
A
OUT
= 80MHz
A
OUT
= 5MHz
00636-B-019
Figure 19. Residual Phase Noise, 30 MHz REFCLK with
REFCLK Multiplier = 10x
DAC CURRENT (mA)
55
0
SFDR (dBc)
54
53
52
51
50
49
48 510152025
00636-B-020
Figure 20. SFDR vs. DAC Current, 59.1 AOUT, 300 MHz
REFCLK with REFCLK Multiplier Bypassed
FREQUENCY (MHz)
620
0
SUPPLY CURRENT (mA)
615
610
605
600
595
590 20 40 60 80 100 120 140
00636-B-021
Figure 21. Supply Current vs. Output Frequency; Variation
Is Minimal as a Percentage and Heavily Dependent on Tuning Word
AD9854
Rev. C | Page 15 of 52
RISE TIME
1.04ns
500ps/DIV 232mV/DIV 50
Ω
INPUT
JITTER
[10.6ps RMS]
–33ps 0ps +33ps
00636-B-022
Figure 22. Typical Comparator Output Jitter, 40 MHz AOUT, 300 MHz RFCLK
with REFCLK Multiplier Bypassed
CH1 500mV
Ω
M 500ps CH1 980mV
00636-B-023
REF1 RISE
1.174ns
C1 FALL
1.286ns
Figure 23. Comparator Rise/Fall Times
FREQUENCY (MHz)
1200
0
AMPLITUDE (mV p-p)
1000
800
600
400
200
0100 200 300 400 500
MINIMUM COMPARATOR
INPUT DRIVE
V
CM
= 0.5V
00636-B-024
Figure 24. Comparator Toggle Voltage Requirement
AD9854
Rev. C | Page 16 of 52
TYPICAL APPLICATIONS
LPF
REFCLK
RF/IF
INPUT
I BASEBAND
COS
LPF
LPF
AD9854
Q BASEBAND
LPF
CHANNEL
SELECT
FILTERS
SIN
00636-B-025
Figure 25. Quadrature Downconversion
LPF
REFCLK
COS
LPF
AD9854
SIN
RF OUTPUT
I BASEBAND
Q BASEBAND
00636-B-026
Figure 26. Direct Conversion Quadrature Upconverter
I
Q
Rx
RF IN
DUAL
8-/10-BIT
ADC
DIGITAL
DEMODULATOR Rx BASEBAND
DIGITAL DATA
OUT
8
8
I/Q MIXER
AND
LOW-PASS
FILTER
VCA
ADC ENCODE
ADC CLOCK FREQUENCY
LOCKED TO Tx CHIP/
SYMBOL/PN RATE
REFERENCE
CLOCK
48
CHIP/SYMBOL/PN
RATE DATA
AD9854
CLOCK
GENERATOR
AGC
00636-B-027
Figure 27. Chip Rate Generator in Spread Spectrum Application
50
Ω
BAND-PASS
FILTER
50
Ω
IOUT
AD9854
FUNDAMENTAL
FC– FO
IMAGE
FCLK
FC + FO
IMAGE
BAND-PASS
FILTER
FC + FO
IMAGE
AD9854
SPECTRUM FINAL OUTPUT
SPECTRUM
AMPLIFIER
00636-B-028
Figure 28. Using an Aliased Image to Generate a High Frequency
AD9854
Rev. C | Page 17 of 52
VCO
LOOP
FILTER
PHASE
COMPARATOR
REFERENCE
CLOCK
FILTER
AD9854
DDS
TUNING
WORD
REF CLK IN
RF
FREQUENCY
OUT
DAC OUT PROGRAMMABLE
"DIVIDE-BY-N" FUNCTION
(WHERE N = 2
48
/TUNING WORD)
00636-B-029
Figure 29. Programmable Fractional Divide-by-N Synthesizer
TUNING
WORD
VCO
LOOP
FILTER
PHASE
COMPARATOR
REF
CLOCK RF
FREQUENCY
OUT
FILTER
AD9854
DDS
DIVIDE-BY-N
00636-B-030
Figure 30. Agile High Frequency Synthesizer
PHASE
SPLITTER
Σ
0.8 TO
2.5GHz AD9854
QUADRATURE
DDS
DDS – LO LO DDS
+ LO
36dB
TYPICAL
SSB
REJECTION 50Ω
V
OUT
AD8346 QUADRATURE
MODULATOR
90
COSINE (DC TO 70MHz)
SINE (DC TO 70MHz)
LO
LO
0
NOTES
FLIP DDS QUADRATURE SIGNALS TO SELECT ALTERNATE SIDEBAND. ADJUST DDS
SINE OR COSINE SIGNAL AMPLITUDE FOR GREATEST SIDEBAND SUPPRESSION.
DDS DAC OUTPUTS MUST BE LOW-PASS FILTERED PRIOR TO USE WITH THE AD8346.
(REFER TO THE TECHNICAL NOTE AT WEBSITE [WWW.ANALOG.COM/DDS])
00636-B-031
Figure 31. Single Sideband Upconversion
REFERENCE
CLOCK
50
Ω
1:1 TRANSFORMER
I.E, MINICIRCUITS T1–1T
FILTER 50
Ω
DIFFERENTIAL
TRANSFORMER-COUPLED
OUTPUT
AD9854
DDS
I
OUT
I
OUT
00636-B-032
Figure 32. Differential Output Connection for Reduction of Common-Mode Signals
AD9854
Rev. C | Page 18 of 52
CLOCK OUT = 200MHz
LPF SIN
LPF
AD9854
COS
REFERENCE
CLOCK
COMPARATORS
A
OUT
= 100MHz
00636-B-032
Figure 33. Clock Frequency Doubler
µ
PROCESSOR/
CONTROLLER
FPGA, ETC.
R
SET
8-BIT PARALLEL OR
SERIAL PROGRAMMING
DATA AND CONTROL
SIGNALS
AD9854
+
CMOS LOGIC "CLOCK" OUT
REFERENCE
CLOCK
300MHz MAX DIRECT
MODE OR 15 TO 75MHz
MAX IN THE 4
×
TO 20
×
CLOCK
MULTIPLIER MODE
2k
Ω
I DAC 1
2
NOTES
I
OUT
= APPROX 20mA MAX WHEN R
SET
= 2k
Ω
SWITCH POSTION 1 PROVIDES COMPLEMENTARY
SINUSOIDAL SIGNALS TO THE COMPARATOR
TO PRODUCE A FIXED 50% DUTY CYCLE FROM THE
COMPARATOR.
SWITCH POSTION 2 PROVIDES THE SAME DUTY CYCLE
USING QUADRATURE SINUSOIDAL SIGNALS TO THE
COMPARATOR OR A DC THRESHOLD VOLTAGE TO
ALLOW SETTING OF THE COMPARATOR DUTY CYCLE
(DEPENDS ON THE Q DAC's CONFIGURATION)
Q DAC OR
CONTROL
DAC
LOW-PASS
FILTER
LOW-PASS
FILTER
00636-B-034
Figure 34. Frequency Agile Clock Generator Applications for the AD9854
AD9854
Rev. C | Page 19 of 52
THEORY OF OPERATION
The AD9854 quadrature output digital synthesizer is a highly
flexible device that addresses a wide range of applications. The
device consists of an NCO with a 48-bit phase accumulator, a
programmable reference clock multiplier, inverse sinc filters,
digital multipliers, two 12-bit/300 MHz DACs, a high speed
analog comparator, and interface logic. This highly integrated
device can be configured to serve as a synthesized LO, agile
clock generator, and FSK/BPSK modulator.
Analog Devices, Inc. provides a technical tutorial about the
operational theory of the functional blocks of the device. The
tutorial includes a technical description of the signal flow
through a DDS device and provides basic applications infor-
mation for a variety of digital synthesis implementations. The
document, “A Technical Tutorial on Digital Signal Synthesis,” is
available from the DDS Technical Library, on the Analog
Devices DDS website at www.analog.com/dds.
MODES OF OPERATION
The AD9854 has five programmable operational modes. To
select a mode, three bits in the control register (parallel Address
1F hex) must be programmed, as described in Table 4.
Table 4. Mode Selection Table
Mode 2 Mode 1 Mode 0 Result
0 0 0 Single Tone
0 0 1 FSK
0 1 0 Ramped FSK
0 1 1 Chirp
1 0 0 BPSK
In each mode, some functions may not be permitted. Table 5
lists the functions and their availability for each mode.
Single-Tone (Mode 000)
This is the default mode when master reset is asserted. It may
also be accessed if it is user programmed into the control
register. The phase accumulator, responsible for generating an
output frequency, is presented with a 48-bit value from
Frequency Tuning Word 1 registers whose default values are
zero. Default values from the remaining applicable registers
further define the single-tone output signal qualities.
The default values after a master reset configure the device
with an output signal of 0 Hertz, 0 phase. At power-up and
reset, the output from the I and Q DACs is a dc value equal to
the midscale output current. This is the default mode amplitude
setting of zero. See the Shaped On/Off Keying section for more
details about the output amplitude control. All or some of the
28 program registers must be programmed to realize a user-
defined output signal.
Figure 35 shows the transition from the default condition
(0 Hz) to a user-defined output frequency (F1).
Table 5. Functions Availabile for Modes
Mode
Function Single-Tone FSK Ramped FSK Chirp BPSK
Phase Adjust 1 ✓ ✓ ✓ ✓ ✓
Phase Adjust 2 ✓
Single-Pin FSK/BPSK or HOLD ✓ ✓ ✓ ✓
Single-Pin Shaped-Keying ✓ ✓ ✓ ✓ ✓
Phase Offset or Modulation ✓ ✓ ✓ ✓
Amplitude Control or Modulation ✓ ✓ ✓ ✓ ✓
Inverse SINC Filter ✓ ✓ ✓ ✓ ✓
Frequency Tuning Word 1 ✓ ✓ ✓ ✓ ✓
Frequency Tuning Word 2 ✓ ✓
Automatic Frequency Sweep ✓ ✓
AD9854
Rev. C | Page 20 of 52
000 (SINGLE TONE)
MODE
F1
TW1
000 (DEFAULT)
0
F1
0
FREQUENCY
MASTER RESET
I/O UPDATE
CLOCK
00636-B-035
Figure 35. Default State to User-Defined Output Transition
As with all Analog Devices DDS devices, the value of the
frequency tuning word is determined by
FTW = (Desired Output Frequency × 2N)/SYSCLK
where:
N is the phase accumulator resolution (48 bits in this instance).
Frequency is expressed in Hz.
FTW (frequency tuning word) is a decimal number.
Once a decimal number has been calculated, it must be
rounded to an integer and then converted to binary format—a
series of 48 binary-weighted 1s or 0s. The fundamental sine
wave DAC output frequency range is from dc to 1/2 SYSCLK.
Changes in frequency are phase continuous, which means that
the first sampled phase value of the new frequency is referenced
in time from the last sampled phase value of the previous
frequency.
The I and Q DACs of the AD9854 are always 90° out of phase.
The 14-bit phase registers do not independently adjust the
phase of each DAC output. Instead, both DACs are affected
equally by a change in phase offset.
The single-tone mode allows the user to control the following
signal qualities:
• Output frequency to 48-bit accuracy
• Output amplitude to 12-bit accuracy
– Fixed, user-defined, amplitude control
– Variable, programmable amplitude control
– Automatic, programmable, single-pin-controlled,
shaped on/off keying
• Output phase to 14-bit accuracy
All of these qualities can be changed or modulated via the 8-bit
parallel programming port at a 100 MHz parallel-byte rate or at
a 10 MHz serial rate. Incorporating this attribute permits FM,
AM, PM, FSK, PSK, and ASK operation in single-tone mode.
Unramped FSK (Mode 001)
When selected, the output frequency of the DDS is a function
of the values loaded into Frequency Tuning Word Register 1
and 2 and the logic level of Pin 29 (FSK/BPSK/HOLD). A logic
low on Pin 29 chooses F1 (Frequency Tuning Word 1, Parallel
Address 4 to 9 hex) and a logic high chooses F2 (Frequency
Tuning Word 2, Parallel Register Address A to F hex). Changes
in frequency are phase continuous and are internally coincident
with the FSK data pin (29); however, there is deterministic
pipeline delay between the FSK data signal and the DAC output.
(Please refer to pipeline delays in Table 1.)
The unramped FSK mode, shown in Figure 36, represents
traditional FSK, Radio Teletype (RTTY), or Teletype (TTY)
transmission of digital data. FSK is a very reliable means of
digital communication; however, it makes inefficient use of the
bandwidth in the RF spectrum. Ramped FSK, shown in
Figure 37, is a method of conserving bandwidth.
Ramped FSK (Mode 010)
This mode is a method of FSK whereby changes from F1 to F2
are not instantaneous but, instead, are accomplished in a
frequency sweep or ramped fashion (the ramped notation
implies that the sweep is linear). While linear sweeping or
frequency ramping is easily and automatically accomplished, it
is only one of many schemes. Other frequency transition
schemes may be implemented by changing the ramp rate and
ramp step size on-the-fly, in piecewise fashion.
AD9854
Rev. C | Page 21 of 52
F1
F2
0
FREQUENCY
MODE
TW1
TW2
FSK DATA (PIN 29)
001 (FSK NO RAMP)
F1
F2
000 (DEFAULT)
0
0
I/O UPDATE CLK
00636-B-036
Figure 36. Traditional FSK Mode
I/O UPDATE CLK
F1
F2
0
FREQUENCY
MODE
TW1
TW2
010 (RAMPED FSK)
F1
F2
000 (DEFAULT)
0
0
REQUIRES A POSITIVE TWOS COMPLEMENT VALUE
RAMP RATE
DFW
FSK DATA (PIN 29)
00636-B-037
Figure 37. Ramped FSK Mode
F1
F2
0
FREQUENCY
MODE
TW1
TW2
FSK DATA
010 (RAMPED FSK)
F1
F2
000 (DEFAULT)
0
0
I/O UPDATE
CLOCK
00636-B-038
Figure 38. Ramped FSK Mode
AD9854
Rev. C | Page 22 of 52
Frequency ramping, whether linear or nonlinear, necessitates
that many intermediate frequencies between F1 and F2 are
output in addition to the primary F1 and F2 frequencies.
Figure 37 and Figure 38 depict the frequency vs. time
characteristics of a linear ramped FSK signal.
Note that in ramped FSK mode, the delta frequency (DFW) is
required to be programmed as a positive twos complement
value. Another requirement is that the lowest frequency (F1)
be programmed in the Frequency Tuning Word 1 register.
The purpose of ramped FSK is to provide better bandwidth
containment than traditional FSK by replacing the instanta-
neous frequency changes with more gradual, user-defined
frequency changes. The dwell time at F1 and F2 can be equal
to, or much greater than, the time spent at each intermediate
frequency. The user controls the dwell time at F1 and F2, the
number of intermediate frequencies, and the time spent at each
frequency. Unlike unramped FSK, ramped FSK requires the
lowest frequency to be loaded into F1 registers and the highest
frequency into F2 registers.
Several registers must be programmed to instruct the DDS on
the resolution of intermediate frequency steps (48 bits) and the
time spent at each step (20 bits). Furthermore, the CLR ACC1
bit in the control register should be toggled (low-high-low)
prior to operation to ensure that the frequency accumulator is
starting from an all zeros output condition. For piecewise,
nonlinear frequency transitions, it is necessary to reprogram
the registers while the frequency transition is in progress to
affect the desired response.
Parallel Register Addresses 1A to 1C hex comprise the 20-bit
ramp rate clock registers. This is a countdown counter that
outputs a single pulse whenever the count reaches zero. The
counter is activated any time a logic level change occurs on
FSK input Pin 29. This counter is run at the system clock rate,
300 MHz maximum. The time period between each output
pulse is given as
(N + 1) × (System Clock Period)
where N is the 20-bit ramp rate clock value programmed by
the user.
The allowable range of N is from 1 to (220 − 1). The output of
this counter clocks the 48-bit frequency accumulator shown in
Figure 39. The ramp rate clock determines the amount of time
spent at each intermediate frequency between F1 and F2. The
counter stops automatically when the destination frequency is
achieved. The dwell time spent at F1 and F2 is determined by
the duration that the FSK input, Pin 29, is held high or low after
the destination frequency has been reached.
FREQUENCY
TUNING
WORD 2
FREQUENCY
TUNING
WORD 1
20-BIT
RAMP RATE
CLOCK
48-BIT DELTA-
FREQUENCY
WORD (TWOS
COMPLEMENT)
FREQUENCY
ACCUMULATOR
PHASE
ACCUMULATOR
INSTANTANEOUS
PHASE OUT
ADDER
FSK (PIN 29)
SYSTEM
CLOCK
00636-B-039
Figure 39. Block Diagram of Ramped FSK Function
Parallel Register Addresses 10 to 15 hex comprise the 48-bit,
twos complement, delta frequency word registers. This 48-bit
word is accumulated (added to the accumulator’s output) every
time it receives a clock pulse from the ramp rate counter. The
output of this accumulator is then added to or subtracted from
the F1 or F2 frequency word, which is then fed to the input of
the 48-bit phase accumulator that forms the numerical phase
steps for the sine and cosine wave outputs. In this fashion, the
output frequency is ramped up and down in frequency,
according to the logic state of Pin 29. The rate at which this
happens is a function of the 20-bit ramp rate clock. Once the
destination frequency is achieved, the ramp rate clock is
stopped, which halts the frequency accumulation process.
Generally speaking, the delta frequency word is a much
smaller value compared to that of the F1 or F2 tuning word.
For example, if F1 and F2 are 1 kHz apart at 13 MHz, the delta
frequency word might be only 25 Hz.
.
AD9854
Rev. C | Page 23 of 52
F1
F2
0
FREQUENCY
MODE
TW1
TW2
FSK DATA
TRIANGLE
BIT
010 (RAMPED FSK)
F1
F2
I/O UPDATE
CLOCK
00636-B-040
Figure 40. Effect of Triangle Bit in Amped FSK Mode
F1
F2
0
FREQUENCY
MODE
TW1
TW2
FSK DATA
F1
F2
000 (DEFAULT)
0
0
010 (RAMPED FSK)
I/O UPDATE
CLOCK
00636-B-041
Figure 41. Effect of Premature Ramped FSK Data
Figure 41 shows that premature toggling causes the ramp to
immediately reverse itself and proceed at the same rate and
resolution back to the originating frequency.
The control register contains a triangle bit at Parallel Register
Address 1F hex. Setting this bit high in Mode 010 causes an
automatic ramp-up and ramp-down between F1 and F2 to
occur without toggling Pin 29, as shown in Figure 40. The logic
state of Pin 29 has no effect once the triangle bit is set high. This
function uses the ramp rate clock time period and the delta
frequency word step size to form a continuously sweeping linear
ramp from F1 to F2 and back to F1 with equal dwell times at
every frequency. Using this function, one can automatically
sweep between any two frequencies from dc to Nyquist.
In the ramped FSK mode, with the triangle bit set high, an auto-
matic frequency sweep begins at either F1 or F2, according to
the logic level on Pin 29 (FSK input pin) when the triangle bit’s
rising edge occurs (Figure 42). If the FSK data bit is high instead
of low, F2, rather than F1, is chosen as the start frequency.
Additional flexibility in the ramped FSK mode is provided in
the ability to respond to changes in the 48-bit delta frequency
word and/or the 20-bit ramp rate counter on the fly during the
ramping from F1 to F2 or vice versa. To create these nonlinear
frequency changes, it is necessary to combine several linear
ramps, in a piecewise fashion, with differing slopes. This is done
by programming and executing a linear ramp at some rate or
slope and then altering the slope (by changing the ramp rate
clock or delta frequency word or both). Changes in slope are
made as often as needed to form the desired nonlinear
frequency sweep response before the destination frequency is
reached. These piecewise changes can be precisely timed using
the 32-bit internal update clock (see the Internal and External
Update Clock section).
AD9854
Rev. C | Page 24 of 52
Nonlinear ramped FSK has the appearance of a chirp function,
as Figure 46 shows. The difference between a ramped FSK
function and a chirp function is that FSK is limited to operation
between F1 and F2. Chirp operation has no F2 limit frequency.
Two additional control bits are available in the ramped FSK
mode that allow more options. CLR ACC1, Register Address
1F hex, if set high, clears the 48-bit frequency accumulator
(ACC1) output with a retriggerable one-shot pulse of one
system clock duration. If the CLR ACC1 bit is left high, a one-
shot pulse is delivered on the rising edge of every update clock.
The effect is to interrupt the current ramp, reset the frequency
back to the start point, F1 or F2, and then continue to ramp up
(or down) at the previous rate. This occurs even when a static
F1 or F2 destination frequency is achieved.
Next, CLR ACC2 Control Bit (Register Address 1F hex) is
available to clear both the frequency accumulator (ACC1) and
the phase accumulator (ACC2). When this bit is set high, the
output of the phase accumulator results in 0 Hz output from the
DDS. As long as this bit is set high, the frequency and phase
accumulators are cleared, resulting in 0 Hz output. To return to
previous DDS operation, CLR ACC2 must be set to logic low.
Chirp (Mode 011)
This mode is also known as pulsed FM. Most chirp systems use
a linear FM sweep pattern, but the AD9854 supports nonlinear
patterns, as well. In radar applications, use of chirp or pulsed
FM allows operators to significantly reduce the output power
needed to achieve the same result as a single-frequency radar
system would produce. Figure 43 shows a very low resolution
nonlinear chirp to demonstrate the different slopes that are
created by varying the time steps (ramp rate) and frequency
steps (delta frequency word).
F2
F1
0
FREQUENCY
MODE
TW1
TW2
FSK DATA
T
RIANGLE BI
T
000 (DEFAULT)
0
0
010 (RAMPED FSK)
F1
F2
00636-B-042
Figure 42. Automatic Linear Ramping Using the Triangle Bit
F1
0
FREQUENCY
010 (RAMPED FSK)
F1
000 (DEFAULT)
0
MODE
TW1
DFW
RAMP RAT
E
I/O UPDATE
CLOCK
00636-B-043
Figure 43. Example of a Nonlinear Chirp
AD9854
Rev. C | Page 25 of 52
The AD9854 permits precise, internally generated linear, or
externally programmed nonlinear, pulsed or continuous FM
over the complete frequency range, duration, frequency resolu-
tion, and sweep direction(s). These are all user programmable.
Figure 44 shows a block diagram of the FM chirp components.
20-BIT
RAMP RATE
CLOCK
48-BIT DELTA-
FREQUENCY
WORD (TWO
S
COMPLEMENT)
FREQUENCY
ACCUMULATOR
PHASE
ACCUMULATOR
OUT
ADDER
SYSTEM
CLOCK
CLR ACC2
CLR ACC1
FREQUENCY
TUNING
WORD 1
HOLD
00636-B-044
Figure 44. FM Chirp Components
Basic FM Chirp Programming Steps
1. Program a start frequency into Frequency Tuning Word 1
(FTW1) at Parallel Register Addresses 4 to 9 hex.
2. Program the frequency step resolution into the 48-bit,
twos complement, delta frequency word (Parallel Register
Addresses 10 to 15 hex).
3. Program the rate of change (time at each frequency) into
the 20-bit ramp rate clock (Parallel Register Addresses
1A to 1C hex).
When programming is complete, an I/O update pulse at Pin 20
engages the program commands.
The necessity for a twos complement delta frequency word is to
define the direction in which the FM chirp moves. If the 48-bit
delta frequency word is negative (MSB is high), then the incre-
mental frequency changes are in a negative direction from
FTW1. If the 48-bit word is positive (MSB is low), then the
incremental frequency changes are in a positive direction.
It is important to note that FTW1 is only a starting point for
FM chirp. There is no built-in restraint requiring a return to
FTW1. Once the FM chirp has begun, it is free to move,
under program control, within the Nyquist bandwidth (dc to
1/2 system clock). Instant return to FTW1 is easily achieved,
though, as explained next.
Two control bits are available in the FM chirp mode that allow
the return to the beginning frequency, FTW1, or to 0 Hz. First,
when the CLR ACC1 Bit (Register Address 1F hex) is set high,
the 48-bit frequency accumulator (ACC1) output is cleared
with a retriggerable one-shot pulse of one system clock dura-
tion. The 48-bit delta frequency word input to the accumulator
is unaffected by CLR ACC1 bit. If the CLR ACC1 bit is held
high, a one-shot pulse is delivered to the frequency accumulator
(ACC1) on every rising edge of the I/O update clock. The effect
is to interrupt the current chirp, reset the frequency back to
FTW1, and continue the chirp at the previously programmed
rate and direction. Clearing the output of the frequency
accumulator in the chirp mode is illustrated in Figure 45.
Shown in the diagram is the I/O update clock, which is either
user supplied or internally generated.
Next, CLR ACC2 Control Bit (Register Address 1F hex) is
available to clear both the frequency accumulator (ACC1) and
the phase accumulator (ACC2). When this bit is set high, the
output of the phase accumulator results in 0 Hz output from the
DDS. As long as this bit is set high, the frequency and phase
accumulators are cleared, resulting in 0 Hz output. To return to
previous DDS operation, CLR ACC2 must be set to logic low.
This bit is useful in generating pulsed FM.
Figure 46 illustrates the effect of the CLR ACC2 bit upon the
DDS output frequency. Note that reprogramming the registers
while the CLR ACC2 bit is high allows a new FTW1 frequency
and slope to be loaded.
Another function that is available only in chirp mode is the
hold function (Pin 29). This function stops the clock signal to
the ramp rate counter, halting any further clocking pulses to
the frequency accumulator, ACC1. The effect is to halt the chirp
at the frequency existing just before hold was pulled high.
When the Pin 29 is returned low, the clock resumes and chirp
continues. During a hold condition, the user may change the
programming registers; however, the ramp rate counter must
resume operation at its previous rate until a count of zero is
obtained before a new ramp rate count can be loaded. Figure 47
shows the hold function’s effect on the DDS output frequency.
AD9854
Rev. C | Page 26 of 52
I/O UPDATE
CLOCK
F1
0
FREQUENCY
MODE
FTW1
DFW
F1
000 (DEFAULT)
0
RAMP RATE RAMP RATE
011 (CHIRP)
DELTA FREQUENCY WORD
CLR ACC1
00636-B-045
Figure 45. Effect of CLR ACC1 in FM Chirp Mode
CLR ACC2
F1
0
FREQUENCY
MODE
TW1
DPW
000 (DEFAULT)
0
RAMP RAT
E
011 (CHIRP)
I/O UPDATE
CLOCK
00636-B-046
Figure 46. Effect of CLR ACC2 in Chirp Mode
HOLD
F1
0
FREQUENCY
MODE
TW1
DFW
000 (DEFAULT)
0
RAMP RATE
011 (CHIRP)
F1
DELTA FREQUENCY WORD
RAMP RATE
I/O UPDATE
CLOCK
00636-B-047
Figure 47. Illustration of Hold Function
AD9854
Rev. C | Page 27 of 52
BPSK DATA
360
0
PHASE
MODE
FTW1
PHASE ADJUST 1
000 (DEFAULT)
0
PHASE ADJUST 2
100 (BPSK)
F1
270 DEGREES
90 DEGREES
I/O UPDATE
CLOCK
00636-B-048
Figure 48. BPSK Mode
The 32-bit automatic I/O update counter may be used to con-
struct complex chirp or ramped FSK sequences. Because this
internal counter is synchronized with the AD9854 system clock,
it allows precisely timed program changes to be invoked. In this
manner, the user is only required to reprogram the desired
registers before the automatic I/O update clock is generated.
In chirp mode, the destination frequency is not directly
specified. If the user fails to control the chirp, the DDS naturally
confines itself to the frequency range between dc and Nyquist.
Unless terminated by the user, the chirp continues until power
is removed.
When the chirp destination frequency is reached, the following
can be done:
• Stop at the destination frequency using the hold pin, or by
loading all zeros into the delta frequency word registers of
the frequency accumulator (ACC1).
• Use the hold function to stop the chirp, then ramp down
the output amplitude using the digital multiplier stages and
the shaped keying pin, Pin 30, or via the program register
control (Addresses 21 to 24 hex).
• Abruptly end the transmission with the CLR ACC2 bit.
• Continue chirp by reversing direction and returning to
the previous, or another, destination frequency in a linear
or user-directed manner. If this involves going down in
frequency, a negative 48-bit delta frequency word (the MSB
is set to 1) must be loaded into Registers 10 hex to 15 hex.
Any decreasing frequency step of the delta frequency word
requires the MSB to be set to logic high.
• Continue chirp by immediately returning to the beginning
frequency (F1) in a saw-toothed fashion and repeat the
previous chirp process. This is where the CLR ACC1
control bit is used. An automatic, repeating chirp can be
set up using the 32-bit update clock to issue the CLR ACC1
command at precise time intervals. Adjusting the timing
intervals or changing the delta frequency word changes the
chirp range. It is incumbent upon the user to balance the
chirp duration and frequency resolution to achieve the
proper frequency range.
BPSK (Mode 100)
Binary, biphase, or bipolar-phase shift keying is a means to
rapidly select between two preprogrammed 14-bit output phase
offsets that identically affect both the I and Q outputs of the
AD9854. The logic state of Pin 29, the BPSK pin, controls the
selection of Phase Adjust Register 1 or 2. When low, Pin 29
selects Phase Adjust Register 1; when high, Phase Adjust
Register 2 is selected. Table 7 illustrates phase changes made
to four cycles of an output carrier.
Basic BPSK Programming Steps
1. Program a carrier frequency into Frequency Tuning
Word 1.
2. Program the appropriate 14-bit phase words in Phase
Adjust Registers 1 and 2.
3. Attach the BPSK data source to Pin 29.
4. Activate the I/O update clock when ready.
Note that for higher-order PSK modulation, the user can select
the single-tone mode and program Phase Adjust Register 1
using the serial or high speed parallel programming bus.
AD9854
Rev. C | Page 28 of 52
USING THE AD9854
INTERNAL AND EXTERNAL UPDATE CLOCK
This update clock function is comprised of a bidirectional
I/O pin, Pin 20, and a programmable 32-bit down counter. For
programming changes to be transferred from the I/O buffer
registers to the active core of the DDS, a clock signal (low-to-
high edge) must be externally supplied to Pin 20 or internally
generated by the 32-bit update clock.
When the user provides an external update clock, it is internally
synchronized with the system clock to prevent partial transfer
of program register information due to violation of data set up
or hold times. This mode gives the user complete control of
when updated program information becomes effective. The
default mode for the update clock is internal (Int Update Clk
control register bit is logic high). To switch to external update
clock mode, the Int Update Clk register bit must be set to logic
low. The internal update mode generates automatic, periodic
update pulses with the time period set by the user.
An internally generated update clock can be established by
programming the 32-bit update clock registers (Address 16 to
19 hex) and setting the Int Update Clk (Address 1F hex) control
register bit to logic high. The update clock down-counter
function operates at 1/2 the rate of the system clock (150 MHz
maximum) and counts down from a 32-bit binary value
(programmed by the user). When the count reaches 0, an auto-
matic I/O update of the DDS output or functions is generated.
The update clock is internally and externally routed on Pin 20
to allow users to synchronize programming of update infor-
mation with the update clock rate. The time period between
update pulses is given as
(N + 1) × (System Clock Period × 2)
where N is the 32-bit value programmed by the user.
The allowable range of N is from 1 to (232 − 1). The internally
generated update pulse output on Pin 20 has a fixed high time
of eight system clock cycles.
Programming the update clock register for values less than five
causes the I/O UD CLK pin to remain high. The update clock
functionality still works; however, the user cannot use the signal
as an indication that data is transferring. This is an effect of the
minimum high pulse time when I/O UD CLK is an output.
SHAPED ON/OFF KEYING
This feature allows the user to control the amplitude vs. time
slope of the I and Q DAC output signals. This function is used
in burst transmissions of digital data to reduce the adverse
spectral impact of short, abrupt bursts of data. Users must
first enable the digital multipliers by setting the OSK EN bit
(Control Register Address 20 hex) to logic high in the control
register. Otherwise, if the OSK EN bit is set low, the digital
multipliers responsible for amplitude control are bypassed and
the I and Q DAC outputs are set to full-scale amplitude. In
addition to setting the OSK EN bit, a second control bit, OSK
INT (also at Address 20 hex), must be set to logic high. Logic
high selects the linear internal control of the output ramp-up or
ramp-down function. A logic low in the OSK INT bit switches
control of the digital multipliers to user-programmable 12-bit
registers allowing users to dynamically shape the amplitude
transition in practically any fashion. These 12-bit registers,
labeled Output Shape Key I and Output Shape Key Q, are
located at Addresses 21 through 24 hex, as listed in Table 7. The
maximum output amplitude is a function of the RSET resistor
and is not programmable when OSK INT is enabled.
ABRUPT ON/OFF KEYING
SHAPED ON/OFF KEYING
ZERO
SCALE
ZERO
S
CALE
FULL
SCALE
FULL
SCALE
00636-B-049
Figure 49. Shaped On/Off Keying
The transition time from zero scale to full scale must also be
programmed. The transition time is a function of two fixed
elements and one variable. The variable element is the
programmable 8-bit ramp rate counter. This is a down counter
that is clocked at the system clock rate (300 MHz max) and that
generates one pulse whenever the counter reaches zero. This
pulse is routed to a 12-bit counter that increments with each
pulse received. The outputs of the 12-bit counter are connected
to the 12-bit digital multiplier. When the digital multiplier has a
value of all zeros at its inputs, the input signal is multiplied by
zero, producing zero scale. When the multiplier has a value of
all ones, the input signal is multiplied by a value of 4095/4096,
producing nearly full scale. There are 4,094 remaining
fractional multiplier values that produce output amplitudes
scaled according to their binary values.
AD9854
Rev. C | Page 29 of 52
12-BIT DIGITAL
MULTIPLIER
12 12
(BYPASS MULTIPLIER)
OSK EN = 0
OSK EN = 1
OSK EN = 0
OSK EN = 1
12
12
DIGITAL
SIGNAL IN
USER-PROGRAMMABLE
12-BIT Q-CHANNEL
MULTIPLIER
"OUTPUT SHAPE
KEY Q MULT"
REGISTER
12 OSK INT = 1
OSK INT = 0
18-BIT RAMP
RATE
COUNTER
SYSTEM
CLOCK
SHAPED ON/OFF
KEYING PIN
SINE DAC
12-BIT
UP/DOWN
COUNTER
DDS DIGITAL
OUTPUT
00636-B-050
Figure 50. Block Diagram of Q-Pathway of the Digital Multiplier Section Responsible for Shaped-Keying Function
The two fixed elements of the transition time are the period of
the system clock (which drives the ramp rate counter) and the
number of amplitude steps (4096). To give an example, assume
that the system clock of the AD9854 is 100 MHz (10 ns period).
If the ramp rate counter is programmed for a minimum count
of three, it takes two system clock periods (one rising edge loads
the count-down value, the next edge decrements the counter
from three to two). If the count-down value is less than three,
the ramp rate counter stalls and, therefore, produces a constant
scaling value to the digital multipliers. This stall condition may
have application by the user.
The relationship of the 8-bit count-down value to the time
period between output pulses is given as
(N + 1) × System Clock Period
where N is the 8-bit count-down value.
It takes 4096 of these pulses to advance the 12-bit up-counter
from zero scale to full scale. Therefore, the minimum shaped
keying ramp time for a 100 MHz system clock is
4096 × 4 × 10 ns = approximately 164 µs.
The maximum ramp time is 4096 × 256 × 10 ns ≈ 10.5 ms.
Finally, changing the logic state of Pin 30, shaped keying,
automatically performs the programmed output envelope
functions when OSK INT is high. A logic high on Pin 30 causes
the outputs to linearly ramp up to full-scale amplitude and to
hold until the logic level is changed to low, causing the outputs
to ramp down to zero scale.
I AND Q DACS
The sine and cosine outputs of the DDS drive the Q and I
DACs, respectively (300 MSPS maximum). Their maximum
output amplitudes are set by the DAC RSET resistor at Pin 56.
These are current-out DACs with a full-scale maximum output
of 20 mA; however, a nominal 10 mA output current provides
best spurious-free dynamic range (SFDR) performance. The
value of RSET = 39.93/IOUT, where IOUT is in amps. DAC output
compliance specifications limit the maximum voltage developed
at the outputs to −0.5 V to +1 V. Voltages developed beyond this
limitation cause excessive DAC distortion and possibly perma-
nent damage. The user must choose a proper load impedance to
limit the output voltage swing to the compliance limits. Both
DAC outputs should be terminated equally for best SFDR,
especially at higher output frequencies where harmonic
distortion errors are more prominent.
Both DACs are preceded by inverse SIN(x)/x filters (also called
inverse sinc filters) that precompensate for DAC output ampli-
tude variations over frequency to achieve flat amplitude
response from dc to Nyquist. Both DACs can be powered down
by setting the DAC PD bit high (Address 1D of the control
register) when not needed. I DAC outputs are designated as
IOUT1 and IOUT1, Pins 48 and 49, respectively. Q DAC
outputs are designated as IOUT2 and IOUT2, Pins 52 and 51,
respectively.
CONTROL DAC
The 12-bit Q DAC can be reconfigured to perform as a control
or auxiliary DAC. The control DAC output can provide dc
control levels to external circuitry, generate ac signals, or enable
duty cycle control of the on-board comparator. When the SRC
Q DAC bit in the control register (Parallel Address 1F hex) is
set high, the Q DAC inputs are switched from internal 12-bit Q
data source (default setting) to external 12-bit, twos comple-
ment data, supplied by the user. Data is channeled through
the serial or parallel interface to the 12-bit Q DAC register
(Address 26 and 27 hex) at a maximum 100 MHz data rate. This
DAC is clocked at the system clock, 300 MSPS (maximum), and
has the same maximum output current capability as that of the
I DAC. The single RSET resistor on the AD9854 sets the full-
scale output current for both DACs. The control DAC can be
separately powered down for power conservation when not
needed by setting the Q DAC power-down bit high (Address
1D hex). Control DAC outputs are designated as IOUT2 and
IOUT2 (Pins 52 and 51, respectively).
AD9854
Rev. C | Page 30 of 52
INVERSE SINC FUNCTION
The inverse sinc function precompensates input data to both
DACs for the SIN(x)/x roll-off characteristic inherent in the
DAC’s output spectrum. This allows wide bandwidth signals
(such as QPSK) to be output from the DACs without appre-
ciable amplitude variations as a function of frequency. The
inverse SINC function may be bypassed to significantly reduce
power consumption, especially at higher clock speeds. When
the Q DAC is configured as a control DAC, the inverse SINC
function does not apply to the Q path.
Inverse SINC is engaged by default and is bypassed by bringing
the Bypass Inv SINC bit high in Control Register 20 (hex), as
shown in Table 7.
REFCLK MULTIPLIER
The REFCLK multiplier is a programmable PLL-based refer-
ence clock multiplier that allows the user to select an integer
clock multiplying value over the range of 4× to 20×. With this
function, users can input as little as 15 MHz at the REFCLK
input to produce a 300 MHz internal system clock. Five bits
in Control Register 1E hex set the multiplier value, as detailed
in Table 6.
The REFCLK multiplier function can be bypassed to allow
direct clocking of the AD9854 from an external clock source.
The system clock for the AD9854 is either the output of the
REFCLK multiplier (if it is engaged) or the REFCLK inputs.
REFCLK may be either a single-ended or differential input by
setting Pin 64, DIFF CLK ENABLE, low or high, respectively.
FREQUENCY NORMALIZED TO SAMPLE RATE
4.0
0 0.1
–0.5
0
dB
3.5
3.0
2.5
2.0
1.5
1.0
0.5
–1.0
–1.5
–2.0
–2.5
–3.0
–3.5
–4.0 0.2 0.3 0.4 0.5
SYSTEM
ISF
SINC
00636-B-051
Figure 51. Inverse SINC Filter Response
PLL Range Bit
The PLL range bit selects the frequency range of the REFCLK
multiplier PLL. For operation from 200 MHz to 300 MHz
(internal system clock rate), the PLL range bit should be set to
Logic 1. For operation below 200 MHz, the PLL range bit
should be set to Logic 0. The PLL range bit adjusts the PLL loop
parameters for best phase noise performance within each range.
Pin 61, PLL FILTER
This pin provides the connection for the external zero compen-
sation network of the PLL loop filter. The zero compensation
network consists of a 1.3 kΩ resistor in series with a 0.01 µF
capacitor. The other side of the network should be connected as
close as possible to Pin 60, AVDD. For optimum phase noise
performance, the clock multiplier can be bypassed by setting
the bypass PLL bit in Control Register Address 1E.
Differential REFCLK Enable
A high level on this pin enables the differential clock inputs,
REFCLK and REFCLK (Pins 69 and 68, respectively). The
minimum differential signal amplitude required is 400 mV p-p,
at the REFCLK input pins. The center point or common-mode
range of the differential signal can range from 1.6 V to 1.9 V.
When Pin 64 (DIFF CLK ENABLE) is tied low, REFCLK
(Pin 69) is the only active clock input. This is referred to as
single-ended mode. In this mode, Pin 68 (REFCLK) should
be tied low or high.
High Speed Comparator—This comparator is optimized for
high speed, a >300 MHz toggle rate, low jitter, sensitive input,
built-in hysteresis, and an output level of 1 V p-p minimum into
50 Ω or CMOS logic levels into high impedance loads. The
comparator can be separately powered down to conserve power.
This comparator is used in clock generator applications to
square up the filtered sine wave generated by the DDS.
Power-Down—Several individual stages may be powered down
to reduce power consumption via the programming registers
while still maintaining functionality of desired stages. These
stages are identified in Table 7, Address 1D hex. Power-down is
achieved by setting the specified bits to logic high. A logic low
indicates that the stages are powered up.
Furthermore, and perhaps most significantly, the inverse sinc
filters and the digital multiplier stages, can be bypassed to
achieve significant power reduction through programming of
the control registers in Address 20 hex. Again, logic high causes
the stage to be bypassed. Of particular importance is the inverse
sinc filter; this stage consumes a significant amount of power.
A full power-down occurs when all four PD bits in Control
Register 1D hex are set to logic high. This reduces power
consumption to approximately 10 mW (3 mA).
.
AD9854
Rev. C | Page 31 of 52
Table 6. REFCLK Multiplier Control Register Values
Mult Value Ref Mult Bit 4 Ref Mult Bit 3 Ref Mult Bit 2 Ref Mult Bit 1 Ref Mult Bit 0
4 0 0 1 0 0
5 0 0 1 0 1
6 0 0 1 1 0
7 0 0 1 1 1
8 0 1 0 0 0
9 0 1 0 0 1
10 0 1 0 1 0
11 0 1 0 1 1
12 0 1 1 0 0
13 0 1 1 0 1
14 0 1 1 1 0
15 0 1 1 1 1
16 1 0 0 0 0
17 1 0 0 0 1
18 1 0 0 1 0
19 1 0 0 1 1
20 1 0 1 0 0
AD9854
Rev. C | Page 32 of 52
PROGRAMMING THE AD9854
The AD9854 register layout, shown in Table 7, contains the
information that programs the chip for the desired function-
ality. While many applications require very little programming
to configure the AD9854, some make use of all 12 accessible
register banks. The AD9854 supports an 8-bit parallel I/O
operation or an SPI-compatible serial I/O operation. All
accessible registers can be written and read back in either I/O
operating mode.
S/P SELECT, Pin 70, is used to configure the I/O mode. Systems
that use the parallel I/O mode must connect the S/P SELECT
pin to VDD. Systems that operate in the serial I/O mode must tie
the S/P SELECT pin to GND.
Regardless of mode, the I/O port data is written to a buffer
memory that does NOT affect operation of the part until the
contents of the buffer memory are transferred to the register
banks. The transfer of information occurs synchronously to the
system clock in one of two ways:
1. Internally, at a rate programmable by the user.
2. Externally, by the user. I/O operations can occur in the
absence of REFCLK but the data cannot be moved from
the buffer memory to the register bank without REFCLK.
See the Internal and External Update Clock section of this
document for detail.
Master RESET—Logic high active, must be held high for a
minimum of 10 system clock cycles. This causes the
communications bus to be initialized and to load the default
values listed in the Internal and External Update Clock section.
Table 7. Register Layout
Parallel
Address
Serial
Address AD9854 Register Layout
Hex Hex Bit 7 Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Bit 0
Default
Value
00 0 Phase Adjust Register No. 1 <13:8> (Bits 15, 14, don’t care) Phase 1 00h
01 Phase Adjust Register No. 1 <7:0> 00h
02 1 Phase Adjust Register No. 2 <13:8:> (Bits 15, 14, don’t care) Phase 2 00h
03 Phase Adjust Register No. 2 <7:0> 00h
04 2 Frequency Tuning Word 1 <47:40> Frequency 1 00h
05 Frequency Tuning Word 1 <39:32> 00h
06 Frequency Tuning Word 1 <31:24> 00h
07 Frequency Tuning Word 1 <23:16> 00h
08 Frequency Tuning Word 1 <15:8> 00h
09 Frequency Tuning Word 1 <7:0> 00h
0A 3 Frequency Tuning Word 2 <47:40> Frequency 2
0B Frequency Tuning Word 2 <39:32> 00h
0C Frequency Tuning Word 2 <31:24> 00h
0D Frequency Tuning Word 2 <23:16> 00h
0E Frequency Tuning Word 2 <15:8> 00h
0F Frequency Tuning Word 2 <7:0> 00h
10 4 Delta Frequency Word <47:40> 00h
11 Delta Frequency Word <39:32> 00h
12 Delta Frequency Word <31:24> 00h
13 Delta Frequency Word <23:16> 00h
14 Delta Frequency Word <15:8> 00h
15 Delta Frequency Word <7:0> 00h
16 5 Update Clock <31:24> 00h
17 Update Clock <23:16> 00h
18 Update Clock <15:8> 00h
19 Update Clock <7:0> 40h
1A 6 Ramp Rate Clock <19:16> (Bits 23, 22, 21, 20, don’t care) 00h
1B Ramp Rate Clock <15:8> 00h
1C Ramp Rate Clock <7:0> 00h
AD9854
Rev. C | Page 33 of 52
Parallel
Address
Serial
Address AD9854 Register Layout
Hex Hex Bit 7 Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Bit 0
Default
Value
1D 7 Don’t Care
CR [31]
Don’t Care Don’t Care Comp PD Reserved,
Always Low
QDAC PD DAC
PD
DIG PD 10h
1E Don’t Care PLL Range Bypass PLL Ref Mult 4 Ref Mult 3 Ref Mult 2 Ref
Mult 1
Ref Mult 0 64h
1F CLR ACC 1 CLR ACC 2 Triangle SRC QDAC Mode 2 Mode 1 Mode
0
INT/EXT
Update Clk
01h
20
Don’t Care Bypass Inv
Sinc
OSK EN OSK INT Don’t Care Don’t Care LSB
First
SDO Active
CR [0]
20h
21 8 Output Shape Key I Mult <11:8> (Bits 15, 14, 13, 12 don’t care) 00h
22 Output Shape Key I Mult <7:0> 00h
23 9 Output Shape Key Q Mult <11:8> (Bits 15, 14, 13, 12 don’t care) 00h
24 Output Shape Key Q Mult <7:0> 00h
25 A Output Shape Key Ramp Rate <7:0> 80h
26 B 00h
27
QDAC <11:8> (Bits 15, 14, 13, 12 don’t care)
QDAC <7:0> (Data is required to be in twos complement format) 0
PARALLEL I/O OPERATION
With the S/P SELECT pin tied high, the parallel I/O mode is
active. The I/O port is compatible with industry-standard DSPs
and microcontrollers. Six address bits, eight bidirectional data
bits, and separate write/read control inputs make up the I/O
port pins.
Parallel I/O operation allows write access to each byte of any
register in a single I/O operation up to 1/10.5 ns. Readback
capability for each register is included to ease designing with
the AD9854. (Reads are not guaranteed at 100 MHz, as they are
intended for software debugging only.)
Parallel I/O operation timing diagrams are shown in Figure 52
and Figure 53.
SERIAL PORT I/O OPERATION
With the S/P SELECT pin tied low, the serial I/O mode is active.
The serial port is a flexible, synchronous serial communications
port allowing easy interface to many industry-standard micro-
controllers and microprocessors. The serial I/O is compatible
with most synchronous transfer formats, including both the
Motorola® 6905/11 SPI and Intel® 8051 SSR protocols. The inter-
face allows read/write access to all 12 registers that configure
the AD9854 and can be configured as a single-pin I/O (SDIO)
or two unidirectional pins for in/out (SDIO/SDO). Data trans-
fers are supported in most significant bit (MSB) first format or
least significant bit (LSB) first format at up to 10 MHz.
When configured for serial I/O operation, most AD9854
parallel port pins are inactive; some are used for the serial I/O.
Table 8 describes pin requirements for serial I/O.
Note that when operating in serial I/O mode, it is best to use the
external I/O update clock mode to avoid an I/O update CLK
during a serial communication cycle. Such an occurrence could
cause incorrect programming due to partial data transfer. To
exit the default internal update mode, program the device for
external update operation at power-up, before starting the
REFCLK signal, but after a master reset. Starting the REFCLK
causes this information to transfer to the register bank, putting
the device in external update mode.
Table 8. Serial I/O Pin Requirements
Pin Number Pin Name Serial I/O Description
1, 2, 3, 4, 5, 6, 7, 8 D[7:0] The parallel data pins are not active; tie to VDD or GND.
14, 15, 16 A[5:3] The parallel address Pins A5, A4, A3 are not active; tie to VDD or GND.
17 A2 IO RESET.
18 A1 SDO.
19 A0 SDIO.
20 I/O UD CLOCK Update Clock. Same functionality for serial mode as parallel mode.
21 WR SCLK.
22 RD CS—Chip Select
AD9854
Rev. C | Page 34 of 52
A<5:0>
D<7:0>
A1
D1
A2
D2
A3
D3
T
RDHOZ
T
RDLOV
T
AHD
T
ADV
SPECIFICATION VALUE DESCRIPTION
T
ADV
T
AHD
T
RDLOV
T
RDHOZ
15ns
5ns
15ns
10ns
ADDRESS TO DATA VALID TIME (MAXIMUM)
ADDRESS HOLD TIME TO RD SIGNAL INACTIVE (MINIMUM)
RD LOW TO OUTPUT VALID (MAXIMUM)
RD HIGH TO DATA THREE-STATE (MAXIMUM)
RD
00636-B-052
Figure 52. Parallel Port Read Timing Diagram
D<7:0> D1 D2 D3
SPECIFICATION VALUE DESCRIPTION
T
ASU
T
DSU
T
ADH
T
DHD
8.0ns
3.0ns ADDRESS SETUP TIME TO WR SIGNAL ACTIVE
DATA SETUP TIME TO WR SIGNAL ACTIVE
0ns
0ns ADDRESS HOLD TIME TO WR SIGNAL INACTIVE
DATA HOLD TIME TO WR SIGNAL INACTIVE
T
WRLOW
T
WRHIGH
T
WR
2.5ns WR SIGNAL MINIMUM LOW TIME
7ns
10.5ns WR SIGNAL MINIMUM HIGH TIME
MINIMUM WRITE TIME
A<5:0> A1 A2 A3
T
ASU
T
AHD
T
WRHIGH
T
WRLOW
T
DHD
T
DSU
T
WR
WR
00636-B-053
Figure 53. Parallel Port Write Timing Diagram
AD9854
Rev. C | Page 35 of 52
GENERAL OPERATION OF THE SERIAL INTERFACE
There are two phases to a serial communication cycle with the
AD9854. Phase 1 is the instruction cycle, which is the writing of
an instruction byte into the AD9854, coincident with the first
eight SCLK rising edges. The instruction byte provides the
AD9854 serial port controller with information regarding the
data transfer cycle, which is Phase 2 of the communication
cycle. The Phase 1 instruction byte defines whether the
upcoming data transfer is a read or write, and the register
address to be acted upon.
The first eight SCLK rising edges of each communication cycle
are used to write the instruction byte into the AD9854. The
remaining SCLK edges are for Phase 2 of the communication
cycle. Phase 2 is the actual data transfer between the AD9854
and the system controller. The number of data bytes transferred
in Phase 2 of the communication cycle is a function of the
register address. (Table 9 describes how many bytes must be
transferred.) The AD9854 internal serial I/O controller expects
every byte of the register being accessed to be transferred. Thus,
the user would want to write between I/O update clocks.
At the completion of any communication cycle, the AD9854
serial port controller expects the next eight rising SCLK edges
to be the instruction byte of the next communication cycle.
In addition, an active high input on the IO RESET pin immedi-
ately terminates the current communication cycle. After IO
RESET returns low, the AD9854 serial port controller requires
the next eight rising SCLK edges to be the instruction byte of
the next communication cycle.
All data input to the AD9854 is registered on the rising edge of
SCLK. All data is driven out of the AD9854 on the falling edge
of SCLK.
Figure 54 and Figure 55 show the general operation of the
AD9854 serial port.
Table 9. Register Address vs. Data Bytes Transferred
Serial Register Address Register Name Bytes Transferred
0 Phase Offset Tuning Word Register No. 1 2
1 Phase Offset Tuning Word Register No. 2 2
2 Frequency Tuning Word No. 1 6
3 Frequency Tuning Word No. 2 6
4 Delta Frequency Register 6
5 Update Clock Rate Register 4
6 Ramp Rate Clock Register 3
7 Control Register 4
8 I Path Digital Multiplier Register 2
9 Q Path Digital Multiplier Register 2
A Shaped On/Off Keying Ramp Rate Register 1
B Q DAC Register 2
INSTRUCTION
CYCLE DATA TRANSFER
INSTRUCTION
BYTE DATA BYTE 1 DATA BYTE 2 DATA BYTE 3
SDIO
00636-B-054
CS
Figure 54. Using SDIO as a Read/Write Transfer
INSTRUCTION
CYCLE DATA TRANSFER
INSTRUCTION
BYTE
S
DIO
DATA TRANSFER
DATA BYTE 1 DATA BYTE 2 DATA BYTE 3
SDO
00636-B-055
CS
Figure 55. Using SDIO as an Input, SDO as an Output
AD9854
Rev. C | Page 36 of 52
INSTRUCTION BYTE
The instruction byte contains the following information.
Table 10. Instruction Byte Information
MSB D6 D5 D4 D3 D2 D1 LSB
R/WX X X A3 A2 A1 A0
R/W—Bit 7 determines whether a read or write data transfer
occurs following the instruction byte. Logic high indicates read
operation. Logic 0 indicates a write operation.
Bits 6, 5, and 4 are dummy bits (don’t care).
A3, A2, A1, A0—Bits 3, 2, 1, and 0 determine which register is
accessed during the data transfer portion of the communica-
tions cycle. See Table 7 for register address details.
SERIAL INTERFACE PORT PIN DESCRIPTIONS
SCLK
Serial Clock (Pin 21). The serial clock pin is used to
synchronize data to and from the AD9854 and to run the
internal state machines. SCLK maximum frequency is 10 MHz.
CS
Chip Select (Pin 22). Active low input that allows more than
one device on the same serial communications line. The SDO
and SDIO pins go to a high impedance state when this input is
high. If driven high during any communications cycle, that
cycle is suspended until CS is reactivated low. Chip Select can
be tied low in systems that maintain control of SCLK.
SDIO
Serial Data I/O (Pin 19). Data is always written to the AD9854
on this pin. However, this pin can be used as a bidirectional
data line. The configuration of this pin is controlled by Bit 0 of
Register Address 20h. The default is Logic 0, which configures
the SDIO pin as bidirectional.
SDO
Serial Data Out (Pin 18). Data is read from this pin for proto-
cols that use separate lines for transmitting and receiving data.
In the case where the AD9854 operates in a single bidirectional
I/O mode, this pin does not output data and is set to a high
impedance state.
IO RESET
Synchronize I/O Port (Pin 17). Synchronizes the I/O port state
machines without affecting the contents of the addressable
registers. An active high input on the IO RESET pin causes the
current communication cycle to terminate. After IO reset
returns low (Logic 0), another communication cycle may begin,
starting with the instruction byte.
NOTES ON SERIAL PORT OPERATION
The AD9854 serial port configuration bits reside in Bit 1 and
Bit 0 of Register Address 20h. It is important to note that the
configuration changes immediately upon a valid I/O update.
For multibyte transfers, writing to this register may occur
during the middle of a communication cycle. Care must be
taken to compensate for this new configuration for the
remainder of the current communication cycle.
The system must maintain synchronization with the AD9854,
or the internal control logic is not able to recognize further
instructions. For example, if the system sends the instruction to
write a 2-byte register, then pulses the SCLK pin for a 3-byte
register (24 additional SCLK rising edges), communication
synchronization is lost. In this case, the first 16 SCLK rising
edges after the instruction cycle will properly write the first two
data bytes into the AD9854, but the next eight rising SCLK
edges are interpreted as the next instruction byte, not the final
byte of the previous communication cycle.
In the case where synchronization is lost between the system
and the AD9854, the IO RESET pin provides a means to
reestablish synchronization without reinitializing the entire
chip. Asserting the IO RESET pin (active high) resets the
AD9854 serial port state machine, terminating the current I/O
operation and putting the device into a state in which the next
eight SCLK rising edges are understood to be an instruction
byte. The IO RESET pin must be deasserted (low) before the
next instruction byte write can begin. Any information that is
written to the AD9854 registers during a valid communication
cycle prior to loss of synchronization remains intact.
SCLK
SDIO
T
PRE
T
DSU
T
SCLKPWH
T
SCLKPWL
T
SCLK
T
DHLD
2ND BIT1ST BIT
SYMBOL MIN DEFINITION
CS SETUP TIME
PERIOD OF SERIAL DATA CLOCK
SERIAL DATA SETUP TIME
SERIAL DATA CLOCK PULSE WIDTH HIGH
SERIAL DATA CLOCK PULSE WIDTH LOW
SERIAL DATA HOLD TIME
T
PRE
T
SCLK
T
DSU
T
SCLKPWH
T
SCLKPWL
T
DHLD
30ns
100ns
30ns
40ns
40ns
0ns
00636-B-056
CS
Figure 56. Timing Diagram for Data Write to AD9854
INSTRUCTION
CYCLE DATA TRANSFER
INSTRUCTION
BYTE
SDIO
DATA TRANSFER
DATA BYTE 1 DATA BYTE 2 DATA BYTE 3
SDO
CS
00636-B-057
Figure 57. Timing Diagram for Read from AD9854
AD9854
Rev. C | Page 37 of 52
MSB/LSB TRANSFERS
The AD9854 serial port can support both most significant bit
(MSB) first or least significant bit (LSB) first data formats. This
functionality is controlled by Bit 1 of Serial Register Bank 20h.
When this bit is set active high, the AD9854 serial port is in
LSB-first format. This bit defaults low, to the MSB-first format.
The instruction byte must be written in the format indicated by
Bit 1 of Serial Register Bank 20h. That is, if the AD9854 is in
LSB-first mode, the instruction byte must be written from least
significant bit to most significant bit.
CONTROL REGISTER DESCRIPTION
The control register is located in the shaded portion of Table 7
at Address 1D through 20 hex. It is composed of 32 bits. Bit 31
is located at the top left position and Bit 0 is located in the lower
right position of the shaded portion. In the text that follows, the
register descriptions have been subdivided to make it easier to
locate the text associated with specific control categories.
CR[31:29] are open.
CR[28] is the comparator power-down bit. When this bit is set
(Logic 1), its signal tells the comparator that a power-down
mode is active. This bit is an output of the digital section and is
an input to the analog section.
CR[27] must always be written to Logic 0. Writing this bit to
Logic 1 causes the AD9854 to stop working until a master reset
is applied.
CR[26] is the Q DAC power-down bit. When this bit is set
(Logic 1), it tells the Q DAC that a power-down mode is active.
CR[25] is the full DAC power-down bit. When this bit is set
(Logic 1), it tells both the I and Q DACs, as well as the
reference, that a power-down mode is active.
CR[24] is the digital power-down bit. When this bit is set
(Logic 1), this signal indicates to the digital section that a
power-down mode is active. Within the digital section, the
clocks are forced to dc, effectively powering down the digital
section. The PLL still accepts the REFCLK signal and continues
to output the higher frequency.
CR[23] is reserved. Write to zero.
CR[22] is the PLL range bit, which controls the VCO gain. The
power-up state of the PLL range bit is Logic 1, higher gain for
frequencies above 200 MHz.
CR[21] is the bypass PLL bit, active high. When this bit is
active, the PLL is powered down and the REFCLK input is used
to drive the system clock signal. The power-up state of the
bypass PLL bit is Logic 1, PLL bypassed.
CR[20:16] bits are the PLL multiplier factor. These bits are the
REFCLK multiplication factor, unless the bypass PLL bit is set.
The PLL multiplier valid range is from 4 to 20, inclusive.
CR[15] is the Clear Accumulator 1 bit. This bit has a one-shot
type function. When this bit is written active, Logic 1, a Clear
Accumulator 1 signal is sent to the DDS logic, resetting the
accumulator value to zero. The bit is then automatically reset,
but the buffer memory is not reset. This bit allows the user to
easily create a saw-toothed frequency sweep pattern with
minimal intervention. This bit is intended for chirp mode only,
but its function is still retained in other modes.
CR[14] is the clear accumulator bit. This bit, active high, holds
both the Accumulator 1 and Accumulator 2 values at zero for as
long as the bit is active. This allows the DDS phase to be
initialized via the I/O port.
CR[13] is the triangle bit. When this bit is set, the AD9854
automatically performs a continuous frequency sweep from F1
to F2 frequencies and back. The effect is a triangular frequency
sweep. When this bit is set, the operating mode must be set to
ramped FSK.
CR[12] is the source Q DAC bit. When this bit is set high, the Q
path DAC accepts data from the Q DAC register.
CR[11:9] are the three bits that describe the five operating
modes of the AD9854:
0h = Single-Tone mode
1h = FSK mode
2h = Ramped FSK mode
3h = Chirp mode
4h = BPSK mode
SDIO D
7
I
7
S
CLK
INSTRUCTION CYCLE DATA TRANSFER CYCLE
I
6
I
5
I
4
I
3
I
0
I
2
I
1
D
6
D
5
D
4
D
3
D
2
D
1
D
0
CS
00636-B-058
Figure 58. Serial Port Write Timing-Clock Stall low
AD9854
Rev. C | Page 38 of 52
SDIO
D
O 7
D
O 6
D
O 5
D
O 4
D
O 3
D
O 2
D
O 1
D
O 0
S
CLK
INSTRUCTION CYCLE
DON'T CARE
SDO
DATA TRANSFER CYCLE
I
7
I
6
I
5
I
4
I
3
I
0
I
2
I
1
CS
00636-B-059
Figure 59. 3-Wire Serial Port Read Timing-Clock Stall Low
D
7
D
6
D
5
D
4
D
3
D
2
D
1
D
0
SDIO
SCLK
INSTRUCTION CYCLE DATA TRANSFER CYCLE
I
7
I
6
I
5
I
4
I
3
I
0
I
2
I
1
CS
00636-B-060
Figure 60. Serial Port Write Timing-Clock Stall High
I
7
I
6
I
5
I
4
I
3
I
0
I
2
I
1
SDIO
SCLK
INSTRUCTION CYCLE DATA TRANSFER CYCLE
D
O 7
D
O 6
D
O 5
D
O 4
D
O 3
D
O 2
D
O 1
D
O 0
CS
00636-B-061
Figure 61. 2-Wire Serial Port Read Timing-Clock Stall High
CR[8] is the internal update active bit. When this bit is set to
Logic 1, the I/O UD CLK pin is an output and the AD9854
generates the I/O UD CLK signal. When set to Logic 0, external
I/O UD CLK functionality is performed and the I/O UD CLK
pin is configured as an input.
CR[7] is reserved. Write to zero.
CR[6] is the inverse sinc filter bypass bit. When this bit is set,
the data from the DDS block goes directly to the output shaped-
keying logic and the clock to the inverse sinc filter is stopped.
Default is clear, filter enabled.
CR[5] is the shaped-keying enable bit. When this bit is set, the
output ramping function is enabled and is performed in
accordance with the CR[4] bit requirements.
CR[4] is the internal/external output shaped-keying control bit.
When this bit is set to Logic 1, the shaped-keying factor is
internally generated and applied to both the I and Q paths.
When cleared (default), the output shaped-keying function is
externally controlled by the user and the shaped-keying factor is
the I and Q output shaped-keying factor register value. The two
registers that are the shaped-keying factors also default low such
that the output is off at power-up and until the device is
programmed by the user.
CR[3:2] are reserved. Write to zero.
CR[1] is the serial port MSB/LSB first bit. Defaults low,
MSB first.
CR[0] is the serial port SDO active bit. Defaults low, inactive.
AD9854
Rev. C | Page 39 of 52
POWER DISSIPATION AND THERMAL CONSIDERATIONS
The AD9854 is a multifunctional, high speed device that targets
a wide variety of synthesizer and agile clock applications. The
numerous innovative features contained in the device each
consume incremental power. If enabled in combination, the safe
thermal operating conditions of the device may be exceeded.
Careful analysis and consideration of power dissipation and
thermal management is a critical element in the successful
application of the AD9854 device.
The AD9854 device is specified to operate within the industrial
temperature range of −40°C to +85°C. This specification is
conditional, however, such that the absolute maximum junction
temperature of 150°C is not exceeded. At high operating
temperatures, extreme care must be taken in the operation of
the device to avoid exceeding the junction temperature, which
results in a potentially damaging thermal condition.
Many variables contribute to the operating junction tempera-
ture within the device, including:
• Package style
• Selected mode of operation
• Internal system clock speed
• Supply voltage
• Ambient temperature
The combination of these variables determines the junction
temperature within the AD9854 for a given set of operating
conditions.
The AD9854 is available in two package styles: a thermally
enhanced, surface-mount package with an exposed heat sink,
and a nonthermally enhanced, surface-mount package. The
thermal impedance of these packages is 16°C/W and 38°C/W,
respectively, measured under still-air conditions.
THERMAL IMPEDANCE
The thermal impedance of a package can be thought of as a
thermal resistor that exists between the semiconductor surface
and the ambient air. The thermal impedance is determined by
the package material and the physical dimensions of the
package. The dissipation of the heat from the package is directly
dependent upon the ambient air conditions and the physical
connection made between the IC package and the PCB.
Adequate dissipation of power from the AD9854 relies on all
power and ground pins of the device being soldered directly to
a copper plane on a PCB. In addition, the thermally enhanced
package of the AD9854ASQ contains a heat sink on the bottom
of the package that must be soldered to a ground pad on the
PCB surface. This pad must be connected to a large copper
plane which, for convenience, may be a ground plane. Sockets
for either package style of the device are not recommended.
JUNCTION TEMPERATURE CONSIDERATIONS
The power dissipation (PDISS) of the AD9854 in a given
application is determined by many operating conditions. Some
of the conditions have a direct relationship with PDISS, such as
supply voltage and clock speed, but others are less deterministic.
The total power dissipation within the device, and its effect on
the junction temperature, must be considered when using the
device. The junction temperature of the device is given by
(Thermal Impedance × Power Consumption) + Ambient Temperature.
Given that the junction temperature should never exceed
150°C, and that the ambient temperature can be 85°C, the
maximum power consumption for the AD9854AST is 1.7 W
and 4.1 W for the AD9854ASQ (thermally-enhanced package).
Factors affecting the power dissipation follow.
Supply Voltage
The supply voltage affects power dissipation and junction
temperature because PDISS equals V × I. Users should design for
3.3 V nominal; however, the device is guaranteed to meet
specifications over the full temperature range and over the
supply voltage range of 3.135 V to 3.465 V.
Clock Speed
Clock speed directly and linearly influences the total power
dissipation of the device, and, therefore, junction temperature.
As a rule, to minimize power dissipation, the user should always
select the lowest internal clock speed possible to support a given
application. Typically, the usable frequency output bandwidth
from a DDS is limited to 40% of the clock rate to keep reason-
able requirements on the output low-pass filter. For the typical
DDS application, the system clock frequency should be 2.5× the
highest desired output frequency.
Mode of Operation
The selected operational mode of operation of the AD9854 has
a great influence on the total power consumption. The AD9854
offers many features and modes, each of which imposes an
additional power requirement. Though the collection of
features contained in the AD9854 target a wide variety of
applications, the device was designed under the assumption that
only a few features would be enabled for any given application.
In fact, enabling multiple features at higher clock speeds may
cause the maximum junction temperature of the die to be
exceeded. This can severely limit the long-term reliability of the
device. Figure 62 and Figure 63 show the power requirements
associated with the individual features of the AD9854. These
graphs should be used as a guide in determining the optimum
application of the AD9854 for reliable operation.
AD9854
Rev. C | Page 40 of 52
As can be seen in Figure 63, the inverse sinc filter function
requires a significant amount of power. As an alternate
approach to maintaining flatness across the output bandwidth,
the digital multiplier function may be used to adjust the output
signal level, at a dramatic savings in power consumption.
Careful planning and management of the feature set minimizes
power dissipation and avoids exceeding junction temperature
requirements within the IC.
Figure 62 shows the supply current consumed by the AD9854
over a range of frequencies for two possible configurations. All
circuits enabled means the output scaling multipliers, the
inverse sinc filter, the Q DAC, and the on-board comparator are
all enabled, while basic configuration means the output scaling
multipliers, the inverse sinc filter, the Q DAC, and the on-board
comparator are all disabled.
FREQUENCY (MHz)
1400
20
SUPPLY CURRENT (mA)
1200
1000
800
600
400
200
060 100 140 180 220 260 300
ALL CIRCUITS ENABLED
BASIC CONFIGURATION
00636-B-062
Figure 62. Current Consumption vs. Clock Frequency
Figure 63 shows the approximate current consumed by each of
four functions.
FREQUENCY (MHz)
20 60 100 140 180 220 260 300
450
SUPPLY CURRENT (mA)
400
350
300
250
200
150
0
100
50
Q DAC
500
INVERSE SINC FILTER
OUTPUT SCALING
MULTIPLIERS
COMPARATOR
00636-B-063
Figure 63. Current Consumption by Function vs. Clock frequency
EVALUATION OF OPERATING CONDITIONS
The first step in applying the AD9854 is to select the internal
clock frequency. Clock frequency selections above 200 MHz
require the use of the thermally enhanced package
(AD9854ASQ); clock frequency selections of 200 MHz and
below may allow the use of the standard plastic surface-mount
package, but more information is needed to make that
determination.
The second evaluation step is to determine the maximum
required operating temperature for the AD9854 in the given
application. Subtract this value from 150°C, which is the
maximum junction temperature allowed for the AD9854. For
the extended industrial temperature range, the maximum
operating temperature is 85°C, which results in a difference of
65°C. This is the maximum temperature gradient that the
device may experience due to power dissipation.
The third evaluation step is to divide this maximum temp-
erature gradient by the thermal impedance, to arrive at the
maximum power dissipation allowed for the application. For
the example so far, 65°C divided by both versions of the
AD9854 packages’ thermal impedances of 38°C/W and 16°C/W,
respectively, yields a total power dissipation limit of 1.7 W and
4.1 W, respectively. This means that for a 3.3 V nominal power
supply voltage, the current consumed by the device under full
operating conditions must not exceed 515 mA in the standard
plastic package and 1242 mA in the thermally enhanced
package. The total set of enabled functions and operating
conditions of the AD9854 application must support these
current consumption limits.
To determine the suitability of a given AD9854 application
vs. the power dissipation requirements, use Figure 62 and
Figure 63. These graphs assume that the AD9854 device is
soldered to a multilayer PCB per the recommended best
manufacturing practices and procedures for the given package
type. This ensures that the specified thermal impedance
specifications are achieved.
AD9854
Rev. C | Page 41 of 52
THERMALLY ENHANCED PACKAGE MOUNTING
GUIDELINES
This section provides general recommendations for mounting
the AD9854ASQ (the thermally enhanced exposed heat sink
package) to printed circuit boards. The exceptional thermal
characteristics of this package depend entirely upon proper
mechanical attachment.
Figure 64 shows the package from the bottom and the dimen-
sions of the exposed heat sink. A solid conduit of solder must be
established between this pad and the surface of the PCB.
C
O
U
N
T
R
Y
14mm10mm
00636-B-064
Figure 64. Package Bottom and Exposed Heat Sink
Figure 65 depicts a general PCB land pattern for an exposed
heat sink device. Note that this pattern is for a 64-lead device,
not an 80-lead, but the relative shapes and dimensions still
apply. In this land pattern, a solid copper plane exists inside the
individual lands for device leads. Also, the solder mask opening
is conservatively dimensioned to avoid any assembly problems.
SOLDER MASK
OPENING
THERMAL LAND
00636-B-065
Figure 65. General PCB Land Pattern
The thermal land itself must be able to distribute heat to an
even larger copper plane such as an internal ground plane.
Vias must be uniformly provided over the entire thermal pad
to connect to this internal plane. A proposed via pattern is
shown in Figure 66. Via holes should be small (12 mils, 0.3 mm)
such that they can be plated and plugged. These provide the
mechanical conduit for heat transfer.
00636-B-066
Figure 66. Proposed Via Pattern
Finally, a proposed stencil design is shown in Figure 67 for
screen solder placement. Note that if vias are not plugged,
wicking occurs, which displaces solder away from the exposed
heat sink, and the necessary mechanical bond is not established.
00636-B-067
Figure 67. Proposed Stencil Design for Screen Solder Placement
AD9854
Rev. C | Page 42 of 52
EVALUATION BOARD
An evaluation board package is available for AD9854 DDS
devices. This package consists of a PCB, software, and
documentation to facilitate bench analysis of the device’s
performance. To ensure optimum dynamic performance from
the device, it is recommended that AD9854 users familiarize
themselves with the operation and performance capabilities of
the device with the evaluation board and use the evaluation
board as a PCB reference design.
EVALUATION BOARD INSTRUCTIONS
The AD9852/AD9854 Revision E evaluation board includes
either an AD9852ASQ or AD9854ASQ IC.
The ASQ package permits 300 MHz operation by virtue of its
thermally enhanced design. This package has a bottom-side
heat slug that must be soldered to the ground plane of the PCB
directly beneath the IC. In this manner, the evaluation board
PCB ground plane layer extracts heat from the AD9852/
AD9854 IC package. If device operation is limited to 200 MHz
and below, the AST package without a heat slug may be used in
customer installations over the full temperature range. The AST
package is less expensive than the ASQ package and those costs
are reflected in the price of the IC.
Evaluation boards for both the AD9852 and AD9854 are
identical except for the installed IC.
To assist in proper placement of the pin header shorting-
jumpers, the instructions refer to direction (left, right, top,
bottom) as well as header pins to be shorted. Pin 1 for each
3-pin header has been marked on the PCB corresponding with
the schematic diagram. When following these instructions,
position the PCB so that the PCB text can be read from left to
right. The board is shipped with the pin headers configuring the
board as follows:
• REFCLK for the AD9852/AD9854 is configured as
differential. The differential clock signals are provided by
the MC100LVEL16D differential receiver.
• Input clock for the MC100LVEL16D is single-ended via
J25. This signal may be 3.3 V CMOS or a 2 V p-p sine wave
capable of driving 50 Ω (R13).
• Both DAC outputs from the AD9852/AD9854 are routed
through the two 120 MHz elliptical LP filters and their
outputs connected to J7 (Q or Control DAC) and J6
(I or Cosine DAC).
• The board is set up for software control via the printer port
connector.
• The DAC’s output currents are configured for 10 mA.
GENERAL OPERATING INSTRUCTIONS
Load the CD software on your PC’s hard disk. The current
software (Version 1.72) supports Windows® 95, Windows 98,
Windows 2000, Windows NT®, and Windows XP.
Connect a printer cable from the PC to the AD9854 evaluation
board printer port connector labeled J11.
Hardware preparation: Use the schematic in conjunction with
these instructions to become acquainted with the electrical
functioning of the evaluation board.
Attach power wires to connector labeled TB1 using the screw-
down terminals. This is a plastic connector that press-fits over a
4-pin header soldered to the board. Table 11 shows connections
to each pin. DUT = device under test.
Table 11. Power Requirements for DUT Pins
AVDD 3.3 V DVDD 3.3 V VCC 3.3 V Ground
For all DUT
analog pins
For all DUT
digital pins
For all other
devices
For all
devices
Attach REFCLK to the clock input, J25.
Clock Input, J25
This is a single-ended input that, for conversion to differential
PECL output, is routed to the MC100LVEL16D. This is accom-
plished by attaching a 2 V p-p clock or sine wave source to J25.
Note that this is a 50 Ω impedance point set by R13. The input
signal is ac-coupled and then biased to the center-switching
threshold of the MC100LVEL16D. To engage the differential
clocking mode of the AD9854, W3 Pins 2 and 3 (the bottom
two pins) must be connected with a shorting jumper.
The signal arriving at the AD9854 is called the reference clock.
When engaging the on-chip PLL clock multiplier, this signal is
the reference clock for the PLL and the multiplied PLL output
becomes the system clock. If the PLL clock multiplier is to be
bypassed, the reference clock supplied by the user is directly
operating the AD9854 and is, therefore, the system clock.
Three-State Control
Switch headers W9, W11, W12, W13, W14, and W15 must be
shorted to allow the provided software to control the AD9854
evaluation board via the printer port connector J11.
Programming
If programming of the AD9854 is not to be provided by the
user’s PC and ADI software, Headers W9, W11, W12, W13,
W14, and W15 should be opened (shorting jumpers removed).
This effectively detaches the PC interface and allows the 40-pin
headers, J10 and J1, to assume control without bus contention.
Input signals on J10 and J1 going to the AD9854 should be
3.3 V CMOS logic levels.
AD9854
Rev. C | Page 43 of 52
Low-Pass Filter Testing
The purpose of 2-pin headers W7 and W10 (associated with J4
and J5) is to allow the two 50 Ω, 120 MHz filters to be tested
during PCB assembly without interference from other circuitry
attached to the filter inputs. Typically, a shorting jumper is
attached to each header to allow the DAC signals to be routed
to the filters. If the user wishes to test the filters, the shorting
jumpers at W7 and W10 should be removed and 50 Ω test
signals applied at J4 and J5 inputs to the 50 Ω elliptic filters.
Users should refer to the schematic provided and the following
sections to properly position the remaining shorting jumpers.
Observing the Unfiltered IOUT1 and the Unfiltered
IOUT2 DAC Signals
The unfiltered DAC outputs may be observed at J5 (the I or
cosine signal) and J4 (the Q or Control DAC signal). The
procedure below simply routes the two 50 Ω terminated analog
DAC outputs to the SMB connectors and disconnects any other
circuitry. The raw DAC outputs may appear as a series of
quantized (stepped) output levels that may not resemble a sine
wave until they are filtered. The default 10 mA output current
develops a 0.5 V p-p signal across the on-board 50 Ω termin-
ation. If the observation equipment offers 50 Ω inputs, the DAC
develops only 0.25 V p-p due to the double termination.
1. Install shorting jumpers at W7 and W10.
2. Remove shorting jumper at W16.
3. Remove shorting jumper from 3-pin header W1.
4. Install shorting jumper on Pins 1 and 2 (bottom two pins)
of 3-pin header W4.
On the AD9852 evaluation board, IOUT2, the control DAC
output, is under user control through the serial or parallel ports.
The 12-bit, twos complement value(s) is/are written to the
control DAC register that sets the IOUT2 output to a static dc
level. Allowable hexadecimal values are 7FF (maximum) to 800
(minimum) with all zeros being midscale. Rapidly changing the
contents of the control DAC register (up to 100 MSPS) allows
IOUT2 to assume any waveform that can be programmed.
Observing the Filtered IOUT1 and the Filtered IOUT2
The filtered I and Q (or control) DAC outputs may be observed
at J6 (the I signal) and J7 (the Q or control signal). This places
the 50 Ω (input and output Z) low-pass filters in the I and Q
(or control) DAC pathways to remove images and aliased
harmonics and other spurious signals above approximately
120 MHz. These I and Q signals appear as nearly pure sine
waves and 90° out of phase with each other. These filters are
designed with the assumption that the system clock speed is at
or near maximum (300 MHz). If the system clock speed is
much less than 300 MHz, for example 200 MHz, it is possible
or inevitable that unwanted DAC products other than the
fundamental signal are passed by the low-pass filters.
If the AD9852 evaluation board is used, any reference to the Q
signal should be interpreted to mean the control DAC.
1. Install shorting jumpers at W7 and W10.
2. Install shorting jumper at W16.
3. Install shorting jumper on Pins 1 and 2 (bottom two pins)
of 3-pin header W1.
4. Install shorting jumper on Pins 1 and 2 (bottom two pins)
of 3-pin header W4.
5. Install shorting jumper on Pins 2 and 3 (bottom two pins)
of 3-pin header W2 and W8.
Observing the Filtered IOUT1 and the Filtered IOUT1
The filtered I DAC outputs may be observed at J6 (the true
signal) and J7 (the complementary signal). This places the
120 MHz low-pass filters in the true and complementary
output paths of the I DAC to remove images and aliased
harmonics and other spurious signals above approximately
120 MHz. These signals appear as nearly pure sine waves and
180 degrees out of phase with each other. If the system clock
speed is much less than 300 MHz, for example 200 MHz, it is
possible or inevitable that unwanted DAC products other than
the fundamental signal are passed by the low-pass filters.
1. Install shorting jumpers at W7 and W10.
2. Install shorting jumper at W16.
3. Install shorting jumper on Pins 2 and 3 (top two pins) of
3-pin header W1.
4. Install shorting jumper on Pins 2 and 3 (top two pins) of
3-pin header W4.
5. Install shorting jumpers on Pins 2 and 3 (bottom two pins)
of 3-pin header W2 and W8.
To Connect the High Speed Comparator
To connect the high speed comparator to the DAC output
signals, either the quadrature filtered output configuration
(AD9854 only) or the complementary filtered output config-
uration outlined in the previous section (for both the AD9854
and the AD9852) can be chosen. Follow Steps 1 through 4 for
either filtered configuration as described previously. Then
install a shorting jumper on Pins 1 and 2 (top two pins) of
3-pin header W2 and W8. This reroutes the filtered signals
away from their output connectors (J6 and J7) and to the 100 Ω
configured comparator inputs. This sets up the comparator for
differential input without control of the comparator output duty
cycle. The comparator output duty cycle should be close to 50%
in this configuration.
The user may elect to change the RSET resistor, R2, from 3.9 kΩ
to 1.95 kΩ to receive a more robust signal at the comparator
inputs. This decreases jitter and extends the comparator
operating range. This may be accomplished by installing a
shorting jumper at W6, which provides a second 3.9 kΩ chip
resistor (R20) in parallel with the provided R2. This boosts the
DAC output current from 10 mA to 20 mA and doubles the
peak-to-peak output voltage developed across the loads.
AD9854
Rev. C | Page 44 of 52
Single-Ended Configuration
To connect the high speed comparator in a single-ended
configuration that allows duty cycle or pulse width control
requires that a dc threshold voltage be present at one of the
comparator inputs. The user may supply this voltage using the
control DAC. A 12-bit, twos complement value is written to the
control DAC register that sets the IOUT2 output to a static dc
level. Allowable hexadecimal values are 7FF (maximum) to 800
(minimum) with all zeros being midscale. The IOUT1 channel
continues to output a filtered sine wave programmed by the
user. These two signals are routed to the comparator using W2
and W8 3-pin header switches. The configuration described in
the section Observing the Filtered IOUT1 and the Filtered
IOUT2 must be used. Follow Steps 1 through 4 in that section,
then install shorting jumpers on Pins 1 and 2 (top two pins) of
the 3-pin header W2 and W8.
The user may elect to change the RSET resistor, R2, from 3.9 kΩ
to 1.95 kΩ to receive a more robust signal at the comparator
inputs. This decreases jitter and extends the comparator
operating range. The user can accomplish this by installing a
shorting jumper at W6, which provides a second 3.9 kΩ chip
resistor (R20) in parallel with the provided R2.
USING THE PROVIDED SOFTWARE
The evaluation software is provided on a CD. This brief set of
instructions should be used in conjunction with the AD9852 or
AD9854 data sheet and the AD9852/AD9854 evaluation board
schematic.
The CD contains the following:
• The AD9852/AD9854 evaluation software
• AD9854 evaluation board instructions
• AD9854 data sheet
• AD9854 evaluation board schematics
• AD9854 PCB layout
Several numerical entries, such as frequency and phase
information, require users to press the Enter key to register the
information. So, for example, if a new frequency is input and
nothing new happens when the load button is pressed, the user
probably neglected to press the Enter key after typing the new
frequency information.
Normal operation of the AD9852/AD9854 evaluation board
begins with a master reset. Many of the default register values
after reset are depicted in the software control panel. The reset
command sets the DDS output amplitude to minimum and
0 Hz, 0 phase-offset as well as other states that are listed in the
AD9852/AD9854 Register Layout table in the data sheet.
The next programming block should be the reference clock and
multiplier because this information is used to determine the
proper 48-bit frequency tuning words that are entered and
calculated later.
The output amplitude defaults to the 12-bit straight binary
multiplier values of the I or cosine multiplier register of 000 hex
and no output (dc) should be seen from the DAC. Set the
multiplier amplitude in the Output Amplitude box to a sub-
stantial value, such as FFF hex. The digital multiplier may be
bypassed by clicking the box Output Amplitude Is Always Full
Scale, but experience has shown that doing so does not result in
best spurious-free dynamic range (SFDR). Best SFDR, as much
as 11 dB better, is obtained by routing the signal through the
digital multiplier and backing off on the multiplier amplitude.
For instance, FC0 hex produces less spurious signal amplitude
than FFF hex. It is an exploitable and repeatable phenomenon
that should be investigated in your application if SFDR must be
maximized. This phenomenon is more readily observed at
higher output frequencies where good SFDR becomes more
difficult to achieve.
Refer to this data sheet and evaluation board schematic to
understand all the functions of the AD9854 available to the user
and to gain an understanding of what the software is doing in
response to programming commands.
SUPPORT
Applications assistance is available for the AD9854, the AD9854
PCB evaluation board, and all other Analog Devices products.
Please call 1-800-ANALOGD or visit www.analog.com.
AD9854
Rev. C | Page 45 of 52
D7
D6
D5
D4
D3
D2
D1
D0
DVDD1
DVDD2
DGND1
DGND2
NC
ADDR5
ADDR4
ADDR3
ADDR2
ADDR1
ADDR0
UPDCLK
U1
AD9854
TOP VIEW
(Not to Scale)
PLLVDD
PLLGND
NC4
NC3
RSET
DACBYPASS
AVDD2
AGND2
IOUT2
IOUT2
AVDD
IOUT1
IOUT1
AGND
GND2
COMPVDD
VINB
VIN
GND
COMPGND
PLLFLT
GND3
NC5
DIFFCLKEN
CLKVDD
CLKGND
GND4
REFCLK
REFCLK
SPSELECT
MRESET
OPTGND
DVDD6
DVDD7
DGND6
DGND7
DGND8
DGND9
DVDD8
DVDD9
COUTGND2
COUTGND
COUTVDD2
COUTVDD
VOUT
NC2
DACDGND2
DACDGND
DACDVDD2
DACDVDD
OSK
FSK/BPSK/HOLD
DGND5
DGND4
DVDD5
DVDD4
DVDD3
RD
DGND3
WR
J6
J8
J16
J17
J18
J19
J20
J21
J22
J24 J23
J14
J13
J12
J11 GND
J15
W6 R2
3.9kΩ
R20
3.9kΩ
AVDD
C45
0.1µF
R1
50Ω
J4
W7
W1 1
GND
GND AVDD
R3
25Ω
W10 W16
D7
D6
D5
D4
D3
D2
D1
D0
DVDD
DVDD
GND
GND
ADDR5
ADDR4
ADDR3
ADDR2
ADDR1
ADDR0
UDCLK
AVDD
AVDD
AVDD
RD
DVDD
DVDD
DVDD
OSK
AVDD
AVDD
AVDD
AVDD
AVDD
DVDD
AVDD
GND
DVDD
W3
R4
1.3kΩ
C1
0.01µF
CLK8
CLK
PMODE
RESET
GND GND
DVDD
GND R13
50Ω
C2
0.01µF
OUT GND
NC3.3V
MC100LVEL16
L5
68nH
VEE
VBB
VCC
U3
Y2
D
D
Q
Q
DVDD
14
78
1
2
3
7
6
C25
10µF
C21
10µF
C24
0.1µFC23
0.1µFC22
0.1µFC27
0.1µFC8
0.1µFC44
0.1µF
GND
DVDD
W5
W18
W19
W20
J10
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
ADR5
ADR4
ADR3
ADR2
ADR1
ADR0
UDCLK
WR
RD
PMODE
OSK
RESET
D7
D6
D5
D4
D3
D2
D0
D1
120MHz LOW-PASS FILTER
120MHz LOW-PASS FILTER
W4
R5
50Ω
W17
R8
2kΩ
1DVDD
R11
50Ω
R12
50Ω
R19
0Ω
R14
0Ω
CLKB
CLK
J3
GND
C37
27pF C38
47pF C39
39pF C40
22pF
W8
1
L1
68nH
L6
82nH
C41
2.2pF C42
12pF C43
8.2pF
C31
22pF
C30
39pF
C5
47pF
C4
27pF
L4
82nH L2
68nH
C32
2.2pF C33
12pF C34
8.2pF GN
D
J6
W2
1
GND
1R7
25Ω
R6
50Ω
FDATA
548
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND GND GND GND
GNDGNDGNDGND
GND
GND
GND
NC = NO CONNECT
L3
68nH
R9
100Ω
GND
R10
100Ω
GND
1
GND
GND
GND
GND
GND
GND
TB1
DVDD
AVDD
VCC
1
2
3
4
GND
GND
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
60
59
58
57
56
55
54
53
52
51
50
49
48
47
46
45
44
43
42
41
GND
GND
80 79 78 77 76 75 74 73 72 71 70 69 68 67 66 65 64 63 62 61
21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40
J26
GND
J1
GND
J5
GND GND
J7
GND
J25
C6
10µFC7
0.1µFC29
0.1µFC9
0.1µFC10
0.1µFC11
0.1µFC13
0.1µF
GND
AVDD
C20
0.1µFC19
0.1µFC18
0.1µFC14
0.1µFC26
0.1µFC28
0.1µF
GND
VCC
C12
0.1µF
C17
0.1µFC16
0.1µF
J2
GND
WR
00636-B-068
Figure 68. Evaluation Board Schematic
AD9854
Rev. C | Page 46 of 52
9
8
7
6
5
4
3
2
12
13
14
15
16
17
18
19
8D
1D
GND: 10
11
1EN
74HC574
C1 VCC: 20
D0
D1
D2
D3
D4
D5
D6
D7
U8
1
3
5
9
11
13
7
74HC14
14
VCC GND
2
4
6
8
10
12
1A
2A
3A
4A
5A
6A
1Y
2Y
3Y
4Y
5Y
6Y
GND
VCC
U5
4
6
8
3
5
9
2
7
1
J11
36PINCONN
GND:[19:30]
11
13
10
12
14
A0
C0
A1
A2
A3
A4
A5
A6
A7
B6
B7
B5
B4
C1
C2
B3
C3
U6
U7
VCC
R15
10kΩ
R16
10kΩ
R17
10kΩ
VCC
VCC
GND: 10
11
1EN
74HC574
C1 VCC: 20
ADDR5
ADDR4
ADDR3
ADDR2
U9
VCC
GND: 10
11
1EN
C1
74HC574
VCC: 20
RESET
UDCLK
PMODE
ORAMP
FDATA
U4
74HC125
GND
1G
1A
1Y
2G
2A
2Y
VCC
4G
4A
4Y
3G
3A
3Y
U2
GND
1
2
3
4
5
6
7
13
12
11
10
9
8
14 VCC
VCC
U10
W11
ADDR1
ADDR0 W14
W12
W13
W9
VCC
R18
10kΩ
GND
W15
VCC RP1
10kΩ
1359
246810
7
1
3
5
9
11
13
7
74HC14
14
VCC GND
2
4
6
8
10
12
1A
2A
3A
4A
5A
6A
1Y
2Y
3Y
4Y
5Y
6Y
GND
VCC
1
3
5
9
11
13
7
74HC14
14
VCC GND
2
4
6
8
10
12
1A
2A
3A
4A
5A
6A
1Y
2Y
3Y
4Y
5Y
6Y
GND
VCC
1
3
5
9
11
13
7
74HC14
14
VCC GND
2
4
6
8
10
12
1A
2A
3A
4A
5A
6A
1Y
2Y
3Y
4Y
5Y
6Y
GND
VCC
9
8
7
6
5
4
3
2
12
13
14
15
16
17
18
19
8D
1D 9
8
7
6
5
4
3
2
12
13
14
15
16
17
18
19
8D
1D
31
32
36
VCC
VCC
VCC
WR
RD
00636-B-069
Figure 69. Evaluation Board Schematic
AD9854
Rev. C | Page 47 of 52
Table 12. AD9852/AD9854 Customer Evaluation Board (AD9852 PCB > U1 = AD9852ASQ, AD9854 PCB > U1 = AD9854ASQ)
No. Quantity REFDES Device Package Value Mfg. Part No.
1 3 C1, C2, C45 Capacitor 0805 0.01 µF
2 21 C7 to C14, C16 to C20, C22 to C24, C26
to C29, C44
Capacitor 0603 0.1 µF
3 2 C4, C37 Capacitor 1206 27 pF
4 2 C5, C38 Capacitor 1206 47 pF
5 3 C6, C21, C25 BCAPT TAJD 10 µF
6 2 C30, C39 Capacitor 1206 39 pF
7 2 C31, C40 Capacitor 1206 22 pF
8 2 C32, C41 Capacitor 1206 2.2 pF
9 2 C33, C42 Capacitor 1206 12 pF
10 2 C34, C43 Capacitor 1206 8.2 pF
11 9 J1 to J7, J25, J26 SMB STR-PC MNT ITT Industries
B51–351–0000220
12 16 J8, J9, J11 to J24 W HOLE
13 1 J10 Dual-row header 40 pins SAMTEC
TSW-120-23-L-D
14 4 L1 to L3, L5 IND-COIL 1008CS 68 nH Coilcraft
1008CS-680XGBB
15 2 L4, L6 IND-COIL 1008CS 82 nH Coilcraft
1008CS-820XGBB
16 2 R1, R5, R6, R11 to R13 Resistor 1206 50 Ω (49.9 Ω, 1%)
17 2 R2, R20 Resistor 1206 3900 Ω
18 2 R3, R7 Resistor 1206 25 Ω (24.9 Ω, 1%)
19 1 R4 Resistor 1206 1300 Ω
20 1 R8 Resistor 1206 2000 Ω
21 2 R9, R10 Resistor 1206 100 Ω
22 4 R15 to R18 Resistor 1206 10 kΩ
23 1 RP1 Resistor network SIP-10P 10 kΩ Bourns 4610X-101-103
24 1 TB1 Terminal
block and pins
4-position Wieland 25.602.2453.0 block
Z5.530.3425.0 pins
25 1 U1 AD9852 or
AD9854
LQFP-80 AD9852ASQ or AD9854ASQ
26 1 U2 74HC125 14 SOIC SN74HC125D
27 1 U3 MC100LVEL16D 8 SOIC MC100LVEL16D
28 4 U4 to U7 74HC14 14 SOIC SN74HC14D
29 3 U8 to U10 74HC574 20 SOIC SN74HC574DW
30 1 J11 36-pin connector AMP 552742-1
31 6 W1 to W4, W8, W17 3-pin jumper SAMTEC
32 10 W6, W7, W9 to W16 2-pin jumper SAMTEC
33 2 Self-tapping
screw
4–40, Philips,
round head
34 4 Rubber bumper Square black 3M SJ-5018SPBL
35 1 AD9852/AD9854 PCB GSO2669 Rev. E
36 2 R14, R19 0 Ω jumper 1206 0 Ω
37 4 Pin socket AMP 5-330808-6
38 1 Y1 (not supplied) XTAL COSC (Not supplied)
AD9854
Rev. C | Page 48 of 52
00636-B-070
Figure 70. Assembly Drawing
00636-B-071
Figure 71. Top Routing Layer, Layer 1
AD9854
Rev. C | Page 49 of 52
00636-B-072
Figure 72. Power Plane Layer, Layer 3
00636-B-073
Figure 73. Ground Plane Layer, Layer 2
AD9854
Rev. C | Page 50 of 52
00636-B-074
Figure 74. Bottom Routing Layer, Layer 4
AD9854
Rev. C | Page 51 of 52
OUTLINE DIMENSIONS
COMPLIANT TO JEDEC STANDARDS MS-026BEC-HD
0.65
BSC 0.38
0.32
0.22
16.00
BSC SQ
61 80 1
21 20
40
41
60
BOTTOM
VIEW
(PINS UP)
HEATSINK INTRUSION
0.0127 MAX
10.00
REF SQ
14.00
BSC SQ
1.60 MAX
VIEW A
SEATING
PLANE
0.75
0.60
0.45
0.15
0.05 0.10
COPLANARITY
1.45
1.40
1.35
7°
3.5°
0°
0.20
0.09
VIEW A
ROTATED 90° CCW
10°
6°
2°
6180
1
21
20 4041
60
TOP VIEW
(PINS DOWN)
Figure 75. 80-Lead Low Profile Quad Flat Package, with Heatsink [LQFP_ED]
(SQ-80-2)
Dimensions shown in millimeters
1.45
1.40
1.35
0.15
0.05
61 60
180
20 41
21 40
TOP VIEW
(PINS DOWN)
PIN 1
SEATING
PLANE
VIEW A
1.60
MAX
0.75
0.60
0.45
0.20
0.09
0.10 MAX
COPLANARITY
VIEW A
ROTATED 90° CCW
SEATING
PLANE
10°
6°
2°
7°
3.5°
0°
14.00
BSC SQ
16.00
BSC SQ
0.65
BSC 0.38
0.32
0.22
COMPLIANT TO JEDEC STANDARDS MS-026-BEC
Figure 76. 80-Lead LQFP
(ST-80-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model Temperature Range Package Description Package Option
AD9854ASQ −40°C to +85°C Thermally Enhanced 80-Lead LQFP_ED SQ-80-2
AD9854AST −40°C to +85°C 80-Lead LQFP ST-80-2
AD9854/PCB Evaluation Board
AD9854
Rev. C | Page 52 of 52
NOTES
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C00636–0–10/04(C)
