16-Bit, 1 MSPS
μModule Data Acquisition System
Data Sheet ADAQ7980/ADAQ7988
Rev. A Document Feedback
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FEATURES
Easy to use
Module data acquisition system
All active components designed by Analog Devices, Inc.
50% PCB area savings
Includes critical passive components
SPI-/QSPI-/MICROWIRE™-/DSP-compatible serial interface
Daisy-chain multiple ADAQ7980/ADAQ7988 devices
Versatile supply configuration with 1.8 V/2.5 V/3 V/5 V
logic interface
High performance
16-bit resolution with no missing codes
Throughput: 1 MSPS (ADAQ7980) and 500 kSPS (ADAQ7988)
INL: ±8 ppm typical and 20 ppm maximum
SNR: 91.5 dB typical at 10 kHz (unity gain)
THD: −105 dB at 10 kHz
Zero error: ±0.06 mV typical (unity gain)
Zero error temperature drift: 1.3 µV/°C maximum
Low power dissipation
21 mW typical at 1 MSPS (ADAQ7980)
16.5 mW typical at 500 kSPS (ADAQ7988)
Flexible power-down modes
Small, 24-lead, 5 mm × 4 mm LGA package
Excellent ESD ratings
3500 V human body model (HBM)
1250 V field-induced charged device model (FICDM)
Wide operating temperature range: −55°C to +125°C
APPLICATIONS
Automated test equipment (ATE)
Battery powered instrumentation
Communications
Data acquisition
Process control
Medical instruments
GENERAL DESCRIPTION
The ADAQ7980/ADAQ7988 are 16-bit analog-to-digital converter
(ADC) Module® data acquisition systems that integrate four
common signal processing and conditioning blocks into a system
in package (SiP) design that supports a variety of applications.
These devices contain the most critical passive components,
eliminating many of the design challenges associated with
traditional signal chains that use successive approximation
register (SAR) ADCs. These passive components are crucial to
achieving the specified device performance.
FUNCTIONAL BLOCK DIAGRAM
REF
GND
V
DD
VIO
SDI
SCK
SDO
CNV
20Ω
V
+
V–
1.8nF
LDO
2.2µF10µF
REF_OUT LDO_OUT
PD_REF
AMP_OUT
PD_AMP
PD_LDO
ADC
ADCN
IN+
IN–
ADAQ7980/
ADAQ7988
15060-001
Figure 1.
The ADAQ7980/ADAQ7988 contain a high accuracy, low power,
16-bit SAR ADC, a low power, high bandwidth, high input
impedance ADC driver, a low power, stable reference buffer,
and an efficient power management block. Housed within a tiny,
5 mm × 4 mm LGA package, these products simplify the design
process for data acquisition systems. The level of system integration
of the ADAQ7980/ADAQ7988 solves many design challenges,
while the devices still provide the flexibility of a configurable
ADC driver feedback loop to allow gain and/or common-mode
adjustments. A set of four device supplies provides optimal system
performance; however, single-supply operation is possible with
minimal impact on device operating specifications.
The ADAQ7980/ADAQ7988 integrate within a compact,
integrated circuit (IC)-like form factor key components
commonly used in data acquisition signal chain designs. The
µModule family transfers the design burden of component
selection, optimization, and layout from designer to device,
shortening overall design time, system troubleshooting, and
ultimately improving time to market.
The serial peripheral interface (SPI)-compatible serial interface
features the ability to daisy-chain multiple devices on a single, 3-
wire bus and provides an optional busy indicator. The user
interface is compatible with 1.8 V, 2.5 V, 3 V, or 5 V logic.
Specified operation of these devices is from −55°C to +125°C.
Table 1. Integrated SAR ADC μModules
Type 500 kSPS 1000 kSPS
16-Bit ADAQ7988 ADAQ7980
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 2 of 49
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications ....................................................................................... 1
General Description ......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
Specifications ..................................................................................... 3
Dual-Supply Configuration ........................................................ 3
Single-Supply Configuration ...................................................... 7
Timing Specifications ................................................................ 11
Absolute Maximum Ratings .......................................................... 13
Thermal Data .............................................................................. 13
Thermal Resistance .................................................................... 13
ESD Caution ................................................................................ 13
Pin Configuration and Function Descriptions ........................... 15
Typical Performance Characteristics ........................................... 17
Terminology .................................................................................... 25
Theory of Operation ...................................................................... 26
Circuit Information .................................................................... 26
Converter Operation .................................................................. 26
Typical Connection Diagram.................................................... 27
ADC Driver Input ...................................................................... 28
Input Protection .......................................................................... 28
Noise Considerations And Signal Settling .............................. 28
PD_AMP Operation .................................................................. 31
Dynamic Power Scaling (DPS) ................................................. 31
Slew Enhancement ..................................................................... 33
Effect of Feedback Resistor on Frequency Response ............ 33
Voltage Reference Input ............................................................ 33
Power Supply ............................................................................... 35
LDO Regulator Current-Limit and Thermal Overload
Protection .................................................................................... 36
LDO Regulator Thermal Considerations ............................... 36
Digital Interface .......................................................................... 37
3-Wire CS Mode Without the Busy Indicator ........................ 38
3-Wire CS Mode with the Busy Indicator ............................... 39
4-Wire CS Mode Without the Busy Indicator ........................ 40
4-Wire CS Mode with the Busy Indicator ............................... 41
Chain Mode Without the Busy Indicator................................ 42
Chain Mode with the Busy Indicator ...................................... 43
Application Circuits ....................................................................... 44
Nonunity Gain Configurations ................................................ 45
Inverting Configuration with Level Shift ................................ 46
Using the ADAQ7980/ADAQ7988 With Active Filters ........ 47
Applications Information .............................................................. 48
Layout .......................................................................................... 48
Evaluating the Performance of the ADAQ7980/ADAQ7988 ... 48
Outline Dimensions ....................................................................... 49
Ordering Guide .......................................................................... 49
REVISION HISTORY
8/2017—Rev. 0 to Rev. A
Changed Integrated Data Acquisition System to μModule,
Subsystem to μModule Data Acquisition System, Subsystems to
μModule Data Acquisition Systems, and DAQ Subsystem to
μModule Data Acquisition System .............................. Throughout
Changes to Features Section and Table 1 Title ............................. 1
Moved General Description Section .............................................. 3
Changes to General Description Section ...................................... 3
Change to 0.1 Hz to 10 Hz Voltage Noise Parameter Heading,
Table 2 ................................................................................................ 4
Change to 0.1 Hz to 10 Hz Voltage Noise Parameter Heading,
Table 4 ................................................................................................. 8
Changes to Human Body Model (HBM) Parameter and
Endnote 4, Table 7 .......................................................................... 14
Change to Figure 27 Caption ........................................................ 21
Changes to Circuit Information Section ..................................... 27
Change to Table 15 Title ................................................................ 45
3/2017—Revision 0: Initial Version
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 3 of 49
SPECIFICATIONS
DUAL-SUPPLY CONFIGURATION
VDD = 3.5 V to 10 V, V+ = 6.3 V to 7.7 V, V− = −2.5 V to −0.2 V, VIO = 1.7 V to 5.5 V, VREF = 5 V, TA = −55°C to +125°C, ADC driver in
a unity-gain buffer configuration, fSAMPLE = 1 MSPS (ADAQ7980), and fSAMPLE = 500 kSPS (ADAQ7988), unless otherwise noted.
Table 2.
Parameter Test Conditions/Comments Min Typ Max Unit
RESOLUTION 16 Bits
SYSTEM ACCURACY
No Missing Codes 16 Bits
Differential Nonlinearity Error (DNL) −14 ±7 +14 ppm1
Integral Nonlinearity Error (INL) −20 ±8 +20 ppm1
Transition Noise 0.6 LSB1 rms
Gain Error TA = 25°C −0.01 ±0.002 +0.01 %FS
Gain Error Temperature Drift 0.1 0.4 ppm/°C
Zero Error TA = 25°C −0.5 ±0.06 +0.5 mV
Zero Error Temperature Drift 0.3 1.3 µV/°C
Common-Mode Rejection Ratio ADC driver configured as difference amplifier 103 130 dB
Power Supply Rejection Ratio
Positive V+ = +6.3 V to +8 V, V− = −2 V 75 105 dB
Negative V+ = +7 V, V− = −1.0 V to −2.5 V 80 110 dB
SYSTEM AC PERFORMANCE
Dynamic Range 92 dB2
VREF = 2.5 V 87 dB2
Total RMS Noise 44.4 µV rms
Oversampled Dynamic Range Oversample dynamic range frequency (fODR) = 10 kSPS 111 dB2
Signal-to-Noise Ratio (SNR) Input frequency (fIN) = 10 kHz 90.5 91.5 dB2
fIN = 10 kHz, VREF = 2.5 V 84.5 86.5 dB2
Spurious-Free Dynamic Range fIN = 10 kHz 106 dB2
Total Harmonic Distortion (THD) fIN = 10 kHz −105 −100 dB2
Signal-to-Noise-and-Distortion Ratio fIN = 10 kHz 90 91 dB2
fIN = 10 kHz, VREF = 2.5 V 84 86 dB2
Effective Number of Bits fIN = 10 kHz 14.65 14.8 Bits
Noise Free Code Resolution 14.1 Bits
SYSTEM SAMPLING DYNAMICS
Conversion Rate
ADAQ7980 VIO ≥ 3.0 V 0 1 MSPS
VIO ≥ 1.7 V 0 833 kSPS
ADAQ7988 VIO ≥ 1.7 V 0 500 kSPS
Transient Response Full-scale step 430 500 ns
−3 dB Input Bandwidth ADC driver RC filter 4.42 MHz
−1 dB Frequency ADC driver RC filter 2.2 MHz
−0.1 dB Frequency ADC driver RC filter 0.67 MHz
0.1 Hz to 10 Hz Voltage Noise 17 µV p-p
Aperture Delay 2.0 ns
Aperture Jitter
2.0
ns
1 LSB means least significant bit. With the 5 V input range, 1 LSB is 76.3 µV and 1 LSB = 15.26 ppm.
2 All specifications in dB are referred to a full-scale input, FSR. Tested with an input signal at 0.5 dB below full scale, unless otherwise specified.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 4 of 49
VDD = 3.5 V to 10 V, V+ = 6.3 V to 7.7 V, V− = −2.5 V to −0.2 V, VIO = 1.7 V to 5.5 V, VREF = 5 V, T A = −55°C to +125°C, ADC driver in
a unity-gain buffer configuration, and fSAMPLE = 1 MSPS (ADAQ7980) and fSAMPLE = 500 kSPS (ADAQ7988), unless otherwise noted.
Table 3.
Parameter Test Conditions/Comments Min Typ Max Unit
REFERENCE
Input Voltage Range Voltage at REF pin 2.4 5.1 V
Load Current
REFOUT
330
µA
Buffer Input
Resistance REF 50 MΩ
Capacitance REF 1 pF
Bias Current 550 800 nA
Offset Voltage TA = 25°C 13 125 µV
Offset Voltage Drift 0.2 1.3 µV/°C
Voltage Noise fIN = 100 kHz 5.2 nV/√Hz
Voltage Noise 1/f Corner Frequency 8 Hz
Current Noise fIN = 100 kHz 0.7 pA/√Hz
0.1 Hz to 10 Hz Voltage Noise 44 nV rms
Linear Output Current REFOUT ±40 mA
Short-Circuit Current REFOUT sinking/sourcing 85/73 mA
ADC DRIVER CHARACTERISTICS
Voltage Range
IN+, IN−, AMP_OUT
0
V
REF
V
Absolute Input Voltage IN+, IN−, AMP_OUT −0.1 +5.1 V
ADCN −0.1 +0.1 V
−3 dB Bandwidth G = +1, VAMP_OUT = 0.02 V p-p 37 MHz
G = +1, VAMP_OUT = 2 V p-p 35 MHz
Bandwidth for 0.1 dB Flatness G = +1, VAMP_OUT = 0.1 V p-p 4 MHz
Slew Rate G = +1, VAMP_OUT = 2 V step 110 V/µs
G = +1, VAMP_OUT = 5 V step 40 V/µs
Input Voltage Noise f = 100 kHz 5.2 nV/√Hz
1/f Corner Frequency 8 Hz
0.1 Hz to 10 Hz Voltage Noise 44 nV rms
Input Current
Noise f = 100 kHz 0.7 pA/√Hz
Bias IN+, IN− 550 800 nA
Offset 2.1 nA
Input Offset Voltage TA = 25°C 13 125 µV
Drift 0.2 1.3 µV/°C
Open-Loop Gain
111
dB
Input Resistance IN+, IN−
Common Mode 50 MΩ
Differential Mode 260 kΩ
Input Capacitance IN+, IN− 1 pF
Input Common-Mode Voltage Range
Specified performance
−0.1
V+ − 1.3V
V
Output Overdrive Recovery Time VIN+ = 10% overdrive, fIN = 10 kHz 500 ns
Linear Output Current ±40 mA
Short-Circuit Current Sinking/sourcing 85/73 mA
DIGITAL INPUTS
Logic Levels
Input Voltage
Low (VIL) VIO > 3.0 V −0.3 +0.3 × VIO V
VIO ≤ 3.0 V −0.3 +0.1 × VIO V
High (V
IH
)
VIO > 3.0 V
0.7 × VIO
VIO + 0.3
V
VIO ≤ 3.0 V 0.9 × VIO VIO + 0.3 V
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 5 of 49
Parameter Test Conditions/Comments Min Typ Max Unit
Input Current
Low (IIL) −1 +1 µA
High (IIH) −1 +1 µA
DIGITAL OUTPUTS
Data Format Serial 16 bits, straight binary
Pipeline Delay Conversion results available
immediately after completed
conversion
VOL ISINK = 500 µA 0.4 V
VOH ISOURCE = −500 µA VIO − 0.3 V
POWER-DOWN SIGNALING
ADC Driver/REF Buffer
PD_AMP, PD_REF Voltage
Low Powered down <2.2 V
High Enabled >2.6 V
Turn-Off Time 50% of PD_AMP, PD_REF to <10% of
enabled quiescent current
1.25 2.75 µs
Turn-On Time Specified performance 2 7.25 µs
Dynamic Power Scaling Period Specified performance 10 µs
Low Dropout (LDO) Regulator
PD_LDO Voltage
Low Powered down 1.06 1.12 1.18 V
High Enabled 1.15 1.22 1.30 V
PD_LDO Logic Hysteresis 100 mV
Turn-Off Time 2.2 µF capacitive load 460 650 µs
Turn-On Time 370 425 µs
POWER REQUIREMENTS
VDD 3.5 5 10 V
LDO Voltage Accuracy ILDO_OUT = 10 mA, TA = 25°C −0.8 +0.8 %
100 µA < ILDO_OUT < 100 mA,
VDD = 3.5 V to 10 V
−1.8 +1.8 %
LDO Line Regulation VDD = 3.5 V to 10 V −0.015 +0.015 %/V
LDO Load Regulation ILDO_OUT = 100 µA to 100 mA 0.002 0.004 %/mA
LDO Start-Up Time VLDO_OUT = 2.5 V 380 µs
LDO Current-Limit Threshold
250
360
460
mA
LDO Thermal Shutdown
Threshold TJ rising 150 °C
Hysteresis 15 °C
LDO Dropout Voltage ILDO_OUT = 10 mA 30 60 mV
ILDO_OUT = 100 mA 200 420 mV
V+ 3.7 7 V− + 10 V
V− V+ − 10 −2 +0.1 V
VIO 1.7 5.5 V
Total Standby Current1, 2 Static, all devices enabled 1.2 1.7 mA
ADC driver, REF buffer disable 56 103 µA
ADC driver, REF buffer, LDO disable 14 23 µA
ADAQ7980 Current Draw
1 MSPS
VIO 0.3 0.34 mA
V+/V− 1.5 2.0 mA
VDD 1.45 1.6 mA
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 6 of 49
Parameter Test Conditions/Comments Min Typ Max Unit
ADAQ7980 Power Dissipation 1 MSPS
V+/V−/VDD 20 36 mW
1 kSPS, dynamic power scaling enabled3 5.8 9 mW
VIO 1.0 1.9 mW
Total
21
37.9
4
mW
ADAQ7988 Current Draw
VIO 0.15 0.17 mA
V+/V− 1.35 1.85 mA
VDD 0.73 0.8 mA
ADAQ7988 Power Dissipation 500 kSPS
V+/V−/VDD 16 26.5 mW
1 kSPS, dynamic power scaling enabled3 5.8 9 mW
VIO 0.5 0.95 mW
Total 16.5 27.54 mW
TEMPERATURE RANGE
Specified Performance TMIN to TMAX −55 +125 °C
1 With all digital inputs forced to VIO or GND as required.
2 During the acquisition phase.
3 Dynamic power scaling duty cycle is 10%.
4 Calculated with the maximum supply differential and not the typical supply values.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 7 of 49
SINGLE-SUPPLY CONFIGURATION
VDD = V+ = 5.0 V, V− = 0 V, VIO = 1.7 V to 5.5 V, VREF = 3.3 V, TA = −55°C to +125°C, the ADC driver in a unity-gain buffer configuration, and
fSAMPLE = 1 MSPS (ADAQ7980) and fSAMPLE = 500 kSPS (ADAQ7988), unless otherwise noted.
Table 4.
Parameter Test Conditions/Comments Min Typ Max Unit
RESOLUTION 16 Bits
SYSTEM ACCURACY
Differential Nonlinearity Error1 −14 ±7 +14 ppm2
Integral Nonlinearity Error1 −20 ±8 +20 ppm2
Transition Noise 0.8 LSB2 rms
Gain Error TA = 25°C −0.013 ±0.002 +0.013 %FS
Gain Error Temperature Drift 0.1 0.4 ppm/°C
Zero Error
T
A
= 25°C
−0.5
±0.06
+0.5
mV
Zero Error Temperature Drift 0.35 1.75 µV/°C
Common-Mode Rejection Ratio 103 133 dB
Power Supply Rejection Ratio
Positive V+ = 4.5 V to 5.5 V, V− = 0 V 75 92 dB
SYSTEM AC PERFORMANCE
Dynamic Range 89 dB3
Total RMS Noise 41.4 µV rms
Oversampled Dynamic Range fODR = 10 kSPS 109 dB3
Signal-to-Noise Ratio Input frequency (fIN) = 10 kHz 87.3 88.7 dB3
Spurious-Free Dynamic Range fIN = 10 kHz 103 dB3
Total Harmonic Distortion fIN = 10 kHz −113 −100 dB3
Signal-to-Noise-and-Distortion Ratio fIN = 10 kHz 87 88.4 dB3
Effective Number of Bits fIN = 10 kHz 14.1 14.4 Bits
Noise Free Code Resolution
13.5
Bits
SYSTEM SAMPLING DYNAMICS
Conversion Rate
ADAQ7980 VIO ≥ 3.0 V 0 1 MSPS
VIO ≥ 1.7 V 0 833 kSPS
ADAQ7988 VIO ≥ 1.7 V 0 500 kSPS
Transient Response Full-scale step 430 500 ns
−3 dB Input Bandwidth ADC driver RC filter 4.42 MHz
−1 dB Frequency ADC driver RC filter 2.2 MHz
−0.1 dB Frequency ADC driver RC filter 0.67 MHz
0.1 Hz to 10 Hz Voltage Noise 17 µV p-p
Aperture Delay 2.0 ns
Aperture Jitter 2.0 ns
1 Nonlinearity guaranteed over input voltage range. Codes below 150 mV are not represented with a unipolar supply configuration.
2 LSB means least significant bit. With the 3.3 V input range, 1 LSB = 50.4 µV, and 1 LSB = 15.26 ppm.
3 All specifications in dB are referred to a full-scale input, FSR. Tested with an input signal at 0.5 dB below full scale, unless otherwise specified.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 8 of 49
VDD = V+ = 5.0 V, V− = 0 V, VIO = 1.7 V to 5.5 V, VREF = 3.3 V, TA = −55°C to +125°C, the ADC driver in a unity-gain buffer configuration, and
fSAMPLE = 1 MSPS (ADAQ7980) and fSAMPLE = 500 kSPS (ADAQ7988), unless otherwise noted.
Table 5.
Parameter Test Conditions/Comments Min Typ Max Unit
REFERENCE
Input Voltage Range Voltage at REF pin 2.4 V+ − 1.3 V
Load Current REFOUT 330 µA
Buffer Input
Resistance REF 50 MΩ
Capacitance REF 1 pF
Bias Current 470 720 nA
Offset Voltage TA = 25°C 9 125 µV
Offset Voltage Drift
0.2
1.5
µV/°C
Voltage Noise fIN = 100kHz 5.9 nV/√Hz
Voltage Noise 1/f Corner Frequency 8 Hz
Current Noise fIN = 100kHz 0.6 pA/√Hz
0.1 Hz to 10 Hz Voltage Noise 54 nV rms
Linear Output Current REFOUT ±40 mA
Short-Circuit Current REFOUT sinking/sourcing 73/63 mA
ADC DRIVER CHARACTERISTICS
Specified Voltage Range IN+, IN−, AMP_OUT 0.15 VREF V
Absolute Input Voltage IN+, IN−, AMP_OUT −0.1 V+ − 1.3 V
ADCN −0.1 +0.1 V
−3 dB Bandwidth G = +1, VAMP_OUT = 0.02 V p-p 31 MHz
G = +1, VAMP_OUT = 2 V p-p 30 MHz
Bandwidth for 0.1 dB Flatness G = +1, VAMP_OUT = 0.1 V p-p 4 MHz
Slew Rate G = +1, VAMP_OUT = 2 V step 31 V/µs
G = +1, VAMP_OUT = 3.15 V step 20 V/µs
Input Voltage Noise f = 100 kHz 5.9 nV/√Hz
1/f Corner Frequency 8 Hz
0.1 Hz to 10 Hz Voltage Noise 54 nV rms
Input Current
Noise f = 100 kHz 0.6 pA/√Hz
Bias IN+, IN− 470 720 nA
Offset 0.4 nA
Input Offset Voltage TA = 25°C 9 125 µV
Open-Loop Gain 109 dB
Input Resistance IN+, IN−
Common Mode 50 MΩ
Differential Mode 260 kΩ
Input Capacitance IN+, IN− 1 pF
Input Common-Mode Voltage Range Specified performance −0.1 V+ − 1.3 V
Output Overdrive Recovery Time VIN+ = 10% overdrive, fIN = 10 kHz 800 ns
Linear Output Current ±40 mA
Short-Circuit Current Sinking/sourcing 73/63 mA
DIGITAL INPUTS
Logic Levels
Input Voltage
Low (VIL) VIO > 3.0 V −0.3 +0.3 × VIO V
VIO ≤ 3.0 V −0.3 +0.1 × VIO V
High (VIH) VIO > 3.0 V 0.7 × VIO VIO + 0.3 V
VIO ≤ 3.0 V 0.9 × VIO VIO + 0.3 V
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 9 of 49
Parameter Test Conditions/Comments Min Typ Max Unit
Input Current
Low (IIL) −1 +1 µA
High (IIH) −1 +1 µA
DIGITAL OUTPUTS
Data Format Serial 16 bits straight binary
Pipeline Delay Conversion results available
immediately after completed
conversion
VOL ISINK = 500 µA 0.4 V
VOH ISOURCE = −500 µA VIO − 0.3 V
POWER-DOWN SIGNALING
ADC Driver/Reference Buffer
PD_AMP, PD_REF Voltage
Low Powered down <1.5 V
High Enabled >1.9 V
Turn-Off Time 50% of PD_AMP, PD_REF to <10% of enabled
quiescent current
0.9 1.25 µs
Turn-On Time Specified performance 2 7.25 µs
Dynamic Power Scaling Period Specified performance 10 µs
LDO
PD_LDO Voltage
Low Powered down 1.06 1.12 1.18 V
High Enabled 1.15 1.22 1.30 V
PD_LDO Logic Hysteresis 100 mV
Turn-Off Time 2.2 µF capacitive load 460 650 µs
Turn-On Time 370 425 µs
POWER REQUIREMENTS
VDD 3.5 5 10 V
LDO Voltage Accuracy ILDO_OUT = 10 mA, TA = 25°C −0.8 +0.8 %
100 µA < ILDO_OUT < 100 mA, VDD = 3.5 V to 10 V −1.8 +1.8 %
LDO Line Regulation VDD = 3.5 V to 10 V −0.015 +0.015 %/V
LDO Load Regulation ILDO_OUT = 100 µA to 100 mA 0.002 0.004 %/mA
LDO Start-Up Time VLDO_OUT = 2.5 V 380 µs
LDO Current-Limit Threshold 250 360 460 mA
LDO Thermal Shutdown
Threshold
T
J
rising
150
°C
Hysteresis 15 °C
LDO Dropout Voltage ILDO_OUT = 10 mA 30 60 mV
ILDO_OUT = 100 mA 200 420 mV
V+ 3.7 5 V− + 10 V
V− V+ − 10 0 +0.1 V
VIO 1.7 5.5 V
Total Standby Current1, 2 Static, all devices enabled 1.1 1.7 mA
ADC driver, REF buffer disabled 50 103 µA
ADC driver, REF buffer, LDO disabled 7 23 µA
ADAQ7980 Current Draw 1 MSPS
VIO 0.3 0.34 mA
V+/V− 1.3 2.0 mA
VDD 1.45 1.6 mA
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 10 of 49
Parameter Test Conditions/Comments Min Typ Max Unit
ADAQ7980 Power Dissipation 1MSPS
V+/V−/VDD 13.75 36 mW
1 kSPS, ADC driver dynamic power scaling enabled3 2.9 9 mW
VIO 1.0 1.9 mW
Total
14.75
37.9
4
mW
ADAQ7988 Current Draw
VIO 0.15 0.17 mA
V+/V− 1.15 1.85 mA
VDD 0.73 0.8 mA
ADAQ7988 Power Dissipation 500 kSPS
V+/V−/VDD 9.4 26.5 mW
1 kSPS, ADC driver dynamic power scaling enabled3 2.9 9 mW
VIO 0.5 0.95 mW
Total
9.9
27.5
4
mW
TEMPERATURE RANGE
Specified Performance
T
MIN
to T
MAX
−55
+125
°C
1 With all digital inputs forced to VIO or GND as required.
2 During the acquisition phase.
3 Dynamic power scaling duty cycle is 10%.
4 Calculated with the maximum supply differential and not the typical supply values.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 11 of 49
TIMING SPECIFICATIONS
VDD = 3.5 V to 10 V, VIO = 1.7 V to 5.5 V, and TA = −55°C to +125°C, unless otherwise noted In addition to Figure 2 and Figure 3, see
Figure 72, Figure 74, Figure 76, Figure 78, Figure 80, and Figure 82 for the additional timing diagrams detailed in Table 6.
Table 6.
Parameter Symbol Min Typ Max Unit
CONVERSION TIME: CNV RISING EDGE TO DATA AVAILABLE tCONV
VIO Above 3.0 V (ADAQ7980) 500 710 ns
VIO Above 1.7 V (ADAQ7980) 500 800 ns
ADAQ7988 500 1200 ns
ACQUISITION PHASE1 tACQ ns
ADAQ7980 290 ns
ADAQ7988 800 ns
TIME BETWEEN CONVERSIONS tCYC
VIO Above 3.0 V (ADAQ7980) 1000 ns
VIO Above 1.7 V (ADAQ7980) 1200 ns
VIO Above 1.7 V (ADAQ7988) 2000 ns
CS MODE
CNV Pulse Width tCNVH 10 ns
SCK Period tSCK
VIO Above 4.5 V 10.5 ns
VIO Above 3.0 V 12 ns
VIO Above 1.7 V 22 ns
CNV or SDI Low to SDO D15 MSB Valid tEN
VIO Above 3.0 V
10
ns
VIO Above 1.7 V 40 ns
CNV or SDI High or Last SCK Falling Edge to SDO High Impedance tDIS 20 ns
SDI Valid Hold Time from CNV Rising Edge tHSDICNV
VIO Above 3.0 V 2 ns
VIO Above 1.7 V 10 ns
CHAIN MODE
SCK Period tSCK
VIO Above 4.5 V 11.5 ns
VIO Above 3.0 V 13 ns
VIO Above 1.7 V 23 ns
SDI Valid Hold Time from CNV Rising Edge tHSDICNV 0 ns
SCK Valid Setup Time from CNV Rising Edge tSSCKCNV 5 ns
SCK Valid Hold Time from CNV Rising Edge tHSCKCNV 5 ns
SDI Valid Setup Time from SCK Falling Edge
t
SSDISCK
2
ns
SDI Valid Hold Time from SCK Falling Edge tHSDISCK 3 ns
SDI High to SDO High (with Busy Indicator) tDSDOSDI
VIO Above 3.0 V 15 ns
VIO Above 1.7 V 22 ns
SCK
Low Time tSCKL
VIO Above 3.0 V 4.5 ns
VIO Above 1.7 V 6 ns
High Time tSCKH
VIO Above 3.0 V 4.5 ns
VIO Above 1.7 V 6 ns
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 12 of 49
Parameter Symbol Min Typ Max Unit
Falling Edge to Data Remains Valid tHSDO 3 ns
Falling Edge to Data Valid Delay tDSDO
VIO Above 4.5 V 9.5 ns
VIO Above 3.0 V 11 ns
VIO Above 1.7 V
21
ns
SDI VALID SETUP TIME From CNV RISING EDGE tSSDICNV 5 ns
1 The acquisition phase is the time available for the ADC sampling capacitors to acquire a new input with the ADC running at a throughput rate of 1 MSPS.
500µA I
OL
500µA I
OH
1.4V
TO SDO C
L
20pF
15060-002
Figure 2. Load Circuit for Digital Interface Timing
X% VIO
1
Y% VIO
1
V
IH2
V
IL2
V
IL2
V
IH2
tDELAY tDELAY
1
FOR VIO ≤ 3.0V, X = 90, AND Y = 10; FOR VIO > 3.0V, X = 70, AND Y = 30.
2
MINIMUM V
IH
AND MAXIMUM V
IL
USED. SEE DIGITAL INPUTS
SPE CI FI CATI ONS I N TABL E 3 OR T ABLE 5.
15060-003
Figure 3. Voltage Levels for Timing
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 13 of 49
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
Table 7.
Parameter Rating
V+ to V− 11 V
V+ to GND −0.3 V to +11 V
V− to GND −11 V to +0.3 V
VDD to GND −0.3 V to +24 V
REF_OUT/VIO to GND −0.3 V to +6 V
IN+/IN−/REF to GND V− − 0.7 V to V+ + 0.7 V
AMP_OUT/ADCN to GND
−0.3 V to V
REF
+ 0.3 V or
±130 mA
Differential Analog Input Voltage
(IN+ − IN−)
±1 V
Digital Input1 Voltage to GND −0.3 V to VIO + 0.3 V
Digital Output2 Voltage to GND −0.3 V to VIO + 0.3 V
Input Current to Any Pin Except Supplies3, 4 ±10 mA
Operating Temperature Range −55°C to +125°C
Storage Temperature Range −65°C to +150°C
Junction Temperature 150°C
ESD
Human Body Model (HBM) 3500 V
Field Induced Charged Device Model
(FICDM)
1250 V
1 The digital input pins include the following: CNV, SDI, and SCK.
2 The digital output pin is SDO.
3 Transient currents of up to 100 mA do not cause SCR latch-up.
4 Condition applies when power is provided to the device.
Stresses at or above those listed under Absolute Maximum Ratings
may cause permanent damage to the product. This is a stress
rating only; functional operation of the product at these or any
other conditions above those indicated in the operational section of
this specification is not implied. Operation beyond the maximum
operating conditions for extended periods may affect product
reliability.
THERMAL DATA
Absolute maximum ratings apply individually only, not in
combination. The ADAQ7980/ADAQ7988 can be damaged
when the junction temperature (TJ) limits are exceeded.
Monitoring ambient temperature does not guarantee that TJ is
within the specified temperature limits. In applications with
high power dissipation and poor thermal resistance, the
maximum ambient temperature (TA) may have to be derated.
In applications with moderate power dissipation and low printed
circuit board (PCB) thermal resistance, the maximum TA can
exceed the maximum limit as long as the junction temperature
is within specification limits. The θJA of the package is based on
modeling and calculation using a 4-layer board. The θJA is highly
dependent on the application and board layout. In applications
where high maximum power dissipation exists, close attention
to thermal board design is required. The θJA value may vary
depending on PCB material, layout, and environmental conditions.
THERMAL RESISTANCE
Thermal resistance values specified in Table 8 were calculated
based on JEDEC specifications and must be used in compliance
with JESD51-12. Because the product contains more than one
silicon device, only the worst case junction temperature is reported.
Table 8. Thermal Resistance
Package Type1, 2 θJA θJC TOP2 ΨJT Unit
CC-24-2 65 103 12.6 ˚C/W
1 These values represent the worst case die junction in the package.
2 Table 8 values were calculated based on the standard JEDEC test conditions
defined in Table 9, unless otherwise specified.
3 For θJC test, 100 µm thermal interface material (TIM) was used. TIM is
assumed to be 3.6 W/mK.
Only use θJA and θJC TOP to compare thermal performance of the
package of the device with other semiconductor packages when
all test conditions listed are similar. One common mistake is to
use θJA and θJC to estimate the junction temperature in the system
environment. Instead, using ΨJT is a more appropriate way to
estimate the worst case junction temperature of the device in the
system environment. First, take an accurate thermal measurement
of the top center of the device (on the mold compound in this
case) while the device operates in the system environment. This
measurement is known in the following equation as TTOP. This
equation can then be used to solve for the worst case TJ in that
given environment as follows:
TJ = ΨJT × P + TTOP
where:
ΨJT is the junction to top thermal characterization number as
specified in data sheet.
P refers to total power dissipation in the chip (W).
TTOP refers to the package top temperature (°C) and is measured
at the top center of the package in the environment of the user.
ESD CAUTION
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 14 of 49
Table 9. Standard JEDEC Test Conditions
Test Conditions θJA θJC θJB
Main Heat Transfer Mode Convection Conduction Conduction
Board Type 2S2P 1S0P 2S2P
Board Thickness 1.6 mm 1.6 mm 1.6 mm
Board Dimension If package length is <27 mm,
76.2 mm × 114.3 mm; otherwise,
101.6 mm × 114.3 mm
If package length is <27 mm,
76.2 mm × 114.3 mm; otherwise,
101.6 mm × 114.3 mm
If package length is <27 mm,
76.2 mm × 114.3 mm; otherwise,
101.6 mm × 114.3 mm
Signal Traces Thickness 0.07 mm 0.07 mm 0.07 mm
PWR/GND Traces Thickness 0.035 mm Not applicable 0.035 mm
Thermal Vias Use thermal vias with 0.3 mm
diameter, 0.025 mm plating, and
1.2 mm pitch whenever a package
has an exposed thermal pad; vias
numbers are maximized to cover
the area of the exposed paddle
Use thermal vias with 0.3 mm
diameter, 0.025 mm plating, and
1.2 mm pitch whenever a package
has an exposed thermal pad; vias
numbers are maximized to cover
the area of the exposed paddle
Use thermal vias with 0.3 mm
diameter, 0.025 mm plating, and
1.2 mm pitch whenever a package
has an exposed thermal pad; vias
numbers are maximized to cover
the area of the exposed paddle
Cold Plate Not applicable Cold plate attaches to either
package top or bottom depending
on the path of least thermal
resistance
Fluid cooled, ring style cold plate
that clamps both sides of the test
board such that heat flows from
package radially in the plane of
the test board
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 15 of 49
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
IN+
CNV
1
IN–
2
AMP_OUT
3
ADCN
4
GND
5
GND
GND
V–
GND
PD_REF
PD_AMP
LDO_OUT
13
SDO
14
SCK
15
SDI
16
VIO
17
ADAQ7980/
ADAQ7988
TOP VIEW
(No t t o Scal e)
GND
18
PD_LDO
19
VDD
20
V+
21
GND
22
REF
23
REF_OUT
24
678910 11 12
15060-004
Figure 4. Pin Configuration
Table 10. Pin Function Descriptions
Pin No. Mnemonic Type1 Description
1 IN+ AI ADC Driver Noninverting Input.
2 IN− AI ADC Driver Inverting Input.
3 AMP_OUT AI, AO ADC Driver Output and ADC Input Before Low-Pass Filter (LPF).
4 ADCN AI Analog Input Ground Sense. Connect this pin to the analog ground plane or to a remote sense ground.
5 to 7,
9, 18,
22
GND P Ground.
8 V− P Negative Power Supply Line for the ADC Driver. This pin requires a 100 nF capacitor to GND for best
operation. Connect this pin to ground for single-supply operation.
10 PD_REF DI Active Low Power-Down Signal for Reference Buffer. When powered down, the reference buffer
output enters a high impedance (high-Z) state.
11 PD_AMP DI Active Low Power-Down Signal for ADC Driver. When powered down, the reference buffer output
enters a high-Z state.
12 LDO_OUT P Regulated 2.5 Output Voltage from On-Board LDO. An internal 2.2 μF bypass capacitor to GND is
provided.
13 CNV DI Convert Input. This input has multiple functions. On its leading edge, it initiates the conversions and
selects the interface mode of the device, chain, or CS mode. In CS mode, it enables the SDO pin
when low. In chain mode, read the data when CNV is high.
14 SDO DO Serial Data Output. The conversion result is output on this pin. SDO synchronizes with SCK.
15 SCK DI Serial Data Clock Input. When the device is selected, the conversion result is shifted out onto SDO by
this clock.
16 SDI DI Serial Data Input. This input provides multiple features. It selects the interface mode of the ADC as
follows.
When SDI is low during the CNV rising edge, chain mode is selected. In this mode, SDI is used as a
data input to daisy-chain the conversion results of two or more ADCs onto a single SDO line. The
digital data level on SDI is output on SDO with a delay of 16 SCK cycles.
When SDI is high during the CNV rising edge, CS mode is selected. In this mode, either SDI or CNV
can enable the serial output signals when low; if SDI or CNV is low when the conversion is complete,
the busy indicator feature is enabled.
17 VIO P Input/Output Interface Digital Power. VIO is nominally at the same supply as the host interface (1.8 V,
2.5 V, 3 V, or 5 V).
19 PD_LDO DI Active Low Power-Down Signal for LDO. When powered down, the LDO output enters a high-Z state.
For a continuously enabled state or for automatic startup, tie PD_LDO to the VDD pin (Pin 20).
20 VDD P Regulator Input Supply. Bypass VDD to GND with a 2.2 μF capacitor.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 16 of 49
Pin No. Mnemonic Type1 Description
21 V+ P Positive Power Supply Line for the ADC Driver and Reference Buffer. This pin can be tied to VDD as
long as headroom for the reference buffer is maintained. This pin requires a 100 nF capacitor to GND
for best operation.
23 REF AI External Reference Signal. REF is the noninverting input of on-board reference buffer. Connect an external
reference source to this pin. A low-pass filter may be required between the reference source and this pin to
band limit noise generated by the reference source.
24 REF_OUT AO Reference Buffer Output. This pin provides access to the buffered reference signal presented to the ADC.
1 AI is analog input, AO is analog output, P is power, DI is digital input, and DO is digital output.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 17 of 49
TYPICAL PERFORMANCE CHARACTERISTICS
VDD = 3.5 V to 10 V, V+ = 6.3 V to 7.7 V, V− = −1.0 V to −2.5 V, VIO = 1.7 V to 5.5 V, VREF = 5 V, T A = 25°C, ADC driver in a unity-gain
buffer configuration, fSAMPLE = 1 MSPS (ADAQ7980), fSAMPLE = 500 kSPS (ADAQ7988), and fIN = 10 kHz, unless otherwise noted.
20
15
10
5
0
–5
–10
–15
–20 060k50k40k30k20k10k
INTEGRAL NONLINEARITY (ppm)
CODE
POSITIVE INL = +4.3ppm
NEGATIVE INL = –5.8ppm
15060-105
Figure 5. Integral Nonlinearity vs. Code, REF = 5 V
20
15
10
5
0
–5
–10
–15
–20
2048 620485204842048320482204812048
INTEGRAL NONLINEARITY (ppm)
CODE
POSITIVE INL = +7.8ppm
NEGATIVE INL = –4.0ppm
15060-106
Figure 6. Integral Nonlinearity vs. Code, V+ = VDD = 5 V, V− = 0 V,
REF = 3.3 V
0
–100
–120
–140
–40
–20
–60
–80
–160 0500k400k300k200k100k 450k350k250k150k50k
AMPLITUDE (dB of FULL SCALE)
FREQUENCY ( Hz )
SNR = 91. 77dB
SINAD = 91.56d B
THD = –104.32d B
SFDR = 105.08d Bc
15060-107
Figure 7. FFT, REF = 5 V
20
15
10
5
0
–5
–10
–15
–20 060k50k
40k30k
20k10k
DIFFERENTIAL NONLINEARITY (ppm)
CODE
POSITIVE DNL = 6.0ppm
NEGATI V E DNL = –6. 4pp m
15060-108
Figure 8. Differential Nonlinearity vs. Code, REF = 5 V
20
15
10
5
0
–5
–10
–15
–20
2048 62048
5204842048320482204812048 CODE
POSITIVE DNL = 6.9ppm
NEGATI V E DNL = –6. 1pp m
DIFFERENTIAL NONLINEARITY (ppm)
15060-109
Figure 9. Differential Nonlinearity vs. Code, V+ = VDD = 5 V, V− = 0 V,
REF = 3.3 V
0
–100
–120
–140
–40
–20
–60
–80
–160 0500k400k300k200k100k 450k350k250k150k50k
AMPLITUDE (dB of FULL SCALE)
FREQUENCY ( Hz )
SNR = 86. 87dB
SINAD = 86.85d B
THD = –110.10d B
SFDR = 103.70d Bc
15060-110
Figure 10. FFT, REF = 2.5 V
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 18 of 49
180000
160000
140000
120000
100000
80000
60000
40000
20000
0
COUNTS
ADC CODE
32785 32786 32787 32788 32789 32790 32791 32792 32793
15060-111
03979
52669
170828
37210
454 1 0
TOTAL COUNT = 262144
Figure 11. Histogram of a DC Input at the Code Center, REF = 5 V
140000
120000
100000
80000
60000
40000
20000
0
COUNTS
ADC CODE
15060-112
TOTAL COUNT = 262144
064
8973
118166 124157
10708
76 0
32790 32791 32792 32793 32794 32795 32796 32797
Figure 12. Histogram of a DC Input at the Code Transition, REF = 5 V
93
88
87
91
92
90
89
86
15.0
14.3
14.1
14.8
14.9
14.7
14.5
14.4
14.2
14.6
14.0
2.4 4.94.43.4 3.92.9
SNR, S INAD (dB)
ENOB (Bit s)
REFERENCE (V)
SNR
SINAD
ENOB
15060-113
Figure 13. SNR, SINAD, and ENOB vs. Reference Voltage
100000
90000
70000
50000
30000
80000
60000
40000
20000
10000
0
COUNTS
ADC CODE
15060-114
TOTAL COUNT = 262144
0123 349 3857
21893
62761
88057
60351
20580
3829 426 17 0
26229
26230
26231
26232
26233
26234
26235
26236
26237
26238
26239
26240
26241
26242
Figure 14. Histogram of a DC Input at the Code Center, REF = 2.5 V
–120
–130
–132
–136
–134
–138
–124
–122
–126
–128
–140
–10 0–2–4–6–8 –1
–3
–5
–7–9
NOISE FLOOR (dB)
INPUT LEVEL (d BFS)
FFT SIZE = 65536
15060-115
Figure 15. Noise Floor vs. Input Level
–95
–120
–105
–100
–110
–115
–125
115
90
105
110
100
95
85
2.25 5.254.253.25 4.753.752.75
THD ( dB)
SF DR ( dB)
REFERENCE VOLTAGE (V)
THD
SFDR
15060-116
Figure 16. THD and SFDR vs. Reference Voltage
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 19 of 49
100
75
90
95
85
80
70
–60
–110
–80
–70
–90
–100
–120
110010
SINAD ( dB)
THD ( dB)
FREQUENCY ( kHz )
SINAD
THD
15060-117
Figure 17. SINAD and THD vs. Frequency
93.0
90.5
92.0
92.5
91.5
91.0
90.0
–55 12545–15 85565 10525–35
SNR, S INAD (dB)
TEMPERATURE (°C)
SNR
SINAD
15060-118
Figure 18. SNR and SINAD vs. Temperature
–104.0
–105.0
–105.2
–105.6
–105.4
–105.8
–104.4
–104.2
–104.6
–104.8
–106.0
–55 125
1056525–15 85
455–35
THD ( dB)
TEMPERATURE (°C)
15060-120
Figure 19. THD vs. Temperature
9
–6
–9
3
6
0
–3
–12 100101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
V+ = +7V
V− = −2V
V
OUT
= 20mV p - p
G = +2
G = +1, R
F
= 0
G = –1
0.1
15060-122
Figure 20. ADC Driver Small Signal Frequency Response for Various Gains
9
–6
–9
3
6
0
–3
–12 100101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
20mV p- p
G = +1
V+ = + 7V , V– = –2V
V+ = +5V, V– = 0V
0.1
15060-123
Figure 21. ADC Driver Small Signal Frequency Response for Various Supply
Voltages
3
–6
0
–3
–9 100101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
2V p-p
G = +1
V+ = +7V
V− = −2V
+125°C
+25°C
–40°C
–55°C
0.1
15060-124
Figure 22. Large Signal Frequency Response for Various Temperatures
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 20 of 49
9
–6
–9
3
6
0
–3
–12 100101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
20mV p- p
G = +1
V+ = +7V
V– = –2V
+125°C
+85°C
+25°C
–40°C
–55°C
0.1
15060-125
Figure 23. ADC Driver Small Signal Frequency Response for Various
Temperatures
6
–6
–9
3
0
–3
–12 100
101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
V+ = +7V
V– = –2V
V
OUT
= 2V p-p
G = +2
G = +1,RF = 0
G = –1
0.1
15060-126
Figure 24. ADC Driver Large Signal Frequency Response for Various Gains
9
–6
–9
3
6
0
–3
–120.1 100101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
G = +1
V+ = +7V
V– = –2V
20mV p- p
200mV p- p
500mV p- p
2V p-p
15060-127
Figure 25. Frequency Response for Various Output Voltages
5
1
0
4
3
2
–1 100101
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
100mV p- p
G = +1
V+ = 7V
V– = –2V
0.1
15060-128
Figure 26. ADC Driver Small Signal 0.1 dB Bandwidth
20
–5
–15
–10
10
15
5
0
–20 010
98
76
5432
1
VOLTAGE NOI SE (µ V)
TIME (Seconds)
15060-227
Figure 27. 0.1 Hz to 10 Hz Voltage Noise
300
350
400
450
500
550
600
650
700
750
800
–40 –25 –10 520 35 50 65 80 95 110 125
QUIESCENT SUPPLY CURRENT (µA)
TEMPERATURE (°C)
V+ = +5V, V– = 0V
V+ = +10V, V– = 0V
15060-130
Figure 28. ADC Driver and Reference Buffer Quiescent Supply Current vs.
Temperature for Various Supplies
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 21 of 49
0.6
0.5
0.4
0.3
0.2
0.1
0010987654321
RECOVERY TIME (µs)
OVE RLO AD DURATION (µ s)
15060-229
G = +1
V+ = +7V
V– = –2V
VIN = 10% OVERDRIVE
Figure 29. Recovery Time vs. Overload Duration
–180
–160
–140
–120
–100
–80
–60
–40
–20
0
–20
0
20
40
60
80
100
120
10 100 1k 10k 100k 1M 10M 100M
OPE N- LO OP PHAS E ( Degrees)
OPEN-LOOP GAIN (dB)
FREQUENCY ( Hz )
GAIN
PHASE
15060-132
Figure 30. ADC Driver Open-Loop Gain and Phase vs. Frequency
0
200
400
600
800
100
1200
1400
–55 –35 –15 525 45 65 85 105 125
ADC DRIVE R
DYNAMIC PO WER S CALING T URN- ON TIME ( ns)
TEMPERATURE (°C)
15060-231
fS
= 100kSPS
V– = 0V
V
REF
= 3.3V
V+ = 10V
V+ = 7V
V+ = 5V
Figure 31. ADC Driver Dynamic Power Scaling Turn-On Time vs. Temperature
for Various Supply Voltages
1200
1000
800
600
400
200
04109
87
6
5
ADC DRIVE R DY NAM IC POWE R S CALI NG
TURN O N TI M E ( ns)
SUPPLY (V)
15060-233
fS
= 100kSPS
V– = 0V
V
REF
= 3.3V
Figure 32. ADC Driver Dynamic Power Scaling Turn On Time vs. Supply
Voltage
0
100
200
300
400
500
600
800
700
0123456
SUPPLY CURRE NT (µ A)
TIME (µs)
15060-242
V– = 0V
V+ = 10V
V+ = 5V
V+ = 4V
Figure 33. Supply Current vs. ADC Driver and Reference Buffer Turn-Off
Response Time for Various Supplies
0
100
200
300
400
500
600
700
800
0123456
SUPPLY CURRE NT (µ A)
TIME (µs)
15060-235
V+ = 5V
V– = 0V
+125°C
+25°C
–40°C
Figure 34. Supply Current vs. ADC Driver and Reference Buffer Turn-Off
Response Time for Various Temperatures
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 22 of 49
–140
–120
–100
–80
–60
–40
–20
0
20
10 100 1k 10k 100k 1M 10M 100M
FREQUENCY (Hz)
+PSRR
CMRR
CMRR, P S RR ( dB)
V+ = 5V, V– = 0V
ΔV+, ΔVCM = 100mV p-p
15060-138
Figure 35. CMRR and PSRR vs. Frequency
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 8.5 9.0 9.5 10.0
POWER- DOWN THRES HOLD ( V )
SUPPLY VOLTAGE F ROM G ROUND (V)
DEVI CE E NABLED
DEVI CE DISABLED
T
A
= –40° C
T
A
= +25°C
T
A
= +125°C
15060-139
Figure 36. ADC Driver and Reference Buffer Power-Down Threshold vs.
Supply Voltage from Ground for Various Temperatures
0.0025
0.0005
0.0020
0.0015
0.0010
0
–55 12545–15 85565 10525
–35
GAIN ERRO R ( % FS)
TEMPERATURE (°C)
15060-140
Figure 37. Gain Error vs. Temperature
90
–30
70
30
50
10
–10
–50
–55 12545–15 85565 10525–35
REFERENCE BUFFER OFFSET VOLTAGE (µV)
TEMPERATURE (°C)
15060-141
Figure 38. Reference Buffer Input Offset Voltage vs. Temperature
1.5
0.7
1.4
1.2
1.3
1.1
0.9
0.6
1.0
0.8
0.5
–55 12545
–15 85565 105
25–35
SUPPLY CURRE NT (mA)
TEMPERATURE (°C)
f
SAMPLE = 0Hz
V+ = 10V
V+ = 5V
V+ = 3.8V
15060-142
Figure 39. ADC Driver and Reference Buffer Static Supply Current vs.
Temperature for Various Supplies
1.9
1.7
1.3
1.5
1.1
0.7
0.9
0.5
–55 12545–15 85565 10525–35
SUPPLY CURRE NT (mA)
TEMPERATURE (°C)
V+ = 10V
V+ = 5V
V+ = 3.8V
15060-143
Figure 40. ADC Driver and Reference Buffer Dynamic Supply Current vs.
Temperature for Various Supplies
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 23 of 49
0.020
0.015
0.005
0.010
0
–55 125
45
–15 85
565 105
25
–35
PD CURRENT (mA)
TEMPERATURE (°C)
PD CURRENT 10V S UP P LY DE LTA
PD CURRENT 5V S UP P LY DE LTA
PD CURRENT 3.8V S UP P LY DELT A
f
SAMPLE
= 0Hz
15060-144
Figure 41. Total ADC Driver and Reference Buffer Power-Down (PD) Current vs.
Temperature
0.05
0.03
–0.01
–0.03
0.01
–0.05
–55 12545–15 85565 10525–35
OFF SET ERROR (mV)
TEMPERATURE (°C)
OF FSE T ERRO R V + = + 7V , V– = –2V
OF FSE T ERRO R V + = + 5V , V– = 0V
15060-145
Figure 42. Offset Error vs. Temperature
1.6
1.4
0.8
0.4
1.2
0.6
0.2
1.0
0
–55 12545–15 85565 10525–35
LDO CURRENT ( mA)
TEMPERATURE (°C)
LDO DY NAM IC CURRENT 10V I NP UT
LDO S TAT IC CURRENT 10V INPUT
15060-146
Figure 43. LDO Current vs. Temperature for Various Supplies
0
0.5
1.0
1.5
2.0
2.5
3.5
3.0
0100k 200k 300k 400k 500k 600k 700k 800k 900k 1000k
TO TAL OPERATI NG CURRENT (mA)
SAMPLE RATE (SPS)
15060-245
V+ = 3.8V, V– = 0V
V+ = 5V , V– = 0V
V+ = 7.7V, V– = 0V
V+ = 10V , V– = 0V
Figure 44. Total Operating Current vs. Sample Rate for Various Supplies
2.60
2.55
2.45
2.50
2.40
–55 12545
–15 85565 10525–35
LDO_OUT (V)
TEMPERATURE (°C)
15060-148
Figure 45. Output Voltage (LDO_OUT) vs. Temperature
2.55
2.53
2.49
2.47
2.51
2.45 01005020 7030 60 90804010
LDO_OUT (V)
LO AD CURRE NT (mA)
15060-149
Figure 46. Output Voltage (LDO_OUT) vs. Load Current (ILOAD)
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 24 of 49
2.45
2.55
2.53
2.51
2.49
2.47
45 6 7 8 9 10
LDO_OUT (V)
V
DD
(V)
15060-248
I
LOAD
= 1mA
I
LOAD
= 10mA
I
LOAD
= 100mA
Figure 47. Output Voltage (LDO_OUT) vs. VDD
–55 125
45–15 85
565 105
25–35 TEMPERATURE (°C)
0.0020
0.0015
0.0005
0.0010
0
LDO PD CURRENT (mA)
f
SAMPLE
= 0Hz
V
DD
= 5V
15060-151
Figure 48. LDO PD Current vs. Temperature
0
0.20
0.16
0.12
0.08
0.04
0.02
0.18
0.14
0.10
0.06
0.0001 0.001 0.01 0.1
LDO DROPOUT VOLTAGE (V)
I
LOAD
(mA)
15060-250
Figure 49. LDO Dropout Voltage vs. Load Current (ILOAD), LDO_OUT = 2.5 V
2.30
2.60
2.55
2.50
2.45
2.40
2.35
2.50 3.00
2.952.902.852.80
2.75
2.702.65
2.602.55
LDO_OUT (V)
V
DD
(V)
15060-251
I
LOAD
= 1mA
I
LOAD
= 10mA
I
LOAD
= 100mA
Figure 50. LDO_OUT vs. VDD in Dropout, LDO_OUT = 2.5 V
–90
–80
–70
–60
–50
–40
–30
–20
–10
0.01 0.1 110 100
ISOLATION (dB)
FREQUENCY (MHz)
V+ = 5V
V– = 0V
V
IN
= 0.5 V p-p
15060-154
Figure 51. Forward/Off Isolation vs. Frequency
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 25 of 49
TERMINOLOGY
Integral Nonlinearity Error (INL)
INL refers to the deviation of each individual code from a line
drawn from negative full scale through positive full scale. The
point used as negative full scale occurs ½ LSB before the first
code transition. Positive full scale is a level 1½ LSB beyond the
last code transition. The deviation is measured from the middle
of each code to the true straight
Differential Nonlinearity Error (DNL)
In an ideal ADC, code transitions are 1 LSB apart. DNL is the
maximum deviation from this ideal value. It is often specified in
terms of resolution for which no missing codes are guaranteed.
Zero Error
The first transition occurs at a level ½ LSB above analog ground
(38.1 µV for the 0 V to 5 V range). The offset error is the
deviation of the actual transition from that point.
Gain Error
The last transition (from 111 … 10 to 111 … 11) occurs for an
analog voltage 1½ LSB below the nominal full scale (4.999886 V
for the 0 V to 5 V range). The gain error is the deviation of the
actual level of the last transition from the ideal level after the
offset is adjusted out.
Spurious-Free Dynamic Range (SFDR)
SFDR is the difference, in decibels (dB), between the rms
amplitude of the input signal and the peak spurious signal.
Effective Number of Bits (ENOB)
ENOB is a measurement of the resolution with a sine wave
input. It is related to SINAD by the following formula:
ENOB = (SINADdB − 1.76)/6.02
ENOB is expressed in bits.
Noise Free Code Resolution
Noise free code resolution is the number of bits beyond which it
is impossible to distinctly resolve individual codes. Calculate it as
follows:
Noise Free Code Resolution = log2(2N/Peak to Peak Noise)
Noise free code resolution is expressed in bits.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of the first five harmonic
components to the rms value of a full-scale input signal and is
expressed in decibels (dB).
Dynamic Range
Dynamic range is the ratio of the rms value of the full scale to
the total rms noise measured with the inputs shorted together.
The value for dynamic range is expressed in decibels (dB). It is
measured with a signal at −60 dBFS to include all noise sources
and DNL artifacts.
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the actual input signal to the
rms sum of all other spectral components below the Nyquist
frequency, excluding harmonics and dc. The value for SNR is
expressed in decibels (dB).
Signal-to-Noise-and-Distortion (SINAD) Ratio
SINAD is the ratio of the rms value of the actual input signal to
the rms sum of all other spectral components below the Nyquist
frequency, including harmonics but excluding dc. The value for
SINAD is expressed in decibels (dB).
Aperture Delay
Aperture delay is the measure of the acquisition performance. It
is the time between the rising edge of the CNV input and when
the input signal is held for a conversion.
Transient Response
Transient response is the time required for the ADC to accurately
acquire its input after a full-scale step function is applied.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 26 of 49
THEORY OF OPERATION
COMP
SW ITCHE S CONT ROL
OUTPUT CODE
CNV
CONTROL
LOGIC
SW+LSB
SW–LSB
IN+
REF
GND
IN–
MSB
MSB
C
C
4C 2C
16,384C
32,768C
CC4C 2C16,384C32,768C
15060-055
Figure 52. ADC Simplified Schematic
CIRCUIT INFORMATION
The ADAQ7980/ADAQ7988 system in package (SiP) is a fast,
low power, precise data acquisition (DAQ) signal chain that
uses a SAR architecture. The μModule data acquisition system
contains a high bandwidth, analog-to-digital converter (ADC)
driver, a low noise reference buffer, a low dropout regulator
(LDO), and a 16-bit SAR ADC, along with critical passive
components required to achieve optimal performance. All active
components in the circuit are designed by Analog Devices, Inc.
The ADAQ7980/ADAQ7988 are capable of converting
1,000,000 samples per second (1 MSPS) and 500,000 samples
per second (500 kSPS), respectively. The ADC powers down
between conversions; therefore, power consumption scales with
sample rate. The ADC driver and reference buffer are capable of
dynamic power scaling, where the power consumption of these
components scales with sample rate. When operating at 1 kSPS,
for example, the ADAQ7980/ADAQ7988 consume 2.9 mW
typically, ideal for battery-powered applications.
The ADAQ7980/ADAQ7988 offer a significant form factor
reduction compared to traditional signal chains while still
providing flexibility to adapt to a wide array of applications.
All three signal pins of the ADC driver are available to the user,
allowing various amplifier configurations. The devices house the
LPF between the driver and the ADC, controlling the signal
chain bandwidth and providing a bill of materials reduction.
The ADAQ7980/ADAQ7988 do not exhibit any pipeline delay
or latency, making them ideal for multiplexed applications.
The ADAQ7980/ADAQ7988 house a reference buffer and the
corresponding decoupling capacitor. The placement of this
decoupling capacitor is vital to achieving peak conversion
performance. Inclusion of this capacitor in the μModule data
acquisition system eliminates this performance hurdle. The
reference buffer is configured for unity gain. By only including
the reference buffer, the user has the flexibility to choose the
reference buffer input voltage that matches the desired analog
input range.
The ADAQ7980/ADAQ7988 interface to any 1.8 V to 5 V digital
logic family. They are housed in a tiny 24-lead LGA that provides
significant space savings and allows flexible configurations.
CONVERTER OPERATION
The ADAQ7980/ADAQ7988 contain a successive approximation
ADC based on a charge redistribution digital-to-analog converter
(DAC). Figure 52 shows the simplified schematic of the ADC.
The capacitive DAC consists of two identical arrays of 16 binary
weighted capacitors, which are connected to the two comparator
inputs.
During the acquisition phase, terminals of the array tied to the
input of the comparator are connected to GND via the internal
switches (SW+ and SW−). All independent switches are connected
to the analog inputs. Therefore, the capacitor arrays are used as
sampling capacitors and acquire the analog signal on the ADC
inputs. When the acquisition phase is completed and the CNV
input goes high, a conversion phase initiates. When the conversion
phase begins, SW+ and SW− open first. The two capacitor arrays
are then disconnected from the ADC input and connected to the
GND input. Therefore, the differential voltage between the ADC
input pins captured at the end of the acquisition phase are
applied to the comparator inputs, causing the comparator to
become unbalanced. By switching each element of the capacitor
array between GND and REF, the comparator input varies by
binary weighted voltage steps (VREF/2, VREF/4 … VREF/65,536).
The control logic toggles these switches, starting with the MSB,
to bring the comparator back into a balanced condition. After
the completion of this process, the devices return to the acquisition
phase, and the control logic generates the ADC output code and
a busy signal indicator signaling the user that the conversion is
complete.
Because the ADAQ7980/ADAQ7988 have an on-board conversion
clock, the serial clock (SCK) is not required for the conversion
process.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 27 of 49
Transfer Functions
The ideal transfer characteristics for the ADAQ7980/ADAQ7988
are shown in Figure 53 and Table 11.
000 ... 000
000 ... 001
000 ... 010
111 ... 101
111 ... 110
111 ... 111
–FSR –FSR + 1LSB
–FSR + 0.5LSB
+FSR – 1 LSB
+FSR – 1.5 LSB
ANALOG INPUT
ADC CODE (STRAIGHT BINARY)
15060-056
Figure 53. ADC Ideal Transfer Function
Table 11. Output Codes and Ideal Input Voltages
Analog Input1
Description VREF = 5 V Digital Output Code (Hex)
FSR – 1 LSB 4.999924 V 0xFFFF2
Midscale + 1 LSB 2.500076 V 0x8001
Midscale 2.5 V 0x8000
Midscale – 1 LSB 2.499924 V 0x7FFF
–FSR + 1 LSB 76.3 μV 0x0001
–FSR 0 V 0x00003
1 The ADAQ7980/ADAQ7988 ADC driver in the unity-gain buffer configuration.
2 This is also the code for an overranged analog input (IN+ − IN− above VREF − VGND).
3 This is also the code for an underranged analog input (IN+ − IN− below VGND).
TYPICAL CONNECTION DIAGRAM
Figure 54 shows an example of the recommended connection
diagram for the ADAQ7980/ADAQ7988 when multiple supplies
are available.
REF
GND
VDD
VIO
SDI
SCK
SDO
CNV
20Ω
V+
V–
1.8nF
10µF
LDO
2.2µF
REF_OUT LDO_OUT
PD_REF
2
AMP_OUT
PD_AMP
2
NEGATIVE
SUPPLY
POSITIVE
SUPPLY
REF
1
PD_LDO
2
ADC
ADCN
IN+
IN–
1.8V TO 5V
100nF
100nF
100nF 2.2µF
0
V TO
VREF
1
SEE THE VOLTAGE REFERENCE INPUT SECTION FOR REFERENCE SELECTION.
2
POWER DOWN PINS CONNECTED TO EITHER DIGITAL HOST OR POSITIVE SUPPLY.
15060-057
Figure 54. Typical Application Diagram with Multiple Supplies
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 28 of 49
ADC DRIVER INPUT
The ADC driver of the ADAQ7980/ADAQ7988 features a −3 dB
bandwidth of 35 MHz and a slew rate of 110 V/μs at G = +1
and VAMP_OUT = 2 V step. It features an input voltage noise of
5.9 nV/√Hz .The driver can operate over a supply voltage range
of 3.8 V to 10 V and consumes only 500 μA of supply current at a
supply difference of 5 V. The low end of the supply range allows
−5% variation of a 4 V supply. The amplifier is unity-gain stable,
and the input structure results in an extremely low input voltage
noise 1/f corner. The ADC driver uses a slew enhancement
architecture, as shown in Figure 55. The slew enhancement
circuit detects the absolute difference between the two inputs. It
then modulates the tail current, ITAIL, of the input stage to boost
the slew rate. The architecture allows a higher slew rate and a
faster settling time with a low quiescent current while maintaining
low noise. The user has access to all three amplifier signal pins,
providing flexibility to adapt to the desired application or
configuration.
IN+
V
IN+
V
IN–
V+
INPUT
STAGE
TO DETECT
ABSOLUTE
VALUE
SLEW ENHANCEMENT CIRCUIT
I
TAIL
IN–
15060-058
Figure 55. ADC Driver Slew Enhancement Circuit
INPUT PROTECTION
The amplifier is fully protected from ESD events, withstanding
human body model ESD events of 4000 V and field induced
charged device model events of 1250 V with no measured
performance degradation. The precision input is protected with
an ESD network between the power supplies and diode clamps
across the input device pair, as shown in Figure 56.
IN+
ESD
ESD
V–
V
+
BIAS
TO THE REST OF THE AMPLIFIER
IN–
ESD
ESD
15060-059
Figure 56. ADC Driver Input Stage and Protection Diodes
For differential voltages more than approximately 1.2 V at room
temperature and 0.8 V at 125°C, the diode clamps begin to
conduct. If large differential voltages must be sustained across
the input terminals, the current through the input clamps must
be limited to less than 10 mA.
External series input resistors that are sized appropriately for the
expected differential overvoltage can provide the needed
protection.
The ESD clamps begin to conduct for input voltages that are
more than 0.7 V above the positive supply and input voltages
more than 0.7 V below the negative supply. If an overvoltage
condition is expected, the input current must be limited to less
than 10 mA.
Along with the ADC driver inputs, protection is also provided on
the ADC input. As shown in Figure 1, the ADAQ7980/ADAQ7988
house an RC filter between the ADC driver and the ADC. The
series resistor in this low-pass filter acts to limit current in an
overvoltage condition. The current sink capability of the reference
buffer works to hold the reference node at its desired value when
the ADC input protection diodes conduct due to an overvoltage
event.
Figure 57 shows an equivalent ADC analog input circuit of the
ADAQ7980/ADAQ7988.
The two diodes, D1 and D2, provide ESD protection for the ADC
inputs. Take care to ensure that the ADC analog input signal
never exceeds the reference value by more than 0.3 V or drops
below ground by more than 0.3 V because this causes diodes to
become forward-biased and start conducting current. These diodes
can handle a forward-biased current greater than or equal to the
short-circuit current of the ADC driver. For instance, these
conditions can occur when the ADC driver positive supply is
greater than the reference value. In such a case (for example, an
input buffer with a short circuit), use the current limitation to
protect the devices.
REF
R
IN
C
IN
IN+
OR IN–
GND
D2C
PIN
D1
15060-060
Figure 57. Equivalent ADC Analog Input Circuit
The analog input structure allows the sampling of the true
differential signal between the ADC input pins. By using these
differential inputs, signals common to both inputs are rejected.
NOISE CONSIDERATIONS AND SIGNAL SETTLING
The ADC driver of the ADAQ7980/ADAQ7988 is ideal for
driving the on-board high resolution SAR ADC. The low input
voltage noise and rail-to-rail output stage of the driver helps to
minimize distortion at large output levels. With its low power of
500 μA, the amplifier consumes power that is compatible with
the low power SAR ADC. Furthermore, the ADC driver supports a
single-supply configuration; the input common-mode range
extends to the negative supply, and 1.3 V below the positive
supply.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 29 of 49
Figure 58 illustrates the primary noise contributors for the
typical gain configurations. The total output noise (vn_out) is the
root sum square of all the noise contributions.
R
G
RS
in–
RF
vn
4kTRS
vn_RS =
4kTRG
vn_RG =
vn_RF =
+ vn_out –
4kTRF
in+
15060-061
Figure 58. Noise Sources in Typical Connection
Calculate the output noise spectral density of the ADC driver by
[ ]
2
2
2
2
2
2
2
_
44
1
4F
n
G
G
F
n
S
n
G
F
F
out
n
RikTR
R
R
vRikTRs
R
R
kTR
v
−+ +
+++
+
+
=
where:
k is the Boltzmann constant.
T is the absolute temperature in degrees Kelvin.
RF and RG are the feedback network resistances, as shown in
Figure 58.
RS is the source resistance, as shown in Figure 58.
in+ and in− represent the amplifier input current noise spectral
density in pA/√Hz.
vn is the amplifier input voltage noise spectral density in
nV/√Hz.
For more information on these calculations, see MT-049 and
MT-050.
Source resistance noise, amplifier input voltage noise (vn), and
the voltage noise from the amplifier input current noise
(in+ × RS) are all subject to the noise gain term (1 + RF/RG).
Figure 59 shows the total referred to input (RTI) noise due
to the amplifier vs. the source resistance. Note that with a
5.9 nV/√Hz input voltage noise and 0.6 pA/√Hz input current
noise, the noise contributions of the amplifier are relatively
small for source resistances from approximately 2.6 kΩ to 47 kΩ.
The Analog Devices, Inc., silicon germanium (SiGe) bipolar
process makes it possible to achieve a low voltage noise. This
noise is much improved compared to similar low power amplifiers
with a supply current in the range of hundreds of microamperes.
1
10
100
1k
100 1k 10k 100k 1M
RTI NOISE (nV/√Hz)
SOURCE RESISTANCE (Ω)
TOTAL NOISE
SOURCE RE S ISTANCE NO ISE
AMPLIFIER NOISE
SOURCE RESISTANCE = 2.6kΩ
SOURCE RESISTANCE = 47kΩ
15060-062
Figure 59. RTI Noise vs. Source Resistance
Keep the noise generated by the driver amplifier, and its
associated passive components, as low as possible to preserve
the SNR and transition noise performance of the ADAQ7980/
ADAQ7988. The analog input circuit of the ADAQ7980/
ADAQ7988 features a one-pole, low-pass filter to band limit the
noise coming from the ADC driver. Because the typical noise of
the ADAQ7980/ADAQ7988 is 44.4 µV rms in the dual-supply
typical configuration, the SNR degradation due to the amplifier is
+
=
−2
dB3
2
)(
2
π
44.4
44.4
log20
N
LOSS
Nef
SNR
where:
f–3 dB is the cutoff frequency of the input filter (4.4 MHz).
N is the noise gain of the amplifier (for example, 1 in a buffer
configuration).
eN is the equivalent input noise voltage of the op amp, in
nV/√Hz.
For multichannel multiplexed applications, the analog input
circuit of the ADAQ7980/ADAQ7988 must settle a full-scale
step onto the capacitor array at a 16-bit level (0.0015%, 15 ppm)
within one conversion period. As shown in Figure 20, the
bandwidth of the ADC driver changes with the gain setting
implemented. The ADC driver must maintain a sufficient
bandwidth to allow the ADC input to settle properly. The RC
time constant of the low-pass filter of the ADAQ7980/ADAQ7988
has been set to settle the anticipated SAR ADC charge redistribution
voltage step from a full-scale ADC input voltage transition within
the minimum acquisition phase of the ADC. The maximum
full-scale step is based upon the maximum reference input
voltage of 5.1 V. The reference sets the maximum analog input
range and subsequently the range of voltages that the ADC can
quantize.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 30 of 49
During the conversion process, the capacitive DAC of the SAR
ADC disconnects from the ADC input. In a multiplexed
application, the multiplexer input channel switches during the
conversion time to provide the maximum settling time. At the
end of the conversion time, the capacitive DAC then connects
back to the input. During this time, the DAC is disconnected
from the ADC input, and a voltage change occurs at the ADC
input node. The voltage step observed at the ADC analog input
resulting from capacitive charge redistribution attenuates due to
the voltage divider created by the parallel combination of the
capacitive DAC and the capacitor in the external low-pass filter.
Calculate the voltage step by
VSTEP = (VREF × 30 pF)/(30 pF + 1800 pF) = VREF × 0.016
For a 5.0 V reference, this results in a maximum step size of
82 mV. To calculate the required filter and ADC driver bandwidth,
determine the number of time constants required to settle this
voltage step within the ADC acquisition phase as follows:
=
+116
2
ln
REF
STEP
TC V
V
N
With the number of time constants known, determine the RC
time constant (τ) by τ = 290 ns/NTC. The minimum acquisition
phase of the ADC is 290 ns. Signals must be fully settled within
this acquisition period.
Calculate the filter bandwidth (BW) by BW = 1/(2π × τ).
The ADC driver small signal bandwidth must always remain
greater than or equal to the bandwidth previously calculated.
When the small signal bandwidth reduces, for example in the
presence of a large voltage gain, increase the acquisition phase
to increase the required system τ. An increase in acquisition
phase results in a reduction of the maximum sample rate.
The method previously described assumes the multiplexer switches
shortly after the conversion begins and that the amplifier and
RC have a large enough bandwidth to sufficiently settle the low-
pass filter capacitor before acquisition begins.
During forward settling, approximately 11 time constants are
required to settle a full-scale step to 16 bits. For the low-pass RC
filter housed in the ADAQ7980/ADAQ7988, the forward settling
time of the filter is 11 × 36 ns ≈ 400 ns, which is much less than
the conversion time of 710 ns/1200 ns, respectively. To achieve
an ADC driver forward settling time of less than 710 ns, maintain
an ADC driver large signal bandwidth of 2.49 MHz. Calculate
this as follows:
ADC Driver Forward Settling Time Constant =
710 ns/ln(216) = 64 ns
Minimum ADC Driver Large Signal Bandwidth =
1/(2 π × 64 ns) = 2.49 MHz
The forward settling does not necessarily have to occur during
the conversion time (before the capacitive DAC gets switched to
the input), but the combined forward and reverse settling time
must not exceed the required throughput rate. Forward settling
is less important for low frequency inputs because the rate of
change of the signal is much lower. The importance of which
bandwidth specification of the ADC driver is used is dependent
upon the type of input. Focus high frequency (>100 kHz) or
multiplexed applications on the large signal bandwidth, and
concentrate lower input frequency applications on the ADC
driver small signal bandwidth when performing the previous
calculations.
CONVERSION
MUX CHANNE L SW IT CH
CNV
ADC THROUGHP UT
t
CYC
t
CONV
t
ACQ
ACQUISITIONACQUISITION
ADC INP UT
NEGATIVE FS
POSITIVE FS
REVERSE
SETTLING
CAPACI TIV E DAC S WI TCH TO ACQ UIRE
FORWARD
SETTLING
15060-063
Figure 60. Multiplexed Application Timing
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 31 of 49
PD_AMP OPERATION
Figure 61 shows the ADC driver and reference buffer shutdown
circuitry. To maintain a low supply current in shutdown mode,
no internal pull-up circuitry exists; therefore, drive the PD_AMP
pin high or low externally and do not leave it floating. Pulling the
PD_AMP pin to ≥1 V below midsupply turns the device off,
reducing the supply current to 2.9 µA for a 5 V supply. When
the amplifier powers down, its output enters a high impedance
state. The output impedance decreases as frequency increases. In
shutdown mode, a forward isolation of −62 dB can be achieved
at 100 kHz (see Figure 51).
V+
V–
PD_AMP
ESD
ESD
2.2Ω
1.8Ω
1.1V
TO ENABL E
AMPLIFIER
15060-073
Figure 61. Shutdown Circuit
ESD clamps protect the PD_AMP pin, as shown in Figure 61.
Voltages beyond the power supplies cause these diodes to conduct.
To protect the PD_AMP pin, ensure that the voltage to this pin
does not exceed 0.7 V above the positive supply or 0.7 V below
the negative supply. If expecting an overvoltage condition, limit
the input current to less than 10 mA with a series resistor.
Table 12 summarizes the threshold voltages for the powered down
and enabled modes for various supplies. For any supply voltage,
pulling the PD_AMP pin to ≥1 V below midsupply turns the
device off.
Table 12. Threshold Voltages for Powered Down and
Enabled Modes
Mode
V+/V−
+4 V/0 V +5 V/0 V +7 V/−2 V
Enabled >+1.4 V >+1.9 V >+1.9 V
Powered Down <+1.0 V <+1.5 V <+1.5 V
DYNAMIC POWER SCALING (DPS)
One of the merits of a SAR ADC is that its power scales with
the sampling rate. This power scaling makes SAR ADCs very
power efficient, especially when running at lower sampling
frequencies. Traditionally, the ADC driver associated with the
SAR ADC consumes constant power, regardless of the sampling
frequency. The ADC driver allows dynamic power scaling. This
feature allows the user to provide a periodic signal to the power-
down pin of the ADC driver that is synchronized to the convert
start signal, thus scaling the system power consumption with
the sample rate.
Figure 62 illustrates the method by which the sampling rate of
the system dynamically scales the quiescent power of the ADC
driver. By providing properly timed signals to the convert start
(CNV) pin of the ADC and the PD_AMP pins of the ADC
driver, both devices run at optimum efficiency.
REF
GND
VDD
VIO
SDI
SCK
SDO
CNV
20Ω
V+
V–
1.8nF
10µF LDO
2.2µF
REF_OUT LDO_OUT
PD_REF
AMP_OUT
PD_AMP
PD_LDO
ADC
ADCN
IN+
IN–
ADAQ7980/
ADAQ7988
TIMING
GENERATOR
15060-065
Figure 62. Power Management Circuitry
Figure 63 illustrates the relative signal timing for power scaling
the ADC driver and the ADC. To prevent degradation in the
performance of the ADC, the ADC driver must have a fully
settled output into the ADC before the activation of the
CNV pin. In this example, the amplifier is switched to full
power mode 3 µs prior to the rising edge of the CNV signal. The
PD_AMP pin of the ADC driver is pulled low when the ADC
input is inactive in between samples. The quiescent current of
the amplifier typically falls to 10% of the normal operating value
within 0.9 µs at a supply difference of 5 V. While in shutdown
mode, the ADC driver output impedance is high.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 32 of 49
SAMPLI NG F REQUENCY = 100kHz
tS = 10µs
ACQUISITION ACQUISITION ACQUISITION
SHUTDOWN
3µs 3µs
POWERED
ON
POWERED
ON
POWERED
ON SHUTDOWN SHUTDOWN
Vf1
tf1tTURNOFF1 tf2
Vf2
tTURNOFF2
tAMP, ON
Vf3
tf3 tTURNOFF3
ADC
MODE
CNV
PD_AMP
CONVERSION CONVERSION CONVERSION
MINIMUM
POWERED ON TIME = 3µs
ADC DRIVE R
OUTPUT
15060-066
Figure 63. Timing Waveforms
Figure 64 shows the quiescent power of the ADC driver with
and without the power scaling. Without power scaling, the
amplifier constantly consumes power regardless of the sampling
frequency, as shown in the following equation.
PQ = IQ × VS
With power scaling, the quiescent power becomes proportional
to the ratio of the amplifier on time (tAMP, ON) and the sampling
time (tS).
PQ = IQ × VS × (tAMP, ON/tS)
Thus, by dynamically switching the driver between shutdown
and full power modes during the sample period, the quiescent
power of the driver scales with the sampling rate.
1.2
1.0
0.4
0.8
0.2
0.6
011M100k100 1k 10k10
QUI E S CE NT CURRENT (mA)
SAMPLI NG FRE QUENCY (
f
S
)
AMP Q UIES CE NT CURRENT NO DPS
AMP Q UIES CE NT CURRENT WI TH DPS
ADC CURRENT DRAW
PD_AMP ON
TIME = 3µs
V+ = 5V
V– = 0V
15060-067
Figure 64. Quiescent Current of the ADC Driver vs. ADC Sampling Frequency
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 33 of 49
SLEW ENHANCEMENT
The ADC driver has an internal slew enhancement circuit that
increases the slew rate as the feedback error voltage increases.
This circuit improves the amplifier settling response for a large
step, as shown in Figure 65. This improvement in settling response
is useful in applications where the multiplexing of multiple input
signals occurs.
–0.5
2.5
2.0
1.5
1.0
0.5
0
0 20 40 60 80 100 120 140
OUTPUT VOLTAGE (V)
TIME (ns)
15060-266
500mV p-p
1V p-p
2V p-p
Figure 65. Step Response with Selected Output Steps
EFFECT OF FEEDBACK RESISTOR ON FREQUENCY
RESPONSE
The amplifiers input capacitance and feedback resistor form a
pole that, for larger value feedback resistors, can reduce phase
margin and contribute to peaking in the frequency response.
Figure 66 shows the peaking for 500 feedback resistors (RF)
when the amplifier is configured in a gain of +2. Figure 66 also
shows how peaking can mitigate with the addition of a small
value capacitor placed across the feedback resistor of the
amplifier.
–6
9
6
3
0
–3
0.1 1 10 100
CLOSED-LOOP GAIN (dB)
FREQUENCY (MHz)
15060-267
8pF
0pF
G = +2
RF, RG = 500Ω
VOUT = 200mV p-p
Figure 66. Peaking Mitigation in Small Signal Frequency Response
VOLTAGE REFERENCE INPUT
The ADAQ7980/ADAQ7988 voltage reference input (REF) is
the noninverting node of the on-board low noise reference
buffer. The reference buffer is included to optimally drive the
dynamic input impedance of the SAR ADC reference node.
Also housed in the ADAQ7980/ADAQ7988 is a 10 F decoupling
capacitor that is ideally laid out within the devices. This decoupling
capacitor is a required piece of the SAR architecture. The
REF_OUT capacitor is not just a bypass capacitor. This capacitor is
part of the SAR ADC that simply cannot fit on the silicon.
During the bit decision process, because the bits are settled in a
few 10s of nanoseconds or faster, the storage capacitor replenishes
the charge of the internal capacitive DAC. As the binary bit
weighted conversion is processed, small chunks of charge are
taken from the 10 µF capacitor. The internal capacitor array is a
fraction of the size of the decoupling capacitor, but this large
value storage capacitor is required to meet the SAR bit decision
settling time.
There is no need for an additional lower value ceramic decoupling
capacitor (for example, 100 nF) between the REF_OUT and
GND pins.
The reference value sets the maximum ADC input voltage that
the SAR capacitor array can quantize. The reference buffer is
set in the unity-gain configuration; therefore, the user sets the
reference voltage value with the REF pin and observes this value
at the REF_OUT pin. The user is responsible for selecting a
reference voltage value that is appropriate for the system under
design. Allowable reference values range from 2.4 V to 5.1 V;
however, do not violate the input common-mode voltage range
specification of the reference buffer.
With the inclusion of the reference buffer, the user can implement a
much lower power reference source than many traditional SAR
ADC signal chains because the reference source drives a high
impedance node instead of the dynamic load of the SAR capacitor
array. Root sum square the reference buffer noise with the
reference source noise to arrive at a total noise estimate. Generally,
the reference buffer has a noise density much less than that of
the reference source.
ADC
RG
BUFFER
VOLTAGE
REFERENCE
OPTIONAL
FILTER
ADAQ7980
15060-072
Figure 67. Voltage Reference with RC Filtering
As shown in Figure 67, place a passive, RC low-pass filter with a
very low cutoff frequency between the reference source and the
REF pin of the ADAQ7980/ADAQ7988 to band limit noise from
the reference source.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 34 of 49
This filtering can be useful, considering the voltage reference
source is usually the dominant contributor to the noise of the
reference input circuit. Filters with extremely low bandwidths
can be used since the reference signal is a dc type signal.
However, because with such low frequency cutoffs, the settling
time at power on is quite large. For example, a single pole, low-
pass filter with a −3 dB bandwidth of 20 Hz has a time constant
of approximately 8 ms.
Just like the ADC driver, the reference buffer features a PD_REF
pin that allows the user to control the power consumption of the
ADAQ7980/ADAQ7988. A timing scheme similar to Figure 63
can be implemented for the PD_REF pin. Also, use the PD_REF
feature during long idle periods where extremely low power
consumption is desired.
Figure 68 shows the reference buffer shutdown circuitry. To
maintain very low supply current in shutdown mode, do not
supply the internal pull-up resistor; therefore, the drive PD_REF
pin high or low externally and do not leave it floating. Pulling the
PD_REF pin to ≥1 V below midsupply turns the device off,
reducing the supply current to 2.9 μA for a 5 V supply. When the
amplifier powers down, its output enters a high impedance
state. The output impedance decreases as frequency increases. In
shutdown mode, a forward isolation of −80 dB can be achieved
at frequencies below 10 kHz (see Figure 51).
V
+
V–
PD_REF
ESD
ESD
2.2Ω
1.8Ω
1.1V
TO ENABLE
AMPLIFIER
15060-064
Figure 68. Reference Buffer Shutdown Circuit
ESD clamps protect the PD_REF pin, as shown in Figure 68.
Voltages beyond the power supplies cause these diodes to conduct.
To protect the PD_REF pin, ensure that the voltage to this pin
does not exceed 0.7 V above the positive supply or 0.7 V below
the negative supply. When expecting an overvoltage condition,
limit the input current to less than 10 mA with a series resistor.
Table 13 summarizes the threshold voltages for the powered down
and enabled modes for various supplies. For any supply voltage,
pulling the PD_REF pin to ≥1 V below midsupply turns the
device off.
Table 13. Threshold Voltages for Powered Down and
Enabled Modes
Mode
V+/V−
+4 V/0 V +5 V/0 V +7 V/−2 V
Enabled >+1.4 V >+1.9 V >+1.9 V
Powered Down <+1.0 V <+1.5 V <+1.5 V
If more than one ADAQ7980/ADAQ7988 is used in a system,
for example, in a daisy-chain configuration, it is possible to use
the reference buffer of one ADAQ7980/ADAQ7988 to provide the
REF_OUT signal for multiple ADAQ7980/ADAQ7988 devices.
Enabling the PD_REF pin of the reference buffer places the
reference buffer output in a high impedance state. The active
reference buffer can drive the subsequent REF_OUT nodes. See
Figure 69 for connection details.
The sample rate of each individual converter determines the
number of ADAQ7980/ADAQ7988 references that can be
chained together. Each ADAQ7980/ADAQ7988 SAR ADC
reference consumes 330 μA of load current at a reference input
of 5 V and with the converter running at 1 MSPS. This current
consumption scales linearly with sample rate. For example,
reducing the sample rate to 100 kSPS reduces the reference
current draw to 33 μA. The active reference buffer must regulate
the cumulative current draw well enough so that the reference
voltage does not change by more than ½ of an LSB. An unperceived
change in reference value manifests as a gain error.
ADC
R
G
BUFFER
VOLTAGE
REFERENCE
OPTIONAL
FILTER
ADAQ7980/
ADAQ7988
DEVICE 1
ADAQ7980/
ADAQ7988
DEVICE 2
ADC
R
G
BUFFER
ADAQ7980/
ADAQ7988
DEVICE 3
ADC
R
G
BUFFER
PD_REF 1 PD_REF 2 PD_REF 3 POWERED
OFF
POWERED
OFF
POWERED
ON
15060-074
Figure 69. Reference Configuration for Multiple ADAQ7980/ADAQ7988 Devices
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 35 of 49
POWER SUPPLY
Power supply bypassing is a critical aspect in the performance
of the ADC driver. A parallel connection of capacitors from
each amplifier power supply pin (V+ and V−) to ground works
best. Smaller value ceramic capacitors offer improved high
frequency response, whereas larger value ceramic capacitors
offer improved low frequency performance.
Paralleling different values and sizes of capacitors helps to ensure
that the power supply pins are provided with a low ac impedance
across a wide band of frequencies. Parralleling is important for
minimizing the coupling of noise into the amplifier—especially
when the amplifier PSRR begins to roll off—because the bypass
capacitors can help lessen the degradation in PSRR
performance.
Place the smallest value capacitor on the same side of the board
as the ADAQ7980/ADAQ7988 and as close as possible to the
amplifier power supply pins. Connect the ground end of the
capacitor directly to the ground plane.
The ADAQ7980/ADAQ7988 feature two other power supply pins:
the input to the LDO regulator that supplies the ADC (VDD) and a
digital input/output interface supply (VIO). VIO allows direct
interface with any logic between 1.8 V and 5.0 V. The ADAQ7980/
ADAQ7988 are independent of power supply sequencing between
VIO and VDD. It is recommended to provide power to VIO and
VDD prior to V+ and V−. In addition, while not required, it is
recommended to place the ADC driver and reference buffer in
power-down by applying a logic low to the PD_AMP and
PD_REF pins during the power-on sequence of the ADAQ7980/
ADAQ7988. The following are the recommended sequences for
applying and removing power to the μModule data acquisition
systems.
The recommended dual-supply, power-on sequence follows:
1. Apply a logic low to PD_AMP, PD_REF, and PD_LDO.
2. Apply a voltage to VIO.
3. Apply a voltage to VDD.
4. Apply a logic high to PD_LDO.
5. Apply a voltage to V+ and V−.
6. Apply a logic high to PD_AMP and PD_REF.
The recommended single-supply, power-on sequence follows:
1. Apply a logic low to PD_AMP, PD_REF, and PD_LDO.
2. Apply a voltage to VIO.
3. Apply a voltage to VDD and V+.
4. Apply a logic high to PD_LDO.
5. Apply a logic high to PD_AMP and PD_REF.
The recommended dual-supply, power-down sequence follows:
1. Apply a logic low to PD_AMP and PD_REF.
2. Remove the voltage from V+ and V−.
3. Apply a logic low to PD_LDO.
4. Remove the voltage from VDD.
5. Remove the voltage from VIO.
The recommended single-supply, power-down sequence follows:
1. Apply a logic low to PD_AMP and PD_REF.
2. Apply a logic low to PD_LDO.
3. Remove the voltage from V+ and VDD.
4. Remove the voltage from VIO.
Additionally, the ADAQ7980/ADAQ7988 are insensitive to power
supply variations over a wide frequency range, as shown in
Figure 70.
80
55 11000
FRE QUENCY ( kHz )
PSRR ( dB)
10 100
75
70
65
60
15060-075
Figure 70. PSRR vs. Frequency
The VDD input is the input of an on-board LDO regulator that
supplies 2.5 V to the SAR ADC. By housing an LDO regulator,
the ADAQ7980/ADAQ7988 provide a wide supply range to the
user. When operating these devices in a single-supply
configuration, tie the V+ and VDD pins together and connect
the V− pin to ground. Refer to Table 4 for the full list of
operating requirements associated with a single-supply system.
The LDO regulator of the ADAQ7980/ADAQ7988 is a 2.5 V,
low quiescent current, linear regulator that operates from 3.5 V
to 10 V and provides up to 100 mA of output current. The LDO
regulator draws a low 180 µA of quiescent current (typical) at
full load. The typical shutdown current consumption is less than
3 µA at room temperature. Typical start-up time for the LDO
regulator is 380 µs.
The ADAQ7980/ADAQ7988 require a small 2.2 µF ceramic
capacitor connected between the VDD pin and ground. Any
quality ceramic capacitors can be used as long as they meet the
minimum capacitance and maximum equivalent series resistance
(ESR) requirements. Ceramic capacitors are manufactured with
a variety of dielectrics, each with different behavior over
temperature and applied voltage. Capacitors must have a
dielectric adequate to ensure the minimum capacitance over the
necessary temperature range and dc bias conditions. X5R or
X7R dielectrics with a voltage rating of 6.3 V to 100 V are
recommended. Y5V and Z5U dielectrics are not recommended
due to their poor temperature and dc bias characteristics.
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 36 of 49
Internally, the LDO regulator consists of a reference, an error
amplifier, a feedback voltage divider, and a positive metal-oxide
semiconductor (PMOS) pass transistor. The PMOS pass device,
which is controlled by the error amplifier, delivers the output
current. The error amplifier compares the reference voltage with
the feedback voltage from the output and amplifies the difference.
If the feedback voltage is lower than the reference voltage, the
gate of the PMOS device pulls lower, allowing more current to
pass and increasing the output voltage. If the feedback voltage is
higher than the reference voltage, the gate of the PMOS device pulls
higher, allowing less current to pass and decreasing the output
voltage.
The LDO regulator uses the PD_LDO pin to enable and disable
the LDO_OUT pin under normal operating conditions. When
PD_LDO is high, LDO_OUT turns on, and when PD_LDO is low,
LDO_OUT turns off. For automatic startup, tie PD_LDO to VDD.
Only apply a logic low to PD_LDO if a logic low is applied to
PD_AMP and PD_REF as well.
LDO REGULATOR CURRENT-LIMIT AND THERMAL
OVERLOAD PROTECTION
The current and thermal overload protection circuits protect
the LDO regulator of the ADAQ7980/ADAQ7988 against damage
due to excessive power dissipation. The LDO regulator current
limits when the output load reaches 360 mA (typical). When
the output load exceeds the current limit threshold, the output
voltage reduces to maintain a constant current limit.
Thermal overload protection is included, which limits the LDO
regulator junction temperature to a maximum of 150°C (typical).
Under extreme conditions (that is, high ambient temperature
and/or high power dissipation), when the junction temperature
starts to rise above 150°C, the output turns off, reducing the
output current to zero. When the junction temperature drops
below 135°C, the output turns on again, and the output current
restores to its operating value.
Consider the case where a hard short circuit from LDO_OUT
to ground occurs. At first, the LDO regulator limits the current
threshold that can be conducted into the short circuit. If self
heating of the junction is enough to cause its temperature to rise
above 150°C, thermal shutdown activates, turning off the output
and reducing the output current to zero. As the junction
temperature cools and drops below 135°C, the output turns on
and conducts the current limit into the short, again causing the
junction temperature to rise above 150°C. This thermal oscillation
between 135°C and 150°C causes a current oscillation between
the maximum current and 0 mA that continues as long as the
short circuit remains at the output.
Current-limit and thermal limit protections protect the device
against accidental overload conditions. For reliable operation,
externally limit the power dissipation of the devices so that the
junction temperature does not exceed 125°C.
LDO REGULATOR THERMAL CONSIDERATIONS
In applications with a low, input to output voltage differential,
the LDO regulator does not dissipate much heat. However, in
applications with high ambient temperature and/or high input
voltage, the heat dissipated in the package may become large
enough to cause the junction temperature of the die to exceed
the specified junction temperature of 125°C.
When the junction temperature exceeds 150°C, the LDO
regulator enters thermal shutdown. It recovers only after the
junction temperature decreases below 135°C to prevent any
permanent damage. Therefore, thermal analysis for the chosen
application is important to guarantee reliable performance over
all conditions. To guarantee specified operation, the junction
temperature of the LDO regulator must not exceed 125°C. To
ensure that the junction temperature stays below this value, the
user must be aware of the parameters that contribute to junction
temperature changes. These parameters include ambient
temperature, power dissipation in the power device, and thermal
resistances between the junction and ambient air (θJA). The θJA
number is dependent on the package assembly compounds used
and the amount of material used to solder the package GND pins
to the PCB.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 37 of 49
DIGITAL INTERFACE
Though the ADAQ7980/ADAQ7988 have a reduced number of
pins, they offer flexibility in their serial interface modes.
The ADAQ7980/ADAQ7988, when in CS mode, are compatible
with SPI, QSPI™, and digital hosts. This interface can use either a
3-wire or 4-wire interface. A 3-wire interface using the CNV,
SCK, and SDO signals minimizes wiring connections useful, for
instance, in isolated applications. A 4-wire interface using the
SDI, CNV, SCK, and SDO signals allows CNV, which initiates
the conversions, to be independent of the readback timing
(SDI). This independence is useful in low jitter sampling or
simultaneous sampling applications.
The ADAQ7980/ADAQ7988, when in chain mode, provide a
daisy-chain feature using the SDI input for cascading multiple
ADCs on a single data line similar to a shift register.
The mode in which these devices operate depends on the SDI
level when the CNV rising edge occurs. To select CS mode, set
SDI high, and to select chain mode, set SDI low. The SDI hold
time is such that when SDI and CNV are connected together,
chain mode is selected.
In either mode, the ADAQ7980/ADAQ7988 offer the flexibility
to optionally force a start bit in front of the data bits. This start
bit can be used as a busy signal indicator to interrupt the digital
host and trigger the data reading. Otherwise, without a busy
indicator, the user must time out the maximum conversion time
prior to readback.
The busy indicator enables
• In CS mode if CNV or SDI is low when the ADC
conversion ends (see Figure 74 and Figure 78).
• In chain mode if SCK is high during the CNV rising edge
(see Figure 82).
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 38 of 49
3-WIRE CS MODE WITHOUT THE BUSY INDICATOR
To connect a single ADAQ7980/ADAQ7988 to an SPI-compatible
digital host, use 3-wire CS mode without the busy indicator.
Figure 71 shows the connection diagram, and Figure 72 shows
the corresponding timing.
With SDI tied to VIO, a rising edge on CNV initiates a conversion,
selects CS mode, and forces SDO to high impedance. After a
conversion initiates, it continues until completion irrespective
of the state of CNV, which is useful, for instance, to bring CNV
low to select other SPI devices, such as analog multiplexers.
However, before the minimum conversion time elapses, return
CNV high and then hold it high for the maximum conversion
time to avoid the generation of a busy signal indicator. When
the conversion completes, the ADAQ7980/ADAQ7988 enter the
acquisition phase and power down.
When CNV goes low, the MSB is output onto SDO. Then, the
remaining data bits clock out by subsequent SCK falling edges.
The data is valid on both SCK edges. Although the rising edge
can capture data, a digital host using the SCK falling edge
allows a faster reading rate if it has an acceptable hold time.
After the 16th SCK falling edge, or when CNV goes high,
whichever is earlier, SDO returns to high impedance.
ADAQ7980/
ADAQ7988
SDOSDI DATA IN
DIGITAL HOST
CONVERT
CLK
V
I
O
CNV
SCK
15060-076
Figure 71. 3-Wire CS Mode Without the Busy Indicator Connection Diagram
(SDI = 1, High)
SDI = 1
t
CNVH
t
CONV
t
CYC
CNV
A
CQUISITION ACQUISITION
t
ACQ
t
SCK
t
SCKL
CONVERSION
SCK
SDO D15 D14 D13 D1 D0
t
EN
t
HSDO
123 14 1516
t
DSDO
t
DIS
t
SCKH
15060-077
Figure 72. 3-Wire CS Mode Without the Busy Indicator Serial Interface Timing (SDI = 1, High)
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 39 of 49
3-WIRE CS MODE WITH THE BUSY INDICATOR
To connect a single ADAQ7980/ADAQ7988 to an SPI-compatible
digital host that has an interrupt input, use 3-wire CS mode
with the busy indicator.
Figure 73 shows the connection diagram, and Figure 74 shows
the corresponding timing.
With SDI tied to VIO, a rising edge on CNV initiates a conversion,
selects CS mode, and forces SDO to high impedance. SDO
stays in high impedance until the completion of the conversion
irrespective of the state of CNV. Prior to the minimum conversion
time, use CNV to select other SPI devices, such as analog
multiplexers; however, return CNV to low before the minimum
conversion time elapses and then hold it low for the maximum
conversion time to guarantee the generation of the busy signal
indicator.
When the conversion completes, SDO goes from high impedance
to low impedance. With a pull-up on the SDO line, use this
transition as an interrupt signal to initiate the data reading
controlled by the digital host. The ADAQ7980/ADAQ7988 then
enter the acquisition phase and power down. The data bits clock
out, MSB first, by subsequent SCK falling edges. The data is
valid on both SCK edges. Although the rising edge can capture the
data, a digital host using the SCK falling edge allows a faster
reading rate if it has an acceptable hold time. After the optional
17th SCK falling edge, or when CNV goes high, whichever is
earlier, SDO returns to high impedance.
If selecting multiple ADAQ7980/ADAQ7988 devices at the
same time, the SDO output pin handles this contention without
damage or induced latch-up. Meanwhile, it is recommended to
keep this contention as short as possible to limit extra power
dissipation.
ADAQ7980/
ADAQ7988
SDOSDI DATA IN
IRQ
DIGITAL HOST
CONVERT
CLK
VIO
VIO
47kΩ
CNV
SCK
15060-078
Figure 73. 3-Wire CS Mode with the Busy Indicator Connection Diagram
(SDI = 1, High)
t
CONV
t
CNVH
t
CYC
ACQUISITION ACQUISITION
t
ACQ
t
SCK
t
SCKH
t
SCKL
CONVERSION
SCK
CNV
SDI = 1
SDO D15 D14 D1 D0
t
HSDO
123 15 1617
t
DSDO
t
DIS
15060-079
Figure 74. 3-Wire CS Mode with the Busy Indicator Serial Interface Timing (SDI = 1, High)
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 40 of 49
4-WIRE CS MODE WITHOUT THE BUSY INDICATOR
To connecting multiple ADAQ7980/ADAQ7988 devices to an
SPI-compatible digital host, use 4-wire CS mode without the
busy indicator.
Figure 75 shows a connection diagram example using two
ADAQ7980/ADAQ7988 devices, and Figure 76 shows the
corresponding timing.
With SDI high, a rising edge on CNV initiates a conversion,
selects CS mode, and forces SDO to high impedance. In this
mode, hold CNV high during the conversion phase and the
subsequent data readback (if SDI and CNV are low, SDO is
driven low). Prior to the minimum conversion time, use SDI to
select other SPI devices, such as analog multiplexers; however,
return SDI to high before the minimum conversion time elapses
and then hold it high for the maximum conversion time to
avoid the generation of the busy signal indicator.
When the conversion completes, the ADAQ7980/ADAQ7988
enter the acquisition phase and power down. Bringing the SDI
input low reads each ADC result, which consequently outputs
the MSB onto SDO. Then, the remaining data bits clock out by
subsequent SCK falling edges. The data is valid on both SCK
edges. Although the rising edge can capture the data, a digital
host using the SCK falling edge allows a faster reading rate if it
has an acceptable hold time. After the 16th SCK falling edge, or
when SDI goes high, whichever is earlier, SDO returns to high
impedance and another ADAQ7980/ADAQ7988 can be read.
DIGITAL HOST
CONVERT
CS2
CS1
CLK
DATA IN
ADAQ7980/
ADAQ7988
SDOSDI
CNV
SCK
ADAQ7980/
ADAQ7988
SDOSDI
CNV
SCK
15060-080
Figure 75. 4-Wire CS Mode Without the Busy Indicator Connection Diagram
t
CONV
t
CYC
A
CQUISITION ACQUISITION
t
ACQ
t
SCK
t
SCKH
t
SCKL
CONVERSION
SCK
CNV
t
SSDICNV
t
HSDICNV
SDO
D15 D13D14 D1 D0 D15 D14 D1 D0
t
HSDO
t
EN
1 2 3 14 15 16 17 18 30 31 32
t
DSDO
t
DIS
SDI(CS1)
SDI(CS2)
15060-081
Figure 76. 4-Wire CS Mode Without the Busy Indicator Serial Interface Timing
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 41 of 49
4-WIRE CS MODE WITH THE BUSY INDICATOR
To connect a single ADAQ7980/ADAQ7988 to an SPI-compatible
digital host that has an interrupt input, and when keeping CNV,
which samples the analog input, independent of the signal used
to select the data reading, use 4-wire CS mode with the busy
indicator. This requirement is particularly important in
applications where low jitter on CNV is a requirement.
Figure 77 shows the connection diagram, and Figure 78 shows
the corresponding timing.
With SDI high, a rising edge on CNV initiates a conversion,
selects CS mode, and forces SDO to high impedance. In this
mode, hold CNV high during the conversion phase and the
subsequent data readback (if SDI and CNV are low, SDO is
driven low). Prior to the minimum conversion time, use SDI to
select other SPI devices, such as analog multiplexers; however,
return SDI low before the minimum conversion time elapses
and then hold it low for the maximum conversion time to
guarantee the generation of the busy signal indicator. When the
conversion completes, SDO goes from high impedance to low
impedance.
With a pull-up resistor on the SDO line, use this transition as an
interrupt signal to initiate the data readback controlled by the
digital host. The ADAQ7980/ADAQ7988 then enter the
acquisition phase and power down. The data bits clock out,
MSB first, by subsequent SCK falling edges. The data is valid on
both SCK edges. Although the rising edge can capture the data,
a digital host using the SCK falling edge allows a faster reading
rate if it has an acceptable hold time. After the optional 17th
SCK falling edge, or SDI going high, whichever is earlier, the
SDO returns to high impedance.
ADAQ7980/
ADAQ7988
SDOSDI DATA IN
IRQ
DIGITAL HOST
CONVERT
CS1
CLK
VIO
47kΩ
CNV
SCK
15060-082
Figure 77. 4-Wire CS Mode with the Busy Indicator Connection Diagram
t
CONV
t
CYC
ACQUISITION
t
SSDICNV
ACQUISITION
t
ACQ
t
SCK
t
SCKH
t
SCKL
CONVERSION
SDI
t
HSDICNV
SCK
CNV
S
DO
t
EN
D15 D14 D1 D0
t
HSDO
1 2 3 15 16 17
t
DSDO
t
DIS
15060-083
Figure 78. 4-Wire CS Mode with the Busy Indicator Serial Interface Timing
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 42 of 49
CHAIN MODE WITHOUT THE BUSY INDICATOR
To daisy-chain multiple ADAQ7980/ADAQ7988 devices on a
3-wire serial interface, use chain mode without the busy indicator.
This feature is useful for reducing component count and wiring
connections, for example, in isolated multiconverter
applications or for systems with a limited interfacing capacity.
Data readback is analogous to clocking a shift register.
Figure 79 shows a connection diagram example using two
ADAQ7980/ADAQ7988 devices, and Figure 80 shows the
corresponding timing.
When SDI and CNV are low, drive SDO low. With SCK low, a
rising edge on CNV initiates a conversion, selects chain mode,
and disables the busy indicator.
In this mode, hold CNV high during the conversion phase and
the subsequent data readback. When the conversion completes,
the MSB is output onto SDO, and the ADAQ7980/ADAQ7988
enter the acquisition phase and power down. The remaining data
bits stored in the internal shift register clock out by subsequent
SCK falling edges. For each ADC, SDI feeds the input of the
internal shift register and it clocks out by the SCK falling edge.
Each ADC in the chain outputs its data MSB first, and 16 × N
clocks are required to read back the N ADCs. The data is valid
on both SCK edges. Although the rising edge can capture the
data, a digital host using the SCK falling edge allows a faster
reading rate and, consequently, more ADAQ7980/ADAQ7988
devices in the chain, if the digital host has an acceptable hold
time. The total readback time can reduce the maximum
conversion rate.
DIGITAL HOST
CONVERT
CLK
DATA IN
ADAQ7980/
ADAQ7988 SDOSDI
CNV
A
SCK
ADAQ7980/
ADAQ7988 SDOSDI
CNV
B
SCK
15060-084
Figure 79. Chain Mode Without the Busy Indicator Connection Diagram
t
CONV
t
CYC
t
SSDISCK
t
SCKL
t
SCK
t
HSDISCK
t
ACQ
ACQUISITION
t
SSCKCNV
ACQUISITION
t
SCKH
CONVERSION
SDO
A
= SDI
B
t
HSCKCNV
SCK
CNV
SDI
A
= 0
SDO
B
t
EN
D
A
15 D
A
14 D
A
13
D
B
15 D
B
14 D
B
13 D
B
1D
B
0D
A
15 D
A
14 D
A
0D
A
1
D
A
1D
A
0
t
HSDO
123 15161714 18 30 31 32
t
DSDO
15060-085
Figure 80. Chain Mode Without the Busy Indicator Serial Interface Timing
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 43 of 49
CHAIN MODE WITH THE BUSY INDICATOR
To daisy-chain multiple ADAQ7980/ADAQ7988 devices on a
3-wire serial interface while providing a busy indicator, use
chain mode with the busy indicator. This feature is useful for
reducing component count and wiring connections, for example,
in isolated multiconverter applications or for systems with a
limited interfacing capacity. Data readback is analogous to
clocking a shift register.
Figure 81 shows a connection diagram example using three
ADAQ7980/ADAQ7988 devices, and Figure 82 shows the
corresponding timing.
When SDI and CNV are low, drive SDO low. With SCK high, a
rising edge on CNV initiates a conversion, selects chain mode,
and enables the busy indicator feature.
In this mode, hold CNV high during the conversion phase and
the subsequent data readback. When all ADCs in the chain
complete their conversions, drive the SDO pin of the ADC
closest to the digital host (see the ADAQ7980/ADAQ7988 ADC
labeled C in Figure 81) high. Use this transition on SDO as a
busy indicator to trigger the data readback controlled by the
digital host. The ADAQ7980/ ADAQ7988 then enter the
acquisition phase and power down. The data bits stored in the
internal shift register clock out, MSB first, by subsequent SCK
falling edges. For each ADC, SDI feeds the input of the internal
shift register and clocks out by the SCK falling edge. Each ADC
in the chain outputs its data MSB first, and 16 × N + 1 clocks are
required to read back the N ADCs. Although the rising edge
can capture the data, a digital host using the SCK falling edge
allows a faster reading rate and, consequently, more
ADAQ7980/ADAQ7988 devices in the chain, if the digital host
has an acceptable hold time.
ADAQ7980/
ADAQ7988
C
SDOSDI DATA IN
IRQ
DIGITAL HOST
CONVERT
CLK
CNV
SCK
ADAQ7980/
ADAQ7988
B
SDOSDI
CNV
SCK
ADAQ7980/
ADAQ7988
A
SDOSDI
CNV
SCK
15060-086
Figure 81. Chain Mode with the Busy Indicator Connection Diagram
t
CONV
t
CYC
t
SSDISCK
t
SCKH
t
SCK
t
HSDISCK
t
ACQ
t
DSDOSDI
t
DSDOSDI
t
DSDODSI
ACQUISITION
t
SSCKCNV
ACQUISITION
t
SCKL
CONVERSION
t
HSCKCNV
SCK
CNV = S DI
A
SDO
A
= SDI
B
SDO
B
= SDI
C
SDO
C
t
EN
D
A
15 D
A
14 D
A
13
D
B
15 D
B
14 D
B
13
D
C
15 D
C
14 D
C
13
D
B
1D
B
0D
A
15 D
A
14 D
A
1D
A
0
D
C
1D
C
0D
B
15 D
B
14 D
A
0D
A
1D
B
0D
B
1D
A
14D
A
15
D
A
1D
A
0
t
HSDO
123 1516174 1819 3132333435474849
t
DSDO
t
DSDOSDI
t
DSDOSDI
15060-087
Figure 82. Chain Mode with Busy Indicator Serial Interface Timing
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 44 of 49
APPLICATION CIRCUITS
Table 14 provides recommended component values at various
gains and the corresponding slew rate, bandwidth, and noise of
a given configuration. As shown in Figure 83, the noise gain,
GN, of an op amp gain block is equal to its noninverting voltage
gain, regardless of whether it is actually used for inverting or
noninverting gain. Thus,
Noninverting GN = RF/RG + 1
Inverting GN = RF/RG + 1
+
–
–
+
NONINVERTING
1R
S
G = G
N
= +5
R
F
1kΩ
R
G
249Ω
R
F
1kΩ
R
G
249Ω
G = –4
G
N
= +5
INVERTING
15060-088
Figure 83. Noise Gain of Both Equals 5
With the ADC driver, a variety of trade-offs can be made to fine
tune its dynamic performance. As with all high speed amplifiers,
parasitic capacitance and inductance around the amplifier can
affect its dynamic response. Often, the input capacitance (due to
the op amp itself, as well as the PCB) has a significant effect.
The feedback resistance, together with the input capacitance,
can contribute to a loss of phase margin, thereby affecting the
high frequency response. A capacitor (CF) in parallel with the
feedback resistor can compensate for this phase loss.
Additionally, any resistance in series with the source creates a
pole with the input capacitance (as well as dampen high
frequency resonance due to package and board inductance and
capacitance). It must also be noted that increasing resistor
values increases the overall noise of the amplifier and that
reducing the feedback resistor value increases the load on the
output stage, thus increasing distortion.
The ADC driver, which has no crossover region, has a wide linear
input range from 100 mV below ground to 1.3 V below positive
rail. The amplifier, when configured as a follower, has a linear
signal range from 150 mV above the negative supply voltage
(limited by the output stage of the amplifier) to 1.3 V below the
positive supply (limited by the amplifier input stage). If the
supply differential between V+ and V− is less than 5 V, t he
linear range of the ADC driver is reduced from 150 mV above
the negative supply voltage to 200 mV above the minus supply
voltage. A 0 V to +4.096 V signal range can be accommodated
with a positive supply as low as +5.4 V and a negative power
supply of −0.2 V. If ground is used as the amplifier negative
supply, at the low end of the input range close to ground, the
ADC driver exhibits substantial nonlinearity, as with any rail-
to-rail output amplifier.
The amplifier drives a one-pole, low-pass filter. This filter limits
the already very low noise contribution from the amplifier to
the SAR ADC.
Table 14. Recommended Component Values
Noise Gain, Noninverting Gain RS (Ω) RF (Ω) RG (Ω) CF (pF)
1 49.9 49.9 Not applicable Not applicable
1.25 49.9 249 1 k 8
2 49.9 499 499 8
5 49.9 1 k 249 8
Table 15. ADAQ7980/ADAQ7988 Performance at Selected Input Frequency with 5 V Reference Value
Results
Input Frequency (kHz) ADC Driver Gain SNR (dB) THD (dB) SINAD (dB) ENOB
1 1 91.9 −106.1 91.5 14.9
10 1 91.5 −105.0 91.0 14.8
20 1 90.7 −103.6 90.1 14.7
50 1 88.3 −99.7 87.6 14.2
100 1 84.5 −93.3 83.3 13.5
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 45 of 49
NONUNITY GAIN CONFIGURATIONS
Figure 84 shows a typical connection diagram and the major dc
error sources. The ideal transfer function (all error sources set
to 0 and infinite dc gain) can be written as
IN
G
F
IP
G
F
OUT
V
R
R
V
R
R
V×
−×
+= 1
(1)
R
G
– V
IN
+
R
S
– V
IP
+
I
B
+
I
B
–+ V
OUT
–
R
F
+ V
OS
–
15060-089
Figure 84. Typical ADC Driver Connection Diagram and DC Error Sources
This function reduces to the following familiar forms for
noninverting and inverting op amp gain expressions.
IP
G
F
OUT V
R
R
V×
+= 1
(2)
(Noninverting gain, VIN = 0 V)
IN
G
F
OUT
V
R
R
V×
−
=
(3)
(Inverting gain, VIP = 0 V)
The total output voltage error is the sum of errors due to the
amplifier offset voltage and input currents. Estimate the output
error due to the offset voltage by the following:
+×
+
−
++
=
G
F
OUTPNOM
P
OFFSET
OUT
R
R
A
V
PSRR
VV
CMRR
VCM
V
V
NOM
ERROR
1
(4)
where:
NOM
OFFSET
V
is the offset voltage at the specified supply voltage,
which is measured with the input and output at midsupply.
VCM is the common-mode voltage.
VP is the power supply voltage.
NOM
p
V
is the specified power supply voltage.
CMRR is the common-mode rejection ratio.
PSRR is the power supply rejection ratio.
A is the dc open-loop gain.
Estimate the output error due to the input currents by the
following:
+−
×
+×−
+×
=
B
G
F
S
B
G
F
G
F
OUT
I
R
R
RI
R
R
RR
VERROR
11)||(
(5)
Note that setting RS equal to RF||RG compensates for the voltage
error due to the input bias current.
Figure 85 shows the ADC driver noninverting gain connection.
The circuit was tested with multiple gain settings and an output
voltage of approximately 5 V p-p for optimum resolution and
noise performance.
REF
GND
VDD
VIO 1.8V TO 5V
100nF
SDI
SCK
SDO
CNV
20Ω
V+
V–
1.8nF
10µF LDO
2.2µF
REF_OUT
LDO_OUT
PD_REF
AMP_OUT
PD_AMP
PD_LDO
ADC
ADCN
IN+
IN–
ADAQ7980/
ADAQ7988
R
G
499Ω
2.2µF100nF REF
1
POSITIVE
SUPPLY
49.9Ω
49.9Ω
50Ω
R
F
499Ω
OPTIONAL C
F
NEGATIVE
SUPPLY
100nF
15060-090
Figure 85. Noninverting ADC Driver, Gain = 2
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 46 of 49
Table 16. Typical Ambient Temperature Performance for the ADAQ7980/ADAQ7988 for Various Gain Configurations (fIN = 10 kHz)
Gain (V/V) SNR (dB) THD (dB) SINAD (dB) SFDR (dB) ENOB (Bits)
−1 88.3 −103.4 88.0 104.5 14.3
−0.25 90.6 −96.9 90.2 102.0 14.7
1 91.5 −105 91.0 106.0 14.8
2 89.7 −103.9 89.3 102.9 14.5
The typical ambient temperature results are listed Table 16.
INVERTING CONFIGURATION WITH LEVEL SHIFT
Configuration of the ADAQ7980/ADAQ7988 to acquire bipolar
inputs is possible. For example, the device configuration can be
made such that ±10 V signals can fit the 0 V to VREF volt input
range. In this example, because a 20 V p-p signal is fit to a
smaller peak-to-peak input range, an inverting configuration
must be selected. Attenuation of the input signal requires an
inverting configuration. This configuration results in an 180o
phase shift due to the inversion.
With the SAR ADC input range being unipolar, a level shift must
be performed to fit a bipolar signal into the unipolar input of
the ADC. This level shift is performed using a difference amplifier
configuration. The resistor ratios selected for the difference
amplifier depend upon the peak-to-peak voltage of the bipolar
input signal and the reference voltage being used for the
μModule data acquisition sysem that sets the full scale of the
ADC conversion range.
G
F
T
S
T
G
F
IN
ADCP
R
R
RR
R
REF
R
R
VBipolar
V
1
ADC
10µF
R
T
R
G
20Ω
R
F
R
S
REF
1.8nF
AMP_OUT
IN–
IN+
BIPOLA
R
SOURCE
15060-091
Figure 86. Difference Amplifier Configuration Used to Fit Bipolar Signals to
the ADAQ7980/ADAQ7988
For both noninverting and inverting gain configurations, it is
often useful to increase the RF value to decrease the load on the
output. Increasing the RF value improves harmonic distortion at
the expense of reducing the bandwidth of the amplifier. Note
that as the gain increases, the small signal bandwidth decreases,
as is expected from the gain bandwidth product relationship. In
addition, the phase margin improves with higher gains, and the
amplifier becomes more stable. As a result, the peaking in the
frequency response is reduced.
The PCB layout configuration and bond pads of the chip often
contribute to stray capacitance. The stray capacitance at the
inverting input forms a pole with the feedback and gain resistors.
This additional pole adds phase shift and reduces phase margin
in the closed-loop phase response, causing instability in the
amplifier and peaking in the frequency response.
To obtain the desired bandwidth, adjust the feedback resistor,
RF. If RF cannot be adjusted, a small capacitor can be placed in
parallel with RF to reduce peaking.
The feedback capacitor, CF, forms a zero with the feedback
resistor, which cancels out the pole formed by the input stray
capacitance and the gain and feedback resistors. For the first
pass in determining the CF value, use the following equation:
RG × CS = RF × CF
where:
RG is the gain resistor.
CS is the input stray capacitance.
RF is the feedback resistor.
CF is the feedback capacitor.
Using this equation, the original closed-loop frequency response of
the amplifier is restored, as if there is no stray input capacitance.
Most often, however, the value of CF is determined empirically.
See Table 14 for recommended values.
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 47 of 49
USING THE ADAQ7980/ADAQ7988 WITH ACTIVE
FILTERS
The low noise and high gain bandwidth of the ADC driver make
it an excellent choice in active filter circuits. Most active filter
literature provides resistor and capacitor values for various filters
but neglects the effect of the finite bandwidth of the op amp on
filter performance; ideal filter response with infinite loop gain is
implied. Unfortunately, real filters do not behave in this manner.
Instead, they exhibit finite limits of attenuation, depending on
the gain bandwidth of the active device. Optimal low-pass filter
performance requires an op amp with high gain bandwidth for
attenuation at high frequencies, and low noise and high dc gain
for low frequency, pass-band performance.
Figure 87 shows the schematic of a second-order, low-pass active
filter and lists typical component values for filters having a
Bessel type response with a gain of 2 and a gain of 5.
IN+
IN–
R
F
R2R1
C1
R
G
C2
V
IN
AMP_OUT
15060-092
Figure 87. Schematic of a Second-Order, Low-Pass Active Filter
Table 17. Typical Component Values for Second-Order,
Low-Pass Active Filter of Figure 87
Gain R1 (Ω) R2 (Ω) RF (Ω) RG (Ω) C1 (nF) C2 (nF)
2 71.5 215 499 499 10 10
5 44.2 365 365 90.9 10 10
Figure 88 is a network analyzer plot of the performance of this
filter.
1k 10k 100k 1M 10M
FREQUENCY (Hz)
50
40
30
20
10
0
–10
–20
–30
–40
–50
GAIN (dB)
G = 2
G = 5
15060-093
Figure 88. Frequency Response of the Filter Circuit of Figure 87 for Two
Different Gains
ADAQ7980/ADAQ7988 Data Sheet
Rev. A | Page 48 of 49
APPLICATIONS INFORMATION
LAYOUT
Heat dissipation from the package can be improved by
increasing the amount of copper attached to the pins of the
ADAQ7980/ADAQ7988. However, a point of diminishing
returns is eventually reached, beyond which an increase in the
copper size does not yield significant heat dissipation benefits.
When designing the PCB, separate and confine the analog and
digital sections to certain areas of the PCB that houses the
ADAQ7980/ADAQ7988. The ADAQ7980/ADAQ7988 pinouts
with all their analog signals on the left side and all their digital
signals on the right side eases this task.
Avoid running digital lines under the devices because these couple
noise onto the die, unless using a ground plane under the
ADAQ7980/ADAQ7988 as a shield. Never run fast switching
signals, such as CNV or clocks, near the analog signal paths.
Avoid crossover of digital and analog signals.
Use at least one ground plane, and it can be common or split
between the digital and analog section. In the latter case, join
the planes underneath the ADAQ7980/ADAQ7988 devices.
Finally, decouple the power supplies (V+, V−, VDD, and VIO)
of the ADAQ7980/ADAQ7988 with low ESR ceramic capacitors
that are placed close to the ADAQ7980/ADAQ7988 and that are
connected using short and wide traces to provide low impedance
paths and reduce the effect of glitches on the power supply lines.
Place the smallest value capacitor on the same side of the board
as the ADAQ7980/ADAQ7988 and as close as possible to the
amplifier power supply pins. Connect the ground end of the
capacitor directly to the ground plane.
See Figure 89 for an example layout of the ADAQ7980/
ADAQ7988 that can save 50% PCB area compared to similar
designs using individual components for each section of the
μModule data acquisition system.
EVALUATING THE PERFORMANCE OF THE
ADAQ7980/ADAQ7988
The evaluation board (EVAL-ADAQ7980SDZ) user guide for
the ADAQ7980/ADAQ7988 outlines the other recommended
layouts for the ADAQ7980/ADAQ7988.
The evaluation board package includes a fully assembled and
tested evaluation board, documentation, and software for
controlling the board from a PC via the separately purchased
EVAL-SDP-CB1Z.
15060-290
ADAQ7980/
ADAQ7988
PIN 1
Figure 89. Example Layout for the ADAQ7980/ADAQ7988
Data Sheet ADAQ7980/ADAQ7988
Rev. A | Page 49 of 49
OUTLINE DIMENSIONS
09-11-2015-A
PKG-004990
BOTTOM VIEW
TOP VIEW
SIDE VIEW
2.08
1.98
1.88
1.65 REF
0.362
0.332
0.302
0.45
0.40
0.35
0.30
0.25
0.20
5.10
5.00
4.90
4.10
4.00
3.90
PIN 1
CORNER ARE A
1
5
612
13
17
18 24
0.50
BSC
0.10
REF
2.00 REF
3.00 REF
PIN 1
INDICATOR
Figure 90. 24-Lead Land Grid Array [LGA]
5 mm × 4 mm Body and 1.98 mm Package Height
(CC-24-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model1 Temperature Range Package Description Package Option
ADAQ7980BCCZ −55°C to +125°C 24-Lead Land Grid Array [LGA] CC-24-2
ADAQ7980BCCZ-RL7
−55°C to +125°C
24-Lead Land Grid Array [LGA]
CC-24-2
ADAQ7988BCCZ −55°C to +125°C 24-Lead Land Grid Array [LGA] CC-24-2
ADAQ7988BCCZ-RL7 −55°C to +125°C 24-Lead Land Grid Array [LGA] CC-24-2
EVAL-ADAQ7980SDZ Evaluation Board
EVAL-SDP-CB1Z Evaluation Controller Board
1 Z = RoHS Compliant Part.
©2017 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D15060-0-8/17(A)
