MIC2207
3mm x 3mm 2MHz 3A PWM Buck
Regulator
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
December 2006 M9999-122006
General Description
The Micrel MIC2207 is a high efficiency PWM buck (step-
down) regulators that provides up to 3A of output current.
The MIC2207 operates at 2MHz and has proprietary
internal compensation that allows a closed loop bandwidth
of over 200KHz.
The low on-resistance internal p-channel MOSFET of the
MIC2207 allows efficiencies over 94%, reduces external
components count and eliminates the need for an
expensive current sense resistor.
The MIC2207 operates from 2.7V to 5.5V input and the
output can be adjusted down to 1V. The devices can
operate with a maximum duty cycle of 100% for use in low-
dropout conditions.
The MIC2207 is available in the exposed pad 12-pin
3mm x 3mm MLF® package with a junction operating
range from –40°C to +125°C.
Features
• 2.7 to 5.5V supply voltage
• 2MHz PWM mode
• Output current to 3A
• >94% efficiency
• 100% maximum duty cycle
• Adjustable output voltage option down to 1V
• Ultra-fast transient response
• Ultra-small external components
Stable with a 1µH inductor and a 4.7µF output capacitor
• Fully integrated 3A MOSFET switch
• Micropower shutdown
• Thermal shutdown and current limit protection
• Pb-free 12-pin 3mm x 3mm MLF® package
• –40°C to +125°C junction temperature range
Applications
• 5V or 3.3V Point of Load Conversion
• Telecom/Networking Equipment
• Set Top Boxes
• Storage Equipment
• Video Cards
• DDR Power Supply
Typical Application
MIC2207
3A 2MHz Buck Regulator 80
82
84
86
88
90
92
94
96
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
3.3VOUT Efficiency
4.5VIN
5VIN 5.5VIN
Micrel, Inc. MIC2207
December 2006 2 M9999-122006
Ordering Information
Part Number Output Voltage(1) Junction Temp. Range Package Lead Finish
MIC2207YML Adj. –40° to +125°C 12-Pin 3mm x 3mm MLF® Pb-free
Note:
1. Other Voltage options available. Contact Micrel for details.
Pin Configuration
BIAS EN
SW
VIN
PGND
SGND
SW
VIN
PGND
PGOOD
5
1
2
3
4
8
FB NC
67
12
11
10
9
EP
12-Pin 3mm x 3mm MLF® (ML)
Pin Description
Pin Number Pin Name Pin Function
1,12 SW Switch (Output): Internal power P-Channel MOSFET output switch
2,11 VIN Supply Voltage (Input): Supply voltage for the source of the internal P-channel
MOSFET and driver.
Requires bypass capacitor to GND.
3,10 PGND Power Ground. Provides the ground return path for the high-side drive current.
4 SGND Signal (Analog) Ground. Provides return path for control circuitry and internal
reference.
5 BIAS Internal circuit bias supply. Must be bypassed with a 0.1µF ceramic capacitor to
SGND.
6 FB Feedback. Input to the error amplifier, connect to the external resistor divider
network to set the output voltage.
7 NC
No Connect. Not internally connected to die. This pin can be tied to any other pin
if desired.
8 EN Enable (Input). Logic level low will shutdown the device, reducing the current
draw to less than 5µA.
9 PGOOD Power Good. Open drain output that is pulled to ground when the output voltage
is within ±7.5% of the set regulation voltage
EP GND Connect to ground.
Micrel, Inc. MIC2207
December 2006 3 M9999-122006
Absolute Maximum Ratings(1)
Supply Voltage (VIN) ...................................................... +6V
Output Switch Voltage (VSW) ......................................... +6V
Output Switch Current (ISW) ........................................... 11A
Logic Input Voltage (VEN)...................................–0.3V to VIN
Storage Temperature (Ts)......................... –60°C to +150°C
ESD Rating(3) ..................................................................2kV
Operating Ratings(2)
Supply Voltage (VIN) .....................................+2.7V to +5.5V
Logic Input Voltage (VEN)........................................0V to VIN
Junction Temperature (TJ).........................–40°C to +125°C
Junction Thermal Resistance
3x3 MLF-12L (θJA) ............................................. 60°C/W
Electrical Characteristics(4)
VIN = VEN = 3.6V; L = 1µH; COUT = 4.7µF; TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C.
Parameter Condition Min Typ Max Units
Supply Voltage Range 2.7 5.5 V
Under-Voltage Lockout
Threshold
(turn-on) 2.45 2.55 2.65 V
UVLO Hysteresis 100 mV
Quiescent Current VFB = 0.9 * VNOM (not switching) 570 900 µA
Shutdown Current VEN = 0V 2 10 µA
[Adjustable] Feedback
Voltage
± 1% ILOAD = 100mA
± 2% (over temperature) ILOAD = 100mA
0.99
0.98
1 1.01
1.02
V
FB pin input current 1 nA
Current Limit in PWM Mode VFB = 0.9 * VNOM 3.5 5 A
Output Voltage
Line Regulation
VOUT > 2V; VIN = VOUT + 500mV to 5.5V; ILOAD = 100mA
VOUT < 2V; VIN = 2.7V to 5.5V; ILOAD = 100mA
0.07 %
Output Voltage
Load Regulation
20mA < ILOAD < 3A 0.2 0.5 %
Maximum Duty Cycle VFB ≤ 0.4V 100 %
PWM Switch ON-Resistance ISW = 50mA; VFB = 0.7VFB_NOM (High Side Switch)
95
200
300
mΩ
Oscillator Frequency 1.8 2 2.2 MHz
Enable Threshold 0.5 0.85 1.3 V
Enable Hysteresis 50 mV
Enable Input Current 0.1 2 µA
Power Good Range ±7 ±10 %
Power Good Resistance IPGOOD = 500µA 145 200 Ω
Over-Temperature Shutdown 160 °C
Over-Temperature Hysteresis 20 °C
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification for packaged product only.
5. Dropout voltage is defined as the input-to-output differential at which the output voltage drops 2% below its nominal value that is initially measured at
a 1V differential. For outputs below 2.7V, the dropout voltage is the input-to-output voltage differential with a minimum input voltage of 2.7V.
Micrel, Inc. MIC2207
December 2006 4 M9999-122006
Typical Characteris t ics
80
82
84
86
88
90
92
94
96
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
3.3VOUT Efficiency
4.5VIN
5VIN 5.5VIN
80
82
84
86
88
90
92
94
96
98
100
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
2.5VOUT Efficiency
3.3VIN
3VIN
3.6VIN
80
82
84
86
88
90
92
94
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
2.5VOUT Efficiency
5.5VIN
5VIN
4.5VIN
75
77
79
81
83
85
87
89
91
93
95
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1.8VOUT Efficiency
3.6VIN
3VIN
3.3VIN
70
72
74
76
78
80
82
84
86
88
90
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1.8VOUT Efficiency
5.5VIN
5VIN
4.5VIN
70
75
80
85
90
95
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1.5VOUT Efficiency
3.6VIN
3VIN
3.3VIN
65
67
69
71
73
75
77
79
81
83
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1.5VOUT Efficiency
5.5VIN
5VIN
4.5VIN
70
72
74
76
78
80
82
84
86
88
90
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1.2VOUT Efficiency
3.6VIN
3VIN
3.3VIN
65
67
69
71
73
75
77
79
81
83
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1.2VOUT Efficiency
5.5VIN
5VIN
4.5VIN
65
67
69
71
73
75
77
79
81
83
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1VOUT Efficiency
3.6VIN
3VIN
3.3VIN
60
65
70
75
80
85
00.511.522.53
EFFICIENCY (%)
OUTPUT CURRENT (A)
MIC2207
1VOUT Efficiency
5.5VIN
5VIN
4.5VIN
0.990
0.995
1.000
1.005
1.010
00.511.522.53
OUTPUT VOLTAGE (V)
OUTPUT CURRENT (A)
Load Regulation
VIN = 3.3V
Micrel, Inc. MIC2207
December 2006 5 M9999-122006
Typical Characteris tics (cont.)
SUPPLY VOLTAGE (V)
0.990
0.992
0.994
0.996
0.998
1.000
1.002
1.004
1.006
1.008
1.010
-40
-20
0
20
40
60
80
100
120
FEEDBACK VOLTAGE (V)
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
VIN = 3.3V
1.500
1.600
1.700
1.800
1.900
2.000
2.100
2.200
2.300
2.400
2.500
-40
-20
0
20
40
60
80
100
120
FREQUENCY (MHz)
TEMPERATURE (°C)
Frequency
vs. Temperature
VIN = 3.3V
0
0.2
0.4
0.6
0.8
1
1.2
012345
FEEDBACK VOLTAGE (V)
SUPPLY VOLTAGE (V)
Feedback Voltage
vs. Supply Voltage
VEN = VIN
0
100
200
300
400
500
600
700
800
900
012345
QUIESCENT CURRENT (µA)
SUPPLY VOLTAGE (V)
Quiescent Current
vs. Supply Voltage
VEN = VIN
70
75
80
85
90
95
100
105
110
115
120
2.73.23.74.24.75.2
P-CHANNEL RDSON (mOhm)
SUPPLY VOLTAGE (V)
RDSON
vs. Supply Voltage
0
20
40
60
80
100
120
140
160
-40
-20
0
20
40
60
80
100
120
P-CHANNEL RDSON (mOhm)
TEMPERATURE (°C)
RDSON
vs. Temperature
3.3VIN
0
0.2
0.4
0.6
0.8
1.0
1.2
2.7 3.2 3.7 4.2 4.7
ENABLE THRESHOLD (V)
SUPPLY VOLTAGE (V)
Enable Threshold
vs. Supply Voltage
0
0.2
0.4
0.6
0.8
1.0
1.2
-40
-20
0
20
40
60
80
100
120
ENABLE THRESHOLD (V)
TEMPERATURE (°C)
Enable Threshold
vs. Temperature
3.3VIN
Micrel, Inc. MIC2207
December 2006 6 M9999-122006
Functional Characteristics
Continuious Current
TIME (200ns/div.)
SWITCH VOLTAGE
(2V/div.)
INDUCTOR CURRENT
(500mA/div.)
VIN = 3.3V
VOUT = 1V
L = 1µH
COUT = 4.7µF
IOUT = 1A
0A
Discontinuous Current
TIME (200ns/div.)
SWITCH VOLTAGE
(2V/div.)
INDUCTOR CURRENT
(200mA/div.)
VIN = 3.3V
VOUT = 1V
L = 1µH
COUT = 4.7µF
IOUT = 30mA
0A
LoadTransient Response
TIME (400µs/div.)
OUTPUT VOLTAGE
(20mV/div.)
OUTPUT CURRENT
(2A/div.)
VIN = 3.3V
VOUT = 1.8V
0A
Output Ripple
TIME (400ns/div.)
SWITCH VOLTAGE
(2V/div.)
OUTPUT VOLTAGE
(10mV/div.)
AC COUPLED
IOUT = 3.0A
Start-UpWaveforms
TIME (40µs/div.)
ENABLE VOLTAGE
(2V/div.)
INDUCTOR CURRENT
(2A/div.)
INPUT CURRENT
(1A/div.)
FEEDBACK VOLTAGE
(1V/div.)
Micrel, Inc. MIC2207
December 2006 7 M9999-122006
Functional Diagram
VIN
VIN
BIAS
EN
SW
SW
FB
PGOOD
PGND
Enable and
Control Logic
PWM
Control
P-Channel
Current Limit
SGND
1.0V
1.0V
Soft
Start
Bias,
UVLO,
Thermal
Shutdown
HSD
EA
MIC2207 Block Diagram
Micrel, Inc. MIC2207
December 2006 8 M9999-122006
Pin Descriptions
VIN
Two pins for VIN provide power to the source of the
internal P-channel MOSFET along with the current
limiting sensing. The VIN operating voltage range is from
2.7V to 5.5V. Due to the high switching speeds, a 10µF
capacitor is recommended close to VIN and the power
ground (PGND) for each pin for bypassing. Please refer
to layout recommendations.
BIAS
The bias (BIAS) provides power to the internal reference
and control sections of the MIC2207. A 10Ω resistor
from VIN to BIAS and a 0.1µF from BIAS to SGND is
required for clean operation.
EN
The enable pin provides a logic level control of the
output. In the off state, supply current of the device is
greatly reduced (typically <1µA). Do not drive the enable
pin above the supply voltage.
FB
The feedback pin (FB) provides the control path to
control the output. For adjustable versions, a resistor
divider connecting the feedback to the output is used to
adjust the desired output voltage. The output voltage is
calculated as follows:
⎟
⎠
⎞
⎜
⎝
⎛+×= 1
R2
R1
VV REFOUT
where VREF is equal to 1.0V.
A feedforward capacitor is recommended for most
designs using the adjustable output voltage option. To
reduce current draw, a 10K feedback resistor is
recommended from the output to the FB pin (R1). Also, a
feedforward capacitor should be connected between the
output and feedback (across R1). The large resistor
value and the parasitic capacitance of the FB pin can
cause a high frequency pole that can reduce the overall
system phase margin. By placing a feedforward
capacitor, these effects can be significantly reduced.
Feedforward capacitance (CFF) can be calculated as
follows:
200kHzR12
1
CFF ××
=
π
SW
The switch (SW) pin connects directly to the inductor
and provides the switching current necessary to operate
in PWM mode. Due to the high speed switching on this
pin, the switch node should be routed away from
sensitive nodes. This pin also connects to the cathode of
the free-wheeling diode.
PGOOD
Power good is an open drain pull down that indicates
when the output voltage has reached regulation. For a
power good low, the output voltage is within ±10% of the
set regulation voltage. For output voltages greater or
less than 10%, the PGOOD pin is high. This should be
connected to the input supply through a pull up resistor.
A delay can be added by placing a capacitor from
PGOOD to ground.
PGND
Power ground (PGND) is the ground path for the
MOSFET drive current. The current loop for the power
ground should be as small as possible and separate
from the Signal ground (SGND) loop. Refer to the layout
considerations fro more details.
SGND
Signal ground (SGND) is the ground path for the biasing
and control circuitry. The current loop for the signal
ground should be separate from the power ground
(PGND) loop. Refer to the layout considerations for more
details.
Micrel, Inc. MIC2207
December 2006 9 M9999-122006
Application Information
The MIC2207 is a 3A PWM non-synchronous buck
regulator. By switching an input voltage supply, and
filtering the switched voltage through an Inductor and
capacitor, a regulated DC voltage is obtained. Figure 1
shows a simplified example of a non-synchronous buck
converter.
Figure 1.
For a non-synchronous buck converter, there are two
modes of operation; continuous and discontinuous.
Continuous or discontinuous refer to the inductor
current. If current is continuously flowing through the
inductor throughout the switching cycle, it is in
continuous operation. If the inductor current drops to
zero during the off time, it is in discontinuous operation.
Critically continuous is the point where any decrease in
output current will cause it to enter discontinuous
operation. The critically continuous load current can be
calculated as follows;
LMH
z
VIN
V
V
V
OUT
OUT
OUT ××
⎥
⎥
⎦
⎤
⎢
⎢
⎣
⎡−
=22
2
Continuous or discontinuous operation determines how
we calculate peak inductor current.
Continuous Operation
Figure 2 illustrates the switch voltage and inductor
current during continuous operation.
Figure 2. Continuous Operation
The output voltage is regulated by pulse width
modulating (PWM) the switch voltage to the average
required output voltage. The switching can be broken up
into two cycles; On and Off.
During the on-time, the high side switch is turned on,
current flows from the input supply through the inductor
and to the output. The inductor current is
Figure 3. On-Time
charged at the rate;
()
L
VV OUTIN −
To determine the total on-time, or time at which the
inductor charges, the duty cycle needs to be calculated.
The duty cycle can be calculated as;
IN
OUT
V
V
D=
and the On time is;
2MHz
D
TON =
Therefore, peak to peak ripple current is;
()
L2MHz
V
V
VV
IIN
OUT
OUTIN
pkpk ×
×−
=
−
Since the average peak to peak current is equal to the
load current. The actual peak (or highest current the
inductor will see in a steady state condition) is equal to
the output current plus ½ the peak to peak current.
()
L2MHz2
V
V
VV
II IN
OUT
OUTIN
OUTpk ××
×−
+=
Figure 4 demonstrates the off-time. During the off-
time, the high-side internal P-channel MOSFET turns off.
Since the current in the inductor has to discharge, the
current flows through the free-wheeling Schottky diode
to the output. In this case, the inductor discharge rate is
(where VD is the diode forward voltage);
Micrel, Inc. MIC2207
December 2006 10 M9999-122006
()
L
VV DOUT +
−
The total off time can be calculated as;
2MHz
D1
TOFF
−
=
Figure 4. Off-Time
Discontinuous Operation
Discontinuous operation is when the inductor current
discharges to zero during the off cycle. Figure 5.
demonstrates the switch voltage and inductor currents
during discontinuous operation.
Figure 5. Discontinuous Operation
When the inductor current (IL) has completely
discharged, the voltage on the switch node rings at the
frequency determined by the parasitic capacitance and
the inductor value. In figure 5, it is drawn as a DC
voltage, but to see actual operation (with ringing) refer to
the functional characteristics.
Discontinuous mode of operation has the advantage
over full PWM in that at light loads, the MIC2207 will skip
pulses as necessary, reducing gate drive losses,
drastically improving light load efficiency.
Efficiency Considerations
Calculating the efficiency is as simple as measuring
power out and dividing it by the power in;
100
P
P
Efficiency
IN
OUT ×=
Where input power (PIN) is;
INININ IVP ×=
and output power (POUT) is calculated as;
OUTOUTOUT IVP ×=
The Efficiency of the MIC2207 is determined by several
factors.
• Rdson (Internal P-channel Resistance)
• Diode conduction losses
• Inductor Conduction losses
• Switching losses
Rdson losses are caused by the current flowing through
the high side P-channel MOSFET. The amount of power
loss can be approximated by;
DIRP 2
OUTDSONSW ××=
Where D is the duty cycle.
Since the MIC2207 uses an internal P-channel
MOSFET, Rdson losses are inversely proportional to
supply voltage. Higher supply voltage yields a higher
gate to source voltage, reducing the Rdson, reducing the
MOSFET conduction losses. A graph showing typical
Rdson vs input supply voltage can be found in the typical
characteristics section of this datasheet.
Diode conduction losses occur due to the forward
voltage drop (VF) and the output current. Diode power
losses can be approximated as follows;
()
D1IVP OUTFD −××=
For this reason, the schottky diode is the rectifier of
choice. Using the lowest forward voltage drop will help
reduce diode conduction losses, and improve efficiency.
Duty cycle, or the ratio of output voltage to input voltage,
determines whether the dominant factor in conduction
losses will be the internal MOSFET or the schottky
diode. Higher duty cycles place the power losses on the
high side switch, and lower duty cycles place the power
losses on the schottky diode.
Inductor conduction losses (PL) can be calculated by
multiplying the DC resistance (DCR) times the square of
the output current;
2
OUTLIDCRP×=
Micrel, Inc. MIC2207
December 2006 11 M9999-122006
Also, be aware that there are additional core losses
associated with switching current in an inductor. Since
most inductor manufacturers do not give data on the
type of material used, approximating core losses
becomes very difficult, so verify inductor temperature
rise.
Switching losses occur twice each cycle, when the
switch turns on and when the switch turns off. This is
caused by a non-ideal world where switching transitions
are not instantaneous, and neither are currents. Figure 6
demonstrates (Or exaggerates…) how switching losses
due to the transitions dissipate power in the switch.
Figure 6. Switching Transition Losses
Normally, when the switch is on, the voltage across the
switch is low (virtually zero) and the current through the
switch is high. This equates to low power dissipation.
When the switch is off, voltage across the switch is high
and the current is zero, again with power dissipation
being low. During the transitions, the voltage across the
switch (VS-D) and the current through the switch (IS-D) are
at midpoint of their excursions and cause the transition
to be the highest instantaneous power point. During
continuous mode, these losses are the highest. Also,
with higher load currents, these losses are higher. For
discontinuous operation, the transition losses only occur
during the “off” transition since the “on” transitions there
is no current flow through the inductor.
Component Selection
Input Capacitor
A 10µF ceramic is recommended on each VIN pin for
bypassing. X5R or X7R dielectrics are recommended for
the input capacitor. Y5V dielectrics lose most of their
capacitance over temperature and are therefore not
recommended. Also, tantalum and electrolytic capacitors
alone are not recommended because of their reduced
RMS current handling, reliability, and ESR increases.
An additional 0.1µF is recommended close to the VIN
and PGND pins for high frequency filtering. Smaller case
size capacitors are recommended due to their lower
ESR and ESL. Please refer to layout recommendations
for proper layout of the input capacitor.
Output Capacitor
The MIC2207 is designed for a 4.7µF output capacitor.
X5R or X7R dielectrics are recommended for the output
capacitor. Y5V dielectrics lose most of their capacitance
over temperature and are therefore not recommended.
In addition to a 4.7µF, a small 0.1µF is recommended
close to the load for high frequency filtering. Smaller
case size capacitors are recommended due to their
lower equivalent series ESR and ESL.
The MIC2207 utilizes type III voltage mode internal
compensation and utilizes an internal zero to
compensate for the double pole roll off of the LC filter.
For this reason, larger output capacitors can create
instabilities. In cases where a 4.7µF output capacitor is
not sufficient, the MIC2208 offers the ability to externally
control the compensation, allowing for a wide range of
output capacitor types and values.
Inductor Selection
The MIC2207 is designed for use with a 1µH inductor.
Proper selection should ensure the inductor can handle
the maximum average and peak currents required by the
load. Maximum current ratings of the inductor are
generally given in two methods; permissible DC current
and saturation current. Permissible DC current can be
rated either for a 40°C temperature rise or a 10% to 20%
loss in inductance. Ensure the inductor selected can
handle the maximum operating current. When saturation
current is specified, make sure that there is enough
margin that the peak current will not saturate the
inductor.
Diode Selection
Since the MIC2207 is non-synchronous, a free-wheeling
diode is required for proper operation. A schottky diode
is recommended due to the low forward voltage drop
and their fast reverse recovery time. The diode should
be rated to be able to handle the average output current.
Also, the reverse voltage rating of the diode should
exceed the maximum input voltage. The lower the
forward voltage drop of the diode the better the
efficiency. Please refer to the layout recommendations to
minimize switching noise.
Feedback Resistors
The feedback resistor set the output voltage by dividing
down the output and sending it to the feedback pin. The
feedback voltage is 1.0V. Calculating the set output
voltage is as follows;
⎟
⎠
⎞
⎜
⎝
⎛+= 1
R2
R1
VV FBOUT
Where R1 is the resistor from VOUT to FB and R2 is the
resistor from FB to GND. The recommended feedback
resistor values for common output voltages are available
Micrel, Inc. MIC2207
December 2006 12 M9999-122006
in the bill of materials on page 19. Although the range of
resistance for the FB resistors is very wide, R1 is
recommended to be 10K. This minimizes the effect the
parasitic capacitance of the FB node.
Feedforward Capacitor (CFF)
A capacitor across the resistor from the output to the
feedback pin (R1) is recommended for most designs.
This capacitor can give a boost to phase margin and
increase the bandwidth for transient response. Also,
large values of feedforward capacitance can slow down
the turn-on characteristics, reducing inrush current. For
maximum phase boost, CFF can be calculated as follows;
R1200kHz2
1
CFF ××
=
π
Bias filter
A small 10Ω resistor is recommended from the input
supply to the bias pin along with a small 0.1µF ceramic
capacitor from bias to ground. This will bypass the high
frequency noise generated by the violent switching of
high currents from reaching the internal reference and
control circuitry. Tantalum and electrolytic capacitors are
not recommended for the bias, these types of capacitors
lose their ability to filter at high frequencies.
Loop Stability and Bode Analysis
Bode analysis is an excellent way to measure small
signal stability and loop response in power supply
designs. Bode analysis monitors gain and phase of a
control loop. This is done by breaking the feedback loop
and injecting a signal into the feedback node and
comparing the injected signal to the output signal of the
control loop. This will require a network analyzer to
sweep the frequency and compare the injected signal to
the output signal. The most common method of injection
is the use of a transformer. Figure 7 demonstrates how a
transformer is used to inject a signal into the feedback
network.
Figure 7. Transformer Injection
A 50Ω resistor allows impedance matching from the
network analyzer source. This method allows the DC
loop to maintain regulation and allow the network
analyzer to insert an AC signal on top of the DC voltage.
The network analyzer will then sweep the source while
monitoring A and R for an A/R measurement. While this
is the most common method for measuring the gain and
phase of a power supply, it does have significant
limitations. First, to measure low frequency gain and
phase, the transformer needs to be high in inductance.
This makes frequencies <100Hz require an extremely
large and expensive transformer. Conversely, it must be
able to inject high frequencies. Transformers with these
wide frequency ranges generally need to be custom
made and are extremely expensive (usually to the tune
of several hundred dollars!). By using an op-amp, cost
and frequency limitations caused by an injection
transformer are completely eliminated. Figure 8
demonstrates using an op-amp in a summing amplifier
configuration for signal injection.
Networ k Analyzer
Source
+8V R1
1k
R3
1k R4
1k
50
Feedback Output
Network
Analyzer
“A” Input
Network
Analyzer
“R” Input MIC922BC5
Figure 8. Op Amp Injection
R1 and R2 reduce the DC voltage from the output to the
non-inverting input by half. The network analyzer is
generally a 50Ω source. R1 and R2 also divide the AC
signal sourced by the network analyzer by half. These
two signals are “summed” together at half of their
original input. The output is then amplified by 2 by R3
and R4 (the 50Ω is to balance the network analyzer’s
source impedance) and sent to the feedback signal. This
essentially breaks the loop and injects the AC signal on
top of the DC output voltage and sends it to the
feedback. By monitoring the feedback “R” and output
“A”, gain and phase are measured. This method has no
minimum frequency. Ensure that the bandwidth of the
op-amp being used is much greater than the expected
bandwidth of the power supply’s control loop. An op-amp
with >100MHz bandwidth is more than sufficient for most
power supplies (which includes both linear and
switching) and are more common and significantly
cheaper than the injection transformers previously
mentioned. The one disadvantage to using the op-amp
injection method, is the supply voltages need to be
below the maximum operating voltage of the op-amp.
Also, the maximum output voltage for driving 50Ω inputs
using the MIC922 is 3V. For measuring higher output
voltages, a 1MΩ input impedance is required for the A
and R channels. Remember to always measure the
Micrel, Inc. MIC2207
December 2006 13 M9999-122006
output voltage with an oscilloscope to ensure the
measurement is working properly. You should see a
single sweeping sinusoidal waveform without distortion
on the output. If there is distortion of the sinusoid, reduce
the amplitude of the source signal. You could be
overdriving the feedback causing a large signal
response.
The following Bode analysis show the small signal loop
stability of the MIC2207. The MIC2207 utilizes a type III
compensation. This is a dominant low frequency pole,
followed by 2 zero’s and finally the double pole of the
inductor capacitor filter, creating a final 20dB/decade roll
off. Bode analysis gives us a few important data points;
speed of response (Gain Bandwidth or GBW) and loop
stability. Loop speed or GBW determines the response
time to a load transient. Faster response times yield
smaller voltage deviations to load steps.
Instability in a control loop occurs when there is gain and
positive feedback. Phase margin is the measure of how
stable the given system is. It is measured by determining
how far the phase is from crossing zero when the gain is
equal to 1 (0dB).
-30
-20
-10
0
10
20
30
40
50
60
GAIN (dB)
FREQUENCY (Hz)
Bode Plot
VIN=3.3V, VOUT=1.8V, IOUT=3A
-105
-70
-35
0
35
70
105
140
175
210
PHASE (°)
100 1k 10k 100k 1M
L=1µH
COUT = 4.7µF
R1 = 10k
R2 = 12.4k
CFF = 82pF
GAIN
PHASE
Typically for 3.3Vin and 1.8Vout at 3A;
• Phase Margin=47 Degrees
• GBW=156KHz
Gain will also increase with input voltage. The following
graph shows the increase in GBW for an increase in
supply voltage.
-30
-20
-10
0
10
20
30
40
50
60
GAIN (dB)
FREQUENCY (Hz)
Bode Plot
VIN=5V, VOUT=1.8V, IOUT=3A
-105
-70
-35
0
35
70
105
140
175
210
PHASE (°)
100 1k 10k 100k 1M
L=1µH
COUT = 4.7µF
R1 = 10k
R2 = 12.4k
CFF = 82pF
GAIN
PHASE
5Vin, 1.8Vout at 3A load;
• Phase Margin=43.1 Degrees
• GBW= 218KHz
Being that the MIC2207 is non-synchronous; the
regulator only has the ability to source current. This
means that the regulator has to rely on the load to be
able to sink current. This causes a non-linear response
at light loads. The following plot shows the effects of the
pole created by the nonlinearity of the output drive
during light load (discontinuous) conditions.
-30
-20
-10
0
10
20
30
40
50
60
GAIN (dB)
FREQUENCY (Hz)
Bode Plot
VIN=3.3V,VOUT=1.8V,IOUT=50mA
-105
-70
-35
0
35
70
105
140
175
210
PHASE (°)
100 1k 10k 100k 1M
L=1µH
COUT = 4.7µF
R1 = 10k
R2 = 12.4k
CFF = 82pF
GAIN
PHASE
3.3Vin, 1.8Vout Iout=50mA;
• Phase Margin=90.5 Degrees
• GBW= 64.4KHz
Feed Forward Capacitor
The feedback resistors are a gain reduction block in the
overall system response of the regulator. By placing a
capacitor from the output to the feedback pin, high
frequency signal can bypass the resistor divider, causing
a gain increase up to unity gain.
-10
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
GAIN (dB)
FREQUENCY (Hz)
Gain and Phase
vs. Frequen c y
0
5
10
15
20
25
PHASE BOOST (°)
100 1k 10k 100k 1M
L=1µH
COUT = 4.7µF
R1 = 10k
R2 = 12.4k
CFF = 82pF
GAIN
PHASE
The graph above shows the effects on the gain and
phase of the system caused by feedback resistors and a
feedforward capacitor. The maximum amount of phase
boost achievable with a feedforward capacitor is
graphed below.
Micrel, Inc. MIC2207
December 2006 14 M9999-122006
0
5
10
15
20
25
30
35
40
45
50
12345
PAHSE BOOST (°)
OUTPUT VOLTAGE (V)
Max. Amount of Phase Boost
Obta ina bl e us ing CFF vs. Output
Voltage
VREF = 1V
By looking at the graph, phase margin can be affected to
a greater degree with higher output voltages.
The next bode plot shows the phase margin of a 1.8V
output at 3A without a feedforward capacitor.
-30
-20
-10
0
10
20
30
40
50
60
GAIN (dB)
FREQUENCY (Hz)
Bode Plot
VIN=3.3V, VOUT=1.8V, IOUT=3A
-105
-70
-35
0
35
70
105
140
175
210
PHASE (°)
100 1k 10k 100k 1M
L=1µH
COUT = 4.7µF
R1 = 10k
R2 = 12.4k
CFF = 0pF
GAIN
PHASE
As you can see the typical phase margin, using the
same resistor values as before without a feedforward
capacitor results in 33.6 degrees of phase margin. Our
prior measurement with a feedforward capacitor yielded
a phase margin of 47 degrees. The feedforward
capacitor has given us a phase boost of 13.4 degrees
(47 degrees – 33.6 Degrees = 13.4 Degrees).
Output Impedance and Transient
response
Output impedance, simply stated, is the amount of
output voltage deviation vs. the load current deviation.
The lower the output impedance, the better.
OUT
OUT
OUT I
V
Z∆
∆
=
Output impedance for a buck regulator is the parallel
impedance of the output capacitor and the MOSFET and
inductor divided by the gain;
COUT
LDSON
TOTAL X
GAIN
XDCRR
Z++
=
To measure output impedance vs. frequency, the load
current must be swept across the frequencies measured,
while the output voltage is monitored. Fig 9 shows a test
set-up to measure output impedance from 10Hz to 1MHz
using the MIC5190 high speed controller.
By setting up a network analyzer to sweep the feedback
current, while monitoring the output of the voltage
regulator and the voltage across the load resistance,
output impedance is easily obtainable. To keep the
current from being too high, a DC offset needs to be
applied to the network analyzer’s source signal. This can
be done with an external supply and 50Ω resistor. Make
sure that the currents are verified with an oscilloscope
first, to ensure the integrity of the signal measurement. It
is always a good idea to monitor the A and R
measurements with a scope while you are sweeping it.
To convert the network analyzer data from dBm to
something more useful (such as peak to peak voltage
and current in our case);
707.0
250Ω1mW10
V
10
dBm
×××
=∆
and peak to peak current;
LOAD
10
dBm
R707.0
250Ω1mW10
I×
×××
=∆
The following graph shows output impedance vs
frequency at 2A load current sweeping the AC current
from 10Hz to 10MHz, at 1A peak to peak amplitude.
0.001
0.01
0.1
1
OUTPUT IMPEDANCE (Ohms)
FREQUENCY (Hz)
Output Impedance
vs. Frequency
100 1k 10k 100k 1M
VOUT=1.8V
L=1µH
COUT=4.7µF + 0.1µ
5VIN
3.3VIN
10
From this graph, you can see the effects of bandwidth
and output capacitance. For frequencies <200KHz, the
output impedance is dominated by the gain and
inductance. For frequencies >200KHz, the output
impedance is dominated by the capacitance. A good
approximation for transient response can be calculated
from determining the frequency of the load step in amps
per second;
π
2
A/sec
=
f
Micrel, Inc. MIC2207
December 2006 15 M9999-122006
Figure 9. Output Impedance Measurement
Then, determine the output impedance by looking at the
output impedance vs frequency graph. Next, calculate
the voltage deviation times the load step;
OUTOUTOUT ZIV ×∆=∆
The output impedance graph shows the relationship
between supply voltage and output impedance. This is
caused by the lower Rdson of the high side MOSFET
and the increase in gain with increased supply voltages.
This explains why higher supply voltages have better
transient response.
COUT
LDSON
TOTAL X
GAIN
XDCRR
Z↑
++↓
=↓
Ripple measurements
To properly measure ripple on either input or output of a
switching regulator, a proper ring in tip measurement is
required. Standard oscilloscope probes
come with a grounding clip, or a long wire
with an alligator clip. Unfortunately, for
high frequency measurements, this
ground clip can pick-up high frequency
noise and erroneously inject it into the
measured output ripple.
The standard evaluation board
accommodates a home made version by
providing probe points for both the input
and output supplies and their respective
grounds. This requires the removing of
the oscilloscope probe sheath and ground
clip from a standard oscilloscope probe
and wrapping a non-shielded bus wire
around the oscilloscope probe. If there
does not happen to be any non shielded
bus wire immediately available, the leads
from axial resistors will work. By maintaining the
shortest possible ground lengths on the oscilloscope
probe, true ripple measurements can be obtained.
Micrel, Inc. MIC2207
December 2006 16 M9999-122006
Recommended Layout / 3A Evaluation Board
Recommended Top Lay out
Recommended Bottom Layout
Micrel, Inc. MIC2207
December 2006 17 M9999-122006
MIC2207 Scheme and B.O.M for 3A Output
MIC2207 Schematic
Item Part Number Description Manufacturer Qty
C1a,C1b C2012JB0J106K
GRM219R60J106KE19
08056D106MAT
10µF Ceramic Capacitor X5R 0805 6.3V
10µF Ceramic Capacitor X5R 0805 6.3V
10µF Ceramic Capacitor X5R 0805 6.3V
TDK
Murata
AVX
2
C2 0402ZD104MAT 0.1µF Ceramic Capacitor X5R 0402 10V AVX 1
C3 C2012JB0J475K
GRM188R60J475KE19
06036D475MAT
4.7µF Ceramic Capacitor X5R 0603 6.3V
4.7µF Ceramic Capacitor X5R 0603 6.3V
4.7µF Ceramic Capacitor X5R 0603 6.3V
TDK
Murata
AVX 1
C4 VJ0402A820KXAA 82pF Ceramic Capacitor 0402 Vishay VT 1
D1 SSA33L 3A Schottky 30V SMA Vishay Semi 1
RLF7030-1R0N6R4 1µH Inductor 8.8mΩ 7.1mm(L) x 6.8mm (W)x 3.2mm(H) TDK 1
744 778 9001 1µH Inductor 12mΩ 7.3mm(L)x7.3mm(W)x3.2mm(H) Wurth Electronik 1
L1
IHLP2525AH-01 1 1µH Inductor 17.5mΩ(L)6.47mmx(W)6.86mmx(H) 1.8mm Vishay Dale 1
R1,R4 CRCW04021002F 10KΩ1% 0402 resistor Vishay Dale 1
R2 CRCW04026651F
CRCW04021242F
CRCW04022002F
CRCW04024022F
6.65kΩ 1% 0402 For 2.5VOUT
12.4kΩ 1% 0402 For 1.8 VOUT
20kΩ 1% 0402 For 1.5 VOUT
40.2kΩ 1% 0402 For 1.2 VOUT
Open For 1.0 VOUT
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale 1
R3 CRCW040210R0F 10Ω1% 0402 resistor Vishay Dale 1
U1 MIC2207YML 2MHz 3A Buck Regulator Micrel 1
Notes:
1. Sumida Tel: 408-982-9660
2. Murata Tel: 949-916-4000
3. Vishay Tel: 402-644-4218
4. Micrel, Inc.: 408-944-0800
Micrel, Inc. MIC2207
December 2006 18 M9999-122006
Package Information
12-Pin MLF® (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 US
A
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implan
t
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.
