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FEATURES
DOperating Input Voltage 10 V to 40 V
DOutput Voltage Tracks External Reference
DProgrammable Fixed-Frequency Up to
100 kHz to 1 MHz Voltage Mode Controller
DInternal Gate Drive Outputs for High-Side
and Synchronous N-Channel MOSFETs
DExternally Synchronizable
DProgrammable Short-Circuit Protection
DThermal Shutdown
D16-Pin PowerPADt Package (θJC = 2°C/W)
DProgrammable Closed-Loop Soft-Start
APPLICATIONS
DDDR Tracking Regulators
DPower Modules
DNetworking Equipment
DIndustrial Servers
CONTENTS
Device Ratings 2
Electrical Characteristics 3
Terminal Information 5
Application Information 7
Design Example 22
Additional References 29
DESCRIPTION
The TPS40056 is part of a family of high-voltage,
wide input, synchronous, step-down converters.
The TPS40056 offers design flexibility with a
variety of user programmable functions, including
soft-start, operating frequency, high-side current
limit, and loop compensation. The TPS40056 is
also synchronizable to an external supply. It
incorporates MOSFET gate drivers for external
N-channel high-side and synchronous rectifier
(SR) MOSFETs. Gate drive logic incorporates
anti-cross conduction circuitry to prevent
simultaneous high-side and synchronous rectifier
conduction. The externally programmable short
circuit protection provides pulse-by-pulse current
limit, as well as hiccup mode operation utilizing an
internal fault counter for longer duration
overloads.
SIMPLIFIED APPLICATION DIAGRAM
VTT
VIN
VTRKIN
UDG−03080
5
13
12
16
15
1
2
3
SYNC
RT
BP5
SGND
VIN
HDRV
SW
BP10
4 EA_REF
11
ILIM
TPS40056PWP
6SS
7 VFB
8 COMP
14BOOST
LDRV 10
PGND 9
+
−
PAD
+
−
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Copyright 2006, Texas Instruments Incorporated
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
TAPACKAGE PART NUMBER
−40°C to 85°CPlastic HTSSOP(PWP)(1) TPS40056PWP
(1) The PWP package is also available taped and reeled. Add an R suffix to the device type
(i.e., TPS40056PWPR). See the application section of the data sheet for PowerPAD
drawing and layout information.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted(2)
TPS40056 UNIT
VIN 45
Input voltage range, VIN
VFB, SS, SYNC, EA_REF −0.3 to 6
Input voltage range, VIN SW −0.3 to 45 V
SW, transient < 50 ns −2.5
V
Output voltage range, VOCOMP, RT, SS −0.3 to 6
Output current, IOUT RT 200 µA
Operating junction temperature range, TJ−40 to 125
Storage temperature, Tstg −55 to 150 °C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260
C
(2) Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only,
and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is
not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN NOM MAX UNIT
Input voltage, VI10 40 V
Operating free-air temperature, TA−40 85 °C
THERMAL
PAD
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
SYNC
RT
BP5
EA_REF
SGND
SS/SD
VFB
COMP
ILIM
VIN
BOOST
HDRV
SW
BP10
LDRV
PGND
PWP PACKAGE(3)(4)
(TOP VIEW)
(3) For more information on the PWP package, refer to TI Technical Brief, Literature No. SLMA002.
(4) PowerPADt heat slug must be connected to SGND (pin 5) or electrically isolated from all other pins.
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ELECTRICAL CHARACTERISTICS
TA = −40°C to 85°C, VIN = 12 Vdc, RT = 90.9 kΩ, fSW = 500 kHz, VEA_REF = 1.25 V, all parameters at zero power dissipation (unless
otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
INPUT SUPPLY
VIN Input voltage range, VIN 10 40 V
OPERATING CURRENT
IDD Quiescent current Output drivers not switching,
VFB = 1.3 V 1.5 3.0 mA
BP5
VBP5 Ouput voltage ILOAD = 1 mA 4.5 5.0 5.5 V
OSCILLATOR/RAMP GENERATOR
fOSC Accuracy 9 V ≤ VIN ≤ 40 V 520 580 640 kHz
VRAMP PWM ramp voltage(1) VPEAK−VVAL 2.0
V
VIH High-level input voltage, SYNC 2 5 V
VIL Low-level input voltage, SYNC 0.8 V
ISYNC Input current, SYNC 5 10 µA
Pulse width, SYNC 50 ns
VRT RT voltage 2.38 2.50 2.58 V
Maximum duty cycle
VFB = 0 V, fSW ≤ 600 kHz 90%
Maximum duty cycle VFB = 0 V, 600 kHz ≤ fSW ≤ 1 MHz 85%
Minumum duty cycle VFB ≥ EA_REF + 0.05 V 0%
SOFT START
ISS Soft-start source current 1.65 2.35 3.05 µA
VSS Soft-start clamp voltage 3.7 V
tDSCH Discharge time CSS = 220 pF 1.6 2.2 2.8
s
tSS Soft-start time CSS = 220 pF, 0 V ≤ VSS ≤ 1.6 V 100 155 205 µs
BP10
VBP10 Ouput voltage 9.0 9.6 10.3 V
ERROR AMPLIFIER
VEA_REF Error amplifier reference input voltage(1)(2) 10 V ≤ VIN ≤ 40 V 0.2 2.5 V
Input offset voltage 0.5 V ≤ VFB ≤ 2.25 V −6 6 mV
Input offset voltage 0.2 V ≤ VFB ≤ 0.5 V −10 0 10 MV
GBW Gain bandwidth 0.2 V ≤ VFB ≤ 0.5 V 1.5 3.5 MHz
GBW Gain bandwidth 0.5 V ≤ VFB ≤ 2.25 V 2.5 5.0 MHz
AVOL Open loop gain 60 80 dB
IOH High-level output source current 1.5 4.0
mA
IOL Low-level output sink current 2.0 4.0 mA
VOH High-level output voltage ISOURCE = 500 µA 3.2 3.5
V
VOL Low-level output voltage ISINK = 500 µA 0.20 0.35 V
IBIAS Input bias current VFB = 1.2 V 100 200 nA
(1) Ensured by design. Not production tested.
(2) Common mode range extends to ground, but not tested below 200 mV.
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ELECTRICAL CHARACTERISTICS
TA = −40°C to 85°C, VIN = 12 Vdc, RT = 90.9 kΩ, fSW = 500 kHz, VEA_REF = 1.25 V all parameters at zero power dissipation (unless
otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
CURRENT LIMIT
ISINK Current limit sink current 8 10 12 µA
Propagation delay to output
VILIM = 11.7 V, VSW = (VILIM − 0.5 V) 300
Propagation delay to output VILIM = 11.7 V, VSW = (VILIM − 2 V) 250 ns
tON Switch leading-edge blanking pulse time(1) 100
ns
tOFF Off time during a fault 7 cycles
VILIM = 11.6 V, TA = 25°C −100 −70 −40
V
OS
Offset voltage SW vs. ILIM VILIM = 11.6 V, 0°C ≤ TA ≤ 85°C−125 −30 mV
VOS
Offset voltage SW vs. ILIM
VILIM = 11.6 V, −40°C ≤ TA ≤ 85°C −125 −15
mV
OUTPUT DRIVER
tLRISE Low-side driver rise time
CLOAD = 2200 pF
48 96
tLFALL Low-side driver fall time CLOAD = 2200 pF 24 48
ns
tHRISE High-side driver rise time
CLOAD = 2200 pF, (HDRV − SW)
48 96 ns
tHFALL High-side driver fall time CLOAD = 2200 pF, (HDRV − SW) 36 72
VOH High-level ouput voltage, HDRV IHDRV = −0.1 A (HDRV − SW) BOOST
−1.5 V BOOST
−1.0 V
VOL Low-level ouput voltage, HDRV IHDRV = 0.1 A (HDRV − SW) 0.75
V
VOH High-level ouput voltage, LDRV ILDRV = −0.1 A BP10
−1.4 V BP10
− 1.0 V
V
VOL Low-level ouput voltage, LDRV ILDRV = 0.1 A 0.5
Minimum controllable pulse width(1) 100 150 ns
SS/SD SHUTDOWN
VSD Shutdown threshold voltage Outputs off 90 125 165
mV
VEN Device active threshold voltage 165 210 260 mV
BOOST REGULATOR
VBOOST Output voltage VIN = 12.0 V 19 20 21 V
SW NODE
ILEAK Leakage current(1) 25 µA
THERMAL SHUTDOWN
TSD
Shutdown temperature(1) 165
°C
TSD Hysteresis(1) 20 °C
UVLO
Input voltage UVLO threshold 8.20 8.75 9.25
V
Input voltage UVLO hysteresis 1.0 V
(1) Ensured by design. Not production tested.
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TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME NO.
I/O
DESCRIPTION
BOOST 14 O Gate drive voltage for the high side N-channel MOSFET. The BOOST voltage is 9 V greater than the input
voltage. A 0.1-µF ceramic capacitor should be connected from this pin to the SW pin.
BP5 3 O5-V reference. This pin should be bypassed to ground with a 0.1-µF ceramic capacitor. This pin may be used
with an external dc load of 1 mA or less.
BP10 11 O10-V reference used for gate drive of the N-channel synchronous rectifier. This pin should be bypassed by a 1-µF
ceramic capacitor. This pin may be used with an external dc load of 1 mA or less.
COMP 8 O Output of the error amplifier, input to the PWM comparator. A feedback network is connected from this pin to the
VFB pin to compensate the overall loop. The comp pin is internally clamped above the peak of the ramp to
improve large signal transient response.
HDRV 13 O Floating gate drive for the high-side N-channel MOSFET. This pin switches from BOOST (MOSFET on) to SW
(MOSFET off).
ILIM 16 I Current limit pin, used to set the overcurrent threshold. An internal current sink from this pin to ground sets a
voltage drop across an external resistor connected from this pin to VCC. The voltage on this pin is compared
to the voltage drop (VIN −SW) across the high side MOSFET during conduction.
EA_REF 4 I Non-inverting input to the error amplifier and used as the reference for the feedback loop.
LDRV 10 O Gate drive for the N-channel synchronous rectifier. This pin switches from BP10 (MOSFET on) to ground
(MOSFET off).
PGND 9 − Power ground reference for the device. There should be a low-impedance path from this pin to the source(s)
of the lower MOSFET(s).
RT 2 I A resistor is connected from this pin to ground to set the internal oscillator and switching frequency.
SGND 5 − Signal ground reference for the device.
SS/SD 6 I
Soft-start programming pin. A capacitor connected from this pin to ground programs the soft-start time. The
capacitor is charged with an internal current source of 2.3 µA. The resulting voltage ramp on the SS pin is used
as a second non-inverting input to the error amplifier. Output voltage regulation is controlled by the SS voltage
ramp until the voltage on the SS pin reaches the internal reference voltage , EA_REF V. Pulling this pin low
disables the controller.
SW 12 I This pin is connected to the switched node of the converter and used for overcurrent sensing.
SYNC 1 I Syncronization input for the device. This pin can be used to synchronize the oscillator to an external master
frequency. If synchronization is not used, connect this pin to SGND.
VFB 7 I Inverting input to the error amplifier. In normal operation the voltage on this pin is equal to the EA_REF reference
voltage.
VIN 15 I Supply voltage for the device.
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FUNCTIONAL BLOCK DIAGRAM
UDG−03081
2
7+
+
6
Clock
Oscillator
13
10
14
N−channel
Driver
12
9
1115
8
1
5
BP10
BP10
7
16
3−Bit Up/Down
Fault Counter
7
7
7
0.7 VREF
1.5 VREF
3.5 VREF
Reference
Voltages
7
Fault
RAMP
7
CLK
7
BP5
4
7
Restart
+
0.7 VREF
7
7
7
Fault
CL SQ
QR
7
CLK
CL
SW
7
HDRV
LDRV
PGND
BOOST
BP10
VIN
SYNC
RT
EA_REF
VFB
SS/SD
COMP
ILIM
SGND
N−channel
Driver
10 V Regulator
7
1.5 VREF
+
3 7
BP5
BP5
7
Restart
+0.7 V
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APPLICATION INFORMATION
The TPS40056 allows the user to optimize the PWM controller to the specific application.
The TPS40056 is the controller of choice for synchronous buck designs, the output of which is required to track
another voltage. It has two quadrant operation and can source or sink output current, providing the best transient
response.
SW NODE RESISTOR AND DIODE
The SW node of the converter will be negative during the dead time when both the upper and lower MOSFETs
are off. The magnitude of this negative voltage is dependent on the lower MOSFET body diode and the output
current which flows during this dead time. This negative voltage could affect the operation of the controller,
especially at low input voltages.
Therefore, a resistor ( 3.3 Ω to 4.7 Ω) and Schottky diode must be placed between the lower MOSFET drain
and pin 12, SW, of the controller as shown in Figure 10. The Schottky diode must have a voltage rating to
accommodate the input voltage and ringing on the SW node of the converter. A 30-V Schottky such as a BAT54
or a 40-V Schottky such as a Zetex ZHCS400 or Vishay SD103AWS are adequate. These components are
shown in Figure 10 as RSW and D2.
SETTING THE SWITCHING FREQUENCY (PROGRAMMING THE CLOCK OSCILLATOR)
The TPS40056 has independent clock oscillator and ramp generator circuits. The clock oscillator serves as the
master clock to the ramp generator circuit. The switching frequency, fSW in kHz, of the clock oscillator is set b y
a single resistor (RT) to ground. The clock frequency is related to RT, in kΩ by Equation (1).
RT+ǒ1
fSW 17.82 10*6*23ǓkW
UVLO OPERATION
The TPS40056 uses fixed UVLO protection. The fixed UVLO monitors the input voltage. The UVLO circuit holds
the soft-start low until the input voltage has exceeded the undervoltage threshold.
TRACKING CONFIGURATION (VOUT TRACKING VIN)
Setting the output, VOUT to track another voltage, VTRKIN, is simply a matter of selecting the proper voltage
divider(s) R4,R5,R1 and R6 as shown in Figure 1. The voltage on the EA_REF input should be in the range of
0.2 V to 2.5 V. If the output voltage is less than 2.5 V, resistor R6 can be omitted. For example in the DDR case,
if the voltage VTRKIN ramps up to 2.5 V and it is desired to have VOUT to track it and come up to 1.25 V, set R4=R5
and omit R6. In general, the output voltage, VOUT, in terms of VTRKIN and the two voltage dividers is shown
in Equation (2).
VOUT +VTRKIN ǒR5
R4 )R5Ǔ ǒR6 )R1
R6 ǓV
(1)
(2)
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4
5
6
7
13
12
11
10
HDRV
SW
BP10
LDRV
EA_REF
SGND
SS
VFB
TPS40056PWP
8 9PGNDCOMP
R3 R1
R5
R4
VOUT
VTRKIN
R6
+
−
UDG−06020
Figure 1. Tracking Configuration, VOUT Tracks VTRKIN
Figure 2
fSW − Switching Frequency − kHz
SWITCHING FREQUENCY
vs
TIMING RESISTANCE
0
100
0200 400 600 800 100
0
200
300
400
500
600
RT − Timing Resistance − kΩ
2
3
06481210 1
4
1
2
7
4
5
6
8
9
10
BP10 AND BP5
vs
INPUT VOLTAGE
VOUT − Output Voltage − V
Figure 3
VIN − Input Voltage − V
BP10
BP5
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APPLICATION INFORMATION
BP5 AND BP10 INTERNAL VOLTAGE REGULATORS
Start-up characteristics of the BP5 and BP10 regulators are shown in Figure 2. Slight variations in the BP5
occurs dependent upon the switching frequency. Variation in the BP10 regulation characteristics is also based
on the load presented by switching the external MOSFETs.
SELECTING THE INDUCTOR VALUE
The inductor value determines the magnitude of ripple current in the output capacitors as well as the load current
at which the converter enters discontinuous mode. Too large an inductance results in lower ripple current but
is physically larger for the same load current. Too small an inductance results in larger ripple currents and a
greater number of (or more expensive output capacitors for) the same output ripple voltage requirement. A good
compromise is to select the inductance value such that the converter doesn’t enter discontinuous mode until
the load approximated somewhere between 10% and 30% of the rated output. The inductance value is
described in equation (3).
L+ǒVIN *VOǓ VO
VIN DI fSW (Henries)
where:.
DVO is the output voltage
D∆I is the peak-to-peak inductor current
CALCULATING THE OUTPUT CAPACITANCE
The output capacitance depends on the output ripple voltage requirement, output ripple current, as well as any
output voltage deviation requirement during a load transient.
The output ripple voltage is a function of both the output capacitance and capacitor ESR. The worst case output
ripple is described in equation (4).
DV+DIƪESR )ǒ1
8 CO fSWǓƫVP*P
The output ripple voltage is typically between 90% and 95% due to the ESR component.
The output capacitance requirement typically increases in the presence of a load transient requirement. During
a step load, the output capacitance must provide energy to the load (light to heavy load step) or absorb excess
inductor energy (heavy to light load step) while maintaining the output voltage within acceptable limits. The
amount of capacitance depends on the magnitude of the load step, the speed of the loop and the size of the
inductor.
(3)
(4)
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APPLICATION INFORMATION
Stepping the load from a heavy load to a light load results in an output overshoot. Excess energy stored in the
inductor must be absorbed by the output capacitance. The energy stored in the inductor is described in
equation (5).
EL+1
2 L I2(Joules)
where:
I2+ƪǒIOHǓ2*ǒIOLǓ2ƫǒ(Amperes)2Ǔ
where:
DIOH is the output current under heavy load conditions
DIOL is the output current under light load conditions
Energy in the capacitor is described in equation (7).
EC+1
2 C V2(Joules)
where:
V2+ƪǒVfǓ2*ǒViǓ2ƫǒVolts2Ǔ
where:
DVf is the final peak capacitor voltage
DVi is the initial capacitor voltage
Substituting equation (6) into equation (5), then substituting equation (8) into equation (7), then setting equation
(7) equal to equation (5), and then solving for CO yields the capacitance described in equation (9).
CO+
L ƪǒIOHǓ2*ǒIOLǓ2ƫ
ƪǒVfǓ2*ǒViǓ2ƫ(Farads)
(5)
(6)
(7)
(8)
(9)
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APPLICATION INFORMATION
PROGRAMMING SOFT START
TPS40056 uses a closed-loop approach to ensure a controlled ramp on the output during start-up. Soft-start
is programmed by charging an external capacitor (CSS) via an internally generated current source. The voltage
on C SS is fed into a separate non-inverting input to the error amplifier (in addition to FB and EA_REF). The loop
is closed on the lower of the CSS voltage or the the external reference voltage EA_REF. Once the CSS voltage
rises above the external reference voltage, regulation is based on the external reference. To ensure a controlled
ramp-up of the output voltage the soft-start time should be greater than the L-CO time constant as described
in equation (10).
tSTART w2p L CO
Ǹ(seconds)
There is a direct correlation between tSTART and the input current required during start-up. The faster tSTART,
the higher the input current required during start-up. This relationship is describe in more detail in the section
titled, Programming the Current Limit which follows. The soft-start capacitance, CSS, is described in
equation (11).
For applications in which the VIN supply ramps up slowly, (typically between 50 ms and 100 ms) it may be
necessary to increase the soft-start time to between approximately 2 ms and 5 ms to prevent nuisance UVLO
tripping. The soft-start time should be longer than the time that the VIN supply transitions between 8 V and 9 V.
CSS +2.3 mA
0.7 V tSTART (Farads)
PROGRAMMING CURRENT LIMIT
The TPS40056 uses a two-tier approach for overcurrent protection. The first tier is a pulse-by-pulse protection
scheme. Current limit is implemented on the high-side MOSFET by sensing the voltage drop across the
MOSFET when the gate is driven high. The MOSFET voltage is compared to the voltage dropped across a
resistor connected from VIN pin to the ILIM pin when driven by a constant current sink. If the voltage drop across
the MOSFET exceeds the voltage drop across the ILIM resistor, the switching pulse is immediately terminated.
The MOSFET remains off until the next switching cycle is initiated.
The second tier consists of a fault counter. The fault counter is incremented on an overcurrent pulse and
decremented on a clock cycle without an overcurrent pulse. When the counter reaches seven (7) a restart is
issued and seven soft-start cycles are initiated. Both the upper and lower MOSFETs are turned off during this
period. The counter is decremented on each soft-start cycle. When the counter is decremented to zero, the
PWM is re-enabled. If the fault has been removed the output starts up normally. If the output is still present the
counter counts seven overcurrent pulses and re-enters the second-tier fault mode. See Figure 3 for typical
overcurrent protection waveforms.
The minimum current limit setpoint (ILIM) depends on tSTART, CO, VO, and the load current at turn-on (IL).
ILIM +ƪǒCO VOǓ
tSTART ƫ)IL(Amperes)
(10)
(11)
(12)
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APPLICATION INFORMATION
The current limit programming resistor (RILIM) is calculated using equation (13). Care must be taken in choosing
the values used for VOS and ISINK in the equation. In order to ensure the output current at the overcurrent level,
the minimum value of ISINK and the maximum value of VOS must be used.
RILIM +IOC RDS(on)[max]
ISINK )VOS
ISINK (W)
where:
DISINK is the current into the ILIM pin and is 8.6 µA, minimum
DIOC is the overcurrent setpoint which is the DC output current plus one-half of the peak inductor current
DVOS is the overcurrent comparator offset and is 30 mV, maximum
UDG−02136
HDRV
CLOCK
VVIN−VSW
SS
7 CURRENT LIMIT TRIPS
(HDRV CYCLE TERMINATED BY CURRENT LIMIT
TRIP) 7 SOFT-START CYCLES
VILIM
tBLANKING
Figure 4. Typical Current Limit Protection Waveforms
SYNCHRONIZING TO AN EXTERNAL SUPPLY
The TPS40056 can be synchronized to an external clock through the SYNC pin. Synchronization occurs on the
falling edge of the SYNC signal. The synchronization frequency should be in the range of 20% to 30% higher
than its programmed free-run frequency. The clock frequency at the SYNC pin replaces the master clock
generated by the oscillator circuit. Pulling the SYNC pin low programs the TPS40056 to freely run at the
frequency programmed by RT.
(13)
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APPLICATION INFORMATION
LOOP COMPENSATION
Voltage-mode buck-type converters are typically compensated using Type III networks. Since the TPS40056
includes no voltage feedforward control, the gain of the PWM modulator must be included. The modulator gain
is described in Figure 5.
AMOD +VIN
VSor AMOD(dB) +20 log ǒVIN
VSǓ
Duty dycle, D, varies from 0 to 1 as the control voltage, VC, varies from the minimum ramp voltage to the
maximum ramp voltage, VS. Also, for a synchronous buck converter, D = VO / VIN. To get the control voltage
to output voltage modulator gain in terms of the input voltage and ramp voltage,
D+VO
VIN +VC
VSor VO
VC+VIN
VS
Calculate the Poles and Zeros
For a buck converter using voltage mode control there is a double pole due to the output L-CO. The double pole
is located at the frequency calculated in equation (16).
fLC +1
2p L CO
Ǹ(Hertz)
There is also a zero created by the output capacitance, CO, and its associated ESR. The ESR zero is located
at the frequency calculated in equation (17).
fZ+1
2p ESR CO(Hertz)
Calculate the value of RBIAS to set the output voltage, VOUT.
RBIAS +VEA_REF R1
VOUT *VEA_REF W
The maximum crossover frequency (0 dB loop gain) is calculated in equation (19).
fC+fSW
4(Hertz)
Typically, fC is selected to be close to the midpoint between the L-CO double pole and the ESR zero. At this
frequency, the control to output gain has a –2 slope (−40 dB/decade), while the Type III topology has a +1 slope
(20 dB/decade), resulting in an overall closed loop –1 slope (−20 dB/decade).
Figure 5 shows the modulator gain, L-C filter, output capacitor ESR zero, and the resulting response to be
compensated.
(14)
(15)
(16)
(17)
(18)
(19)
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APPLICATION INFORMATION
Figure 5
VC
PWM MODULATOR RELATIONSHIPS
VS
D = V C / VS
MODULATOR GAIN
vs
SWITCHING FREQUENCY
Modulator Gain − dB
Figure 6
fSW − Switching Frequency − Hz
100 1 k 10 k 100 k
ESR Zero, + 1
LC Filter, − 2
AMOD = VIN / VS
Resultant, − 1
A Type III topology, shown in Figure 7, has two zero-pole pairs in addition to a pole at the origin. The gain and
phase boost of a T ype III topology is shown in Figure 8. The two zeros are used to compensate the L-CO double
pole and provide phase boost. The double pole is used to compensate for the ESR zero and provide controlled
gain roll-off. In many cases the second pole can be eliminated and the amplifier’s gain roll-off used to roll-off
the overall gain at higher frequencies.
RBIAS
Figure 7. Type III Compensation Configuration
UDG−03099
+
R1
R3
C3
C1 R2
78
EA_REF
VOUT COMP
VFB
C2
(optional)
Figure 8. Type III Compensation Gain and Phase
GAIN
180°
−90°
−270°PHASE
+ 1
− 1
− 1
0 dB
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APPLICATION INFORMATION
The poles and zeros for a type III network are described in equations (20).
fZ1 +1
2p R2 C1 (Hertz) fZ2 +1
2p R1 C3 (Hertz)
fP1 +1
2p R2 C2 (Hertz) fP2 +1
2p R3 C3 (Hertz)
The value of R1 is somewhat arbitraty, but influences other component values. A value between 50 kΩ and
100 kΩ usually yields reasonable values.
The unity gain frequency is described in equation (21)
fC+1
2p R1 C2 G(Hertz)
where G is the reciprocal of the modulator gain at fC.
The modulator gain as a function of frequency at fC, is described in equation (22).
AMOD(f) +AMOD ǒfLC
fCǓ2and G +1
AMOD(f)
Minimum Load Resistance
Care must be taken not to load down the output of the error amplifier with the feedback resistor, R2, that is too
small. The error amplifier has a finite output source and sink current which must be considered when sizing R2.
Too small a value does not allow the output to swing over its full range.
R2(MIN) +VC(max)
ISOURCE (min) +3.45 V
2mA +1725 W
CALCULATING THE BOOST AN BP10 BYPASS CAPACITOR
The BOOST capacitance provides a local, low impedance source for the high-side driver . The BOOST capacitor
should be a good quality, high-frequency capacitor. The size of the bypass capacitor depends on the total gate
charge of the MOSFET and the amount of droop allowed on the bypass capacitor. The BOOST capacitance
is described in equation (24).
CBOOST +Qg
DV(Farads)
The 10-V reference pin, BP10V needs to provide energy for both the synchronous MOSFET and the high-side
MOSFET via the BOOST capacitor. Neglecting any efficiency penalty, the BP10V capacitance is described in
equation (25).
CBP10 +ǒQgHS )QgSRǓ
DV(Farads)
(20)
(21)
(22)
(23)
(24)
(25)
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APPLICATION INFORMATION
dv/dt Induced Turn−On
MOSFETs are susceptible to dv/dt turn-on particularly in high-voltage (VDS) applications. The turn-on is caused
by the capacitor divider that is formed by CGD and CGS. High dv/dt conditions and drain-to-source voltage, on
the MOSFET causes current flow through CGD and causes the gate-to-source voltage to rise. If the
gate-to-source voltage rises above the MOSFET threshold voltage, the MOSFET turns on, resulting in large
shoot-through currents. Therefore, the SR MOSFET should be chosen so that the CGD capacitance is smaller
than the CGS capacitance.
High Side MOSFET Power Dissipation
The power dissipated in the external high-side MOSFET is comprised of conduction and switching losses. The
conduction losses are a function of the IRMS current through the MOSFET and the RDS(on) of the MOSFET. The
high-side MOSFET conduction losses are defined by equation (26).
PCOND +ǒIRMSǓ2 RDS(on) ǒ1)TCR ƪTJ*25ƫǓ(Watts)
where:
DTCR is the temperature coefficient of the MOSFET RDS(on)
The TCR varies depending on MOSFET technology and manufacturer but is typically ranges between
.0035 ppm/_C and .010 ppm/_C.
The IRMS current for the high side MOSFET is described in equation (27).
IRMS +IO d
ǸǒAmperesRMSǓ
The switching losses for the high-side MOSFET are descibed in equation (28).
PSW(fsw) +ǒVIN IOUT tSWǓ fSW (Watts)
where:
DIO is the DC output current
DtSW is the switching rise time, typically < 20 ns
DfSW is the switching frequency
Typical switching waveforms are shown in Figure 8.
(26)
(27)
(28)
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APPLICATION INFORMATION
UDG−0213
9
∆
I
ANTI−CROSS
CONDUCTION SYNCHRONOUS
RECTIFIER ON
BODY DIODE
CONDUCTION
BODY DIODE
CONDUCTION
HIGH SIDE ON
ID1
I
D2
IO
SW
0
}
d 1−d
Figure 9. Inductor Current and SW Node Waveforms
The maximum allowable power dissipation in the MOSFET is determined by equation (29).
PT+ǒTJ*TAǓ
qJA (Watts)
where:
PT+PCOND )PSW(fsw) (Watts)
and θJA is the package thermal impedance.
Synchronous Rectifier MOSFET Power Dissipation
The power dissipated in the synchronous rectifier MOSFET is comprised of three components: RDS(on)
conduction losses, body diode conduction losses, and reverse recovery losses. RDS(on) conduction losses can
be found using equation (32) and the RMS current through the synchronous rectifier MOSFET is described in
equation (31).
IRMS +IO 1*d
ǸǒAmperesRMSǓ
The body-diode conduction losses are due to forward conduction of the body diode during the anti−cross
conduction delay time. The body diode conduction losses are described by equation (32).
PDC +2 IO VF tDELAY fSW (Watts)
where:
DVF is the body diode forward voltage
DtDELAY is the total delay time just before the SW node rises.
(29)
(30)
(31)
(32)
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APPLICATION INFORMATION
The 2-multiplier is used because the body-diode conducts twice each cycle (once on the rising edge and once
on the falling edge). The reverse recovery losses are due to the time it takes for the body diode to recovery from
a forward bias to a reverse blocking state. The reverse recovery losses are described in equation (33).
PRR +0.5 QRR VIN fSW (Watts)
where:
DQRR is the reverse recovery charge of the body diode
The total synchronous rectifier MOSFET power dissipation is described in equation (34).
PSR +PDC )PRR )PCOND (Watts)
TPS40056 POWER DISSIPATION
The power dissipation in the TPS40056 is largely dependent on the MOSFET driver currents and the input
voltage. The driver current is proportional to the total gate charge, Qg, of the external MOSFETs. Driver power
(neglecting external gate resistance, refer to [2] can be calculated from equation (35).
PD+Qg VDR fSW (Watts)
And the total power dissipation in the TPS40056, assuming the same MOSFET is selected for both the high-side
and synchronous rectifier is described in equation (36).
PT+ǒ2 PD
VDR )IQǓ VIN (Watts)
or
PT+ǒ2 Qg fSW )IQǓ VIN (Watts)
where:
DIQ is the quiescent operating current (neglecting drivers)
The maximum power capability of the device’s PowerPad package is dependent on the layout as well as air flow.
The thermal impedance from junction to air, assuming 2 oz. copper trace and thermal pad with solder and no
air flow.
θJA = 36.51°C/W
The maximum allowable package power dissipation is related to ambient temperature by equation (29).
Substituting equation (29) into equation (37) and solving for fSW yields the maximum operating frequency for
the TPS4005x. The result is described in equation (38).
fSW +ǒƪǒTJ*TAǓ
ǒqJA VDDǓƫ*IQǓ
ǒ2 QgǓ(Hz)
(33)
(34)
(35)
(36)
(37)
(38)
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LAYOUT CONSIDERATIONS
The PowerPADt package
The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD
derives its name and low thermal impedance from the large bonding pad on the bottom of the device. For
maximum thermal performance, the circuit board must have an area of solder-tinned-copper underneath the
package. The dimensions of this area depends on the size of the PowerPAD package. For a 16-pin TSSOP
(PWP) package the area is 5 mm x 3.4 mm [3].
Thermal vias connect this area to internal or external copper planes and should have a drill diameter suf ficiently
small so that the via hole is effectively plugged when the barrel of the via is plated with copper. This plug is
needed to prevent wicking the solder away from the interface between the package body and the solder-tinned
area under the device during solder reflow. Drill diameters of 0.33 mm (13 mils) works well when 1-oz copper
is plated at the surface of the board while simultaneously plating the barrel of the via. If the thermal vias are not
plugged when the copper plating is performed, then a solder mask material should be used to cap the vias with
a diameter equal to the via diameter of 0.1 mm minimum. This capping prevents the solder from being wicked
through the thermal vias and potentially creating a solder void under the package. Refer to PowerPAD Thermally
Enhanced Package[3] and the mechanical illustration at the end of this document for more information on the
PowerPAD package.
Thermal Pad
6,60 mm
6,20 mm
4,50 mm
4,30 mm
101
X: Minimum PowerPAD = 1.8 mm
X
Y: Minimum PowerPAD = 1.4 mm
Y
Figure 10. PowerPAD Dimensions
MOSFET Packaging
MOSFET package selection depends on MOSFET power dissipation and the projected operating conditions.
In general, for a surface-mount applications, the DPAK style package provides the lowest thermal impedance
(θJA) and, therefore, the highest power dissipation capability. However, the effectiveness of the DPAK depends
on proper layout and thermal management. The θJA specified in the MOSFET data sheet refers to a given
copper area and thickness. In most cases, a lowest thermal impedance of 40°C/W requires one square inch
of 2-ounce copper on a G−10/FR−4 board. Lower thermal impedances can be achieved at the expense of board
area. Please refer to the selected MOSFET’s data sheet for more information regarding proper mounting.
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LAYOUT CONSIDERATIONS
Grounding and Circuit Layout Considerations
The TPS4005x provides separate signal ground (SGND) and power ground (PGND) pins. It is important that
circuit grounds are properly separated. Each ground should consist of a plane to minimize its impedance if
possible. The high power noisy circuits such as the output, synchronous rectifier, MOSFET driver decoupling
capacitor (BP10), and the input capacitor should be connected to PGND plane at the input capacitor.
Sensitive nodes such as the FB resistor divider, RT, and ILIM should be connected to the SGND plane. The
SGND plane should only make a single point connection to the PGND plane.
Component placement should ensure that bypass capacitors (BP10 and BP5) are located as close as possible
to their respective power and ground pins. Also, sensitive circuits such as FB, R T and ILIM should not be located
near high dv/dt nodes such as HDRV, LDRV, BOOST, and the switch node (SW).
The SW pin Schottky diode, D2 in Figure 10, should be placed close to the TPS40056 with short, wide traces
to pins 9 and 12.
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DESIGN EXAMPLE
DInput Voltage: 10 Vdc to 14.4 Vdc
DOutput voltage: 1.25 V ±1% (1.2375 ≤VO≤1.2625)
DOutput current: 8 A (maximum, steady state), 10 A (surge, 10ms duration, 10% duty cycle maximum)
DOutput ripple: 33 mVP-P at 8 A
DOutput load response: 0.1 V => 10% to 90% step load change, from 1 A to 7 A
DOperating temperature: −40°C to 85°C
DfSW=170 kHz
1. Calculate maximum and minimum duty cycles
dMIN +VO(min)
VIN(max) +1.2375
14.4 +0.086 dMAX +VO(max)
VIN(min) +1.2625
10 +0.126
2. Select switching frequency
The switching frequency is based on the minimum duty cycle ratio and the propagation delay of the current limit
comparator. I n order to maintain current limit capability, the on time of the upper MOSFET, tON, must be greater
than 400 ns (see Electrical Characteristics table). Therefore
VO(min)
VIN(max) +tON
TSW or
1
TSW +fSW +ȧ
ȧ
ȧ
ȧ
ȡ
Ȣ
ǒVO(min)
VIN(max)Ǔ
TON ȧ
ȧ
ȧ
ȧ
ȣ
Ȥ
Using 450 ns to provide margin,
fSW +0.086
450 ns +191 kHz
Since the oscillator can vary by 10%, decrease fSW, by 10%
fSW +0.9 191 kHz +172 kHz
and therefore choose a frequency of 170 kHz.
3. Select ∆I
In this case ∆I is chosen so that the converter enters discontinuous mode at 20% of nominal load.
DI+IO 2 0.2 +8 2 0.2 +3.2 A
(39)
(40)
(41)
(42)
(43)
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DESIGN EXAMPLE
4. Calculate the power losses
Power losses in the high-side MOSFET (Si7860DP) at 14.4-VIN where switching losses dominate can be
calculated from equation (27).
IRMS +IO d
Ǹ+8 0.086
Ǹ+2.35 A
substituting (27) into (26) yields
PCOND +2.352 0.008 (1)0.007 (150 *25)) +0.083 W
and from equation (28), the switching losses can be determined.
PSW(fsw) +ǒVIN IO tSWǓ fSW +14.4 V 8A 20 ns 170 kHz +0.39 W
The MOSFET junction temperature can be found by substituting equation (30) into equation (29)
TJ+ǒPCOND )PSWǓ qJA )TA+(0.083 )0.39) 40 )85 +90OC
5. Calculate synchronous rectifier losses
The synchronous rectifier MOSFET has two loss components, conduction, and diode reverse recovery losses.
The conduction losses are due to IRMS losses as well as body diode conduction losses during the dead time
associated with the anti-cross conduction delay.
The IRMS current through the synchronous rectifier from (31)
IRMS +IO 1*d
Ǹ+8 1*0.126
Ǹ+7.48 ARMS
The synchronous MOSFET conduction loss from (26) is:
PCOND +7.482 0.008 (1)0.007(150 *25)) +0.83 W
The body diode conduction loss from (32) is:
PDC
+
2
IO
VFD
tDELAY
fSW
+
2
8.0 A
0.8 V
100 ns
170 kHz
+
0.218 W
The body diode reverse recovery loss from (33) is:
PRR +0.5 QRR VIN fSW +0.5 30 nC 14.4 V 170 kHz +0.037 W
The total power dissipated in the synchronous rectifier MOSFET from (34) is:
PSR +PRR )PCOND )PDC +0.037 )0.83 )0.218 +1.085 W
The junction temperature of the synchronous rectifier at 85°C is:
TJ+PSR qJA )TA+(1.085) 40 )85 +128oC
In typical applications, paralleling the synchronous rectifier MOSFET with a Schottky rectifier increases the
overall converter efficiency by approximately 2% due to the lower power dissipation during the body diode
conduction and reverse recovery periods.
(44)
(45)
(46)
(47)
(48)
(49)
(50)
(51)
(52)
(53)
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DESIGN EXAMPLE
6. Calculate the inductor value
The inductor value is calculated from equation (3).
L+(14.4 *1.25 V) 1.25 V
14.4 V 3.2 A 170 kHz +2.1 mH
A 2.9-µH Coev DXM1306−2R9 or 2.6-µH Panasonic ETQ−P6F2R9LFA can be used.
7. Setting the switching frequency
The clock frequency is set with a resistor (RT) from the RT pin to ground. The value of RT can be found from
equation (1), with fSW in kHz.
RT+ǒ1
fSW 17.82 10*6*23ǓkW+307 kWNuse 309 kW
8. Calculating the output capacitance (CO)
In this example the output capacitance is determined by the load response requirement of ∆V = 0.1 V for a 1 A
to 7 A step load. CO can be calculated using (9)
CO+2.9 m ǒ(8A
)2*(1A
)2Ǔ
ǒ(1.25)2*(1.15)2Ǔ+761 mF
Using (4) we can calculate the ESR required to meet the output ripple requirements.
33 mV +3.2 AǒESR )1
8 761 mF 170 kHzǓ
ESR +10.3 mW*1.0 mW+9.3 mW
For this design example two (2) Panasonic SP EEFUEOD471R capacitors, (2.0 V, 470 µF, 12 mΩ) are used.
9. Calculate the soft-start capacitor (CSS)
This design requires a soft−start time (tSTART) of 1 ms. CSS can be calculated on (11)
CSS +2.3 mA
0.7 V 1ms+3.29 nF +3300 pF
(54)
(55)
(56)
(57)
(58)
(59)
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DESIGN EXAMPLE
10. Calculate the current limit resistor (RILIM)
The current limit set point depends on tSTART, VO,CO and ILOAD at start-up as shown in equation (12). For this
design,
ILIM u940 mF 1.25 V
1ms )8.0 A +9.2 A
For this design, set ILIM for 11.0 ADC minimum. From equation (13), with IOC equal to the DC output surge current
plus one-half the ripple current of 3.2 A and RDS(on) is increased 30% (1.3 * 0.008) to allow for MOSFET heating.
RILIM +12.6 A 0.0104W
8.6 mA)(0.03)
8.6 mA+15.24 kW*3.5 kW+11.74 kW^11.8 W
11. Calculate loop compensation values
Calculate the DC modulator gain (AMOD) from equation (14)
AMOD +12
2+6.0 AMOD(dB) +20 log (6)+15.6 dB
Calculate the output filter L-CO poles and CO ESR zeros from (16) and (17)
fLC +1
2pL CO
Ǹ+1
2p2.9 mH 940 mF
Ǹ+3.05 kHz
and
fZ+1
2p ESR CO+1
2p 0.006 940 mF+28.2 kHz
Select the close-loop 0 dB crossover frequency, fC. For this example fC = 20 kHz.
Select the double zero location for the T ype III compensation network at the output filter double pole at 3.05 kHz.
Select the double pole location for the Type III compensation network at the output capacitor ESR zero at
28.2 kHz.
The amplifier gain at the crossover frequency of 20 kHz is determined by the reciprocal of the modulator gain
AMOD at the crossover frequency from equation (22).
AMOD(f) +AMOD ǒfLC
fCǓ2
+6 ǒ3.05 kHz
20 kHz Ǔ2
+0.14
And also from equation (22).
G+1
AMOD(f) +1
0.14 +7.14
Choose R1 = 100 kΩ
(60)
(61)
(62)
(63)
(64)
(65)
(66)
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DESIGN EXAMPLE
The poles and zeros for a type III network are described in equations (20) and (21).
fZ2 +1
2p R1 C3 NC3 +1
2p 100 kW 3.05 kHz +522 pF, choose 560 pF
fP2 +1
2p R3 C3 NR3 +1
2p 560 pF 28.2 kHz +10.08 kW, choose 10 kW
fC+1
2p R1 C2 GNC2 +1
2p 100 kW 7.14 20 kHz +11.1 pF, choose 10 pF
fP1 +1
2p R2 C2 NR2 +1
2p 10 pF 28.2 kHz +564 kW, choose 562 kW
fZ1 +1
2p R2 C1 NC1 +1
2p 562 kW 3.05 kHz +92.9 pF, choose 100 pF
Calculate the value of RBIAS from equation (17) with R1 = 100 kΩ. Since the output of 1.25-V is within the
EA_REF input specification of 0.5 V to 1.5 V, an RBIAS resistor is not required.
CALCULATING THE BOOST AND BP10V BYPASS CAPACITANCE
The size of the bypass capacitor depends on the total gate charge of the MOSFET being used and the amount
of droop allowed on the bypass cap. The BOOST capacitance for the Si7860DP, allowing for a 0.5 voltage droop
on the BOOST pin from equation (24) is:
CBOOST +Qg
DV+18 nC
0.5 V +36 nF
and the BP10V capacitance from (25) is
CBP(10 V) +QgHS )QgSR
DV+2 Qg
DV+36 nC
0.5 V +72 nF
For this application, a 0.1-µF capacitor is used for the BOOST bypass capacitor and a 1.0-µF capacitor is used
for the BP10V bypass.
Figure 10 s h o w s c omponent selection for the 10-V to 14.4-V to 1.25-V at 8 A dc-to-dc converter specified in the
design example.
REFERENCES
1. Balogh, Laszlo, Design and Application Guide for High Speed MOSFET Gate Drive Circuits, Texas
Instruments/Unitrode Corporation, Power Supply Design Seminar, SEM−1400 Topic 2.
2. PowerPAD Thermally Enhanced Package Texas Instruments, Semiconductor Group, Technical Brief: TI
Literature No. SLMA002
(67)
(68)
(69)
(70)
(71)
(72)
(73)
SLVS612 − APRIL 2006
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330 µF330 µF
165 kΩ
100 pF
1.0 µF
50 V
0.1 µF
3.3 Ω
2.9 µH
22 µF
50 V 22 µF
50 V
10 kΩ
100 kΩ470 µF470 µF
VTT
VIN
1.0 µF
1.0 µF
UDG−03100
560 pF
11.8 kΩ
10 pF
100 pF
3300 pF
562 kΩ
VTRKIN
Si7860DP
Si7860DP
R4
10 kΩ
R5
10 kΩ
1
2
3
4
16
15
14
13
ILIM
VIN
BOOST
HDRV
SYNC
RT
BP5
EA_REF
TPS40056PWP
5
6
7
8
12
11
10
9
SW
BP10
LDRV
PGND
SGND
SS
VFB
COMP
+
−
+
−
D1
PWP
C2
C1
R2
CSS
RT
R3
R1
C3
D2 RSW
Figure 11. 12-V to 1.25-V at 8-A DC-to-DC Converter (DDR) Design Example