AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 1
SwitchReg™
General Description
The AAT1121 SwitchReg is a member of
AnalogicTech's Total Power Management IC™
(TPMIC™) product family. It is a 1.5MHz step-
down converter with an input voltage range of
2.7V to 5.5V and output as low as 0.6V. Its low
supply current, small size, and high switching fre-
quency make the AAT1121 the ideal choice for
portable applications.
The AAT1121 delivers 250mA of load current, while
maintaining a low 30μA no load quiescent current.
The 1.5MHz switching frequency minimizes the size
of external components, while keeping switching
losses low. The AAT1121 feedback and control
delivers excellent load regulation and transient
response with a small output inductor and capacitor.
The AAT1121 is available in a Pb-free, 8-pin,
2x2mm TDFN package and is rated over the -40°C
to +85°C temperature range.
Features
•V
IN Range: 2.7V to 5.5V
•V
OUT Range: 0.6V to VIN
• 250mA Max Output Current
• Up to 96% Efficiency
•30μA Typical Quiescent Current
• 1.5MHz Switching Frequency
• Soft-Start Control
• Over-Temperature and Current Limit
Protection
• 100% Duty Cycle Low-Dropout Operation
•<1μA Shutdown Current
• Small External Components
• Ultra-Small TDFN22-8 Package
• Temperature Range: -40°C to +85°C
Applications
• Bluetooth™ Headsets
• Cellular Phones
• Digital Cameras
• Handheld Instruments
• Portable Music Players
• USB Devices
Typical Application
3.0μH
L1
R
1
118kΩ
R
2
59kΩ
C
1
4.7µF C
2
4.7µF
EN FB
VP
VIN
LX
PGND
GND
AAT1121
V
IN
V
O
= 1.8V 250m
A
Pin Descriptions
Pin Configuration
TDFN22-8
(Top View)
GND
FB
VP
VIN
EN
N/C
PGND
LX
3
4
1
2
6
5
8
7
Pin # Symbol Function
1 VP Input power pin; connected to the source of the P-channel MOSFET.
Connect to the input capacitor.
2 VIN Input bias voltage for the converter.
3 GND Non-power signal ground pin.
4 FB Feedback input pin. Connect this pin to an external resistive divider for
adjustable output.
5 N/C No connect.
6 EN Enable pin. A logic high enables normal operation. A logic low shuts down
the converter.
7 LX Switching node. Connect the inductor to this pin. It is connected internally to
the drain of both high- and low-side MOSFETs.
8 PGND Input power return pin; connected to the source of the N-channel MOSFET.
Connect to the output and input capacitor return.
EP Exposed paddle (bottom): connect to ground directly beneath the package.
AAT1121
1.5MHz, 250mA Step-Down Converter
21121.2006.04.1.0
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 3
Absolute Maximum Ratings1
Thermal Information
Symbol Description Value Units
PDMaximum Power Dissipation 2 W
θJA Thermal Resistance250 °C/W
Symbol Description Value Units
VIN Input Voltage and Bias Power to GND 6.0 V
VLX LX to GND -0.3 to VIN + 0.3 V
VOUT FB to GND -0.3 to VIN + 0.3 V
VEN EN to GND -0.3 to 6.0 V
TJOperating Junction Temperature Range -40 to 150 °C
TLEAD Maximum Soldering Temperature (at leads, 10 sec) 300 °C
1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at condi-
tions other than the operating conditions specified is not implied. Only one Absolute Maximum Rating should be applied at any one time.
2. Mounted on an FR4 board.
Electrical Characteristics1
VIN = 3.6V, TA= -40°C to +85°C, unless otherwise noted; typical values are TA= 25°C.
Symbol Description Conditions Min Typ Max Units
VIN Input Voltage 2.7 5.5 V
VIN Rising 2.6 V
VUVLO UVLO Threshold Hysteresis 250 mV
VIN Falling 2.0 V
VOUT Output Voltage Tolerance2IOUT = 0 to 250mA, -3.0 3.0 %
VIN = 2.7V to 5.5V
VOUT Output Voltage Range 0.6 VIN V
IQQuiescent Current No Load 30 μA
ISHDN Shutdown Current EN = GND 1.0 μA
ILIM P-Channel Current Limit 600 mA
RDS(ON)H High-Side Switch On Resistance 0.59 Ω
RDS(ON)L Low-Side Switch On Resistance 0.42 Ω
ILXLEAK LX Leakage Current VIN = 5.5V, VLX = 0 to VIN 1.0 μA
ΔVLinereg/ΔVIN Line Regulation VIN = 2.7V to 5.5V 0.2 %/V
VFB Feedback Threshold Voltage Accuracy VIN = 3.6V 0.597 0.606 0.615 V
IFB FB Leakage Current VOUT = 1.0V 0.2 μA
FOSC Oscillator Frequency 1.5 MHz
TSStartup Time From Enable to Output 100 μs
Regulation
TSD Over-Temperature Shutdown Threshold 140 °C
THYS Over-Temperature Shutdown Hysteresis 15 °C
VEN(L) Enable Threshold Low 0.6 V
VEN(H) Enable Threshold High 1.4 V
IEN Input Low Current VIN = VEN = 5.5V -1.0 1.0 μA
AAT1121
1.5MHz, 250mA Step-Down Converter
41121.2006.04.1.0
1. The AAT1121 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured
by design, characterization, and correlation with statistical process controls.
2. Output voltage tolerance is independent of feedback resistor network accuracy.
Typical Characteristics
DC Load Regulation
(V
OUT
= 3.0V; L = 4.7µH)
Output Current (mA)
Output Error (%)
-1.0
-0.5
0.0
0.5
1.0
0.1 1 10 100 1000
V
IN
= 5.0V
V
IN
= 4.2V V
IN
= 3.6V
Efficiency vs. Load
(V
OUT
= 3.0V; L = 4.7µH)
Output Current (mA)
Efficiency (%)
40
50
60
70
80
90
100
0.1 1 10 100 1000
V
IN
= 3.6V
V
IN
= 4.2V
V
IN
= 5.0V
DC Load Regulation
(V
OUT
= 1.8V; L = 3.3µH)
Output Current (mA)
Output Error (%)
-1.0
-0.5
0.0
0.5
1.0
0.1 1 10 100 1000
V
IN
= 4.2V
V
IN
= 3.6V
V
IN
= 2.7V
Efficiency vs. Load
(VOUT = 1.8V; L = 3.3µH)
Output Current (mA)
Efficiency (%)
40
50
60
70
80
90
100
0.1 1 10 100 1000
VIN = 3.6V
VIN = 2.7V
VIN = 4.2V
DC Load Regulation
(V
OUT
= 1.2V; L = 1.5µH)
Output Current (mA)
Output Error (%)
-1.0
-0.5
0.0
0.5
1.0
0.1 1 10 100 1000
V
IN
= 3.6V
V
IN
= 4.2V
V
IN
= 2.7V
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 5
Typical Characteristics
No Load Quiescent Current vs. Input Voltage
Input Voltage (V)
Supply Current (µA)
10
15
20
25
30
35
40
45
50
2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5
85°C
25°C
-40°C
Frequency Variation vs. Input Voltage
Input Voltage (V)
Frequency Variation (%)
-4.0
-3.0
-2.0
-1.0
0.0
1.0
2.0
2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5
V
OUT
= 1.8V
V
OUT
= 3.0V
Switching Frequency Variation
vs. Temperature
(VIN = 3.6V; VOUT = 1.8V)
Temperature (°
°C)
Variation (%)
-10.0
-8.0
-6.0
-4.0
-2.0
0.0
2.0
4.0
6.0
8.0
10.0
-40 -20 0 20 40 60 80 100
Output Voltage Error vs. Temperature
(VIN = 3.6V; VOUT = 1.8V; IOUT = 250mA)
Temperature (°
°C)
Output Error (%)
-3.0
-2.0
-1.0
0.0
1.0
2.0
3.0
-40 -20 0 20 40 60 80 100
Line Regulation
(VOUT = 1.8V)
Input Voltage (V)
Accuracy (%)
-0.3
-0.2
-0.1
0.0
0.1
0.2
0.3
0.4
0.5
0.6
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0
IOUT = 250mA
IOUT = 10mA
IOUT = 0mA
IOUT = 50mA
IOUT = 150mA
Soft Start
(V
IN
= 3.6V; V
OUT
= 1.8V;
I
OUT
= 250mA; C
FF
= 100pF)
Enable and Output Voltage
(top) (V)
Inductor Current
(bottom) (A)
Time (100µs/div)
-5.0
-4.0
-3.0
-2.0
-1.0
0.0
1.0
2.0
3.0
4.0
5.0
-0.2
-0.4
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
V
EN
V
O
I
L
AAT1121
1.5MHz, 250mA Step-Down Converter
61121.2006.04.1.0
Typical Characteristics
Line Response
(V
OUT
= 1.8V @ 250mA; C
FF
= 100pF)
Output Voltage
(top) (V)
Input Voltage
(bottom) (V)
Time (25µs/div)
1.50
1.55
1.60
1.65
1.70
1.75
1.80
1.85
1.90
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
7.0
V
O
V
IN
Load Transient Response
(10mA to 250mA; V
IN
= 3.6V; V
OUT
= 1.8V; C
OUT
= 4.7µF)
Output Voltage
(top) (V)
Load and Inductor Current
(bottom) (200mA/div)
Time (25µs/div)
1.2
1.3
1.4
1.5
1.6
1.7
1.8
1.9
2.0
-0.2
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
V
O
I
LX
I
O
Load Transient Response
(10mA to 250mA; V
IN
= 3.6V; V
OUT
= 1.8V;
C
OUT
= 4.7µF; C
FF
= 100pF)
Output Voltage
(top) (V)
Load and Inductor Current
(bottom) (200mA/div)
Time (25µs/div)
1.2
1.3
1.4
1.5
1.6
1.7
1.8
1.9
2.0
V
O
I
LX
I
O
N-Channel R
DS(ON)
vs. Input Voltage
Input Voltage (V)
R
DS(ON)L
(mΩ
Ω
)
300
350
400
450
500
550
600
650
700
750
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0
120°C100°C
85°C
25°C
P-Channel R
DS(ON)
vs. Input Voltage
Input Voltage (V)
R
DS(ON)H
(mΩ
Ω
)
300
400
500
600
700
800
900
1000
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0
120°C 100°C
85°C
25°C
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 7
Typical Characteristics
Output Ripple
(V
IN
= 3.6V; V
OUT
= 1.8V; I
OUT
= 250mA)
Output Voltage
(AC Coupled) (top) (V)
Inductor Current
(bottom) (A)
Time (200ns/div)
-120
-100
-80
-60
-40
-20
0
20
40
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
V
O
I
L
Output Ripple
(V
IN
= 3.6V; V
OUT
= 1.8V; I
OUT
= 1mA)
Output Voltage
(AC Coupled) (top) (mV)
Inductor Current
(bottom) (A)
Time (2µs/div)
-120
-100
-80
-60
-40
-20
0
20
40
-0.01
0.00
0.01
0.02
0.03
0.04
0.05
0.06
0.07
V
O
I
L
AAT1121
1.5MHz, 250mA Step-Down Converter
81121.2006.04.1.0
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 9
Functional Description
The AAT1121 is a high performance 250mA,
1.5MHz monolithic step-down converter designed
to operate with an input voltage range of 2.7V to
5.5V. The converter operates at 1.5MHz, which
minimizes the size of external components. Typical
values are 3.3μH for the output inductor and 4.7μF
for the ceramic output capacitor.
The device is designed to operate with an output
voltage as low as 0.6V. Power devices are sized for
250mA current capability while maintaining over
90% efficiency at full load. Light load efficiency is
maintained at greater than 80% down to 1mA of
load current.
At dropout, the converter duty cycle increases to
100% and the output voltage tracks the input volt-
age minus the RDS(ON) drop of the P-channel high-
side MOSFET.
A high-DC gain error amplifier with internal com-
pensation controls the output. It provides excellent
transient response and load/line regulation. Soft
start eliminates any output voltage overshoot when
the enable or the input voltage is applied.
Functional Block Diagram
EN
LX
Err
Amp
Logic
DH
DL
PGND
VP
FB
GND
Voltage
Reference
INPUT
VIN
AAT1121
1.5MHz, 250mA Step-Down Converter
10 1121.2006.04.1.0
Control Loop
The AAT1121 is a 250mA current mode step-down
converter. The current through the P-channel
MOSFET (high side) is sensed for current loop
control, as well as short-circuit and overload pro-
tection. A fixed slope compensation signal is added
to the sensed current to maintain stability for duty
cycles greater than 50%. The peak current mode
loop appears as a voltage-programmed current
source in parallel with the output capacitor.
The output of the voltage error amplifier programs
the current mode loop for the necessary peak
switch current to force a constant output voltage for
all load and line conditions. Internal loop compen-
sation terminates the transconductance voltage
error amplifier output. The error amplifier reference
is fixed at 0.6V.
Soft Start / Enable
Soft start increases the inductor current limit point in
discrete steps when the input voltage or enable
input is applied. It limits the current surge seen at
the input and eliminates output voltage overshoot.
When pulled low, the enable input forces the
AAT1121 into a low-power, non-switching state. The
total input current during shutdown is less than 1μA.
Current Limit and
Over-Temperature Protection
For overload conditions, the peak input current is lim-
ited. As load impedance decreases and the output
voltage falls closer to zero, more power is dissipated
internally, raising the device temperature. Thermal
protection completely disables switching when inter-
nal dissipation becomes excessive, protecting the
device from damage. The junction over-temperature
threshold is 140°C with 15°C of hysteresis.
Under-Voltage Lockout
Internal bias of all circuits is controlled via the VIN
power. Under-voltage lockout (UVLO) guarantees
sufficient VIN bias and proper operation of all inter-
nal circuits prior to activation.
Applications Information
Inductor Selection
The step-down converter uses peak current mode
control with slope compensation to maintain stability
for duty cycles greater than 50%. The output induc-
tor value must be selected so the inductor current
down slope meets the internal slope compensation
requirements. The internal slope compensation for
the adjustable and low-voltage fixed versions of the
AAT1121 is 0.45A/μsec. This equates to a slope
compensation that is 75% of the inductor current
down slope for a 1.8V output and 3.0μH inductor.
This is the internal slope compensation for the
AAT1121. When externally programming to 3.0V,
the calculated inductance is 5.0μH.
In this case, a standard 4.7μH value is selected.
For most designs, the AAT1121 operates with an
inductor value of 1μH to 4.7μH. Table 1 displays
inductor values for the AAT1121 with different output
voltage options.
Manufacturer's specifications list both the inductor
DC current rating, which is a thermal limitation, and
the peak current rating, which is determined by the
saturation characteristics. The inductor should not
show any appreciable saturation under normal load
conditions. Some inductors may meet the peak and
average current ratings yet result in excessive
losses due to a high DCR. Always consider the
losses associated with the DCR and its effect on
the total converter efficiency when selecting an
inductor.
0.75 ⋅ V
O
L = =
≈
1.67
⋅ V
O
= 1.67 ⋅ 3.0V = 5.0µH
m
0.75
⋅
V
O
0.45A
µsec
A
µsec
A
A
µsec
0.75 ⋅ V
O
m = = = 0.45
L
0.75 ⋅ 1.8V
3.0µH
A
µsec
Table 1: Inductor Values.
The 3.0μH CDRH2D09 series inductor selected
from Sumida has a 150mΩDCR and a 470mA DC
current rating. At full load, the inductor DC loss is
9.375mW which gives a 2.08% loss in efficiency for
a 250mA, 1.8V output.
Input Capacitor
Select a 4.7μF to 10μF X7R or X5R ceramic capac-
itor for the input. To estimate the required input
capacitor size, determine the acceptable input rip-
ple level (VPP) and solve for CIN. The calculated
value varies with input voltage and is a maximum
when VIN is double the output voltage.
Always examine the ceramic capacitor DC voltage
coefficient characteristics when selecting the prop-
er value. For example, the capacitance of a 10μF,
6.3V, X5R ceramic capacitor with 5.0V DC applied
is actually about 6μF.
The maximum input capacitor RMS current is:
The input capacitor RMS ripple current varies with
the input and output voltage and will always be less
than or equal to half of the total DC load current.
for VIN = 2 x VO
The term appears in both the input
voltage ripple and input capacitor RMS current
equations and is a maximum when VOis twice VIN.
This is why the input voltage ripple and the input
capacitor RMS current ripple are a maximum at
50% duty cycle.
The input capacitor provides a low impedance loop
for the edges of pulsed current drawn by the
AAT1121. Low ESR/ESL X7R and X5R ceramic
capacitors are ideal for this function. To minimize
stray inductance, the capacitor should be placed as
closely as possible to the IC. This keeps the high
frequency content of the input current localized,
minimizing EMI and input voltage ripple.
The proper placement of the input capacitor (C1)
can be seen in the evaluation board layout in
Figure 2.
A laboratory test set-up typically consists of two
long wires running from the bench power supply to
the evaluation board input voltage pins. The induc-
tance of these wires, along with the low-ESR
ceramic input capacitor, can create a high Q net-
work that may affect converter performance. This
problem often becomes apparent in the form of
excessive ringing in the output voltage during load
transients. Errors in the loop phase and gain meas-
urements can also result.
Since the inductance of a short PCB trace feeding
the input voltage is significantly lower than the
power leads from the bench power supply, most
applications do not exhibit this problem.
⎛⎞
· 1
-
⎝⎠
V
O
V
IN
V
O
V
IN
I
O
RMS(MAX)
I2
=
⎛⎞
· 1
-
= D
· (1 - D) = 0.5
2
=
⎝⎠
V
O
V
IN
V
O
V
IN
1
2
⎛⎞
I
RMS
= I
O
· · 1
-
⎝⎠
V
O
V
IN
V
O
V
IN
C
IN(MIN)
= 1
⎛⎞
- ESR
·
4
·
F
S
⎝⎠
V
PP
I
O
⎛⎞
· 1
-
= for VIN = 2 × VO
⎝⎠
VO
VIN
VO
VIN
1
4
⎛⎞
· 1
-
⎝⎠
V
O
V
IN
C
IN
=
V
O
V
IN
⎛⎞
- ESR
·
F
S
⎝⎠
V
PP
I
O
Output Voltage (V) L1 (μH)
1.0 1.5
1.2 2.2
1.5 2.7
1.8 3.0
2.5 3.9
3.0 4.7
3.3 5.6
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 11
In applications where the input power source lead
inductance cannot be reduced to a level that does
not affect the converter performance, a high ESR
tantalum or aluminum electrolytic should be placed
in parallel with the low ESR, ESL bypass ceramic.
This dampens the high Q network and stabilizes
the system.
Output Capacitor
The output capacitor limits the output ripple and
provides holdup during large load transitions. A
4.7μF to 10μF X5R or X7R ceramic capacitor typi-
cally provides sufficient bulk capacitance to stabi-
lize the output during large load transitions and has
the ESR and ESL characteristics necessary for low
output ripple. For enhanced transient response
and low temperature operation application, a 10μF
(X5R, X7R) ceramic capacitor is recommended to
stabilize extreme pulsed load conditions.
The output voltage droop due to a load transient is
dominated by the capacitance of the ceramic out-
put capacitor. During a step increase in load cur-
rent, the ceramic output capacitor alone supplies
the load current until the loop responds. Within two
or three switching cycles, the loop responds and
the inductor current increases to match the load
current demand. The relationship of the output
voltage droop during the three switching cycles to
the output capacitance can be estimated by:
Once the average inductor current increases to the
DC load level, the output voltage recovers. The
above equation establishes a limit on the minimum
value for the output capacitor with respect to load
transients.
The internal voltage loop compensation also limits
the minimum output capacitor value to 4.7μF. This
is due to its effect on the loop crossover frequency
(bandwidth), phase margin, and gain margin.
Increased output capacitance will reduce the
crossover frequency with greater phase margin.
The maximum output capacitor RMS ripple current
is given by:
Dissipation due to the RMS current in the ceramic
output capacitor ESR is typically minimal, resulting in
less than a few degrees rise in hot-spot temperature.
Adjustable Output Resistor Selection
Resistors R1 and R2 of Figure 1 program the output
to regulate at a voltage higher than 0.6V. To limit the
bias current required for the external feedback resis-
tor string while maintaining good noise immunity, the
suggested value for R2 is 59kΩ. Decreased resistor
values are necessary to maintain noise immunity on
the FB pin, resulting in increased quiescent current.
Table 2 summarizes the resistor values for various
output voltages.
With enhanced transient response for extreme
pulsed load application, an external feed-forward
capacitor, (C3 in Figure 1), can be added.
Table 2: Adjustable Resistor Values For
Step-Down Converter.
R2 = 59kΩΩR2 = 221kΩΩ
VOUT (V) R1 (kΩΩ) R1 (kΩΩ)
0.8 19.6 75
0.9 29.4 113
1.0 39.2 150
1.1 49.9 187
1.2 59.0 221
1.3 68.1 261
1.4 78.7 301
1.5 88.7 332
1.8 118 442
1.85 124 464
2.0 137 523
2.5 187 715
3.3 267 1000
⎛⎞
⎝⎠
R1 = -1
·
R2 = - 1
·
59kΩ = 267kΩ
V
OUT
V
REF
⎛⎞
⎝⎠
3.3V
0.6V
1
23
V
OUT
· (V
IN(MAX)
- V
OUT
)
RMS(MAX)
IL
·
F
·
V
IN(MAX)
=·
·
C
OUT
=
3
·
ΔI
LOAD
V
DROOP
·
F
S
AAT1121
1.5MHz, 250mA Step-Down Converter
12 1121.2006.04.1.0
Thermal Calculations
There are three types of losses associated with
the AAT1121 step-down converter: switching loss-
es, conduction losses, and quiescent current loss-
es. Conduction losses are associated with the
RDS(ON) characteristics of the power output switch-
ing devices. Switching losses are dominated by
the gate charge of the power output switching
devices. At full load, assuming continuous conduc-
tion mode (CCM), a simplified form of the losses is
given by:
IQis the step-down converter quiescent current.
The term tsw is used to estimate the full load step-
down converter switching losses.
For the condition where the step-down converter is
in dropout at 100% duty cycle, the total device dis-
sipation reduces to:
Since RDS(ON), quiescent current, and switching
losses all vary with input voltage, the total losses
should be investigated over the complete input
voltage range.
Given the total losses, the maximum junction tem-
perature can be derived from the θJA for the
TDFN22-8 package which is 50°C/W.
T
J(MAX)
=
P
TOTAL
·
Θ
JA
+ T
AMB
P
TOTAL
= I
O
2
· R
DSON(H)
+ I
Q
· V
IN
P
TOTAL
I
O
2
· (R
DSON(H)
· V
O
+ R
DSON(L)
· [V
IN
- V
O
])
V
IN
=
+ (t
sw
· F · I
O
+ I
Q
) · V
IN
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 13
Figure 1: AAT1121 Schematic.
R1
Adj.
R2
59kΩ
L1
4.7μF
C2
4.7μF
C1
VP
1
GND
3
N/C
5
EN
6
LX
7
PGND
8
VIN
2
FB
4
AAT1121
U1
VIN
GND
+VOUT
GND
C3
(optional)
100pF
LX
Layout
The suggested PCB layout for the AAT1121 is
shown in Figures 2, 3, and 4. The following guide-
lines should be used to help ensure a proper layout.
1. The input capacitor (C1) should connect as
closely as possible to VP (Pin 1), PGND (Pin 8),
and GND (Pin 3)
2. C2 and L1 should be connected as closely as
possible. The connection of L1 to the LX pin
should be as short as possible. Do not make the
node small by using narrow trace. The trace
should be kept wide, direct and short.
3. The feedback pin (Pin 4) should be separate
from any power trace and connect as closely as
possible to the load point. Sensing along a
high-current load trace will degrade DC load
regulation. Feedback resistors should be
placed as closely as possible to the FB pin (Pin
4) to minimize the length of the high imped-
ance feedback trace. If possible, they should
also be placed away from the LX (switching
node) and inductor to improve noise immunity.
4. The resistance of the trace from the load return
to PGND (Pin 8) and GND (Pin 3) should be
kept to a minimum. This will help to minimize
any error in DC regulation due to differences in
the potential of the internal signal ground and
the power ground.
5. A high density, small footprint layout can be
achieved using an inexpensive, miniature, non-
shielded, high DCR inductor.
AAT1121
1.5MHz, 250mA Step-Down Converter
14 1121.2006.04.1.0
Figure 2: AAT1121 Evaluation Board Figure 3: Exploded View of AAT1121
Top Side Layout. Evaluation Board Top Side Layout.
Figure 4: AAT1121 Evaluation Board
Bottom Side Layout.
Step-Down Converter Design Example
Specifications
VO= 1.8V @ 250mA, Pulsed Load ΔILOAD = 200mA
VIN = 2.7V to 4.2V (3.6V nominal)
FS= 1.5MHz
TAMB = 85°C
1.8V Output Inductor
(use 3.3μH; see Table 1)
For Sumida inductor CDRH2D09-3R0, 3.3μH, DCR = 150mΩ.
1.8V Output Capacitor
VDROOP = 0.1V
1
23
1 1.8V · (4.2V - 1.8V)
3.0µH · 1.5MHz · 4.2V
23
RMS
IL1 · F
S
· V
IN(MAX)
= ·
·
3 · ΔI
LOAD
V
DROOP
· F
S
3 · 0.2A
0.1V · 1.5MHz
C
OUT
= = = 4µF; use 4.7µF
· = 66mArms
·
(V
O
) · (V
IN(MAX)
- V
O
)
=
P
esr
= esr · I
RMS2
= 5mΩ · (66mA)
2
= 21.8µW
V
O
V
O
1.8
V
1.8V
ΔI
L1
=
⋅ 1 - = ⋅ 1 - = 228m
A
L1 ⋅ F
V
IN
3.0µH ⋅ 1.5MHz
4.2V
I
PKL1
= I
O
+ ΔI
L1
= 250mA + 114mA = 364mA
2
P
L1
= I
O
2
⋅ DCR = 250mA
2
⋅ 150mΩ = 9.375mW
⎛
⎝⎞
⎠
⎛
⎝⎞
⎠
L1 = 1.67 ⋅ V
O2
= 1.67 ⋅ 1.8V = 3µH
µsec
A
µsec
A
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 15
Input Capacitor
Input Ripple VPP = 25mV
AAT1121 Losses
T
J(MAX)
= T
AMB
+ Θ
JA
· P
LOSS
= 85°C + (50°C/W) · 34.5mW = 86.7°C
P
TOTAL
+ (t
sw
· F · I
O
+ I
Q
) · V
IN
I
O
2
· (R
DSON(HS)
· V
O
+ R
DSON(LS)
· [V
IN
-V
O
]
)
V
IN
=
=
+ (5ns · 1.5MHz · 0.2A + 30µA) · 4.2V = 34.5m
W
0.2
2
· (0.7
Ω
·
1.8V + 0.7Ω
·
[4.2V - 1.8V])
4.2V
I
O
RMS
I
P = esr
·
I
RMS
2
= 5mΩ
·
(0.1A)
2
= 0.05mW
2
= = 0.1Arms
C
IN
= = = 1.38µF; use 4.7µF
1
⎛⎞
- ESR
·
4
·
F
S
⎝⎠
V
PP
I
O
1
⎛⎞
- 5mΩ
·
4
·
1.5MHz
⎝⎠
25mV
0.2A
AAT1121
1.5MHz, 250mA Step-Down Converter
16 1121.2006.04.1.0
Table 3: Evaluation Board Component Values.
Table 4: Suggested Inductors and Suppliers.
Inductance Max DC DCR Size (mm)
Manufacturer Part Number (μH) Current (mA) (mΩΩ) LxWxH Type
Sumida CDRH2D09-1R5 1.5 730 88 3.0x3.0x1.0 Shielded
Sumida CDRH2D09-2R2 2.2 600 115 3.0x3.0x1.0 Shielded
Sumida CDRH2D09-2R5 2.5 530 135 3.0x3.0x1.0 Shielded
Sumida CDRH2D09-3R0 3 470 150 3.0x3.0x1.0 Shielded
Sumida CDRH2D09-3R9 3.9 450 180 3.0x3.0x1.0 Shielded
Sumida CDRH2D09-4R7 4.7 410 230 3.0x3.0x1.0 Shielded
Sumida CDRH2D09-5R6 5.6 370 260 3.0x3.0x1.0 Shielded
Sumida CDRH2D11-1R5 1.5 900 54 3.2x3.2x1.2 Shielded
Sumida CDRH2D11-2R2 2.2 780 78 3.2x3.2x1.2 Shielded
Sumida CDRH2D11-3R3 3.3 600 98 3.2x3.2x1.2 Shielded
Sumida CDRH2D11-4R7 4.7 500 135 3.2x3.2x1.2 Shielded
Taiyo Yuden NR3010 1.5 1200 80 3.0x3.0x1.0 Shielded
Taiyo Yuden NR3010 2.2 1100 95 3.0x3.0x1.0 Shielded
Taiyo Yuden NR3010 3.3 870 140 3.0x3.0x1.0 Shielded
Taiyo Yuden NR3010 4.7 750 190 3.0x3.0x1.0 Shielded
FDK MIPWT3226D-1R5 1.5 1200 90 3.2x2.6x0.8 Chip shielded
FDK MIPWT3226D-2R2 2.2 1100 100 3.2x2.6x0.8 Chip shielded
FDK MIPWT3226D-3R0 3 1000 120 3.2x2.6x0.8 Chip shielded
FDK MIPWT3226D-4R2 4.2 900 140 3.2x2.6x0.8 Chip shielded
Output Voltage R2 = 59kΩΩR2 = 221kΩΩ1
VOUT (V) R1 (kΩΩ) R1 (kΩΩ) L1 (μH)
0.62— — 1.5
0.8 19.6 75 1.5
0.9 29.4 113 1.5
1.0 39.2 150 1.5
1.1 49.9 187 1.5
1.2 59.0 221 1.5
1.3 68.1 261 1.5
1.4 78.7 301 2.2
1.5 88.7 332 2.7
1.8 118 442 3.0/3.3
1.85 124 464 3.0/3.3
2.0 137 523 3.0/3.3
2.5 187 715 3.9/4.2
3.3 267 1000 5.6
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 17
1. For reduced quiescent current, R2 = 221kΩ.
2. R2 is opened, R1 is shorted.
AAT1121
1.5MHz, 250mA Step-Down Converter
18 1121.2006.04.1.0
Table 5: Surface Mount Capacitors.
Value Voltage Temp. Case
Manufacturer Part Number (μF) Rating Co. Size
Murata GRM118R60J475KE19B 4.7 6.3 X5R 0603
Murata GRM188R60J106ME47D 10 6.3 X5R 0603
Ordering Information
Package Information
TDFN22-8
All dimensions in millimeters.
2.00
±
0.05
Detail "A"
Detail "B"Index Area
(D/2 x E/2)
7.5°
±
7.5°
0.05
±
0.05
2.00
±
0.05
0.45
±
0.05
0.16 MIN
0.35
±
0.10
Pin 1 Indicator
(optional)
0.23
±
0.05
Top View
Side View
Detail "A"
Detail "B"
Bottom View
0.23
±
0.05
0.075
±
0.075
0.1 REF
(optional)
0.85 MAX
Option A:
C0.30 (4x) max
Chamfered corner
Option B:
R0.30 (4x) max
Round corner
All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means
semiconductor products that are in compliance with current RoHS standards, including
the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more
information, please visit our website at http://www.analogictech.com/pbfree.
Output Voltage Package Marking1Part Number (Tape and Reel)2
0.6V TDFN22-8 RWXYY AAT1121IPS-0.6-T1
AAT1121
1.5MHz, 250mA Step-Down Converter
1121.2006.04.1.0 19
1. XYY = assembly and date code.
2. Sample stock is generally held on all part numbers listed in BOLD.
AAT1121
1.5MHz, 250mA Step-Down Converter
20 1121.2006.04.1.0
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Phone (408) 737-4600
Fax (408) 737-4611
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