LM4960 Piezoelectric Speaker Driver General Description Key Specifications The LM4960 utilizes a switching regulator to drive a dual audio power amplifier. It delivers 24VP-P mono-BTL to a ceramic speaker with less than 1.0% THD+N while operating on a 3.0V power supply. The LM4960's switching regulator is a current-mode boost converter operating at a fixed frequency of 1.6MHz. Boomer audio power amplifiers were designed specifically to provide high quality output power with a minimal amount of external components. The LM4960 does not require output coupling capacitors or bootstrap capacitors, and therefore is ideally suited for mobile phone and other low voltage applications where minimal power consumption is a primary requirement. The LM4960 features a low-power consumption externally controlled micropower shutdown mode. Additionally, the LM4960 features and internal thermal shutdown protection mechanism along with a short circuit protection. The LM4960 is unity-gain stable and can be configured by external gain-setting resistors. VOUT @ VDD = 3.0 THD+N 1% Power supply range Switching Frequency 24VP-P (typ) 3.0 to 7V 1.6MHz (typ) Features Low current shutdown mode "Click and pop" suppression circuitry Low Quiescent current Unity-gain stable audio amplifiers External gain configuration capability Thermal shutdown protection circuitry Wide input voltage range (3.0V - 7V) 1.6MHz switching frequency Applications Mobile phone PDA's Connection Diagram LM4960SQ 20076582 Top View Order Number LM4960SQ See NS Package Number Boomer(R) is a registered trademark of National Semiconductor Corporation. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. 200765 SNAS221B Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 Typical Application 20076581 FIGURE 1. Typical Audio Amplifier Application Circuit 2 Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 Absolute Maximum Ratings (Note 1, Note 2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. Supply Voltage (VDD) Supply Voltage (V1) (Pin 27 referred to GND) Storage Temperature Input Voltage Power Dissipation (Note 3) ESD Susceptibility (Note 4) ESD Susceptibility (Note 5) Junction Temperature Thermal Resistance 8.5V 18V -65C to +150C -0.3V to VDD + 0.3V Internally limited 2000V 200V 150C JA (LLP) C/W See AN-1187 'Leadless Leadframe Packaging (LLP).' Operating Ratings Temperature Range TMIN TA TMAX -40C TA +85C 3.0V VDD 7V Supply Voltage (VDD) 9.6V V1 16V Supply Voltage (V1) Electrical Characteristics VDD = 3.0V (Note 1, Note 2) The following specifications apply for VDD = 3V, AV = 10, RL = 800nF+20, V1 = 12V unless otherwise specified. Limits apply for TA = 25C. LM4960 Symbol Parameter Conditions Typical (Note 6) Limit (Note 7, Note 8) Units (Limits) IDD Quiescent Power Supply Current VIN = GND, No Load 85 150 mA (max) ISD Shutdown Current VSHUTDOWN = GND (Note 9) 30 100 A (max) VOS Output Offset Voltage 5 40 mV (max) VSDIH Shutdown Voltage Input High 2 V (max) VSDIL Shutdown Voltage Input Low 0.4 V (min) TWU Wake-up Time TSD Thermal Shutdown Temperature VO Output Voltage THD = 1% (max); f = 1kHz THD+N Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz OS Output Noise A-Weighted Filter, VIN = 0V PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz VFB Feedback Pin Reference Voltage CB = 0.22F 50 ms 170 150 190 C (min) C (max) 24 20 VP-P (min) 0.04 % 90 V 55 50 1.23 Electrical Characteristics VDD = 5.0V dB (min) V (max) (Note 1, Note 2) The following specifications apply for VDD = 5V, AV = 10, RL = 800nF+20 unless otherwise specified. Limits apply for TA = 25C. Symbol Parameter Conditions LM4960 Typical (Note 6) IDD Quiescent Power Supply Current VIN = GND, No Load 45 ISD Shutdown Current VSHUTDOWN = GND (Note 9) 55 VSDIH Shutdown Voltage Input High Copyright (c) 1999-2012, Texas Instruments Incorporated Limit (Note 7, Note 8) Units (Limits) mA (max) 100 A (max) 2 V (max) 3 LM4960 Symbol Parameter Conditions LM4960 Typical (Note 6) VSDIL Shutdown Voltage Input Low TWU Wake-up Time CB = 0.22F Limit (Note 7, Note 8) Units (Limits) 0.4 V (min) 170 150 190 C (min) C (max) 24 20 VP-P (min) 50 ms TSD Thermal Shutdown Temperature VO Output Voltage THD = 1% (max); f = 1kHz RL = Ceramic Speaker THD+N Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz 0.04 % OS Output Noise A-Weighted Filter, VIN = 0V 90 V PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 60 dB (min) VFB Feedback Pin Reference Voltage 1.23 V (max) Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified. Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good indication of device performance. Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, JA, and the ambient temperature, TA. The maximum allowable power dissipation is PDMAX = (TJMAX - TA) / JA or the given in Absolute Maximum Ratings, whichever is lower. For the LM4960 typical application (shown in Figure 1) with VDD = 12V, RL = 4 stereo operation the total power dissipation is 3.65W. JA = 35C/W. Note 4: Human body model, 100pF discharged through a 1.5k resistor. Note 5: Machine Model, 220pF-240pF discharged through all pins. Note 6: Typicals are measured at 25C and represent the parametric norm. Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level). Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to VDD for minimum shutdown current. 4 Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 Typical Performance Characteristics THD+N vs Frequency VDD = 3V, V1 = 9.6V, V0 = 3Vrms 20076514 THD+N vs Frequency VDD = 3V, V1 = 15V, V0 = 3Vrms 20076516 THD+N vs Frequency VDD = 5V, V1 =12V, V0 = 3Vrms 20076518 Copyright (c) 1999-2012, Texas Instruments Incorporated THD+N vs Frequency VDD = 3V, V1 = 12V, V0 = 3Vrms 20076515 THD+N vs Frequency VDD = 5V, V1 = 9.6V, V0 = 3Vrms 20076517 THD+N vs Frequency VDD = 5V, V1 =15V, V0 = 3Vrms 20076519 5 LM4960 THD+N vs Output Power VDD = 3V, V1 = 9.6V, f = 100Hz, 1kHz, 10kHz THD+N vs Output Power VDD = 3V, V1 = 12V, f = 100Hz, 1kHz, 10kHz 20076520 THD+N vs Output Power VDD = 3V, V1 = 15V, f = 100Hz, 1kHz, 10kHz 20076521 THD+N vs Output Power VDD = 5V, V1 = 9.6V, f = 100Hz, 1kHz, 10kHz 20076522 THD+N vs Output Power VDD = 5V, V1 = 12V, f = 100Hz, 1kHz, 10kHz THD+N vs Output Power VDD = 5V, V1 = 15V, f = 100Hz, 1kHz, 10kHz 20076524 6 20076523 20076525 Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 Power Dissipation vs Output Voltage VDD = 3V, from top to bottom: V1 = 15V, V1 = 12V, V1 = 9.6V Power Dissipation vs Output Voltage VDD = 5V, from top to bottom: V1 = 15V, V1 = 12V, V1 = 9.6V 20076509 20076510 Supply Current vs Supply Voltage from top to bottom: VDD = 15V, VDD = 12V, VDD = 9.6V Power Supply Rejection Ratio VDD = 3V 20076511 20076513 Power Supply Rejection Ratio VDD = 5V 20076512 Copyright (c) 1999-2012, Texas Instruments Incorporated 7 LM4960 Application Information BRIDGE CONFIGURATION EXPLANATION The Audio Amplifier portion of the LM4960 has two internal amplifiers allowing different amplifier configurations. The first amplifier's gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain, inverting configuration. The closedloop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the second amplifier's gain is fixed by the two internal 20k resistors. Figure 1 shows that the output of amplifier one serves as the input to amplifier two. This results in both amplifiers producing signals identical in magnitude, but out of phase by 180. Consequently, the differential gain for the Audio Amplifier is AVD = 2 *(Rf/Ri) By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as "bridged mode" is established. Bridged mode operation is different from the classic single-ended amplifier configuration where one side of the load is connected to ground. A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides differential drive to the load, thus doubling the output swing for a specified supply voltage. Four times the output power is possible as compared to a singleended amplifier under the same conditions. This increase in attainable output power assumes that the amplifier is not current limited or clipped. In order to choose an amplifier's closed-loop gain without causing excessive clipping, please refer to the Audio Power Amplifier Design section. The bridge configuration also creates a second advantage over single-ended amplifiers. Since the differential outputs, Vo1 and Vo2, are biased at half-supply, no net DC voltage exists across the load. This eliminates the need for an output coupling capacitor which is required in a single supply, single-ended amplifier configuration. Without an output coupling capacitor, the half-supply bias across the load would result in both increased internal IC power dissipation and also possible loudspeaker damage. AMPLIFIER POWER DISSIPATION Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or single-ended. A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation. Since the amplifier portion of the LM4960 has two operational amplifiers, the maximum internal power dissipation is 4 times that of a single-ended amplifier. The maximum power dissipation for a given BTL application can be derived from Equation 1. PDMAX(AMP) = 4(VDD)2 / (22ZL) (1) where ZL = Ro1 + Ro2 +1/2fc BOOST CONVERTER POWER DISSIPATION At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be determined by power dissipation within the LM2731 FET switch. The switch power dissipation from ON-time conduction is calculated by Equation 2. PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON) (2) where DC is the duty cycle. There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation. TOTAL POWER DISSIPATION The total power dissipation for the LM4960 can be calculated by adding Equation 1 and Equation 2 together to establish Equation 3: PDMAX(TOTAL) = [4*(VDD)2/22ZL] + [DC x IIND(AVE)2 xRDS(ON)] (3) The result from Equation 3 must not be greater than the power dissipation that results from Equation 4: PDMAX = (TJMAX - TA) / JA (4) For the LQA28A, JA = 59C/W. TJMAX = 125C for the LM4960. Depending on the ambient temperature, TA, of the system surroundings, Equation 4 can be used to find the maximum internal power dissipation supported by the IC packaging. If the result of Equation 3 is greater than that of Equation 4, then either the supply voltage must be increased, the load impedance increased or TA reduced. For the typical application of a 3V power supply, with V1 set to 12V and a 800nF + 20 load, the maximum ambient temperature possible without violating the maximum junction temperature is approximately 118C provided that device operation 8 Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 is around the maximum power dissipation point. Thus, for typical applications, power dissipation is not an issue. Power dissipation is a function of output power and thus, if typical operation is not around the maximum power dissipation point, the ambient temperature may be increased accordingly. Refer to the Typical Performance Characteristics curves for power dissipation information for lower output levels. EXPOSED-DAP PACKAGE PCB MOUNTING CONSIDERATIONS The LM4960's exposed-DAP (die attach paddle) package (LD) provides a low thermal resistance between the die and the PCB to which the part is mounted and soldered. The low thermal resistance allows rapid heat transfer from the die to the surrounding PCB copper traces, ground plane, and surrounding air. The LD package should have its DAP soldered to a copper pad on the PCB. The DAP's PCB copper pad may be connected to a large plane of continuous unbroken copper. This plane forms a thermal mass, heat sink, and radiation area. Further detailed and specific information concerning PCB layout, fabrication, and mounting an LD (LLP) package is found in National Semiconductor's Package Engineering Group under application note AN1187. SHUTDOWN FUNCTION In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to provide a quick, smooth transition into shutdown. Another solution is to use a single-pole, single-throw switch, and a pull-up resistor. One terminal of the switch is connected to GND. The other side is connected to the two shutdown pins and the terminal of the pull-up resistor. The remaining resistance terminal is connected to VDD. If the switch is open, then the external pull-up resistor connected to VDD will enable the LM4960. This scheme guarantees that the shutdown pins will not float thus preventing unwanted state changes. PROPER SELECTION OF EXTERNAL COMPONENTS Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC converters, is critical for optimizing device and system performance. Consideration to component values must be used to maximize overall system quality. The best capacitors for use with the switching converter portion of the LM4960 are multi-layer ceramic capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency, which makes them optimum for high frequency switching converters. When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor manufacturer's data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from Taiyo-Yuden, AVX, and Murata. POWER SUPPLY BYPASSING As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply rejection. The capacitor location on both V1 and VDD pins should be as close to the device as possible. SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is dictated by the choice of external components shown in Figure 1. The input coupling capacitor, Ci, forms a first order high pass filter which limits low frequency response. This value should be chosen based on needed frequency response for a few distinct reasons. High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value capacitor is needed to couple in low frequencies without severe attenuation. But ceramic speakers used in portable systems, whether internal or external, have little ability to reproduce signals below 100Hz to 150Hz. Thus, using a high value input capacitor may not increase actual system performance. In addition to system cost and size, click and pop performance is affected by the value of the input coupling capacitor, Ci. A high value input coupling capacitor requires more charge to reach its quiescent DC voltage (nominally 1/2 VDD). This charge comes from the output via the feedback and is apt to create pops upon device enable. Thus, by minimizing the capacitor value based on desired low frequency response, turn-on pops can be minimized. SELECTING BYPASS CAPACITOR FOR AUDIO AMPLIFIER Besides minimizing the input capacitor value, careful consideration should be paid to the bypass capacitor value. Bypass capacitor, CB, is the most critical component to minimize turn-on pops since it determines how fast the amplifer turns on. The slower the amplifier's outputs ramp to their quiescent DC voltage (nominally 1/2 VDD), the smaller the turn-on pop. Choosing CB equal to 1.0F along with a small value of Ci (in the range of 0.039F to 0.39F), should produce a virtually clickless and popless shutdown function. Although the device will function properly, (no oscillations or motorboating), with CB equal to 0.1F, the device will be much more susceptible to turn-on clicks and pops. Thus, a value of CB equal to 1.0F is recommended in all but the most cost sensitive designs. SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER The LM4960 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic speakers, a typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor (Cf2) will be needed as shown in Figure 2 to bandwidth limit the amplifier. This feedback capacitor creates a low pass filter that eliminates possible high frequency oscillations. Care should be taken when calculating the -3dB frequency because an incorrect combination of Rf and Cf2 will cause rolloff before the desired frequency Copyright (c) 1999-2012, Texas Instruments Incorporated 9 LM4960 SELECTING OUTPUT CAPACITOR (CO) FOR BOOST CONVERTER A single 4.7F to 10F ceramic capacitor will provide sufficient output capacitance for most applications. If larger amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500 kHz because of significant ringing and temperature rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability. In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce ringing, switching losses, and output voltage ripple. SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nominal value of 4.7F, but larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other circuitry. SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER The output voltage is set using the external resistors R1 and R2 (see Figure 1). A value of approximately 13.3k is recommended for R2 to establish a divider current of approximately 92A. R1 is calculated using the formula: R1 = R2 X (V2/1.23 - 1) (5) FEED-FORWARD COMPENSATION FOR BOOST CONVERTER Although the LM4960's internal Boost converter is internally compensated, the external feed-forward capacitor Cf is required for stability (see Figure 1). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz should be approximately 6kHz. Cf1 can be calculated using the formula: Cf1 = 1 / (2 X R1 X fz) (6) SELECTING DIODES The external diode used in Figure 1 should be a Schottky diode. A 20V diode such as the MBR0520 is recommended. The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used. DUTY CYCLE The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined as: Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW This applies for continuous mode operation. INDUCTANCE VALUE The first question we are usually asked is: "How small can I make the inductor." (because they are the largest sized component and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be delivered because the energy stored during each switching cycle is: E = L/2 X (lp)2 10 Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 Where "lp" is the peak inductor current. An important point to observe is that the LM4960 will limit its switch current based on peak current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load. Conversely, using too little inductance may limit the amount of load current which can be drawn from the output. Best performance is usually obtained when the converter is operated in "continuous" mode at the load current range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays "continuous" over a wider load current range. To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10H inductor) will be analyzed. We will assume: VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V Since the frequency is 1.6MHz (nominal), the period is approximately 0.625s. The duty cycle will be 62.5%, which means the ONtime of the switch is 0.390s. It should be noted that when the switch is ON, the voltage across the inductor is approximately 4.5V. Using the equation: V = L (di/dt) We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/s during the ON-time. Using these facts, we can then show what the inductor current will look like during operation: 20076583 FIGURE 2. 10H Inductor Current 5V - 12V Boost (LM4960) During the 0.390s ON-time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFF-time. This is defined as the inductor "ripple current". It can also be seen that if the load current drops to about 33mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and continuous operation will be maintained at the typical load current values. MAXIMUM SWITCH CURRENT The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in a graph in the typical performance characterization section which shows typical values of switch current as a function of effective (actual) duty cycle. CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP) As shown in Figure 2 which depicts inductor current, the load current is related to the average inductor current by the relation: ILOAD = IIND(AVG) x (1 - DC) (7) Where "DC" is the duty cycle of the application. The switch current can be found by: ISW = IIND(AVG) + 1/2 (IRIPPLE) (8) Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency: IRIPPLE = DC x (VIN-VSW) / (f x L) Copyright (c) 1999-2012, Texas Instruments Incorporated (9) 11 LM4960 Combining all terms, we can develop an expression which allows the maximum available load current to be calculated: ILOAD(max) = (1-DC)x(ISW(max)-DC(VIN-VSW))/fL (10) The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses of the FET and diode. DESIGN PARAMETERS VSW AND ISW The value of the FET "ON" voltage (referred to as VSW in equations 7 thru 10) is dependent on load current. A good approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current. FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V. The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical Performance Characteristics curves. INDUCTOR SUPPLIERS Recommended suppliers of inductors for the LM4960 include, but are not limited to Taiyo-Yuden, Sumida, Coilcraft, Panasonic, TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation at peak currents. A suitable core type must be used to minimize core (switching) losses, and wire power losses must be considered when selecting the current rating. PCB LAYOUT GUIDELINES High frequency boost converters require very careful layout of components in order to get stable operation and low noise. All components must be as close as possible to the LM4802 device. It is recommended that a 4-layer PCB be used so that internal ground planes are available. Some additional guidelines to be observed: 1. Keep the path between L1, D1, and Co extremely short. Parasitic trace inductance in series with D1 and Co will increase noise and ringing. 2. The feedback components R1, R2 and Cf 1 must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace. 3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative sides of capacitors Cs1 and Co. GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION This section provides practical guidelines for mixed signal PCB layout that involves various digital/analog power and ground traces. Designers should note that these are only "rule-of-thumb" recommendations and the actual results will depend heavily on the final layout. Power and Ground Circuits For 2 layer mixed signal design, it is important to isolate the digital power and ground trace paths from the analog power and ground trace paths. Star trace routing techniques (bringing individual traces back to a central point rather than daisy chaining traces together in a serial manner) can have a major impact on low level signal performance. Star trace routing refers to using individual traces to feed power and ground to each circuit or even device. This technique will take require a greater amount of design time but will not increase the final price of the board. The only extra parts required may be some jumpers. Single-Point Power / Ground Connection The analog power traces should be connected to the digital traces through a single point (link). A "Pi-filter" can be helpful in minimizing high frequency noise coupling between the analog and digital sections. It is further recommended to place digital and analog power traces over the corresponding digital and analog ground traces to minimize noise coupling. Placement of Digital and Analog Components All digital components and high-speed digital signals traces should be located as far away as possible from analog components and circuit traces. Avoiding Typical Design / Layout Problems Avoid ground loops or running digital and analog traces parallel to each other (side-by-side) on the same PCB layer. When traces must cross over each other do it at 90 degrees. Running digital and analog traces at 90 degrees to each other from the top to the bottom side as much as possible will minimize capacitive noise coupling and crosstalk. 12 Copyright (c) 1999-2012, Texas Instruments Incorporated LM4960 Physical Dimensions inches (millimeters) unless otherwise noted LLP, Plastic, Quad Order Number LM4960SQ NS Package Number SQA28A Copyright (c) 1999-2012, Texas Instruments Incorporated 13 Notes Copyright (c) 1999-2012, Texas Instruments Incorporated IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46C and to discontinue any product or service per JESD48B. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. 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