LT1578CS8#PBF - Linear Technology

LT1578/LT1578-2.5

1.5A, 200kHz Step-Down

Switching Regulator

3.3V Buck Converter

Efficiency vs Load Current

■

1.5A Switch Current

■

High Efficiency—Low Loss 0.2Ω Switch

■

Constant 200kHz Switching Frequency

■

4V to 15V Input VoltageRange

■

Minimum Output: 1.21V

■

Low Supply Current: 1.9mA

■

Low Shutdown Current: 20µA

■

Easily Synchronizable Up to 400kHz

■

Cycle-by-Cycle Current Limit

■

Reduced EMI Generation

■

Low Thermal Resistance SO-8 Package

■

Uses Small Low Value Inductors

The LT

1578 is a 200kHz monolithic buck mode switching

regulator. A 1.5A switch is included on the die along with

all the necessary oscillator, control and logic circuitry. The

topology is current mode for fast transient response and

good loop stability. The LT1578 is a modified version of the

LT1507 that has been optimized for noise sensitive appli-

cations. It will operate over a 4V to 15V input range.

In addition, the reference voltage has been lowered to al-

low the device to produce output voltages down to 1.2V.

Quiescent current has been reduced by a factor of two.

Switch on resistance has been reduced by 30%. Switch tran-

sition times have been slowed to reduce EMI generation.

The oscillator frequency has been reduced to 200kHz to

maintain high efficiency over a wide output current range.

The pinout has been changed to improve PC layout by al-

lowing the high current, high frequency switching circuitry

to be easily isolated from low current, noise sensitive con-

trol circuitry. The new SO-8 package includes a fused

ground lead that significantly reduces the thermal resistance

of the device to extend the ambient operating temperature

range. Standard surface mount external parts can be used

including the inductor and capacitors.

■

Portable Computers

■

Battery-Powered Systems

■

Battery Chargers

■

Distributed Power Systems

, LTC and LT are registered trademarks of Linear Technology Corporation.

LOAD CURRENT (A)

EFFICIENCY (%)

50 0.25 0.50 0.75 1.00

1578 TA02

1.25 1.50

OUT

= 3.3V

= 5V

L = 25µH

DESCRIPTIO

FEATURES

APPLICATIO S

BOOST

LT1578

SHDN

OUTPUT**

3.3V, 1.25A

* RIPPLE CURRENT RATING ≥ I

OUT

** INCREASE L1 TO 30µH FOR LOAD

CURRENTS ABOVE 0.6A AND TO

60µH ABOVE 1A

SEE APPLICATIONS INFORMATION

INPUT

5V TO 15V

1578 TA01

0.33µF

100pF D1

1N5818

100µF, 10V

SOLID

TANTALUM

C3*

10µF TO

50µF

OPEN = ON

1N914

L1**

15µH

GND V

4.99k

8.66k

TYPICAL APPLICATION

LT1578/LT1578-2.5

PARAMETER CONDITIONS MIN TYP MAX UNITS

Feedback Voltage 1.195 1.21 1.225 V

All Conditions ●1.18 1.24 V

Sense Voltage (Fixed 2.5) 2.46 2.5 2.54 V

All Conditions ●2.44 2.56 V

Sense Pin Resistance 5.7 9.5 13.7 kΩ

Reference Voltage Line Regulation 4.3V ≤ V

≤ 15V ●0.01 0.03 %/V

Feedback Input Bias Current ●0.5 2 µA

Error Amplifier Voltage Gain (Notes 2, 10) 200 400

Error Amplifier Transconductance (Note 10) ∆I (V

) = ±10µA 800 1050 1300 µMho

●400 1700 µMho

Pin to Switch Current Transconductance 1.5 A/V

Error Amplifier Source Current V

= 1.1V ●40 110 190 µA

Error Amplifier Sink Current V

= 1.4V ●50 130 200 µA

Pin Switching Threshold Duty Cycle = 0 0.8 V

Pin High Clamp 2.1 V

Switch Current Limit V

Open, V

= 1.1V, DC ≤ 50% ●1.5 2 3.5 A

Slope Compensation (Note 8) DC = 80% 0.3 A

Switch On Resistance (Note 7) I

= 1.5A 0.2 0.35 Ω

●0.45 Ω

Maximum Switch Duty Cycle V

= 1.1V 90 94 %

●86 94 %

Minimum Switch Duty Cycle (Note 9) 8%

Switch Frequency V

Set to Give 50% Duty Cycle 180 200 220 kHz

●160 240 kHz

Switch Frequency Line Regulation 4.3V

≤ V

≤ 15V ●0 0.15 %/V

Frequency Shifting Threshold on FB Pin ∆f = 10kHz ●0.4 0.74 1.0 V

Minimum Input Voltage (Note 3) ●4.0 4.3 V

Minimum Boost Voltage (Note 4) I

≤ 1.5A ●2.3 3.0 V

ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

(Note 1)

Input Voltage .......................................................... 16V

BOOST Pin Above Input Voltage ............................. 10V

SHDN Pin Voltage..................................................... 7V

SENSE Pin Voltage .................................................... 4V

FB Pin Voltage (Adjustable Part)............................ 3.5V

FB Pin Current (Adjustable Part)............................ 1mA

SYNC Pin Voltage ..................................................... 7V

Operating Junction Temperature Range

LT1578C............................................... 0°C to 125° C

LT1578I ........................................... –40°C to 125°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

The ● denotes specifications which apply over the full operating tempera-

ture range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.

ELECTRICAL CHARACTERISTICS

Consult factory for Military grade parts.

ORDER PART

NUMBER

LT1578CS8

LT1578IS8

LT1578CS8-2.5

LT1578IS8-2.5

S8 PART MARKING

1578

1578I

TOP VIEW

S8 PACKAGE

8-LEAD PLASTIC SO

BOOST

GND

SHDN

FB/SENSE

SYNC

=80°C/W WITH FUSED GROUND PIN

CONNECTED TO GROUND PLANE OR

LARGE LANDS 157825

578I25

LT1578/LT1578-2.5

PARAMETER CONDITIONS MIN TYP MAX UNITS

Boost Current (Note 5) I

= 0.5A ●918 mA

= 1.5A ●27 50 mA

Supply Current (Note 6) ●1.9 2.7 mA

Shutdown Supply Current V

SHDN

= 0V, V

≤ 15V, V

= 0V, V

Open 20 50 µA

●75 µA

Lockout Threshold V

Open ●2.34 2.42 2.50 V

Shutdown Thresholds V

Open Device Shutting Down ●0.13 0.37 0.60 V

Device Starting Up ●0.25 0.45 0.7 V

Synchronization Threshold 1.5 2.2 V

Synchronizing Range 250 400 kHz

SYNC Pin Input Resistance 40 kΩ

Note 1: Absolute Maximum Ratings are those values beyond which the life

of a device may be impaired.

Note 2: Gain is measured with a V

swing equal to 200mV above the

switching threshold level to 200mV below the upper clamp level.

Note 3: Minimum input voltage is not measured directly, but is guaranteed

by other tests. It is defined as the voltage where internal bias lines are still

regulated so that the reference voltage and oscillator frequency remain

constant. Actual minimum input voltage to maintain a regulated output will

depend on output voltage and load current. See Applications Information.

Note 4: This is the minimum voltage across the boost capacitor needed to

guarantee full saturation of the internal power switch.

Note 5: Boost current is the current flowing into the boost pin with the pin

held 5V above input voltage. It flows only during switch on time.

Note 6: Input supply current is the bias current drawn by the input pin

with switching disabled.

Note 7: Switch on resistance is calculated by dividing V

to V

voltage

by the forced current (1.5A). See Typical Performance Characteristics for

the graph of switch voltage at other currents.

Note 8: Slope compensation is the current subtracted from the switch

current limit at 80% duty cycle. See Maximum Output Load Current in the

Applications Information section for further details.

Note 9: Minimum on-time is 400ns typical. For a 200kHz operating

frequency this means the minimum duty cycle is 8%. In frequency

foldback mode, the effective duty cycle will be less than 8%.

Note 10: Transconductance and voltage gain refer to the internal amplifier

exclusive of the voltage divider. To calculate gain and transconductance

referred to the sense pin on the fixed voltage parts, divide values shown by

the ratio 2.5/1.21.

TYPICAL PERFORMANCE CHARACTERISTICS

Switch Voltage Drop

JUNCTION TEMPERATURE (°C)

–50

1.23

1.22

1.21

1.20

1.19 100

1576 G03

–25 0 25 50 75 125

FEEDBACK VOLTAGE (V)

SWITCH CURRENT (A)

SWITCH VOLTAGE (V)

0.5

0.4

0.3

0.2

0.1

00.25 0.50 0.75 1.00

1576 G01

1.25 1.50

125°C

–20°C

25°C

Feedback Pin VoltageSwitch Peak Current Limit

DUTY CYCLE (%)

SWITCH PEAK CURRENT (A)

2.5

2.0

1.5

1.0

0.5

080

1576 G02

20 40 60 100

TYPICAL

MINIMUM

The ● denotes specifications which apply over the full operating tempera-

ture range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.

ELECTRICAL CHARACTERISTICS

LT1578/LT1578-2.5

TYPICAL PERFORMANCE CHARACTERISTICS

JUNCTION TEMPERATURE (°C)

–50

0100

1576 G04

–25 0 25 50 75 125

SHDN PIN CURRENT (µA)

AT 2.44V STANDBY THRESHOLD

(CURRENT FLOWS OUT OF PIN)

Shutdown Pin Bias Current

(VSHDN = Lockout Threshold)

JUNCTION TEMPERATURE (°C)

–50

180

160

140

120

100

0100

1576 G05

–25 0 25 50 75 125

SHDN PIN CURRENT (µA)

CURRENT REQUIRED TO FORCE

SHUTDOWN (FLOWS OUT OF PIN).

AFTER SHUTDOWN, CURRENT

DROPS TO A FEW µA

JUNCTION TEMPERATURE (°C)

–50

SHUTDOWN PIN VOLTAGE (V)

100

1576 G06

050

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0–25 25 75 125

START-UP

SHUTDOWN

Shutdown Supply Current

INPUT VOLTAGE (V)

INPUT SUPPLY CURRENT (µA)

1576 G08

510 15

SHDN

= 0V

FREQUENCY (Hz)

GAIN (µMho)

PHASE (DEG)

2000

1500

1000

500

–500

200

150

100

–50

10 1k 10k 1M

1576 G09

100 100k

GAIN

PHASE

ERROR AMPLIFIER EQUIVALENT CIRCUIT

OUT

570k

OUT

2.4pF

LOAD

= 50Ω

1 × 10

–3

)(

Error Amplifier Transconductance

JUNCTION TEMPERATURE (°C)

–50

SHUTDOWN PIN VOLTAGE (V)

2.46

2.45

2.44

2.43

2.42

2.41

2.40 25 75

1576 G07

–25 0 50 100 125

STANDBY

FEEDBACK VOLTAGE (V)

SWITCHING FREQUENCY (kHz)

OR CURRENT (µA)

2.0

1576 G12

0.5 1.0 1.5

250

200

150

100

0FEEDBACK PIN CURRENT

SWITCHING FREQUENCY

Shutdown Supply Current

JUNCTION TEMPERATURE (°C)

–50

TRANSCONDUCTANCE (µMho)

100

1576 G11

050

1600

1400

1200

1000

800

600

400

200

0–25 25 75 125

Error Amplifier Transconductance Frequency Foldback

Shutdown Thresholds

Standby Thresholds

SHUTDOWN VOLTAGE (V)

INPUT SUPPLY CURRENT (µA)

1576 G010

0.1 0.2 0.3 0.4

= 10V

Shutdown Pin Bias Current

(VSHDN = Shutdown Threshold)

LT1578/LT1578-2.5

TYPICAL PERFORMANCE CHARACTERISTICS

Kool Mµ is a registered trademark of Magnetics, Inc.

Metglas is a registered trademark of AlliedSignal, Inc.

JUNCTION TEMPERATURE (°C)

–50

240

220

200

180

160 100

1576 G13

–25 0 25 50 75 125

FREQUENCY (kHz)

Switching Frequency

INPUT VOLTAGE (V)

OUTPUT CURRENT (A)

0.6

0.8

1.0

1578 G15

0.4

0.2

0912

1.2

L = 60µH

1.4

1.6

L = 30µH

L = 15µH

Maximum Output Current

at VOUT = 5V

LOAD CURRENT (mA)

INPUT VOLTAGE (V)

4.50

4.25

4.00

3.75

3.50 10 100 1000

1576 G14

Minimum Input Voltage to Start

with 3.3V Output

INPUT VOLTAGE (V)

OUTPUT CURRENT (A)

0.6

0.8

1.0

1578 G16

0.4

0.2

06 8 10 12

1.2

1.4

1.6

L = 30µH

L = 15µH

L = 60µH

Maximum Output Current

at VOUT = 3.3V

INPUT VOLTAGE (V)

OUTPUT CURRENT (A)

0.6

0.8

1.0

1578 G17

0.4

0.2

06 8 10 12

1.2

1.4

1.6

L = 30µH

L = 15µH

L = 60µH

Maximum Output Current

at VOUT = 2.5V

BOOST Pin Current VC Pin Shutdown Threshold

SWITCH CURRENT (A)

BOOST PIN CURRENT (mA)

00.25 0.50 0.75 1.00

1576 G20

1.25 1.50

JUNCTION TEMPERATURE (°C)

–50

1.0

0.8

0.6

0.4

0.2

0100

1576 G21

–25 0 25 50 75 125

THRESHOLD VOLTAGE (V)

FEEDBACK PIN VOLTAGE (V)

SWITCH CURRENT LIMIT (A)

0.5

1.0

1.5

2.0

3.0

0.2 0.4 0.6 0.8

1578 G19

1.0 1.2

2.5

Switch Current Limit Foldback

LT1578/LT1578-2.5

PIN FUNCTIONS

UUU

(Pin 1): The switch pin is the emitter of the on-chip

power NPN switch. This pin is driven up to the input pin

voltage during switch on time. Inductor current drives the

switch pin negative during switch off time. Negative volt-

age is clamped with the external catch diode. Maximum

negative switch voltage allowed is –0.8V.

(Pin 2): This is the collector of the on-chip power NPN

switch. This pin powers the internal circuitry and internal

regulator. At NPN switch on and off, high dI/dt edges occur

through this pin. Keep the external bypass and catch diode

close to this pin. Trace inductance in this path will create

a voltage spike at switch off, adding to the V

voltage

across the internal NPN.

BOOST (Pin 3): The BOOST pin is used to provide a drive

voltage, higher than the input voltage, to the internal

bipolar NPN power switch. Without this added voltage, the

typical switch voltage loss would be about 1.5V. The

additional boost voltage allows the switch to saturate with

its voltage drop approximating that of a 0.2Ω FET struc-

ture. Efficiency improves from 75% for conventional bipo-

lar designs to >88% for the LT1578.

GND (Pin 4): The GND pin connection needs consideration

for two reasons. First, it acts as the reference for the

regulated output, so load regulation will suffer if the

“ground” end of the load is not at the same voltage as the

GND pin of the IC. This condition will occur when load

current or other currents flow through metal paths be-

tween the GND pin and the load ground point. Keep the

ground path short between the GND pin and the load and

use a ground plane when possible. The second consider-

ation is EMI caused by GND pin current spikes. Internal

capacitance between the V

pin and the GND pin creates

very narrow (<10ns) current spikes in the GND pin. If the

GND pin is connected to system ground with a long metal

trace, this trace may radiate EMI. Keep the path between

the input bypass and the GND pin short. The GND pin of the

SO-8 package is directly attached to the internal tab. This

pin should be attached to a large copper area to improve

thermal resistance.

(Pin 5): The V

pin is the output of the error amplifier

and the input to the peak switch current comparator. It is

normally used for frequency compensation, but can do

double duty as a current clamp or control loop override.

This pin sits at about 1V for very light loads and 2V at

maximum load. It can be driven to ground to shut off the

regulator, but if driven high, current must be limited to

4mA.

FB/SENSE (Pin 6): The feedback pin is used to set output

voltage using an external voltage divider that generates

1.21V at the pin with the desired output voltage. The fixed

voltage (2.5V) parts have the divider included on the chip

and the FB pin is used as a sense pin, connected directly

to the 2.5V output. Three additional functions are per-

formed by the FB pin. When the pin voltage drops below

0.7V, the switch current limit and the switching frequency

are reduced and the external sync function is disabled. See

Feedback Pin Function section in Applications Information

for details.

SHDN (Pin 7): The shutdown pin is used to turn off the

regulator and to reduce input drain current to a few

microamperes. Actually, this pin has two separate thresh-

olds, one at 2.42V to disable switching, and a second at

0.4V to force complete micropower shutdown. The 2.42V

threshold functions as an accurate undervoltage lockout

(UVLO). This can be used to prevent the regulator from

operating until the input voltage has reached a predeter-

mined level.

SYNC (Pin 8): The SYNC pin is used to synchronize the

internal oscillator to an external signal. It is directly logic

compatible and can be driven with any signal between

10% and 90% duty cycle. The synchronizing range is

equal to

initial

operating frequency, up to 400kHz. When

not used, this pin should be grounded. See Synchronizing

section in Applications Information for details.

LT1578/LT1578-2.5

BLOCK DIAGRAM

and output capacitor, then an abrupt 180° shift will occur.

The current fed system will have 90° phase shift at a much

lower frequency, but will not have the additional 90° shift

until well beyond the LC resonant frequency. This makes

it much easier to frequency compensate the feedback loop

and also gives much quicker transient response.

High switch efficiency is attained by using the BOOST pin

to provide a voltage to the switch driver which is higher

than the input voltage, allowing the switch to saturate. This

boosted voltage is generated with an external capacitor

and diode. Two comparators are connected to the shut-

down pin. One has a 2.42V threshold for undervoltage

lockout and the second has a 0.4V threshold for complete

shutdown.

The LT1578 is a constant frequency, current mode buck

converter. This means that there is an internal clock and

two feedback loops that control the duty cycle of the power

switch. In addition to the normal error amplifier, there is a

current sense amplifier that monitors switch current on a

cycle-by-cycle basis. A switch cycle starts with an oscilla-

tor pulse which sets the R

flip-flop to turn the switch on.

When switch current reaches a level set by the inverting

input of the comparator, the flip-flop is reset and the

switch turns off. Output voltage control is obtained by

using the output of the error amplifier to set the switch

current trip point. This technique means that the error

amplifier commands current to be delivered to the output

rather than voltage. A voltage fed system will have low

phase shift up to the resonant frequency of the inductor

–

–+

INPUT

2.9V BIAS

REGULATOR

200kHz

OSCILLATOR

FREQUENCY

SHIFT CIRCUIT

LOCKOUT

COMPARATOR

GND

1578 BD

SLOPE COMP

0.025Ω

INTERNAL

CURRENT SENSE

AMPLIFIER DC

VOLTAGE GAIN = 35

SYNC

SHDN

SHUTDOWN

COMPARATOR

CURRENT

COMPARATOR

ERROR

AMPLIFIER

= 1000µMho

FOLDBACK

CURRENT

LIMIT

CLAMP

BOOST

FLIP-FLOP DRIVER

CIRCUITRY

0.8V

POWER

SWITCH

1.21V2.42V

–

0.4V

3.5µA

Figure 1. Block Diagram

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

Figure 2. Frequency and Current Limit Foldback

–

1.21V

GND

TO SYNC CIRCUIT

1578 F02

TO FREQUENCY

SHIFTING

1k R4

OUTPUT

ERROR

AMPLIFIER

1.4V Q1

LT1578

FEEDBACK PIN FUNCTIONS

The feedback (FB) pin on the LT1578 is used to set output

voltage and provide several overload protection features.

The first part of this section deals with selecting resistors

to set output voltage and the remaining part talks about

foldback frequency and current limiting created by the FB

pin. Please read both parts before committing to a final

design. The fixed 2.5V LT1578-2.5 has internal divider

resistors and the FB pin, renamed SENSE, is connected

directly to the 2.5V output.

The suggested value for the output divider resistor (see

Figure 2) from FB to ground (R2) is 5k or less, and a

formula for R1 is shown below. The output voltage error

caused by ignoring the input bias current on the FB pin is

less than 0.25% with R2 = 5k. Please read the following

if divider resistors are increased above the suggested

values.

RRV

OUT

12121

121

=−

()

More Than Just Voltage Feedback

The feedback pin is used for more than just output voltage

sensing. It also reduces switching frequency and current

limit when output voltage is very low (see the Frequency

Foldback graph in Typical Performance Characteristics).

This is done to control power dissipation in both the IC and

the external diode and inductor during short-circuit con-

ditions. A shorted output requires the switching regulator

to operate at very low duty cycles, and the average current

through the diode and inductor is equal to the short-circuit

current limit of the switch (typically 2A for the LT1578,

folding back to less than 0.77A). Minimum switch on time

limitations would prevent the switcher from attaining a

sufficiently low duty cycle if switching frequency were

maintained at 200kHz, so frequency is reduced by about

5:1 when the feedback pin voltage drops below 0.7V (see

Frequency Foldback graph). This does not affect operation

with normal load conditions; one simply sees a gear shift

in switching frequency during start-up as the output

voltage rises.

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

In addition to lower switching frequency, the LT1578 also

operates at lower switch current limit when the feedback

pin voltage drops below 0.7V. Q2 in Figure 2 performs this

function by clamping the V

pin to a voltage less than its

normal 2.1V upper clamp level. This

foldback current limit

greatly reduces power dissipation in the IC, diode and

inductor during short-circuit conditions. External synchro-

nization is also disabled to prevent interference with

foldback operation. Again, it is nearly transparent to the

user under normal load conditions. The only loads that may

be affected are current sources, such as lamps and mo-

tors, that maintain high load current with output voltage

less than 50% of final value. In these rare situations the

feedback pin can be clamped above 0.7V to defeat foldback

current limit.

Caution:

clamping the feedback pin means

that frequency shifting will also be defeated, so a combina-

tion of high input voltage and dead shorted output may

cause the LT1578 to lose control of current limit.

The internal circuitry which forces reduced switching

frequency also causes current to flow out of the feedback

pin when output voltage is low. The equivalent circuitry is

shown in Figure 2. Q1 is completely off during normal

operation. If the FB pin falls below 0.7V, Q1 begins to

conduct current and reduces frequency at the rate of

approximately 1kHz/µA. To ensure adequate frequency

foldback (under worst-case short-circuit conditions), the

external divider Thevinin resistance must be low enough

to pull 35µA out of the FB pin with 0.5V on the pin (R

DIV

≤

14.3k).

The net result is that reductions in frequency and

current limit are affected by output voltage divider imped-

ance. Although divider impedance is not critical, caution

should be used if resistors are increased beyond the

suggested values and short-circuit conditions will occur

with high input voltage

. High frequency pickup will

increase and the protection accorded by frequency and

current foldback will decrease.

MAXIMUM OUTPUT LOAD CURRENT

Maximum load current for a buck converter is limited by

the maximum switch current rating (I

) of the LT1578.

This current rating is 1.5A up to 50% duty cycle (DC),

decreasing to 1.3A at 80% duty cycle. This is shown

graphically in Typical Performance Characteristics and as

shown in the formula below:

= 1.5A for DC ≤ 50%

= 1.67 – 0.18 (DC) – 0.32(DC)

for 50% < DC < 90%

DC = Duty cycle = V

OUT

Example: with V

OUT

= 5V, V

= 8V; DC = 5/8 = 0.625, and;

SW(MAX)

= 1.67 – 0.18 (0.625) – 0.32(0.625)

= 1.43A

Current rating decreases with duty cycle because the

LT1578 has internal slope compensation to prevent cur-

rent mode subharmonic switching. For more details, read

Application Note 19. The LT1578 is a little unusual in this

regard because it has nonlinear slope compensation which

gives better compensation with less reduction in current

limit.

Maximum load current would be equal to maximum

switch current

for an infinitely large inductor

, but with

finite inductor size, maximum load current is reduced by

one-half peak-to-peak inductor current. The following

formula assumes continuous mode operation, implying

that the term on the right is less than one-half of I

OUT(MAX)

Continuous Mode

For the conditions above and L = 15µH,

OUT MAX

()

−

=−

()

−

()











()

=−=

143 58 5

2 15 10 200 10 8

143 031 112

.••

...

At V

= 15V, duty cycle is 33%, so I

is just equal to a fixed

1.5A, and I

OUT(MAX)

is equal to:

15 515 5

2 15 10 200 10 15

15 056 094

.••

.. .

−

()

−

()











()

=− =

−

P−

()

−

()

()()( )

VVV

LfV

OUT IN OUT

LT1578/LT1578-2.5

Note that there is less load current available at the higher

input voltage because inductor ripple current increases.

This is not always the case. Certain combinations of

inductor value and input voltage range may yield lower

available load current at the lowest input voltage due to

reduced peak switch current at high duty cycles. If load

current is close to the maximum available, please check

maximum available current at both input voltage

extremes. To calculate actual peak switch current with a

given set of conditions, use:

VVV

LfV

SW PEAK OUT OUT IN OUT

(

)

=+ −

()

()()( )

For lighter loads where discontinuous operation can be

used, maximum load current is equal to:

OUT(MAX)

Discontinuous mode

Example: with L = 5µH, V

OUT

= 5V, and V

IN(MAX

) = 15V,

OUT MAX

()

−

()











()

−

()

1 5 200 10 5 10 15

2 5 15 5 034

236

.•• .

The main reason for using such a tiny inductor is that it is

physically very small, but keep in mind that peak-to-peak

inductor current will be very high. This will increase output

ripple voltage. If the output capacitor has to be made larger

to reduce ripple voltage, the overall circuit could actually

wind up larger.

CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR

For most applications the output inductor will fall in the

range of 15µH to 60µH. Lower values are chosen to reduce

APPLICATIONS INFORMATION

WUUU

physical size of the inductor. Higher values allow more

output current because they reduce peak current seen by

the LT1578 switch, which has a 1.5A limit. Higher values

also reduce output ripple voltage, and reduce core loss.

Graphs in the Typical Performance Characteristics section

show maximum output load current versus inductor size

and input voltage.

When choosing an inductor you might have to consider

maximum load current, core and copper losses, allowable

component height, output voltage ripple, EMI, fault cur-

rent in the inductor, saturation, and of course, cost. The

following procedure is suggested as a way of handling

these somewhat complicated and conflicting requirements.

1. Choose a value in microhenries from the graphs of

maximum load current and core loss. Choosing a small

inductor may result in discontinuous mode operation

at lighter loads, but the LT1578 is designed to work

well in either mode. Keep in mind that lower core loss

means higher cost, at least for closed core geometries

like toroids.

Assume that the average inductor current is equal to

load current and decide whether or not the inductor

must withstand continuous fault conditions. If maxi-

mum load current is 0.5A, for instance, a 0.5A inductor

may not survive a continuous 1.5A overload condition.

Dead shorts will actually be more gentle on the induc-

tor because the LT1578 has foldback current limiting.

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current

can be significantly higher than output current, espe-

cially with smaller inductors and lighter loads, so don’t

omit this step. Powdered iron cores are forgiving

because they saturate softly, whereas ferrite cores

saturate abruptly. Other core materials fall somewhere

in between. The following formula assumes continu-

ous mode of operation, but it errs only slightly on the

high side for discontinuous mode, so it can be used for

all conditions.

IfLV

VVV

OUT IN OUT

()()()( )

()

−

()

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

VVV

fLV

PEAK OUT OUT IN OUT

=+ −

()

()()( )

= Maximum input voltage

f = Switching frequency, 200kHz

3. Decide if the design can tolerate an “open” core geom-

etry like a rod or barrel, with high magnetic field

radiation, or whether it needs a closed core like a toroid

to prevent EMI problems. One would not want an open

core next to a magnetic storage media, for instance!

This is a tough decision because the rods or barrels are

temptingly cheap and small and there are no helpful

guidelines to calculate when the magnetic field radia-

tion will be a problem.

4. Start shopping for an inductor (see representative

surface mount units in Table 1) which meets the

requirements of core shape, peak current (to avoid

saturation), average current (to limit heating), and fault

current (if the inductor gets too hot, wire insulation will

melt and cause turn-to-turn shorts). Keep in mind that

all good things like high efficiency, low profile, and high

temperature operation will increase cost, sometimes

dramatically. Get a quote on the cheapest unit first to

calibrate yourself on price, then ask for what you really

want.

5. After making an initial choice, consider the secondary

things like output voltage ripple, second sourcing, etc.

Use the experts in the Linear Technology’s applica-

tions department if you feel uncertain about the final

choice. They have experience with a wide range of

inductor types and can tell you about the latest devel-

opments in low profile, surface mounting, etc.

Table 1

SERIES CORE

VENDOR/ VALUE DC CORE RESIS- MATER- HEIGHT

PART NO. (

H) (Amps) TYPE TANCE(

Ω

) IAL (mm)

Coiltronics

CTX15-2 15 1.7 Tor 0.059 KMµ6.0

CTX33-2 33 1.4 Tor 0.106 KMµ6.0

CTX68-4 68 1.2 Tor 0.158 KMµ6.4

CTX15-1P 15 1.4 Tor 0.087 52 4.2

CTX33-2P 33 1.3 Tor 0.126 52 6.0

CTX68-4P 68 1.1 Tor 0.238 52 6.4

Sumida

CDRH74-150 15 1.47 SC 0.081 Fer 4.5

CDH115-330 33 1.68 SC 0.082 Fer 5.2

CDRH125-680 68 1.5 SC 0.12 Fer 6

CDH74-330 33 1.45 SC 0.17 Fer 5.2

Coilcraft

DO3308P-153 15 2 SC 0.12 Fer 3

DO3316P-333 33 2 SC 0.1 Fer 5.21

DO3316P-683 68 1.4 SC 0.18 Fer 5.21

Pulse

PE-53602 35 1.4 Tor 0.166 Fer 9.1

PE-53604 73 1.3 Tor 0.290 Fer 9.1

PE-53632 22 2.7 Tor 0.063 Fer 9.1

PE-53633 40 2.7 Tor 0.085 Fer 10

Gowanda

SMP3316-152K 15 3.5 SC 0.041 Fer 6

SMP3316-332K 33 2.3 SC 0.092 Fer 6

SMP3316-682K 68 1.7 SC 0.178 Fer 6

Tor = Toroid

SC = Semi-closed geometry

Fer = Ferrite core material

52 = Type 52 powdered iron core material

KMµ = Kool Mµ

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

Output Capacitor Ripple Current (RMS):

IVVV

LfV

RIPPLE RMS OUT IN OUT

(

)

()

−

()

()()( )

029.

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now

becoming available in smaller case sizes. These are tempt-

ing for switching regulator use because of their very low

ESR. Unfortunately, the ESR is so low that it can cause

loop stability problems. Solid tantalum capacitor’s ESR

generates a loop “zero” at 5kHz to 50kHz that is instrumen-

tal in giving acceptable loop phase margin. Ceramic

capacitors remain capacitive to beyond 300kHz and usu-

ally resonate with their ESL before their ESR provides any

damping. They are appropriate for input bypassing be-

cause of their high ripple current ratings and tolerance of

turn-on surges.

OUTPUT RIPPLE VOLTAGE

Figure 3 shows a typical output ripple voltage waveform

for the LT1578. Ripple voltage is determined by the high

frequency impedance of the output capacitor, and ripple

current through the inductor. Peak-to-peak ripple current

through the inductor into the output capacitor is:

IVVV

VLf

POUT IN OUT

-P =

()

−

()

()()()

For high frequency switchers, the sum of ripple current

slew rates may also be relevant and can be calculated

from:

ΣdI

Output Capacitor

The output capacitor is normally chosen by its Effective

Series Resistance (ESR), because this is what determines

output ripple voltage. To get low ESR takes

volume

, so

physically smaller capacitors have high ESR. The ESR

range for typical LT1578 applications is 0.05Ω to 0.2Ω. A

typical output capacitor is an AVX type TPS, 100µF at 10V,

with a guaranteed ESR less than 0.1Ω. This is a “D” size

surface mount solid tantalum capacitor. TPS capacitors

are specially constructed and tested for low ESR, so they

give the lowest ESR for a given volume. The value in

microfarads is not particularly critical, and values from

22µF to greater than 500µF work well, but you cannot

cheat mother nature on ESR. If you find a tiny 22µF solid

tantalum capacitor, it will have high ESR, and output ripple

voltage will be terrible. Table 2 shows some typical solid

tantalum surface mount capacitors.

Table 2. Surface Mount Solid Tantalum Capacitor ESR

and Ripple Current

E Case Size ESR (Max.,

Ω

) Ripple Current (A)

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

AVX TAJ 0.7 to 0.9 0.4

D Case Size

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

C Case Size

AVX TPS 0.2 (typ) 0.5 (typ)

Many engineers have heard that solid tantalum capacitors

are prone to failure if they undergo high surge currents.

This is historically true, and type TPS capacitors are

specially tested for surge capability, but surge ruggedness

is not a critical issue with the

output

capacitor. Solid

tantalum capacitors fail during very high

turn-on

surges,

which do not occur at the output of regulators. High

discharge

surges, such as when the regulator output is

dead shorted, do not harm the capacitors.

Unlike the input capacitor, RMS ripple current in the

output capacitor is normally low enough that ripple cur-

rent rating is not an issue. The current waveform is

triangular with a typical value of 200mA

RMS

. The formula

to calculate this is:

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

CATCH DIODE

The suggested catch diode (D1) is a 1N5818 Schottky, or

its Motorola equivalent, MBR130. It is rated at 1A average

forward current and 30V reverse voltage. Typical forward

voltage is 0.42V at 1A. The diode conducts current only

during switch off time. Peak reverse voltage is equal to

regulator input voltage. Average forward current in normal

operation can be calculated from:

2µs/DIV 1578 F03

Peak-to-peak output ripple voltage is the sum of a

triwave

created by peak-to-peak ripple current times ESR, and a

square

wave created by parasitic inductance (ESL) and

ripple current slew rate. Capacitive reactance is assumed

to be small compared to ESR or ESL.

V I ESR ESL dI

RIPPLE

()( )

()

P-P

Example: with V

=10V, V

OUT

= 5V, L = 30µH, ESR = 0.1Ω,

ESL = 10nH:

RIPPLE

P-P

()

−

()











()()

+









=+=

−

510 5

10 30 10 200 10 042

30 10 033 10

0 42 0 1 10 10 0 33 10

0 042 0 003 45

••

•.•

.. • .•

IIVV

D AVG OUT IN OUT

(

)

=−

()

This formula will not yield values higher than 1A with

maximum load current of 1.25A unless the ratio of input to

output voltage exceeds 5:1. The only reason to consider a

larger diode is the worst-case condition of a high input

voltage and

overloaded

(not shorted) output. Under short-

circuit conditions, foldback current limit will reduce diode

current to less than 1A, but if the output is overloaded and

does not fall to less than 1/3 of nominal output voltage,

foldback will not take effect. With the overloaded condi-

tion, output current will increase to a typical value of 1.8A,

determined by peak switch current limit of 2A. With

= 15V, V

OUT

= 4V (5V overloaded) and I

OUT

= 1.8A:

D AVG

()

=−

()

1 8 15 4

15 132

This is safe for short periods of time, but it would be

prudent to check with the diode manufacturer if continu-

ous operation under these conditions must be tolerated.

BOOST␣ PIN␣ CONSIDERATIONS

For most applications, the boost components are a 0.33µF

capacitor and a 1N914 or 1N4148 diode. The anode is

connected to the regulated output voltage and this gener-

ates a voltage across the boost capacitor nearly identical

to the regulated output. In certain applications, the anode

may instead be connected to the unregulated input volt-

age. This could be necessary if the regulated output

voltage is very low (< 3V) or if the input voltage is less than

6V. Efficiency is not affected by the capacitor value, but the

capacitor should have an ESR of less than 1Ω to ensure

that it can be recharged fully under the worst-case condi-

tion of minimum input voltage. Almost any type of film or

ceramic capacitor will work fine.

WARNING!

Peak voltage on the BOOST pin is the sum of

unregulated input voltage plus the voltage across the

OUT

= 1A

INDUCTOR

CURRENT

AT I

OUT

= 1A

OUT

= 50mA

INDUCTOR

CURRENT

AT I

OUT

= 50mA

20mV/DIV

200mA/DIV

20mV/DIV

200mA/DIV

Figure 3. LT1578 Ripple Voltage Waveform

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

boost capacitor. This normally means that peak BOOST

pin voltage is equal to input voltage plus output voltage,

but

when the boost diode is connected to the regulator

input, peak BOOST pin voltage is equal to twice the input

voltage. Be sure that BOOST pin voltage does not exceed

its maximum rating

For nearly all applications, a 0.33µF boost capacitor works

just fine, but for the curious, more details are provided

here. The size of the boost capacitor is determined by

switch drive current requirements. During switch on time,

drain current on the capacitor is approximately I

OUT

/ 50. At

peak load current of 1.25A, this gives a total drain of 25mA.

Capacitor ripple voltage is equal to the product of on time

and drain current divided by capacitor value;

∆V = (t

)(25mA/C). To keep capacitor ripple voltage to

less than 0.5V (a slightly arbitrary number) at the worst-

case condition of t

= 4.7µs, the capacitor needs to be

0.24µF. Boost capacitor ripple voltage is not a critical

parameter, but if the minimum voltage across the capaci-

tor drops to less than 3V, the power switch may not

saturate fully and efficiency will drop. An

approximate

formula for absolute minimum capacitor value is:

–

2.42V

0.4V

GND

LT1578

INPUT

1578 F04

OUTPUT

SHDN

STANDBY

TOTAL

SHUTDOWN

3.5µA

Figure 4. Undervoltage Lockout

CIVV

fV V

MIN OUT OUT IN

OUT

()( )

()

−

()

//50

f = Switching frequency

OUT

= Regulated output voltage

= Minimum input voltage

This formula can yield capacitor values substantially less

than 0.24µF, but it should be used with caution since it

does not take into account secondary factors such as

capacitor series resistance, capacitance shift with tem-

perature and output overload.

SHUTDOWN FUNCTION AND

UNDERVOLTAGE LOCKOUT

Figure 4 shows how to add undervoltage lockout (UVLO)

to the LT1578. Typically, UVLO is used in situations where

the input supply is

current limited

, or has a relatively high

source resistance. It is particularly useful for input sup-

plies with foldback current limiting. A switching regulator

draws constant power from the source, so source current

increases as source voltage drops. This looks like a

negative resistance load to the source and can cause the

source to current limit and latch under low source voltage

LT1578/LT1578-2.5

conditions. UVLO helps prevent the regulator from oper-

ating at source voltages where these problems might

occur.

Threshold voltage for lockout is about 2.42V. A 3.5µA bias

current flows

out

of the pin at threshold. This internally

generated current is used to force a default high state on

the shutdown pin if the pin is left open. When low shut-

down current is not an issue, the error due to this current

can be minimized by making R

10k or less. If shutdown

current is an issue, R

can be raised to 100k, but the error

due to initial bias current and changes with temperature

should be considered.

RRV V

VR A

HI LO IN

()

=−

()

−

()

242

242 35

to 100k 25k suggested

..µ

= Minimum input voltage

Keep the connections from the resistors to the shutdown

pin short and make sure that interplane or surface capaci-

tance to the switching nodes are minimized. If high resis-

tor values are used, the shutdown pin should be bypassed

with a 1000pF capacitor to prevent coupling problems

from the switch node. If hysteresis is desired in the

undervoltage lockout point, a resistor R

can be added to

the output node. Resistor values can be calculated from:

RRV VV V

RRV V

LO IN OUT

FB HI OUT

=−+

()

[]

−

()

()( )

242 1

242 35

∆∆

∆

25k suggested for R

Input voltage at which switching stops as input

voltage descends to trip level

∆V = Hysteresis in input voltage level

Example: output voltage is 5V, switching is to stop if input

voltage drops below 12V and should not restart unless

APPLICATIONS INFORMATION

WUUU

input rises back to 13.5V. ∆V is therefore 1.5V and

= 12V. Let R

= 25k.

Rk k

=−+

()

[]

−

()

25 12 2 42 15 5 1 15

242 25 35

25 10 35

233 111

111 5 1 5 370

../ .

SWITCH NODE CONSIDERATIONS

For maximum efficiency, switch rise and fall times are

made as short as possible. To prevent radiated EMI and

high frequency resonance problems, proper layout of the

components connected to the switch node is essential. B

field (magnetic) radiation is minimized by keeping catch

diode, switch pin, and input bypass capacitor leads as

short as possible. E field radiation is kept low by minimiz-

ing the length and area of all traces connected to the switch

pin and BOOST pin. A ground plane should always be used

under the switcher circuitry to prevent interplane cou-

pling. A suggested layout for the critical components is

shown in Figure 5. Note that the feedback resistors and

compensation components are kept as far as possible

from the switch node. Also note that the high current

ground path of the catch diode and input capacitor are kept

very short and separate from the analog ground line.

The high speed switching current path is shown schemati-

cally in Figure 6. Minimum lead length in this path is

essential to ensure clean switching and low EMI. The path

including the switch, catch diode, and input capacitor is

the only one containing nanosecond rise and fall times. If

you follow this path on the PC layout, you will see that it is

irreducibly short. If you move the diode or input capacitor

away from the LT1578, get your resumé in order. The

other paths contain only some combination of DC and

200kHz triwave, so are much less critical.

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

Figure 6. High Speed Switching Path

Figure 5. Suggested Layout for LT1578

VOUT

VIN

BOOST FB

SYNC

SHDN

GND

1578 F05

GND

KEEP INPUT

CAPACITOR

AND CATCH

DIODE CLOSE

TO REGULATOR

AND TERMINATE

THEM TO THE

SAME POINT

CONNECT OUTPUT

CAPACITOR DIRECTLY

TO HEAVY GROUND

TAKE OUTPUT DIRECTLY FROM END

OF OUTPUT CAPACITOR TO AVOID

PARASITIC RESISTANCE AND

INDUCTANCE (KELVIN CONNECTION)

MINIMIZE AREA

OF CONNECTIONS

TO SWITCH NODE

AND BOOST NODE

GROUND RING NEED NOT BE AS SHOWN

(NORMALLY EXISTS AS INTERNAL PLANE)

MINIMIZE SIZE

OF FEEDBACK PIN

CONNECTIONS

TO AVOID PICKUP

TERMINATE

FEEDBACK

RESISTORS AND

COMPENSATION

COMPONENTS

DIRECTLY TO

SWITCHER

GROUND PIN

CCRC

1578 F06

HIGH

FREQUENCY

CIRCULATING

PATH

LOAD

SWITCH NODE

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

PARASITIC RESONANCE

Resonance or “ringing” may sometimes be seen on the

switch node (see Figure 7). Very high frequency ringing

following the switch voltage rise time is caused by switch/

diode/input capacitance lead inductance and diode ca-

pacitance. Schottky diodes have very high “Q” junction

capacitance that can ring for many cycles when excited at

high frequency. If total lead length for the input capacitor,

diode and switch path is 1 inch, the inductance will be

approximately 25nH. At switch off, this will produce a

spike across the NPN output device in addition to the input

voltage. At higher currents this spike can be in the order of

10V to 20V or higher with a poor layout, potentially

exceeding the absolute max switch voltage. The path

around switch, catch diode and input capacitor must be

kept as short as possible to ensure reliable operation.

When looking at this, a >100MHz oscilloscope must be

used, and waveforms should be observed on the leads of

the package. This switch off spike will also cause the SW

node to go below ground. The LT1578 has special circuitry

inside which mitigates this problem, but negative voltages

over 1V lasting longer than 10ns should be avoided. Note

that 100MHz oscilloscopes are barely fast enough to see

the details of the falling edge overshoot in Figure 7.

A second, much lower frequency ringing is seen during

switch off time if load current is low enough to allow the

inductor current to fall to zero during part of the switch off

time (see Figure 8). Switch and diode capacitance reso-

nate with the inductor to form damped ringing at 1MHz to

10 MHz. This ringing is not harmful to the regulator and it

has not been shown to contribute significantly to EMI. Any

attempt to damp it with an RC snubber will slightly degrade

efficiency.

INPUT BYPASSING AND VOLTAGE RANGE

Input Bypass Capacitor

Step-down converters draw current from the input supply

in pulses. The average height of these pulses is equal to

load current, and the duty cycle is equal to V

OUT

. Rise

and fall times of the current are very fast. A local bypass

capacitor across the input supply is necessary to ensure

proper operation of the regulator and minimize the ripple

current fed back into the input supply.

The capacitor also

forces switching current to flow in a tight local loop,

minimizing EMI

Do not cheat on the ripple current rating of the input

bypass capacitor, but also do not be overly concerned with

the value in microfarads

. The input capacitor is intended

to absorb all the switching current ripple, which can have

an RMS value as high as one half of the load current. Ripple

current ratings on the capacitor must be observed to

ensure reliable operation. In many cases it is necessary to

parallel two capacitors to obtain the required ripple rating.

Both capacitors must be of the same value and manufac-

turer to guarantee power sharing. The actual value of the

capacitor in microfarads is not particularly important

Figure 7. Switch Node Response

Figure 8. Discontinuous Mode Ringing

5V/DIV

50mA/DIV

50ns/DIV 1578 F07

1µs/DIV 1578 F08

INDUCTOR

CURRENT

SWITCH NODE

VOLTAGE

RISE AND FALL

WAVEFORMS ARE

SUPERIMPOSED

(PULSE WIDTH IS

NOT

350ns)

5V/DIV

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

because at 200kHz, any value above 15µF is essentially

resistive. RMS ripple current rating is the critical param-

eter. Actual RMS current can be calculated from:

IIVVVV

RIPPLE RMS OUT OUT IN OUT IN

()

=−

()

The term inside the radical has a maximum value of 0.5

when input voltage is twice output, and stays near 0.5 for

a relatively wide range of input voltages. It is common

practice therefore to simply use the worst-case value and

assume that RMS ripple current is one half of load current.

At maximum output current of 1.5A for the LT1578, the

input bypass capacitor should be rated at 0.75A ripple

current. Note however, that there are many secondary

considerations in choosing the final ripple current rating.

These include ambient temperature, average versus peak

load current, equipment operating schedule, and required

product lifetime. For more details, see Application Notes

19 and 46, and Design Note 95.

Input Capacitor Type

Some caution must be used when selecting the type of

capacitor used at the input to regulators. Aluminum

electrolytics are lowest cost, but are physically large to

achieve adequate ripple current rating, and size con-

straints (especially height) may preclude their use.

Ceramic capacitors are now available in larger values, and

their high ripple current and voltage rating make them

ideal for input bypassing. Cost is fairly high and footprint

may also be somewhat large. Solid tantalum capacitors

would be a good choice, except that they have a history of

occasional spectacular failures when they are subjected to

large current surges during power-up. The capacitors can

short and then burn with a brilliant white light and lots of

nasty smoke. This phenomenon occurs in only a small

percentage of units, but it has led some OEMs to forbid

their use in high surge applications. The input bypass

capacitors of regulators can see these high surges when

a battery or high capacitance source is connected. Several

manufacturers have developed a line of solid tantalum

capacitors specially tested for surge capability (AVX TPS

series for instance, see Table 3), but even these units may

fail if the input voltage surge approaches the maximum

voltage rating of the capacitor. AVX recommends derating

capacitor voltage by 2:1 for high surge applications. The

highest voltage rating is 50V, so 25V may be a practical

input voltage upper limit when using solid tantalum ca-

pacitors for input bypassing.

Larger capacitors may be necessary when the input volt-

age is very close to the minimum specified on the data

sheet. Small voltage dips during switch on time are not

normally a problem, but at very low input voltage they may

cause erratic operation because the input voltage drops

below the minimum specification. Problems can also

occur if the input-to-output voltage differential is near

minimum. The amplitude of these dips is normally a

function of capacitor ESR and ESL because the capacitive

reactance is small compared to these terms. ESR tends to

be the dominate term and is inversely related to physical

capacitor size within a given capacitor type.

SYNCHRONIZING

The SYNC pin is used to synchronize the internal oscillator

to an external signal. The SYNC input must pass from a

logic level low, through the maximum synchronization

threshold with a duty cycle between 10% and 90%. The

input can be driven directly from a logic level output. The

synchronizing range is equal to

initial

operating frequency

up to 400kHz. This means that

minimum

practical sync

frequency is equal to the worst-case

high

self-oscillating

frequency (250kHz), not the typical operating frequency of

200kHz. Caution should be used when synchronizing

above 280kHz because at higher sync frequencies the

amplitude of the internal slope compensation used to

prevent subharmonic switching is reduced. This type of

subharmonic switching only occurs at input voltages less

than twice output voltage. Higher inductor values will tend

to eliminate this problem. See Frequency Compensation

section for a discussion of an entirely different cause of

subharmonic switching before assuming that the cause is

insufficient slope compensation. Application Note 19 has

more details on the theory of slope compensation.

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

At power-up, when V

is being clamped by the FB pin (see

Figure 2, Q2), the sync function is disabled. This allows the

frequency foldback to operate in the shorted output con-

dition. During normal operation, switching frequency is

controlled by the internal oscillator until the FB pin reaches

0.7V, after which the SYNC pin becomes operational. If no

synchronization is required, this pin should be connected

to ground.

THERMAL CALCULATIONS

Power dissipation in the LT1578 chip comes from four

sources: switch DC loss, switch AC loss, boost circuit

current, and input quiescent current. The following formu-

las show how to calculate each of these losses. These

formulas assume continuous mode operation, so they

should not be used for calculating efficiency at light load

currents.

Switch loss:

PRI V

Vns I V f

SW SW OUT OUT

IN OUT IN

()( )

()()()

Boost current loss:

PVI

BOOST OUT OUT

()

50/

Quiescent current loss:

PV V

Q IN OUT

OUT

=



+















()

−−

055 10 16 10

0 004

.• .•

= Switch resistance (≈0.2Ω)

60ns = Equivalent switch current/voltage overlap time

f = Switch frequency

Example: with V

= 10V, V

OUT

= 5V and I

OUT

= 1A:

BOOST

()()()

+





()( )







=+ =

()( )

=



+



+

()( )

−

−−

02 1 5

10 60 10 1 10 200 10

01 012 022

5150

10 005

10 0 55 10 5 1 6 10 5 0 004

.••

.. .

.• .• .

.. 02W

Total power dissipation is 0.22 + 0.05 + 0.02 = 0.29W.

Thermal resistance for LT1578 package is influenced by

the presence of internal or backside planes. With a full

plane under the SO package, thermal resistance will be

about 80°C/W. No plane will increase resistance to about

120°C/W. To calculate die temperature, add in worst-case

ambient temperature:

= T

+ θ

TOT

)

With the SO-8 package (θ

= 80°C/W), at an ambient

temperature of 50°C,

= 50 + 80 (0.29) = 73.2°C

Die temperature is highest at low input voltage, so use

lowest continuous input operating voltage for thermal

calculations.

FREQUENCY COMPENSATION

Loop frequency compensation of switching regulators

can be a rather complicated problem because the reactive

components used to achieve high efficiency also intro-

duce multiple poles into the feedback loop. The inductor

and output capacitor on a conventional step-down con-

verter actually form a resonant tank circuit that can exhibit

peaking and a rapid 180° phase shift at the resonant

frequency. By contrast, the LT1578 uses a “current mode”

architecture to help alleviate the phase shift created by the

inductor. The basic connections are shown in Figure 9.

Figure 10 shows a Bode plot of the phase and gain of the

power section of the LT1578, measured from the V

pin to

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

the output. Gain is set by the 1.5A/V transconductance of

the LT1578 power section and the effective complex

impedance from output to ground. Gain rolls off smoothly

above the 160Hz pole frequency set by the 100µF output

capacitor. Phase drop is limited to about 85°. Phase

recovers and gain levels off at the zero frequency (≈16kHz)

set by capacitor ESR (0.1Ω).

Error amplifier transconductance phase and gain are shown

in Figure 11. The error amplifier can be modeled as a

transconductance of 1000µMho, with an output imped-

ance of 570kΩ in parallel with 2.4pF. In all practical

applications, the compensation network from the V

pin to

ground has a much lower impedance than the output

impedance of the amplifier at frequencies above 200Hz.

This means that the error amplifier characteristics them-

selves do not contribute excess phase shift to the loop, and

the phase/gain characteristics of the error amplifier sec-

tion are completely controlled by the external compensa-

tion network.

In Figure 12, full loop phase/gain characteristics are

shown with a compensation capacitor of 100pF, giving the

error amplifier a pole at 2.8kHz, with phase rolling off to

90° and staying there. The overall loop has a gain of 66dB

at low frequency, rolling off to unity-gain at 58kHz. The

phase plot shows a two-pole characteristic until the ESR

of the output capacitor brings it back to single pole above

16kHz. Phase margin is about 77° at unity-gain.

FREQUENCY (Hz)

GAIN (µMho)

PHASE (DEG)

2000

1500

1000

500

–500

200

150

100

–50

10 1k 10k 1M

1578 F11

100 100k

GAIN

PHASE

OUT

570k

OUT

2.4pF

ERROR AMPLIFIER EQUIVALENT CIRCUIT

LOAD

= 50Ω

1 × 10

–3

)(

–

1.21V

VSW

LT1578

GND

1578 F09

OUTPUT

ESR

ERROR

AMPLIFIER

CURRENT MODE

POWER STAGE

gm = 1.5A/V

Figure 10. Response from VC Pin to Output

FREQUENCY (Hz)

GAIN (dB)

PHASE (DEG)

–20

–40

–80

–120

100 1k

1578 F07

10k 100k

GAIN

PHASE

= 10V

OUT

= 5V

OUT

= 500mA

Figure 12. Overall Loop Characteristics

FREQUENCY (Hz)

LOOP GAIN (dB)

LOOP PHASE (DEG)

–20

180

135

–45

10 1k 10k 1M

1578 F12

100 100k

= 10V

OUT

= 5V

OUT

= 500mA

OUT

= 100µF

10V, AVX TPS

= 100pF

L = 30µH

PHASE

GAIN

Figure 9. Model for Loop Response Figure 11. Error Amplifier Gain and Phase

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

Analog experts will note that around 7kHz, phase dips

close to the zero phase margin line. This is typical of

switching regulators, especially those that operate over a

wide range of loads. This region of low phase is not a

problem as long as it does not occur near unity-gain. In

practice, the variability of output capacitor ESR tends to

dominate all other effects with respect to loop response.

Variations in ESR

will

cause unity-gain to move around,

but at the same time phase moves with it so that adequate

phase margin is maintained over a very wide range of ESR

(≥ ±3:1).

What About a Resistor in the Compensation Network?

It is common practice in switching regulator design to add

a “zero” to the error amplifier compensation to increase

loop phase margin. This zero is created in the external

network in the form of a resistor (R

) in series with the

compensation capacitor. Increasing the size of this resis-

tor generally creates better and better loop stability, but

there are two limitations on its value. First, the combina-

tion of output capacitor ESR and a large value for R

may

cause loop gain to stop rolling off altogether, creating a

gain margin problem. An approximate formula for R

where gain margin falls to zero is:

R Loop V

G G ESR

COUT

MP MA

Gain =1

()

()()()()

121.

= Transconductance of power stage = 1.5A/V

= Error amplifier transconductance = 1(10

–3

)

ESR = Output capacitor ESR

1.21 = Reference voltage

With V

OUT

= 5V and ESR = 0.1Ω, a value of 27.5k for R

would yield zero gain margin, so this represents an upper

limit. There is a second limitation however which has

nothing to do with theoretical small signal dynamics. This

resistor sets high frequency gain of the error amplifier,

including the gain at the switching frequency. If the

switching frequency gain is high enough, an excessive

amout of output ripple voltage will appear at the V

resulting in improper operation of the regulator. In a

marginal case,

subharmonic

switching occurs, as

evidenced by alternating pulse widths seen at the switch

node. In more severe cases, the regulator squeals or

hisses audibly even though the output voltage is still

roughly correct. None of this will show on a Bode plot

since this is an amplitude insensitive measurement.

Tests

have shown that if ripple voltage on the V

is held to less

than 100mV

P-P

, the LT1578 will generally be well behaved

The formula below will give an estimate of V

ripple

voltage when R

is added to the loop, assuming that R

large compared to the reactance of C

at 200kHz.

VR G V V ESR

VLf

C RIPPLE C MA IN OUT

()

()( )

−

()()()

121.

= Error amplifier transconductance (1000µMho)

If a series compensation resistor of 15k gave the best

overall loop response, with adequate gain margin, the

resulting V

pin ripple voltage with V

= 10V, V

OUT

= 5V,

ESR = 0.1Ω, L = 30µH, would be:

VkV

C RIPPLE

()

−

()

−

()()()

()

()()

15 1 10 10 5 01 121

10 30 10 200 10 0 151

•..

•• .

This ripple voltage is high enough to possibly create

subharmonic switching. In most situations a compromise

value (<10k in this case) for the resistor gives acceptable

phase margin and no subharmonic problems. In other

cases, the resistor may have to be larger to get acceptable

phase response, and some means must be used to control

ripple voltage at the V

pin. The suggested way to do this

is to add a capacitor (C

) in parallel with the R

network

on the V

pin. The pole frequency for this capacitor is

typically set at one-fifth of the switching frequency so that

it provides significant attenuation of the switching ripple,

but does not add unacceptable phase shift at the loop

unity-gain frequency. With R

= 15k,

CfR kpF

()()()

()

2 200 10 15 265

ππ•

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

One way to check switching regulator loop stability is by

pulse loading the regulator output while observing the

transient response at the output, using the circuit shown

in Figure 13. The regulator loop is “hit” with a small

transient AC load current at a relatively low frequency,

50Hz to 1kHz. This causes the output to jump a few

millivolts, then settle back to the original value, as shown

in Figure 14. A well behaved loop will settle back cleanly,

whereas a loop with poor phase or gain margin will “ring”

as it settles. The

number

of rings indicates the degree of

stability, and the

frequency

of the ringing shows the

approximate unity-gain frequency of the loop.

Amplitude

of the signal is not particularly important, as long as the

amplitude is not so high that the loop behaves nonlinearly.

How Do I Test Loop Stability?

The “standard” compensation for LT1578 is a 100pF

capacitor for C

, with R

= 0. While this compensation will

work for most applications, the “optimum” value for loop

compensation components depends, to various extents,

on parameters which are not well controlled. These in-

clude

inductor value

(±30% due to production tolerance,

load current and ripple current variations),

output capaci-

tance

(±20% to ±50% due to production tolerance,

temperature, aging and changes at the load),

output

capacitor ESR

(±200% due to production tolerance,

temperature and aging), and finally,

DC input voltage and

output load current

. This makes it important for the

designer to check out the final design to ensure that it is

“robust” and tolerant of all these variations.

0.2ms/DIV 1578 F14

10mV/DIV

OUT

= 500mA

BEFORE FILTER

OUT

= 500mA

AFTER FILTER

LOAD PULSE

THROUGH 50Ω

f ≈ 780Hz

5A/DIV

OUT

= 50mA

AFTER FILTER

Figure 14. Loop Stability Check

OSCILLOSCOPE

SYNC

ADJUSTABLE

DC LOAD

ADJUSTABLE

INPUT SUPPLY

100Hz TO 1kHz

100mV TO 1V

P-P

100µF TO

1000µF

RIPPLE FILTER

1578 F13

TO X1

OSCILLOSCOPE

PROBE

3300pF 330pF

50Ω

470Ω4.7k

SWITCHING

REGULATOR

Figure 13. Loop Stability Test Circuit

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

The output of the regulator contains both the desired low

frequency transient information and a reasonable amount

of high frequency (200kHz) ripple. The ripple makes it

difficult to observe the small transient, so a two-pole,

100kHz filter has been added. This filter is not particularly

critical; even if it attenuated the transient signal slightly,

this wouldn’t matter because amplitude is not critical.

After verifying that the setup is working correctly, start

varying load current and input voltage to see if you can find

any combination that makes the transient response look

suspiciously “ringy.” This procedure may lead to an ad-

justment for best loop stability or faster loop transient

response. Nearly always you will find that loop response

looks better if you add in several kΩ for R

. Do this only

if necessary, because as explained before, R

above 1k

may require the addition of C

to control V

pin ripple.

If everything looks OK, use a heat gun and cold spray on

the circuit (especially the output capacitor) to bring out

any temperature-dependent characteristics.

Keep in mind that this procedure does not take initial

component tolerance into account. You should see fairly

clean response under all load and line conditions to ensure

that component variations will not cause problems. One

note here: according to Murphy, the component most

likely to be changed in production is the output capacitor,

because that is the component most likely to have manu-

facturer variations (in ESR) large enough to cause prob-

lems. It would be a wise move to lock down the sources of

the output capacitor in production. Also, try varying com-

ponent values by a factor of 2 and see if the behavior is still

acceptable. Double and halve the values of R

and C

and

output capacitors. If the regulator still works correctly, it

will likely be good in production.

A possible exception to the “clean response” rule is at very

light loads, as evidenced in Figure 14 with I

LOAD

= 50mA.

Switching regulators tend to have dramatic shifts in loop

response at very light loads, mostly because the inductor

current becomes discontinuous. One common result is very

slow but stable characteristics. A second possibility is low

phase margin, as evidenced by ringing at the output with

transients. The good news is that the low phase margin at

light loads is not particularly sensitive to component varia-

tion, so if it looks reasonable under a transient test, it will

probably not be a problem in production. Note that

fre-

quency

of the light load ringing may vary with component

tolerance but phase margin generally hangs in there.

POSITIVE-TO-NEGATIVE CONVERTER

The circuit in Figure 15 is a classic positive-to-negative

topology using a grounded inductor. It differs from the

standard approach in the way the IC chip derives its

feedback signal. Because the LT1578 accepts only posi-

tive feedback signals, the ground pin must be tied to the

regulated negative output. A resistor divider to ground or,

in this case, the sense pin, then provides the proper

feedback voltage for the chip.

Figure 15. Positive-to-Negative Converter

OUTPUT**

–5V, 0.5A

INPUT

5.5V TO

15V

1578 F15

0.33µF

1N5818

100µF

10V TANT

×2

15.8k

4.99k

10µF TO

50µF

1N4148

L1*

15µH

BOOST

LT1578

GND V

* INCREASE L1 TO 30µH OR 60µH FOR HIGHER CURRENT APPLICATIONS.

SEE APPLICATIONS INFORMATION

** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE

AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION

Inverting regulators differ from buck regulators in the

basic switching network. Current is delivered to the output

square waves with a peak-to-peak amplitude much

greater than load current

. This means that

maximum load

current will be significantly less than the LT1578’s 1.5A

maximum switch current, even with large inductor values

The buck converter in comparison, delivers current to the

output as a triangular wave superimposed on a DC level

equal to load current, and load current can approach 1.5A

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

with large inductors. Output ripple voltage for the positive-

to-negative converter will be much higher than a buck

converter. Ripple current in the output capacitor will also

be much higher. The following equations can be used to

calculate operating conditions for the positive-to-negative

converter.

Maximum load current:

IVV

VVfL

VV VV

MAX

PIN OUT

OUT IN OUT IN

OUT IN OUT F

−

()( )

()()()













()

−

()

+−

()

2035

035

= Maximum rated switch current

= Minimum input voltage

OUT

= Output voltage

= Catch diode forward voltage

0.35 = Switch voltage drop at 1.5A

Example: with V

IN(MIN)

= 5.5V, V

OUT

= 5V, L = 30µH,

= 0.5V, I

= 1.5A: I

MAX

= 0.6A. Note that this equation

does not take into account that maximum rated switch

current (I

) on the LT1578 is reduced slightly for duty

cycles above 50%. If duty cycle is expected to exceed 50%

(input voltage less than output voltage), use the actual I

value from the Electrical Characteristics table.

Operating duty cycle:

DC VV

VVV

OUT F

IN OUT F

−+ +03.

(This formula uses an average value for switch loss, so it

may be several percent in error.)

With the conditions above:

DC =+

−++=

505

55 03 5 05 51

.. . %

This duty cycle is close enough to 50% that I

can be

assumed to be 1.5A.

OUTPUT DIVIDER

If the adjustable part is used, the resistor connected to

OUT

(R2) should be set to approximately 5k. R1 is

calculated from:

RRV

OUT

12121

121

=−

()

INDUCTOR VALUE

Unlike buck converters, positive-to-negative converters

cannot use large inductor values to reduce output ripple

voltage. At 200kHz, values larger than 75µH make almost

no change in output ripple. The graph in Figure 16 shows

peak-to-peak output ripple voltage for a 5V to –5V con-

verter versus inductor value. The criteria for choosing the

INDUCTOR SIZE (µH)

OUTPUT RIPPLE VOLTAGE (mV

P-P

)

150

120

060

1578 F16

15 30 45 75

DISCONTINUOUS

LOAD

= 0.25A

DISCONTINUOUS

LOAD

= 0.1A

5V TO –5V CONVERTER

OUTPUT CAPACITOR’S

ESR = 0.1Ω

CONTINUOUS

LOAD

> 0.38A

Figure 16. Ripple Voltage on Positive-to-Negative Converter

LT1578/LT1578-2.5

APPLICATIONS INFORMATION

WUUU

For the example above, with maximum load current of

0.25A:

CONT

()()

()

55 15

455555505 038

.. .

This says that discontinuous mode can be used and the

minimum inductor needed is found from:

MIN

()( )







()

25 025

200 10 1 5

•..µ

In practice, the inductor should be increased by about 30%

over the calculated minimum to handle losses and varia-

tions in value. This suggests a minimum inductor of 7.3µH

for this application, but looking at the ripple voltage chart

shows that output ripple voltage could be reduced by a fac-

tor of two by using a 30µH inductor. There is no rule of thumb

here to make a final decision. If modest ripple is needed and

the larger inductor does the trick, this is probably the best

solution. If ripple is noncritical use the smaller inductor. If

ripple is extremely critical, a second stage filter may have

to be added in any case, and the lower value of inductance

can be used. Keep in mind that the output capacitor is the

other critical factor in determining output ripple voltage.

Ripple shown on the graph (Figure 16) is with a capacitor’s

ESR of 0.1Ω. This is

reasonable for AVX type TPS “D” or

“E” size surface mount solid tantalum capacitors, but the

final capacitor chosen must be looked at carefully for ESR

characteristics.

inductor is therefore typically based on ensuring that peak

switch current rating is not exceeded. This gives the

lowest value of inductance that can be used, but in some

cases (lower output load currents) it may give a value that

creates unnecessarily high output ripple voltage. A com-

promise value is often chosen that reduces output ripple.

As you can see from the graph,

large

inductors will not

give arbitrarily low ripple, but

small

inductors can give

high ripple.

The difficulty in calculating the minimum inductor size

needed is that you must first know whether the switcher

will be in continuous or discontinuous mode at the critical

point where switch current is 1.5A. The first step is to use

the following formula to calculate the load current where

the switcher must use continuous mode. If your load

current is less than this, use the discontinuous mode

formula to calculate the minimum inductor value needed.

If the load current is higher, use the continuous mode

formula.

Output current where continuous mode is needed:

IVI

VV VV V

CONT IN P

IN OUT IN OUT F

()()

()

Minimum inductor discontinuous mode:

LVI

MIN OUT OUT

()()

()( )

Minimum inductor continuous mode:

LVV

fV V I I VV

MIN IN OUT

IN OUT P OUT OUT F

()( )

()

−++

()

























LT1578/LT1578-2.5

Ripple Current in the Input and Output Capacitors

Positive-to-negative converters have high ripple current in

both the input and output capacitors. For long capacitor

lifetime, the RMS value of this current must be less than

the high frequency ripple current rating of the capacitor.

The following formula will give an

approximate

value for

RMS ripple current.

This formula assumes continuous

conduction mode and a large inductor value

. Small induc-

tors will give somewhat higher ripple current, especially in

discontinuous mode. The exact formulas are very com-

plex and appear in Application Note 44, pages 30 and 31.

For our purposes here, a simple fudge factor (ff) is added.

The value for ff is about 1.2 for load currents above 0.38A

(in continuous conduction mode) and L ≥10µH. It in-

creases to about 2.0 for smaller inductors at lower load

currents (in discontinuous conduction mode).

Capacitor ff I V

OUT OUT

IRMS =

()( )

ff = Fudge factor (1.2 to 2.0)

APPLICATIONS INFORMATION

WUUU

Diode Current

Average

diode current is equal to load current.

Peak

diode

current will be considerably higher.

Peak diode current:

Continuous

IVV

LfV V

Discontinuous V

OUT IN OUT

IN OUT

OUT

Mode

Mode = 2I

OUT

()

()( )

()()

()

()( )

()()

Keep in mind that during start-up and output overloads,

the average diode current may be much higher than with

normal loads. Care should be used if diodes rated less than

1A are used, especially if continuous overload conditions

must be tolerated.

LT1578/LT1578-2.5

Dimensions in inches (millimeters) unless otherwise noted.

PACKAGE DESCRIPTION

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.016 – 0.050

(0.406 – 1.270)

0.010 – 0.020

(0.254 – 0.508)× 45°

0°– 8° TYP

0.008 – 0.010

(0.203 – 0.254)

SO8 1298

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

TYP

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

BSC

1234

0.150 – 0.157**

(3.810 – 3.988)

8765

0.189 – 0.197*

(4.801 – 5.004)

0.228 – 0.244

(5.791 – 6.197)

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

LT1578/LT1578-2.5

PART NUMBER DESCRIPTION COMMENTS

LT1074/LT1076 Step-Down Switching Regulators 40V Input, 100kHz, 5A and 2A

LTC1174 High Efficiency Step-Down and Inverting DC/DC Converter 0.5A, 150kHz Burst ModeTM Operation

LT1370 High Efficiency DC/DC Converter 42V, 6A, 500kHz Switch

LT1371 High Efficiency DC/DC Converter 35V, 3A, 500kHz Switch

LT1372/LT1377 500kHz and 1MHz High Efficiency 1.5A Switching Regulators Boost Topology

LT1376 High Efficiency Step-Down Switching Regulator 25V, 1.5A, 500kHz Switch

LT1507 High Efficiency Step-Down Switching Regulator 15V, 1.5A, 500kHz Switch

LT1676/LT1776 High Efficiency Step-Down Switching Regulators 7.4V to 60V Input, 100kHz/200kHz

LTC1772 SOT-23 Low Voltage Step-Down DC/DC Controller 550kHz, Drives PFET, 6-Lead SOT-23 Package; up to 4.5A Output Current

LTC1735 High Efficiency Step-Down Converter Synchronous Buck Controller Drives External MOSFETs

LT1777 Low Noise Step-Down Switching Regulator 48V Input, Internally Limited dV/dt, Programmable di/dt

Burst Mode is a trademark of Linear Technology Corporation.

1578f LT/TP 0100 4K • PRINTED IN USA

 LINEAR T ECHNOLOGY CORPORATION 1999

TYPICAL APPLICATION

Dual Output SEPIC␣ Converter

The circuit in Figure 17 generates both positive and

negative 5V outputs with a single piece of magnetics. The

inductor L1 is a 33µH surface mount inductor from

Coiltronics. It is manufactured with two identical windings

that can be connected in series or parallel. The topology for

the 5V output is a standard buck converter. The –5V

topology would be a simple flyback winding coupled to the

buck converter if C4 were not present. C4 creates the

SEPIC (Single-Ended Primary Inductance Converter) to-

pology which improves regulation and reduces ripple

current in L1. Without C4, the voltage swing on L1B

compared to L1A would vary due to relative loading and

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

●

FAX: (408) 434-0507

●

www.linear-tech.com

BOOST

LT1578

OUTPUT

–5V

†

* L1 IS A SINGLE CORE WITH TWO WINDINGS

COILTRONICS CTX33-2

** AVX TSPD107M010

†

IF LOAD CAN GO TO ZERO, AN OPTIONAL

PRELOAD OF 1k TO 5k MAY BE USED TO

IMPROVE LOAD REGULATION

INPUT

TO 15V

GND

1578 F17

0.33µF

100pF D1

1N5818

C1**

100µF

10V TANT

C5**

100µF

10V TANT

22µF

35V TANT

C4**

100µF

1N914

15.8k

4.99k

1N5818

L1A*

33µH

L1B*

GND

SHDN V

RELATED PARTS

Figure 17. Dual Output SEPIC Converter

coupling losses. C4 provides a low impedance path to

maintain an equal voltage swing in L1B, improving regu-

lation. In a flyback converter, during switch on time, all the

converter’s energy is stored in L1A only, since no current

flows in L1B. At switch off, energy is transferred by

magnetic coupling into L1B, powering the –5V rail. C4

pulls L1B positive during switch on time, causing current

to flow, and energy to build in L1B and C4. At switch off,

the energy stored in both L1B and C4 supply the –5V rail.

This reduces the current in L1A and changes L1B current

waveform from square to triangular. For details on this

circuit see Design Note 100.