MIC2172/3172
100kHz 1.25A Switching Regulators
Micrel Inc. • 2180 Fort une Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel .com
April 2006 M9999-041806
(408) 955-1690
General Description
The MIC2172 and MIC3172 are complete 100kHz SMPS
current-mode controllers with internal 65V 1.25A power
switches. The MIC2172 features external frequency
synchronization or frequency adjustment, while the
MIC3172 features an enable/shutdown control input.
Although primarily intended for voltage step-up
applications, the floating switch architecture of the
MIC2172/3172 makes it practical for step-down, inverting,
and Cuk configurations as well as isolated topologies.
Operating from 3V to 40V, the MIC2172/3172 draws only
7mA of quiescent current making it attractive for battery
operated supplies.
The MIC3172 is for applications that require on/off control
of the regulator. The MIC3172 is externally shutdown by
applying a TTL low signal to EN (enable). W hen disabled,
the MIC3172 draws only leakage current (typically less
than 1µA). EN must be high for normal operation. For
applicat ions n ot r equ ir in g contr ol , EN must be tied to VIN or
TTL high.
The MIC2172 is for applications requiring two or more
SMPS re gulators that op erate from the s ame input su pply.
The MIC2172 features a SYNC input which allows locking
of its internal oscillator to an external reference. This
makes it possible to avoid the audible beat frequencies
that result from the unequal oscillator frequencies of
indepe nde nt SM PS regulator s .
A reference signal can be supplied by one MIC2172
designated as a master. To insure locking of the slave’s
oscillators, the reference oscillator frequency must be
higher than the slave’s. The master MIC2172’s oscillator
frequency is increased up to 135kHz by connecting a
resistor from SYNC to ground (see applications
information).
The MIC2172/3172 is available in an 8-pin plastic DIP or
SOIC for –40°C to +85°C operation.
Features
1.25A, 65V internal switch rating
3V to 40V input voltage range
Current-mode operation
Internal cycle-by-cycle current limit
Thermal shutdown
Low external parts count
Operates in most switching topologies
7mA quiescent current (operating)
<1µA quiescent current, shutdown mode (MIC3172)
TTL shutdown compatibility (MIC3172)
External frequency synchronization (MIC2172)
External frequency trim (MIC2172)
Fits most LT1172 sockets (see applications info)
Applications
Laptop/palmtop computers
Toys
Hand-held instruments
Off-line converter up to 50W (requires external power
switch)
Predriver for higher power capability
Master/slave configurations (MIC2172)
___________________________________________________________________________________________________________
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Typical Applications
Figure 1. MIC2172 5V to 12V Boost Converter
Figure 2. MIC3172 Flyback Converter
Ordering Information
Part Number
Standard Pb-Free
Junction Temp. Range Package
MIC 2172BN MIC 2172Y N –40°C to +85°C 8-pin plastic DIP
MIC2172BM MIC2172YM –40°C to +85°C 8-pin SOIC
MIC 3172BN MIC 3172Y N –40°C to +85°C 8-pin plastic DIP
MIC3172BM MIC3172YM –40°C to +85°C 8-pin SOIC
Note:
1. Other Voltage available. Contact Micrel for details.
Pin Configuration
8-pin DIP (N) 8-pin SOIC (M)
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Pin Description
Pin Number Pin Name Pin Function
1 S GND Signal Ground: Internal analog circuit ground. Connect directly to the input filter
capacitor for proper operation (see applications info). Keep separate from power
grounds.
2 COMP Frequency Compensation: Output of transconductance type error amplifier.
Primary function is for loop stabilization. Can also be used for output voltage
soft-start and curr ent lim it tailo ring.
3 FB Feedback: Inverting input of error amplifier. Connect to external resistive divider
to set power supply output voltage.
4 (MIC2172) SYNC Synchronization/Frequency Adjust: Capacitively coupled input signal greater
than device’s free running frequency (up to 135kHz) will lock device’s oscillator
on falling edge. Oscillator frequency can be trimmed up to 135kHz by adding a
resistor to ground. If unused, pin must float (no connection).
4 (MIC3172) EN Enable: Apply TTL high or connect to VIN to enable the regulator. Apply TTL low
or connect to ground to disable the regulator. Device draws only leakage current
(<1µA) when disabled.
5 VIN Supply Voltage: 3.0V to 40V
6 P GND 2 Power Ground #2: One of two NPN power switch emitters with 0.3 current
sense resistor in series. Required. Connect to external inductor or input voltage
ground depending on circuit topology.
7 VSW Power Switch Collector: Collector of NPN switch. Connect to external inductor or
input voltage depending on circuit topology.
8 P GND 1 Power Ground #1: One of two NPN power switch emitters with 0.3 current
sense resistor in series. Optional. For maximum power capability connect to P
GND 2. Floating pin reduces current limit by a factor of two.
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Absolute Maximum Ratings MIC2172
Input Voltage.................................................................. 40V
Switch Voltage............................................................... 65V
Sync Current............................................................... 50mA
Feedback Voltage (Transient, 1ms) ............................ ±15V
Operating Temperature Range
8-pin PDIP.................................................–40 to +85°C
8-pin SOIC ................................................–40 to +85°C
Junction Temperature................................–55°C to +150°C
Thermal Resistance
θJA 8-pin PDIP................................................. 130°C/W
θJA 8-pin SOIC.................................................120°C/W
Storage Temperature ................................–65°C to +150°C
Soldering (10 sec.) ...................................................+300°C
Electrical Characteristics MIC2172
Note 1, 3. Unless otherwise specified, VIN = 5V.
Parameter Condition Min Typ Max Units
Reference Section Pin 2 tied to pin 3
Feedback Voltage (VFB) 1.220
1.214 1.240 1.264
1.274 V
V
Feedback Voltage Line
Regulation 3V VIN 40V 0.03 %/V
Feedback Bias Current
(IFB) 310 750
1100 nA
nA
Error Amplifier Section
Transconductance
(ICOMP/VFB) ICOMP = ±25µA 3.0
2.4 3.9 6.0
7.0 µA/mV
µA/mV
Voltage Gain
(VCOMP/VFB) 0.9V VCOMP 1.4V 500 800 2000 V/V
Output Current VCOMP = 1.5V 125
100 175 350
400 µA
µA
Output Swing High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V 1.8
0.25 2.1
0.35 2.3
0.52 V
V
Compensation Pin
Threshold Duty Cycle = 0 0.8
0.6 0.9 1.08
1.25 V
V
Output Switch Section
ON Resistance ISW = 1A, VFB = 0.8V 0.76 1
1.1
Current Limit Duty Cycle = 50%, TJ 25°C
Duty Cycle = 50%, TJ < 25°C
Duty Cycle = 80%, Note 2
1.25
1.25
1
3
3.5
2.5
A
A
A
Breakdown Voltage (BV) 3V VIN 40V
ISW = 5mA 65 75 V
Notes:
1. Exceeding the absolute maximum rat i ng may damage the device.
2. The device is not guaranteed to function outside its operat i ng rating.
3. Devices are ESD sensit i ve. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification f or packaged product only.
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Typical Characteris tics MI C2172 (cont)
Parameter Condition Min Typ Max Units
Oscillator Section
Frequency (fO) 88
85 100 112
115 kHz
kHz
Duty Cycle [δ(max)]
80 89
95 %
Sync Coupling Capacitor
Required for Frequency
Lock
VPP = 3.0V
VPP = 40V 22
2.2 51
4.7 120
10 pF
pF
Input Supply Voltage Section
Minimum Operating
Voltage 2.7
3.0 V
Quiescent Current (IQ) 3V VIN 40V, VCOMP = 0.6V, ISW = 0 7 9 mA
Supply Current Increase
(IIN) ISW = 1A, VCOMP = 1.5V 9 20 mA
Electrical Characteristics MIC3172
Note 1, 3. Unless otherwise specified, VIN = 5V.
Parameter Condition Min Typ Max Units
Reference Section Pin 2 tied to pin 3
Feedback Voltage (VFB) 1.224
1.214 1.240 1.264
1.274 V
V
Feedback Voltage Line
Regulation 3V VIN 40V 0.07 %/V
Feedback Bias Current
(IFB) 310 750
1100 nA
nA
Error Amplifier Section
Transconductance
(ICOMP/VFB) ICOMP = ±25µA 3.0
2.4 3.9 6.0
7.0 µA/mV
µA/mV
Voltage Gain
(VCOMP/VFB) 0.9V VCOMP 1.4V 500 800 2000 V/V
Output Current VCOMP = 1.5V 125
100 175 350
400 µA
µA
Output Swing High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V 1.8
0.25 2.1
0.35 2.3
0.52 V
V
Compensation Pin
Threshold Duty Cycle = 0 0.8
0.6 0.9 1.08
1.25 V
V
Output Switch Section
ON Resistance ISW = 1A, VFB = 0.8V 0.76 1
1.1
Current Limit Duty Cycle = 50%, TJ 25°C
Duty Cycle = 50%, TJ < 25°C
Duty Cycle = 80%, Note 2
1.25
1.25
1
3
3.5
2.5
A
A
A
Breakdown Voltage (BV) 3V VIN 40V
ISW = 5mA 65 75 V
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Typical Characteris tics MI C3172 (cont)
Parameter Condition Min Typ Max Units
Oscillator Section
Frequency (fO) 88
85 100 112
115 kHz
kHz
Duty Cycle [δ(max)]
80 89
95 %
Sync Coupling Capacitor
Required for Frequency
Lock
VPP = 3.0V
VPP = 40V 22
2.2 51
4.7 120
10 pF
pF
Input Supply Voltage Section and Enable Section
Minimum Operating
Voltage 2.7
3.0 V
Quiescent Current (IQ) 3V VIN 40V, VCOMP = 0.6V, ISW = 0 7 9 mA
Supply Current Increase
(IIN) ISW = 1A, VCOMP = 1.5V 9 20 mA
Enable Input Threshold 0.4 1.2 2.4 V
Enable Input Current VEN = 0V
VEN = 2.4V –1 0
2 1
10 µA
µA
Bold type denotes specifications applicabl e to the full operat i ng temperature range.
Note 1. Devices are ESD sens iti ve. Handli ng precautions required.
Note 2. For duty cycles (δ) between 50% and 95%, minimum guaranteed switch current is given by ICL = 0.833 (2- δ) for the MIC3172.
Note 3. Specification for packaged product only.
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Typical Characteris t ics
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Typical Characteris tics (cont.)
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Functional Characteristics
MIC2172 Block Diagram
MIC3172 Block Diagram
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Functional Description
Refer to “Block Diagram MIC2172” and “Block Diagram
MIC3172.”
Internal Po w er
The MIC2172/3172 operates when VIN is 2.6V (and
VEN 2.0V for the MIC3172). An internal 2.3V regulator
supplies biasing to all internal circuitry including a
precision 1.24V band gap reference.
The enable control (MIC3172 only) enables or disables
the internal regulator which supplies power to all other
internal circuitry.
PWM Operation
The 100kHz oscillator generates a signal with a duty
cycle of approximately 90%. The current-mode
compar ator o utpu t is us ed t o reduce th e d uty cycle wh en
the current amplifier output voltage exceeds the error
amplifier output voltage. The resulting PWM signal
controls a driver which supplies base current to output
transistor Q1.
Current Mode Advantages
The MIC2172 /3172 op erates in curr ent mode rather than
voltage mode. There are three distinct advantages to
this technique. Feedback loop compensation is greatly
simplified because inductor current sensing removes a
pole from the closed loop response. Inherent cycle-by-
cycle current limiting greatly improves the power switch
reliability and provides automatic output current limiting.
Finall y, curre nt-mode oper ation pro vides autom atic input
voltag e feed f orward whic h pre vents inst antaneous input
voltage changes from disturbing the output voltage
setting.
Anti-Saturation
The anti-s aturation diode ( D1) increases the us able dut y
cycle range of the MIC2172/3172 by eliminating the
base to collector stored charge which would delay Q1’s
turnoff.
Compensation
Loop stability compensation of the MIC2172/3172 can
be accomplished by connecting an appropriate network
from either COMP to circuit ground (Typical
Applications) or COMP to FB.
The error amplifier output (COMP) is also useful for soft
start and current limiting. Because the error amplifier
output is a transconductanc e type, the o utpu t impedanc e
is rel atively high which m eans the outp ut voltage c an be
easily clamped or adjusted externally.
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Application Information
Using the MIC3172 Enable Control (New Designs)
For new designs requiring enable/shutdown control,
connect EN to a TTL or CMOS control signal (figure 3).
The very low driver current requirement ensures
compatibility regardless of the driver or gate used.
Figure 3. MIC3172 TTL Enable/Shutdown
Using the MIC3172 in LT1172 Applications
The MIC3172 can be used in most original LT1172
applications by adapting the MIC3172’s
enable/shutdown feature to the existing LT1172 circuit.
Unlike the LT1172 which can be shutdown by reducing
the voltage on pin 2 (VC) belo w 0.15V, the MIC 3172 has
a dedicated enable/shutdown pin. To replace the
LT1172 with the MIC3172, determine if the LT1172’s
shutdown feature is used.
Circuits without Shutdown
If the shutd own feat ure is not be ing used, connect EN to
VIN to continuously enable the MIC3172 or use an
MIC2172 with SYNC open (figure 4).
Figure 4. MIC2172/3172 Always Enabled
Circuits with Shutdown
If shutdo wn was used in the or iginal LT1172 a pplicati on,
connect EN to a logic gate that produces a TTL logic-
level output signal that matches the shutdown signal.
The MIC3172 will be enabled by a logic-high input and
shutdown with a logic-low input (figure 5). The actual
components perform ing the functions of U1 and Q1 may
vary according to the original application.
Figure 5. Adapting to the LT1172 Socket
By using the MIC3172, U1 and Q1 sho wn in f igure 5 can
be eliminated, reducing the total components count.
Synchronizing the MIC2172
Using several unsynchronized switching regulators in the
same circuit will cause beat frequencies to appear on the
inputs and outputs. These beat frequencies can be very
low making them difficult to filter.
Micrel’s MIC2172 can be synchronized to a single
master frequency avoiding the possibility of undesirable
beat frequencies in multiple regulator circuits. The
master frequency can be an external oscillator or a
designated master MIC2172. The master frequency
should be 1 .05 to 1.2 0 tim es the slav e’s 1 00kHz nom inal
frequency to guarantee synchronization.
Figure 6. Master/Slave Synchronization
Figure 6 shows a typical application where several
MIC2172s operate from the same supply voltage. U1’s
oscill ator fr equenc y is incre ased above U2 ’s and U3 ’s b y
connecting a resistor from SYNC to ground. U2-SYNC
and U3-SYNC are capacitively coupled to the master’s
output (VSW). The slaves lock to the negative (falling
edge) of U1’s output waveform.
Figure 7. External Synchronization
Care must be exercised to insure that the master
MIC2172 is always operating in continuous mode.
Figure 7 shows how one or more MIC2172s can be
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locked to an external reference frequency. The slaves
lock to the negative (falling edge) of the external
reference waveform.
Soft Start
A diode-coupled capacitor from COMP to circuit ground
slows the output voltage rise at turn on (figure 8).
Figure 8. Soft Start
The additional time it takes for the error amplifier to
charge th e capacitor c orresponds to the tim e it takes the
output to reach regulation. Diode D1 discharges C1
when VIN is removed.
Another soft start circuit is shown in figure 8A. The
circuit us es c apac it or C1 to contr ol th e out put r ise time b y
providing feedback from the output to the FB pin. The
output vol tage starts to rise when the MI C3172 re gulator
starts switching. T his is the dv/dt of the output will forc e
a current through capacitor C1, which flows through the
lower feedback resistor, R2, increasing the voltage on
the FB pin. This increased voltage on the FB pin
reduces the d uty cycle at th e VSW pin, lim iting the tur n- on
time of the output. Increasing the value of C1 causes
the output voltage to rise more slowly. Diode D1 is
reverse biased in normal operation and prevents C1
from appearing in parallel with the upper voltage divider
resistor, which would affect stability and transient
response. Zener diode D2 clamps the voltage seen by
the feedback pin and provides a discharge path for C1
when the power supp ly is turned of f.
Figure 8a. Additional Soft Start Circuit
This circuit only limits the dv/dt of the output when the
boost con verter is runn ing. It will n ot decrease th e dv/dt
or the i nitial inrush c aused by app lying the input voltag e.
Figure 8B shows the turn-on without a soft start circuit
and Figure 8C shows how the soft start circuit reduces
inrush and prevents output voltage overshoot.
Figure 8b. Without Soft Start
Figure 8c. With Soft Start
Current Limit
For designs demanding less output current than the
MIC2172/ 3172 is c apable of deliv ering, P G ND 1 can be
left open reducing the current capability of Q1 by one-
half.
Figure 9. Current Limit
Alternatively, the maximum current limit of the
MIC2172/3172 can be reduced by adding a voltage
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clamp to the COMP output (figure 9). This feature can be
useful in applications requiring either a complete
shutdown of Q1’s switching action or a form of current
fold-bac k lim iting. This use of the COMP o utput do es not
disable the oscillator, amplifiers or other circuitry,
therefore the supply current is never less than
approximately 5mA.
Thermal Management
Although the MIC2172/3172 family contains thermal
protection circuitry, for best reliability, avoid prolonged
operation with junction temperatures near the rated
maximum.
The junction temperature is determined by first
calculating the power dissipation of the device. For the
MIC2172/3172, the total power dissipation is the sum of
the device operating losses and power switch losses.
The device operating losses are the dc losses
associated with biasing all of the internal functions plus
the losses of the power switch driver circuitry. The dc
losses are calculated from the supply voltage (VIN) and
device supply current (IQ). The MIC2172/3172 supply
current is almost constant regardless of the supply
voltage (see “Electrical Characteristics”). The driver
section l osses (not inc ludin g the s witch) ar e a func tion of
supply voltage, power switch current, and duty cycle.
()
()
+
+=
+50
δ0.004
IVI VP SWINQINdriverbias
where:
P(bias+driver) = device operating losses
VIN = supply voltage
IQ = quiescent supply current
ISW = power switch current
(see “Design Hints: Switch Current Calculations”)
δ = duty cycle
FOUT
INFOUT VV VVV
δ
+
±+
=
V
OUT = output vo lta ge
V
F = D1 forward voltage drop
As a practical example refer to figure 1.
VIN = 5.0V
IQ = 0.006A
ISW = 0.625A
δ = 60% (0 .6)
Then:
()
()
()
0.068WP50 0.60.004
0.62550.0065P
driverbias
driverbias
=
+
+×=
+
+
Power switch dissipation calculations are greatly
simplified by making two assumptions which are usually
fairly accurate. First, the majority of losses in the power
switch are du e to o n-losses . T o find thes e loss es, as sign
a resistance value to the collector/emitter terminals of
the device using the saturation voltage versus collector
current curves (see Typical Performance
Characteristics). Power switch losses are calculated by
modeling the switch as a resistor with the switch duty
cycle modifying the average power dissipation.
PSW = (ISW)2 RSW δ
From the Typical performance Characteristics:
RSW = 1
Then:
PSW = (0.625)2 × 1 × 0.6
PSW = 0.234W
P(total) = 0.068 + 0.234
P(total) = 0.302W
The junction temperature for any semiconductor is
calculated using the following:
TJ = TA + P(total) θJA
Where:
TJ = junction temperature
TA = ambient temperature (maximum)
P(total) = total power dissipation
θJA = junction to ambient thermal resistance
For the practical example:
TA = 70°C
θJA = 130°C/W (for plastic DIP)
Then:
TJ = 70 + 0.30 130
TJ = 109°C
This junction temperature is below the rated maximum of
150°C.
Grounding
Refer to figure 10. Heavy lines indicate high current
paths.
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Figure 10. Single Point Ground
A single point ground is strongly recommended for
proper operation.
The signal ground, compensation network ground, and
feedback network connections are sensitive to minor
voltage variations. The input and output capacitor
grounds and power ground conductors will exhibit
voltage drop when carrying large currents. Keep the
sensitive circuit ground traces separate from the power
ground traces. Small voltage variations applied to the
sensitive circuits can prevent the MIC2172/3172 or any
switching regulator from functioning properly.
Applications and Design Hints
Access to both the collector and emitter(s) of the NPN
power switch makes the MIC2172/3172 extremely
versati le and suitable f or us e in most PW M power s up p l y
topologies.
Boost Conversion
Refer to figure 11 for a typical boost conversion
application where a +5V logic supply is available but
+12V at 0.14A is required.
Figure 11. 5V to 12V Boost Converter
The first step in designing a boost converter is
determ ining whether induct or L1 will caus e the con verter
to operate in either continuous or discontinuous mode.
Discontinuous mode is preferred because the feedback
control of the converter is simpler.
When L1 discharges its current completely during the
MIC2172/3172’s off-time, it is operating in discontinuous
mode.
L1 is operating in continuous mode if it does not
discharge completely before the MIC2172/3172 power
switch is turned on again.
Discontinuo us Mode Des ig n
Given the m aximum output current, s olve equati on (1) to
determine whether the device can operate in
discontinuous mode without initiating the internal device
current limit.
OUT
IN
CL
OUT V
δ V
2
I
I
(1)
FOUT
INFOUT VV VVV
δ
+
±+
= (1a)
Where:
ICL = internal switch current limit
ICL = 1.25A when δ < 50%
ICL = 0.833 (2 – δ) when δ 50%
(Refer to Electrical Characteristics.)
IOUT = maximum output current
VIN = minimum input voltage
δ = duty cycle
VOUT = required outp u t vo lta ge
VF = D1 forward voltage drop
For the example in figure 11.
IOUT = 0.14A
ICL = 1.147A
VIN = 4.75V (minimum)
δ = 0.623
VOUT = 12.0V
VF = 0.6V
Then:
0.141AI12
0.6234.75
2
1.147
I
OUT
OUT
××
This value is greater than the 0.14A output current
requirement so we can proceed to find the inductance
value of L1.
()
SWOUT
2
IN f P 2
δ V
L1 (2)
Where:
POUT = 12 0.14 = 1.68W
fSW = 1105kHz (100kHz)
For our practical example:
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()
5
2
1011.6820.6234.75
L1 ×××
×
IL1 26.062µH (use 27µH)
Equation (3) solves for L1’s maximum current value.
L1
T V
IONIN
L1(peak) = (3)
Where:
TON = δ / fSW = 6.23×10-6 sec
6
6
L1(peak) 1027 106.234.75
I
×
××
=
IL1(peak) = 1.096A
Use a 27µH inductor with a peak current rating of at
least 1.4A.
Flyback Conversion
Flyback converter topology may be used in low power
applications where voltage isolation is required or
whenever the input voltage can be less than or greater
than the output voltage. As with the step-up converter
the inductor (transformer primary) current can be
continuous or discontinuous. Discontinuous operation is
recommended.
Figure 12 shows a practical flyback converter design
using the MIC317 2.
Switch Operation
During Q1’s on time ( Q1 is the internal N PN transistor
see block diagrams), energy is stored in T1’s primary
inductance. During Q1’s off time, stored energy is
partially discharged into C4 (output filter capacitor).
Careful selection of a low ESR capacitor for C4 may
provide satisfactory output ripple voltage making
additional filter stages unnecessary.
C1 (input capacitor) m ay be reduced or eliminat ed if the
MIC3172 is located near a low impedance voltage
source.
Output Diode
The output di ode allo ws T1 to stor e energy in its prim ar y
inductance (D2 nonconducting) and release energy into
C4 (D2 conducting). The low forward voltage drop of a
Schottky diode minimizes power loss in D2.
Frequency Compensation
A simple frequency compensation network consisting of
R3 and C2 prevents output oscillations.
High impedance output stages (transconductance type)
in the MIC2172/ 3172 often perm it simp lified lo op-st abil ity
solutions to be connected to circuit ground, although a
more conventional technique of connecting the
components from the error amplifier output to its
inverti ng inp ut is also pos s i ble.
Voltage Clipper
Care must be taken to minimize T1’s leakage
inductance, otherwise it may be necessary to
incorpora te the voltage c lipper cons isting of D1, R4, a nd
C3 to avoid second breakdown (failure) of the
MIC3172’s power NPN Q1.
Enable/Shutdown
The MIC3172 includes the enable/shutdown feature.
W hen the de vic e is s h utdo wn, total s up ply curren t is les s
than 1µA. This is ideal for battery applications where
portions of a system are powered only when needed. If
this feature is not required, simply connect EN to VIN or
to a TTL high voltage.
Discontinuo us Mode Des ig n
When designing a discontinuous flyback converter, first
determine whether the device can safely handle the
peak primary current demand placed on it by the output
power. Equation (8) finds the maximum duty cycle
required for a given input voltage and output power. If
the duty cycle is greater than 0.8, discontinuous
operation cannot be used.
IN(min)CL
OUT
VIP 2
δ (8)
For a practical example let:
POUT = 5.0V × 0.25A = 1.25W
VIN = 4.0V to 6.0V
ICL = 1.25A when δ < 50%
Then:
41.25
1.252
δ
×
×
δ 0.5 (50%) Use 0.55.
The s lightly hig her dut y c ycle valu e is used to overco me
circuit i neff iciencies. A few iter ations of equati on (8) m ay
be required if the duty cycle is found to be greater than
50%.
Calculate the maximum transformer turns ratio a, or
NPRI/NSEC, that will guarantee safe operation of the
MIC2172/3172 power switch.
SEC
IN(max)CECE VVF V
a±
(9)
Where:
a = transformer maximum turns ratio
VCE = power switch collector to emitter maximum
voltage
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FCE = safety derating factor (0.8 for most
commercial and industrial applications)
VIN(max) = maximum input voltage
VSEC = transformer secondary voltage (VOUT + VF)
For the practical example:
VCE = 65V max. for the MIC2172/3172
FCE = 0.8
VSEC = 5.6V
Then:
5.6 6.00.865
a±×
a 8.2143
Next, calculate the maximum primary inductance
required t o store the need ed output energ y with a power
switch duty cycle of 55%.
OUT
2
ON
2
IN(min)SW
PRI PT Vf 0.5
L (10)
Where:
LPRI = maximum primary inductance
fSW = device switching frequency (100kHz)
VIN(min) = minimum input voltage
TON = power switch on time
Then:
()
1.25 105.54.01010.5
L2
625
PRI
××××
LPRI 19.23µH
Use an 18µH primary inductance to overcome circuit
inefficiencies.
To complete the design the inductance value of the
secondary is found which will guarantee that the energy
stored in the transformer during the power switch on
time will be completed discharged into the output during
the off-time. This is necessary when operating in
discontinuous-mode.
OUT
2
OFF
2
SECSW
SEC PT Vf 0.5
L (11)
Where:
LSEC = maximum secondary inductance
TOFF = power switch off time
Then:
()
1.25 104.55.61010.5
L2
625
SEC
×××××
LSEC 25.4µH
Figure 12. MIC3172 5V 0.25A Flyback Converter
Micrel MIC2172/3172
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Finally, recalculate the transformer turns ratio to insure
that it is less than the value earlier found in equation (9).
SEC
PRI
L
L
a (12)
Then:
5
5
102.54 101.8
a
×
×
a 0.84 Use 0.8 (same as 1:1.25).
This ratio is les s than th e ratio ca lcula ted in eq uat ion ( 9).
When specifying the transf ormer it is necessar y to k now
the primary peak current which must be withstood
without satur at ing the transf ormer core.
PRI
ONIN(min)
PEAK(pri) LT V
I=
So:
18µ8105.54.0
I6
PEAK(pri)
××
= (13)
IPEAK(pri) = 1.22A
Now find the minimum reverse voltage requirement for
the output rectifier. This rectifier must have an average
current rating greater than the maximum output current
of 0.25A.
()
aa
BR
OUTIN(max)
BR FVV
V+
(14)
Where:
VBR = output rectifier maximum peak reverse
voltage rating
a = transformer turns ratio (0.8)
FBR = reverse voltage safety derating factor (0.8)
Then:
()
15.625VV0.80.8 0.85.06.0
V
BR
BR
×
×+
A 1N5817 will safely handle voltage and current
requirements in this example.
Forward Converters
Micrel’s MIC2172/3172 can be used in several circuit
configurations to generate an output voltage which is
less than the i nput vo ltage (buc k or step-down topo logy) .
Figure 13 shows the MIC3172 in a voltage step-down
applicat ion. Becaus e of the internal arc hitecture of these
devices, more external components are required to
implem ent a s tep-down r egulat or than with other devic es
offered by Micrel (refer to the LM257x or LM457x family
of buck switchers). However, for step-down conversion
requiring a tr ansf ormer (f orward), th e MIC2 172/31 72 i s a
good choice.
A 12V to 5V step-down converter using transformer
isolation (forward) is shown in figure 14. Unlike the
isolated flyback converter which stores energy in the
primary inductance during the controller’s on-time and
releases it to the load during the off-time, the forward
converter transfers energy to the output during the on-
time, using the off-time to reset the transformer core. In
the application shown, the transformer core is reset by
the tertiary winding discharging T1’s peak magnetizing
current through D2.
For most forward converters the duty cycle is limited to
50%, allowing the transform er f lux to reset with only two
times the input voltage appearing across the power
switch. Although during normal operation this circuit’s
duty cycle is well below 50%, the MIC2172 (and
MIC3172) has a maximum duty cycle capability of 90%.
If 90% was required during operation (start-up and high
load currents), a complete reset of the transformer
during the off-time would require the voltage across the
power switch to be ten times the input voltage. This
would limit the input voltage to 6V or less for forward
converter applications.
To prevent core saturation, the application given here
uses a duty cycle limiter consisting of Q1, C4 and R3.
Whenever the MIC3172 exceeds a duty cycle of 50%,
T1’s reset winding current turns Q1 on. This action
reduces th e duty cyc le of the MIC3 172 un til T 1 is able t o
reset during each cycle.
Fluorescent Lamp Supply
An extremely useful application of the MIC3172 is
generating an ac voltage for fluorescent lamps used as
liquid crystal display back lighting in portable computers.
Figure 15 shows a complete power supply for lighting a
fluorescent lamp. Transistors Q1 and Q2 together with
capacitor C2 form a Royer oscillator. The Royer
oscillator generates a sine wave whose frequency is
determ ined b y th e ser i es L/ C c irc uit c omprised of T1 and
C2. Assum ing that the MIC3172 and L1 are absent, and
the transistors’ emitters are grounded, circuit operation is
described in “Oscillator Operation.”
Oscillator Operation
Resistor R2 provides initial base current that turns
transistor Q1 on and impresses the input voltage across
one half of T1’s primary winding (Pri 1). T1’s feedback
winding provides additional base drive (positive
feedback) to Q 1 f or c ing it well in to s aturat ion f or a period
determined by the Pri 1/C2 time constant. Once the
voltage across C2 has reached its maximum circuit
value, Q1’s collector current will no longer increase.
Since T1 is in s er ies with Q1 , th is drop in pr imar y curr ent
causes the flux in T1 to change and because of the
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mutual coupling to t he feedback winding further reduc es
primary current eventually turning Q1 off. The primary
windings now change state with the feedback winding
forcing Q2 on repeating the alternate half cycle exactly
as with Q1. This action produces a sinusoidal voltage
wave form; whose amplitude is proportional to the input
voltage, across T1’s primary winding which is stepped
up and capacitively coupled to the lamp.
Lamp Current Regulation
Initial ionization (lighting) of the fluorescent lamp
requires several times the ac voltage across it than is
required to sustain current through the device. The
current through the lamp is sampled and regulated by
the MIC3172 to achieve a given intensity. The MIC3172
uses L1 to maintain a constant average current through
the transistor emitters. This current controls the voltage
amplitude of the Royer osc illator a nd mainta ins the lam p
current. During the negative half cycle, lamp current is
rectified by D3. During the positive half cycle, lamp
current is rectif ied by D2 th rough R4 and R 5. R3 and C5
filter the voltage dropped across R4 and R5 to the
MIC3172’s feedback pin. The MIC3172 maintains a
constant lamp curr ent by adjusting its duty cycle to k eep
the feedback voltage at 1.2 4V. The intensity of the la mp
is adjusted using potentiometer R5. The MIC3172
adjusts its duty cycle accordingly to bring the average
voltage across R4 and R5 back to 1.24V.
On/Off Control
Especially important for battery powered applications,
the lamp can be remotely or automatically turned off
using the MIC3172’s EN pin. The entire circuit draws
less than 1µA while shutdown.
Efficiency
To obtain maximum circuit efficiency careful selection of
Q1 and Q2 for low c ollecto r to em itter saturation voltage
is a m us t. Induc tor L1 sh ou ld b e c hos e n f or minim al c or e
and copper losses at the switching frequency of the
MIC3172, and T1 should be carefully constructed from
magnetic materials optimized for the output power
required at the Royer oscillator frequency. Suitable
inductors may be obtained from Coiltronics, Inc., tel:
(407) 241-7876.
Figure 13. Step-Down or Buck Regulator
Micrel MIC2172/3172
April 2006 19 M9999-041806
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Figure 14. 12V to 5V Forward Converter
Figure 15. LCD Backlight Fluorescent Lamp Supply
Micrel MIC2172/3172
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Package Information
8-Pin Plastic DIP (N)
8-Pin SOIC (M)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 US
A
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The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any t ime without notification to the customer.
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t
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