Sample & Buy Product Folder Support & Community Tools & Software Technical Documents LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 LMV221 50-MHz to 3.5-GHz 40-dB Logarithmic Power Detector for CDMA and WCDMA 1 Features 3 Description * * * * * * * * The LMV221 is a 40-dB RF power detector intended for use in CDMA and WCDMA applications. The device has an RF frequency range from 50 MHz to 3.5 GHz. It provides an accurate temperature and supply-compensated output voltage that relates linearly to the RF input power in dBm. The circuit operates with a single supply from 2.7 V to 3.3 V. 1 2.7-V to 3.3-V Supply Voltage 40-dB Linear in dB Power Detection Range 0.3-V to 2-V Output Voltage Range Shutdown Multi-Band Operation from 50 MHz to 3.5 GHz 0.5-dB Accurate Temperature Compensation External Configurable Output Filter Bandwidth 2.5 mm x 2.2 mm x 0.8 mm 6-pin WSON Package 2 Applications * * * * UMTS/CDMA/WCDMA RF Power Control GSM/GPRS RF Power Control PA Modules IEEE 802.11b, g (WLAN) The LMV221 has an RF power detection range from -45 dBm to -5 dBm and is ideally suited for direct use in combination with a 30-dB directional coupler. Additional low-pass filtering of the output signal can be achieved by means of an external resistor and capacitor. shows a detector with an additional output low pass filter. The filter frequency is set with RS and CS. shows a detector with an additional feedback low pass filter. Resistor RP is optional and lowers the trans-impedance gain (RTRANS). The filter frequency is set with CP//CTRANS and RP//RTRANS. The device is active for Enable = High, otherwise it is in a low power consumption shutdown mode. To save power and prevent discharge of an external filter capacitance, the output (OUT) is high-impedance during shutdown. Device Information(1) PART NUMBER LMV221 PACKAGE BODY SIZE (NOM) WSON (6) 2.50 mm x 2.20 mm (1) For all available packages, see the orderable addendum at the end of the data sheet. space space Typical Application: Output RC Low Pass Filter RF COUPLER ANTENNA PA space RF Typical Application: Feedback (R)C Low Pass Filter COUPLER 50 : 50 : VDD VDD RFIN 2 1 6 OUT 4 3 5 RS RFIN REF 2 + CS LMV221 EN ANTENNA PA ADC - GND Copyright (c) 2016, Texas Instruments Incorporated 1 6 OUT LMV221 EN 4 3 + RP 5 REF CP ADC - GND Copyright (c) 2016, Texas Instruments Incorporated 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Table of Contents 1 2 3 4 5 6 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 6.1 6.2 6.3 6.4 6.5 6.6 6.7 7 1 1 1 2 3 4 Absolute Maximum Ratings ...................................... 4 ESD Ratings.............................................................. 4 Recommended Operating Conditions....................... 4 Thermal Information .................................................. 4 2.7-V DC and AC Electrical Characteristics.............. 5 Timing Requirements .............................................. 11 Typical Characteristics ............................................ 12 Detailed Description ............................................ 23 7.1 Overview ................................................................. 23 7.2 Functional Block Diagram ....................................... 23 7.3 Feature Description................................................. 23 7.4 Device Functional Modes........................................ 30 8 Application and Implementation ........................ 31 8.1 Application Information............................................ 31 8.2 Typical Applications ............................................... 34 9 Power Supply Recommendations...................... 38 10 Layout................................................................... 39 10.1 Layout Guidelines ................................................ 39 10.2 Layout Example .................................................... 41 11 Device and Documentation Support ................. 42 11.1 11.2 11.3 11.4 Community Resources.......................................... Trademarks ........................................................... Electrostatic Discharge Caution ............................ Glossary ................................................................ 42 42 42 42 12 Mechanical, Packaging, and Orderable Information ........................................................... 42 4 Revision History NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision C (March 2013) to Revision D Page * Added Device Information and Pin Configuration and Functions sections, ESD Ratings table and Thermal Information table, Feature Description, Device Functional Modes, Application and Implementation, Power Supply Recommendations, Layout, Device and Documentation Support, and Mechanical, Packaging, and Orderable Information sections................................................................................................................................................................ 1 * Changed RJA value from 86.6C/W to 100.4C/W................................................................................................................. 4 Changes from Revision B (March 2013) to Revision C * 2 Page Changed layout of National Semiconductor data sheet to TI format.................................................................................... 41 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 5 Pin Configuration and Functions NGF Package 6-Pin WSON Top View VDD 1 RFIN 2 GND 3 DAP (GND) 6 OUT 5 REF 4 EN Pin Functions PIN NUMBER NAME TYPE 1 VDD 3 GND 2 RFIN Analog input 4 EN Logic input 5 REF 6 OUT DAP GND Power supply Output DESCRIPTION Positive supply voltage Power ground RF input signal to the detector, internally terminated with 50 . The device is enabled for EN = high, and brought to a low-power shutdown mode for EN = Low. Reference output, for differential output measurement (without pedestal). Connected to inverting input of output amplifier. Ground referenced detector output voltage (linear in dB) -- Ground (must be connected) Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 3 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings over operating free-air temperature range (unless otherwise noted) (1) MIN MAX UNIT 3.6 V 10 dBm 400 mV SUPPLY VOLTAGE VDD - GND RF INPUT Input power DC voltage Enable input voltage VSS - 0.4 V < VEN < VDD + 0.4 V Junction temperature (2) Maximum lead temperature (soldering, 10 seconds) Storage temperature, Tstg (1) (2) -65 150 C 260 C 150 C Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. The maximum power dissipation is a function of TJ(MAX) , RJA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/RJA. All numbers apply for packages soldered directly into a PC board. 6.2 ESD Ratings VALUE V(ESD) (1) (2) Electrostatic discharge Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) 2000 Charged-device model (CDM), per JEDEC specification JESD22C101 (2) 2000 Machine model 200 UNIT V JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. 6.3 Recommended Operating Conditions over operating free-air temperature range (unless otherwise noted) MIN NOM MAX UNIT Supply voltage 2.7 3.3 V Ambient temperature -40 85 C RF frequency RF input power (1) (1) 50 3500 MHz -45 -5 dBm -58 -18 dBV Power in dBV = dBm + 13 when the impedance is 50 . 6.4 Thermal Information LVM221 THERMAL METRIC (1) NGF (WSON) UNIT 6 PINS RJA Junction-to-ambient thermal resistance (2) 100.4 C/W RJC(top) Junction-to-case (top) thermal resistance 120 C/W RJB Junction-to-board thermal resistance 7 C/W JT Junction-to-top characterization parameter 69.6 C/W JB Junction-to-board characterization parameter 6.9 C/W RJC(bot) Junction-to-case (bottom) thermal resistance 69.9 C/W (1) (2) 4 For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics, SPRA953. The maximum power dissipation is a function of TJ(MAX) , RJA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/RJA. All numbers apply for packages soldered directly into a PC board. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 6.5 2.7-V DC and AC Electrical Characteristics Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated. (1) MIN (2) TYP (3) MAX (2) Active mode: EN = high, no signal present at RFIN 6.5 7.2 8.5 Active mode: EN = high, no signal present at RFIN TA = -40C to 85C 5 PARAMETER TEST CONDITIONS UNIT SUPPLY INTERFACE IDD Supply current Shutdown: EN = low, no signal present at RFIN 10 0.5 Shutdown: EN = low, no signal present at RFIN TA = -40C to 85C EN = Low: PIN = 0 dBm TA = -40C to 85C mA 3 4 A (4) 10 LOGIC ENABLE INTERFACE VLOW EN logic low input level (shutdown mode) TA = -40C to 85C VHIGH EN logic high input level TA = -40C to 85C IEN Current into EN pin TA = -40C to 85C 0.6 V 1 A 60 1.1 V RF INPUT INTERFACE RIN (1) (2) (3) (4) Input resistance 40 47.1 2.7-V DC and AC Electrical Characteristics values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ > TA. All limits are ensured by test or statistical analysis. Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and also depend on the application and configuration. The typical values are not tested and are not specified on shipped production material. All limits are ensured by design and measurements which are performed on a limited number of samples. Limits represent the mean 3-sigma values. The typical value represents the statistical mean value. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 5 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 2.7-V DC and AC Electrical Characteristics (continued) Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated.(1) PARAMETER TEST CONDITIONS MIN (2) TYP (3) MAX (2) 16 40 UNIT OUTPUT INTERFACE From positive rail, sourcing, VREF = 0 V, IOUT = 1 mA VOUT Output voltage swing From positive rail, sourcing, VREF = 0 V, IOUT = 1 mA TA = -40C to 85C 50 mV From negative rail, sinking, VREF = 2.7 V, IOUT = 1 mA 14 From negative rail, sinking, VREF = 2.7 V, IOUT = 1 mA TA = -40C to 85C Output short circuit current IOUT 3 Sourcing, VREF = 0 V, VOUT = 2.6 V TA = -40C to 85C 2.7 Sinking, VREF = 2.7 V, VOUT = 0.1 V 3 Sinking, VREF = 2.7 V, VOUT = 0.1 V TA = -40C to 85C 2.7 Small signal bandwidth No RF input signal. Measured from REF input current to VOUT RTRANS Output amplifier transimpedance gain No RF input signal, from IREF to VOUT, DC SR Slew rate 50 Sourcing, VREF = 0 V, VOUT = 2.6 V BW Output impedance (4) IOUT,SD Output leakage current in shutdown mode 5.4 mA 5.7 450 35 42.7 Positive, VREF from 2.7 V to 0 V 3 4.1 Positive, VREF from 2.7 V to 0 V TA = -40C to 85C 2.7 Negative, VREF from 0 V to 2.7 V 3 Negative, VREF from 0 V to 2.7 V TA = -40C to 85C 2.7 No RF input signal, EN = High, DC measurement ROUT 40 kHz 55 V/s 4.2 0.6 No RF input signal, EN = High, DC measurement TA = -40C to 85C 5 6 EN = Low, VOUT = 2 V 21 EN = Low, VOUT = 2 V TA = -40C to 85C k 300 500 nA RF DETECTOR TRANSFER = 50 MHz, PIN= -5 dBm = 50 MHz, PIN= -5 dBm TA = -40C to 85 1.76 1.67 = 900 MHz, PIN= -5 dBm = 900 MHz, PIN= -5 dBm TA = -40C to 85C 1.75 1.67 = 1855 MHz, PIN= -5 dBm VOUT,MAX Maximum output voltage (4) = 1855 MHz, PIN= -5 dBm TA = -40C to 85C 1.53 1.42 6 Submit Documentation Feedback V 1.57 1.4 1.33 = 3500 MHz, PIN= -5 dBm = 3500 MHz, TA = -40C to 85C 1.68 1.49 = 3000 MHz, PIN= -5 dBm = 3000 MHz, PIN= -5 dBm TA = -40C to 85C 1.82 1.61 = 2500 MHz, PIN= -5 dBm = 2500 MHz, PIN= -5 dBm TA = -40C to 85C 1.83 1.48 1.28 1.21 1.36 Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 2.7-V DC and AC Electrical Characteristics (continued) Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated.(1) MIN (2) TYP (3) MAX (2) Minimum output voltage (pedestal) No input signal 175 250 350 No input signal, TA = -40C to 85C 142 388 Pedestal variation over temperature No input signal, relative to 25C TA = -40C to 85C -20 20 PARAMETER VOUT,MIN TEST CONDITIONS = 50 MHz, PIN from -45 dBm to -5 dBm = 50 MHz, PIN from -45 dBm to -5 dBm TA = -40C to 85C 1.37 Output voltage range (4) = 1855 MHz, PIN from -45 dBm to -5 dBm TA = -40C to +85C 1.34 1.24 1.14 = 50 MHz KSLOPE PINT Logarithmic slope (4) Logarithmic intercept (4) en Output referred noise (5) vN (4) Output referred noise 1.3 1.07 1.2 1.01 0.96 1.09 39 40.5 = 900 MHz 36.7 38.5 40 = 1855 MHz 34.4 35.7 37.1 = 2500 MHz 32.6 33.8 35.2 = 3000 MHz 31 32.5 34 = 3500 MHz 30 31.9 33.5 = 50 MHz -50.4 -49.4 -48.3 = 900 MHz -54.1 -52.8 -51.6 = 1855 MHz -53.2 -51.7 -50.2 = 2500 MHz -51.8 -50 -48.3 = 3000 MHz -51.1 -48.9 -46.6 = 3500 MHz -49.6 -46.8 -44.1 PIN = -10 dBm at 10 kHz 1.5 Integrated over frequency band, 1 kHz to 6.5 kHz 100 Integrated over frequency band, 1 kHz to 6.5 kHz TA = -40C to 85C PSRR (5) Power supply rejection ratio (5) V 1.12 = 3500 MHz, PIN from -45 dBm to -5 dBm = 3500 MHz, PIN from -45 dBm to -5 dBm TA = -40C to 85C 1.37 1.2 = 3000 MHz, PIN from -45 dBm to -5 dBm = 3000 MHz, PIN from -45 dBm to -5 dBm TA = -40C to 85C 1.47 1.3 = 2500 MHz, PIN from -45 dBm to -5 dBm = 2500 MHz, PIN from -45 dBm to -5 dBm TA = -40C to 85C 1.52 1.4 = 1855 MHz, PIN from -45 dBm to -5 dBm VOUT 42 mV/dB dBm V/Hz 150 PIN = -10 dBm, = 1800 MHz PIN = -10 dBm, = 1800 MHz TA = -40C to 85C mV 1.44 = 900 MHz, PIN from -45 dBm to -5 dBm = 900 MHz, PIN from -45 dBm to -5 dBm TA = -40C to 85C UNIT VRMS 60 55 dB This parameter is ensured by design and/or characterization and is not tested in production. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 7 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 2.7-V DC and AC Electrical Characteristics (continued) Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated.(1) PARAMETER TEST CONDITIONS MIN (2) TYP (3) MAX (2) UNIT POWER MEASUREMENT PERFORMANCE = 50 MHz -40 dBm PIN -10 dBm -0.6 = 50 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -1.1 = 900 MHz -40 dBm PIN -10 dBm -0.7 = 900 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm ELC Log conformance error (4) = 1855 MHz -40 dBm PIN -10 dBm -0.4 = 1855 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -1.1 = 2500 MHz -40 dBm PIN -10 dBm = 2500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm = 3000 MHz -40 dBm PIN -10 dBm = 3000 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm = 3500 MHz -40 dBm PIN -10 dBm = 3500 MHz, TA = -40C to 85C E1 8 dB Measurement error for a 1-dB Input power step (4) 0.37 0.46 0.48 1.1 0.51 1.1 1.34 0.56 -1.73 1.6 2.72 2.7 -1.1 0.4 1.4 -1 0.38 1.27 = 1855 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -1.1 0.44 1.31 = 2500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -1.1 0.48 1.15 = 3000 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -1.2 0.5 0.98 = 3500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -1.2 0.62 0.85 dB = 50 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.06 0.069 = 900 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.056 0.056 = 1855 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.069 0.069 = 2500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.084 0.084 = 3000 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.092 0.092 = 3500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.1 0.1 Submit Documentation Feedback dB 0.56 -0.87 -1.2 1.1 0.24 -0.43 -1 1.3 0.84 = 900 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm EVOT 0.53 -2 = 50 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm Variation over temperature (4) -1.24 0.56 dB Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 2.7-V DC and AC Electrical Characteristics (continued) Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated.(1) PARAMETER E10 dB TEST CONDITIONS MIN (2) -0.65 0.57 = 900 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.75 0.58 -0.88 0.72 -0.86 0.75 = 3000 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.85 0.77 = 3500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm -0.76 0.74 = 1855 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm Measurement Error for a 10-dB Input power step (4) = 2500 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm = 50 MHz, -40C < TA < 25C (4) = 900 MHz, -40C < TA < 25C -40 dBm PIN -10 dBm (4) -7 -15 -13.4 -14.1 -13.4 -10.5 ST Temperature sensitivity -12.3 -13.1 -14.7 0.42 mdB/C -7.6 -15.9 0.63 -8.5 -18 = 3500 MHz, -40 dBm PIN -10 dBm = 3500 MHz, 25C < TA < 85C -40 dBm PIN -10 dBm (4) -0.2 -7.1 = 3000 MHz, -40 dBm PIN -10 dBm = 3000 MHz, 25C < TA < 85C -40 dBm PIN -10 dBm (4) -1.1 -6.7 = 2500 MHz,-40 dBm PIN -10 dBm = 2500 MHz, 25C < TA < 85C -40 dBm PIN -10 dBm (4) 1.2 -6.7 =1855 MHz, -40 dBm PIN -10 dBm =1855 MHz, 25C < TA < 85C -40 dBm PIN -10 dBm (4) 8 0.5 = 900 MHz, -40 dBm PIN -10 dBm = 900 MHz, 25C < TA < 85C -40 dBm PIN -10 dBm (4) 5.2 -11.7 = 50 MHz, -40 dBm PIN -10 dBm = 50 MHz, 25C < TA < 85C -40 dBm PIN -10 dBm (4) mdB/C -1.8 = 3500 MHz, -40 dBm PIN -10 dBm = 3500 MHz, -40C < TA < 25C -40 dBm PIN -10 dBm (4) 2.3 -4.1 = 3000 MHz, -40 dBm PIN -10 dBm = 3000 MHz, -40C < TA < 25C -40 dBm PIN -10 dBm (4) 1.5 -5.9 = 2500 MHz,-40 dBm PIN -10 dBm = 2500 MHz, -40C < TA < 25C -40 dBm PIN -10 dBm (4) 1 -6 = 1855 MHz, -40 dBm PIN -10 dBm = 1855 MHz, -40C < TA < 25C -40 dBm PIN -10 dBm (4) UNIT dB = 900 MHz,-40 dBm PIN -10 dBm Temperature sensitivity MAX (2) = 50 MHz, TA = -40C to 85C -40 dBm PIN -10 dBm = 50 MHz, -40 dBm PIN -10 dBm ST TYP (3) 1 -9.5 -21.2 2.5 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 9 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 2.7-V DC and AC Electrical Characteristics (continued) Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated.(1) PARAMETER TEST CONDITIONS MIN (2) = 50 MHz, PIN = -10 dBm = 50 MHz, -40C < TA < 25C PIN = -10 dBm (4) -15.8 ST Temperature sensitivity (4) -14.9 -14.5 ST Temperature sensitivity (4) -12 -12.4 -13.7 -14.6 -15.2 -6.5 -12.2 -16.5 -7.9 -13.5 -18.1 -9 -8.85 = 900 MHz -6.1 = 900 MHz, MIN at TA = -40C to 85C -9.3 = 1855 MHz = 1855 MHz, TA = -40C to 85C -5.5 -8.3 = 2500 MHz = 2500 MHz, TA = -40C to 85C -4.2 = 3000 MHz, TA = -40C to 85C -3.7 -5.4 = 3500 MHz = 3500 MHz, TA = -40C to 85C Submit Documentation Feedback dBm -6 = 3000 MHz 10 mdB/C -5.9 = 50 MHz, TA = -40C to 85C Maximum input power for ELC = 1 dB (4) -5.6 -10.8 = 50 MHz PMAX -5 -10 = 3500 MHz, PIN = -10 dBm = 3500 MHz, 25C < TA < 85C PIN = -10 dBm (4) -5.3 -9.4 = 3000 MHz, PIN = -10 dBm = 3000 MHz, 25C < TA < 85C PIN = -10 dBm (4) 5.3 -8.9 = 2500 MHz, PIN = -10 dBm = 2500 MHz, 25C < TA < 85C PIN = -10 dBm (4) 3.3 -3.4 = 1855 MHz, PIN = -10 dBm = 1855 MHz, 25C < TA < 85C PIN = -10 dBm (4) 1.3 -13 = 900 MHz, PIN = -10 dBm = 900 MHz, 25C < TA < 85C PIN = -10 dBm (4) mdB/C -4.9 = 50 MHz, PIN = -10 dBm = 50 MHz, 25C < TA < 85C PIN = -10 dBm (4) 2 -6.6 = 3500 MHz, PIN = -10 dBm = 3500 MHz, -40C < TA < 25C PIN = -10 dBm (4) 2.2 -7.4 = 3000 MHz, PIN = -10 dBm = 3000 MHz, -40C < TA < 25C PIN = -10 dBm (4) -0.75 -14.2 = 2500 MHz, PIN = -10 dBm = 2500 MHz, -40C < TA < 25C PIN = -10 dBm (4) UNIT -6 = 1855 MHz, PIN = -10 dBm = 1855 MHz, -40C < TA < 25C PIN = -10 dBm (4) MAX (2) -8.3 = 900 MHz, PIN = -10 dBm = 900 MHz, -40C < TA < 25C PIN = -10 dBm (4) TYP (3) -2.7 -7.2 Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 2.7-V DC and AC Electrical Characteristics (continued) Unless otherwise specified, all limits are ensured at TA = 25C, VDD = 2.7 V, RF input frequency = 1855-MHz continuous wave (CW), modulated.(1) PARAMETER TEST CONDITIONS MIN (2) = 50 MHz TYP (3) MAX (2) = 50 MHz, TA = -40C to 85C -38.9 = 900 MHz -44.2 = 900 MHz, MIN at TA = -40C to 85C -42.9 = 1855 MHz PMIN Minimum input power for ELC = 1 dB (4) -42.9 = 1855 MHz, TA = -40C to 85C -41.2 = 2500 MHz -40.4 = 2500 MHz, TA = -40C to 85C -38.4 = 3000 MHz, TA = -40C to 85C -35.8 = 3500 MHz -35.3 = 3500 MHz, TA = -40C to 85C -31.9 = 50 MHz 34.5 = 50 MHz, TA = -40C to 85C 31.5 = 900 MHz 38.1 = 900 MHz, MIN at TA = -40C to 85C 34.4 = 1855 MHz Dynamic range for ELC = 1 dB (4) dBm -38.6 = 3000 MHz DR UNIT -40.3 = 1855 MHz, TA = -40C to 85C 37.4 34 = 2500 MHz = 2500 MHz, TA = -40C to 85C 33.8 = 3000 MHz = 3000 MHz, TA = -40C to 85C 34.8 32.4 = 3500 MHz = 3500 MHz, TA = -40C to 85C dB 36.1 32.7 26.2 6.6 Timing Requirements MIN Turnon time, no signal at PIN, low-high transition EN, VOUT to 90% (1) tON (1) (2) 10 12 Rise time , PIN = no signal to 0 dBm, VOUT from 10% to 90% TA = -40C to 85C Fall time (2), PIN = no signal to 0 dBm, VOUT from 90% to 10% TA = -40C to 85C UNIT s 2 (2) Fall time (2), PIN = no signal to 0 dBm, VOUT from 90% to 10% tF MAX 8 Turnon time, no signal at PIN, low-high transition EN, VOUT to 90% (1) TA = -40C to 85C Rise time (2), PIN = no signal to 0 dBm, VOUT from 10% to 90% tR NOM 12 s 2 12 s All limits are ensured by design and measurements, which are performed on a limited number of samples. Limits represent the mean 3-sigma values. The typical value represents the statistical mean value. This parameter is ensured by design and/or characterization and is not tested in production. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 11 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 6.7 Typical Characteristics Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 10 10 85C 25C -40C SUPPLY CURRENT (mA) 8 8 ISUPPLY (mA) 85C 6 4 6 25C -40C 4 2 2 0 2.2 2.5 2.8 3.1 0 650 3.4 700 SUPPLY VOLTAGE (V) 900 45 -40C 45 LOGARITHMIC SLOPE (mV/dB) 2.0 1855 MHz 1.6 1.6 900 MHz VOUT (V) 850 Figure 2. Supply Current vs Enable Voltage 2.0 1.2 1.2 2500 MHz 0.8 0.8 3000 MHz 0.4 3500 MHz 0.4 0.0 25C 40 40 85C 3535 30 30 25 25 0.0 -60 -50 -40 -30 -20 -10 0 10M 10 RF INPUT POWER (dBm) 1G 10G Figure 4. Log Slope vs Frequency -38 -38 2.0 -40C VOUT (V) -42 -42 -46 -46 25C -50 -50 100M FREQUENCY (Hz) Figure 3. Output Voltage vs RF Input Power LOGARITHMIC INTERCEPT (dBm) 800 VENABLE (mV) Figure 1. Supply Current vs Supply Voltage RFIN = - 5 dBm 1.6 RFIN = -15 dBm 1.2 RFIN = -25 dBm 0.8 RFIN = -35 dBm 85C 0.4 RFIN = -45 dBm -54 -54 10M 10M 100M 1G 100M 1G FREQUENCY (Hz) 10G 10G 0.0 10M 100M 1G 10G FREQUENCY (Hz) Figure 5. Log Intercept vs Frequency 12 750 Submit Documentation Feedback Figure 6. Output Voltage vs Frequency Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Typical Characteristics (continued) 2.0 1.6 1.5 -40C 1.0 25C 1.2 1.0 0.5 0.0 85C 0.8 -0.5 0.6 -1.0 0.4 -45 -2.0 -25 -15 -5 1.5 -40C 1.4 1.0 -0.5 -1.0 -45 -2.5 -35 -25 -15 -5 5 RF INPUT POWER (dBm) 2.0 1.5 -40C 1.4 1.2 1.0 0.5 25C 1.0 0.0 85C 0.8 -0.5 0.6 -1.0 0.4 -1.5 85C 25C 0.2 -2.0 -40C -45 Figure 8. Mean Output Voltage and Log Conformance Error vs RF Input Power -25 -15 -5 2.5 1.8 2.0 1.6 1.5 0.5 85C 1.0 0.0 0.8 -0.5 0.6 -1.0 -1.5 85C 0.2 0.0 -55 5 -2.0 25C -40C -2.5 -45 RF INPUT POWER (dBm) -35 -25 -15 -5 5 RF INPUT POWER (dBm) 2500 MHz 2.0 2.5 1.8 2.0 1.6 1.5 1.4 1.0 -40C 25C 1.2 0.5 1.0 0.0 0.8 -0.5 0.6 -1.0 85C 0.4 -1.5 85C 0.2 -40C -2.0 25C LOG CONFORMANCE ERROR (dB) Figure 9. Mean Output Voltage and Log Conformance Error vs RF Input Power -2.5 -35 -25 -15 -5 5 Figure 10. Mean Output Voltage and Log Conformance Error vs RF Input Power 2.0 2.5 1.8 2.0 1.5 1.6 1.4 VOUT (V) 1855 MHz -45 1.0 -40C 25C 1.2 0.4 -2.5 -35 2.0 1.4 VOUT (V) 2.5 1.8 LOG CONFORMANCE ERROR (dB) 2.0 1.6 VOUT (V) -2.0 -40C 900 MHz Figure 7. Mean Output Voltage and Log Conformance Error vs RF Input Power VOUT (V) -1.5 85C 25C RF INPUT POWER (dBm) 0.0 -55 0.0 85C 0.6 0.0 -55 50 MHz 0.0 -55 0.5 0.8 0.2 5 1.0 25C 1.2 0.4 -2.5 -35 2.0 1.6 LOG CONFORMANCE ERROR (dB) 0.0 -55 -1.5 85C 25C -40C 0.2 2.5 1.8 -40C 1.2 1.0 25C 0.5 1.0 0.0 0.8 -0.5 -1.0 0.6 85C 0.4 0.0 -55 -1.5 85C 0.2 -40C -45 RF INPUT POWER (dBm) -2.0 25C LOG CONFORMANCE ERROR (dB) VOUT (V) 1.4 2.0 LOG CONFORMANCE ERROR (dB) 2.5 1.8 VOUT (V) 2.0 LOG CONFORMANCE ERROR (dB) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. -2.5 -35 -25 -15 -5 5 RF INPUT POWER (dBm) 3000 MHz 3500 MHz Figure 11. Mean Output Voltage and Log Conformance Error vs RF Input Power Figure 12. Mean Output Voltage and Log Conformance Error vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 13 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 50 MHz 900 MHz Figure 13. Log Conformance Error (Mean 3 Sigma) vs RF Input Power 1855 MHz 2500 MHz Figure 15. Log Conformance Error (Mean 3 Sigma) vs RF Input Power 3000 MHz Figure 16. Log Conformance Error (Mean 3 Sigma) vs RF Input Power 3500 MHz Figure 17. Log Conformance Error (Mean 3 Sigma) vs RF Input Power 14 Figure 14. Log Conformance Error (Mean 3 Sigma) vs RF Input Powert Figure 18. Log Conformance Error (Mean 3 Sigma) vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 1.5 1.5 1.0 TEMPERATURE DRIFT ERROR (dB) TEMPERATURE DRIFT ERROR (dB) 1.5 -40C 1.0 0.5 0.5 0.0 0.0 -0.5 -0.5 -1.0 85C -1.5 -1.0 -1.5 1.5 1.0 -40C 1.0 0.5 0.5 0.0 0.0 -0.5 -0.5 -1.0 85C -1.5 -1.0 -1.5 -55 -45 -35 -25 -15 -5 5 -55 -45 RF INPUT POWER (dBm) 50 MHz -15 -5 5 Figure 20. Mean Temperature Drift Error vs RF Input Power 1.5 1.5 TEMPERATURE DRIFT ERROR (dB) TEMPERATURE DRIFT ERROR (dB) -25 900 MHz Figure 19. Mean Temperature Drift Error vs Rf Input Power At 50 MHz 1.5 1.0 -40C 1.0 0.5 0.5 0.0 0.0 -0.5 -0.5 -1.0 85C -1.5 -1.0 1.5 1.0 1.0 -55 -45 -35 -25 -15 -5 -40C 0.5 0.5 0.0 0.0 -0.5 -0.5 -1.0 85C -1.5 -1.0 -1.5 -1.5 5 -55 -45 -35 -25 -15 -5 5 RF INPUT POWER (dBm) RF INPUT POWER (dBm) 2500 MHz 1855 MHz Figure 21. Mean Temperature Drift Error vs RF Input Power Figure 22. Mean Temperature Drift Error vs RF Input Powert 1.5 1.5 TEMPERATURE DRIFT ERROR (dB) TEMPERATURE DRIFT ERROR (dB) -35 RF INPUT POWER (dBm) 1.5 1.0 1.0 0.5 0.5 -40C 0.0 0.0 -0.5 -0.5 -1.0 85C -1.5 -1.0 -1.5 1.5 1.0 1.0 0.5 0.5 -40C 0.0 0.0 -0.5 -0.5 -1.0 -1.5 -1.0 85C -1.5 -55 -45 -35 -25 -15 -5 5 -55 RF INPUT POWER (dBm) -45 -35 -25 -15 -5 5 RF INPUT POWER (dBm) 3000 MHz 3500 MHz Figure 23. Mean Temperature Drift Error vs RF Input Power Figure 24. Mean Temperature Drift Error vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 15 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 50 MHz 900 MHz Figure 25. Temperature Drift Error (Mean 3 Sigma) vs RF Input Power 1855 MHz 2500 MHz Figure 27. Temperature Drift Error (Mean 3 Sigma) vs RF Input Power 3000 MHz Figure 28. Temperature Drift Error (Mean 3 Sigma) vs RF Input Power 3500 MHz Figure 29. Temperature Drift Error (Mean 3 Sigma) vs RF Input Power 16 Figure 26. Temperature Drift Error (Mean 3 Sigma) vs RF Input Power Figure 30. Temperature Drift Error (Mean 3 Sigma) vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 0.3 0.3 0.2 0.2 1 dB POWER STEP ERROR (dB) 1 dB POWER STEP ERROR (dB) 0.3 -40C 0.1 0.1 0.0 0.0 -0.1 -0.1 -0.2 25C 85C -0.3 -0.2 -0.3 -55 -45 -35 -25 -15 -5 0.3 0.2 0.2 0.1 0.1 0.0 0.0 -55 Figure 31. Error For 1-dB Input Power Step vs RF Input Power 5 Figure 32. Error For 1-dB Input Power Step vs RF Input Power 0.3 1 dB POWER STEP ERROR (dB) 0.3 1 dB POWER STEP ERROR (dB) -45 -35 -25 -15 -5 RF INPUT POWER (dBm) 900 MHz 50 MHz 0.3 0.2 -40C 0.2 0.1 0.1 0.0 0.0 85C 25C -0.1 -0.1 -0.2 -0.2 -0.3 0.3 0.2 -40C 0.2 0.1 0.1 0.0 0.0 25C -0.1 -0.1 -0.2 85C -0.3 -0.2 -0.3 -0.3 -55 -45 -35 -25 -15 -5 -55 5 RF INPUT POWER (dBm) -45 -35 -25 -15 -5 RF INPUT POWER (dBm) 5 2500 MHz 1855 MHz Figure 33. Error For 1-dB Input Power Step vs RF Input Power Figure 34. Error For 1-dB Input Power Step vs RF Input Power 0.3 0.3 0.3 0.2 0.2 1 dB POWER STEP ERROR (dB) 1 dB POWER STEP ERROR (dB) 85C -0.3 -0.2 RF INPUT POWER (dBm) -40C 0.1 0.1 0.0 0.0 -0.1 -0.1 -0.2 25C 85C -0.3 -0.2 -0.3 25C -0.1 -0.1 -0.2 -0.3 5 -40C -55 -45 -35 -25 -15 -5 RF INPUT POWER (dBm) 0.3 0.2 -40C 0.2 0.1 0.1 0.0 0.0 25C -0.1 -0.1 -0.2 85C -0.3 -0.2 -0.3 5 -55 -45 -35 -25 -15 -5 5 RF INPUT POWER (dBm) 3000 MHz 3500 MHz Figure 35. Error For 1-dB Input Power Step vs RFInput Power Figure 36. Error For 1-dB Input Power Step vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 17 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 1.00 1.00 10 dB POWER STEP ERROR (dB) 10 dB POWER STEP ERROR (dB) -40C 0.75 1.00 0.50 0.75 0.50 0.25 0.25 0.00 0.00 -0.25 -0.25 -0.50 -0.75 -0.50 -1.00 25C 85C -0.75 -1.00 0.75 1.00 0.50 0.75 0.50 0.25 0.25 0.00 0.00 -0.25 -0.25 -0.50 -0.75 -0.50 -1.00 -40C 25C 85C -0.75 -1.00 -60 -50 -40 -30 -20 -10 -60 0 -50 -40 -30 -20 -10 RF INPUT POWER (dBm) RF INPUT POWER (dBm) 900 MHz 50 MHz Figure 37. Error For 10-dB Input Power Step vs RF Input Power Figure 38. Error For 10-dB Input Power Step vs RF Input Power 1.00 1.00 25C 10 dB POWER STEP ERROR (dB) 10 dB POWER STEP ERROR (dB) -40C 0.75 1.00 0.50 0.75 0.50 0.25 0.25 0.00 0.00 -0.25 -0.25 -0.50 -0.75 -0.50 -1.00 25C 85C -0.75 -1.00 0.75 1.00 0.50 0.75 0.50 0.25 0.25 0.00 0.00 -0.25 -0.25 -0.50 -0.75 -0.50 -1.00 -40C 85C -0.75 -1.00 -60 -50 -40 -30 -20 -10 0 -60 -50 RF INPUT POWER (dBm) -30 -20 -10 0 2500 MHz Figure 39. Error For 10-dB Input Power Step vs RF Input Power Figure 40. Error For 10-dB Input Power Step vs RF Input Power 1.00 1.00 0.75 1.00 0.50 0.75 0.50 0.25 0.25 0.00 0.00 -0.25 -0.25 -0.50 -0.75 -0.50 -1.00 10 dB POWER STEP ERROR (dB) -40C 10 dB POWER STEP ERROR (dB) -40 RF INPUT POWER (dBm) 1855 MHz 25C 85C -0.75 -1.00 -40C 0.75 1.00 0.50 0.75 0.50 0.25 0.25 0.00 0.00 -0.25 -0.25 -0.50 -0.75 -0.50 -1.00 25C 85C -0.75 -1.00 -60 -50 -40 -30 -20 -10 0 -60 RF INPUT POWER (dBm) -50 -40 -30 -20 -10 0 RF INPUT POWER (dBm) 3000 MHz 3500 MHz Figure 41. Error For 10-dB Input Power Step vs RF Input Power 18 0 Figure 42. Error For 10-dB Input Power Step vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Typical Characteristics (continued) 20 20 15 20 10 15 10 55 0 0 -5 -10 -5 -15 -10 -20 TEMPERATURE SENSITIVITY (mdB/C) TEMPERATURE SENSITIVITY (mdB/C) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. -40C -30C 85C -15 -20 -55 -45 -35 -25 -15 -5 15 20 10 15 10 5 5 00 -5 -5 -10 -40C -30C -15 -10 -20 85C -15 -20 5 -55 RF INPUT POWER (dBm) -25 -15 -5 5 900 MHz Figure 43. Mean Temperature Sensitivity vs RF Input Power Figure 44. Mean Temperature Sensitivity vs RF Input Power 20 20 15 20 10 15 10 5 5 00 -5 -5 -10 -15 -10 -20 TEMPERATURE SENSITIVITY (mdB/C) TEMPERATURE SENSITIVITY (mdB/C) -35 RF INPUT POWER (dBm) 50 MHz -40C -30C 85C -15 15 20 10 15 10 5 5 00 -5 -5 -10 -15 -10 -20 -40C -30C 85C -15 -20 -20 -55 -45 -35 -25 -15 -5 -55 5 RF INPUT POWER (dBm) -45 -35 -25 -15 -5 5 RF INPUT POWER (dBm) 1855 MHz 2500 MHz Figure 45. Mean Temperature Sensitivity vs RF Input Power Figure 46. Mean Temperature Sensitivity vs RF Input Power 20 TEMPERATURE SENSITIVITY (mdB/C) 20 TEMPERATURE SENSITIVITY (mdB/C) -45 15 20 10 15 510 5 00 -5 -5 -10 -15 -10 -20 -40C -30C 85C -15 15 20 10 15 10 5 5 00 -5 -5 -10 -40C 85C -30C -15 -10 -20 -15 -20 -20 -55 -45 -35 -25 -15 -5 -55 5 -45 -35 -25 -15 -5 5 RF INPUT POWER (dBm) RF INPUT POWER (dBm) 3500 MHz 3000 MHz Figure 47. Mean Temperature Sensitivity vs RF Input Power Figure 48. Mean Temperature Sensitivity vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 19 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 50 MHz 900 MHz Figure 49. Temperature Sensitivity (Mean 3 Sigma) vs RF Input Power 1855 MHz 2500 MHz Figure 51. Temperature Sensitivity (Mean 3 Sigma) vs RF Input Power 3000 MHz Figure 52. Temperature Sensitivity (Mean 3 Sigma) vs RF Input Power 3500 MHz Figure 53. Temperature Sensitivity (Mean 3 Sigma) vs RF Input Power 20 Figure 50. Temperature Sensitivity (Mean 3 Sigma) vs RF Input Power Figure 54. Temperature Sensitivity (Mean 3 Sigma) vs RF Input Power Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 2.5 IS-95 1.6 2.0 1.41.8 1.6 1.21.4 1.2 1.01.0 0.8 0.80.6 0.4 0.60.2 0.0 0.4 WCDMA 64 CH CW 1.5 2.5 2.0 1.0 1.5 1.0 0.5 0.5 0.0 0.0 -0.5 -1.0 -0.5 -1.5 -2.0 -1.0 -2.5 -1.5 WCDMA 64 ch IS-95 0.2 -2.0 0.0 -45 -35 -25 -15 -5 RF INPUT POWER (dBm) 2.0 IS-95 1.6 2.0 1.41.8 1.6 1.21.4 1.2 1.0 1.00.8 0.6 0.8 0.4 0.2 0.6 0.0 1.5 2.5 1.0 2.0 1.5 1.0 0.5 0.5 0.0 0.0 -0.5 -1.0 -0.5 -1.5 -2.0 -1.0 -2.5 -1.5 WCDMA 64 CH CW 0.4 IS-95 -2.0 WCDMA 64 ch -2.5 0.0 5 -55 900 MHz -45 -35 -25 -15 -5 RF INPUT POWER (dBm) 5 1855 MHz Figure 55. Output Voltage and Log Conformance Error vs RR Input Power for Various Modulation Types Figure 56. Output Voltage and Log Conformance Error vs RF Input Power for Various Modulation Types 100 10 9 75 100 5075 50 25 25 0 0 -25 -25 -50 -75 -50 -100 R 8 VOLTAGE NOISE (V/ Hz) RF INPUT IMPEDANCE ( ) CW 0.2 -2.5 -55 2.5 2.0 1.8 VOUT (V) VOUT (V) 2.0 LOG CONFORMANCE ERROR (dB) CW LOG CONFORMANCE ERROR (dB) 2.0 1.8 X 7 6 5 4 3 2 -75 1 -100 100M 1G 0 10 10G 100 FREQUENCY (Hz) Figure 57. RF Input Impedance vs Frequency (Resistance and Reactance) 10k 100k 1M Figure 58. Output Noise Spectrum vs Frequency 80 270 100k GAIN 70 80 60 70 60 50 50 4040 30 3020 10 20 0 TRANS IMPEDANCE GAIN PSRR (dB) 1k FREQUENCY (Hz) 225 180 10k 135 PHASE 90 PHASE () 10M 45 1k 0 -45 10 100 100 0 10 100 1k 10k 100k FREQUENCY (Hz) 1M 1k 10k 100k 1M -90 10M FREQUENCY (Hz) Figure 59. Power Supply Rejection Ratio vs Frequency Figure 60. Output Amplifier Gain and Phase vs Frequency Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 21 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VDD = 2.7V, TA = 25C, measured on a limited number of samples. 60 25C 85C 50 -40C 40 30 20 10 0 0.0 0.5 1.0 1.5 2.0 25C SINKING OUTPUT CURRENT (mA) SOURCING OUTPUT CURRENT (mA) 60 2.5 -40C 40 30 20 10 0 0.0 3.0 0.5 1.0 1.5 VOUT (V) 2.5 3.0 Figure 62. Sinking Output Current vs Output Voltage 2.70 0.08 -40C -40C 2.68 2.70 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 2.0 VOUT (V) Figure 61. Sourcing Output Current vs Output Voltage 25C 2.66 2.68 85C 2.66 2.64 2.64 2.62 2.62 2.60 0 1 2 3 4 5 0.06 25C 85C 0.04 0.02 0.00 0 SOURCING CURRENT (mA) 1 2 3 4 5 SINKING CURRENT (mA) Figure 63. Output Voltage vs Sourcing Current 22 85C 50 Figure 64. Output Voltage vs Sinking Current Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 7 Detailed Description 7.1 Overview The LMV221 is a versatile logarithmic RF power detector suitable for use in power measurement systems. The LMV221 is particularly well suited for CDMA and UMTS applications. It produces a DC voltage that is a measure for the applied RF power. The core of the LMV221 is a progressive compression LOG detector consisting of four gain stages. Each of these saturating stages has a gain of approximately 10 dB and therefore achieves about 10 dB of the detector dynamic range. The five diode cells perform the actual detection and convert the RF signal to a DC current. This DC current is subsequently supplied to the transimpedance amplifier at the output, which converts it into an output voltage. In addition, the amplifier provides buffering of and applies filtering to the detector output signal. To prevent discharge of filtering capacitors between OUT and GND in shutdown, a switch is inserted at the amplifier input that opens in shutdown to realize a high impedance output of the device. 7.2 Functional Block Diagram B2 REF VDD A1 RTRANS en en EN C2 I/I VREF RFIN B1 GND C1 V-V 10 dB 10 dB 10 dB A2 OUT + + - en 10 dB RIN Copyright (c) 2016, Texas Instruments Incorporated 7.3 Feature Description 7.3.1 Characteristics of the LMV221 The LMV221 is a logarithmic RF power detector with approximately 40-dB dynamic range. This dynamic range plus its logarithmic behavior make the LMV221 ideal for various applications such as wireless transmit power control for CDMA and UMTS applications. The frequency range of the LMV221 is from 50 MHz to 3.5 GHz, which makes it suitable for various applications. The LMV221 transfer function is accurately temperature compensated. This makes the measurement accurate for a wide temperature range. Furthermore, the LMV221 can easily be connected to a directional coupler because of its 50- input termination. The output range is adjustable to fit the ADC input range. The detector can be switched into a power saving shutdown mode for use in pulsed conditions. 7.3.2 Accurate Power Measurement The power measurement accuracy achieved with a power detector is not only determined by the accuracy of the detector itself, but also by the way it is integrated into the application. In many applications some form of calibration is employed to improve the accuracy of the overall system beyond the intrinsic accuracy provided by the power detector. For example, for LOG-detectors calibration can be used to eliminate part to part spread of the LOG-slope and LOG-intercept from the overall power measurement system, thereby improving its power measurement accuracy. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 23 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Feature Description (continued) Calibration techniques can be used to improve the accuracy of a power measurement system beyond the intrinsic accuracy of the power detector itself. LOG-Conformance Error and Temperature Drift Error discuss power measurement systems using LOG-detectors, specifically the LMV221, but the more generic concepts can also be applied to other power detectors. Other factors influencing the power measurement accuracy, such as the resolution of the ADC reading the detector output signal, are not considered here because these factors are not fundamentally due to the power detector. 7.3.2.1 Concept of Power Measurements Power measurement systems generally consists of two clearly distinguishable parts with different functions: 1. A power detector device, that generates a DC output signal (voltage) in response to the power level of the (RF) signal applied to its input. 2. An estimator that converts the measured detector output signal into a (digital) numeric value representing the power level of the signal at the detector input. This conceptual configuration is shown in Figure 65. FEST MODEL PIN FDET VOUT PEST PARAMETERS Figure 65. Generic Concept of a Power Measurement System The core of the estimator is usually implemented as a software algorithm, receiving a digitized version of the detector output voltage. Its transfer FEST from detector output voltage to a numerical output must be equal to the inverse of the detector transfer FDET from (RF) input power to DC output voltage. If the power measurement system is ideal, that is, if no errors are introduced into the measurement result by the detector or the estimator, the measured power PEST(the output of the estimator) and the actual input power PIN must be identical. In that case, the measurement error E, the difference between the two, should be identically zero: E = PEST - PIN { 0 oe PEST = FEST[FDET(PIN)] = PIN -1 oe FEST(VOUT) = FDET(VOUT) (1) From Equation 1 it follows that one would design the FEST transfer function to be the inverse of the FDET transfer function. In practice the power measurement error is not zero, due to the following effects: * The detector transfer function is subject to various kinds of random errors that result in uncertainty in the detector output voltage; the detector transfer function is not exactly known. * The detector transfer function might be too complicated to be implemented in a practical estimator. The function of the estimator is then to estimate the input power PIN, that is, to produce an output PEST such that the power measurement error is, on average, minimized, based on the following information: 1. Measurement of the not-completely-accurate detector output voltage VOUT 2. Knowledge about the detector transfer function FDET, for example the shape of the transfer function, the types of errors present (part-to-part spread, temperature drift), and so forth. Obviously the total measurement accuracy can be optimized by minimizing the uncertainty in the detector output signal (select an accurate power detector), and by incorporating as much accurate information about the detector transfer function into the estimator as possible. 24 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Feature Description (continued) The knowledge about the detector transfer function is condensed into a mathematical model for the detector transfer function, consisting of: * A formula for the detector transfer function; and * Values for the parameters in this formula. The values for the parameters in the model can be obtained in various ways. They can be based on measurements of the detector transfer function in a precisely controlled environment (parameter extraction). If the parameter values are separately determined for each individual device, errors like part-to-part spread are eliminated from the measurement system. Errors may occur when the operating conditions of the detector (for example, the temperature) become significantly different from the operating conditions during calibration (for example, room temperature). Examples of simple estimators for power measurements that result in a number of commonly used metrics for the power measurement error are discussed in LOG-Conformance Error, Temperature Drift Error, Temperature Compensation and Temperature Drift Error. 7.3.2.2 LOG-Conformance Error Probably the simplest power measurement system that can be realized is obtained when the LOG-detector transfer function is modeled as a perfect linear-in-dB relationship between the input power and output voltage: VOUT,MOD = FDET,MOD(PIN) = KSLOPE(PIN PINTERCEPT) where * KSLOPE represents the LOG-slope and PINTERCEPT the LOG-intercept (2) The estimator based on Equation 2 implements the inverse of the model equation, that is: PEST = FEST(VOUT) = VOUT + PINTERCEPT KSLOPE (3) The resulting power measurement error, the LOG-conformance error, is thus equal to: ELCE = PEST - PIN = = VOUT - (PIN - PINTERCEPT ) KSLOPE VOUT - VOUT,MOD KSLOPE (4) The most important contributions to the LOG-conformance error are generally: * The deviation of the actual detector transfer function from an ideal logarithm (the transfer function is nonlinear in dB). * Drift of the detector transfer function over various environmental conditions, most importantly temperature; KSLOPE and PINTERCEPT are usually determined for room temperature only. * Part-to-part spread of the (room temperature) transfer function. The latter component is conveniently removed by means of calibration, that is, if the LOG slope and LOGintercept are determined for each individual detector device (at room temperature). This can be achieved by measurement of the detector output voltage (at room temperature) for a series of different power levels in the LOG-linear range of the detector transfer function. The slope and intercept can then be determined by means of linear regression. An example of this type of error and its relationship to the detector transfer function is shown in Figure 66. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 25 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 2.0 2.5 1.8 2.0 1.6 1.5 -40C VOUT (V) 1.4 1.2 1.0 0.5 25C 1.0 0.0 85C 0.8 -0.5 0.6 -1.0 0.4 0.2 0.0 -55 -1.5 85C 25C -2.0 -40C -45 LOG CONFORMANCE ERROR (dB) Feature Description (continued) -2.5 -35 -25 -15 -5 5 RF INPUT POWER (dBm) Figure 66. LOG-Conformance Error and LOG-Detector Transfer Function In the center of the dynamic range of the detector, the LOG-conformance error is small, especially at room temperature; in this region the transfer function closely follows the linear-in-dB relationship while KSLOPE and PINTERCEPT are determined based on room temperature measurements. At the temperature extremes the error in the center of the range is slightly larger due to the temperature drift of the detector transfer function. The error rapidly increases toward the top and bottom end of the detector's dynamic range; here the detector saturates and its transfer function starts to deviate significantly from the ideal LOG-linear model. The detector dynamic range is usually defined as the power range for which the LOG conformance error is smaller than a specified amount. Often an error of 1 dB is used as a criterion. 7.3.2.3 Temperature Drift Error A more accurate power measurement system can be obtained if the first error contribution, due to the deviation from the ideal LOG-linear model, is eliminated. This is achieved if the actual measured detector transfer function at room temperature is used as a model for the detector, instead of the ideal LOG-linear transfer function used in the previous section. The formula used for such a detector is: VOUT,MOD = FDET(PIN,TO) where * TO represents the temperature during calibration (room temperature). (5) The transfer function of the corresponding estimator is thus the inverse of this: -1 PEST = FDET[VOUT(T),T0] where * VOUT(T) represents the measured detector output voltage at the operating temperature T. (6) The resulting measurement error is only due to drift of the detector transfer function over temperature and can be expressed as in Equation 7: -1 EDRIFT (T,T0) = PEST - PIN = FDET[VOUT(T),T0] - PIN -1 -1 = FDET[VOUT(T),T0] - FDET[VOUT(T),T)] (7) 26 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Feature Description (continued) Unfortunately, the (numeric) inverse of the detector transfer function at different temperatures makes this expression rather impractical. However, because the drift error is usually small VOUT(T) is only slightly different from VOUT(TO). This means that Equation 8 can be applied: EDRIFT(T,T0) | EDRIFT(T0,T0) w -1 -1 + (T - T0) w7 {FDET[VOUT(T),T0] - FDET[VOUT(T),T]} (8) This expression is easily simplified by taking the following considerations into account: * The drift error at the calibration temperature E(TO,TO) equals zero (by definition). * The estimator transfer FDET(VOUT,TO) is not a function of temperature; the estimator output changes over temperature only due to the temperature dependence of VOUT. * The actual detector input power PIN is not temperature dependent (in the context of this expression). * The derivative of the estimator transfer function to VOUT equals approximately 1/KSLOPE in the LOG-linear region of the detector transfer function (the region of interest). Taking into account the preceding considerations, the simplified expression would be: w -1 FDET[VOUT(T),T0] w7 -1 w VOUT(T) w = (T T0) FDET[VOUT(T),T0] wVOUT w7 EDRIFT (T,T0) | (T T0) | VOUT(T) VOUT(T0) KSLOPE (9) Equation 9 is very similar to Equation 4 determined previously. The only difference is that instead of the output of the ideal LOG-linear model, the actual detector output voltage at the calibration temperature is now subtracted from the detector output voltage at the operating temperature. Figure 67 depicts an example of the drift error. TEMPERATURE DRIFT ERROR (dB) 1.5 1.5 1.0 -40C 1.0 0.5 0.5 0.0 0.0 -0.5 -0.5 -1.0 85C -1.5 -1.0 -1.5 -55 -45 -35 -25 -15 -5 5 RF INPUT POWER (dBm) Figure 67. Temperature Drift Error of the LMV221 at = 1855 MHz In agreement with the definition, the temperature drift error is zero at the calibration temperature. Further, the main difference with the LOG-conformance error is observed at the top and bottom end of the detection range; instead of a rapid increase the drift error settles to a small value at high and low input power levels due to the fact that the detector saturation levels are relatively temperature independent. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 27 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Feature Description (continued) In a practical application it may not be possible to use the exact inverse detector transfer function as the algorithm for the estimator. For example, it may require too much memory or too much factory calibration time. However, using the ideal LOG-linear model in combination with a few extra data points at the top and bottom end of the detection range -- where the deviation is largest -- can already significantly reduce the power measurement error. 7.3.2.3.1 Temperature Compensation A further reduction of the power measurement error is possible if the operating temperature is measured in the application. For this purpose, the detector model used by the estimator should be extended to cover the temperature dependency of the detector. Because the detector transfer function is generally a smooth function of temperature (the output voltage changes gradually over temperature), the temperature is in most cases adequately modeled by a first-order or secondorder polynomial (see Equation 10). VOUT,MOD = FDET(PIN,T0)[1 + (T-T0)TC1(PIN) + (T-T0)2TC2(PIN) + O(T3)] (10) The required temperature dependence of the estimator, to compensate for the detector temperature dependence can be approximated similarly: -1 PEST = FDET[VOUT(T),T0]{1 + (T-T0)S1[VOUT(T)] + + (T-T0)2S2[VOUT(T)] + O(T3)} -1 | FDET[VOUT(T),T0]{1 + (T-T0)S1[VOUT(T)]} (11) The last approximation results from the fact that a first-order temperature compensation is usually sufficiently accurate. For second and higher-order compensation a similar approach can be followed. Ideally, the temperature drift could be completely eliminated if the measurement system is calibrated at various temperatures and input power levels to determine the temperature sensitivity S1. In a practical application, however, that is usually not possible due to the associated high costs. The alternative is to use the average temperature drift in the estimator, instead of the temperature sensitivity of each device individually. In this way it is possible to eliminate the systematic (reproducible) component of the temperature drift without the need for calibration at different temperatures during manufacturing. What remains is the random temperature drift, which differs from device to device. (see Figure 68). The graph at the left of Figure 68 schematically represents the behavior of the drift error versus temperature at a certain input power level for a large number of devices. Figure 68. Elimination of the Systematic Component from the Temperature Drift 28 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Feature Description (continued) The mean drift error represents the reproducible (systematic) part of the error, while the mean 3 sigma limits represent the combined systematic plus random error component. Obviously the drift error must be zero at calibration temperature T0. If the systematic component of the drift error is included in the estimator, the total drift error becomes equal to only the random component, as shown in the graph at the right of Figure 68. A significant reduction of the temperature drift error can be achieved in this way only if: * The systematic component is significantly larger than the random error component (otherwise the difference is negligible). * The operating temperature is measured with sufficient accuracy. It is essential for the effectiveness of the temperature compensation to assign the appropriate value to the temperature sensitivity S1. Two different methods can be followed to determine this parameter: 1. Determination of a single value to be used over the entire operating temperature range. 2. Division of the operating temperature range in segments and use of separate values for each of the segments. For the first method, the accuracy of the extracted temperature sensitivity increases when the number of measurement temperatures increases. Linear regression to temperature can then be used to determine the two parameters of the linear model for the temperature drift error: the first order temperature sensitivity S1 and the best-fit (room temperature) value for the power estimate at T0: FDET[VOUT(T),T0]. Note that to achieve an overall (over all temperatures) minimum error, the room temperature drift error in the model can be non-zero at the calibration temperature (which is not in agreement with the strict definition). The second method does not have this drawback but is more complex. In fact, segmentation of the temperature range is a form of higher-order temperature compensation using only a first-order model for the different segments: one for temperatures below 25C, and one for temperatures above 25C. The mean (or typical) temperature sensitivity is the value to be used for compensation of the systematic drift error component. Figure 69 and Figure 70 show the temperature drift error without and with temperature compensation using two segments. With compensation the systematic component is completely eliminated; the remaining random error component is centered around zero. Note that the random component is slightly larger at -40C than at +85C. Figure 69. Temperature Drift Error without Temperature Compensation Figure 70. Temperature Drift Error With Temperature Compensation In a practical power measurement system, temperature compensation is usually only applied to a small power range around the maximum power level for two reasons: 1. The various communication standards require the highest accuracy in this range to limit interference. 2. The temperature sensitivity itself is a function of the power level it becomes impractical to store a large number of different temperature sensitivity values for different power levels. The 2.7-V DC and AC Electrical Characteristics specifies the temperature sensitivity for the aforementioned two segments at an input power level of -10 dBm (near the top-end of the detector dynamic range). The typical value represents the mean which is to be used for calibration. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 29 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Feature Description (continued) 7.3.2.3.2 Differential Power Errors Many third-generation communication systems contain a power control loop through the base station and mobile unit that require frequent updates to the transmit-power level by a small amount (typically 1 dB). For such applications it is important that the actual change of the transmit power is sufficiently close to the requested power change. The error metrics in this data sheet that describe the accuracy of the detector for a change in the input power are E1 dB (for a 1-dB change in the input power) and E10 dB (for a 10-dB step, or ten consecutive steps of 1 dB). Because it can be assumed that the temperature does not change during the power step the differential error equals the difference of the drift error at the two involved power levels: E1dB(PIN,T)= EDRIFT(PIN+1dB,T) - EDRIFT(PIN,T) E10dB(PIN,T)= EDRIFT(PIN+10dB,T) - EDRIFT(PIN,T) (12) NOTE The step error increases significantly when one (or both) power levels in the above expression are outside the detector dynamic range. For E10 dB this occurs when PIN is less than 10 dB below the maximum input power of the dynamic range, PMAX. 7.4 Device Functional Modes 7.4.1 Shutdown To save power, the LMV221 can be brought into a low power-shutdown mode. The device is active for EN = high (VEN > 1.1 V) and in the low power-shutdown mode for EN = low (VEN < 0.6 V). In this state the output of the LMV221 is switched to a high impedance mode. Using the shutdown function, care must be taken not to exceed the absolute maximum ratings. Forcing a voltage to the enable input that is 400 mV higher than VDD or 400 mV lower than GND damages the device, and further operation is not ensured. The absolute maximum ratings can also be exceeded when the enable EN is switched to high (from shutdown to active mode) while the supply voltage is low (off). This must be prevented at all times. A possible solution to protect the device is to add a resistor of 100 k in series with the enable input. 7.4.1.1 Output Behavior in Shutdown In order to save power, the LMV221 can be used in pulsed mode so that it is active to perform the power measurement only during a fraction of the time. During the remaining time the device is in low-power shutdown. Applications using this approach usually require that the output value is available at all times, including when the LMV221 is in shutdown. The settling time in active mode, however, must not become excessively large. This can be achieved by the combination of the LMV221 and a low pass output filter (see Figure 75). In active mode, the filter capacitor CS is charged to the output voltage of the LMV221, which in this mode has a low output impedance to enable fast settling. During shutdown mode, the capacitor should preserve this voltage. Discharge of CS through any current path must therefore be avoided in shutdown. The output impedance of the LMV221 becomes high in shutdown, thus, the discharge current cannot flow from the capacitor top plate, through RS and the device OUT pin to GND. This is detected by the internal shutdown mechanism of the output amplifier and by the switch depicted in Figure 79. Additionally, the ADC input impedance must be high to prevent a possible discharge path through the ADC. 30 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI's customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality. 8.1 Application Information 8.1.1 Functionality and Applications of RF Power Detectors 8.1.1.1 Functionality of RF Power Detectors An RF power detector is a device that produces a DC output voltage in response to the RF power level of the signal applied to its input. A wide variety of power detectors can be distinguished, each having certain properties that suit a particular application. This section provides an overview of the key characteristics of power detectors, and discusses the most important types of power detectors. The functional behavior of the LMV221 is discussed in detail. 8.1.1.1.1 Key Characteristics of RF Power Detectors Power detectors are used to accurately measure the power of a signal inside the application. The attainable accuracy of the measurement is therefore dependent upon the accuracy and predictability of the detector transfer function from the RF input power to the DC output voltage. Certain key characteristics determine the accuracy of RF detectors and they are classified accordingly: * Temperature Stability * Dynamic Range * Waveform Dependency * Transfer Shape Generally, the transfer function of RF power detectors is slightly temperature dependent. This temperature drift reduces the accuracy of the power measurement, because most applications are calibrated at room temperature. In such systems, the temperature drift significantly contributes to the overall system power measurement error. The temperature stability of the transfer function differs for the various types of power detectors. Generally, power detectors that contain only one or few semiconductor devices (diodes, transistors) operating at RF frequencies attain the best temperature stability. The dynamic range of a power detector is the input power range for which it creates an accurately reproducible output signal. What is considered accurate is determined by the applied criterion for the detector accuracy; the detector dynamic range is thus always associated with certain power measurement accuracy. This accuracy is usually expressed as the deviation of its transfer function from a certain predefined relationship, such as linear in dB for LOG detectors and square-law transfer (from input RF voltage to DC output voltage) for mean-square detectors. For LOG-detectors, the dynamic range is often specified as the power range for which its transfer function follows the ideal linear-in-dB relationship with an error smaller than or equal to 1 dB. Again, the attainable dynamic range differs considerably for the various types of power detectors. According to its definition, the average power is a metric for the average energy content of a signal and is not directly a function of the shape of the signal in time. In other words, the power contained in a 0-dBm sine wave is identical to the power contained in a 0-dBm square wave or a 0-dBm WCDMA signal; all these signals have the same average power. Depending on the internal detection mechanism, though, power detectors may produce a slightly different output signal in response to the aforementioned waveforms, even though their average power level is the same. This is due to the fact that not all power detectors strictly implement the definition formula for signal power, being the mean of the square of the signal. Most types of detectors perform some mixture of peak detection and average power detection. A waveform independent detector response is often desired in applications that exhibit a large variety of waveforms, such that separate calibration for each waveform becomes impractical. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 31 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Application Information (continued) The shape of the detector transfer function from the RF input power to the DC output voltage determines the required resolution of the ADC connected to it. The overall power measurement error is the combination of the error introduced by the detector, and the quantization error contributed by the ADC. The impact of the quantization error on the overall transfer's accuracy is highly dependent on the detector transfer shape, as shown in Figure 71 and Figure 72. 2 2 AV1 VOUT (V) VOUT (V) AV AV AV2 0 -60 AP AP RF INPUT POWER (dBm) 0 0 -60 Figure 71. Convex Detector Transfer Function AP AP RF INPUT POWER (dBm) 0 Figure 72. Linear Transfer Function Figure 71 and Figure 72 shows two different representations of the detector transfer function. In both Figure 71 and Figure 72 the input power along the horizontal axis is displayed in dBm because most applications specify power accuracy requirements in dBm (or dB). Figure 71 shows a convex detector transfer function, while the transfer function on the right hand side is linear (in dB). The slope of the detector transfer function -- the detector conversion gain - is of key importance for the impact of the quantization error on the total measurement error. If the detector transfer function slope is low, a change, P, in the input power results only in a small change of the detector output voltage, such that the quantization error is relatively large. On the other hand, if the detector transfer function slope is high, the output voltage change for the same input power change will be large, such that the quantization error is small. Figure 72 has a very low slope at low input power levels, resulting in a relatively large quantization error. Therefore, to achieve accurate power measurement in this region, a highresolution ADC is required. On the other hand, for high input power levels the quantization error are very small due to the steep slope of the curve in this region. For accurate power measurement in this region, a much lower ADC resolution is sufficient. Figure 71 has a constant slope over the power range of interest, such that the required ADC resolution for a certain measurement accuracy is constant. For this reason, the LOG-linear curve in Figure 71 generally leads to the lowest ADC resolution requirements for certain power measurement accuracy. 32 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Application Information (continued) 8.1.1.1.2 Types of RF Power Detectors Three different detector types are distinguished based on the four characteristics previously discussed: * Diode Detector * (Root) Mean Square (R)MS) Detector * Logarithmic Detectors 8.1.1.1.2.1 Diode Detector A diode is one of the simplest types of RF detectors. As depicted in Figure 73, the diode converts the RF input voltage into a rectified current. This unidirectional current charges the capacitor. The RC time constant of the resistor and the capacitor determines the amount of filtering applied to the rectified (detected) signal. Z0 D VREF RS CS VOUT Copyright (c) 2016, Texas Instruments Incorporated Figure 73. Diode Detector The advantages and disadvantages can be summarized as follows: * The temperature stability of the diode detectors is generally very good because they contain only one semiconductor device that operates at RF frequencies. * The dynamic range of diode detectors is poor. The conversion gain from the RF input power to the output voltage quickly drops to very low levels when the input power decreases. Typically a dynamic range of 20 dB to 25 dB can be achieved with this type of detector. * The response of diode detectors is waveform dependent. As a consequence of this dependency, for example, its output voltage for a 0-dBm WCDMA signal is different than for a 0-dBm unmodulated carrier. This is due to the fact that the diode measures peak power instead of average power. The relation between peak power and average power is dependent on the wave shape. * The transfer shape of diode detectors puts high requirements on the resolution of the ADC that reads their output voltage. Especially at low input power levels a very high ADC resolution is required to achieve sufficient power measurement accuracy (See Figure 71). 8.1.1.1.2.2 (Root) Mean Square (R)MS) Detector This type of detector is particularly suited for the power measurements of RF modulated signals that exhibits large peak-to-average power ratio variations. This is because its operation is based on direct determination of the average power and not - like the diode detector - of the peak power. The advantages and disadvantages can be summarized as follows: * The temperature stability of (R)MS detectors is almost as good as the temperature stability of the diode detector; only a small part of the circuit operates at RF frequencies, while the rest of the circuit operates at low frequencies. * The dynamic range of (R)MS detectors is limited. The lower end of the dynamic range is limited by internal device offsets. * The response of (R)MS detectors is highly waveform independent. This is a key advantage compared to other types of detectors in applications that employ signals with high peak-to-average power variations. For example, the (R)MS detector response to a 0-dBm WCDMA signal and a 0-dBm unmodulated carrier is essentially equal. * The transfer shape of R(MS) detectors has many similarities with the diode detector and is therefore subject to similar disadvantages with respect to the ADC resolution requirements (see Figure 72). Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 33 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Application Information (continued) 8.1.1.1.2.3 Logarithmic Detectors The transfer function of a logarithmic detector has a linear in dB response, which means that the output voltage changes linearly with the RF power in dBm. This is convenient because most communication standards specify transmit power levels in dBm as well. The advantages and disadvantages can be summarized as follows: * The temperature stability of the LOG detector transfer function is generally not as good as the stability of diode and R(MS) detectors. This is because a significant part of the circuit operates at RF frequencies. * The dynamic range of LOG detectors is usually much larger than that of other types of detectors. * Because LOG detectors perform a kind of peak detection their response is wave form dependent, similar to diode detectors. * The transfer shape of LOG detectors puts the lowest possible requirements on the ADC resolution (See Figure 72). 8.2 Typical Applications RF power detectors can be used in a wide variety of applications. Figure 74 shows the LMV221 in a transmit power-control system, and Figure 82 measures the voltage standing wave ratio (VSWR). 8.2.1 Application With Transmit Power Control Loop The key benefit of a transmit power control loop circuit is that it makes the transmit power insensitive to changes in the power amplifier (PA) gain control function, such as changes due to temperature drift. When a control loop is used, the transfer function of the PA is eliminated from the overall transfer function. Instead, the overall transfer function is determined by the power detector. The overall transfer function accuracy depends thus on the RF detector accuracy. The LMV221 is especially suited for this application, due to the accurate temperature stability of its transfer function. Figure 74 shows a block diagram of a typical transmit power control system. The output power of the PA is measured by the LMV221 through a directional coupler. The measured output voltage of the LMV221 is filtered and subsequently digitized by the ADC inside the baseband chip. The baseband adjusts the PA output power level by changing the gain control signal of the RF VGA accordingly. With an input impedance of 50 , the LMV221 can be directly connected to a 30-dB directional coupler without the need for an additional external attenuator. The setup can be adjusted to various PA output ranges by selection of a directional coupler with the appropriate coupling factor. B A S E B A N D RF COUPLER PA VGA ANTENNA 50 : GAIN RS OUT RFIN ADC CS LOGIC LMV221 EN GND Copyright (c) 2016, Texas Instruments Incorporated Figure 74. Transmit Power Control System 34 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Typical Applications (continued) 8.2.1.1 Design Requirements Some of the design requirements for this logarithmic RMS power detector include: Table 1. Design Parameters DESIGN PARAMETER EXAMPLE VALUE Supply voltage 2.7 V RF input frequency (unmodulated continuous wave) 1855 MHz Minimum input power for ELC = 1 dB -42.9 dBm Maximum input power for ELC = 1 dB -5.5 dBm Maximum output voltage, PIN = -5 dBm 1.61 V 8.2.1.2 Detailed Design Procedure 8.2.1.2.1 Detector Interfacing For optimal performance of the LMV221 device, it is important that all its pins are connected to the surrounding circuitry in the appropriate way. Starting from the Functional Block Diagram the function of each pin is elaborated in the following sections. The details of the electrical interfacing are separately discussed for each pin. Output filtering options and the differences between single ended and differential interfacing with an ADC are also discussed in detail in the following subsections. 8.2.1.2.1.1 RF Input RF parts typically use a characteristic impedance of 50 . To comply with this standard the LMV221 has an input impedance of 50 . Using a characteristic impedance other then 50 causes a shift of the logarithmic intercept with respect to the value given in the 2.7-V DC and AC Electrical Characteristics. This intercept shift can be calculated according to Equation 13. 2 RSOURCE * PINT-SHIFT = 10 LOG (c) RSOURCE + 50 (13) The intercept shifts to higher power levels for RSOURCE > 50 , and shifts to lower power levels for RSOURCE < 50 . 8.2.1.2.1.2 Output and Reference The possible filtering techniques that can be applied to reduce ripple in the detector output voltage are discussed in Filtering. In addition, two different topologies to connect the LMV221 to an ADC are elaborated. 8.2.1.2.1.2.1 Filtering The output voltage of the LMV221 is a measure for the applied RF signal on the RF input pin. Usually, the applied RF signal contains AM modulation that causes low frequency ripple in the detector output voltage. CDMA signals, for instance, contain a large amount of amplitude variations. Filtering of the output signal can be used to eliminate this ripple. The filtering can either be achieved by a low pass output filter or a low pass feedback filter. Those two techniques are shown in Figure 75 and Figure 76. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 35 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com VDD VDD RS RFIN 2 1 6 RS OUT RFIN + 2 1 6 OUT + ADC EN 4 5 3 ADC RP CS LMV221 REF CS LMV221 EN - 4 5 3 GND REF - GND Copyright (c) 2016, Texas Instruments Incorporated Copyright (c) 2016, Texas Instruments Incorporated Figure 75. Low Pass Output Filter Figure 76. Low Pass Feedback Filter Depending on the system requirements one of the these filtering techniques can be selected. The low pass output filter has the advantage that it preserves the output voltage when the LMV221 is brought into shutdown. This is elaborated in Output Behavior in Shutdown. In the feedback filter, resistor RP discharges capacitor CP in shutdown and therefore changes the output voltage of the device. A disadvantage of the low pass output filter is that the series resistor RS limits the output drive capability. This may cause inaccuracies in the voltage read by an ADC when the ADC input impedance is not significantly larger than RS. In that case, the current flowing through the ADC input induces an error voltage across filter resistor RS. The low pass feedback filter does not have this disadvantage. NOTE Note that adding an external resistor between OUT and REF reduces the transfer gain (LOG-slope and LOG-intercept) of the device. The internal feedback resistor sets the gain of the transimpedance amplifier. The filtering of the low pass output filter is achieved by resistor RS and capacitor CS. The -3 dB bandwidth of this filter can then be calculated by: -3 dB = 1 / 2RSCS. The bandwidth of the low pass feedback filter is determined by external resistor RP in parallel with the internal resistor RTRANS, and external capacitor CP in parallel with internal capacitor CTRANS (see Figure 79). The -3 dB bandwidth of the feedback filter can be calculated by -3 dB = 1 / 2 (RP//RTRANS) (CP + CTRANS). The bandwidth set by the internal resistor and capacitor (when no external components are connected between OUT and REF) equals -3 dB = 1 / 2 RTRANS CTRANS = 450 kHz. 8.2.1.2.1.3 Interface to the ADC The LMV221 can be connected to the ADC with a single-ended or a differential topology. The single ended topology connects the output of the LMV221 to the input of the ADC and the reference pin is not connected. In a differential topology, both the output and the reference pins of the LMV221 are connected to the ADC. The topologies are depicted in Figure 77 and Figure 78. VDD VDD RS RFIN 2 1 6 OUT RFIN + 2 1 6 OUT + ADC ADC RP LMV221 EN 4 3 5 CP REF RP LMV221 EN - GND 4 3 5 REF CP - GND Copyright (c) 2016, Texas Instruments Incorporated Figure 77. Single-Ended Application Copyright (c) 2016, Texas Instruments Incorporated Figure 78. Differential Application The differential topology has the advantage that it is compensated for temperature drift of the internal reference voltage. This can be explained by looking at the transimpedance amplifier of the LMV221 (Figure 79). 36 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 REF CTRANS RTRANS IDET - VREF OUT + + - Copyright (c) 2016, Texas Instruments Incorporated Figure 79. Output Stage of the LMV221 Equation 14 shows that the output of the amplifier is set by the detection current IDET multiplied by the resistor RTRANS plus the reference voltage VREF: VOUT = IDET RTRANS + VREF where * IDET represents the detector current that is proportional to the RF input power. (14) Equation 14 shows that temperature variations in VREF are also present in the output VOUT. In case of a single ended topology the output is the only pin that is connected to the ADC. The ADC voltage for single ended is thus: VADC = IDET RTRANS + VREF (15) A differential topology also connects the reference pin, which is the value of reference voltage VREF. The ADC reads VOUT - VREF: VADC = VOUT - VREF = IDET RTRANS (16) Equation 16 no longer contains the reference voltage VREF anymore. Temperature variations in this reference voltage are therefore not measured by the ADC. 8.2.1.3 Application Curves 2.0 2.0 1855 MHz 2.0 1.2 1.2 50 MHz 2500 MHz 0.8 0.8 RFIN = -15 dBm 1.2 RFIN = -25 dBm 0.8 RFIN = -35 dBm 3000 MHz 0.4 0.4 0.4 0.0 RFIN = -45 dBm 4000 MHz 0.0 -60 1.6 900 MHz VOUT (V) VOUT (V) 1.6 1.6 RFIN = - 5 dBm -50 -40 -30 -20 -10 0 10 0.0 10M 100M 1G 10G FREQUENCY (Hz) RF INPUT POWER (dBm) Figure 80. Output Voltage vs RF Input Power Figure 81. Output Voltage vs Frequency Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 37 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 8.2.2 Application With Voltage Standing Wave Ratio (VSWR) Measurement Transmission in RF systems requires matched termination by the proper characteristic impedance at the transmitter and receiver side of the link. In wireless transmission systems, however, matched termination of the antenna can rarely be achieved. The part of the transmitted power that is reflected at the antenna bounces back toward the PA and may cause standing waves in the transmission line between the PA and the antenna. These standing waves can attain unacceptable levels that may damage the PA. A VSWR measurement is used to detect such an occasion. It acts as an alarm function to prevent damage to the transmitter. VSWR is defined as the ratio of the maximum voltage divided by the minimum voltage at a certain point on the transmission line: 1+ _*_ VSWR = 1 - _*_ where * = VREFLECTED / VFORWARD denotes the reflection coefficient. (17) This means that to determine the VSWR, both the forward (transmitted) and the reflected power levels must be measured. This can be accomplished by using two LMV221 RF power detectors according to Figure 82. A directional coupler is used to separate the forward and reflected power waves on the transmission line between the PA and the antenna. One secondary output of the coupler provides a signal proportional to the forward power wave, the other secondary output provides a signal proportional to the reflected power wave. The outputs of both RF detectors that measure these signals are connected to a microcontroller or baseband that calculates the VSWR from the detector output signals. RF COUPLER ANTENNA PA MICRO CONTROLLER VDD RFIN EN 1 2 REVERSE POWER 6 OUT ADC1 LMV221 4 3 RP1 5 CP1 REF GND RFIN 2 TRANSMITTED POWER EN 1 6 OUT ADC2 LMV221 4 3 5 RP2 CP2 REF GND Copyright (c) 2016, Texas Instruments Incorporated Figure 82. VSWR Application 9 Power Supply Recommendations The LMV221 is designed to operate from an input voltage supply range from 2.7 V to 3.3 V. This input voltage must be well regulated. Enable voltage levels lower than 400 mV below GND could lead to incorrect operation of the device. Also, the resistance of the input supply rail must be low enough to ensure correct operation of the device. 38 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 10 Layout 10.1 Layout Guidelines As with any other RF device, careful attention must be paid to the board layout. If the board layout is not properly designed, unwanted signals can easily be detected or interference picked up. Electrical signals (voltages and currents) need a finite time to travel through a trace or transmission line. RF voltage levels at the generator side and at the detector side can therefore be different. This is not only true for the RF strip line, but for all traces on the PCB. Signals at different locations or traces on the PCB are in a different phase of the RF frequency cycle. Phase differences in, for example, the voltage across neighboring lines, may result in crosstalk between lines due to parasitic capacitive, or inductive coupling. This crosstalk is further enhanced by the fact that all traces on the PCB are susceptible to resonance. The resonance frequency depends on the trace geometry. Traces are particularly sensitive to interference when the length of the trace corresponds to a quarter of the wavelength of the interfering signal or a multiple thereof. 10.1.1 Supply Lines Because the PSRR of the LMV221 is finite, variations of the supply can result in some variation at the output. This can be caused among others by RF injection from other parts of the circuitry or the on/off switching of the PA. 10.1.1.1 Positive Supply (VDD) In order to minimize the injection of RF interference into the LMV221 through the supply lines, the phase difference between the PCB traces connecting to VDD and GND must be minimized. A suitable way to achieve this is to short both connections for RF. This can be done by placing a small decoupling capacitor between the VDD and GND. It must be placed as close to the device VDD and GND pins as possible as shown in Figure 85. Be aware that the resonance frequency of the capacitor itself must be above the highest RF frequency used in the application, because the capacitor acts as an inductor above its resonance frequency. Low frequency-supply voltage variations due to PA switching might result in a ripple at the output voltage. The LMV221 has a PSRR of 60 dB for low frequencies. 10.1.1.2 Ground (GND) The LMV221 must have a ground plane free of noise and other disturbing signals. It is important to separate the RF ground return path from the other grounds. This is due to the fact that the RF input handles large voltage swings. A power level of 0 dBm causes a voltage swing larger than 0.6 VPP over the internal 50- input resistor, resulting in a significant RF return current toward the source. Therefore, TI recommends that the RF ground return path not be used for other circuits in the design. The RF path must be routed directly back to the source without loops. 10.1.2 RF Input Interface The LMV221 is designed to be used in RF applications having a characteristic impedance of 50 . To achieve this impedance, the input of the LMV221 must be connected via a 50- transmission line. Transmission lines can be easily created on PCBs using microstrip or (grounded) coplanar waveguide (GCPW) configurations. For more details about designing microstrip or GCPW transmission lines, TI recommends a microwave designer handbook is recommended. 10.1.3 Microstrip Configuration One way to create a transmission line is to use a microstrip configuration. A cross section of the configuration is shown in Figure 83, assuming a two-layer PCB. Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 39 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com Layout Guidelines (continued) METAL CONDUCTOR W FR4 PCB H GROUND PLANE Figure 83. Microstrip Configuration A conductor (trace) is placed on the topside of a PCB. The bottom side of the PCB has a fully copper ground plane. The characteristic impedance of the microstrip transmission line is a function of the width W, height H, and the dielectric constant r. Characteristics such as height and the dielectric constant of the board have significant impact on transmission line dimensions. A 50- transmission line may result in impractically wide traces. A typical 1.6-mm thick FR4 board results in a trace width of 2.9 mm, for instance. This is impractical for the LMV221 because the pad width of the 6-pin WSON package is 0.25 mm. The transmission line has to be tapered from 2.9 mm to 0.25 mm. Significant reflections and resonances in the frequency transfer function of the board may occur due to this tapering. 10.1.4 GCPW Configuration A transmission line in a (grounded) coplanar waveguide (GCPW) configuration gives more flexibility in terms of trace width. The GCPW configuration is constructed with a conductor surrounded by ground at a certain distance, S, on the top side. Figure 84 shows a cross section of this configuration. The bottom side of the PCB is a ground plane. The ground planes on both sides of the PCB must be firmly connected to each other by multiple vias. The characteristic impedance of the transmission line is mainly determined by the width W and the distance S. In order to minimize reflections, the width W of the center trace must match the size of the package pad. The required value for the characteristic impedance can subsequently be realized by selection of the proper gap width S. METAL CONDUCTOR S W S H FR4 PCB GROUND PLANE Figure 84. GCPW Configuration 40 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 LMV221 www.ti.com SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 Layout Guidelines (continued) 10.1.5 Reference (REF) The REF pin can be used to compensate for temperature drift of the internal reference voltage as described in Interface to the ADC. The REF pin is directly connected to the inverting input of the transimpedance amplifier. Thus, RF signals and other spurious signals couple directly through to the output. Introduction of RF signals can be prevented by connecting a small capacitor between the REF pin and ground. The capacitor must be placed close to the REF pin as depicted in Figure 85. 10.1.6 Output (OUT) The OUT pin is sensitive to crosstalk from the RF input, especially at high power levels. The ESD diode between OUT and VDD may rectify the crosstalk, but may add an unwanted inaccurate DC component to the output voltage. The board layout must minimize crosstalk between the detectors input RFIN and the output of the detector. Using an additional capacitor connected between the output and the positive supply voltage (VDD pin) or GND can prevent this. For optimal performance this capacitor must be placed as close as possible to the OUT pin. 10.2 Layout Example Figure 85. Recommended LMV221 Board Layout Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 41 LMV221 SNWS018D - DECEMBER 2006 - REVISED JUNE 2016 www.ti.com 11 Device and Documentation Support 11.1 Community Resources The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of Use. TI E2ETM Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help solve problems with fellow engineers. Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and contact information for technical support. 11.2 Trademarks E2E is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. 11.3 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 11.4 Glossary SLYZ022 -- TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. 42 Submit Documentation Feedback Copyright (c) 2006-2016, Texas Instruments Incorporated Product Folder Links: LMV221 PACKAGE OPTION ADDENDUM www.ti.com 19-Oct-2017 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (C) Device Marking (4/5) LMV221SD/NOPB NRND WSON NGF 6 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 85 A96 LMV221SDX/NOPB NRND WSON NGF 6 4500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 85 A96 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based flame retardants must also meet the <=1000ppm threshold requirement. (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. 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Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 19-Oct-2017 Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 6-Apr-2016 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LMV221SD/NOPB WSON NGF 6 1000 178.0 12.4 2.8 2.5 1.0 8.0 12.0 Q1 LMV221SDX/NOPB WSON NGF 6 4500 330.0 12.4 2.8 2.5 1.0 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 6-Apr-2016 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LMV221SD/NOPB WSON NGF 6 1000 210.0 185.0 35.0 LMV221SDX/NOPB WSON NGF 6 4500 367.0 367.0 35.0 Pack Materials-Page 2 MECHANICAL DATA NGF0006A www.ti.com IMPORTANT NOTICE Texas Instruments Incorporated (TI) reserves the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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