LM3402/LM3402HV
0.5A Constant Current Buck Regulator for Driving High
Power LEDs
General Description
The LM3402/02HV are monolithic switching regulators de-
signed to deliver constant currents to high power LEDs. Ideal
for automotive, industrial, and general lighting applications,
they contain a high-side N-channel MOSFET switch with a
current limit of 735 mA (typical) for step-down (Buck) regu-
lators. Hysteretic control with controlled on-time coupled with
an external resistor allow the converter output voltage to
adjust as needed to deliver a constant current to series and
series - parallel connected arrays of LEDs of varying number
and type, LED dimming by pulse width modulation (PWM),
broken/open LED protection, low-power shutdown and ther-
mal shutdown complete the feature set.
Features
nIntegrated 0.5A N-channel MOSFET
nV
IN
Range from 6V to 42V (LM3402)
nV
IN
Range from 6V to 75V (LM3402HV)
n500 mA Output Current Over Temperature
nCycle-by-Cycle Current Limit
nNo Control Loop Compensation Required
nSeparate PWM Dimming and Low Power Shutdown
nSupports all-ceramic output capacitors and
capacitor-less outputs
nThermal shutdown protection
nMSOP-8 Package
Applications
nLED Driver
nConstant Current Source
nAutomotive Lighting
nGeneral Illumination
nIndustrial Lighting
Typical Application
20192101
October 2006
LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs
© 2006 National Semiconductor Corporation DS201921 www.national.com
Connection Diagram
20192102
8-Lead Plastic MSOP-8 Package
NS Package Number MUA08A
Ordering Information
Order Number Package Type NSC Package Drawing Supplied As
LM3402MM
MSOP-8 MUA08A
1000 units on tape and reel
LM3402MMX 3500 units on tape and reel
LM3402HVMM 1000 units on tape and reel
LM3402HVMMX 3500 units on tape and reel
Pin Descriptions
Pin(s) Name Description Application Information
1 SW Switch pin Connect this pin to the output inductor and Schottky diode.
2 BOOT MOSFET drive bootstrap pin Connect a 10 nF ceramic capacitor from this pin to SW.
3 DIM Input for PWM dimming Connect a logic-level PWM signal to this pin to enable/disable the
power FET and reduce the average light output of the LED array.
4 GND Ground pin Connect this pin to system ground.
5 CS Current sense feedback pin Set the current through the LED array by connecting a resistor from
this pin to ground.
6 RON On-time control pin A resistor connected from this pin to VIN sets the regulator
controlled on-time.
7 VCC Output of the internal 7V linear
regulator
Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor
with X5R or X7R dielectric.
8 VIN Input voltage pin Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V
(LM3402HV).
LM3402/LM3402HV
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Absolute Maximum Ratings
(LM3402)(Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN to GND -0.3V to 45V
BOOT to GND -0.3V to 59V
SW to GND -1.5V
BOOT to VCC -0.3V to 45V
BOOT to SW -0.3V to 14V
VCC to GND -0.3V to 14V
DIM to GND -0.3V to 7V
CS to GND -0.3V to 7V
RON to GND -0.3V to 7V
Junction Temperature 150˚C
Storage Temp. Range -65˚C to 125˚C
ESD Rating (Note 2) 2kV
Soldering Information
Lead Temperature (Soldering,
10sec) 260˚C
Infrared/Convection Reflow (15sec) 235˚C
Operating Ratings
(LM3402) (Note 1)
V
IN
6V to 42V
Junction Temperature Range −40˚C to +125˚C
Thermal Resistance θ
JA
(Note 3) 200˚C/W
Absolute Maximum Ratings
(LM3402HV)(Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN to GND -0.3V to 76V
BOOT to GND -0.3V to 90V
SW to GND -1.5V
BOOT to VCC -0.3V to 76V
BOOT to SW -0.3V to 14V
VCC to GND -0.3V to 14V
DIM to GND -0.3V to 7V
CS to GND -0.3V to 7V
RON to GND -0.3V to 7V
Junction Temperature 150˚C
Storage Temp. Range -65˚C to 125˚C
ESD Rating (Note 2) 2kV
Soldering Information
Lead Temperature (Soldering,
10sec) 260˚C
Infrared/Convection Reflow (15sec) 235˚C
Operating Ratings
(LM3402HV) (Note 1)
V
IN
6V to 75V
Junction Temperature Range −40˚C to +125˚C
Thermal Resistance θ
JA
(Note 3) 200˚C/W
LM3402/LM3402HV
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Electrical Characteristics V
IN
= 24V unless otherwise indicated. Typicals and limits appearing in plain type
apply for T
A
=T
J
= +25˚C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range.
Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.
LM3402
Symbol Parameter Conditions Min Typ Max Units
SYSTEM PARAMETERS
t
ON-1
On-time 1 V
IN
= 10V, R
ON
= 200 kΩ2.1 2.75 3.4 µs
t
ON-2
On-time 2 V
IN
= 40V, R
ON
= 200 kΩ490 650 810 ns
LM3402HV
Symbol Parameter Conditions Min Typ Max Units
SYSTEM PARAMETERS
t
ON-1
On-time 1 V
IN
= 10V, R
ON
= 200 kΩ2.1 2.75 3.4 µs
t
ON-2
On-time 2 V
IN
= 70V, R
ON
= 200 kΩ290 380 470 ns
LM3402/LM3402HV
Symbol Parameter Conditions Min Typ Max Units
REGULATION AND OVER-VOLTAGE COMPARATORS
V
REF-REG
CS Regulation Threshold CS Decreasing, SW turns on 194 200 206 mV
V
REF-0V
CS Over-voltage Threshold CS Increasing, SW turns off 300 mV
I
CS
CS Bias Current CS = 0V 0.1 µA
SHUTDOWN
V
SD-TH
Shutdown Threshold R
ON
/ SD Increasing 0.3 0.7 1.05 V
V
SD-HYS
Shutdown Hysteresis R
ON
/ SD Decreasing 40 mV
OFF TIMER
t
OFF-MIN
Minimum Off-time CS = 0V 300 ns
INTERNAL REGULATOR
V
CC-REG
V
CC
Regulated Output 6.6 77.4 V
V
IN-DO
V
IN
-V
CC
Dropout I
CC
= 5 mA, 6.0V <V
IN
<8.0V 300 mV
V
CC-BP-TH
V
CC
Bypass Threshold V
IN
Increasing 8.8 V
V
CC-BP-HYS
V
CC
Bypass Hysteresis V
IN
Decreasing 225 mV
V
CC-Z-6
V
CC
Output Impedance
(0 mA <I
CC
<5 mA)
V
IN
=6V 55 Ω
V
CC-Z-8
V
IN
=8V 50
V
CC-Z-24
V
IN
= 24V 0.4
V
CC-LIM
V
CC
Current Limit (Note 3) V
IN
= 24V, V
CC
=0V 16 mA
V
CC-UV-TH
V
CC
Under-voltage Lock-out
Threshold
V
CC
Increasing 5.25 V
V
CC-UV-HYS
V
CC
Under-voltage Lock-out
Hysteresis
V
CC
Decreasing 150 mV
V
CC-UV-DLY
V
CC
Under-voltage Lock-out
Filter Delay
100 mV Overdrive 3 µs
I
IN-OP
I
IN
Operating Current Non-switching, CS = 0V 600 900 µA
I
IN-SD
I
IN
Shutdown Current RON / SD = 0V 90 180 µA
CURRENT LIMIT
I
LIM
Current Limit Threshold 530 735 940 mA
DIM COMPARATOR
V
IH
Logic High DIM Increasing 2.2 V
V
IL
Logic Low DIM Decreasing 0.8 V
I
DIM-PU
DIM Pull-up Current DIM = 1.5V 75 µA
LM3402/LM3402HV
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Electrical Characteristics V
IN
= 24V unless otherwise indicated. Typicals and limits appearing in plain type
apply for T
A
=T
J
= +25˚C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range.
Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. (Continued)
LM3402/LM3402HV (Continued)
Symbol Parameter Conditions Min Typ Max Units
N-MOSFET AND DRIVER
R
DS-ON
Buck Switch On Resistance I
SW
= 200mA, BOOT-SW = 6.3V 0.7 1.5 Ω
V
DR-UVLO
BOOT Under-voltage
Lock-out Threshold
BOOT–SW Increasing 1.7 34V
V
DR-HYS
BOOT Under-voltage
Lock-out Hysteresis
BOOT–SW Decreasing 400 mV
THERMAL SHUTDOWN
T
SD
Thermal Shutdown Threshold 165 ˚C
T
SD-HYS
Thermal Shutdown
Hysteresis
25 ˚C
THERMAL RESISTANCE
θ
JA
Junction to Ambient MUA Package 200 ˚C/W
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics.
Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩresistor into each pin.
Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
Note 4: Typical specifications represent the most likely parametric norm at 25˚C operation.
LM3402/LM3402HV
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Typical Performance Characteristics
V
REF
vs Temperature (V
IN
= 24V) V
REF
vs V
IN
, LM3402 (T
A
= 25˚C)
20192129 20192130
V
REF
vs V
IN
, LM3402HV (T
A
= 25˚C) Current Limit vs Temperature (V
IN
= 24V)
20192131 20192132
Current Limit vs V
IN
, LM3402 (T
A
= 25˚C) Current Limit vs V
IN
, LM3402HV (T
A
= 25˚C)
20192133 20192134
LM3402/LM3402HV
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Typical Performance Characteristics (Continued)
T
ON
vs V
IN
,
R
ON
= 100 kΩ(T
A
= 25˚C)
T
ON
vs V
IN
,
(T
A
= 25˚C)
20192135 20192136
T
ON
vs V
IN
,
(T
A
= 25˚C)
T
ON
vs R
ON
, LM3402
(T
A
= 25˚C)
20192137 20192144
T
ON
vs R
ON
, LM3402HV
(T
A
= 25˚C)
V
CC
vs V
IN
(T
A
= 25˚C)
20192138 20192139
LM3402/LM3402HV
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Typical Performance Characteristics (Continued)
V
O-MAX
vs f
SW
, LM3402
(T
A
= 25˚C)
V
O-MIN
vs f
SW
, LM3402
(T
A
= 25˚C)
20192140 20192141
V
O-MAX
vs f
SW
, LM3402HV
(T
A
= 25˚C)
V
O-MIN
vs f
SW
, LM3402HV
(T
A
= 25˚C)
20192142 20192143
LM3402/LM3402HV
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Block Diagram
20192103
Application Information
THEORY OF OPERATION
The LM3402 and LM3402HV are buck regulators with a wide
input voltage range, low voltage reference, and a fast output
enable/disable function. These features combine to make
them ideal for use as a constant current source for LEDs with
forward currents as high as 500 mA. The controlled on-time
(COT) architecture is a combination of hysteretic mode con-
trol and a one-shot on-timer that varies inversely with input
voltage. Hysteretic operation eliminates the need for small-
signal control loop compensation. When the converter runs
in continuous conduction mode (CCM) the controlled on-time
maintains a constant switching frequency over the range of
input voltage. Fast transient response, PWM dimming, a low
power shutdown mode, and simple output overvoltage pro-
tection round out the functions of the LM3402/02HV.
CONTROLLED ON-TIME OVERVIEW
Figure 1 shows the feedback system used to control the
current through an array of LEDs. A voltage signal, V
SNS
,is
created as the LED current flows through the current setting
resistor, R
SNS
, to ground. V
SNS
is fed back to the CS pin,
where it is compared against a 200 mV reference, V
REF
. The
on-comparator turns on the power MOSFET when V
SNS
falls
below V
REF
. The power MOSFET conducts for a controlled
on-time, t
ON
, set by an external resistor, R
ON
, and by the
input voltage, V
IN
. On-time is governed by the following
equation:
At the conclusion of t
ON
the power MOSFET turns off for a
minimum off-time, t
OFF-MIN
, of 300 ns. Once t
OFF-MIN
is
complete the CS comparator compares V
SNS
and V
REF
again, waiting to begin the next cycle.
LM3402/LM3402HV
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Application Information (Continued)
The LM3402/02HV regulators should be operated in continu-
ous conduction mode (CCM), where inductor current stays
positive throughout the switching cycle. During steady-state
operationin the CCM, the converter maintains a constant
switching frequency, which can be selected using the follow-
ing equation:
V
F
= forward voltage of each LED, n = number of LEDs in
series
AVERAGE LED CURRENT ACCURACY
The COT architecture regulates the valley of ∆V
SNS
, the AC
portion of V
SNS
. To determine the average LED current
(which is also the average inductor current) the valley induc-
tor current is calculated using the following expression:
In this equation t
SNS
represents the propagation delay of the
CS comparator, and is approximately 220 ns. The average
inductor/LED current is equal to I
L-MIN
plus one-half of the
inductor current ripple, ∆i
L
:
I
F
=I
L
=I
L-MIN
+∆i
L
/2
Detailed information for the calculation of ∆i
L
is given in the
Design Considerations section.
MAXIMUM OUTPUT VOLTAGE
The 300 ns minimum off-time limits on the maximum duty
cycle of the converter, D
MAX
, and in turn ,the maximum
output voltage V
O(MAX)
is determined by the following equa-
tions:
The maximum number of LEDs, n
MAX
, that can be placed in
a single series string is governed by V
O(MAX)
and the maxi-
mum forward voltage of the LEDs used, V
F(MAX)
, using the
expression:
At low switching frequency the maximum duty cycle and
output voltage are higher, allowing the LM3402/02HV to
regulate output voltages that are nearly equal to input volt-
age. The following equation relates switching frequency to
maximum output voltage.
20192105
FIGURE 1. Comparator and One-Shot
LM3402/LM3402HV
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Application Information (Continued)
MINIMUM OUTPUT VOLTAGE
The minimum recommended on-time for the LM3402/02HV
is 300 ns. This lower limit for t
ON
determines the minimum
duty cycle and output voltage that can be regulated based on
input voltage and switching frequency. The relationship is
determined by the following equation:
HIGH VOLTAGE BIAS REGULATOR
The LM3402/02HV contains an internal linear regulator with
a 7V output, connected between the VIN and the VCC pins.
The VCC pin should be bypassed to the GND pin with a 0.1
µF ceramic capacitor connected as close as possible to the
pins of the IC. VCC tracks VIN until VIN reaches 8.8V
(typical) and then regulates at 7V as VIN increases. Opera-
tion begins when VCC crosses 5.25V.
INTERNAL MOSFET AND DRIVER
The LM3402/02HV features an internal power MOSFET as
well as a floating driver connected from the SW pin to the
BOOT pin. Both rise time and fall time are 20 ns each
(typical) and the approximate gate charge is 3 nC. The
high-side rail for the driver circuitry uses a bootstrap circuit
consisting of an internal high-voltage diode and an external
10 nF capacitor, C
B
.V
CC
charges C
B
through the internal
diode while the power MOSFET is off. When the MOSFET
turns on, the internal diode reverse biases. This creates a
floating supply equal to the V
CC
voltage minus the diode
drop to drive the MOSFET when its source voltage is equal
to V
IN
.
FAST SHUTDOWN FOR PWM DIMMING
The DIM pin of the LM3402/02HV is a TTL logic compatible
input for low frequency PWM dimming of the LED. A logic low
(below 0.8V) at DIM will disable the internal MOSFET and
shut off the current flow to the LED array. While the DIM pin
is in a logic low state the support circuitry (driver, bandgap,
VCC) remains active in order to minimize the time needed to
turn the LED array back on when the DIM pin sees a logic
high (above 2.2V). A 75 µA (typical) pull-up current ensures
that the LM3402/02HV is on when DIM pin is open circuited,
eliminating the need for a pull-up resistor. Dimming fre-
quency, f
DIM
, and duty cycle, D
DIM
, are limited by the LED
current rise time and fall time and the delay from activation of
the DIM pin to the response of the internal power MOSFET.
In general, f
DIM
should be at least one order of magnitude
lower than the steady state switching frequency in order to
prevent aliasing.
PEAK CURRENT LIMIT
The current limit comparator of the LM3402/02HV will en-
gage whenever the power MOSFET current (equal to the
inductor current while the MOSFET is on) exceeds 735 mA
(typical). The power MOSFET is disabled for a cool-down
time that is 10x the steady-state on-time. At the conclusion of
this cool-down time the system re-starts. If the current limit
condition persists the cycle of cool-down time and restarting
will continue, creating a low-power hiccup mode, minimizing
thermal stress on the LM3402/02HV and the external circuit
components.
OVER-VOLTAGE/OVER-CURRENT COMPARATOR
The CS pin includes an output over-voltage/over-current
comparator that will disable the power MOSFET whenever
V
SNS
exceeds 300 mV. This threshold provides a hard limit
for the output current. Output current overshoot is limited to
300 mV / R
SNS
by this comparator during transients.
The OVP/OCP comparator can also be used to prevent the
output voltage from rising to V
O(MAX)
in the event of an
output open-circuit. This is the most common failure mode
for LEDs, due to breaking of the bond wires. In a current
regulator an output open circuit causes V
SNS
to fall to zero,
commanding maximum duty cycle. Figure 2 shows a method
using a zener diode, Z1, and zener limiting resistor, R
Z
,to
limit output voltage to the reverse breakdown voltage of Z1
plus 200 mV. The zener diode reverse breakdown voltage,
V
Z
, must be greater than the maximum combined V
F
of all
LEDs in the array. The maximum recommended value for R
Z
is1kΩ.
As discussed in the Maximum Output Voltage section, there
is a limit to how high V
O
can rise during an output open-
circuit that is always less than V
IN
. If no output capacitor is
used, the output stage of the LM3402/02HV is capable of
withstanding V
O(MAX)
indefinitely, however the voltage at the
output end of the inductor will oscillate and can go above V
IN
or below 0V. A small (typically 10 nF) capacitor across the
LED array dampens this oscillation. For circuits that use an
output capacitor, the system can still withstand V
O(MAX)
in-
definitely as long as C
O
is rated to handle V
IN
. The high
current paths are blocked in output open-circuit and the risk
of thermal stress is minimal, hence the user may opt to allow
the output voltage to rise in the case of an open-circuit LED
failure.
LM3402/LM3402HV
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Application Information (Continued)
LOW POWER SHUTDOWN
The LM3402/02HV can be switched to a low power state
(I
IN-SD
= 90 µA) by grounding the RON pin with a signal-level
MOSFET as shown in Figure 3. Low power MOSFETs like
the 2N7000, 2N3904, or equivalent are recommended de-
vices for putting the LM3402/02HV into low power shutdown.
Logic gates can also be used to shut down the LM3402/
02HV as long as the logic low voltage is below the over
temperature minimum threshold of 0.3V. Noise filter circuitry
on the RON pin can cause a few pulses with a longer on-time
than normal after RON is grounded or released. In these
cases the OVP/OCP comparator will ensure that the peak
inductor or LED current does not exceed 300 mV / R
SNS
.
THERMAL SHUTDOWN
Internal thermal shutdown circuitry is provided to protect the
IC in the event that the maximum junction temperature is
exceeded. The threshold for thermal shutdown is 165˚C with
a 25˚C hysteresis (both values typical). During thermal shut-
down the MOSFET and driver are disabled.
20192112
FIGURE 2. Output Open Circuit Protection
20192113
FIGURE 3. Low Power Shutdown
LM3402/LM3402HV
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Design Considerations
SWITCHING FREQUENCY
Switching frequency is selected based on the tradeoffs be-
tween efficiency (better at low frequency), solution size/cost
(smaller at high frequency), and the range of output voltage
that can be regulated (wider at lower frequency.) Many ap-
plications place limits on switching frequency due to EMI
sensitivity. The on-time of the LM3402/02HV can be pro-
grammed for switching frequencies ranging from the 10’s of
kHz to over 1 MHz. The maximum switching frequency is
limited only by the minimum on-time requirement.
LED RIPPLE CURRENT
Selection of the ripple current, ∆i
F
, through the LED array is
analogous to the selection of output ripple voltage in a
standard voltage regulator. Where the output ripple in a
voltage regulator is commonly ±1% to ±5% of the DC output
voltage, LED manufacturers generally recommend values
for ∆i
F
ranging from ±5% to ±20% of I
F
. Higher LED ripple
current allows the use of smaller inductors, smaller output
capacitors, or no output capacitors at all. The advantages of
higher ripple current are reduction in the solution size and
cost. Lower ripple current requires more output inductance,
higher switching frequency, or additional output capacitance.
The advantages of lower ripple current are a reduction in
heating in the LED itself and greater range of the average
LED current before the current limit of the LED or the driving
circuitry is reached.
BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS
The buck converter is unique among non-isolated topologies
because of the direct connection of the inductor to the load
during the entire switching cycle. By definition an inductor
will control the rate of change of current that flows through it,
and this control over current ripple forms the basis for com-
ponent selection in both voltage regulators and current regu-
lators. A current regulator such as the LED driver for which
the LM3402/02HV was designed focuses on the control of
the current through the load, not the voltage across it. A
constant current regulator is free of load current transients,
and has no need of output capacitance to supply the load
and maintain output voltage. Referring to the Typical Appli-
cation circuit on the front page of this datasheet, the inductor
and LED can form a single series chain, sharing the same
current. When no output capacitor is used, the same equa-
tions that govern inductor ripple current, ∆i
L
, also apply to the
LED ripple current, ∆i
F
. For a controlled on-time converter
such as LM3402/02HV the ripple current is described by the
following expression:
A minimum ripple voltage of 25 mV is recommended at the
CS pin to provide good signal-to-noise ratio (SNR). The CS
pin ripple voltage, ∆V
SNS
, is described by the following:
∆V
SNS
=∆i
F
xR
SNS
BUCK CONVERTERS WITH OUTPUT CAPACITORS
A capacitor placed in parallel with the LED or array of LEDs
can be used to reduce the LED current ripple while keeping
the same average current through both the inductor and the
LED array. This technique is demonstrated in Design Ex-
ample 1. With this topology the output inductance can be
lowered, making the magnetics smaller and less expensive.
Alternatively, the circuit could be run at lower frequency but
keep the same inductor value, improving the efficiency and
expanding the range of output voltage that can be regulated.
Both the peak current limit and the OVP/OCP comparator
still monitor peak inductor current, placing a limit on how
large ∆i
L
can be even if ∆i
F
is made very small. A parallel
output capacitor is also useful in applications where the
inductor or input voltage tolerance is poor. Adding a capaci-
tor that reduces ∆i
F
to well below the target provides head-
room for changes in inductance or V
IN
that might otherwise
push the peak LED ripple current too high.
Figure 4 shows the equivalent impedances presented to the
inductor current ripple when an output capacitor, C
O
, and its
equivalent series resistance (ESR) are placed in parallel with
the LED array. The entire inductor ripple current flows
through R
SNS
to provide the required 25 mV of ripple voltage
for proper operation of the CS comparator.
To calculate the respective ripple currents the LED array is
represented as a dynamic resistance, r
D
. LED dynamic re-
sistance is not always specified on the manufacturer’s
datasheet, but it can be calculated as the inverse slope of
the LED’s V
F
vs. I
F
curve. Note that dividing V
F
by I
F
will give
an incorrect value that is 5x to 10x too high. Total dynamic
resistance for a string of n LEDs connected in series can be
calculated as the r
D
of one device multiplied by n. Inductor
ripple current is still calculated with the expression from Buck
Regulators without Output Capacitors. The following equa-
tions can then be used to estimate ∆i
F
when using a parallel
capacitor:
The calculation for Z
C
assumes that the shape of the induc-
tor ripple current is approximately sinusoidal.
Small values of C
O
that do not significantly reduce ∆i
F
can
also be used to control EMI generated by the switching
action of the LM3402/02HV. EMI reduction becomes more
important as the length of the connections between the LED
and the rest of the circuit increase.
20192115
FIGURE 4. LED and C
O
Ripple Current
LM3402/LM3402HV
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Design Considerations (Continued)
INPUT CAPACITORS
Input capacitors at the VIN pin of the LM3402/02HV are
selected using requirements for minimum capacitance and
rms ripple current. The input capacitors supply pulses of
current approximately equal to I
F
while the power MOSFET
is on, and are charged up by the input voltage while the
power MOSFET is off. Switching converters such as the
LM3402/02HV have a negative input impedance due to the
decrease in input current as input voltage increases. This
inverse proportionality of input current to input voltage can
cause oscillations (sometimes called ‘power supply interac-
tion’) if the magnitude of the negative input impedance is
greater the the input filter impedance. Minimum capacitance
can be selected by comparing the input impedance to the
converter’s negative resistance; however this requires accu-
rate calculation of the input voltage source inductance and
resistance, quantities which can be difficult to determine. An
alternative method to select the minimum input capacitance,
C
IN(MIN)
, is to select the maximum voltage ripple which can
be tolerated. This value,∆v
IN(MAX)
, is equal to the change in
voltage across C
IN
during the converter on-time, when C
IN
supplies the load current. C
IN(MIN)
can be selected with the
following:
A good starting point for selection of C
IN
is to use an input
voltage ripple of 5% to 10% of V
IN
. A minimum input capaci-
tance of 2x the C
IN(MIN)
value is recommended for all
LM3402/02HV circuits. To determine the rms current rating,
the following formula can be used:
Ceramic capacitors are the best choice for the input to the
LM3402/02HV due to their high ripple current rating, low
ESR, low cost, and small size compared to other types.
When selecting a ceramic capacitor, special attention must
be paid to the operating conditions of the application. Ce-
ramic capacitors can lose one-half or more of their capaci-
tance at their rated DC voltage bias and also lose capaci-
tance with extremes in temperature. A DC voltage rating
equal to twice the expected maximum input voltage is rec-
ommended. In addition, the minimum quality dielectric which
is suitable for switching power supply inputs is X5R, while
X7R or better is preferred.
RECIRCULATING DIODE
The LM3402/02HV is a non-synchronous buck regulator that
requires a recirculating diode D1 (see the Typical Application
circuit) to carrying the inductor current during the MOSFET
off-time. The most efficient choice for D1 is a Schottky diode
due to low forward drop and near-zero reverse recovery
time. D1 must be rated to handle the maximum input voltage
plus any switching node ringing when the MOSFET is on. In
practice all switching converters have some ringing at the
switching node due to the diode parasitic capacitance and
the lead inductance. D1 must also be rated to handle the
average current, I
D
, calculated as:
I
D
=(1–D)xI
F
This calculation should be done at the maximum expected
input voltage. The overall converter efficiency becomes
more dependent on the selection of D1 at low duty cycles,
where the recirculating diode carries the load current for an
increasing percentage of the time. This power dissipation
can be calculated by checking the typical diode forward
voltage, V
D
, from the I-V curve on the product datasheet and
then multiplying it by I
D
. Diode datasheets will also provide a
typical junction-to-ambient thermal resistance, θ
JA
, which
can be used to estimate the operating die temperature of the
Schottky. Multiplying the power dissipation (P
D
=I
D
xV
D
)by
θ
JA
gives the temperature rise. The diode case size can then
be selected to maintain the Schottky diode temperature
below the operational maximum.
Design Example 1: LM3402
The first example circuit will guide the user through compo-
nent selection for an architectural accent lighting application.
A regulated DC voltage input of 24V ±10% will power a
single 1W white LED at a forward current of 350 mA ±5%.
The typical forward voltage of a 1W InGaN LED is 3.5V,
hence the estimated average output voltage will be 3.7V.
The objective of this application is to place the complete
current regulator and LED in the compact space formerly
occupied by an MR16 halogen light bulb. (The LED will be on
a separate metal-core PCB.) Switching frequency will be as
fast as the 300 ns t
ON
limit allows, with the emphasis on
space savings over efficiency. Efficiency cannot be ignored,
however, as the confined space with little air-flow requires a
maximum temperature rise of 40˚C in each circuit compo-
nent. A complete bill of materials can be found in Table 1 at
the end of this datasheet.
LM3402/LM3402HV
www.national.com 14
Design Example 1: LM3402 (Continued)
R
ON
and t
ON
To select R
ON
the expression relating t
ON
to input voltage
from the Controlled On-time Overview section can be re-
written as:
Minimum on-time occurs at the maximum V
IN
, which is 24V
x 110% = 26.4V. R
ON
is therefore calculated as:
R
ON
=(300x10
-9
x 26.4) / 1.34 x 10
-10
= 59105 Ω
The closest 1% tolerance resistor is 59.0 kΩ. The switching
frequency of the circuit can then be found using the equation
relating R
ON
to f
SW
:
f
SW
= 3.7 / (59000 x 1.34 x 10
-10
) = 468 kHz
USING AN OUTPUT CAPACITOR
The inductor will be the largest component used in this
design. Because the application does not require any PWM
dimming, an output capacitor can be used to greatly reduce
the inductance needed without worry of slowing the potential
PWM dimming frequency. The total solution size will be
reduced by using an output capacitor and small inductor as
opposed to one large inductor.
OUTPUT INDUCTOR
Knowing that an output capacitor will be used, the inductor
can be selected for a larger current ripple. The desired
maximum value for ∆i
L
is ±30%, or 0.6 x 350 mA = 210
mA
P-P
. Minimum inductance is selected at the maximum
input voltage. Re-arranging the equation for current ripple
selection yields the following:
L
MIN
= [(26.4 – 3.7) x 300 x 10
-9
] / (0.6 x 0.35) = 32.4 µH
The closest standard inductor value is 33 µH. Off-the-shelf
inductors rated at 33 µH are available from many magnetics
manufacturers.
Inductor datasheets should contain three specifications
which are used to select the inductor. The first of these is the
average current rating, which for a buck regulator is equal to
the average load current, or I
F
. The average current rating is
given by a specified temperature rise in the inductor, nor-
mally 40˚C. For this example, the average current rating
should be greater than 350 mA to ensure that heat from the
inductor does not reduce the lifetime of the LED or cause the
LM3402 to enter thermal shutdown.
The second specification is the tolerance of the inductance
itself, typically ±10% to ±30% of the rated inductance. In this
example an inductor with a tolerance of ±20% will be used.
With this tolerance the typical, minimum, and maximum in-
ductor current ripples can be calculated:
∆i
L(TYP)
= [(26.4 – 3.7) x 300 x 10
-9
]/33x10
-6
= 206 mA
P-P
∆i
L(MIN)
= [(26.4 – 3.7) x 300 x 10
-9
]/39.6x10
-6
= 172 mA
P-P
∆i
L(MAX)
= [(26.4 – 3.7) x 300 x 10
-9
]/26.4x10
-6
= 258 mA
P-P
The third specification for an inductor is the peak current
rating, normally given as the point at which the inductance
drops off by a given percentage due to saturation of the core.
The worst-case peak current occurs at maximum input volt-
age and at minimum inductance, and can be determined
with the equation from the Design Considerations section:
20192119
FIGURE 5. Schematic for Design Example 1
LM3402/LM3402HV
www.national.com15
Design Example 1: LM3402
(Continued)
I
L(PEAK)
= 0.35 + 0.258/2=479mA
For this example the peak current rating of the inductor
should be greater than 479 mA. In the case of a short circuit
across the LED array, the LM3402 will continue to deliver
rated current through the short but will reduce the output
voltage to equal the CS pin voltage of 200 mV. Worst-case
peak current in this condition is equal to:
∆i
L(LED-SHORT)
= [(26.4 – 0.2) x 300 x 10
-9
]/26.4x10
-6
= 298 mA
P-P
I
L(PEAK)
= 0.35 + 0.149 = 499 mA
In the case of a short at the switch node, the output, or from
the CS pin to ground the short circuit current limit will engage
at a typical peak current of 735 mA. In order to prevent
inductor saturation during these short circuits the inductor’s
peak current rating must be above 735 mA. The device
selected is an off-the-shelf inductor rated 33 µH ±20% with a
DCR of 96 mΩand a peak current rating of 0.82A. The
physical dimensions of this inductor are 7.0 x 7.0 x 4.5 mm.
R
SNS
The current sensing resistor value can be determined by
re-arranging the expression for average LED current from
the LED Current Accuracy section:
R
SNS
= 0.74Ω,t
SNS
= 220 ns
Sub-1Ωresistors are available in both 1% and 5% tolerance.
A 1%, 0.75Ωresistor will give the best accuracy of the
average LED current. To determine the resistor size the
power dissipation can be calculated as:
P
SNS
=(I
F
)
2
xR
SNS
P
SNS
= 0.35
2
x0.75=92mW
Standard 0805 size resistors are rated to 125 mW and will be
suitable for this application.
To select the proper output capacitor the equation from Buck
Regulators with Output Capacitors is re-arranged to yield the
following:
The target tolerance for LED ripple current is ±5% or 10%
P-P
=35mA
P-P
, and the LED datasheet gives a typical value for
r
D
of 1.0Ωat 350 mA. The required capacitor impedance to
reduce the worst-case inductor ripple current of 258 mA
P-P
is
therefore:
Z
C
= [0.035 / (0.258 - 0.035] x 1.0 = 0.157Ω
A ceramic capacitor will be used and the required capaci-
tance is selected based on the impedance at 468 kHz:
C
O
= 1/(2 x πx 0.157 x 4.68 x 10
5
) = 2.18 µF
This calculation assumes that impedance due to the equiva-
lent series resistance (ESR) and equivalent series induc-
tance (ESL) of C
O
is negligible. The closest 10% tolerance
capacitor value is 2.2 µF. The capacitor used should be rated
to 10V or more and have an X7R dielectric. Several manu-
facturers produce ceramic capacitors with these specifica-
tions in the 0805 case size. A typical value for ESR of 1 mΩ
can be read from the curve of impedance vs. frequency in
the product datasheet.
INPUT CAPACITOR
Following the calculations from the Input Capacitor section,
∆v
IN(MAX)
will be 1%
P-P
= 240 mV. The minimum required
capacitance is:
C
IN(MIN)
= (0.35 x 300 x 10
-9
) / 0.24 = 438 nF
In expectation that more capacitance will be needed to pre-
vent power supply interaction a 1.0 µF ceramic capacitor
rated to 50V with X7R dielectric in a 1206 case size will be
used. From the Design Considerations section, input rms
current is:
I
IN-RMS
= 0.35 x Sqrt(0.154 x 0.846) = 126 mA
Ripple current ratings for 1206 size ceramic capacitors are
typically higher than 1A, more than enough for this design.
RECIRCULATING DIODE
The first parameter for D1 which must be determined is the
reverse voltage rating. Schottky diodes are available at re-
verse ratings of 30V and 40V, often in the same package,
with the same forward current rating. To account for ringing a
40V Schottky will be used.
The next parameters to be determined are the forward cur-
rent rating and case size. In this example the low duty cycle
(D = 3.7 / 24 = 15%) requires the recirculating diode D1 to
carry the load current much longer than the internal power
MOSFET of the LM3402. The estimated average diode cur-
rent is:
I
D
= 0.35 x 0.85 = 298 mA
Schottky diodes are available at forward current ratings of
0.5A, however the current rating often assumes a 25˚C
ambient temperature and does not take into account the
application restrictions on temperature rise. A diode rated for
higher current may be needed to keep the temperature rise
below 40˚C.To determine the proper case size, the dissipa-
tion and temperature rise in D1 can be calculated as shown
in the Design Considerations section. V
D
for a small case
LM3402/LM3402HV
www.national.com 16
Design Example 1: LM3402
(Continued)
size such as SOD-123 in a 40V, 0.5A Schottky diode at 350
mA is approximately 0.4V and the θ
JA
is 206˚C/W. Power
dissipation and temperature rise can be calculated as:
P
D
= 0.298 x 0.4 = 119 mW
T
RISE
= 0.119 x 206 = 24.5˚C
According to these calculations the SOD-123 diode will meet
the requirements. Heating and dissipation are among the
factors most difficult to predict in converter design. If pos-
sible, a footprint should be used that is capable of accepting
both SOD-123 and a larger case size, such as SMA. A larger
diode with a higher forward current rating will generally have
a lower forward voltage, reducing dissipation, as well as
having a lower θ
JA
, reducing temperature rise.
C
B
and C
F
The bootstrap capacitor C
B
should always be a 10 nF ce-
ramic capacitor with X7R dielectric. A 25V rating is appropri-
ate for all application circuits. The linear regulator filter ca-
pacitor C
F
should always be a 100 nF ceramic capacitor,
also with X7R dielectric and a 25V rating.
EFFICIENCY
To estimate the electrical efficiency of this example the
power dissipation in each current carrying element can be
calculated and summed. This term should not be confused
with the optical efficacy of the circuit, which depends upon
the LEDs themselves.
Total output power, P
O
, is calculated as:
P
O
=I
F
xV
O
= 0.35 x 3.7 = 1.295W
Conduction loss, P
C
, in the internal MOSFET:
P
C
=(I
F2
xR
DSON
)xD=(0.35
2
x 1.5) x 0.154 = 28 mW
Gate charging and VCC loss, P
G
, in the gate drive and linear
regulator:
P
G
=(I
IN-OP
+f
SW
xQ
G
)xV
IN
P
G
=(600x10
-6
+ 468000x3x10
-9
)x24=48mW
Switching loss, P
S
, in the internal MOSFET:
P
S
=0.5xV
IN
xI
F
x(t
R
+t
F
)xf
SW
P
S
=0.5x24x0.35x(40x10
-9
) x 468000 = 78 mW
AC rms current loss, P
CIN
, in the input capacitor:
P
CIN
=I
IN(rms)2
x ESR = (0.126)
2
x 0.006 = 0.1 mW (negli-
gible)
DCR loss, P
L
, in the inductor
P
L
=I
F2
x DCR = 0.35
2
x 0.096 = 11.8 mW
Recirculating diode loss, P
D
= 119 mW
Current Sense Resistor Loss, P
SNS
=92mW
Electrical efficiency, η=P
O
/(P
O
+ Sum of all loss terms) =
1.295 / (1.295 + 0.377) = 77%
DIE TEMPERATURE
T
LM3402
=(P
C
+P
G
+P
S
)xθ
JA
T
LM3402
= (0.028 + 0.05 + 0.078) x 200 = 31˚C
Design Example 2: LM3402HV
The second example application is an RGB backlight for a
flat screen monitor. A separate boost regulator provides a
60V ±5% DC input rail that feeds three LM3402HV current
regulators to drive one series array each of red, green, and
blue 1W LEDs. The target for average LED current is 350
mA ±5% in each string. The monitor will adjust the color
temperature dynamically, requiring fast PWM dimming of
each string with external, parallel MOSFETs. 1W green and
blue InGaN LEDs have a typical forward voltage of 3.5V,
however red LEDs use AlInGaP technology with a typical
forward voltage of 2.9V. In order to match color properly the
design requires 14 green LEDs, twice as many as needed
for the red and blue LEDs. This example will follow the
design for the green LED array, providing the necessary
information to repeat the exercise for the blue and red LED
arrays. The circuit schematic for Design Example 2 is the
same as the Typical Application on the front page. The bill of
materials (green array only) can be found in Table 2 at the
end of this datasheet.
OUTPUT VOLTAGE
Green Array: V
O(G)
= 14 x 3.5 + 0.2 = 49.2V
Blue Array: V
O(B)
= 7 x 3.5 + 0.2 = 24.7V
Red Array: V
O(R)
= 7 x 2.9 + 0.2 = 20.5V
R
ON
and t
ON
A compromise in switching frequency is needed in this ap-
plication to balance the requirements of magnetics size and
efficiency. The high duty cycle translates into large conduc-
tion losses and high temperature rise in the IC. For best
response to a PWM dimming signal this circuit will not use an
output capacitor; hence a moderate switching frequency of
300 kHz will keep the inductance from becoming so large
that a custom-wound inductor is needed. This design will use
only surface mount components, and the selection of off-the-
shelf SMT inductors for switching regulators is poor at 1000
µH and above. R
ON
is selected from the equation for switch-
ing frequency as follows:
R
ON
= 49.2 / (1.34 x 10
-10
x3x10
5
) = 1224 kΩ
The closest 1% tolerance resistor is 1.21 MΩ. The switching
frequency and on-time of the circuit can then be found using
the equations relating R
ON
and t
ON
to f
SW
:
f
SW
= 49.2 / (1210000 x 1.34 x 10
-10
) = 303 kHz
LM3402/LM3402HV
www.national.com17
Design Example 2: LM3402HV
(Continued)
t
ON
= (1.34 x 10
-10
x 1210000) / 60 = 2.7 µs
USING AN OUTPUT CAPACITOR
This application is dominated by the need for fast PWM
dimming, requiring a circuit without any output capacitance.
OUTPUT INDUCTOR
In this example the ripple current through the LED array and
the inductor are equal. Inductance is selected to give the
smallest ripple current possible while still providing enough
∆v
SNS
signal for the CS comparator to operate correctly.
Designing to a desired ∆v
SNS
of 25 mV and assuming that
the average inductor current will equal the desired average
LED current of 350 mA yields the target current ripple in the
inductor and LEDs:
∆i
F
=∆i
L
=∆v
SNS
/R
SNS
,R
SNS
=V
SNS
/I
F
∆i
F
= 0.025 / 0.57 = 43.8 mA
With the target ripple current determined the inductance can
be chosen:
L
MIN
= [(60 – 49.2) x 2.7 x 10
-6
] / (0.044) = 663 µH
The closest standard inductor value is 680 µH. As with the
previous example, the average current rating should be
greater than 350 mA. Separation between the LM3402HV
drivers and the LED arrays mean that heat from the inductor
will not threaten the lifetime of the LEDs, but an overheated
inductor could still cause the LM3402HV to enter thermal
shutdown.
The inductance itself of the standard part chosen is ±20%.
With this tolerance the typical, minimum, and maximum in-
ductor current ripples can be calculated:
∆i
F(TYP)
= [(60 - 49.2) x 2.7 x 10
-6
]/680x10
-6
=43mA
P-P
∆i
F(MIN)
= [(60 - 49.2) x 2.7 x 10
-6
]/816x10
-6
=36mA
P-P
∆i
F(MAX)
= [(60 - 49.2) x 2.7 x 10
-6
]/544x10
-6
=54mA
P-P
The peak LED/inductor current is then estimated:
I
L(PEAK)
=I
L
+[∆i
L(MAX)
]/2
I
L(PEAK)
= 0.35 + 0.027 = 377 mA
In the case of a short circuit across the LED array, the
LM3402HV will continue to deliver rated current through the
short but will reduce the output voltage to equal the CS pin
voltage of 200 mV. Worst-case peak current in this condition
would be equal to:
∆i
F(LED-SHORT)
= [(63 – 0.2) x 2.7 x 10
-6
]/544x10
-6
= 314 mA
P-P
I
F(PEAK)
= 0.35 + 0.156 = 506 mA
In the case of a short at the switch node, the output, or from
the CS pin to ground the short circuit current limit will engage
at a typical peak current of 735 mA. In order to prevent
inductor saturation during these fault conditions the induc-
tor’s peak current rating must be above 735 mA. A 680 µH
off-the shelf inductor rated to 1.2A (peak) and 0.72A (aver-
age) with a DCR of 1.1Ωwill be used for the green LED
array.
R
SNS
A preliminary value for R
SNS
was determined in selecting ∆i
L
.
This value should be re-evaluated based on the calculations
for ∆i
F
:
Sub-1Ωresistors are available in both 1% and 5% tolerance.
A 1%, 0.56Ωdevice is the closest value, and a 0.125W, 0805
size device will handle the power dissipation of 69 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% toler-
ance requirement. From the expression for LED current
accuracy:
I
F
= 0.19 / 0.56 + 0.043/2=361mA,3%above 350 mA
INPUT CAPACITOR
Following the calculations from the Input Capacitor section,
∆v
IN(MAX)
will be 1%
P-P
= 600 mV. The minimum required
capacitance is:
C
IN(MIN)
= (0.35 x 2.7 x 10
-6
) / 0.6 = 1.6 µF
In expectation that more capacitance will be needed to pre-
vent power supply interaction a 2.2 µF ceramic capacitor
rated to 100V with X7R dielectric in an 1812 case size will be
used. From the Design Considerations section, input rms
current is:
I
IN-RMS
= 0.35 x Sqrt(0.82 x 0.18) = 134 mA
Ripple current ratings for 1812 size ceramic capacitors are
typically higher than 2A, more than enough for this design.
RECIRCULATING DIODE
The input voltage of 60V ±5% requires Schottky diodes with
a reverse voltage rating greater than 60V. Some manufac-
turers provide Schottky diodes with ratings of 70, 80 or 90V;
however the next highest standard voltage rating is 100V.
Selecting a 100V rated diode provides a large safety margin
for the ringing of the switch node and also makes cross-
referencing of diodes from different vendors easier.
LM3402/LM3402HV
www.national.com 18
Design Example 2: LM3402HV
(Continued)
The next parameters to be determined are the forward cur-
rent rating and case size. In this example the high duty cycle
(D = 49.2 / 60 = 82%) places less thermals stress on D1 and
more on the internal power MOSFET of the LM3402. The
estimated average diode current is:
I
D
= 0.361 x 0.18 = 65 mA
A Schottky with a forward current rating of 0.5A would be
adequate, however at 100V the majority of diodes have a
minimum forward current rating of 1A. To determine the
proper case size, the dissipation and temperature rise in D1
can be calculated as shown in the Design Considerations
section. V
D
for a small case size such as SOD-123F in a
100V, 1A Schottky diode at 350 mA is approximately 0.65V
and the θ
JA
is 88˚C/W. Power dissipation and temperature
rise can be calculated as:
P
D
= 0.065 x 0.65 = 42 mW
T
RISE
= 0.042 x 88 = 4˚C
C
B
AND C
F
The bootstrap capacitor C
B
should always be a 10 nF ce-
ramic capacitor with X7R dielectric. A 25V rating is appropri-
ate for all application circuits. The linear regulator filter ca-
pacitor C
F
should always be a 100 nF ceramic capacitor,
also with X7R dielectric and a 25V rating.
EFFICIENCY
To estimate the electrical efficiency of this example the
power dissipation in each current carrying element can be
calculated and summed. Electrical efficiency, η, should not
be confused with the optical efficacy of the circuit, which
depends upon the LEDs themselves.
Total output power, P
O
, is calculated as:
P
O
=I
F
xV
O
= 0.361 x 49.2 = 17.76W
Conduction loss, P
C
, in the internal MOSFET:
P
C
=(I
F2
xR
DSON
)xD=(0.361
2
x 1.5) x 0.82 = 160 mW
Gate charging and VCC loss, P
G
, in the gate drive and linear
regulator:
P
G
=(I
IN-OP
+f
SW
xQ
G
)xV
IN
P
G
=(600x10
-6
+3x10
5
x3x10
-9
)x60=90mW
Switching loss, P
S
, in the internal MOSFET:
P
S
=0.5xV
IN
xI
F
x(t
R
+t
F
)xf
SW
P
S
= 0.5 x 60 x 0.361 x 40 x 10
-9
x3x10
5
= 130 mW
AC rms current loss, P
CIN
, in the input capacitor:
P
CIN
=I
IN(rms)2
x ESR = (0.134)
2
x 0.006 = 0.1 mW (negli-
gible)
DCR loss, P
L
, in the inductor
P
L
=I
F2
x DCR = 0.35
2
x 1.1 = 135 mW
Recirculating diode loss, P
D
=42mW
Current Sense Resistor Loss, P
SNS
=69mW
Electrical efficiency, η=P
O
/(P
O
+ Sum of all loss terms) =
17.76 / (17.76 + 0.62) = 96%
Temperature Rise in the LM3402HV IC is calculated as:
T
LM3402
=(P
C
+P
G
+P
S
)xθ
JA
= (0.16 + 0.084 + 0.13) x
200 = 74.8˚C
Layout Considerations
The performance of any switching converter depends as
much upon the layout of the PCB as the component selec-
tion. The following guidelines will help the user design a
circuit with maximum rejection of outside EMI and minimum
generation of unwanted EMI.
COMPACT LAYOUT
Parasitic inductance can be reduced by keeping the power
path components close together and keeping the area of the
loops that high currents travel small. Short, thick traces or
copper pours (shapes) are best. In particular, the switch
node (where L1, D1, and the SW pin connect) should be just
large enough to connect all three components without ex-
cessive heating from the current it carries. The LM3402/
02HV operates in two distinct cycles whose high current
paths are shown in Figure 6:
LM3402/LM3402HV
www.national.com19
Layout Considerations (Continued)
The dark grey, inner loop represents the high current path
during the MOSFET on-time. The light grey, outer loop rep-
resents the high current path during the off-time.
GROUND PLANE AND SHAPE ROUTING
The diagram of Figure 6 is also useful for analyzing the flow
of continuous current vs. the flow of pulsating currents. The
circuit paths with current flow during both the on-time and
off-time are considered to be continuous current, while those
that carry current during the on-time or off-time only are
pulsating currents. Preference in routing should be given to
the pulsating current paths, as these are the portions of the
circuit most likely to emit EMI. The ground plane of a PCB is
a conductor and return path, and it is susceptible to noise
injection just as any other circuit path. The continuous cur-
rent paths on the ground net can be routed on the system
ground plane with less risk of injecting noise into other
circuits. The path between the input source and the input
capacitor and the path between the recirculating diode and
the LEDs/current sense resistor are examples of continuous
current paths. In contrast, the path between the recirculating
diode and the input capacitor carries a large pulsating cur-
rent. This path should be routed with a short, thick shape,
preferably on the component side of the PCB. Multiple vias
in parallel should be used right at the pad of the input
capacitor to connect the component side shapes to the
ground plane. A second pulsating current loop that is often
ignored is the gate drive loop formed by the SW and BOOT
pins and capacitor CB. To minimize this loop at the EMI it
generates, keep CB close to the SW and BOOT pins.
CURRENT SENSING
The CS pin is a high-impedance input, and the loop created
by R
SNS
,R
Z
(if used), the CS pin and ground should be
made as small as possible to maximize noise rejection. R
SNS
should therefore be placed as close as possible to the CS
and GND pins of the IC.
REMOTE LED ARRAYS
In some applications the LED or LED array can be far away
(several inches or more) from the LM3402/02HV, or on a
separate PCB connected by a wiring harness. When an
output capacitor is used and the LED array is large or
separated from the rest of the converter, the output capacitor
should be placed close to the LEDs to reduce the effects of
parasitic inductance on the AC impedance of the capacitor.
The current sense resistor should remain on the same PCB,
close to the LM3402/02HV.
20192128
FIGURE 6. Buck Converter Current Loops
LM3402/LM3402HV
www.national.com 20
Layout Considerations (Continued)
TABLE 1. BOM for Design Example 1
ID Part Number Type Size Parameters Qty Vendor
U1 LM3402 LED Driver MSOP-8 40V, 0.5A 1 NSC
L1 SLF7045T-330MR82 Inductor 7.0x7.0 x4.5mm 33µH, 0.82A, 96mΩ1 TDK
D1 CMHSH5-4 Schottky Diode SOD-123 40V, 0.5A 1 Central Semi
Cf VJ0805Y104KXXAT Capacitor 0805 100nF 10% 1 Vishay
Cb VJ0805Y103KXXAT Capacitor 0805 10nF 10% 1 Vishay
Cin C3216X7R1H105M Capacitor 1206 1µF 50V 1 TDK
Co C2012X7R1A225M Capacitor 0805 2.2 µF 10V 1 TDK
Rsns ERJ6BQFR75V Resistor 0805 0.75Ω1% 1 Panasonic
Ron CRCW08055902F Resistor 0805 59.0 kΩ1% 1 Vishay
TABLE 2. BOM for Design Example 2
ID Part Number Type Size Parameters Qty Vendor
U1 LM3402HV LED Driver MSOP-8 75V, 0.5A 1 NSC
L1 DO5022P-684 Inductor 18.5x15.2 x7.1mm 680µH, 1.2A, 1.1Ω1 Coilcraft
D1 CMMSH1-100 Schottky Diode SOD-123F 100V, 1A 1 Central Semi
Cf VJ0805Y104KXXAT Capacitor 0805 100nF 10% 1 Vishay
Cb VJ0805Y103KXXAT Capacitor 0805 10nF 10% 1 Vishay
Cin C4532X7R2A225M Capacitor 1812 2.2µF 100V 1 TDK
Rsns ERJ6BQFR56V Resistor 0805 0.56Ω1% 1 Panasonic
Ron CRCW08051214F Resistor 0805 1.21MΩ1% 1 Vishay
LM3402/LM3402HV
www.national.com21
Physical Dimensions inches (millimeters) unless otherwise noted
8-Lead MSOP Package
NS Package Number MUA08A
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves
the right at any time without notice to change said circuitry and specifications.
For the most current product information visit us at www.national.com.
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LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs
