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FEATURES DESCRIPTION
APPLICATIONS
Vin V 1
U
o
p to 15 V / 400 mA
V 3
U
o
p to 30 V / 20 mA
V 2
U
o
p to 12 V / 20 mA
V 4
3.3
o
V
Vcom
Boost
Converter
Positive Charge
Pump
Negative
Charge Pump
Vcom Buffer
Linear Regulator
Controller
TPS6510x
2.7 V to 5.8 V
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
TRIPLE OUTPUT LCD SUPPLY WITH LINEAR REGULATOR AND VCOM BUFFER
•2.7-V to 5.8-V Input Voltage Range
The TPS6510x series offers a compact and smallpower supply solution that provides all three voltages•1.6-MHz Fixed Switching Frequency
required by thin film transistor (TFT) LCD displays.•3 Independent Adjustable Outputs
The auxiliary linear regulator controller can be used•Main Output of up to 15 V With < 1% Output
to generate a 3.3-V logic power rail for systemsVoltage Accuracy
powered by a 5-V supply rail only.•Virtual Sychronous Converter Technology
The main output, V
O
1, is a 1.6-MHz, fixed-frequency•Negative Regulated Charge Pump Driver V
O
2
PWM boost converter providing the source drivevoltage for the LCD display. The device is available•Positive Charge Pump Converter V
O
3
in two versions with different internal switch current•Integrated VCOM Buffer
limits to allow the use of a smaller external inductor•Auxiliary 3.3-V Linear Regulator Controller
when lower output power is required. TheTPS65100/01 has a typical switch current limit of 2.3•Internal Soft Start
A, and the TPS65105 has a typical switch current•Internal Power-On Sequencing
limit of 1.37 A. A fully integrated adjustable charge•Fault Detection of all Outputs (TPS65100/05)
pump doubler/tripler provides the positive LCD gatedrive voltage. An externally adjustable negative•No Fault Detection (TPS65101)
charge pump provides the negative gate drive•Thermal Shutdown
voltage. Due to the high 1.6-MHz switching•Available in TSSOP-24 and QFN-24
frequency of the charge pumps, inexpensive andPowerPAD™ Packages
small 220-nF capacitors can be used.
The TPS6510x series has an integrated VCOMbuffer to power the LCD backplane. For LCD panels•TFT LCD Displays for Notebooks
powered by 5 V only, the TPS6510x series has a•TFT LCD Displays for Monitors
linear regulator controller using an external transistorto provide a regulated 3.3-V output for the digital•Portable DVD Players
circuits. For maximum safety, the TPS65100/05 goes•Tablet PCs
into shutdown as soon as one of the outputs is out of•Car Navigation Systems
regulation. The device can be enabled again by•Industrial Displays
toggling the input or the enable (EN) pin to GND.The TPS65101 does not enter shutdown when oneof the outputs is below its power good threshold.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of TexasInstruments semiconductor products and disclaimers thereto appears at the end of this data sheet.PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Copyright © 2003–2006, Texas Instruments IncorporatedProducts conform to specifications per the terms of the TexasInstruments standard warranty. Production processing does notnecessarily include testing of all parameters.
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TYPICAL APPLICATION CIRCUIT
VIN
COMP
VCOMIN
EN
ENR
C1+
C1−
DRV
FB2
REF
FB4
BASE
SW
SW
FB1
SUP
C2+
C2−/MODE
OUT3
FB3
VCOM
PGND
PGND
GND
TPS65100
D1
C3
Vo3
up to 30 V/20 mA
C7
C8
C10
Q1
BCP68
D2
D3
C1
R3
R4
C11
C6
C9
R5
R6
C4
C2
C5
R2
R1
C12
C11
10 nF
R7
R8
VI
2.7 to 5.8 V
22 µF
L1
4.7 µHVO1
Up to 15 V/350 mA
VO1
0.22 µF
0.22 µF
0.22 µF
VO2
Up to 12 V/20 mA
220 nF VI
1 µF4.7 µF
VO4
3.3 V
0.22 µF
22 µF
0.22 µF
Vcom
1 µF
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foamduring storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
PACKAGE
(1) (2)LINEAR REGULATOR MINIMUM SWITCH PACKAGET
A
OUTPUT VOLTAGE CURRENT LIMIT MARKINGTSSOP QFN
3.3 V 1.6 A TPS65100PWP TPS65100RGE TPS65100-40 °C to
3.3 V 1.6 A TPS65101PWP TPS65101RGE TPS6510185 °C
3.3 V 0.96 A TPS65105PWP TPS65105RGE TPS65105
(1) The PWP and RGE packages are available taped and reeled. Add an R suffix to the device type (TPS65100PWPR) to order the devicetaped and reeled. The PWPR package has quantities of 2000 devices per reel, and the the RGER package has 3000 devices per reel.Without suffix the PWP package only, is shipped in tubes with 60 devices per tube.(2) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TIWeb site at www.ti.com .
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ABSOLUTE MAXIMUM RATINGS
DISSIPATION RATINGS
RECOMMENDED OPERATING CONDITIONS
ELECTRICAL CHARACTERISTICS
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
over operating free-air temperature range (unless otherwise noted)
(1)
UNIT
Voltages on pin VIN
(2)
-0.3 V to 6 VVoltages on pin V
O
1, SUP, PG
(2)
-0.3 V to 15.5 VVoltages on pin EN, MODE, ENR
(2)
-0.3 V to V
I
+ 0.3 VVoltage on VCOMIN 14 VVoltage on pin SW
(2)
20 VContinuous power dissipation See Dissipation Rating TableOperating junction temperature range -40 °C to 150 °CStorage temperature range -65 °C to 150 °CLead temperature (soldering, 10 sec) 260 °C
(1) Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratingsonly, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operatingconditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.(2) All voltage values are with respect to network ground terminal.
T
A
≤25 °C T
A
= 70 °C T
A
= 85 °CPACKAGE R Θ
JA
POWER RATING POWER RATING POWER RATING
24-Pin TSSOP 30.13 C °/W (PWP soldered) 3.3 W 1.83 W 1.32 W24-Pin QFN 30 C °/W 3.3 W 1.8 W 1.3 W
MIN TYP MAX UNIT
VIN Input voltage range 2.7 5.8 VL Inductor
(1)
4.7 µHT
A
Operating free-air temperature -40 85 °CT
J
Operating junction temperature -40 125 °C
(1) See the application information section for further information.
V
in
= 3.3 V, EN = VIN, V
O
1 = 10 V, T
A
= -40 °C to 85 °C, typical values are at T
A
= 25 °C (unless otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
SUPPLY CURRENT
V
i
Input voltage range 2.7 5.8 VENR = VCOMIN = GND, V
O
3 = 2 x V
O
1, 0.7 0.9 mAI
Q
Quiescent current into VIN
Boost converter not switchingV
O
1 = SUP = 10 V, V
O
3 = 2 x V
O
1 1.7 2.7Charge pump quiescentI
QCharge
mAcurrent into SUP
V
O
1 = SUP = 10 V, V
O
3 = 3 x V
O
1 3.9 6I
QVCOM
VCOM quiescent current into ENR = GND, V
O
1 = SUP = 10 V 750 1300 µASUPI
QEN
LDO controller quiescent ENR = VIN, EN = GND 300 800 µAcurrent into VINI
SD
Shutdown current into VIN EN = ENR = GND 1 10 µAV
UVLO
Undervoltage lockout VIN falling 2.2 2.4 Vthreshold
Thermal shutdown Temperature rising 160 °C
LOGIC SIGNALS EN, ENR
V
IH
High-level input voltage 1.5 VV
IL
Low-level input voltage 0.4 V
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TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
ELECTRICAL CHARACTERISTICS (continued)V
in
= 3.3 V, EN = VIN, V
O
1 = 10 V, T
A
= -40 °C to 85 °C, typical values are at T
A
= 25 °C (unless otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
I
Ileak
Input leakage current EN = GND or VIN 0.01 0.1 µA
MAIN BOOST CONVERTER
V
O
1 Output voltage range 5 15 VMinimum input to outputV
O
1 - V
I
1 Vvoltage differenceV
REF
Reference voltage 1.205 1.213 1.219 VV
FB
Feedback regulation voltage 1.136 1.146 1.154 VI
FB
Feedback input bias current 10 100 nAV
O
1 = 10 V, I
sw
= 500 mA 195 290N-MOSFET on-resistancer
DS(ON)
mΩ(Q1)
V
O
1 = 5 V, I
sw
= 500 mA 285 420TPS65100, TPS65101 1.6 2.3 2.6 AN-MOSFET switch currentI
LIM
limit (Q1)
TPS65105 0.96 1.37 1.56 AV
O
1 = 10 V, I
sw
= 100 mA 9 15P-MOSFET on-resistancer
DS(ON)
Ω(Q2)
V
O
1 = 5 V, I
sw
= 100 mA 14 22Maximum P-MOSFET peakI
MAX
1 Aswitch currentI
leak
Switch leakage current V
sw
= 15 V 1 10 µA0°C≤T
A
≤85 °C 1.295 1.6 2.1f
SW
Oscillator frequency MHz-40 °C≤T
A
≤85 °C 1.191 1.6 2.1Line regulation 2.7 V ≤V
I
≤5.7 V, I
load
= 100 mA 0.012 %/VLoad regulation 0 mA ≤I
O
≤300 mA 0.2 %/A
NEGATIVE CHARGE PUMP V
O
2
V
O
2 Output voltage range -2 VV
ref
Reference voltage 1.205 1.213 1.219 VV
FB
Feedback regulation voltage -36 0 36 mVI
FB
Feedback input bias current 10 100 nAQ8 P-Channel switch r
DS(ON)
4.3 8r
DS(ON)
I
O
= 20 mA ΩQ9 N-Channel switch r
DS(ON)
2.9 4.4I
O
Maximum output current 20 mALine regulation 7 V ≤V
O
1≤15 V, I
load
= 10 mA, V
O
2 = -5 V 0.09 %/VLoad regulation 1 mA ≤I
O
≤20 mA, V
O
2 = -5 V 0.126 %/mA
POSITIVE CHARGE PUMP V
O
3
V
O
3 Output voltage range 30 VV
ref
Reference voltage 1.205 1.213 1.219 VV
FB
Feedback regulation voltage 1.187 1.214 1.238 VI
FB
Feedback input bias current 10 100 nAQ3 P-Channel switch r
DS(ON)
9.9 15.5Q4 N-Channel switch r
DS(ON)
1.1 1.8r
DS(ON)
I
O
= 20 mA ΩQ5 P-Channel switch r
DS(ON)
4.6 8.5Q6 N-Channel switch r
DS(ON)
1.2 2.2D1 – D4 Shottky diodeV
d
I
D1-D4
= 40 mA 610 720 mVforward voltageI
O
Maximum output current 20 mALine regulation 10 V ≤V
O
1≤15 V, I
load
= 10 mA, V
O
3 = 27 V 0.56 %/VLoad regulation 1 mA ≤I
O
≤20 mA, V
O
3 = 27 V 0.05 %/mA
LINEAR REGULATOR CONTROLLER V
O
4
V
O
4 Output voltage 4.5 V ≤V
I
≤5.5V, 10 mA ≤I
O
≤500 mA 3.2 3.3 3.4 V
4
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DEVICE INFORMATION
FB2
REF
GND
DRV
C1−
C1+
1
2
3
4
5
6
18
17
16
15
14
13
7 8 910 11 12
19
20212223
24
Exposed
Thermal Die*
COMP
ENR
EN
FB1
FB4
BASE
VIN
SW
SW
PGND
PGND
SUP
C2−/MODE
C2+
OUT3
FB3
VCOMIN
VCOM
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
FB1
FB4
BASE
VIN
SW
SW
PGND
PGND
SUP
VCOM
VCOMIN
FB3
EN
ENR
COMP
FB2
REF
GND
DRV
C1−
C1+
C2−/MODE
C2+
OUT3
Thermal PAD*
PWP PACKAGE
TOP VIEW RGE PACKAGE
TOP VIEW
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
ELECTRICAL CHARACTERISTICS (continued)V
in
= 3.3 V, EN = VIN, V
O
1 = 10 V, T
A
= -40 °C to 85 °C, typical values are at T
A
= 25 °C (unless otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
V
I
- V
O
4 - V
BE
≥0.5 V
(1)
13.5 19I
BASE
Maximum base drive current mAV
I
- V
O
4 - V
BE
≥0.75 V
(1)
20 27Line regulation 4.75 V ≤V
I
≤5.5 V, I
load
= 500 mA 0.186 %/VLoad regulation 1 mA ≤I
O
≤500 mA, V
I
= 5 V 0.064 %/AStart up current V
O
4≤0.8 V 11 20 25 mA
VCOM BUFFER
V
cm
Common mode input range 2.25 (V
O
1)-2 VV
os
Input offset voltage I
O
= 0 mA -25 +25 mVI
O
=±25 mA -30 37I
O
=±50 mA -45 55DC Load regulation mVI
O
=±100 mA -72 85I
O
=±150 mA -97 110I
B
VCOMIN Input bias current -300 -30 300 nAV
O
1 = 15 V 1.2 AI
peak
Peak output current V
O
1 = 10 V 0.65 AV
O
1 = 5 V 0.15 A
FAULT PROTECTION THRESHOLDS
-8.75%V
(th, Vo1)
V
O
1 Rising -12 -6 VV
O
1Shutdown thresholdV
(th, Vo2)
V
O
2 Rising -13 -9% V
O
2 -5 VV
(th, Vo3)
V
O
3 Rising -11 -8% V
O
3 -5 V
(1) With V
I
= supply voltage of the TPS6510x, V
O
4 = output voltage of the regulator, V
BE
= basis emitter voltage of external transistor
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TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
DEVICE INFORMATION (continued)Terminal Functions
TERMINAL
I/O DESCRIPTIONNO. NO.NAME
(PWP) (RGE)
VIN 4 7 I Input voltage pin of the deviceEnable pin of the device. This pin should be terminated and not be left floating. A logicEN 24 3 I
high enables the device and a logic low shuts down the device.COMP 22 1 Compensation pin for the main boost converter. A small capacitor is connected to thispin.
Positive input terminal of the VCOM buffer. When the VCOM buffer is not used, thisVCOMIN 11 14 I
terminal can be connected to GND to reduce the overall quiescent current of the IC.Enable pin of the linear regulator controller. This pin should be terminated and not beENR 23 2 I left floating. Logic high enables the regulator and a logic low puts the regulator inshutdown.C1+ 16 19 Positive terminal of the charge pump flying capacitorC1- 17 20 Negative terminal of the charge pump flying capacitorDRV 18 21 O External charge pump driverFB2 21 24 I Feedback pin of negative charge pumpREF 20 23 O Internal reference output typically 1.23 VFB4 2 5 I Feedback pin of the linear regulator controller. The linear regulator controller is set toa fixed output voltage of 3.3 V or 3 V depending on the version.BASE 3 6 O Base drive output for the external transistorGND 19 22 GroundPGND 7, 8 10, 11 Power groundVCOM 10 13 O VCOM buffer outputFB3 12 15 I Feedback pin of positive charge pumpOUT3 13 16 O Positive charge pump outputNegative terminal of the charge pump flying capacitor and charge pump MODE pin. Ifthe flying capacitor is connected to this pin, the converter operates in a voltage triplerC2-/MODE 15 18
mode. If the charge pump needs to operate in a voltage doubler mode, the flyingcapacitor is removed and the C2-/MODE pin should be connected to GND.Positive terminal for the charge pump flying capacitor. If the device runs in voltageC2+ 14 17
doubler mode, this pin should be left open.Supply pin of the positive, negative charge pump, boost converter gate drive circuit,and VCOM buffer. This pin should be connected to the output of the main boostSUP 9 12 I converter and cannot be connected to any other voltage source. For performancereasons, it is not recommended for a bypass capacitor to be connected directly to thispin.FB1 1 4 I Feedback pin of the boost converterSW 5, 6 8, 9 I Switch pin of the boost converterPowerPAD™/ The PowerPad or exposed thermal die needs to be connected to the power groundThermal Die pins (PGND)
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1.6-MHz
Oscillator
D
S
VFB
1.146 V
Comparator
GM Amplifier Sawtooth
Generator
VFB
1.146 V
D
S
D
S
D
S
D
S
Vref
1.213 V
Vin SUP
~1 V
D
S
SUP
SUP
D
S
D
S
SUP
C1−
C1+
Vo3
C2+
C2−
D
S
Vref
1.214 V
Soft Start
Vref
1.213 V
SUP
ENR BASE
FB4
REF
FB2
DRV
COMP
FB1
VIN SW SW
FB3
VCOM
VCOMIN GND PGND PGND
Q1
Q3
Q4
Q5
Q6
Q7
D
S
D
S
Q8
Q9
Q10
Q11
Q12
SUP
EN
VCOM
Buffer
Linear
Regulator
Controller
Reference
Output
Negative
Charge Pump
Positive
Charge Pump
Main boost
converter
Vref
0 V
D1
D4
D2
D3
Disable
GM Amplifier
Low Gain
FB1
FB2
FB3
D
S
Q2
SUP
Bias Vref = 1.213 V
Thermal Shutdown
Start-Up Sequencing
Undervoltage Detection
Short-Circuit Protection
Current Limit
and
Soft Start
Control Logic
Gate Drive Circuit
SUP
(VO)
Current
Control
Gain Select
(Doubler or
Tripple Mode)
Soft Start
Iref = 20 mA Short Circuit
Detect Soft Start
Current
Control
Soft Start
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
FUNCTIONAL BLOCK DIAGRAM
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TYPICAL CHARACTERISTICS
Table of Graphs
10
20
30
40
50
60
70
80
90
100
1 10 100 1 k
Vo1 = 6 V
IL − Load Current − mA
Efficiency − %
Vo1 = 15 V
Vo1 = 10 V
VI = 3.3 V
Vo2, Vo3 = No Load, Switching
10
20
30
40
50
60
70
80
90
100
1 10 100 1 k
IL − Load Current − mA
Efficiency − %
Vo1 = 15 V
Vo1 = 10 V
VI = 5 V
Vo2, Vo3 = No Load, Switching
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
FIGURE
Main Boost Converter
Efficiency, main boost converter V
O
1 vs Load current 1ηEfficiency, main boost converter V
O
1 vs Load current 2Efficiency vs Input voltage 3f
sw
Switching frequency vs Free-air temperature 4r
DS(on)
r
DS(on)
N-Channel main switch Q1 vs Free-air temperature 5PWM operation, continuous mode 6PWM operation at light load 7Load transient response, C
O
= 22 µF 8Load transient response, C
O
= 2 x 22 µF 9Power-up sequencing 10Soft start V
O
1 11
Negative Charge Pump
I
max
V
O
2 Maximum load current vs Output voltage V
O
1 12
Positive Charge Pump
I
max
V
O
3 Maximum load current vs Output voltage V
O
1 (doubler mode) 13I
max
V
O
3 Maximum load current vs Output voltage V
O
1 (tripler mode) 14
EFFICIENCY EFFICIENCY EFFICIENCYvs vs vsLOAD CURRENT LOAD CURRENT INPUT VOLTAGE
Figure 1. Figure 2. Figure 3.
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VSW
10 V/div
VO
50 mV/div
VI = 3.3 V
VO = 10 V/300 mA
IL
1 A/div
250 ns/div
100
150
200
250
300
350
−40 −20 0 20 40 60 80 100
TA − Free-Air Temperature − °C
Vo1 = 5 V
Vo1 = 15 V
Vo1 = 10 V
− N−Channel Main Switch − mΩrDS(on)
1.3
1.4
1.5
1.6
1.7
1.8
1.9
−40 −20 0 20 40 60 80 100
TA − Free-Air Temperature − °C
Switching Frequency − MHz
VI = 2.7 V
VI = 3.3 V
VI = 5.8 V
Vo1
200 mV/div
VI = 3.3 V
Vo1 = 10 V, CO= 22 µF
IO
50 mA to 250 mA
100 µs/div
Vo1
100 mV/div
VI = 3.3 V
Vo1 = 10 V, CO= 2*22 µF
IO
50 mA to 250 mA
100 µs/div
VSW
10 V/div
VO
50 mV/div
VI = 3.3 V
VO = 10 V/10 mA
IL
500 mA/div
250 ns/div
Vo1
5 V/div
VI = 3.3 V
VO = 10 V,
IO = 300 mA
500 µs/div
II
500
mA/div
Vo1
5 V/div
VI = 3.3 V
VO = 10 V,
VCOM CI = 1 nF
500 µs/div
Vo2
5 V/div
Vo3
10 V/div
VCOM
2 V/div
0
0.02
0.04
0.06
0.08
0.10
0.12
0.14
0.16
0.18
0.20
8.8 9.8 10.8 11.8 12.8 13.8 14.8
Vo1 − Output V oltage − V
− Output Current − AIO
Vo2 = −8 V
TA = −40°C
TA = 25°C
TA = 85°C
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
SWITCHING FREQUENCY r
DS(on)
N-CHANNEL MAIN SWITCH PWM OPERATION CONTINUOUSvs vs MODEFREE-AIR TEMPERATURE FREE-AIR TEMPERATURE
Figure 4. Figure 5. Figure 6.
PWM OPERATION AT LIGHT LOAD LOAD TRANSIENT RESPONSE LOAD TRANSIENT RESPONSE
Figure 7. Figure 8. Figure 9.
POWER-UP SEQUENCING SOFT START V
O
1 V
O
2 MAXIMUM LOAD CURRENT
Figure 10. Figure 11. Figure 12.
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0
0.02
0.04
0.06
0.08
0.10
0.12
910 11 12 13 14 15
Vo1 − Output V oltage − V
− Output Current − AIO
Vo3 = 28 V (Tripler Mode)
TA = 25°C
TA = 85°C
TA = −40°C
DETAILED DESCRIPTION
Main Boost Converter
VCOM Buffer
Enable and Power On Sequencing (EN, ENR)
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
V
O
3 MAXIMUM LOAD CURRENT V
O
3 MAXIMUM LOAD CURRENT
Figure 13. Figure 14.
The TPS6510x series consists of a main boost converter operating with a fixed switching frequency of 1.6 MHzto allow for small external components. The boost converter output voltage V
O
1 is also the input voltage,connected via the pin SUP, for the positive and negative charge pumps and the bias supply for the VCOMbuffer. The linear regulator controller is independent from this system with its own enable pin. This allows thelinear regulator controller to continue to operate while the other supply rails are disabled or in shutdown due to afault condition on one of their outputs. See the functional block diagram for more information.
The main boost converter operates with PWM and a fixed switching frequency of 1.6 MHz. The converter uses aunique fast response, voltage mode controller scheme with input voltage feedforward. This achieves excellentline and load regulation (0.2% A load regulation typical) and allows the use of small external components. Toadd higher flexibility to the selection of external component values the device uses external loop compensation.Although the boost converter looks like a nonsynchronous boost converter topology operating in discontinuousmode at light load, the TPS6510x series maintains continuous conduction even at light load currents. This isaccoplished using the Virtual Synchronous Converter Technology for improved load transient response. Thisarchitecture uses an external Schottky diode and an integrated MOSFET in parallel connected between SW andSUP (see the functional block diagram). The integrated MOSFET Q2 allows the inductor current to becomenegative at light load conditions. For this purpose, a small integrated P-channel MOSFET with typically 10 Ωr
DSon
is sufficient. When the inductor current is positive, the external Schottky diode with the lower forwardvoltage conducts the current. This causes the converter to operate with a fixed frequency in continuousconduction mode over the entire load current range. This avoids the ringing on the switch pin as seen with astandard nonsynchronous boost converter and allows a simpler compensation for the boost converter.
VCOMIN is the input of the VCOM buffer. If the VCOM buffer is not required for certain applications, it ispossible to shut down the VCOM buffer by statically connecting VCOMIN to ground, reducing the overallquiescent current. The VCOM buffer features soft start avoiding a large voltage drop at V
O
1 during start-up. TheVCOMIN cannot be pulled dynamically to ground during operation.
The device has two enable pins. These pins should be terminated and not left floating to prevent unpredictableoperation. Pulling the enable pin (EN) high enables the device and starts the power on sequencing with the mainboost converter V
O
1 coming up first then the negative and positive charge pump and the VCOM buffer. If theVCOMIN pin is held low, the VCOM buffer remains disabled. The linear regulator has an independent enable pin(ENR). Pulling this pin low disables the regulator, and pulling this pin high enables this regulator.
If the enable pin EN is pulled high, the device starts its power on sequencing. The main boost converter startsup first with its soft start. If the output voltage has reached 91.25% of its output voltage, the negative charge
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Positive Charge Pump
Negative Charge Pump
Linear Regulator Controller
Soft Start
Fault Protection
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
DETAILED DESCRIPTION (continued)pump comes up next. The negative charge pump starts with a soft start and when the output voltage hasreached 91% of the nominal value, the positive charge pump comes up with a soft start. The VCOM buffer isenabled as soon as the positive charge pump has reached its nominal value and VCOMIN is greater thantypically 1.0 V. Pulling the enable pin low shuts down the device. Depended on load current and outputcapacitance, each of the outputs goes down.
The TPS6510x series has a fully regulated integrated positive charge pump generating V
O
3. The input voltagefor the charge pump is applied to the SUP pin that is equal to the output of the main boost converter V
O
1. Thecharge pump is capable of supplying a minimum load current of 20 mA. Depending on the voltage differencebetween V
O
1 and V
O
3 higher load currents are possible. See Figure 13 and Figure 14.
The TPS6510x series has a regulated negative charge pump using two external Schottky diodes. The inputvoltage for the charge pump is applied to the SUP pin that is connected to the output of the main boostconverter V
O
1. The charge pump inverts the main boost converter output voltage and is capable of supplying aminimum load current of 20 mA. Depending on the voltage difference between V
O
1 and V
O
2, higher loadcurrents are possible. See Figure 12.
The TPS6510x series includes a linear regulator controller to generate a 3.3-V rail which is useful when thesystem is powered from a 5-V supply. The regulator is independent from the other voltage rails of the device andhas its own enable (ENR). Since most of the systems require this voltage rail to come up first it is recommendedto use a R-C delay on EN. This delays the start-up of the main boost converter which will reduce the inrushcurrent as well.
The main boost converter as well as the charge pumps, linear regulator, and VCOM buffer have an internal softstart. This avoids heavy voltage drops at the input voltage rail or at the output of the main boost converter V
O
1during start-up caused by high inrush currents. See Figure 10 and Figure 11. During softstart of the main boostconverter V
O
1 the internal current limit threshold is increased in three steps. The device starts with the first stepwhere the current limit is set to 2/5 of the typical current limit (2/5 of 2.3A) for 1024 clock cycles then increasedto 3/5 of the current limit for 1024 clock cycles and the 3rd step is the full current limit. The TPS65101 has anextended softstart time where each step is 2048 clock cycles.
All the outputs of the TPS65100/05 have short circuit detection where the device enters shutdown. TheTPS65101, as an exception, does not enter shutdown in case one of the outputs falls below its power goodthreshold. The main boost converter has overvoltage and undervoltage protection. If the output voltage V
O
1 risesabove the overvoltage protection threshold of typically 5% of V
O
1, then the device stops switching but remainsoperational. When the output voltage falls below this threshold again, then the converter continues operation.When the output voltage falls below power good threshold of typically 8.75% of V
O
1, in case of a short circuitcondition, then the TPS65100/05 goes into shutdown. Because there is a direct pass from the input to the outputthrough the diode, the short circuit condition remains. If this condition needs to be avoided, a fuse at the input oran output disconnect using a single transistor and resistor is required. The negative and positive charge pumpshave an undervoltage lockout to protect the LCD panel of possible latch-up conditions in case of a short circuitcondition or faulty operation. When the negative output voltage is typically above 9.5% of its output voltage(closer to ground), then the device enters shutdown. When the positive charge pump output voltage V
O
3 isbelow 8% typ of its output voltage, then the device goes into shutdown as well. See the electrical characteristicstable under fault protection thresholds. The device can be enabled again by toggling the enable pin (EN) below0.4 V or by cycling the input voltage below the UVLO of 1.7 V. The linear regulator reduces the output current totypical 20 mA under a short circuit condition when the output voltage is typically < 1 V. See the functional blockdiagram. The linear regulator does not go into shutdown under a short-circuit condition.
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Thermal Shutdown
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
DETAILED DESCRIPTION (continued)
A thermal shutdown is implemented to prevent damage due to excessive heat and power dissipation. Typically,the thermal shutdown threshold is 160 °C. If this temperature is reached, the device goes into shutdown. Thedevice can be enabled by toggling the enable pin to low and back to high or by cycling the input voltage to GNDand back to V
I
again.
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APPLICATION INFORMATION
BOOST CONVERTER DESIGN PROCEDURE
DVout VDVin
Vout VDVsw 10 V 0.8 V 3.3 V
10 V 0.8 V 0.5 V 0.73
iLVin VswD
fsL(3.3 V 0.5 V) 0.73
1.6 MHz 4.2 H304 mA
Iswpeak IL
iL
21.11 A 304 mA
21.26 A
Inductor Selection
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
The first step in the design procedure is to calculate the maximum possible output current of the main boostconverter under certain input and output voltage conditions. The following example is for a 3.3-V to 10-Vconversion:
V
in
= 3.3 V, V
out
= 10 V, Switch voltage drop V
sw
= 0.5 V, Schottky diode forward voltage V
D
= 0.8 V1. Duty cycle:
2. Average inductor current:
3. Inductor peak-to-peak ripple current:
4. Peak switch current:
The integrated switch, the inductor, and the external Schottky diode must be able to handle the peak switchcurrent. The calculated peak switch current has to be equal to or lower than the minimum N-MOSFET switchcurrent limit as specified in the electrical characteristics table (1.6 A for the TPS65100/01 and 0.96 A for theTPS65105). If the peak switch current is higher, then the converter cannot support the required load current.This calculation must be done for the minimum input voltage where the peak switch current is highest. Thecalculation includes conduction losses like switch r
DSon
(0.5 V) and diode forward drop voltage losses (0.8 V).Additional switching losses, inductor core and winding losses, etc., require a slightly higher peak switch currentin the actual application. The above calculation still allows a for good design and component selection.
Several inductors work with the TPS6510x series. Especially with the external compensation, the performancecan be adjusted to the specific application requirements. The main parameter for inductor selection is thesaturation current of the inductor which should be higher than the peak switch current as calculated above withadditional margin to cover for heavy load transients and extreme start-up conditions. Another method is tochoose the inductor with a saturation current at least as high as the minimum switch current limit of 1.6 A for theTPS65100/01 and 0.96 A for the TPS65105. The different switch current limits allow selection of a physicallysmaller inductor when less output current is required. The second important parameter is inductor dc resistance.Usually, the lower the DC resistance, the higher the efficiency. However, inductor DC resistance, is not the onlyparameter determining the efficiency. Especially for a boost converter where the inductor is the energy storageelemen,t the type and material of the inductor influences the efficiency as well. Especially at the high switchingfrequency of 1.6 MHz, inductor core losses, proximity effects, and skin effects become more important. Usually,an inductor with a larger form factor yields higher efficiency. The efficiency difference between different inductorscan vary between 2% to 10%. For the TPS6510x series, inductor values between 3.3 µH and 6.8 µH are a goodchoice but other values can be used as well. Possible inductors are shown in Table 1 .
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Output Capacitor Selection
Vout Iout
Cout 1
fsIpL
Vout VdVinIpESR
Input Capacitor Selection
Rectifier Diode Selection
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
APPLICATION INFORMATION (continued)Table 1. Inductor Selection
DEVICE INDUCTOR VALUE COMPONENT SUPPLIER DIMENSIONS / mm ISAT/DCR
4.7 µH Coilcraft DO1813P-472HC 8,89 x 6,1 x 5 2.6 A/54 m Ω4.2 µH Sumida CDRH5D28 4R2 5,7 x 5,7 x 3 2.2 A/23 m Ω4.7 µH Sumida CDC5D23 4R7 6 x 6 x 2,5 1.6 A/48 m ΩTPS65100
3.3 µH Wuerth Elektronik 744042003 4,8 x 4,8 x 2 1.8 A/65 m Ω4.2 µH Sumida CDRH6D12 4R2 6,5 x 6,5 x 1,5 1.8 A/60 m Ω3.3 µH Sumida CDRH6D12 3R3 6,5 x 6,5 x 1,5 1.9 A/50 m Ω3.3 µH Sumida CDPH4D19 3R3 5,1 x 5,1 x 2 1.5 A/26 m Ω3.3 µH Coilcraft DO1606T-332 6,5 x 5,2 x 2 1.4 A/120 m ΩTPS65105 3.3 µH Sumida CDRH2D18/HP 3R3 3,2 x 3,2 x 2 1.45 A/69 m Ω4.7 µH Wuerth Elektronik 744010004 5,5 x 3,5 x 1 1.0 A/260 m Ω3.3 µH Coilcraft LPO6610-332M 6,6 x 5,5 x 1 1.3 A/160 m Ω
For the best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have alow ESR value, but depending on the application, tantalum capacitors can be used as well. A 22- µF ceramicoutput capacitor works for most of the applications. Higher capacitor values can be used to improve the loadtransient regulation. See Table 2 for selection of the output capacitor. The output voltage ripple can becalculated as:
with:
IP = Peak switch current as calculated in the previous section with I
SW(peak)
.L = Selected inductor valueI
OUT
= Normal load currentf
s
= Switching frequencyV
d
= Rectifier diode forward voltage (typical 0.3 V)C
OUT
= Selected output capacitorESR = Output capacitor ESR value
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 22- µF ceramic input capacitoris sufficient for most of the applications. For better input voltage filtering, this value can be increased. SeeTable 2 and the typical applications for input capacitor recommendations.
Table 2. Input and Output Capacitors Selection
CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER COMMENTS
22 µF/1210 16 V Taiyo Yuden EMK325BY226MM C
out
22 µF/1206 6.3 V Taiyo Yuden JMK316BJ226 C
in
To achieve high efficiency, a Schottky diode should be used. The voltage rating should be higher than themaximum output voltage of the converter. The average forward current should be equal to the average inductorcurrent of the converter. The main parameter influencing the efficiency of the converter is the forward voltageand the reverse leakage current of the diode; both should be as low as possible. Possible diodes are: OnSemiconductor MBRM120L, Microsemi UPS120E, and Fairchild Semiconductor MBRS130L.
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Converter Loop Design and Stability
Design Procedure Quick Steps
Setting the Output Voltage and Selecting the Feedforward Capacitor
Vout 1.146 V 1R1
R2
R1
430 kΩ
SW
SW
FB1
SUP
C2+
C2−/MODE
D1
C8
6.8 pF C4
22 µF
C2 0.22 µFR2
56 kΩ
VO1
Up to 10 V/150 mA
ƒz1
2C8 R1
TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
The TPS6510x series converter loop can be externally compensated and allows access to the internaltransconductance error amplifier output at the COMP pin. A small feedforward capacitor across the upperfeedback resistor divider speeds up the circuit as well. To test the converter stability and load transientperformance of the converter, a load step from 50 mA to 250 mA is applied, and the output voltage of theconverter is monitored. Applying load steps to the converter output is a good tool to judge the stability of such aboost converter.
1. Select the feedback resistor divider to set the output voltage.2. Select the feedforward capacitor to place a zero at 50 kHz.3. Select the compensation capacitor on pin COMP. The smaller the value, the higher the low frequency gain.4. Use a 50-k Ωpotentiometer in series to C
c
and monitor V
out
during load transients. Fine tune the loadtransient by adjusting the potentiometer. Select a resistor value that comes closest to the potentiometerresistor value. This needs to be done at the highest V
in
and highest load current since the stability is mostcritical at these conditions.
The output voltage is set by the external resistor divider and is calculated as:
Across the upper resistor a bypass capacitor is required to speed up the circuit during load transients as shownin Figure 15 .
Figure 15. Feedforward Capacitor
Together with R1 the bypass capacitor C8 sets a zero in the control loop at approximately 50 kHz:
A value closest to the calculated value should be used. Larger feedforward capacitor values reduce the loadregulation of the converter and cause load steps as shown in Figure 16 .
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Load Step
Compensation
VIN
COMP
RC
15 kΩ
CC
1 nF
fz1
2Cc Rc
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
Figure 16. Load Step Caused By A Too Large Feedforward Capacitor Value
The regulator loop can be compensated by adjusting the external components connected to the COMP pin. TheCOMP pin is connected to the output of the internal transconductance error amplifier. A typical compensationscheme is shown in Figure 17 .
Figure 17. Compensation Network
The compensation capacitor C
C
adjusts the low frequency gain and the resistor value adjusts the high frequencygain. The formula below calculates at what frequency the resistor increases the high frequency gain.
Lower input voltages require a higher gain and therefore a lower compensation capacitor value. A good start isC
C
= 1 nF for a 3.3-V input and C
C
= 2.2 nF for a 5-V input. If the device operates over the entire input voltagerange from 2.7 V to 5.8 V, a large compensation capacitor up to 10 nF is recommended. Figure 18 shows theload transient with a larger compensation capacitor, and Figure 19 shows a smaller compensation capacitor.
Figure 18. C
c
= 4.7 nF
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TPS65100 , TPS65101
TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
Figure 19. C
c
= 1 nF
Lastly, R
c
needs to be selected. A good practice is to use a 50-k Ωpotentiometer and adjust the potentiometerfor best load transient where no oscillations should occur. These tests have to be done at the highest V
in
andhighest load current because converter stability is most critical under these conditions. Figure 20 ,Figure 21 , andFigure 22 show the fine tuning of the loop with R
c
.
Figure 20. Overcompensated (Damped Oscillation), R
c
Is Too Large
Figure 21. Undercompensated (Loop Is Too Slow), R
c
Is Too Small
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V = -V x
OUT REF = -1.213 V x
RR
33
RR
44
R3 = R4 x = R4 x
|VOUT||VOUT|
VREF 1.213
TPS65100 , TPS65101TPS65105
SLVS496C – SEPTEMBER 2003 – REVISED APRIL 2006
Figure 22. Optimum, R
c
Is Ideal
Negative Charge Pump
The negative charge pump provides a regulated output voltage by inverting the main output voltage V
O
1. Thenegative charge pump output voltage is set with external feedback resistors.
The maximum load current of the negative charge pump depends on the voltage drop across the externalSchottky diodes, the internal on resistance of the charge pump MOSFETS Q8 and Q9, and the impedance ofthe flying capacitor C12. When the voltage drop across these components is larger than