LT1110
1
D
U
ESCRIPTIO
The LT1110 is a versatile micropower DC-DC converter.
The device requires only three external components to
deliver a fixed output of 5V or 12V. The very low minimum
supply voltage of 1.0V allows the use of the LT1110 in
applications where the primary power source is a single
cell. An on-chip auxiliary gain block can function as a low
battery detector or linear post regulator.
The 70kHz oscillator allows the use of surface mount
inductors and capacitors in many applications. Quiescent
current is just 300µA, making the device ideal in remote or
battery powered applications where current consumption
must be kept to a minimum.
The device can easily be configured as a step-up or
step-down converter, although for most step-down
applications or input sources greater than 3V, the LT1111
is recommended. Switch current limiting is user-adjustable
by adding a single external resistor. Unique reverse battery
protection circuitry limits reverse current to safe, non-
destructive levels at reverse supply voltages up to 1.6V.
S
FEATURE
■
Operates at Supply Voltages From 1.0V to 30V
■
Works in Step-Up or Step-Down Mode
■
Only Three External Off-the-Shelf Components
Required
■
Low-Battery Detector Comparator On-Chip
■
User-Adjustable Current Limit
■
Internal 1A Power Switch
■
Fixed or Adjustable Output Voltage Versions
■
Space-Saving 8-Pin MiniDIP or S8 Package
U
S
A
O
PPLICATI
■
Pagers
■
Cameras
■
Single-Cell to 5V Converters
■
Battery Backup Supplies
■
Laptop and Palmtop Computers
■
Cellular Telephones
■
Portable Instruments
■
Laser Diode Drivers
■
Hand-Held Inventory Computers
Micropower
DC-DC Converter
Adjustable and Fixed 5V, 12V
LOAD CURRENT (mA)
0
EFFICIENCY (%)
50
60
70
80
85
90
10 20 30 40
LT1110 • TA02
75
65
55
5152535
V
IN
= 1.50V
V
IN
= 1.25V
V
IN
= 1.00V
U
A
O
PPLICATITYPICAL
Efficiency
All Surface Mount
Single Cell to 5V Converter
LT1110 • TA01
+
GND SW2
SENSE
SW1
LIM
I
IN
V
54
1
3
8
LT1110-5
1.5V
AA CELL*
*ADD 10 F DECOUPLING CAPACITOR IF BATTERY
IS MORE THAN 2" AWAY FROM LT1110.
µ
15µF
TANTALUM
5V
MBRS120T3
SUMIDA
CD54-470K
47µH
OPERATES WITH CELL VOLTAGE 1.0V
≥
2
LT1110
2
WU
U
PACKAGE/ORDER I FOR ATIO
A
U
G
W
A
W
U
W
ARBSOLUTEXI T
I
S
Supply Voltage, Step-Up Mode................................ 15V
Supply Voltage, Step-Down Mode ........................... 36V
SW1 Pin Voltage...................................................... 50V
SW2 Pin Voltage.........................................–0.5V to V
IN
Feedback Pin Voltage (LT1110) .............................. 5.5V
Switch Current........................................................ 1.5A
Maximum Power Dissipation ............................. 500mW
Operating Temperature Range ..................... 0°C to 70°C
Storage Temperature Range ..................–65°C to 150°C
Lead Temperature (Soldering, 10 sec.)................. 300°C
ORDER PART
NUMBER
LT1110CN8
LT1110CN8-5
LT1110CN8-12
1110
11105
11012
Consult factory for Industrial and Military grade parts.
T
JMAX
= 90°C, θ
JA
= 150°C/W
1
2
3
4
8
7
6
5
TOP VIEW
I
LIM
V
IN
SW1
SW2
FB (SENSE)*
SET
A0
GND
N8 PACKAGE
8-LEAD PLASTIC DIP
*FIXED VERSIONS
T
JMAX
= 90°C, θ
JA
= 130°C/W
1
2
3
4
8
7
6
5
TOP VIEW
FB (SENSE)*
SET
A0
GND
I
LIM
V
IN
SW1
SW2
S8 PACKAGE
8-LEAD PLASTIC SOIC
*FIXED VERSIONS
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
I
Q
Quiescent Current Switch Off ●300 µA
V
IN
Input Voltage Step-Up Mode ●1.15 12.6 V
1.0 12.6 V
Step-Down Mode ●30 V
Comparator Trip Point Voltage LT1110 (Note 1) ●210 220 230 mV
V
OUT
Output Sense Voltage LT1110-5 (Note 2) ●4.75 5.00 5.25 V
LT1110-12 (Note 2) ●11.4 12.00 12.6 V
Comparator Hysteresis LT1110 ●48 mV
Output Hysteresis LT1110-5 ●90 180 mV
LT1110-12 ●200 400 mV
f
OSC
Oscillator Frequency ●52 70 90 kHz
DC Duty Cycle Full Load (V
FB
< V
REF
)●62 69 78 %
t
ON
Switch ON Time ●7.5 10 12.5 µs
I
FB
Feedback Pin Bias Current LT1110, V
FB
= 0V ●70 150 nA
I
SET
Set Pin Bias Current V
SET
= V
REF
●100 300 nA
V
AO
AO Output Low I
AO
= –300µA, V
SET
= 150mV ●0.15 0.4 V
Reference Line Regulation 1.0V ≤ V
IN
≤ 1.5V ●0.35 1.0 %/V
1.5V ≤ V
IN
≤ 12V ●0.05 0.1 %/V
ELECTRICAL C CHARA TERISTICS
TA = 25°C, VIN = 1.5V, unless otherwise noted.
S8 PART MARKING
LT1110CS8
LT1110CS8-5
LT1110CS8-12
LT1110
3
V
CESAT
Switch Saturation Voltage V
IN
= 1.5V, I
SW
= 400mA 300 400 mV
Step-Up Mode ●600 mV
V
IN
= 1.5V, I
SW
= 500mA 400 550 mV
●750 mV
V
IN
= 5V, I
SW
= 1A 700 1000 mV
A
V
A2 Error Amp Gain R
L
= 100kΩ (Note 3) ●1000 5000 V/V
I
REV
Reverse Battery Current (Note 4) 750 mA
I
LIM
Current Limit 220Ω Between I
LIM
and V
IN
400 mA
Current Limit Temperature – 0.3 %/°C
Coefficient
I
LEAK
Switch OFF Leakage Current Measured at SW1 Pin 1 10 µA
V
SW2
Maximum Excursion Below GND I
SW1
≤ 10µA, Switch Off – 400 – 350 mV
ELECTRICAL C CHARA TERISTICS
TA = 25°C, VIN = 1.5V, unless otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Note 3: 100kΩ resistor connected between a 5V source and the AO pin.
Note 4: The LT1110 is guaranteed to withstand continuous application of
+1.6V applied to the GND and SW2 pins while V
IN
, I
LIM
, and SW1 pins are
grounded.
The ● denotes the specifications which apply over the full operating
temperature range.
Note 1: This specification guarantees that both the high and low trip point
of the comparator fall within the 210mV to 230mV range.
Note 2: This specification guarantees that the output voltage of the fixed
versions will always fall within the specified range. The waveform at the
sense pin will exhibit a sawtooth shape due to the comparator hysteresis.
CCHARA TERISTICS
UW
AT
Y
P
I
CALPER
F
O
RC
E
Oscillator Frequency Oscillator Frequency Switch On Time
INPUT VOLTAGE (V)
0
64
FREQUENCY (KHz)
66
70
72
74
78
80
36912
LT1110 • TPC02
15 18 21
62
60 24 27 30
68
76
TEMPERATURE (°C)
ON TIME (µs)
–50 –25 0 25
LT1110 • TPC03
50 75 100
14
13
12
11
10
9
8
7
TEMPERATURE (°C)
–50
40
OSCILLATOR FREQUENCY (KHz)
50
60
70
80
90
100
–25 0 25 50
LT1110 • TPC01
75 100
LT1110
4
I
SWITCH
(A)
ON VOLTAGE (V)
0 0.2 0.4 0.6
LT1110 • TPC07
0.8 1.0
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
V
IN
= 12V
CCHARA TERISTICS
UW
AT
Y
P
I
CALPER
F
O
RC
E
Saturation Voltage
Duty Cycle Switch Saturation Voltage Step-Up Mode
Switch On Voltage Minimum/Maximum Frequency vs
Step-Down Mode On Time Quiescent Current
Maximum Switch Current vs Maximum Switch Current vs
Quiescent Current RLIM Step-Up RLIM Step-Down
TEMPERATURE (°C)
DUTY CYCLE (%)
–50 –25 0 25
LT1110 • TPC04
50 75 100
78
76
74
72
70
68
66
64
62
60
58
TEMPERATURE (°C)
V
CESAT
(mV)
–50 – 25 0 25
LT1110 • TPC05
50 75 100
500
450
400
350
300
250
200
150
100
50
0
V
IN
= 1.5V
I
SW
= 500mA
I (A)
0
0
V (V)
0.2
0.4
0.6
1.2
1.4
0.2 0.4 0.8 1.2
LT1110 • TPC06
1.0
1.4 1.6
SWITCH
CESAT
V
IN
= 1.0V
V = 1.2V
IN
V
IN
= 1.5V
V
IN
= 5.0V
V = 2.0V
IN
0.6 1.0
0.8
V
IN
= 3.0V
INPUT VOLTAGE (V)
QUIESCENT CURRENT (µA)
0
LT1110 • TPC09
3
400
380
360
340
320
280
260
6
240
220
200
300
912151821242730
TEMPERATURE (°C)
QUIESCENT CURRENT (µA)
–50
LT1110 • TPC10
500
450
400
350
250
–25
150
100
200
300
0 25 50 75 100
RLIM (Ω)
SWITCH CURRENT (A)
10
LT1110 • TPC11
100
1.5
1.3
1.1
0.9
1000
0.7
0.5
0.3
0.1
STEP-UP MODE
VIN ≤ 5V
RLIM (Ω)
SWITCH CURRENT (A)
10
LT1110 • TPC12
100
1.5
1.3
1.1
0.9
1000
0.7
0.5
0.3
0.1
STEP-DOWN MODE
VIN = 12V
SWITCH ON TIME (µs)
OSCILLATOR FREQUENCY (KHz)
79
LT1110 • TPC08
10
100
90
80
70
60
50
40 13
0°C ≤ T
A
≤ 70°C
811 12
95
85
75
65
55
45
LT1110
5
W
IDAGRA
B
L
O
C
K
LT1110
CCHARA TERISTICS
UW
AT
Y
P
I
CALPER
F
O
RC
E
Set Pin Bias Current FB Pin Bias Current Reference Voltage
I
LIM
(Pin 1): Connect this pin to V
IN
for normal use. Where
lower current limit is desired, connect a resistor between
I
LIM
and V
IN
. A 220Ω resistor will limit the switch current
to approximately 400mA.
V
IN
(Pin 2): Input supply voltage.
SW1 (Pin 3):
Collector of power transistor. For step-up
mode connect to inductor/diode. For step-down mode
connect to V
IN
.
SW2 (Pin 4):
Emitter of power transistor. For step-up
mode connect to ground. For step-down mode connect to
inductor/diode. This pin must never be allowed to go more
than a Schottky diode drop below ground.
GND (Pin 5): Ground.
AO (Pin 6): Auxiliary Gain Block (GB) output. Open collector,
can sink 300µA.
SET (Pin 7): GB input. GB is an op amp with positive input
connected to SET pin and negative input connected to
220mV reference.
FB/SENSE (Pin 8): On the LT1110 (adjustable) this pin
goes to the comparator input. On the LT1110-5 and
LT1110-12, this pin goes to the internal application resistor
that sets output voltage.
PI
U
FU
U
C
U
S
O
TI
TEMPERATURE (°C)
BIAS CURRENT (nA)
–50
LT1110 • TPC13
160
140
120
100
60
–25
20
0
40
80
0 25 50 75 100
TEMPERATURE (°C)
V
REF
(mV)
–50 –25 0 25
LT1110 • TPC15
50 75 100
226
224
222
220
218
216
214
212
TEMPERATURE (°C)
BIAS CURRENT (nA)
–50
120
100
90
70
40
–25
10
0
30
60
0 25 50 75 100
20
50
80
110
LT1110 • TPC14
LT1110 • BD01
IN
V
GND
SET
AO
GAIN BLOCK/ERROR AMP
220mV
REFERENCE A1
A2
DRIVER
+
–
FB
SW1
SW2
LIM
I
OSCILLATOR
COMPARATOR
Q1
LT1110
6
-
LT1110
U
OPER
O
ATI
LT1110
The LT1110 is a gated oscillator switcher. This type
architecture has very low supply current because the
switch is cycled only when the feedback pin voltage drops
below the reference voltage. Circuit operation can best be
understood by referring to the LT1110 block diagram
above. Comparator A1 compares the FB pin voltage with
the 220mV reference signal. When FB drops below
220mV, A1 switches on the 70kHz oscillator. The driver
amplifier boosts the signal level to drive the output NPN
power switch Q1. An adaptive base drive circuit senses
switch current and provides just enough base drive to
ensure switch saturation without overdriving the switch,
resulting in higher efficiency. The switch cycling action
raises the output voltage and FB pin voltage. When the FB
voltage is sufficient to trip A1, the oscillator is gated off. A
small amount of hysteresis built into A1 ensures loop
stability without external frequency compensation. When
the comparator is low the oscillator and all high current
circuitry is turned off, lowering device quiescent current to
just 300µA for the reference, A1 and A2.
The oscillator is set internally for 10µs ON time and 5µs
OFF time, optimizing the device for step-up circuits where
V
OUT
≈ 3V
IN
, e.g., 1.5V to 5V. Other step-up ratios as well
as step-down (buck) converters are possible at slight
losses in maximum achievable power output.
A2 is a versatile gain block that can serve as a low battery
detector, a linear post regulator, or drive an under voltage
lockout circuit. The negative input of A2 is internally
connected to the 220mV reference. An external resistor
divider from V
IN
to GND provides the trip point for A2. The
AO output can sink 300µA (use a 47k resistor pull up to
+5V). This line can signal a microcontroller that the battery
voltage has dropped below the preset level. To prevent the
gain block from operating in its linear region, a 2MΩ
resistor can be connected from AO to SET. This provides
positive feedback.
A resistor connected between the I
LIM
pin and V
IN
adjusts
maximum switch current. When the switch current ex-
ceeds the set value, the switch is turned off. This feature
is especially useful when small inductance values are used
with high input voltages. If the internal current limit of 1.5A
is desired, I
LIM
should be tied directly to V
IN
. Propagation
delay through the current limit circuitry is about 700ns.
In step-up mode, SW2 is connected to ground and SW1
drives the inductor. In step-down mode, SW1 is con-
nected to V
IN
and SW2 drives the inductor. Output voltage
is set by the following equation in either step-up or step-
down modes where R1 is connected from FB to GND and
R2 is connected from V
OUT
to FB.
The LT1110-5 and LT1110-12 fixed output voltage ver-
sions have the gain setting resistors on-chip. Only three
external components are required to construct a 5V or 12V
output converter. 16µA flows through R1 and R2 in the
LT1110-5, and 39µA flows in the LT1110-12. This current
represents a load and the converter must cycle from time
to time to maintain the proper output voltage. Output
ripple, inherently present in gated oscillator designs, will
typically run around 90mV for the LT1110-5 and 200mV
for the LT1110-12 with the proper inductor/capacitor
selection. This output ripple can be reduced considerably
by using the gain block amp as a pre-amplifier in front of
the FB pin. See the Applications section for details.
W
IDAGRA
B
L
O
C
K
-5, -12
LT1110
A1
LT1110 • BD02
IN
V
GND
SET
AO
A2
220mV
REF OSCILLATOR
DRIVER
+
–
R1
SW1
SW2
LIM
I
R2
300kΩSENSE LT1110-5:
LT1110-12: R1 = 13.8kΩ
R2 = 5.6kΩ
GAIN BLOCK/ERROR AMP
COMPARATOR
Q1
U
OPER
O
ATI
-5, -12
VmV
R
R
OUT =
()
+
220 2
1101.()
LT1110
7
Inductor Selection — General
A DC-DC converter operates by storing energy as mag-
netic flux in an inductor core, and then switching this
energy into the load. Since it is flux, not charge, that is
stored, the output voltage can be higher, lower, or oppo-
site in polarity to the input voltage by choosing an appro-
priate switching topology. To operate as an efficient en-
ergy transfer element, the inductor must fulfill three re-
quirements. First, the inductance must be low enough for
the inductor to store adequate energy under the worst
case condition of minimum input voltage and switch ON
time. The inductance must also be high enough so maxi-
mum current ratings of the LT1110 and inductor are not
exceeded at the other worst case condition of maximum
input voltage and ON time. Additionally, the inductor core
must be able to store the required flux; i.e., it must not
saturate
. At power levels generally encountered with
LT1110 based designs, small surface mount ferrite core
units with saturation current ratings in the 300mA to 1A
range and DCR less than 0.4Ω (depending on application)
are adequate. Lastly, the inductor must have sufficiently
low DC resistance so excessive power is not lost as heat
in the windings. An additional consideration is Electro-
Magnetic Interference (EMI). Toroid and pot core type
inductors are recommended in applications where EMI
must be kept to a minimum; for example, where there are
sensitive analog circuitry or transducers nearby. Rod core
types are a less expensive choice where EMI is not a
problem. Minimum and maximum input voltage, output
voltage and output current must be established before an
inductor can be selected.
Inductor Selection — Step-Up Converter
In a step-up, or boost converter (Figure 4), power gener-
ated by the inductor makes up the difference between
input and output. Power required from the inductor is
determined by
PV VV I
L OUT D IN MIN OUT
=+
()()
–()01
where V
D
is the diode drop (0.5V for a 1N5818 Schottky).
U
S
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PPLICATI
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I FOR ATIO
Energy required by the inductor per cycle must be equal or
greater than
P
f
L
OSC
()02
in order for the converter to regulate the output.
When the switch is closed, current in the inductor builds
according to
It V
Re
LIN Rt
L
() '–()
–'
=
103
where R' is the sum of the switch equivalent resistance
(0.8Ω typical at 25°C) and the inductor DC resistance.
When the drop across the switch is small compared to V
IN
,
the simple lossless equation
It V
L
t
LIN
()
=()04
can be used. These equations assume that at t = 0,
inductor current is zero. This situation is called “discon-
tinuous mode operation” in switching regulator parlance.
Setting “t” to the switch ON time from the LT1110 speci-
fication table (typically 10µs) will yield I
PEAK
for a specific
“L” and V
IN
. Once I
PEAK
is known, energy in the inductor
at the end of the switch ON time can be calculated as
ELI
LPEAK
=1
205
2
()
E
L
must be greater than P
L
/f
OSC
for the converter to deliver
the required power. For best efficiency I
PEAK
should be
kept to 1A or less. Higher switch currents will cause
excessive drop across the switch resulting in reduced
efficiency. In general, switch current should be held to as
low a value as possible in order to keep switch, diode and
inductor losses at a minimum.
As an example, suppose 12V at 120mA is to be generated
from a 4.5V to 8V input. Recalling equation (01),
P V V V mA mW
L=+
()()
=12 0 5 4 5 120 960 06.–. .()
Energy required from the inductor is
P
f
mW
kHz J
L
OSC
==
960
70 13 7 07.. ()µ
LT1110
8
U
S
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PPLICATI
WU
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I FOR ATIO
Picking an inductor value of 47µH with 0.2Ω DCR results
in a peak switch current of
IVemA
PEAK
s
H
=−
=
−•
45
10 1 862 08
10 10
47
.
..()
.
W
Wm
m
Substituting I
PEAK
into Equation 05 results in
EHAJ
L
=
()( )
=
1
2
47 0 862 17 5 09
2
µµ...()
Since 17.5µJ > 13.7µJ, the 47µH inductor will work. This
trial-and-error approach can be used to select the opti-
mum inductor. Keep in mind the switch current maximum
rating of 1.5A. If the calculated peak current exceeds this,
an external power transistor can be used.
A resistor can be added in series with the I
LIM
pin to invoke
switch current limit. The resistor should be picked such
that the calculated I
PEAK
at minimum V
IN
is equal to the
Maximum Switch Current (from Typical Performance
Characteristic curves). Then, as V
IN
increases, switch
current is held constant, resulting in increasing efficiency.
Inductor Selection — Step-Down Converter
The step-down case (Figure 5) differs from the step-up in
that the inductor current flows through the load during
both the charge and discharge periods of the inductor.
Current through the switch should be limited to ~800mA
in this mode. Higher current can be obtained by using an
external switch (see Figure 6). The I
LIM
pin is the key to
successful operation over varying inputs.
After establishing output voltage, output current and input
voltage range, peak switch current can be calculated by the
formula
II
DC
VV
VV V
PEAK OUT OUT D
IN SW D
=++
210
–()
where DC = duty cycle (0.69)
V
SW
= switch drop in step-down mode
V
D
= diode drop (0.5V for a 1N5818)
I
OUT
= output current
V
OUT
= output voltage
V
IN
= minimum input voltage
V
SW
is actually a function of switch current which is in turn
a function of V
IN
, L, time and V
OUT
. To simplify, 1.5V can
be used for V
SW
as a very conservative value.
Once I
PEAK
is known, inductor value can be derived from
LVVV
It
IN MIN SW OUT
PEAK ON
=•
–– ()11
where t
ON
= switch ON time (10µs).
Next, the current limit resistor R
LIM
is selected to give
I
PEAK
from the R
LIM
Step-Down Mode curve. The addition
of this resistor keeps maximum switch current constant as
the input voltage is increased.
As an example, suppose 5V at 250mA is to be generated
from a 9V to 18V input. Recalling Equation (10),
ImA mA
PEAK
=
()
+
+
=
2 250
069
505
91505 498 12
.
.
–. . .( )
Next, inductor value is calculated using Equation (11)
LmA sH=•=
9155
498 10 50 13
–.– .()µµ
Use the next lowest standard value (47µH).
Then pick R
LIM
from the curve. For I
PEAK
= 500mA,
R
LIM
= 82Ω.
Inductor Selection — Positive-to-Negative Converter
Figure 7 shows hookup for positive-to-negative conver-
sion. All of the output power must come from the inductor.
In this case,
PV VI
L OUT D OUT
=+
()()
|| . ()14
In this mode the switch is arranged in common collector
or step-down mode. The switch drop can be modeled as
a 0.75V source in series with a 0.65Ω resistor. When the
LT1110
9
switch closes, current in the inductor builds according to
IV
Re
LLRt
L
+
()
=
'–()
–'
115
where R' = 0.65Ω + DCR
L
V
L
= V
IN
– 0.75V
As an example, suppose –5V at 75mA is to be generated
from a 4.5V to 5.5V input. Recalling Equation (14),
PVVmAmW
L
=− +
()()
=||..()5 0 5 75 413 16
Energy required from the inductor is
P
f
mW
kHz J
L
OSC ==
413
70 59 17.. ()µ
Picking an inductor value of 56µH with 0.2Ω DCR results
in a peak switch current of
IVV
emA
PEAK
s
H
=
()
+
()
=
•
45 075
065 02 1 621 18
085 10
56
.–.
..
–.()
–.
ΩΩ
Ωµ
µ
Substituting I
PEAK
into Equation (04) results in
EHAJ
L
=
()( )
=
1
2
56 0 621 10 8 19
2
µµ...()
Since 10.8µJ > 5.9µJ, the 56µH inductor will work.
With this relatively small input range, R
LIM
is not usually
necessary and the I
LIM
pin can be tied directly to V
IN
. As in
the step-down case, peak switch current should be limited
to ~800mA.
Capacitor Selection
Selecting the right output capacitor is almost as important
as selecting the right inductor. A poor choice for a filter
capacitor can result in poor efficiency and/or high output
ripple. Ordinary aluminum electrolytics, while inexpensive
and readily available, may have unacceptably poor Equiva-
lent Series Resistance (ESR) and ESL (inductance). There
are low ESR aluminum capacitors on the market specifi-
cally designed for switch mode DC-DC converters which
work much better than general-purpose units. Tantalum
capacitors provide still better performance at more ex-
pense. We recommend OS-CON capacitors from Sanyo
Corporation (San Diego, CA). These units are physically
quite small and have extremely low ESR. To illustrate,
Figures 1, 2 and 3 show the output voltage of an LT1110
based converter with three 100µF capacitors. The peak
switch current is 500mA in all cases. Figure 1 shows a
Sprague 501D, 25V aluminum capacitor. V
OUT
jumps by
over 120mV when the switch turns off, followed by a drop
in voltage as the inductor dumps into the capacitor. This
works out to be an ESR of over 240mΩ. Figure 2 shows the
same circuit, but with a Sprague 150D, 20V tantalum
capacitor replacing the aluminum unit. Output jump is
now about 35mV, corresponding to an ESR of 70mΩ.
Figure 3 shows the circuit with a 16V OS-CON unit. ESR is
now only 20mΩ.
5 s/DIV
50mV/DIV
LT1110 • TA19
µ
Figure 1. Aluminum
5 s/DIV
50mV/DIV
LT1110 • TA20
µ
Figure 2. Tantalum
5 s/DIV
50mV/DIV
LT1110 • TA21
µ
Figure 3. OS-CON
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LT1110
10
Diode Selection
Speed, forward drop, and leakage current are the three
main considerations in selecting a catch diode for LT1110
converters. General purpose rectifiers such as the 1N4001
are
unsuitable
for use in
any
switching regulator applica-
tion. Although they are rated at 1A, the switching time of
a 1N4001 is in the 10µs-50µs range. At best, efficiency will
be severely compromised when these diodes are used; at
worst, the circuit may not work at all. Most LT1110 circuits
will be well served by a 1N5818 Schottky diode, or its
surface mount equivalent, the MBRS130T3. The combina-
tion of 500mV forward drop at 1A current, fast turn ON and
turn OFF time, and 4µA to 10µA leakage current fit nicely
with LT1110 requirements. At peak switch currents of
100mA or less, a 1N4148 signal diode may be used. This
diode has leakage current in the 1nA-5nA range at 25°C
and lower cost than a 1N5818. (You can also use them to
get your circuit up and running, but beware of destroying
the diode at 1A switch currents.)
Step-Up (Boost Mode) Operation
A step-up DC-DC converter delivers an output voltage
higher than the input voltage. Step-up converters are
not
short circuit protected since there is a DC path from input
to output.
The usual step-up configuration for the LT1110 is shown
in Figure 4. The LT1110 first pulls SW1 low causing V
IN
–
V
CESAT
to appear across L1. A current then builds up in L1.
At the end of the switch ON time the current in L1 is
1
:
IV
Lt
PEAK IN ON
=()20
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Immediately after switch turn off, the SW1 voltage pin
starts to rise because current cannot instantaneously stop
flowing in L1. When the voltage reaches V
OUT
+ V
D
, the
inductor current flows through D1 into C1, increasing
V
OUT
. This action is repeated as needed by the LT1110 to
keep V
FB
at the internal reference voltage of 220mV. R1
and R2 set the output voltage according to the formula
VR
RmV
OUT
=+
()
1
2
1
220 21.()
Step-Down (Buck Mode) Operation
A step-down DC-DC converter converts a higher voltage
to a lower voltage. The usual hookup for an LT1110 based
step-down converter is shown in Figure 5.
LT1110 • TA15
GND
SW2
SW1
LIM
I
IN
V
R3
220
FB
V
OUT
+
C2
+
C1
D1
1N5818
V
IN
R2
R1
L1
Ω
LT1110
Figure 5. Step-Down Mode Hookup
When the switch turns on, SW2 pulls up to V
IN
– V
SW
. This
puts a voltage across L1 equal to V
IN
– V
SW
– V
OUT
,
causing a current to build up in L1. At the end of the switch
ON time, the current in L1 is equal to
IVVV
Lt
PEAK IN SW OUT ON
=−− .()22
When the switch turns off, the SW2 pin falls rapidly and
actually goes below ground. D1 turns on when SW2
reaches 0.4V below ground.
D1
MUST BE A SCHOTTKY
DIODE
. The voltage at SW2 must never be allowed to go
below –0.5V. A silicon diode such as the 1N4933 will allow
SW2 to go to –0.8V, causing potentially destructive power
Figure 4. Step-Up Mode Hookup.
L1
LT1110 • TA14
GND SW2
SW1
LIM
I
IN
V
D1
R3*
LT1110
+
V
OUT
R2
R1
C1
* = OPTIONAL
V
IN
FB
Note 1: This simple expression neglects the effects of switch and coil
resistance. This is taken into account in the “Inductor Selection” section.
LT1110
11
Converter” section with the following conservative ex-
pression for V
SW
:
VVV V
SW R SAT
=+ ≈
1
09 24.. ()
R2 provides a current path to turn off Q1. R3 provides base
drive to Q1. R4 and R5 set output voltage.
Inverting Configurations
The LT1110 can be configured as a positive-to-negative
converter (Figure 7), or a negative-to-positive converter
(Figure 8). In Figure 7, the arrangement is very similar to
a step-down, except that the high side of the feedback is
referred to ground. This level shifts the output negative. As
in the step-down mode, D1 must be a Schottky diode,
andV
OUT
should be less than 6.2V. More negative out-
put voltages can be accommodated as in the prior section.
LT1110 • TA03
–V
OUT
+
C2
+
C1
D1
1N5818
+V
IN
R1
R2
L1
GND
SW2
SW1
LIM
I
IN
V
R3
FB
LT1110
Figure 7. Positive-to-Negative Converter
In Figure 8, the input is negative while the output is
positive. In this configuration, the magnitude of the input
voltage can be higher or lower than the output voltage. A
level shift, provided by the PNP transistor, supplies proper
polarity feedback information to the regulator.
dissipation inside the LT1110. Output voltage is deter-
mined by
VR
RmV
OUT
=+
()
1
2
1
220 23.()
R3 programs switch current limit. This is especially im-
portant in applications where the input varies over a wide
range. Without R3, the switch stays on for a fixed time
each cycle. Under certain conditions the current in L1 can
build up to excessive levels, exceeding the switch rating
and/or saturating the inductor. The 220Ω resistor pro-
grams the switch to turn off when the current reaches
approximately 800mA. When using the LT1110 in step-
down mode, output voltage should be limited to 6.2V or
less. Higher output voltages can be accommodated by
inserting a 1N5818 diode in series with the SW2 pin
(anode connected to SW2).
Higher Current Step-Down Operation
Output current can be increased by using a discrete PNP
pass transistor as shown in Figure 6. R1 serves as a
current limit sense. When the voltage drop across R1
equals a V
BE
, the switch turns off. For temperature com-
pensation a Schottky diode can be inserted in series with
the I
LIM
pin. This also lowers the maximum drop across R1
to V
BE
– V
D
, increasing efficiency. As shown, switch
current is limited to 2A. Inductor value can be calculated
based on formulas in the “Inductor Selection Step-Down
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Figure 6. Q1 Permits Higher-Current Switching.
LT1110 Functions as Controller.
LT1110 • TA16
D1
1N5821
+
+
V
OUT
V
IN
25V
MAX
L1
R1
0.3Ω
R2
220
Q1
MJE210 OR
ZETEX ZTX789A
R3
330
R4
R5
C1
V
OUT
= 220mV
(
1 +
)
R4
R5
LT1110
GND SW2
SW1
V
IN
I
L
FB
C2
Figure 8. Negative-to-Positive Converter
L1
LT1110 • TA04
GND SW2
FB
SW1
LIM
I
IN
V
D1
AO
+V
OUT
R2 V
OUT
= 220mV + 0.6V
R1
R2
( )
R1
2N3906
–V
IN
+
C1
LT1110
+
C2
LT1110
12
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Using the I
LIM
Pin
The LT1110 switch can be programmed to turn off at a set
switch current, a feature not found on competing devices.
This enables the input to vary over a wide range without
exceeding the maximum switch rating or saturating the
inductor. Consider the case where analysis shows the
LT1110 must operate at an 800mA peak switch current
with a 2.0V input. If V
IN
rises to 4V, peak current will rise
to 1.6A, exceeding the maximum switch current rating.
With the proper resistor selected (see the “Maximum
Switch
Current vs R
LIM
” characteristic), the switch current
will be limited to 800mA, even if the input voltage
increases.
Another situation where the I
LIM
feature is useful occurs
when the device goes into continuous mode operation.
This occurs in step-up mode when
VV
VV DC
OUT DIODE
IN SW
+
−<−1
125.()
When the input and output voltages satisfy this relation-
ship, inductor current does not go to zero during the
switch OFF time. When the switch turns on again, the
current ramp starts from the non-zero current level in the
inductor just prior to switch turn on. As shown in Figure 9,
the inductor current increases to a high level before the
comparator turns off the oscillator. This high current can
cause excessive output ripple and requires oversizing the
output capacitor and inductor. With the I
LIM
feature,
however, the switch current turns off at a programmed
level as shown in Figure 10, keeping output ripple to a
minimum.
Figure 11 details current limit circuitry. Sense transistor
Q1, whose base and emitter are paralleled with power
switch Q2, is ratioed such that approximately 0.5% of Q2’s
collector current flows in Q1’s collector. This current is
passed through internal 80Ω resistor R1 and out through
the I
LIM
pin. The value of the external resistor connected
between I
LIM
and V
IN
set the current limit. When sufficient
switch current flows to develop a V
BE
across R1 + R
LIM
, Q3
turns on and injects current into the oscillator, turning off
the switch. Delay through this circuitry is approximately
800ns. The current trip point becomes less accurate for
switch ON times less than 3µs. Resistor values program-
ming switch ON time for 800ns or less will cause spurious
response in the switch circuitry although the device will
still maintain output regulation.
LT1110 • TA05
I
OFF
L
ON
SWITCH
Figure 9. No Current Limit Causes Large Inductor
Current Build-Up
LT1110 • TA06
I
ON
L
OFF
SWITCH
PROGRAMMED CURRENT LIMIT
Figure 10. Current Limit Keeps Inductor Current Under Control
LT1110 • TA17
SW2
SW1
Q2
DRIVER
OSCILLATOR
VIN
ILIM
R1
80Ω
(INTERNAL)
RLIM
(EXTERNAL)
Q1
Q3
Figure 11. LT1110 Current Limit Circuitry
Using the Gain Block
The gain block (GB) on the LT1110 can be used as an error
amplifier, low battery detector or linear post regulator. The
gain block itself is a very simple PNP input op amp with an
open collector NPN output. The negative input of the gain
block is tied internally to the 220mV reference. The posi-
tive input comes out on the SET pin.
LT1110
13
Arrangement of the gain block as a low battery detector is
straightforward. Figure 12 shows hookup. R1 and R2 need
only be low enough in value so that the bias current of the
SET input does not cause large errors. 33kΩ for R2 is
adequate. R3 can be added to introduce a small amount of
hysteresis. This will cause the gain block to “snap” when
the trip point is reached. Values in the 1M-10M range are
optimal. The addition of R3 will change the trip point,
however.
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Output ripple of the LT1110, normally 90mV at 5V
OUT
can
be reduced significantly by placing the gain block in front
of the FB input as shown in Figure 13. This effectively
reduces the comparator hysteresis by the gain of the gain
block. Output ripple can be reduced to just a few millivolts
using this technique. Ripple reduction works with step-
down or inverting modes as well. For this technique to be
effective, output capacitor C1 must be large, so that each
switching cycle increases V
OUT
by only a few millivolts.
1000µF is a good starting value.
Figure 13. Output Ripple Reduction Using Gain Block
Figure 12. Setting Low Battery Detector Trip Point
LT1110 • TA07
V
BAT
R1
R2
220mV
REF
SET
GND
V
IN
LT1110
47k
+5V
TO 
PROCESSOR
R1 = V
LB
– 220mV
( )
4.33µA
V
LB
= BATTERY TRIP POINT
+
–
AO
R3
R2 = 33kΩ
R3 = 2MΩ
Table 2. Capacitor Manufacturers
MANUFACTURER PART NUMBERS
Sanyo Video Components OS-CON Series
2001 Sanyo Avenue
San Diego, CA 92173
619-661-6835
Nichicon America Corporation PL Series
927 East State Parkway
Schaumberg, IL 60173
708-843-7500
Sprague Electric Company 150D Solid Tantalums
Lower Main Street 550D Tantalex
Sanford, ME 04073
207-324-4140
Matsuo 267 Series
714-969-2491 Surface Mount
Table 3. Transistor Manufacturers
MANUFACTURER PART NUMBERS
Zetex ZTX Series
Commack, NY FZT Series
516-543-7100 Surface Mount
Table 1. Inductor Manufacturers
MANUFACTURER PART NUMBERS
Coiltronics International CTX100-4 Series
984 S.W. 13th Court Surface Mount
Pompano Beach, FL 33069
305-781-8900
Sumida Electric Co. USA CD54
708-956-0666 CDR74
CDR105
Surface Mount
LT1110
14
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SA
O
PPLICATITYPICAL
All Surface Mount
Flash Memory VPP Generator
1.5V Powered Laser Diode Driver
L1
2.2 H
LT1110 • TA13
µ
C1
100 F
OS-CON
µ
1N5818
4.7k
10
1k*
R1
+
TOSHIBA
TOLD-9211
Ω
220Ω
2N3906
12
3
7
45
6
8
22nF
1.5V
ADJUST R1 FOR CHANGE IN LASER OUTPUT POWER
TOKO 262LYF-0076M
*
1N4148
2Ω
GND SW2
SET
SW1
LIM
I
IN
V
FB
AO
LT1110
LASER DIODE CASE COMMON TO +BATTERY TERMINAL
170mA CURRENT DRAIN FROM 1.5V CELL (50mA DIODE)
NO OVERSHOOT
•
•
•
✝
✝
MJE210
0.22 F
CERAMIC
µ
1.5V Powered Laser Diode Driver
LT1110 • TA18
GND SW2
SENSE
SW1
LIM
I
IN
V
LT1110CS8-12
*L1= SUMIDA CD105-470M
47µF
20V
L1*
47µH
V
PP
12V
120MA
22µF
10k
+5V
±10%
+
+
1k
MBRS12OT3
MMBT4403
= PROGRAM
= SHUTDOWN
1
0MMBF170
LT1110
15
LT1110 • TA10
GND SW2
SENSE
SW1
LIM
I
IN
V
LT1110-5
9V
*L1 = COILCRAFT 1812LS-473
10µF
MBRL120
5V
40mA
220
L1*
47µH
+
LT1110 • TA11
GND SW2
FB
SW1
LIM
I
IN
V
LT1110
1.5V
AA OR
AAA
CELL
= MBRL120
= COILCRAFT 1812LS-823
4.7µF
L1*
82µH
+5V
3mA
4.7µF
+10V
3mA
490k
*L1
4.7µF
11k
+
+
+
LT1110 • TA09
GND SW2
SENSE
SW1
LIM
I
IN
V
LT1110-5
3V
2x
AA CELL
*L1 = COILCRAFT 1812LS-473
10µF
MBRL120
L1*
47µH
5V
40mA
220
+
LT1110 • TA12
GND SW2
SENSE
SW1
LIM
I
IN
V
LT1110
1.5V
AA OR
AAA
CELL
= MBRL120
= COILCRAFT 1812LS-823
4.7µF
L1*
82µH
+5V
4mA
4.7µF
–5V
4mA
4.7µF
+
+
*L1
+
U
SA
O
PPLICATITYPICAL
All Surface Mount
3V to 5V Step-Up Converter
All Surface Mount
9V to 5V Step-Down Converter
All Surface Mount
1.5V to +10V, +5V Dual Output Step-Up Converter
All Surface Mount
1.5V to ±5V Dual Output Step-Up Converter
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LT1110
16
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PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead Plastic DIP
S8 Package
8-Lead Plastic SOIC
LINEAR TECHNOLOGY CORPORATION 1994
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900
●
FAX
: (408) 434-0507
●
TELEX
: 499-3977
LT/GP 0594 2K REV B • PRINTED IN USA
0.016 – 0.050
0.406 – 1.270
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
1234
0.150 – 0.157*
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm).
0.009 – 0.015
(0.229 – 0.381)
0.300 – 0.320
(7.620 – 8.128)
0.325 +0.025
–0.015
+0.635
–0.381
8.255
()
0.045 ± 0.015
(1.143 ± 0.381)
0.100 ± 0.010
(2.540 ± 0.254)
0.065
(1.651)
TYP
0.045 – 0.065
(1.143 – 1.651)
0.130 ± 0.005
(3.302 ± 0.127)
0.020
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.125
(3.175)
MIN
12 34
8765
0.250 ± 0.010
(6.350 ± 0.254)
0.400
(10.160)
MAX