Application Note 86 - Linear Technology

Application Note 86

AN86-1

A Standards Lab Grade 20-Bit DAC with 0.1ppm/°C Drift

The Dedicated Art of Digitizing One Part Per Million

Jim Williams

J. Brubaker

P. Copley

J. Guerrero

F. Oprescu

January 2001

INTRODUCTION

Significant progress in high precision, instrumentation

grade D-to-A conversion has recently occurred. Ten years

ago 12-bit D-to-A converters (DACs) were considered

premium devices. Today, 16-bit DACs are available and

increasingly common in system design. These are true

precision devices with less than 1LSB linearity error and

1ppm/°C drift.

Nonetheless, there are DAC applications

that require even higher performance. Automatic test

equipment, instruments, calibration apparatus, laser trim-

mers, medical electronics and other applications often

require DAC accuracy beyond 16 bits. 18-bit DACs have

been produced in circuit assembly form, although they are

expensive and require frequent calibration. 20 and even

23+ (0.1ppm!) bit DACs are represented by manually

switched Kelvin-Varley dividers. These devices, although

amazingly accurate, are large, slow and extremely costly.

Their use is normally restricted to standards labs.

useful development would be a practical, 20-bit (1ppm)

DAC that is easily constructed and does not require

frequent calibration.

20-Bit DAC Architecture

Figure 1 diagrams the architecture of a 20-bit (1ppm) DAC.

This scheme is based on the availability of a true 1ppm

analog-to-digital converter with scale and zero drifts be-

low 0.02ppm/°C. This device, the LTC

2400, is used as a

feedback element in a digitally corrected loop to realize a

20-bit DAC.

In practice, the “slave” 20-bit DAC’s output is monitored

by the “master” LTC2400 A-to-D, which feeds digital

information to a code comparator. The code comparator

, LTC and LT are registered trademarks of Linear Technology Corporation.

Note 1: See Appendix A, “A History of High Accuracy Digital-to-Analog

Conversion,” for a review of high accuracy digital-to-analog conver-

sion.

Note 2: Consult Appendix C, “Verifying Data Converter Linearity to

1ppm,” for discussion on Kelvin-Varley dividers. Also, see Appendix A,

“A History of High Accuracy Digital-to-Analog Conversion.”

Note 3: The LTC2400 analog-to-digital converter is profiled in

Appendix␣ B, “The LTC2400—A Monolithic 24-Bit Analog-to-Digital

Converter.”

Note 4: D-to-A converters have been placed in loops to make A-to-D

converters for a long time. Here, an A-to-D converter feeds back a loop

to form a D-to-A converter. There seems a certain justified symmetry

to this development. Turnabout is indeed fair play.

differences the user input word with the LTC2400 output,

presenting a corrected code to the slave DAC. In this

fashion, the slave DAC’s drifts and nonlinearity are con-

tinuously corrected by the loop to an accuracy determined

by the A-to-D converter and V

REF4

. The sole DAC require-

ment is that it be monotonic. No other components in the

loop need to be stable.

REF

DIGITAL

CODE

COMPARATOR

20-BIT

USER

INPUT

CODE

AN86 F01

20-BIT

(1PPM)

ANALOG

OUTPUT

CORRECTED

CODE

LTC2400

“MASTER”

A-TO-D

20-BIT

“SLAVE”

DAC

FEEDBACK

CODE V

OUT

Figure 1. Conceptual Loop-Based 20-Bit DAC. Digital

Comparison Allows A-to-D to Correct DAC Errors. LTC2400

A-to-D’s Low Uncertainty Characteristics Permit 1ppm

Output Accuracy

Application Note 86

AN86-2

This loop has a number of desireable attributes. As men-

tioned, accuracy limitations are set by the A-to-D con-

verter and its reference. No other components need be

stable. Additionally, loop behavior averages low order bit

indexing and jitter, obviating the loop’s inherent small-

signal instability. Finally, classical remote sensing may be

used or digitally based sensing is possible by placing the

A-to-D converter at the load. The A-to-D’s SO-8 package

and lack of external components makes this digitally

incarnated Kelvin sensing scheme practical.

Circuitry Details

Figure 2 is a detailed schematic of the 1ppm DAC. The

slave DAC is comprised of two DACs. The upper 16 bits of

the code comparator’s output are fed to a 16-bit DAC

(“MSB DAC”), while the lower bits are converted by a

separate DAC (“LSB DAC”). Although a total of 32 bits are

presented to the two DACs, there are 8 bits of overlap,

assuring loop capture under all conditions. The composite

Note 5: One wonders what Lord Kelvin’s response would be to the

digizatation of his progeny. Such uncertainties are the residue of progress.

Note 6: The code comparator is detailed in Appendix D, “A Processor

Based Code Comparator.”

DACs’ resultant 24-bit resolution provides 4 bits of index-

ing range below the 20th bit, ensuring a stable LSB of

1ppm of scale. A1 and A2 transform the DAC’s output

currents into voltages, which are summed at A3. A3’s

scaling is arranged so that the correction loop can always

capture and correct any combination of zero- and full-

scale errors. A3’s output, the circuit output, feeds the

LTC2400 A-to-D. The LT

1010 provides buffering to drive

loads and cables. The A-to-D’s digital output is differenced

against the input word by the code comparator, which

produces a corrected code. This corrected code is applied

to the MSB and LSB DACs, closing a feedback loop.

The

loop’s integrity is determined by A-to-D converter and

voltage reference errors.

The resistor and diodes at the

5V powered A-to-D protect it from inadvertent A3 outputs

(power up, transient, lost supply, etc.). A4 is a reference

inverter and A5 provides a clean ground potential to both

DACs.

–15V

REF

(SEE APPENDIX I

FOR OPTIONS)

20-BIT

USER

INPUT

CODE

AN86 F02

CORRECTED

CODE OUT—

24-BIT WORD

COMMAND

INPUTS FROM

CODE COMPARATOR

* = 1% METAL FILM

LTC1599

MSB DAC

100pF

OUT

–

24-BIT

FEEDBACK

CODE 1k

1N5817

REF

DATA

INPUTS

5V 5V

CODE

COMPARATOR

(SEE APPENDIX D

FOR DETAILS)

LTC2400

A-TO-D

SERIAL

DIGITAL

OUT IN

REF

–

LT1001

–

LT1001

–

LT1001

15V

–

LT1001 LT1010

2k*

4.12M*

OUTPUT

AMPLIFIER

OUTPUT

RREF R

REF

CLVL CLR

2k*

0.1µF

LTC1599

LSB DAC

100pF

OUT

DATA

INPUTS

CLVL

–

LT1001

REFML

BYTE

–

CLR

Figure 2. Detail of 1ppm DAC. Composite DAC Is Comprised of Two DAC Values Summed

at Output Amplifier. LTC2400 A-to-D and Code Comparator Furnish Stabilizing Feedback

Note 7: Voltage reference options are discussed in Appendix I,

“Voltage References.” For tutorial on the LTC2400, refer to

Appendix␣ B.

Application Note 86

AN86-3

Note 8: Establishing and maintaining confidence in a 1ppm linearity

measurement is uncomfortably close to the state of the art. The

technique used is shown in Appendix C, “Verifying Data Converter

Linearity to 1ppm.”

Note 9: Noise measurement considerations appear in Appendix H,

“Microvolt Level Noise Measurement.”

Note 10: Measuring DAC settling time to 1ppm is by no means

straightforward, even at the relatively slow speed involved here. See

Appendix G, “Measuring DAC Settling Time.”

Linearity Considerations

A-to-D linearity determines overall DAC linearity. The

A-to-D has about ±2ppm nonlinearity. In applications

where this error is permissible, it may be ignored. If 1ppm

linearity is required, it is obtainable by correcting the

residual linearity error with software techniques. Details

on LTC2400 linearity and this feature are presented in

Appendices D and E.

DC Performance Characteristics

Figure 3 is a plot of linearity vs output code. The data

shows linearity is within 1ppm over all codes.

Output

noise, measured in a 0.1Hz to 10Hz bandpass, is seen in

Figure 4 to be about 0.2LSB.

This measurement is

somewhat corrupted by equipment limitations, which set

a noise floor of about 0.2µV.

Dynamic Performance

The A-to-D’s conversion rate combines with the loop’s

sampled data characteristic and slow amplifiers to dictate

relatively slow DAC response. Figure 5’s slew response

requires about 150 microseconds.

Figure 6 shows full-scale DAC settling time to within 1ppm

(±5µV) requires about 1400 milliseconds. A smaller step

(Figure 7) of 500µV needs only 100 milliseconds to settle

within 1ppm.

Conclusion

Summarized 1ppm DAC specifications appear in Figure 9.

These specifications should be considered guidelines, as

the options and variations noted will affect performance.

Consult the appropriate appendices for design specifics

and trade-offs.

Figure 4. Output Noise Indicates Less Than 1µV,

About 0.2LSB. Measurement Noise Floor, Due

to Equipment Limitations, Is 0.2µV

Figure 3. Linearity Plot Shows No

Error Outside 1ppm for All Codes

DIGITAL INPUT CODE

0262,144 524,288 786,432 1,048,576

LINEARITY ERROR (ppm)

AN86 F03

1.0

0.5

–0.5

–1.0

INDICATED MEASUREMENT RESULTS

SHADED REGION DELINEATES

MEASUREMENT UNCERTAINTY DUE TO

DAC OUPUT NOISE AND

INSTRUMENTATION LIMITATIONS

500nV/DIV

2s/DIV AN86 F04.tif

Application Note 86

AN86-4

Figure 5. DAC Output Full-Scale Slew Characteristics Figure 6. High Resolution Settling Detail After a Full-Scale Step.

Settling Time Is 1400 Milliseconds to Within 1ppm (±5µV)

Figure 8. Summarized Specifications for the 20-Bit DAC

Figure 7. Small Step Settling Time Measures 100

Milliseconds to Within 1ppm (±5µV) for a 500µV

Transition

Note: This Application Note was derived from a manuscript originally

prepared for publication in EDN magazine.

PARAMETER SPECIFICATION

Resolution 1ppm

Full-Scale Error 4ppm of V

REF

(Trimmable to 1ppm by V

REF

Adjustment)

Full-Scale Error Drift 0.04ppm/°C Exclusive of Reference

(0.1ppm/°C with LTZ1000A Reference

)

Offset Error 0.5ppm

Offset Error Drift 0.01ppm/°C

Nonlinearity ±2ppm, Trimmable to Less Than 1ppm

Output Noise 0.2ppm

(≈0.9µV, 0.1Hz to 10Hz BW)

Slew Rate 0.033V/µs

Settling Time—Full-Scale Step 1400 Milliseconds

Settling Time—500µV Step 100 Milliseconds

Output Voltage Range 0V to 5V. For Other Ranges See Note 3

Note 1: See Appendix I

Note 2: See Appendix E

Note 3: See Appendices E and F

1V/DIV

50µs/DIV AN86 F05.tif

5µV/DIV

500ms/DIV AN86 F06.tif

5µV/DIV

50ms/DIV AN86 F07.tif

Application Note 86

AN86-5

REFERENCES

1. Linear Technology Corporation, “LTC2400 Data Sheet,”

Linear Technology Corporation, January 1999.

2. Linear Technology Corporation, “LTC2410 Data Sheet,”

Linear Technology Corporation, April 2000.

3. Keithley Instruments, “Low Level Measurements,”

Keithley Instruments, 1984.

4. Williams, J., “Testing Linearity of the LTC2400 24-Bit

A/D Converter,” Linear Technology Corporation, De-

sign Solution 11, November 1999.

5. Seebeck, T. Dr., “Magnetische Polarisation der Metalle

und Erze durch Temperatur-Differenz,” Abhaandlungen

der Preussischen Akademic der Wissenschaften

(1822–1823), pp. 265–373.

6. Williams, J., “Component and Measurement Advances

Ensure 16-Bit DAC Settling Time,” Linear Technology

Corporation, Application Note 74, July 1998.

7. Lee, M., “Understanding and Applying Voltage Refer-

ences,” Linear Technology Corporation, Application

Note 82, November 1999.

8. Williams, J., “Applications Considerations and Cir-

cuits for a New Chopper-Stabilized Op Amp,” Linear

Technology Corporation, Application Note 9, August

1987.

9. Huffman, B., “Voltage Reference Circuit Collection,”

Linear Technology Corporation, Application Note 42,

June 1991.

10. Spreadbury, P. J., “The Ultra-Zener—A Portable Re-

placement for the Weston Cell?” IEEE Transactions on

Instrumentation and Measurement, Vol. 40, No. 2,

April 1991, pp. 343–346.

11. Williams, J., “Thermocouple Measurement,” Linear

Technology Corporation, Application Note 28, Febru-

ary 1988.

12. Hueckel, J. H., “Input Connection Practices for Differ-

ential Amplifiers,” Neff Inst. Corporation, Duarte, Cali-

fornia.

13. Gould Inc., “Elimination of Noise in Low Level Cir-

cuits,” Gould Inc., Instrument Systems Division, Cleve-

land, Ohio.

14. Williams, J., “Prevent Low Level Amplifier Problems,”

Electronic Design, February 15, 1975, p. 62.

15. Pascoe, G., “The Choice of Solders for High Gain

Devices,” New Electronics (Great Britain), February 6,

1977.

16. Pascoe, G., “The Thermo-E.M.F. of Tin-Lead Alloys,”

Journal Phys. E, December 1976.

17. Brokaw, A. P., “Designing Sensitive Circuits? Don’t

Take Grounds for Granted,” EDN, October 5, 1975,

p.␣ 44.

18. Morrison, R., “Noise and Other Interfering Signals,”

John Wiley and Sons, 1992.

19. Morrison, R., “Grounding and Shielding Techniques

in Instrumentation,” Wiley-Interscience, 1986.

20. Vishay Inc., “Vishay Foil Resistors,” Vishay Inc., 1999.

Application Note 86

AN86-6

APPENDIX A

A HISTORY OF HIGH ACCURACY

DIGITAL-TO-ANALOG CONVERSION

People have been converting digital-to-analog quantities

for a long time. Probably among the earliest uses was the

summing of calibrated weights (Figure A1, left center) in

weighing applications. Early electrical digital-to-analog

conversion inevitably involved switches and resistors of

different values, usually arranged in decades. The applica-

tion was often the calibrated balancing of a bridge or

reading, via null detection, some unknown voltage. The

most accurate resistor-based DAC of this type is Lord

Kelvin’s Kelvin-Varley divider (Figure, large box). Based

on switched resistor ratios, it can achieve ratio accuracies

of 0.1ppm (23+ bits) and is still widely employed in

standards laboratories.

High speed digital-to-analog con-

version resorts to electronically switching the resistor

network. Early electronic DACs were built at the board level

using discrete precision resistors and germanium transis-

tors (Figure, center foreground, is a 12-bit DAC from a

Minuteman missile D-17B inertial navigation system, circa

1962). The first electronically switched DACs available as

standard product were probably those produced by

Pastoriza Electronics in the mid 1960s. Other manufactur-

ers followed and discrete- and monolithically-based modu-

lar DACs (Figure, right and left) became popular by the

1970s. The units were often potted (Figure, left) for

ruggedness, performance or to (hopefully) preserve pro-

prietary knowledge. Hybrid technology produced smaller

package size (Figure, left foreground). The development of

Si-Chrome resistors permitted precision monolithic DACs

such as the LTC1595 (Figure, immediate foreground). In

keeping with all things monolithic, the cost-performance

trade off of modern high resolution IC DACs is a bargain.

Think of it! A 16-bit DAC in an 8-pin IC package. What Lord

Kelvin would have given for a credit card and LTC’s phone

number.

Figure A1. Historically Significant Digital-to-Analog Converters Include: Weight Set (Center Left), 23+ Bit Kelvin-Varley Divider

(Large Box), Hybrid, Board and Modular Types, and the LTC1595 IC (Foreground). Where Will It All End?

AN86 FA01.tif

Note 1: See Appendix C, “Verifying Data Converter Linearity to 1ppm,”

for details on Kelvin-Varley Dividers.

Application Note 86

AN86-7

APPENDIX B

THE LTC2400—A MONOLITHIC 24-BIT

ANALOG-TO-DIGITAL CONVERTER

The LTC2400 is a micropower 24-bit A-to-D converter

with an integrated oscillator, 4ppm nonlinearity and 0.3ppm

RMS noise. It uses delta-sigma technology to provide

extremely high stability. The device can be configured for

better than 110dB rejection at 50Hz or 60Hz ±2%, or it can

be driven by an external oscillator for a user defined

rejection frequency in the range 1Hz to 120Hz.

This ultraprecision A-to-D converter in an SO-8 pin pack-

age forms the heart of the 20-bit DAC described in the text.

It is significant that the device is used here as a circuit

component

rather than in the traditional standalone role

accorded precision A-to-D converters. This freedom, in

keeping with the IC’s economy and ease of use, is a

noteworthy opportunity. Alert designers will recognize

this development and capitalize on it. Key specifications

for the A-to-D are given in Figure B1.

PARAMETER CONDITIONS

Resolution (No Missing Codes) 0.1V ≤ V

REF

≤ V

24 Bits

Integral Nonlinearity V

REF

= 2.5V 2ppm of V

REF

= 5V 4ppm of V

REF

Offset Error 2.5V ≤ V

REF

≤ V

0.5ppm of V

REF

Offset Error Drift 2.5V ≤ V

REF

≤ V

0.01ppm of V

REF

/°C

Full-Scale Error 2.5V ≤ V

REF

≤ V

4ppm of V

REF

Full-Scale Error Drift 2.5V ≤ V

REF

≤ V

0.02ppm of V

REF

/°C

Total Unadjusted Error V

REF

= 2.5V 5ppm of V

REF

= 5V 1ppm of V

REF

Output Noise 1.5µV

RMS

Normal Mode Rejection 110dB (Min)

60Hz ±2%

Normal Mode Rejection 110dB (Min)

50Hz ±2

Input Voltage Range 0.125V • V

REF

to 1.125V • V

REF

Reference Voltage Range 0.1V ≤ V

REF

≤ V

Supply Voltage 2.7V ≤ V

≤ 5.5V

Supply Current

Conversion Mode CS = 0V 200µA

Sleep Mode CS = V

20µA

Figure B1. Key Specifications for LTC2400 A-to-D Converter.

High Linearity and Extreme Stability Allow Realization of 1ppm DAC

Application Note 86

AN86-8

APPENDIX C

VERIFYING DATA CONVERTER LINEARITY TO 1PPM

Help from the Nineteenth Century

INTRODUCTION

Verifying 1ppm linearity of the DAC and the analog-to-

digital converter used to construct it requires special

considerations. Testing necessitates some form of volt-

age source that produces equal amplitude output steps for

incremental digital inputs. Additionally, for measurement

confidence, it is desirable that the source be substantially

more linear than the 1ppm requirement. This is, of course,

a stringent demand and painfully close to the state of the

art.

The most linear “D to A” converter is also one of the oldest.

Lord Kelvin’s Kelvin-Varley divider (KVD), in its most

developed form, is linear to 0.1ppm. This manually switched

device features ten million individual dial settings ar-

ranged in seven decades. It may be thought of as a

3-terminal potentiometer with fixed “end-to-end” resis-

tance and a 7-decade switched wiper position (Figure C1).

R = 100k

AN86 FC01

SEVEN-DECADE SWITCHED

WIPER POSITION PERMITS

SETTING TO 0.1ppm LINEARITY

Figure C1. Conceptual Kelvin-Varley Divider

AN86 FC02

OUT

INPUT

10k 2k 400Ω80Ω

80Ω

COMMON

Figure C2. A 4-Decade Kelvin-Varley Divider. Additional

Decades Are Implemented By Opening Last Switch, Deleting

Two Associated 80Ω Values and Continuing ÷ 5 Resistor Chains

The actual construction of a 0.1ppm KVD is more artistry

and witchcraft than science. The market is relatively small,

the number of vendors few and resultant price high. If

$13,000 for a bunch of switches and resistors seems

offensive, try building and certifying your own KVD. Figure

C2 shows a detailed schematic.

The KVD shown has a 100kΩ input impedance. A conse-

quence of this is that wiper output resistance is high and

varies with setting. As such, a very low bias current

follower is required to unload the KVD without introducing

significant error. Now, our KVD looks like Figure␣ C3. The

LT1010 output buffer allows driving cables and loads and,

more subtly, maintains the amplifier’s high open-loop

gain.

Figure C3. KVD with Buffer Gives Output Drive Capability

KVD

KVD = ELECTRO SCIENTIFIC INDUSTRIES RV-722,

FLUKE 720A OR JULIE RESEARCH LABS VDR-307

0.1µF

10k

INPUT

AN86 FC03

OUTPUT

–

LT1010LTC1152

Application Note 86

AN86-9

Approach and Error Considerations

This schematic is deceptively simple. In practice, con-

struction details are crucial. Parasitic thermocouples

(Seebeck effect), layout, grounding, shielding, guarding,

cable choice and other issues affect achievable perfor-

mance.

In fact, as good as the chopper-stabilized LTC1152

is with respect to drift, offset, bias current and CMRR,

selection is required if we seek sub-ppm nonlinearity

performance. Figure C4, an error budget analysis, details

some of the selection criteria.

Note 1: See Appendix J, “Cables, Connections, Solder, Layout,

Component Choice, Terror and Arcana,” for relevant tutorial.

Note 2: See above Footnote.

The buffer is tested with Figure C5’s circuit. As the KVD is

run through its entire range, the floating null detector must

remain well within 1ppm (5µV), preferably below 0.5ppm.

This test ensures that all error sources, particularly I

and

CMRR, whose effects vary with operating point, are ac-

counted for. Measured performance indicates the sum of

all errors called out in Figure C4 is well within desired

limits.

In Figure C6, we add offset trim, a stable voltage source

and a second KVD to drive the main KVD. Additionally, an

ensemble of three HP3458A voltmeters monitor the out-

put.

The offset trim bleeds a small current into the main KVD

ground return, producing a few microvolts of offset-trim

range. This functionally trims out all sources of zero error

(amplifier offsets, parasitic thermocouple mismatches

and the like), permitting a true zero volt output when the

main KVD is set to all zeros.

The voltmeters, specified for <0.1ppm nonlinearity on the

10V range, “vote” on the source’s output.

Circuitry Details

Figure C7 is a more detailed schematic. It is similar to

Figure C6 but highlights issues and concerns. The ground-

ing scheme is single point, preventing mixing of return

currents and the attendant errors. The shielded cables

used for interconnections between the KVDs and voltme-

ters should be specified for low thermal activity. Keithley

type SC-93 and Guildline #SCW are suitable. Crush type

copper lugs (as opposed to soldered types) provide lower

parasitic thermocouple activity at KVD and DVM connec-

tion points. However, they must be kept clean to prevent

oxidation, thus avoiding excessive thermal voltages.

copper deoxidant (Caig Labs “Deoxit” D100L) is quite

effective for maintaining such cleanliness. Low thermal

lugs and jacks, preterminated to cables, are also available

(Hewlett-Packard 11053, 11174A) and convenient.

Thermal baffles enclosing KVD and DVM connections tend

Figure C4. Error Budget Analysis for the KVD Buffer.

Selection Permits ≈0.4ppm Predicted Linearity Error

0.1µF

LT1010 OUTPUT

10k

KVD R

= 100k

≈ 30k

WORST-CASE

OUTPUT

RESISTANCE

AN86 FC04

ERROR

SOURCE

OS∆T

CMRR

FINITE GAIN

WORST-CASE

SPEC

5µV

0.05µV/°C

50pA

110dB

140dB

REALISTIC SELECTION

TARGET

0.5µV

0.05µV/°C

10pA

140dB

ERROR IN

PPM

0.1

0.01/°C

0.1

–

LTC1152

Figure C5. Determining Buffer Error By Measuring Input-Output

Deviation with Floating Microvolt Null Detector. Technique

Permits Evaluation of Fixed and Operating Point Induced Errors

AN86 FC05

0.1µF

–

LTC1152

KVD

FLOATING, BATTERY-POWERED

µV NULL DETECTOR

HP-419A

LT1010 OUTPUT

10k

Application Note 86

AN86-10

to thermally equilibrate their associated banana jack ter-

minals, minimizing residual parasitic thermocouple activ-

ity. Additionally, restrict the number of connections in the

signal path. Necessary connections should be matched in

number and material so that differential cancellation oc-

curs. Complying with this guideline may necessitate delib-

erate introduction of solder-copper junctions (marked “X”

on Figure␣ C7) to obtain optimum differential cancellation.

This is normally facilitated by simply breaking the appro-

priate wire or PC trace and soldering it. Ensure that the

introduced thermocouples temperature track the junc-

tions they are supposed to cancel. This is usually accom-

plished by locating all junctions within close physical

proximity.

The noise filtering capacitor at the main KVD is a low

leakage type, with its metal case driven by the output

buffer to guard out surface leakage.

When studying the approach used, it is essential to differ-

entiate between linearity and absolute accuracy. This

eliminates concerns with absolute standards, permitting

certain freedoms in the measurement scheme. In particu-

lar, although single-point grounding was used, remote

sensing was not. This is a deliberate choice, made to

minimize the number of potential error-causing parasitic

thermocouples in the signal path. Similarly, a ratiometric

reference connection between the KVD LTZ1000A voltage

source and the voltmeters was not utilized for the same

reason. In theory, a ratiometric connection affords lower

drift. In practice, the resultant introduced parasitic ther-

mocouples obviate the desired advantage. Additionally,

the aggregate stability of the LTZ1000A reference and the

voltmeter references (also, incidentally, LTZ1000A based)

is comfortably inside 0.1ppm for periods of 10 minutes.

This is more than enough time for a 10-point linearity

measurement.

Note 3: See Appendix J, “Cables, Connections, Solder, Layout,

Component Choice, Terror and Arcana,” for further discussion.

Note 4: The LTZ1000A reference is detailed in Appendix I, “ Voltage

References.”

Figure C6. Simplified Sub-ppm Linearity Voltage Source

KVD

0.1µF

10k

AN86 FC06

–

LT1010

STABLE

VOLTAGE

SOURCE

(LTZ1000A

BASED) LTC1150

MAIN

KVD

OFFSET

TRIM

0.1µF

OUTPUT

0.000000V

5.000000V

WIRE

–

HP3458A

LT1010LTC1152

20k

–V

ADJUST FOR

5.000000V AT A

Application Note 86

AN86-11

Construction

Figures C8 and C9 are photographs of the voltage source

and the reference-buffer box internal construction. The

figure captions annotate some significant features.

Results

This KVD-based, high linearity voltage source has been in

use in our lab for nearly two years. During this period, the

total linearity uncertainty defined by the source and its

monitoring voltmeters has been just 0.3ppm (see Figure

Figure C7. Complete Sub-ppm Linearity Voltage Source

KVD

0.1µF

10k

AN86 FC07

–

LT1010

STABLE

VOLTAGE

SOURCE

(LTZ1000A

BASED) LTC1150

10k

MAIN

KVD

OFFSET

TRIM

HIGH QUALITY

GROUND

0.1µF

OUTPUT

0.000000V

5.000000V

–

HP3458A

LT1010LTC1152

20k

–V

10k

2µF

WIRE

CASE

= SOLDER-COPPER JUNCTION.

PLACEMENT AND NUMBER AS REQUIRED

2µF = POLYSTYRENE. COMPONENT RSCH. CORP.

USE LOW THERMAL, LOW TRIBOELECTRIC

SHIELDED CABLE FOR KVD AND DVM CONNECTIONS.

SEE TEXT

ADJUST FOR

5.000000V AT A

SEE APPENDIX F

FOR CIRCUIT DETAILS

OF LTZ1000A

C10’s measurement regime). This is more than 3 times

better than the desired 1ppm performance, promoting

confidence in our measurements.

Acknowledgments

The author is indebted to Lord Kelvin and to Warren Little

of the C. S. Draper Laboratory (née M. I. T. Instrumentation

Laboratory) standards lab. Warren taught me, with great

patience, the wonders of KVDs some thirty years ago and

I am still trading on his efforts.

Note 5: The author, wholly unenthralled by web surfing, has spent

many delightful hours “surfing the Kelvin.” This activity consists of

dialing various Kelvin-Varley divider settings and noting monitoring

A-to-D agreement within 1ppm. This is astonishingly nerdy behavior,

but thrills certain types.

Application Note 86

AN86-12

Figure C8. The Sub-ppm Linearity Voltage Source. Box Upper Right Is LTZ1000A Based Reference and Buffers. Upper Left Is Offset

Trim. Reference and Main Kelvin-Varley Dividers Are Photo Center—Upper and Center-Middle, Respectively. Three HP3458 DVMs

(Photo Lower) Monitor Output. Computer (Left Foreground) Aids Linearity Calculations

AN86 FC08.tif

Application Note 86

AN86-13

Figure C9. Reference-Buffer Box Construction. LTZ1000A Reference Circuitry Is Photo Lower Left, Buffer Amplifiers Photo Center. Note

Capacitor Case Bootstrap Connection (Photo Center—Right). Single Point Ground Mecca Appears Photo Upper Left. Power Supply

(Photo Top) Mounts Outside Box, Minimizing Magnetic Field Disturbance

Figure C10. Testing Regime for the High Linearity Voltage Source

1. VERIFY KVD LINEARITY BY INTERCOMPARISON AND INDEPENDENT CAL. LAB.

2. TAKE WORST-CASE VOLTMETER ENSEMBLE DEVIATIONS OVER 0V TO 5V, EVERY 0.5V

3. 100 RUNS (10 PER DAY, ONCE PER HOUR)

4. INDICATED RESULT IS 0.3ppm NONLINEARITY

AN86 FC10

AN86 FC09.tif

Application Note 86

AN86-14

APPENDIX D

A PROCESSOR-BASED CODE COMPARATOR

The code comparator enforces the loop by setting the

slave DAC inputs to the code that equalizes the user input

and the LTC2400 A-to-D output. This action is more fully

described on page one of the text.

Figure D1 is the code comparator’s digital hardware. It is

composed of three input data latches and a PIC-16C5X

processor. Inputs include user data (e.g., DAC inputs),

linearity curvature correction (via DIP switches), convert

command (“DA WR”) and a selectable filter time constant.

An output (“DAC RDY”) indicates when the DAC output is

settled to the user input value. Additional outputs and an

input control and monitor the analog section (text Fig-

ure␣ 2) to effect loop closure. Note that although a total of

32 bits are presented to the two 16-bit slave DACs, there

are 8 bits of overlap, allowing a total dynamic range of 24

bits. This provides 4 bits of indexing range below the 20th

bit, ensuring a stable LSB of 1ppm of scale. The 8-bit

overlap assures the loop will always be able to capture the

correct output value.

The processor is driven by software code, authored by

Florin Oprescu, which is described below.

10k

MCLR

CLK

GND

0.1µF

U2 74HCT574

CLK

GND

0.1µF

4MHz

CRYSTAL

U3 74HCT574

CLK

GND

0.1µF

AN86 FD01

U4 74HCT574

USER INPUT

ADC SCK

ADC SDO

LSB OE

MID OE

MSB OE

LSB WR

MSB WR

10k

MCLR

CLKIN

CLKIN2

MSB D7

LSB D0

DAC RDY

DAC LD

MLBYTE

D7 MSB

DAC

COMMANDS

TO ANALOG

SECTION

TO LTC2400

IN ANALOG

SECTION

U1 PIC16C5X

10k

DAC LD

MLBYTE

MSB WR

LSB WR

ADC SDO

ADC SCK

20pF

USER

DATA

INPUT

MSB

DAC RDY

DAC WR

LSB

1ppm

2ppm

4ppm

FILTER

Figure D1. Code Comparator Hardware. User Control Lines Are at Left, Analog Section Connections Appear at Right Side

Application Note 86

AN86-15

;20bit DAC code comparator

;

;***************************************************

; *

; Filename: dac20.asm *

; Date 12/4/2000 *

; File Version: 1.1 *

; *

; Author: Florin Oprescu *

; Company: Linear Technology Corp. *

; *

;***************************************************

;

; Variables

;============

; uses 17 bytes of RAM as follows:

;

; {UB2, UB1, UB0} user input word buffer

;———————————————————————————————————————

; 24 bits unsigned integer (3 bytes):

;

; The information is transferred from the external input register

; into {UB2, UB1, UB0} whenever a “user input update” event

; is detected by testing the timer0 content. Following the data

; transfer, the UIU (“user input update”) flag is set and the DAC

; ready flags RDY and RDY2 are cleared. UB0 uses the same physical

; location as U0. The user input double buffering is necessary

; because the loop error corresponding to the current ADC reading

; must be calculated using the previous user input value.

; The old user input value can be replaced by the new user input

; value only after the loop error calculation.

; The worst case minimum response time to an UIU event must be

; calculated. The user shall not update the external input register

; at intervals shorter than this response time. For the moment the

; program can not block the user access to the external input

; register during a read operation. In such a situation the result

; of the read operation can be very wrong.

;

; UB0 - least significant byte. Same physical location as U0

;

; UB1 - second byte.

;

; UB2 - most significant byte.

;

; {U2, U1, U0} user input word

;——————————————————————————————

; 24 bits unsigned integer (3 bytes):

;

; The information is transferred from {UB2, UB1, UB0[7:4], [0000]}

; into {U2, U1, U0} whenever the UIU flag is found set within the

; CComp (“code comparator”) procedure. The UIU flag is reset

; following the data transfer.

Application Note 86

AN86-16

;

; U0 - least significant byte of current DAC input

; The 4 least significant bits U0[3:0] are set

; to zero.

;

; U1 - second byte of current DAC input

;

; U2 - most significant byte of current DAC input

;

; {CON} control byte

;————————————————————

; (1 byte):

;

; The 3 least significant bits CON[2:0] represent the ADC linearity

; correction factor transferred from UB[2:0] when the UIU flag

; is found set within the CComp procedure - at the same time as the

; {U2, U1, U0} variable is updated.

;

; The effect of CON[2:0] is additive and its weight is as follows:

;

; CON[0] = 1 linearity correction effect is about 1ppm

; CON[1] = 1 linearity correction effect is about 2ppm

; CON[2] = 1 linearity correction effect is about 4ppm

;

; The LTC2400 has a typical 4ppm INL error therefore the default

; curvature correction value can be set at CON[2:0] = 100

;

; CON[3] is the control loop integration factor transferred from

; UB[3] when the UIU flag is found set within the CComp procedure.

; If CON[3]=0, after the control loop error becomes less than 4ppm

; the error correction gain is reduced from 1 to 1/4

; If CON[3]=1, after the control loop error becomes less than 4ppm

; the error correction gain is reduced from 1 to 1/16

;

; CON[7] is used as the UIU (“user input update”) flag. It is set

; when {UB2, UB1, UB0} is updated and it is reset when {U2, U1, U0}

; and CON[3:0] are updated.

;

; CON[6] is used as the RDY (“DAC ready”) flag. It is set when

; the DAC loop error becomes less than 4ppm and it is reset when

; the UIU flag is set.

;

; CON[5] is used as the RDY2 (“DAC ready twice”) flag. It is set

; whenever the DAC loop error becomes less than 4ppm and the RDY

; flag has been previously set. It is reset when the UIU flag is set.

;

; The bit CON[4] is not used and is always set to 0.

;

Application Note 86

AN86-17

; {ADC3, ADC2, ADC1, ADC0} formatted ADC conversion result

;—————————————————————————————————————————————————————————

; 32 bits signed integer (4 bytes).

;

; The ADC reading is necessary only for the calculation of the control

; loop error and in order to save RAM space, it can share the same

; physical space as the loop error variable.

;

; The LTC2400 output format is offset binary. It must be converted

; to 2’s complement before any arithmetic operation. A number of

; possible codes are not valid LTC2400 output codes. If such codes

; are detected it can be inferred that a serial transfer error has

; occurred, the data must be discarded and a new conversion must

; be started. For all LTC2400 devices B31=0 and B30=0 always. In

; addition, with the exception of some early samples of the device

; the sequence B[29:28]=00 should not occur in a valid transaction.

;

; ADC0 - least significant byte

; contains ADC output bits B11(MSBIT) to B4 (LSBIT)

;

; ADC1 - second byte

; contains ADC output bits B19(MSBIT) to B12 (LSBIT)

;

; ADC2 - third byte

; contains ADC output bits B27(MSBIT) to B20 (LSBIT)

;

; ADC3 - most significant byte

; contains ADC output bits ~B29(as 7 MSBITS for

; 2’s complement sign extension) and B28 (LSBIT)

;

; {ADCC} ADC curvature correction

;————————————————————————————————

; 8 bits unsigned integer (1 byte)

;

; The LTC2400 transfer characteristic has a typical INL of about

; 4ppm and a parabolic shape symmetric with respect to mid-scale.

; This error can be corrected to a first and second order and

; ADDC contains the magnitude of this correction.

;

; {ER3, ER2, ER1, ER0} control loop error value

;——————————————————————————————————————————————

; signed integer (4 bytes)

;

; Contains the value of the current control loop error calculated

; as the difference between the previous user input and the last

; ADC reading. It is used to adjust the Low_DAC setting. Uses the

; same physical location as {ADC3, ADC2, ADC1, ADC0}:

;

; ER0 - least significant byte, same location as ADC0

;

; ER1 - second byte, same location as ADC1

Application Note 86

AN86-18

;

; ER2 - third byte, same location as ADC2

;

; ER3 - most significant byte, same location as ADC3

;

; {DL3, DL2, DL1, DL0} Low DAC control value

;———————————————————————————————————————————

; signed integer (4 bytes):

;

; Contains the Low_DAC setting in a 2’s complement, 32 bit

; format. Must be initialized to 0!

;

; DL0 - least significant byte - used for Low_DAC

; control

;

; DL1 - second byte - used for Low_DAC control after

; conversion to offset binary format {DL1, DL0}

;

; DL2 - third byte - may be only 00 or FF

;

; DL3 - most significant byte - may be only 00 or FF

;

; {INDX} Index variable for various program functions

;————————————————————————————————————————————————————

; 1 byte.

;

; {TMPV} Temporary variable for various program functions

;————————————————————————————————————————————————————————

; 1 byte.

;

; Algorithm

;===========

;

; After each ADC conversion cycle the processor calculates the control

; loop error value as the difference between the desired output and

; the latest conversion result. Than it updates the DACs command

; such as to reduce the error magnitude. A new ADC conversion cycle

; is started following the DACs update operation.

;

; In order to maintain adequate control loop stability it is necessary

; for the DACs and the associated amplifiers to settle to better than

; 20 bits accuracy before the ADC starts sampling the system output. For

; an LTC2400 based system this settling time is 66ms.

;

; Initialization:

; Initializes the PIC controller and the hardware interface

; and starts the Scan procedure.

;

Application Note 86

AN86-19

; 1. Load ADC control port with default values

; SCKAD = 0

; SDOAD = 1

; 2. Set ADC control port I and O pins

; SCKAD = output

; SDOAD = input

; 3. Load register control port with default values

; NCSR[2:0] = 111

; NCSD[1:0] = 11

; ADDAC = 1

; NLDAC = 1

; DACRDY = 0

; 4. Set register control port in output mode

; 5. Set data bus to default value DBUS[7:0]=0x00

; 6. Set data bus in output mode

; 7. Initialize internal registers and variables:

; OPTION = 0x2F

; Timer0 used as counter is incremented by low-to-high edge

; Prescaler works with watch dog timer in div128 mode

; CON = 0x80

; Simulate a UIU “user interface update” event to force

; the update of both Low_DAC and High_DAC

; {DL3, DL2, DL1, DL0} = 0

; { U2, U1, U0} = 0

; 8. Update hardware using the initialized variables

; 9. Start new ADC conversion by reading and discarding

; 32 serial bits.

; 10.Start the Scan procedure

;

; Scan:

; Continuously looks for “user input update” events. When

; a “user input update” event is detected updates the

; input buffer {UB2, UB1, UB0}, resets timer, sets UIU flag

; and resets RDY and RDY2 flags.

;

; The active low write signal for the external input register

; (which is the same as the user interface NWR input signal)

; is driven by the user and it is connected to the counter

; input of Timer0. The Timer0 is used in counter mode without a

; prescaler and it increments whenever a low-to-high transition

; is detected at its input. This is the same transition which

; latches in the input register a new user command.

; Because of the PIC controller timing constraints, this write

; signal must be maintained low for at least 2*Tosc + 20ns

; where Tosc is the maximum PIC clock period. When a 4 MHz

; clock is used for the PIC processor, the low time must be

; longer than about 520ns.

;

; 1. Test for “user input update” events by testing the Timer0

; value.

; If Timer0>0 an UIU event has occurred. Reset the timer

; and answer Yes.

; If Timer0=0 answer No.

Application Note 86

AN86-20

; 1.1 If Yes, read input latch into {UB2, UB1, UB0},

; reset DACRDY output line, set UIU flag and

; and reset RDY and RDY2 flags (CON[7:5]=100)

; Than continue

; 1.2 If No continue

;

; Continuously looks for the ADC end of conversion event. When

; the “end of conversion” is detected it reads the 28 most

; significant bits from the ADC and it constructs the ADC

; result {ADC3, ADC2, ADC1, ADC0} in 2’s complement format

; If ADC3[1] == 0 => ADC3[7:1]=1111 111

; If ADC3[1] == 1 => ADC3[7:1]=0000 000

; For very early LTC2400 samples only, it is possible

; to obtain as a valid 0 conversion result ADC3[1:0]=00

; In this case:

; If ADC3[1:0] == 0 => ADC3=0

; It also calculates the first (x1) and second (x2) order ADC

; curvature correction ADCC as follows:

; x1 = {0x00, 0x80} -

; -abs({ADC3, ADC2, ADC1, ADC0}/(2^16)-{0x00, 0x80})

; x2 = {0x00, 0x40} -

; -abs({0x00,{0,ADC2[6:0]},ADC1,ADC0}/(2^16)-{0x00,0x40})

; ADCC = floor((x1 + x2/2) * {00000 CON[2:0]} / 4 )

; The actual implementation uses only the least significant

; byte of x without any substantial additional error.

; Thus the above relation can be modified as follows:

; ADCC = floor((abs(ADC2) + abs({ADC2[6],ADC2[6:0]})/2) *

; * {00000 CON[2:0]} / 4 )

; The maximum correction range is about 7ppm INL at mid

; scale for CON[2:0] = 111.

;

; 2. Test for ADC “end of conversion” event by testing the

; value of the ADC_SDO signal.

; If ADC_SDO = LOW answer Yes.

; If ADC_SDO = HIGH answer No.

; 2.1 If Yes read 28 most significant bits from the ADC,

; update {ADC3, ADC2, ADC1, ADC0} and calculate the

; curvature correction byte ADCC. Than start the CComp

; procedure.

; It should be noticed that while reading the first 28

; most significant bits from the ADC the controller

; generates 27 serial clock pulses. An additional 5 serial

; clock pulses (for a total of 32) are necessary to restart

; the conversion.

; 2.2 If No restart the Scan procedure.

;

; CComp:

; Calculates the current control loop error as:

;

; error = current_user_input - ADC_reading +

; + new_user_input_LSB - current_user_input_LSB

;

Application Note 86

AN86-21

; The curvature correction is included in the ADC

; conversion result and is always positive therefore:

;

; ADC_reading = {ADC3, ADC2, ADC1, ADC0} +

; + { 0, 0, 0, ADCC}

;

; The term “new_user_input_LSB - current_user_input_LSB”

; represents the residue of the new user command which

; is added to the Low_DAC.

;

; {ER3, ER2, ER1, ER0} =

; = {0, U2, U1, U0} - {ADC3, ADC2, ADC1, ADC0} -

; - { 0, 0, 0, ADCC} +

; + {0, 0, 0, UB0} - { 0, 0, 0, U0} =

;

; = {0, U2, U1, UB0} - {ADC3, ADC2, ADC1, ADC0} -

; - { 0, 0, 0, ADCC}.

;

; The loop error {ER3, ER2, ER1, ER0} is a 32 bit signed number

; and the weight of the least significant bit is 1/16ppm of

; the ADC reference voltage. A 4ppm error value is represented

; as {0, 0, 0, 0x40}.

;

; The ADC output noise is dominated by thermal noise and has a

; white distribution. The control loop noise can be reduced by

; the square root of N by averaging N successive ADC readings.

; The obvious penalty is a slow settling time. Due to the

; limited amount of RAM available a direct implementation

; of this noise reduction strategy is difficult. In an equivalent

; implementation, when the absolute value of the loop error

; {ER3, ER2, ER1, ER0} decreases below a certain threshold, the

; gain of the error correction loop can be decreased. The default

; threshold is set at a very conservative 4ppm. This value must

; always be larger than the peak noise level of the ADC. A very

; quiet implementation can probably operate with a threshold of

; 2ppm. If CON[3]=0 the gain of the error correction loop is

; decreased from 1 to 1/4. If CON[3]=1 the gain of the error

; correction loop is decreased from 1 to 1/16.

;

; The High_DAC is always controlled by the 16 most significant

; bits of the most recent user command {UB2, UB1}

;

; The Low_DAC is controlled by the {DL3, DL2, DL1, DL0}

; variable which integrates the control loop error. Under

; correct operating condition abs({DL3, DL2, DL1, DL0})<2^15.

; In order to avoid roll-overs during large transients the

; {DL3, DL2, DL1, DL0} must be clamped within the +/- 2^15 range.

; The 16 bit Low_DAC can than be controlled by {DL1, DL0}

; after conversion to offset binary format.

;

Application Note 86

AN86-22

; The DACRDY output line reflects the state of the

; internal RDY2 flag.

;

; After the updates are completed we must start a new ADC

; conversion by completing the serial transfer.

;

; 1. Test if UIU flag is set

; 1.1 If Yes, move UB[3:0] into CON[3:0]

; and {UB0[7:4], 0000} into U0. The last ADC result

; is curvature corrected using the previous CON[3:0] value!.

; 2. Calculate {ER3, ER2, ER1, ER0}.

; 3. Test if UIU flag is set

; 3.1 If Yes, move {UB2, UB1} into {U2, U1} and

; clear UIU, RDY and RDY2 flags (CON[7:5]=000 )

; 3.2 If No, test if abs({ER3, ER2, ER1, ER0}) < 4ppm

; 3.2.1 If Yes, test if CON[6]=1 (RDY flag)

; 3.2.1.1 If Yes, set RDY2 flag (CON[5]=1 )

; 3.2.1.2 If No, set RDY flag (CON[6]=1 )

; and test if CON[3]=0 (filter flag)

; 3.2.1.3 If Yes, {ER3, ER2, ER1, ER0} =

; = {ER3, ER2, ER1, ER0}/4

; 3.2.1.4 If No, {ER3, ER2, ER1, ER0} =

; = {ER3, ER2, ER1, ER0}/16

; 3.2.2 If No, clear UIU, RDY and RDY2

; flags (CON[7:5]=000 )

; 4 {DL3, DL2, DL1, DL0} = {DL3, DL2, DL1, DL0} +

; +{ER3, ER2, ER1, ER0}.

; 5. Update High_DAC, Low_DAC and DACRDY output line

; 6. Read the 4 least significant bits from ADC and start

; a new conversion

; 7. Restart the Scan procedure

;

; Hardware resources

;====================

;

; Uses 8 input/output pins, 9 output pins, 1 input pin and 1

; counter input pin

;

; DBUS[7:0] data bus

;———————————————————

; 8 bit bi-directional data bus is used to read the 20 bit input

; command IC[19:0], the one bit filter selection FS[0] and the 3 bit

; curvature correction selection CC[2:0]. It is also used to write

; the 16 bit Low_DAC command LDAC[15:0] and the 16 bit High_DAC

; command HDAC[15:0].

;

; assigned to PIC port C[7:0]

;

Application Note 86

AN86-23

; The data format for the read and write operations is as follows:

;

; DBUS[ 7:0] = IC[19:12] when NCSR[2] = 0

; DBUS[ 7:0] = IC[11: 4] when NCSR[1] = 0

; DBUS[ 7:0] = {IC[3:0], FS[0], CC[2:0]} when NCSR[0] = 0

; LDAC[ 7:0] = DBUS[7:0] when NCSD[0] = 0 and ADDAC = 0

; LDAC[15:8] = DBUS[7:0] when NCSD[0] = 0 and ADDAC = 1

; HDAC[ 7:0] = DBUS[7:0] when NCSD[1] = 0 and ADDAC = 0

; HDAC[15:7] = DBUS[7:0] when NCSD[1] = 0 and ADDAC = 1

;

; NCSR[2:0] active low output enable controls for input registers

;————————————————————————————————————————————————————————————————

; 3 output lines used to selectively enable the three 8-bit input

; registers in order to read the user updated DAC command, the 3

; curvature correction bits and the one filter control bit.

;

; NCSR[0] enables the low input byte (LSB) and is assigned to port B[0]

;

; NCSR[1] enables the second input byte and is assigned to port B[1]

;

; NCSR[2] enables the high input byte (MSB) and is assigned to port B[2]

;

; NCSD[1:0] active low input enable controls for the DACs

;————————————————————————————————————————————————————————

; 2 output lines used to selectively enable the two DACs

;

; NCSD[0] enables the Low_DAC and is assigned to port B[3]

;

; NCSD[1] enables the High_DAC and is assigned to port B[4]

;

; ADDAC DAC address control

;——————————————————————————

; output line. A low enables a write operation to the low byte of

; Low_DAC or High_DAC. A high enables a write operation to the high

; byte of Low_DAC or High_DAC.

;

; ADDAC is assigned to port B[5]

;

Application Note 86

AN86-24

; NLDAC active low DAC load control

;——————————————————————————————————

; output line. A high to low transition on this line updates the

; Low_DAC and High_DAC output values

;

; NLDAC is assigned to port B[6]

;

; DACRDY active high ready output signal

;———————————————————————————————————————

; output line. Indicates that the control loop error has been

; within a +/- 4ppm range for at least 250 ms

;

; DACRDY is assigned to port B[7]

;

; SCKAD external serial clock line for the ADC

;—————————————————————————————————————————————

; output line. ADC external serial clock. An external 10Kohm

; pull-down resistor is necessary on this line for correct

; power-up configuration.

;

; SCKAD is assigned to port A[0]

;

; SDOAD serial data line from ADC

;————————————————————————————————

; input line. Used to read ADC serial data.

;

; SDOAD is assigned to port A[1]

;

; NWRUI active low user interface write control

;——————————————————————————————————————————————

; input line. The user must bring this line low in order to update

; the DAC input value. A minimum low and high time is required !

;

; NWRUI is assigned to TOCKI counter input pin

;

Application Note 86

AN86-25

; The spare I/O pins A[3:2] are configured as outputs and maintained LOW.

;

;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;;

list p=16c55A ; list directive to define processor

#include <p16c5x.inc> ; processor specific variable definitions

__CONFIG _CP_OFF & _WDT_ON & _XT_OSC

;VARIABLE DEFINITIONS

;====================

UB0 EQU H’0008' ; user input word buffer LSB

UB1 EQU H’0009' ; user input word buffer second byte

UB2 EQU H’000A’ ; user input word buffer MSB

U0 EQU H’0008' ; user input word LSB

U1 EQU H’000B’ ; user input word second byte

U2 EQU H’000C’ ; user input word MSB

CON EQU H’000D’ ; control byte

ADC0 EQU H’000E’ ; ADC conversion result LSB

ADC1 EQU H’000F’ ; ADC conversion result second byte

ADC2 EQU H’0010' ; ADC conversion result third byte

ADC3 EQU H’0011' ; ADC conversion result MSB

ADCC EQU H’0012' ; ADC curvature correction byte

ER0 EQU H’000E’ ; control loop error LSB

ER1 EQU H’000F’ ; control loop error second byte

ER2 EQU H’0010' ; control loop error third byte

ER3 EQU H’0011' ; control loop error MSB

DL0 EQU H’0013' ; Low_DAC LSB

DL1 EQU H’0014' ; Low_DAC second byte

DL2 EQU H’0015' ; Low_DAC third byte

DL3 EQU H’0016' ; Low_DAC MSB

INDX EQU H’0017' ; index variable

TMPV EQU H’0018' ; temporary variable

#define OPRDF 0x2F ; OPTION register default value

#define CONDF 0x80 ; CON register default value

Application Note 86

AN86-26

;HARDWARE ASSIGNMENT DEFINITIONS

;===============================

#define DBUS PORTC ; 8bit I/O data bus

#define REGCN PORTB ; register control port

#define REGDF 0x7F ; register control port default value

#define NCSR0 PORTB,0 ; LSB input register active low output enable

#define NCSR1 PORTB,1 ; second byte input register active low output enable

#define NCSR2 PORTB,2 ; MSB input register active low output enable

#define NCSD0 PORTB,3 ; Low_DAC active low write enable

#define NCSD1 PORTB,4 ; High_DAC active low write enable

#define ADDAC PORTB,5 ; address bit for Low_DAC and High_DAC

#define NLDAC PORTB,6 ; active low load control for Low_DAC and High_DAC

#define DACRDY PORTB,7 ; 20bit_DAC ready indicator

#define ADCCN PORTA ; ADC control port

#define ADCTR 0x02 ; ADC control port configuration

; SDOAD input, the rest outputs

#define ADCDF 0x02 ; ADC control port default value

#define SCKAD PORTA,0 ; ADC external serial clock

#define SDOAD PORTA,1 ; ADC serial data output

;THE CODE

;===============================

RESET ORG 0x1FF ; processor reset vector

goto start

ORG 0x000 ;Initialization procedure

;————————————————————————

start movlw ADCDF ;write ADC control port default value

movwf ADCCN ;

movlw ADCTR ;set the I and O pin states for the

tris ADCCN ;ADC control port

;

movlw REGDF ;write register control port default value

movwf REGCN ;

clrw ;set register control port pins as

tris REGCN ;output only

;

movwf DBUS ;set DBUS default value of 0

tris DBUS ;set DBUS as output

;

movlw OPRDF ;set OPTION register default value

option ;

;

Application Note 86

AN86-27

clrf TMR0 ;

btfss STATUS,NOT_TO ;if this is not a power-on reset

movwf TMR0 ;load Timer0 with a nonzero value

;to force an initial read of the

;external input register

;

clrf DL3 ;initialize {DL3, DL2, DL1, DL0}=0

clrf DL2 ;

clrf DL1 ;

clrf DL0 ;

clrf U2 ;initialize {U2, U1, U0}=0

clrf U1 ;

clrf U0 ;

;

movlw CONDF ;set CON variable default value

movwf CON ;

;prepare to trigger a new ADC conversion

;after completing a hardware update

movlw 0x20 ;read and discard 32 serial bits from

movwf INDX ;the ADC

;

goto iupdt ;go to the hardware update function

;ADC output buffer flush function

;————————————————————————————————

fladc movlw 0x20 ;reads and discards 32 serial bits from

movwf INDX ;the ADC

;ADC dummy serial read function

;——————————————————————————————

;reads and discards the number of serial

;bits indicated by the INDX variable

rddmy bsf SCKAD ;low-to-high ADC serial clock edge

bcf SCKAD ;high-to-low ADC serial clock edge

decfsz INDX,1 ;test if we read enough bits

goto rddmy ;if No, read one more bit

btfss SDOAD ;if Yes test that a new conversion has started

goto fladc ;if No, there is an interface problem. Flush the

;ADC output buffer and start a new conversion

goto scan ;if Yes restart the scan procedure

;external input register read function

;—————————————————————————————————————

rduiu movlw 0xFF ;input register read routine

tris DBUS ;set data bus in read mode (input)

bcf NCSR0 ;output enable for input reg. LSB

nop ;wait for data bus to settle

movf DBUS,0 ;read input reg. LSB

bsf NCSR0 ;output disable for input reg. LSB

bcf NCSR1 ;output enable for input reg. second byte

movwf UB0 ;store input reg. LSB into input buffer

movf DBUS,0 ;read input reg. second byte

bsf NCSR1 ;output disable for input reg. second byte

Application Note 86

AN86-28

bcf NCSR2 ;output enable for input reg. MSB

movwf UB1 ;store input reg. second byte into input buffer

movf DBUS,0 ;read input reg. MSB

movwf UB2 ;store input reg. MSB into input buffer

clrw ;terminate input reg. read operation

bsf NCSR2 ;output disable for input reg. MSB

tris DBUS ;return data bus to write mode

clrf TMR0 ;clear Timer0 to continue wait for a UIU event

bcf DACRDY ;signal user that input data has been read

bsf CON,7 ;set UIU flag

bcf CON,6 ;clear RDY flag

bcf CON,5 ;clear RDY2 flag

;scan procedure

;——————————————

;monitors UIU and end-of-conversion events

scan movf TMR0,1 ;test if Timer0 = 0

btfss STATUS,Z ;if Timer0=0 no UIU has occurred, skip next

goto rduiu ;a user interface update has occurred

;go and read the new DAC input data

btfsc SDOAD ;test ADC end of conversion signal

goto scan ;conversion not ready, rescan

;ADC serial read function

;————————————————————————

rdadc movlw 0x1B ;ADC conversion is done, read first 28 bits

movwf INDX ;the first bit must be “0” to get here

;so do not bother with it

rdbit bsf SCKAD ;low-to-high ADC serial clock edge

bcf SCKAD ;high-to-low ADC serial clock edge

bcf STATUS,C ;move ADC output bit to carry. First clear carry

btfsc SDOAD ;read ADC output bit

bsf STATUS,C ;if ADC output is “1” set carry

rlf ADC0,1 ;load carry as msb of ADC result

rlf ADC1,1 ;and shift left all 4 bytes of the ADC result

rlf ADC2,1 ;

rlf ADC3,1 ;

decfsz INDX,1 ;test if all 28 bits have been read

goto rdbit ;if not, continue

;

;we have skipped the first ADC bit (ADC bit31=0)

;which has been tested as =0 when we detected the

;end of conversion.

;we have read 27 additional bits and have generated

;27 clock pulses. To restart the conversion we must

;produce the 5 remaining clock pulses

Application Note 86

AN86-29

;verify validity of ADC serial data and format it

;————————————————————————————————————————————————

btfsc ADC3,2 ;test if the ADC bit30 is “0”

goto fladc ;if not there is an interface problem. Flush the

;ADC output buffer and start a new conversion

;if yes, put the ADC result in 2’s complement form

movlw 0x03 ;first clear the 6 most significant bits in ADC3

andwf ADC3,1 ;

btfsc STATUS,Z ;tests for the [ADC_B29,ADC_B28]=00 ADC output

goto rdend ;if Yes the formatting is completed.

;in very early LTC2400 samples the ADC output code

;[ADC_B29,ADC_B28]=00 is valid

; goto fladc ;for current LTC2400 devices improved error

;detection capability is obtained if the

;previous line is replaced with this line.

;The replacement is not mandatory.

;For current LTC2400 parts the output code

;[ADC_B29,ADC_B28]=00 is not valid thus it may

;be assumed that an ADC interface error has

;occurred. Flush the ADC output buffer and start

;a new conversion

movlw 0x02 ;if No, convert ADC3 in 2’s complement form

btfss ADC3,1 ;

movlw 0xFE ;

xorwf ADC3,1 ;

;curvature correction calculator

;———————————————————————————————

;first order curvature correction multiplier

;use ADC2[7:0] as a 2’s complement number

rdend movf ADC2,0 ;calculate abs(ADC2)

btfsc ADC2,7 ;if ADC2[7]=0 w = ADC2

comf ADC2,0 ;else w = !ADC2

movwf ADCC ;ADCC=w=abs(ADC2)

;second order curvature correction multiplier

;use ADC2[6:0] as a 2’s complement number

movf ADC2,0 ;calculate abs(ADC2[6:0])

btfsc ADC2,6 ;if ADC2[6]=0 w = ADC2

comf ADC2,0 ;else w = !ADC2

movwf TMPV ;TMPV=w=abs(ADC2[6:0])

rrf TMPV,0 ;w=TMPV/2 in order to scale the second order

;curvature correction

andlw 0x1f ;clear 3 MSB of w to complete calculation

addwf ADCC,0 ;w=abs(ADC2)+abs(ADC2[6:0])/2

movwf TMPV ;TMPV contains the curvature correction multiplier

;

clrf ADCC ;

bcf STATUS,C ;clear carry for div-by-2 operation

btfsc CON,2 ;if CON[2]=1

Application Note 86

AN86-30

movwf ADCC ;ADCC=ADCC+abs(ADC2)

rrf TMPV,1 ;TMPV=TMPV/2

movf TMPV,0 ;

bcf STATUS,C ;clear carry for div-by-2 operation

btfsc CON,1 ;if CON[1]=1

addwf ADCC,1 ;ADCC=ADCC+abs(ADC2)/2

rrf TMPV,1 ;TMPV=TMPV/2

movf TMPV,0 ;

btfsc CON,0 ;if CON[0]=1

addwf ADCC,1 ;ADCC=ADCC+abs(ADC2)/4

;code comparator procedure

;—————————————————————————

ccomp btfss CON,7 ;if the UIU flag is clear

goto ercalc ;skip CON[3:0] and U0 update

movlw 0xF0 ;else update CON[3:0]

andwf CON,1 ;clear CON[3:0]

movlw 0x0F ;extract new CON[3:0]

andwf UB0,0 ;from input buffer

iorwf CON,1 ;and load it

movlw 0xF0 ;move UB[7:4] to U0[7:4]

andwf UB0,1 ;UB0 and U0 use the same

;physical location

;calculate control loop error

;————————————————————————————

ercalc comf ADCC,1 ;ADCC 1’s complement

comf ADC0,1 ;ADC0 1’s complement

movlw 0x02 ;add carry-in for ADCC and for ADC0

;2’s complement conversion

clrf TMPV ;prepare carry-out accumulator

addwf UB0,0 ;w=carry-in + UB0

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

addwf ADCC,0 ;w=carry-in + UB0 - ADCC

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

addwf ADC0,1 ;ER0=UB0 - ADC0 - ADCC

;has same location as ADC0

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

Application Note 86

AN86-31

comf ADC1,1 ;ADC1 1’s complement

movlw 0xff ;w=0xff (1’s complement of ADCC second byte)

addwf TMPV,0 ;w=0xff + carry-in

clrf TMPV ;prepare carry-out accumulator

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

addwf U1,0 ;w=0xff + carry-in + UB1

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

addwf ADC1,1 ;ER1=U1 - ADC1 - 0 + carry-in

;has same location as ADC1

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

comf ADC2,1 ;ADC2 1’s complement

movlw 0xff ;w=0xff (1’s complement of ADCC third byte)

addwf TMPV,0 ;w=0xff + carry-in

clrf TMPV ;prepare carry-out accumulator

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

addwf U2,0 ;w=0xff + carry-in + UB2

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

addwf ADC2,1 ;ER2=U2 - ADC2 - 0 + carry-in

;has same location as ADC2

btfsc STATUS,C ;if there is a carry-out

incf TMPV,1 ;accumulate it

comf ADC3,1 ;ADC3 1’s complement

decf TMPV,1 ;ADCC 2’s complement term. The next

;carry-in is not useful - discard

movf TMPV,0 ;w=carry-in

addwf ADC3,1 ;ER3= 0 - ADC3 - 0 + carry-in

;has same location as ADC3

btfsc CON,7 ;test if the UIU flag is set

goto lduiu ;go to U1, U2 update

Application Note 86

AN86-32

;error comparator

;————————————————

;calculate absolute value of loop error and

;compare loop error magnitude with the 4ppm

;threshold

movf ER3,0 ;W = ER3

btfsc ER3,7 ;test if {ER3, ER2, ER1, ER0} < 0

comf ER3,0 ;if yes W = -ER3

btfss STATUS,Z ;test if W=0

goto nrdy ;if not absolute error >= 4ppm

movf ER2,0 ;W = ER2

btfsc ER3,7 ;test if {ER3, ER2, ER1, ER0} < 0

comf ER2,0 ;if yes W = -ER2

btfss STATUS,Z ;test if W=0

goto nrdy ;if not absolute error >= 4ppm

movf ER1,0 ;W = ER1

btfsc ER3,7 ;test if {ER3, ER2, ER1, ER0} < 0

comf ER1,0 ;if yes W = -ER1

btfss STATUS,Z ;test if W=0

goto nrdy ;if not absolute error >= 4ppm

movf ER0,0 ;W = ER0

btfsc ER3,7 ;test if {ER3, ER2, ER1, ER0} < 0

comf ER0,0 ;if yes W = -ER0

andlw 0xC0 ;keep only W[7:6] which are bits >= 4ppm

btfss STATUS,Z ;test if W[7:6]=0

goto nrdy ;if not absolute error >= 4ppm

;if we are here the absolute loop error is

;less than 4 ppm. Set the flags and

;scale the loop error.

btfsc CON,6 ;test if RDY flag is already set

bsf CON,5 ;if Yes, set RDY2 flag

bsf CON,6 ;set RDY flag in any case

Application Note 86

AN86-33

;error scaling

;—————————————

;reduce error correction value for loop

;damping and ADC noise reduction

btfsc CON,3 ;test if CON[3]=0

goto div4 ;if Yes ER0=ER0/4

;if No ER0=ER0/16

rrf ER0,1 ;*1/2

rrf ER0,0 ;*1/2

andlw 0x3F ;clear 2 most significant bits

btfsc ER3,7 ;if {ER3, ER2, ER1, ER0} < 0

iorlw 0xC0 ;set 2 most significant bits

movwf ER0 ;ER0=ER0/4

div4 rrf ER0,1 ;*1/2

rrf ER0,0 ;*1/2

andlw 0x3F ;clear 2 most significant bits

btfsc ER3,7 ;if {ER3, ER2, ER1, ER0} < 0

iorlw 0xC0 ;set 2 most significant bits

movwf ER0 ;ER0=ER0/4

goto eracc ;go to error accumulator

;load latest user input

;——————————————————————

lduiu movf UB1,0 ;

movwf U1 ;U1=UB1

movf UB2,0 ;

movwf U2 ;U2=UB2

nrdy movlw 0x1F ;

andwf CON,1 ;clear UIU, RDY and RDY2 flags

;error accumulator

;—————————————————

;adds the current loop error to the

;previous Low_DAC control value

;{DL3, DL2, DL1, DL0}={DL3, DL2, DL1, DL0}+

; +{ER3, ER2, ER1, ER0}

eracc movf ER0,0 ;the carry-in is 0

clrf TMPV ;clear carry-in accumulator

addwf DL0,1 ;DL0=DL0+ER0

btfsc STATUS,C ;if there is a carryout

incf TMPV,1 ;accumulate in carry-in

movf TMPV,0 ;load carry-in

clrf TMPV ;clear carry-in accumulator

addwf ER1,0 ;W=ER1+carry-in

btfsc STATUS,C ;if there is a carryout

incf TMPV,1 ;accumulate in carry-in

addwf DL1,1 ;DL1=DL1+ER1

btfsc STATUS,C ;if there is a carryout

incf TMPV,1 ;accumulate in carry-in

movf TMPV,0 ;load carry-in

clrf TMPV ;clear carry-in accumulator

addwf ER2,0 ;W=ER2+carry-in

Application Note 86

AN86-34

btfsc STATUS,C ;if there is a carryout

incf TMPV,1 ;accumulate in carry-in

addwf DL2,1 ;DL2=DL2+ER2

btfsc STATUS,C ;if there is a carryout

incf TMPV,1 ;accumulate in carry-in

movf TMPV,0 ;load carry-in

addwf ER3,0 ;W=ER3+carry-in

addwf DL3,1 ;DL3=DL3+ER3

;Low_DAC control truncation

;——————————————————————————

;limits the {DL3, DL2, DL1, DL0} range to

; abs({DL3, DL2, DL1, DL0}) < 2^15 by

;truncation

btfss STATUS,Z ;test if DL3=0

goto negpot ;if No, DL may be negative

;if Yes, DL is positive

;test for overflow (>= 2^15)

movf DL2,1 ;

btfss STATUS,Z ;test if DL2=0

goto ovflow ;if No, DL >= 2^15, must truncate

;if Yes continue testing for overflow

btfsc DL1,7 ;test if DL1[7]=1

goto ovflow ;if No, DL >= 2^15, must truncate

goto updt ;if Yes we are done with DL range control

ovflow clrf DL3 ;if we arrive here DL >= 2^15. Must

clrf DL2 ;truncate to DL=2^15-1 => DL3=DL2=0

movlw 0xFF ;and DL1=0xEF, DL0=0xFF

movwf DL0 ;

movwf DL1 ;

bcf DL1,7 ;

goto updt ;done with overflow correction

Application Note 86

AN86-35

udflow clrf DL1 ;if we arrive here DL < -2^15. Must

bsf DL1,7 ;truncate to DL=-2^15-1 => DL3=DL2=0xFF

clrf DL0 ;and DL1=0x80, DL0=0

movlw 0xFF ;

movwf DL3 ;

movwf DL2 ;

goto updt ;done with underflow correction

negpot btfss DL3,7 ;DL may be negative. Test if DL3[7]=1

goto ovflow ;if No, DL > 2^15, must truncate

incf DL3,0 ;if Yes, DL <0.

btfss STATUS,Z ;test if DL3=FF

goto udflow ;if No, DL < -2^15, must truncate

incf DL2,0 ;if Yes continue testing for underflow

btfss STATUS,Z ;test if DL2=FF

goto udflow ;if No, DL < -2^15, must truncate

;if Yes continue testing for underflow

btfss DL1,7 ;test if DL1[7]=0

goto udflow ;if No, DL < -2^15, must truncate

;if Yes we are done with DL range control

;Hardware update function

;————————————————————————

;Low_DAC and High_DAC update

;

;This is the hardware update function

;entry point for normal operation.

;

updt movlw 0x05 ;prepare to generate the last 5 ADC external

movwf INDX ;serial clock pulses

;when going to restart the scan procedure

;at the end of the hardware update function

;This will trigger a new ADC conversion.

;

;This is the hardware update function

;entry point for initial operation.

;The INDX variable has been initialized

;before to 0x2F which will generate

;32 serial clock pulses to the ADC thus

;starting a new conversion

;

Application Note 86

AN86-36

iupdt clrw ;set the data bus in write mode

tris DBUS ;

bcf ADDAC ;set DAC address for LSB

movf U1,0 ;load High_DAC LSB

movwf DBUS ;

bcf NCSD1 ;write to High_DAC

bsf NCSD1 ;

movf DL0,0 ;load Low_DAC LSB

movwf DBUS ;

bcf NCSD0 ;write to Low_DAC

bsf NCSD0 ;

bsf ADDAC ;set DAC address for MSB

movf U2,0 ;load High_DAC MSB

movwf DBUS ;

bcf NCSD1 ;write to High_DAC

bsf NCSD1 ;

movlw 0x80 ;change DL1 to offset binary

xorwf DL1,0 ;load Low_DAC MSB

movwf DBUS ;

bcf NCSD0 ;write to Low_DAC

bsf NCSD0 ;

bcf NLDAC ;load Low_DAC and High_DAC

bsf NLDAC ;

;DACRDY output update

btfsc CON,5 ;test if RDY2 flag is set

bsf DACRDY ;if Yes, set DACRDY output

btfss CON,5 ;if No

bcf DACRDY ;and only if No, clear DACRDY output

;

clrwdt ;clear watch dog timer

;

goto rddmy ;generate the necessary number

;of ADC serial clock pulses in order

;to start a new conversion

END ; directive ‘end of program’

Application Note 86

AN86-37

APPENDIX E

LINEARITY AND OUTPUT RANGE OPTIONS

The LTC2400 used as the feedback A-to-D element in the

DAC has a typical ±2ppm residual nonlinearity. Figure E1’s

lower curve shows this, along with the first order correc-

tion necessary (upper curve) to get nonlinearity inside

1ppm (center curve). If true 1ppm performance is neces-

sary, the software based correction described in Appendix␣ D

can be utilized. The software generates the desired “in-

verted bowl” correction characteristic. The correction may

be set to complement the residual nonlinearity character-

istics of any individual LTC2400 via DIP switches at the

code comparator.

Note 1: The strata is becoming rarified when “error contribution” is

delineated in fractional parts-per-million and the yearly drift rate noted.

OUTPUT CODE (DECIMAL)

0 8,338,608 16,777,215

LINEARITY ERROR (ppm)

AN86 FE01

–2

–4

–6

–8

–10

CORRECTION

CHARACTERISITIC

INHERENT

ERROR

IDEAL CORRECTED ERROR

Figure E1. LTC2400 A-to-D Inherent Residual Linearity Error (Lower),

Correction Characteristic (Upper) and Resultant Corrected Linearity

(Center). Correction Ensures <1ppm Nonlinearity

The LTC2410 offers another approach to improved linear-

ity. This LTC2400 variant has improved linearity but speci-

fies a maximum 2.5V input range. Figure E2 divides the

DAC output with a precision resistor ratio set, allowing

LTC2410 use while maintaining the 5V full-scale output.

The disadvantage of this approach is the ratio set’s addi-

tional 0.1ppm/°C and 5ppm/year error contribution.

Fig-

ure E3 is similar, although the ratio set’s new value permits

a 10V full-scale output.

FROM

DAC OUTPUT

AMPLIFIER

DIGITAL OUTPUT

TO CODE COMPARATOR

OUTPUT

0V TO 5V

AN86 E02

*10ppm RATIO SET

VISHAY VHD-200

0.1ppm/°C

LTC2410

(PARTIAL)

(2.5V

MAX)

FROM

DAC OUTPUT

AMPLIFIER

DIGITAL OUTPUT

TO CODE COMPARATOR

OUTPUT

0V TO 10V

AN86 E03

2.5k

*10ppm RATIO SET

VISHAY VHD-200

0.1ppm/°C

7.5k

LTC2410

(PARTIAL)

(2.5V

MAX)

Figure E2. Precision Resistor Ratio Set Divides DAC Output,

Permitting Higher Inherent Linearity LTC2410 Utilization.

Disadvantage Is 5ppm/Yr and 0.1ppm/°C Additional Drift Terms

Figure E3. Similar to Figure E5, Except 3:1 Ratio Set Permits

10V Output While Accomodating LTC2410’s 2.5V Input

Application Note 86

AN86-38

APPENDIX F

OUTPUT STAGES

Some applications may require outputs other than the text

circuit’s 0V to 5V range. The simplest variation is a bipolar

output, shown in Figure F1. The circuit, a summing in-

verter, subtracts the DAC output from a reference to obtain

a bipolar output. Resistor and reference values may be

varied to obtain different output excursions. The LT1010

output buffer provides drive capability and the chopper

stabilized amplifier maintains 0.05µV/°C stability. The

resistors introduce a 0.3ppm/°C error contribution

Figure F2 yields voltage gain by dividing the DAC output

prior to its application to the feedback A-to-D. In this case,

the 1:1 divider ratio sets a 10V output, assuming an A-to-D

reference of 5V. As in Figure F1, the resistors add a slight

temperature error, about 0.1ppm/°C for the ratio set

specified.

Figure F3 uses active devices for voltage outputs as high

as ±100V. The discrete high voltage stage is driven in

closed-loop fashion by a chopper stabilized amplifier. Q1

and Q2 furnish voltage gain, and feed the Q3-Q4 emitter

follower outputs. Q5 and Q6 set current limit at 25mA by

diverting output drive when voltages across the 27Ω

shunts become too high. The local 1M-50k feedback pairs

set stage gain at 20, allowing LTC1152 drives to cause full

Note 1: See Note 1 in Appendix E.

Note 2: See above footnote.

±120V output swing. The local feedback reduces stage

gain-bandwidth, making dynamic control easier. This stage

is relatively simple to frequency compensate because only

Q1 and Q2 contribute voltage gain. Additionally, the high

voltage transistors have large junctions, resulting in low

s, and no special high frequency roll-off precautions are

needed. Because the stage inverts, feedback is returned to

the amplifier’s positive input. Frequency compensation is

achieved by rolling off the amplifier with the local 0.005µF-

10k pair.

Heating and voltage coefficient errors are minimized in the

feedback term by using four individual resistors. Trim-

ming involves selecting the indicated resistor for exactly

100.0000V output with the DAC at full scale.

Figure F4 increases output current capability with a cur-

rent gain stage inside the DAC output amplifier’s feedback

loop. This stage replaces the LT1010 150mA buffer shown

in the text. The figure shows two options, differing in

output capacity. It is worth noting that as output current

rises, wiring resistance becomes a large potential error

term. For example, at only 10mA output, 0.001Ω of wiring

resistance introduces 10µV drop—a 2ppm error. Because

of this, heavy loads should be supplied via short, highly

conductive paths and remote sensing employed.

AN86 FF01

–

LTC1152

–5V

REFERENCE

0V TO 5V

FROM DAC

OUTPUT

AMPLIFIER

20k*

*= VISHAY TYPE VHP-100 MATCHED SET

10k*

LT1010 OUTPUT

–2.5V TO 2.5V

10k*

FROM

DAC OUTPUT

AMPLIFIER

TO LTC2400/LTC2410

A-TO-D INPUT

OUTPUT

0V TO 10V

AN86 F02

*VISHAY VHD-200

RATIO SET

0.1ppm/°C

USE 7.5k-2.5k VALUES

FOR LTC2410

Figure F1. Precision Resistors and Chopper Stabilized

Output Amplifier Allow Bipolar DAC Output. Trade-Off Is

≈0.3ppm/°C Additional Resistor Based Error

Figure F2. ×2 Voltage Gain Obtained By

Feedback Division at A-to-D. Slight Increase in

Overall Temperature Coefficient Results

Application Note 86

AN86-39

–

LTC1152

15V

–15V

0.005µF

10k

19.994k**

SELECT

***

TYPICAL

6Ω

INPUT FROM DAC

27Ω

OUTPUT

AN86 FF03

27Ω

330Ω510Ω

–125V

1µF

1M50k

330Ω510Ω

125V

1N4148

2N2222

2N2907

2N3440*

2N5415

100k**100k**100k**100k**

2N3440

1µF

***

HEAT SINK

VISHAY VHP-100 RESISTOR 0.01%

1% METAL FILM RESISTOR

2N5415*

AN86 FF04

0.1µF

(OPTIONAL—

FOR INCREASED

LOAD STABILITY)

FROM TEXT FIGURE 2

(PARTIAL)

FROM

MSB

AND LSB

DACs

–

LT1001

LT1206: 250mA OUTPUT

LT1210: 1.1A OUTPUT

Figure F3. High Voltage Output Stage Delivers ±100V at 25mA. Multiple

Feedback Resistors Minimize Dissipation and Voltage Coefficient Effects

Figure F4. LT1206/LT1210 Output Stages Supply 250mA and 1.1A Loads,

Respectively. Remote Sensing Is Usually Necessary to Compensate IR Drops

Application Note 86

AN86-40

APPENDIX G

MEASURING DAC SETTLING TIME

Measuring the 20-bit DAC’s output settling time is a

challenging task. Although the time scale involved is

relatively slow, the 5µV LSB step size presents problems.

The issue reduces to obtaining a great deal of gain without

inducing overdrive in the monitoring oscilloscope. Such

overdrive will corrupt the measurement, rendering dis-

played results meaningless.

Figure G1 is a solution. The DAC output is resistively

balanced against a precision variable reference supply,

adjustable in 0.5µV steps.

The circuit’s remainder consti-

tutes a clamped, distributed gain of 2000 amplifier. Diode

clamping, placed at each gain stage input, prevents satu-

ration from occurring even with large DAC-reference sup-

ply imbalances. The distributed gain allows 10kHz

bandwidth while maintaining clamping effectiveness. The

monitoring oscilloscope, operating at 5mV or 10mV/DIV

(5µV to 10µV at the DAC output), can readily discern 5µV

settling without incurring deleterious overdrive.

Note 1: See Appendix C for details on such a supply.

Layout and construction of this circuit requires care.

Figure G2 shows construction details. A linear layout

minimizes parasitic feedback paths, preventing oscilla-

tion. The DAC input step is fully shielded, preventing

feedthrough to various sensitive points within the ampli-

fier. Finally, the entire circuit is built into a shielded

enclosure to minimize effects of stray RF and pick up.

The circuit is tested by applying a test step that settles

much faster than the DAC. Figure G3 uses a mercury

wetted reed relay based pulse generator to supply the step.

The unit noted is commercially produced, although similar

results are obtainable with standard mercury based reed

relays. When the relay opens the circuit’s output settles

essentially instantaneously (Figure G4) relative to DAC

speed and settling time amplifier bandwidth.

Figure G1’s response is tested by grounding one of its

inputs and driving the other with the pulse generator.

Figure G5 shows settling to within 1ppm (±5µV) in 2ms.

This is much faster than the DAC settles, lending confi-

dence to text Figures 6 and 7 indicated results.

5.1k OUTPUT

–

LT1008

5.1k

20pF

–

LT1008

5.1k

AN86 FG01

5pF

–

LT1008

FROM

DAC

OUTPUT

0V TO 5V

0V TO –5V

IN 0.5µV STEPS

FROM BUFFERED

KVD BIASED FROM

–5V REFERENCE

20k*

VISHAY VHD-200 RATIO SET 10ppm MATCHING

1N4148

1N5712

CONNECT OUTPUT DIRECTLY TO

OSCILLOSCOPE. DO NOT USE CABLE

TO SET ZERO, GROUND BOTH INPUTS

AND ADJUST “ZERO” FOR V

OUT

= <1mV

20k*

10k

ZERO

–15V15V

10Ω909k

5.1k

5pF

–

LT1008

5pF

5.1k

Figure G1. Clamped, Distributed Gain-of-2000 Amplifier Permits

DAC Settling Time Measurement Without Saturation Effects

Application Note 86

AN86-41

Figure G2. Settling Time Amplifier Construction. Bandwidth Is Only 10kHz, Although Gain of 2000 Necessitates Layout Care to Avoid

Parasitic Feedback Induced Oscillation. Input (Photo Lower Left) Is Fully Shielded, Preventing Radiative Feedthrough to Amplifier.

Enclosure Shields Circuit from Stray RF and Pickup

AN86 FG02.tif

Application Note 86

AN86-42

Figure G3. Reed Relay Based Pulser Supplies Clean Step to Test Settling Time Circuit

OUTPUT

FROM

PULSE

GENERATOR

AN86 FG03

50Ω

Hg REED RELAY

TO DAC INPUT OF

SETTLE CIRCUIT

(GROUND SETTLE CIRCUIT

REFERENCE INPUT)

TEKTRONIX

067-0608-00

PULSE

25ms

30ms

Figure G5. Settling Time Circuit Responds to Test Step with 2ms Settling to ±1ppm (±5µV)

HORIZ = 2ms/DIV AN86 FE03.tif

10µV/DIV

Figure G4. Mercury Wetted Reed Relay Opens in 5 Nanoseconds, Settles Quickly to Zero.

500MHz Ring-Off Derives from Source-Termination Impedance Mismatch

AN86 FG03.tif

Application Note 86

AN86-43

APPENDIX H

MICROVOLT LEVEL NOISE MEASUREMENT

Verifying DAC output noise requires a quiet, high gain

amplifier at the oscilloscope. Figure H1 shows one way to

take the measurement. The input preamplifier, operating

at a gain of 1000, supplies a high pass cutoff at 0.1Hz. It

drives the oscilloscope via a 10Hz discrete low pass filter.

The oscilloscope, set to 1mV/DIV, indicates 1µV/DIV

referred to the preamplifier input. Figure H2 indicates DAC

output noise well below an LSB, about 0.9µV. Equipment

limitations set measurement noise floor at 0.2µV.

Figure H3 shows the noise measurement test setup. Note

that the signal levels involved dictate a completely shielded,

coaxial path from breadboard to oscilloscope.

Figure H4 lists some applicable high sensitivity amplifiers

suitable for the noise measurement. Current generation

oscilloscopes rarely have greater than 2mV/DIV sensitiv-

ity, although older instruments offer more capability. The

figure lists representative preamplifiers and oscilloscope

plug-ins suitable for noise measurement. These units

Note 1: In our work we have found Tektronix types 453, 453A, 454,

454A, 547 and 556 excellent choices. Their pristine trace presentation

is ideal for discerning small signals of interest against a noise floor

limited background.

feature wideband, low noise performance. It is particularly

significant that many of these instruments are no longer

produced. This is in keeping with current instrumentation

trends, which emphasize digital signal acquisition as

opposed to analog measurement capability.

The monitoring oscilloscope should have exceptional trace

clarity. In the latter regard high quality analog oscillo-

scopes are unmatched. The exceptionally small spot size

of these instruments is well-suited to low level noise

measurement.

The digitizing uncertainties and raster

scan limitations of DSOs impose display resolution penal-

ties. Many DSO displays will not even register the fine

structure of the noise waveform.

FROM

DAC

OUTPUT

AN86 FH01

10µF

PREAMPLIFIER

A = 1000

HIGH PASS

CUTOFF = 0.1Hz

OSCILLOSCOPE

1mv/DIV = 1µV/DIV

1.6k (10Hz)

Figure H2. Indicated DAC Output Noise in a 0.1Hz to 10Hz

Bandpass Is Below 1µV, About 0.2LSB. Equipment

Limitations Set Measurement Noise Floor at 0.2µV

Figure H1. Microvolt Noise Measurement Necessitates High

Gain Preamplifier for Oscilloscope. Preamplifier and Discrete

Filter Set 0.1Hz to 10Hz Measurement Bandpass

500nV/DIV

2s/DIV AN86 F04.tif

Application Note 86

AN86-44

Figure H3. Noise Measurement Test Setup Includes Shielded DAC Breadboard (Foreground), Preamplifier

(Left) and Low Pass Filter Attached to Oscilloscope (Center). Measurement Path Is Fully Coaxial

AN86 FH03.tif

Application Note 86

AN86-45

APPENDIX I

VOLTAGE REFERENCES

Figure I1 lists some voltage reference options for use with

the DAC. The self-contained types are convenient and

easily applied. The LM199A and the LTZ1000A require

external circuitry but offer higher performance. All choices

must be trimmed to establish absolute DAC accuracy. The

LTZ1000A offers the highest stability and is discussed

below.

Figure I2 shows the LTZ1000A and its support circuitry.

A1 senses LTZ1000A die temperature and accordingly

controls the IC heater via the 2N3904. A2 controls refer-

ence current. The Zener reference is sensed via Kelvin

connections, minimizing voltage drop effects. A single

point ground eliminates return current mixing and the

attendant errors that would be produced.

Figure I3 offers choices for reference buffering. All employ

a chopper stabilized amplifier augmented with a buffer

output stage. Buffer error is extremely low, as noted in

Appendix C’s discussion. I3a, a simple unity-gain stage,

transmits the input to the output with low error and

minimal reference loading. I3b takes moderate gain, al-

lowing a 7V reference input to produce (in this case) 10V

at the output. I3c offers two ways to get 5V from the

nominal 7V input. A precision divider lightly loads

the reference in one case; the 5V output is taken at the

LT1010. Reference loading is avoided by placing the

divider at the output (optional case shown) and driving the

A-to-D reference input from the divider output, which is

permissible.

INITIAL TEMPERATURE LONG-TERM

TYPE VOLTAGE ACCURACY DRIFT STABILITY COMMENTS

LTZ1000A 7.2V Minimum 7V 0.05ppm/°C 4ppm/Yr Typical Highest Stability Zener Available. Requires

Maximum 7.5V External Heater Control and Reference Buffer Circuitry

LM199A 6.95V 2% 0.5ppm/°C 10ppm/Yr Typical Self-Contained, Including Heater Control Circuitry.

Zener Output Is Unbuffered

LT1021 5V, 7V, 10V 0.05V (7V) 2ppm/°C (7V) 20ppm/kHr Noncumulative Fully Self-Contained. Trimmable

LT1027 5V 0.02% 2ppm/°C 20ppm/kHr Noncumulative Fully Self-Contained. Trimmable

LT1236 5V, 10V 0.05% 5ppm/°C 20ppm/kHr Noncumulative Fully Self-Contained. Trimmable

Figure I1. Reference Choices Compared for Output Voltage, Accuracy

and Stability. Highest Stability Types Require External Circuitry

INSTRUMENT MODEL MAXIMUM

TYPE MANUFACTURER NUMBER BANDWIDTH SENSITIVITY/GAIN AVAILABILITY COMMENTS

Differential Amplifier Tektronix 1A7/1A7A 500kHz/1MHz 10µV/DIV Secondary Market Requires 500 Series Mainframe,

Settable Bandstops

Differential Amplifier Tektronix 7A22 1MHz 10µV/DIV Secondary Market Requires 7000 Series Mainframe,

Settable Bandstops

Differential Amplifier Tektronix 5A22 1MHz 10µV/DIV Secondary Market Requires 5000 Series Mainframe,

Settable Bandstops

Differential Amplifier Tektronix ADA-400A 1MHz 10µV/DIV Current Production Standalone with Optional Power

Supply, Settable Bandstops

Differential Amplifier Tektronix AM-502 1MHz 100,000 Secondary Market Standalone with Optional Power

Supply, Settable Bandstops

Differential Amplifier Preamble 1822 10MHz Gain = 1000 Current Production Standalone, Settable Bandstops

Differential Amplifier Stanford Research SR-560 1MHz Gain = 50000 Current Production Standalone, Settable Bandstops,

Systems Battery or Line Operation

Figure H4. Some Applicable High Sensitivity, Low Noise Amplifiers.

Trade-Offs Include Compatibility, Sensitivity and Availability

Application Note 86

AN86-46

Figure I2. 7V Reference Includes A1 Heater Control Amplifier, A2 Zener Current Regulator

and LTZ1000A Zener. Note Zener Kelvin Connections and Single Point Ground

Figure I3. Chopper Stabilized Reference Buffer Options Include Unity Gain (a),

10V (b) and 5V (c) Output. Trimming Is Required for Absolute Accuracy

–

AN86 FI02

LT1013

1N4148

SINGLE

POINT

GROUND

15V

ZENER + SENSE

ZENER – SENSE

ZENER – FORCE

HEATER RETURN

R1 THROUGH R5:

VISHAY VHP-100 0.1%

HEATER,

UNLABELED

AND

ZENER = LTZ1000A

PIN NUMBERS APPEAR

FOR INDIVIDUAL

LTZ1000A ELEMENTS

HEATER

1N4148

–

LT1013

70k

0.022µF

120Ω

70k

TEMP

SENSOR

13k

10k

2N3904

0.1µF

–

LTC1150

0.1µF

10k

OUT = 5V NOMINAL

22k*

TO ZENER

SENSE

50k*

LT1010

OUT = 5V NOMINAL

OPTIONAL, DELETE INPUT DIVIDER.

SEE TEXT

5k**

2.2k**

–

LTC1150

0.1µF

5k**

OUT = 10V NOMINAL

TO ZENER

SENSE

LT1010

12.9k**

–

LTC1150

0.1µF

10k

OUT = 7.2V NOMINAL

TO ZENER

SENSE

LT1010

VISHAY VHP-100 – 1%

VISHAY VHD-200 RATIO SET – 0.1%

AN86 I03

Application Note 86

AN86-47

APPENDIX J

CABLES, CONNECTIONS, SOLDER, COMPONENT

CHOICE, TERROR AND ARCANA

Subtle parasitic effects can have pronounced and seem-

ingly inexplicable effects on low level circuit performance.

Perhaps the most prevalent detractor to microvolt level

circuitry is unintended thermocouples. Considerable dis-

cussion for dealing with thermocouples appeared in Ap-

pendix C and should be considered preliminary to this

section’s material.

In 1822, Thomas Seebeck, an Estonian physician, acci-

dentally joined semicircular pieces of bismuth and copper

(Figure J1) while studying thermal effects on galvanic

arrangements. A nearby compass indicated a magnetic

disturbance. Seebeck experimented repeatedly with dif-

ferent metal combinations at various temperatures, noting

relative magnetic field strengths. Curiously, he did not

believe that electric current was flowing and preferred to

describe the effect as “thermomagnetism.” He published

his results in a paper, “Magnetische Polarisation der

Metalle und Erze durch Temperatur-Differenz” (see Refer-

ences).

In low drift circuits, unwanted thermocouples are prob-

ably the primary source of error. Connectors, switches,

relay contacts, sockets, wire and even solder are all

candidates for thermal EMF generation. It is relatively clear

that connectors and sockets can form thermal junctions.

However, it is not at all obvious that junctions of wire from

different manufacturers can easily generate 200nV/°C—

four times a precision amplifier’s drift specification! Fig-

ure J2 shows a plot obtained for such a wire junction. Even

solder can become an error term at low levels, creating a

junction with copper or Kovar wires or PC traces (see

Figure J3). Figure J4 lists thermocouple potentials for

some common materials found in electronic assemblies.

JUNCTION

COMPASS

BISMUTH

AN86 FJ01

COPPER

Figure J1. The Arrangement for Dr. Seebeck’s

Accidental Discovery of “Thermomagnetism”

Subsequent investigation has shown the “Seebeck Effect”

to be fundamentally electrical in nature, repeatable and

quite useful. Thermocouples, by far the most common

transducer, are Seebeck’s descendants. Unfortunately,

unintended and unwanted thermocouples are also

Seebeck’s progeny.

DEGREES CENTIGRADE

25 30 35 40 45

MICROVOLTS REFERRED TO 25°C

AN86 FJ02

3.0

2.8

2.6

2.4

2.2

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

Figure J2. Thermal EMF Generated by Two “Identical”

Copper Wires Due to Oxidation and Impurities

SOLDER-COPPER JUNCTION DIFFERENTIAL TERMPERATURE

SOURCE: NEW ELECTRONICS 2/6/77

010 20 30 40 50

THERMALLY PRODUCED VOLTAGE

IN MICROVOLTS

AN86 FJ03

100

–50

–100

SLOPE ≈ 1.5µV/°C BELOW 25°C

SLOPE ≈ 160nV/°C BELOW 25°C

64% Sn/36% Pb

60% Cd/40% Sn

Figure J3. Solder-Copper Thermal EMFs. Cd/Sn Has Notably

Lower Activity but Is Toxic, Not Available and Not Recommended

Application Note 86

AN86-48

The unusually energetic response of Cu-CuO necessitated

the treatment described in Appendix C (Figure C7 and

associated text) for cleaning DVM and Kelvin-Varley di-

vider connections. Readers finding this figure’s informa-

tion seemingly academic should be awakened by Figure

J5. This chart lists thermoelectric potentials for com-

monly employed laboratory connectors. Thermocouple

activity of some types is more than 20 times greater than

others. Be careful!

Minimizing thermal EMF induced errors is possible if

judicious attention is given to circuit board layout. In

general, it is good practice to limit the number of junctions

in the signal path. Avoid connectors, sockets, switches

and other potential error sources to the extent possible. In

some cases this will not be possible. In these instances,

attempt to balance the number and type of junctions in the

signal path so that differential cancellation occurs. Doing

this may involve deliberately creating and introducing

junctions to offset unavoidable junctions. This can be a

tricky procedure. Repeated deliberate temperature excur-

sions may be necessary to determine the optimal number

and placement of added junctions. Experimentation, tem-

pered by a healthy reserve of patience and abundance of

time, is required. This practice, borrowed from standards

lab procedures, can be quite effective in reducing thermal

EMF originated drifts. Figure J6 shows a simple example

where a nominally unnecessary resistor is included to

promote such thermal balancing. For remote signal sources

connectors may be unavoidable. In these cases, choose a

connector specified for relatively low thermal EMF activity

and ensure a similarly balanced approach in routing sig-

nals through the connector along the circuit board and to

circuitry. If some imbalance is unavoidable, deliberately

introduce an intentional counterbalancing junction. In all

cases maintain the differencing junctions in close physical

proximity, which will keep them at the same temperature.

Avoid drafts and temperature gradients, which can intro-

duce thermal imbalances and cause problems. Figure J7

shows the LTC1150 set up in a test circuit to measure its

temperature stability. The lead lengths of the resistors

connected to the amplifier’s inputs are identical. The

thermal capacity each input sees is also balanced because

of the symmetrical connection of the resistors and their

identical size. Thus, thermal EMF induced shifts are equal

in phase and amplitude and cancellation occurs. Very

slight air currents can still affect even this arrangement.

Figure J8 shows a strip chart of output noise with the

circuit covered by a small styrofoam cup (HANDI-KUP

Company Model H8-S) and with no cover in “still” air. This

data illustrates why it is often prudent to enclose low level

circuitry inside some form of thermal baffle.

Figure J4. Thermoelectric Potentials for Various Materials

Indicates Inadvisability of Mixing Materials in Signal Path.

Cu-Cu Connections (Chart Top) Must Be Kept Clean or

5000:1 Degradation Occurs As They Oxidize (Chart Bottom)

MATERIALS POTENTIAL (µV/°C)

Cu-Cu < 0.2

Cu-Ag 0.3

Cu-Au 0.3

Cu-Cd/Sn 0.3

Cu-Pb/Sn 1 to 3

Cu-Kovar 40

Cu-Si 400

Cu-CuO 1000

Source: Low Level Measurements,

Keithley Instruments, 1984

(see References)

HANDI-KUP is a trademark of WinCup.

Application Note 86

AN86-49

THERMOELECTRIC

CONNECTION TYPE DESCRIPTION POTENTIAL (µV/°C)

BNC-BNC Mate 0.4

BNC-Banana Adapter 0.35

BNC-BNC “Barrel” Adapter 0.4

Male/Female Banana Mate Sample #1 0.35

Male/Female Banana Mate Sample #2 1.1

Male/Female Banana Mate (Type Specified 0.07

for Low Thermal Activity) Sample #3

Copper Lug-Copper Banana Binding Post 0.08

Copper Lug-Standard Banana Binding Post 0.5

Plated Lug-Copper Banana Binding Post 1.7

Figure J5. Measured Thermoelectric Potentials for Some Common Laboratory Connectors.

Pronounced Difference Between “Banana” Samples Is Due to Manufacturer’s Materials Choice. Note

That Copper Lug/Copper Banana Post Has 20× Lower Activity Than Plated Lug/Copper Banana Post

Application Note 86

AN86-50

–

LEAD WIRE/SOLDER/COPPER

TRACE JUNCTION

OUTPUT

AN86 FJ06

DELIBERATE SPLICE MAY BE DESIRABLE

TO BALANCE OTHER JUNCTIONS

NOMINALLY UNNECESSARY

RESISTOR USED TO

THERMALLY BALANCE OTHER

INPUT RESISTOR

RESISTOR LEAD, SOLDER,

COPPER TRACE JUNCTION

Figure J6. Typical Thermal Layout Considerations Emphasize Minimizing and Differencing

Parasitic Thermocouples. Thermal Mass at Amplifier Inputs Should Be Equal, Allowing

Differenced Parasitic Thermocouple Outputs to Arrive Matched In Phase and Amplitude

–

50k

100Ω

AN86 FJ06

50k

LTC1150

× 1000

Figure J7. Amplifier Drift Test Circuit. Thermal EMFs and Thermal

Capacity at Each Input Must Be Similar for Cancellation to Occur

Thermal EMFs are the most likely, but not the only,

potential low level error source. Electrostatic and electro-

magnetic shielding may be required. Power supply trans-

former fields are notorious sources of errors often

mistakenly attributed to amplifier DC drift and noise. A

transformer’s magnetic field impinging on a PC trace can

easily generate microvolts across that conductor in accor-

dance with well known magnetic theory. The circuit cannot

distinguish between this spurious signal and the desired

input. Attempts to eliminate the problem by rolling off

circuit response may work, but often the filtered version of

the undesired pickup masquerades as an unstable DC

term. The most direct approach is to use shielded trans-

formers but careful layout may be equally effective and

less costly. A circuit that requires the transformer to be

close by to achieve a good quality grounding scheme may

be disturbed by the transformer’s magnetic field. An RF

choke connected across a scope probe can determine the

presence and relative intensity of transformer fields, aid-

ing layout experimentation.

Another source of parasitic error is stray leakage current.

Such leakage currents must be prevented from influenc-

ing circuit operation. The simplest way to do this is to

connect leakage sensitive points via teflon standoffs.

Because the points never contact the PC board, stray

leakage currents do not affect them. Although this ap-

proach is effective, its implementation may not be accept-

able in production. Guarding is another technique for

minimizing board leakage effects. The guard is a PC trace

completely encircling the leakage sensitive points. This

trace is driven at a potential equal to that of the point,

preventing leakage to the “guarded” point. On PC boards,

the guard should enclose the node(s) to be protected.

Guarding was used to eliminate the effects of capacitor

surface leakage in Appendix C’s Figure C7.

Application Note 86

AN86-51

Figure J8. Effect of Thermal Baffle on Low Frequency Amplifier Noise in “Still” Air. Amplifier Is Covered

By Small Cup in Upper Trace, Uncovered in Lower Trace. Instability Worsens If Air Movement Increases

AN86 FJ08.tif

Application Note 86

AN86-52

an86f LT/TP 0101 4K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 2001

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

●

FAX: (408) 434-0507

●