Low Power Programmable
Temperature Controller
TMP01
Rev. E
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FEATURES
−55°C to +125°C (−67°F to +257°F) operation
±1.0°C accuracy over temperature (typ)
Temperature-proportional voltage output
User-programmable temperature trip points
User-programmable hysteresis
20 mA open-collector trip point outputs
TTL/CMOS compatible
Single-supply operation (4.5 V to 13.2 V)
PDIP, SOIC, and TO-99 packages
APPLICATIONS
Over/under temperature sensor and alarm
Board-level temperature sensing
Temperature controllers
Electronic thermostats
Thermal protection
HVAC systems
Industrial process control
Remote sensors
FUNCTIONAL BLOCK DIAGRAM
VPTAT
V+
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
2.5V SENSOR
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
VREF
SET
HIGH
SET
LOW
GND
UNDER
OVER
R1
R2
R3
0
0333-001
Figure 1.
GENERAL DESCRIPTION
The TMP01 is a temperature sensor that generates a voltage
output proportional to absolute temperature and a control
signal from one of two outputs when the device is either above
or below a specific temperature range. Both the high/low
temperature trip points and hysteresis (overshoot) band are
determined by user-selected external resistors. For high volume
production, these resistors are available on board.
The TMP01 consists of a band gap voltage reference combined
with a pair of matched comparators. The reference provides
both a constant 2.5 V output and a voltage proportional to
absolute temperature (VPTAT) which has a precise temperature
coefficient of 5 mV/K and is 1.49 V (nominal) at 25°C. The
comparators compare VPTAT with the externally set tempera-
ture trip points and generate an open-collector output signal
when one of their respective thresholds has been exceeded.
Hysteresis is also programmed by the external resistor chain
and is determined by the total current drawn out of the 2.5 V
reference. This current is mirrored and used to generate a
hysteresis offset voltage of the appropriate polarity after a
comparator has been tripped. The comparators are connected
in parallel, which guarantees that there is no hysteresis overlap
and eliminates erratic transitions between adjacent trip zones.
The TMP01 utilizes proprietary thin-film resistors in conjunc-
tion with production laser trimming to maintain a temperature
accuracy of ±1°C (typical) over the rated temperature range,
with excellent linearity. The open-collector outputs are capable
of sinking 20 mA, enabling the TMP01 to drive control relays
directly. Operating from a 5 V supply, quiescent current is only
500 A (max).
The TMP01 is available in 8-pin mini PDIP, SOIC, and TO-99
packages.
TMP01
Rev. E | Page 2 of 20
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications ....................................................................................... 1
Functional Block Diagram .............................................................. 1
General Description ......................................................................... 1
Revision History ............................................................................... 2
Specifications ..................................................................................... 3
TMP01EST, TMP01FP, TMP01FS ............................................. 3
TMP01FJ ........................................................................................ 4
Absolute Maximum Ratings ............................................................ 5
Typical Performance Characteristics ............................................. 6
Theory of Operation ........................................................................ 8
Temperature Hysteresis ............................................................... 8
Programming the TMP01 ........................................................... 8
Understanding Error Sources ..................................................... 9
Safety Considerations in Heating and Cooling System
Design ............................................................................................ 9
Applications Information .............................................................. 10
Self-Heating Effects .................................................................... 10
Buffering the Voltage Reference ............................................... 10
Preserving Accuracy Over Wide Temperature Range
Operation .................................................................................... 10
Thermal Response Time ........................................................... 10
Switching Loads with the Open-Collector Outputs .............. 11
High Current Switching ............................................................ 12
Buffering the Temperature Output Pin ................................... 13
Differential Transmitter ............................................................. 13
4 mA to 20 mA Current Loop .................................................. 13
Temperature-to-Frequency Converter .................................... 14
Isolation Amplifier ..................................................................... 15
Out-of-Range Warning .............................................................. 15
Translating 5 mV/K to 10 mV/°C ............................................ 16
Translating VPTAT to the Fahrenheit Scale ........................... 16
Outline Dimensions ....................................................................... 17
Ordering Guide .......................................................................... 18
REVISION HISTORY
7/09—Rev. D to Rev. E
Updated Format .................................................................. Universal
Updated Outline Dimensions ....................................................... 18
Changes to Ordering Guide .......................................................... 19
1/02—Rev. C: Rev. D
Edits to General Descriptions Section ........................................... 1
Edits to Specifications Section ........................................................ 2
Edits to Wafer Test Limits Section.................................................. 4
Edits to Dice Characteristics Section ............................................. 4
Edits to Ordering Guide .................................................................. 5
7/93—Revision 0: Initial Version
TMP01
Rev. E | Page 3 of 20
SPECIFICATIONS
TMP01ES, TMP01FP, TMP01FS
PDIP and SOIC packages. V+ = 5 V, GND = O V, −40°C ≤ TA ≤ +85°C, unless otherwise noted.
Table 1.
Parameter Symbol Conditions Min Typ Max Unit
INPUTS SET HIGH, SET LOW
Offset Voltage VOS 0.25 mV
Offset Voltage Drift TCVOS 3 μV/°C
Input Bias Current, E Grade IB 25 50 nA
Input Bias Current, F Grade IB 25 100 nA
OUTPUT VPTAT
Output Voltage VPTAT TA = 25°C, no load 1.49 V
Scale Factor1TCVPTAT 5 mV/K
Temperature Accuracy, E Grade TA = 25°C, no load −1.5 ±0.5 1.5 °C
Temperature Accuracy, F Grade TA = 25°C, no load −3 ±1.0 3 °C
Temperature Accuracy, E Grade 10°C < TA < 40°C, no load ±0.75 °C
Temperature Accuracy, F Grade 10°C < TA < 40°C, no load ±1.5 °C
Temperature Accuracy, E Grade −40°C < TA < 85°C, no load −3.0 ±1 3.0 °C
Temperature Accuracy, F Grade −40°C < TA < 85°C, no load −5.0 ±2 5.0 °C
Temperature Accuracy, E Grade −55°C < TA < 125°C, no load ±1.5 °C
Temperature Accuracy, F Grade ΔVPTAT −55°C < TA < 125°C, no load ±2.5 °C
Repeatability Error2 0.25 Degree
Long-Term Drift Error3, 4
0.25 0.5 Degree
Power Supply Rejection Ratio PSRR TA = 25°C, 4.5 V ≤ V+ ≤ 13.2 V ±0.02 ±0.1 %/V
OUTPUT VREF
Output Voltage, E Grade VREF TA = 25°C, no load 2.495 2.500 2.505 V
Output Voltage, F Grade VREF TA = 25°C, no load 2.490 2.500 2.510 V
Output Voltage, E Grade VREF −40°C < TA < 85°C, no load 2.490 2.500 2.510 V
Output Voltage, F Grade VREF −40°C < TA < 85°C, no load 2.485 2.500 2.515 V
Output Voltage, E Grade VREF −55°C < TA < 125°C, no load 2.5 ± 0.01 V
Output Voltage, F Grade VREF −55°C < TA < 125°C, no load 2.5 ± 0.015 V
Drift TCVREF −10 ppm/°C
Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.01 ±0.05 %/V
Load Regulation 10 μA ≤ IVREF ≤ 500 μA ±0.1 ±0.25 %/mA
Output Current, Zero Hysteresis IVREF 7 μA
Hysteresis Current Scale Factor1
SFHYS 5.0 μA/°C
Turn-On Settling Time To rated accuracy 25 μs
OPEN-COLLECTOR OUTPUTS OVER, UNDER
Output Low Voltage VOL ISINK = 1.6 mA 0.25 0.4 V
V
OL ISINK = 20 mA 0.6 V
Output Leakage Current IOH V+ = 12 V 1 100 μA
Fall Time tHL See Figure 2 40 ns
POWER SUPPLY
Supply Range V+ 4.5 13.2 V
Supply Current ISY Unloaded, +V = 5 V 400 500 μA
I
SY Unloaded, +V = 13.2 V 450 800 μA
Power Dissipation PDISS +V = 5 V 2.0 2.5 mW
1 K = °C + 273.15.
2 Maximum deviation between 25°C readings after temperature cycling between −55°C and +125°C.
3 Guaranteed but not tested.
4 Observed in a group sample over an accelerated life test of 500 hours at 150°C.
TMP01
Rev. E | Page 4 of 20
V
+
1kΩ
20pF
00333-002
Figure 2. Test Load
TMP01FJ
TO-99 metal can package. V+ = 5 V, GND = 0 V, −40°C ≤ TA ≤ +85°C, unless otherwise noted.
Table 2.
Parameter Symbol Conditions Min Typ Max Unit
INPUTS SET HIGH, SET LOW
Offset Voltage VOS 0.25 mV
Offset Voltage Drift TCVOS 3 μV/°C
Input Bias Current, F Grade IB 25 100 nA
OUTPUT VPTAT
Output Voltage VPTAT TA = 25°C, no load 1.49 V
Scale Factor1 TCVPTAT 5 mV/K
Temperature Accuracy, F Grade TA = 25°C, no load −3 ±1.0 3 °C
Temperature Accuracy, F Grade 10°C < TA < 40°C, no load ±1.5 °C
Temperature Accuracy, F Grade −40°C < TA < 85°C, no load −5.0 ±2 5.0 °C
Temperature Accuracy, F Grade ΔVPTAT −55°C < TA < 125°C, no load ±2.5 °C
Repeatability Error2 0.25 Degree
Long-Term Drift Error3, 4 0.25 0.5 Degree
Power Supply Rejection Ratio PSRR TA = 25°C, 4.5 V ≤ V+ ≤ 13.2 V ±0.02 ±0.1 %/V
OUTPUT VREF
Output Voltage, F Grade VREF TA = 25°C, no load 2.490 2.500 2.510 V
Output Voltage, F Grade VREF −40°C < TA < 85°C, no load 2.485 2.500 2.515 V
Output Voltage, F Grade VREF −55°C < TA < 125°C, no load 2.5 ± 0.015 V
Drift TCVREF −10 ppm/°C
Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.01 ±0.05 %/V
Load Regulation 10 μA ≤ IVREF ≤ 500 μA ±0.1 ±0.25 %/mA
Output Current, Zero Hysteresis IVREF 7 μA
Hysteresis Current Scale Factor1 SFHYS 5.0 μA/°C
Turn-On Settling Time To rated accuracy 25 μs
OPEN-COLLECTOR OUTPUTS OVER, UNDER
Output Low Voltage VOL ISINK = 1.6 mA 0.25 0.4 V
V
OL ISINK = 20 mA 0.6 V
Output Leakage Current IOH V+ = 12 V 1 100 μA
Fall Time tHL See Figure 2 40 ns
POWER SUPPLY
Supply Range V+ 4.5 13.2 V
Supply Current ISY Unloaded, +V = 5 V 400 500 μA
I
SY Unloaded, +V = 13.2 V 450 800 μA
Power Dissipation PDISS +V = 5 V 2.0 2.5 mW
1K = °C + 273.15.
2Maximum deviation between 25°C readings after temperature cycling between −55°C and +125°C.
3Guaranteed but not tested.
4Observed in a group sample over an accelerated life test of 500 hours at 150°C.
TMP01
Rev. E | Page 5 of 20
ABSOLUTE MAXIMUM RATINGS
Table 3.
Parameter Rating
Maximum Supply Voltage −0.3 V to +15 V
Maximum Input Voltage (SET HIGH, SET LOW) −0.3 V to V+ +0.3 V
Maximum Output Current (VREF, VPTAT) 2 mA
Maximum Output Current (Open-Collector
Outputs)
50 mA
Maximum Output Voltage (Open-Collector
Outputs)
15 V
Operating Temperature Range −55°C to +150°C
Die Junction Temperature 150°C
Storage Temperature Range −65°C to +150°C
Lead Temperature (Soldering 60 sec) 300°C
Digital inputs and outputs are protected; however, permanent
damage may occur on unprotected units from high energy
electrostatic fields. Keep units in conductive foam or packaging
at all times until ready to use. Use proper antistatic handling
procedures.
Remove power before inserting or removing units from their
sockets.
Table 4.
Package Type θJA θJC Unit
8-Lead PDIP (N-8) 103143 °C/W
8-Lead SOIC (R-8) 158243 °C/W
8-Pin TO-99 Can (H-08) 1501
18 °C/W
1 θJA is specified for device in socket (worst-case conditions).
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
2 θJA is specified for device mounted on PCB.
ESD CAUTION
TMP01
Rev. E | Page 6 of 20
TYPICAL PERFORMANCE CHARACTERISTICS
20515010
SUPPLY VOLTAGE (V)
SUPPLY CURRENT ( µA)
550
350
400
375
450
425
475
500
525
+25°C
+125°C
+85°C
–55°C
–40°C
00333-003
Figure 3. Supply Current vs. Supply Voltage
5.0
3.0
4.5
3.5
4.0
125–75 –50 –25 1007550250
TEM P ERATURE (°C)
MINIMUM SUPPLY VOLTAGE (V)
00333-004
Figure 4. Minimum Supply Voltage vs. Temperature
2.0
1.0
0.5
–2.0
1.5
–1.0
–1.5
–0.5
0
125–75 –50 –25 10075
V+ = 5V
50250
TEM P ERATURE (°C)
VPTAT ERROR (°C)
00333-005
Figure 5. VPTAT Accuracy vs. Temperature
2.508
2.506
2.504
2.496
2.500
2.498
2.502
125–75 –50 –25 10075
V+ = 5V
50250
TEMPERAT URE (°C)
VREF (V)
00333-006
Figure 6. VREF Accuracy vs. Temperature
6
0
3
1
2
5
4
5010 4003020
V
C
= 15V
V+ = 5V
T
A
= 25°C
I
C
(mA)
V
CE
(V)
00333-007
Figure 7. Open-Collector Output (OVER, UNDER) Saturation Voltage vs.
Output Current
X – 3σ
X + 3σ
2.510
2.490
2.496
2.492
2.494
2.502
2.498
2.500
2.504
2.506
2.508
1000200 8000 400 600
X
CURVES NOT NORMAL IZE D
EXT RAP OLATED F ROM OPERAT ING LIFE DAT A
T = HOURS O F O P E RATION AT 125°C; V + = 5V
VREF (V)
0
0333-008
Figure 8. VREF Long Term Drift Accelerated by Burn-In
TMP01
Rev. E | Page 7 of 20
100 1M1k 100k10k
–20
100
40
20
0
60
80
FREQUENCY (Hz )
PSRR (dB)
V+ = 5V
I
VREF
= 10µ A
00333-009
Figure 9. VREF Power Supply Rejection vs. Frequency
1.0
0.1
0.01
OFFSET VOLTAGE (mV)
V+ = 5V
I
VREF
= 7. 5µA
00333-010
Figure 10. Set High, Set Low Input Offset Voltage vs. Temperature
8
0
2
1
4
3
5
6
7
–0.4 –0.24
–0.32 0–0.16 0.16–0.08 0.08
OFFSET (mV)
NUMBER OF DEVICES
V+ = 5V
T
A
= 25°C
I
VREF
= 5µA
00333-011
Figure 11. Comparator Input Offset Distribution
7.276.2 6.8
6.6 86.4 7.87.67.4
REFERENCE CURRENT ( µ A)
NUMBER O F DEVICES
10
0
2
1
4
3
5
6
7
8
9
V+ = 5V
T
A
= 25°C
00333-012
Figure 12. Zero Hysteresis Current Distribution
TMP01
Rev. E | Page 8 of 20
THEORY OF OPERATION
The TMP01 is a linear voltage-output temperature sensor, with
a window comparator that can be programmed by the user to
activate one of two open-collector outputs when a predeter-
mined temperature setpoint voltage has been exceeded. A low
drift voltage reference is available for setpoint programming.
The temperature sensor is basically a very accurate, temperature
compensated, band gap-type voltage reference with a buffered
output voltage proportional to absolute temperature (VPTAT),
accurately trimmed to a scale factor of 5 mV/K.
The low drift 2.5 V reference output VREF is easily divided
externally with fixed resistors or potentiometers to accurately
establish the programmed heat/cool setpoints, independent of
temperature. Alternatively, the setpoint voltages can be supplied
by other ground referenced voltage sources such as user-
programmed DACs or controllers. The high and low setpoint
voltages are compared to the temperature sensor voltage, thus
creating a two-temperature thermostat function. In addition,
the total output current of the reference (IVREF) determines the
magnitude of the temperature hysteresis band. The open
collector outputs of the comparators can be used to control a
wide variety of devices.
VPTAT
V+
WINDOW
COMPARATOR
TEMPERATURE
OUTPUT
HYSTERESIS
CURRENT
CURRENT
MIRROR
HYSTERESIS
VOLTAGE
ENABLE
TMP01
VREF
SET
HIGH
SET
LOW
GND
UNDER
OVER
8
5
6
7
1
4
3
2
VOLTAGE
REFERENCE
AND
SENSOR
1kΩ
I
HYS
00333-013
Figure 13. Detailed Block Diagram
TEMPERATURE HYSTERESIS
The temperature hysteresis is the number of degrees beyond
the original setpoint temperature that must be sensed by the
TMP01 before the setpoint comparator is reset and the output
disabled. Figure 14 shows the hysteresis profile. The hysteresis
is programmed by the user by setting a specific load on the
reference voltage output VREF. This output current IVREF is also
called the hysteresis current, which is mirrored internally and
fed to a buffer with an analog switch.
HYSTERESIS
HIGH
HYSTERESIS
LOW
LO
HI
OUTPUT
VOLTAGE
OVE R, UNDER
TEMPERATURE
HYSTERESIS HIGH =
HYSTERES I S LOW
T
SETLOW
T
SETHIGH
00333-014
Figure 14. TMP01 Hysteresis Profile
After a temperature setpoint is exceeded and a comparator
tripped, the buffer output is enabled. The output is a current
of the appropriate polarity that generates a hysteresis offset volt-
age across an internal 1000  resistor at the comparator input.
The comparator output remains on until the voltage at the
comparator input, now equal to the temperature sensor voltage
VPTAT summed with the hysteresis offset, returns to the
programmed setpoint voltage. The comparator then returns
low, deactivating the open-collector output and disabling the
hysteresis current buffer output. The scale factor for the
programmed hysteresis current is:
IHYS = IVREF = 5 µA/°C + 7 µA
Thus, since VREF = 2.5 V, with a reference load resistance
of 357 k or greater (output current 7 A or less), the temper-
ature setpoint hysteresis is zero degrees. Larger values of load
resistance only decrease the output current below 7 A and
have no effect on the operation of the device. The amount of
hysteresis is determined by selecting a value of load resistance
for VREF.
PROGRAMMING THE TMP01
In the basic fixed setpoint application utilizing a simple resistor
ladder voltage divider, the desired temperature setpoints are
programmed in the following sequence:
1. Select the desired hysteresis temperature.
2. Calculate the hysteresis current IVREF.
3. Select the desired setpoint temperatures.
4. Calculate the individual resistor divider ladder values
needed to develop the desired comparator setpoint voltages
at SET HIGH and SET LOW.
TMP01
Rev. E | Page 9 of 20
The hysteresis current is readily calculated. For example, for
2 degrees of hysteresis, IVREF = 17 A. Next, the setpoint
voltages, VSETHIGH and VSETLOW, are determined using the VPTAT
scale factor of 5 mV/K = 5 mV/(°C + 273.15), which is 1.49 V
for 25°C. Then, calculate the divider resistors, based on those
setpoints. The equations used to calculate the resistors are
VSETHIGH = (TSETHIGH + 273.15) (5 mV/°C)
VSETLOW = (TSETLOW + 273.15) (5 mV/°C)
R1 (kΩ) = (VVREF − VSETHIGH)/IVREF = (2.5 V − VSETHIGH)/IVREF
R2 (kΩ) = (VSETHIGH − VSETLOW)/IVREF
R3 (kΩ) = VSETLOW/IVREF
VPTAT
V+
1
2
3
4
8
7
6
5
TMP01
V
VREF
= 2. 5V
V
SETHIGH
V
SETLOW
GND
UNDER
OVER
(V
VREF
– V
SETHIGH
)/I
VREF
= R1
(V
SETHIGH
– V
SETLOW
)/I
VREF
= R2
V
SETLOW
/I
VREF
= R3
I
VREF
00333-015
Figure 15. TMP01 Setpoint Programming
The total R1 + R2 + R3 is equal to the load resistance needed to
draw the desired hysteresis current from the reference, or IVREF.
The formulas shown above are also helpful in understanding
the calculation of temperature setpoint voltages in circuits other
than the standard two-temperature thermostat. If a setpoint
function is not needed, the appropriate comparator should be
disabled. SET HIGH can be disabled by tying it to V+, SET
LOW by tying it to GND. Either output can be left
unconnected.
V
PTAT
K
°C
°F
1.09 1.24 1.991.8651.741.6151.491.365
218 248 398373348323298273
–67 –25 257200 21215010050 77320
–55 –25 1251007550250–18
0
0333-016
Figure 16. Temperature—VPTAT Scale
UNDERSTANDING ERROR SOURCES
The accuracy of the VPTAT sensor output is well characterized
and specified; however, preserving this accuracy in a heating or
cooling control system requires some attention to minimizing
the various potential error sources. The internal sources of
setpoint programming error include the initial tolerances and
temperature drifts of the reference voltage VREF, the setpoint
comparator input offset voltage and bias current, and the
hysteresis current scale factor. When evaluating setpoint
programming errors, remember that any VREF error
contribution at the comparator inputs is reduced by the
resistor divider ratios. The comparator input bias current
(inputs SET HIGH, SET LOW) drops to less than 1 nA (typ)
when the comparator is tripped. This can account for some
setpoint voltage error, equal to the change in bias current times
the effective setpoint divider ladder resistance to ground.
The thermal mass of the TMP01 package and the degree of
thermal coupling to the surrounding circuitry are the largest
factors in determining the rate of thermal settling, which
ultimately determines the rate at which the desired temperature
measurement accuracy may be reached. Thus, allow sufficient
time for the device to reach the final temperature. The typical
thermal time constant for the plastic package is approximately
140 seconds in still air. Therefore, to reach the final temperature
accuracy within 1%, for a temperature change of 60 degrees, a
settling time of 5 time constants, or 12 minutes, is necessary.
The setpoint comparator input offset voltage and zero hyster-
esis current affect setpoint error. While the 7 A zero hysteresis
current allows the user to program the TMP01 with moderate
resistor divider values, it does vary somewhat from device to
device, causing slight variations in the actual hysteresis obtained
in practice. Comparator input offset directly impacts the pro-
grammed setpoint voltage and thus the resulting hysteresis
band, and must be included in error calculations.
External error sources to consider are the accuracy of the pro-
gramming resistors, grounding error voltages, and the overall
problem of thermal gradients. The accuracy of the external
programming resistors directly impacts the resulting setpoint
accuracy. Thus, in fixed-temperature applications, the user
should select resistor tolerances appropriate to the desired
programming accuracy. Resistor temperature drift must be
taken into account also. This effect can be minimized by
selecting good quality components, and by keeping all com-
ponents in close thermal proximity. Applications requiring high
measurement accuracy require great attention to detail
regarding thermal gradients. Careful circuit board layout,
component placement, and protection from stray air currents
are necessary to minimize common thermal error sources.
Also, the user should take care to keep the bottom of the set-
point programming divider ladder as close to GND (Pin 4) as
possible to minimize errors due to IR voltage drops and coup-
ling of external noise sources. In any case, a 0.1 F capacitor for
power supply bypassing is always recommended at the chip.
SAFETY CONSIDERATIONS IN HEATING AND
COOLING SYSTEM DESIGN
Designers should anticipate potential system fault conditions,
which may result in significant safety hazards, which are outside
the control of and cannot be corrected by the TMP01-based
circuit. Observe governmental and industrial regulations
regarding safety requirements and standards for such designs
where applicable.
TMP01
Rev. E | Page 10 of 20
APPLICATIONS INFORMATION
SELF-HEATING EFFECTS
In some applications, the user should consider the effects of
self-heating due to the power dissipated by the open-collector
outputs, which are capable of sinking 20 mA continuously.
Under full load, the TMP01 open-collector output device is
dissipating
PDISS = 0.6 V × .020A = 12 mW
which in a surface-mount SOIC package accounts for a
temperature increase due to self-heating of
∆T = PDISS × θJA = .012 W × 158°C/W = 1.9°C
This self-heating effect directly affects the accuracy of the
TMP01 and will, for example, cause the device to activate
the OVER output 2 degrees early.
Bonding the package to a moderate heat sink limits the self-
heating effect to approximately:
∆T = PDISS × θJC = .012 W × 43°C/W = 0.52°C
which is a much more tolerable error in most systems. The
VREF and VPTAT outputs are also capable of delivering
sufficient current to contribute heating effects and should not
be ignored.
BUFFERING THE VOLTAGE REFERENCE
The reference output VREF is used to generate the temper-
ature setpoint programming voltages for the TMP01 and also
to determine the hysteresis temperature band by the reference
load current IVREF. The on-board output buffer amplifier is
typically capable of 500 A output drive into as much as 50 pF
load (maximum). Exceeding this load affects the accuracy
of the reference voltage, could cause thermal sensing errors
due to dissipation, and may induce oscillations. Selection of
a low drift buffer functioning as a voltage follower with high
input impedance ensures optimal reference accuracy, and
does not affect the programmed hysteresis current. Amplifiers
which offer the low drift, low power consumption, and low cost
appropriate to this application include the OP295, and members
of the OP90, OP97, OP177 families, and others as shown in the
following applications circuits.
With excellent drift and noise characteristics, VREF offers a
good voltage reference for data acquisition and transducer
excitation applications as well. Output drift is typically better
than −10 ppm/°C, with 315 nV/√Hz (typ) noise spectral density
at 1 kHz.
PRESERVING ACCURACY OVER WIDE
TEMPERATURE RANGE OPERATION
The TMP01 is unique in offering both a wide range temper-
ature sensor and the associated detection circuitry needed
to implement a complete thermostatic control function in
one monolithic device. While the voltage reference, setpoint
comparators, and output buffer amplifiers have been carefully
compensated to maintain accuracy over the specified temper-
ature range, the user has an additional task in maintaining the
accuracy over wide operating temperature ranges in the
application.
Since the TMP01 is both sensor and control circuit, in many
applications it is possible that the external components used to
program and interface the device may be subjected to the same
temperature extremes. Thus, it may be necessary to locate
components in close thermal proximity to minimize large
temperature differentials, and to account for thermal drift
errors, such as resistor matching tempcos, amplifier error drift,
and the like, where appropriate. Circuit design with the TMP01
requires a slightly different perspective regarding the thermal
behavior of electronic components.
THERMAL RESPONSE TIME
The time required for a temperature sensor to settle to a speci-
fied accuracy is a function of the thermal mass of the sensor,
and the thermal conductivity between the sensor and the object
being sensed. Thermal mass is often considered equivalent to
capacitance.
Thermal conductivity is commonly specified using the symbol
Q, and can be thought of as the reciprocal of thermal resistance.
It is commonly specified in units of degrees per watt of power
transferred across the thermal joint. Thus, the time required
for the TMP01 to settle to the desired accuracy is dependent
on the package selected, the thermal contact established in that
particular application, and the equivalent power of the heat
source. In most applications, the settling time is probably best
determined empirically.
TMP01
Rev. E | Page 11 of 20
SWITCHING LOADS WITH THE OPEN-COLLECTOR
OUTPUTS
In many temperature sensing and control applications, some
type of switching is required. Whether it be to turn on a heater
when the temperature goes below a minimum value or to turn
off a motor that is overheating, the open-collector outputs
OVER and UNDER can be used. For the majority of
applications, the switches used need to handle large currents on
the order of 1 A and above. Because the TMP01 is accurately
measuring temperature, the open-collector outputs should
handle less than 20 mA of current to minimize self-heating.
The OVER and UNDER outputs should not drive the equip-
ment directly. Instead, an external switching device is required
to handle the large currents. Some examples of these are relays,
power MOSFETs, thyristors, IGBTs, and Darlingtons.
Figure 17 through Figure 21 show a variety of circuits where the
TMP01 controls a switch. The main consideration in these
circuits, such as the relay in Figure 17, is the current required to
activate the switch.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
R1
R2
R3
MOTOR
SHUTDOWN
2604-12-311
COTO
IN4001
OR EQUIV.
12V
0
0333-017
Figure 17. Reed Relay Drive
It is important to check the particular relay to ensure that the
current needed to activate the coil does not exceed the TMP01’s
recommended output current of 20 mA. This is easily deter-
mined by dividing the relay coil voltage by the specified coil
resistance. Keep in mind that the inductance of the relay creates
large voltage spikes that can damage the TMP01 output unless
protected by a commutation diode across the coil, as shown.
The relay shown has a contact rating of 10 W maximum. If
a relay capable of handling more power is desired, the larger
contacts probably require a commensurately larger coil, with
lower coil resistance and thus higher trigger current. As the
contact power handling capability increases, so does the current
needed for the coil. In some cases, an external driving transistor
should be used to remove the current load on the TMP01.
Power FETs are popular for handling a variety of high current
dc loads. Figure 18 shows the TMP01 driving a p-channel
MOSFET transistor for a simple heater circuit. When the out-
put transistor turns on, the gate of the MOSFET is pulled down
to approximately 0.6 V, turning it on. For most MOSFETs, a
gate-to-source voltage, or Vgs, on the order of −2 V to −5 V
is sufficient to turn the device on.
Figure 19 shows a similar circuit for turning on an n-channel
MOSFET, except that now the gate to source voltage is positive.
For this reason, an external transistor must be used as an
inverter so that the MOSFET turns on when the UNDER
output pulls down.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
NC = NO CONNECT
TMP01
R1
R2
R3
NC
NC IRFR9024
OR EQUIV.
HEATING
ELEMENT
2.4kΩ (12V)
1.2kΩ (6V)
5%
V+
+
00333-018
Figure 18. Driving a P-Channel MOSFET
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
NC = NO CO NNECT
TMP01
R1
R2
R3
NC
NC
IRF130
2N1711
4.7kΩ
V+
4.7kΩHEATING
ELEMENT
00333-019
Figure 19. Driving an N-Channel MOSFET
Isolated gate bipolar transistors (IGBT) combine many of the
benefits of power MOSFETs with bipolar transistors, and are
used for a variety of high power applications. Because IGBTs
have a gate similar to MOSFETs, turning on and off the devices
is relatively simple as shown in Figure 20.
The turn-on voltage for the IGBT shown (IRGBC40S) is
between 3.0 V and 5.5 V. This part has a continuous collector
current rating of 50 A and a maximum collector-to-emitter
voltage of 600 V, enabling it to work in very demanding
applications.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
NC = NO CON NE CT
TMP01
R1
R2
R3
NC
NC
IRGBC40S
2N1711
4.7kΩ
V+
4.7kΩ
MOTOR
CONTROL
00333-020
Figure 20. Driving an IGBT
TMP01
Rev. E | Page 12 of 20
Thus, the output taken from the collector of Q2 is identical
to the output of the TMP01. By picking a transistor that can
accommodate large amounts of current, many high power
devices can be switched.
The last class of high power devices discussed here are
thyristors, which includes SCRs and Triacs. Triacs are a useful
alternative to relays for switching ac line voltages. The 2N6073A
shown in Figure 21 is rated to handle 4A (rms). The opto-
isolated MOC3011 Triac features excellent electrical isolation
from the noisy ac line and complete control over the high power
Triac with only a few additional components.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR Q1
TMP01
R1
R2
R3
V+
4.7kΩ
2N1711
I
C
00333-022
TEMPERATURE
SENS OR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
LOAD AC
WINDOW
COMPARATOR
NC = NO CON NE C T
TMP01
R1
R2
R3
NC
NC
V+ = 5V
300Ω
6
5
4
1
2
3
150Ω
2N6073A
MOC9011
00333-021
Figure 22. An External Resistor Minimizes Self-Heating
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR Q1
TMP01
R1
R2
R3
V+
4.7kΩ
2N1711
Q2
2N1711
I
C
4.7kΩ
00333-023
Figure 21. Controlling the 2N6073A Triac
HIGH CURRENT SWITCHING
Internal dissipation due to large loads on the TMP01 outputs
causes some temperature error due to self-heating. External
transistors remove the load from the TMP01, so that virtually
no power is dissipated in the internal transistors and no self-
heating occurs. Figure 22 through Figure 24 show a few
examples using external transistors. The simplest case, using a
single transistor on the output to invert the output signal is
shown in Figure 22. When the open collector of the TMP01
turns on and pulls the output down, the external transistor Q1
base is pulled low, turning off the transistor. Another transistor
can be added to reinvert the signal as shown in Figure 23. Now,
when the output of the TMP01 is pulled down, the first transis-
tor, Q1, turns off and its collector goes high, which turns Q2 on,
pulling its collector low.
Figure 23. Second Transistor Maintains Polarity of TMP01 Output
An example of a higher power transistor is a standard Darlington
configuration as shown in Figure 24. The part chosen, TIP-110,
can handle 2 A continuous which is more than enough to
control many high power relays. In fact, the Darlington itself
can be used as the switch, similar to MOSFETs and IGBTs.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
R1
R2
R3
V+
4.7kΩ
2N1711
RELAY
12
V
TIP-110
I
C
4.7kΩ
MOTOR
SWITCH
00333-024
Figure 24. Darlington Transistor Can Handle Large Currents
TMP01
Rev. E | Page 13 of 20
BUFFERING THE TEMPERATURE OUTPUT PIN
The VPTAT sensor output is a low impedance dc output voltage
with a 5 mV/K temperature coefficient, that is useful in multiple
measurement and control applications. In many applications,
this voltage needs to be transmitted to a central location for
processing. The buffered VPTAT voltage output is capable of
500 A drive into 50 pF (maximum).
Consider external amplifiers for interfacing VPTAT to external
circuitry to ensure accuracy, and to minimize loading which
could create dissipation-induced temperature sensing errors.
An excellent general-purpose buffer circuit using the OP177 is
shown in Figure 25. It is capable of driving over 10 mA, and
remains stable under capacitive loads of up to 0.1 F. Other
interfacing ideas are also provided in this section.
DIFFERENTIAL TRANSMITTER
In noisy industrial environments, it is difficult to send an
accurate analog signal over a significant distance. However,
by sending the signal differentially on a wire pair, these errors
can be significantly reduced. Because the noise is picked up
equally on both wires, a receiver with high common-mode
input rejection can be used to cancel out the noise very effec-
tively at the receiving end. Figure 26 shows two amplifiers used
to send the signal differentially, and an excellent differential
receiver, the AMP03, which features a common-mode rejection
ratio of 95 dB at dc and very low input and drift errors.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
OP177
R1
R2
R3 VPTAT V
OUT
C
L
V
+
V–
V+
100Ω
10kΩ
0.1µF
0
0333-025
Figure 25. Buffer VPTAT to Handle Difficult Loads
4 mA TO 20 mA CURRENT LOOP
Another common method of transmitting a signal over long
distances is to use a 4 mA to 20 mA loop, as shown in Figure 27.
An advantage of using a 4 mA to 20 mA loop is that the
accuracy of a current loop is not compromised by voltage drops
across the line. One requirement of 4 mA to 20 mA circuits is
that the remote end must receive all of its power from the loop,
meaning that the circuit must consume less than 4 mA.
Operating from 5 V, the quiescent current of the TMP01 is
500 A maximum, and the OP90s is 20 A maximum, totaling
less than 4 mA. Although not shown, the open collector outputs
and temperature setting pins can be connected to do any local
control of switching.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
1/2
OP297
1/2
OP297
AMP03
R1
R2
R3 VPTAT
V
OUT
V
+
50Ω
10kΩ
10kΩ
50ΩV–
V+
10kΩ
00333-026
Figure 26. Send the Signal Differentially for Noise Immunity
TMP01
Rev. E | Page 14 of 20
The current is proportional to the voltage on the VPTAT
output, and is calibrated to 4 mA at a temperature of −40°C, to
20 mA for +85°C. The main equation governing the operation
of this circuit gives the current as a function of VPTAT
⎟
⎠
⎞
⎜
⎝
⎛⎟
⎠
⎞
⎜
⎝
⎛+
+
×
−
×
=2
5
1
13
3
2
5
6
1
R
R
RR
RVREF
R
RVPTAT
R
IOUT
The resulting temperature coefficient of the output current is
128 A/°C.
5
8
1
4
2N1711
VREF
GND
V+
VPTAT
TMP01
OP90
4–20mA
5V TO 13.2V
7
4
3
6
2
R
L
R6
100Ω
R1
243kΩ
R2
39.2kΩ
R3
100kΩ
R5
100kΩ
00333-027
Figure 27. 4mA to 20 mA Current Loop
To determine the resistor values in this circuit, first note that
VREF remains constant over temperature. Thus, the ratio of
R5 over R2 must give a variation of IOUT from 4 mA to 20 mA
as VPTAT varies from 1.165 V at −40°C to 1.79 V at +85°C.
The absolute value of the resistors is not important, only the
ratio. For convenience, 100 k is chosen for R5. Once R2 is
calculated, the value of R3 and R1 is determined by substituting
4 mA for IOUT and 1.165 V for VPTAT and solving. The final
values are shown in the circuit. The OP90 is chosen for this
circuit because of its ability to operate on a single supply and its
high accuracy. For initial accuracy, a 10 k trim potentiometer
can be included in series with R3, and the value of R3 lowered
to 95 k. The potentiometer should be adjusted to produce an
output current of 12.3 mA at 25°C.
TEMPERATURE-TO-FREQUENCY CONVERTER
Another common method of transmitting analog information
is to convert a voltage to the frequency domain. This is easily
done with any of the low cost monolithic voltage-to-frequency
converters (VFCs) available, which feature a robust, open-
collector digital output. A digital signal is immune to noise
and voltage drops because the only important information is
the frequency. As long as the conversions between temperature
and frequency are done accurately, the temperature data can be
successfully transmitted.
A simple circuit to do this combines the TMP01 with an AD654
VFC, as shown in Figure 28. The AD654 outputs a square wave
that is proportional to the dc input voltage according to the
following equation:
T
IN
OUT CRR
V
F)21(10 +
=
By simply connecting the VPTAT output to the input of the
AD654, the 5 mV/°C temperature coefficient gives a sensitivity
of 25 Hz/°C, centered around 7.5 kHz at 25°C. The trimming
resistor R2 is needed to calibrate the absolute accuracy of the
AD654. For more information on that part, consult the AD654
data sheet. Finally, the AD650 can be used to accurately convert
the frequency back to a dc voltage on the receiving end.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
VPTAT
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
R1
R2
R3
V
+
F
OUT
V+
V+
4
3
1
2
5
AD654
OSC
78 6
R1
1.8kΩ
R2
500Ω
5kΩ
C
T
0.1µF
00333-028
Figure 28. Temperature-to-Frequency Converter
TMP01
Rev. E | Page 15 of 20
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
ISOLATION
BARRIER
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
OP290
R1
R2
R3
V+
604kΩ100kΩ
R1
470kΩ
V+
IN4148 I1I2
6
5
3
4
1
2
2.5V
V+
REF43 4
62
1.16V TO 1.7V
100kΩ
680pF
OP290
7
4
3
6
2
V+
OP90
7
4
3
6
2
IL300XC
680pF
00333-029
Figure 29. Isolation Amplifier
ISOLATION AMPLIFIER
In many industrial applications, the sensor is located in an envi-
ronment that needs to be electrically isolated from the central
processing area. Figure 29 shows a simple circuit that uses an
8-pin optoisolator (IL300XC) that can operate across a 5,000 V
barrier. IC1 (an OP290 single-supply amplifier) is used to drive
the LED connected between Pin 1 and Pin 2. The feedback
actually comes from the photodiode connected from Pin 3 to
Pin 4. The OP290 drives the LED such that there is enough
current generated in the photodiode to exactly equal the current
derived from the VPTAT voltage across the 470 k resistor.
On the receiving end, an OP90 converts the current from the
second photodiode to a voltage through its feedback resistor R2.
Note that the other amplifier in the dual OP290 is used to buffer
the 2.5 V reference voltage of the TMP01 for an accurate, low
drift LED bias level without affecting the programmed hyster-
esis current. A REF43 (a precision 2.5 V reference) provides an
accurate bias level at the receiving end.
To understand this circuit, it helps to examine the overall
equation for the output voltage. First, the current (I1) in the
photodiode is set by
k470
V5.2
1
VPTAT
I−
=
Note that the IL300XC has a gain of 0.73 (typical) with a
minimum and maximum of 0.693 and 0.769, respectively.
Because this is less than 1.0, R2 must be larger than R1 to
achieve overall unity gain. To show this, the full equation is
=−= 22
5.2 RIVVOUT
VPTAT
VPTATV
V⎜
⎜
⎝
⎛−
−k470
5.2
7.05.2 =
⎟
⎟
⎠
⎞k644
A trim is included for R2 to correct for the initial gain accuracy
of the IL300XC. To perform this trim, simply adjust for an
output voltage equal to VPTAT at any particular temperature.
For example, at room temperature, VPTAT = 1.49 V, so adjust
R2 until VOUT = 1.49 V as well. Both the REF43 and the OP90
operate from a single supply, and contribute no significant error
due to drift.
In order to avoid the accuracy trim, and to reduce board space,
complete isolation amplifiers are available, such as the high
accuracy AD202.
OUT-OF-RANGE WARNING
By connecting the two open-collector outputs of the TMP01
together into a wired-OR configuration, a temperature out-
of-range warning signal is generated. This can be useful in
sensitive equipment calibrated to work over a limited temper-
ature range.
R1, R2, and R3 in Figure 30 are chosen to give a temperature
range of 10°C around room temperature (25°C). Thus, if the
temperature in the equipment falls below 15°C or rises above
35°C, the OVER or UNDERoutput, respectively, goes low and
turns the LED on. The LED may be replaced with a simple pull-
up resistor to give a logic output for controlling the instrument,
or any of the switching devices discussed above can be used.
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
VREF VPTAT
VPTAT
LED
1
2
3
4
8
7
6
5
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
TMP01
R1
7.5kΩ
R2
.99kΩ
R3
71.5kΩ
V
+
200Ω
00333-030
Figure 30. Out-of-Range Warning
TMP01
Rev. E | Page 16 of 20
TRANSLATING 5 mV/K TO 10 mV/°C
A useful circuit shown in Figure 31 translates the VPTAT
output voltage, which is calibrated in Kelvins, into an output
that can be read directly in degrees Celsius on a voltmeter
display.
To accomplish this, an external amplifier is configured as a
differential amplifier. The resistors are scaled so the VREF
voltage exactly cancels the VPTAT voltage at 0.0°C.
5
1
+15V
–15V
10pF
V
OUT
= (10mV/° C)
(V
OUT
= 0. 0V @ T = 0.0° C)
VPTAT
VREF
TMP01
OP177
7
4
3
6
2
100kΩ
100kΩ
105kΩ4.22kΩ
4.12kΩ487Ω
00333-031
Figure 31. Translating 5 mV/K to 10 mV/°C
However, the gain from VPTAT to the output is two, so that
5 mV/K becomes 10 mV/°C. Thus, for a temperature of 80°C,
the output voltage is 800 mV. Circuit errors will be due prima-
rily to the inaccuracies of the resistor values. Using 1% resistors,
the observed error was less than 10 mV, or 1°C. The 10 pF
feedback capacitor helps to ensure against oscillations. For
better accuracy, an adjustment potentiometer can be added in
series with either 100 k resistor.
TRANSLATING VPTAT TO THE FAHRENHEIT SCALE
A similar circuit to the one shown in Figure 31 can be used
to translate VPTAT into an output that can be read directly in
degrees Fahrenheit, with a scaling of 10 mV/°F. Only unity gain
or less is available from the first stage differentiating circuit, so
the second amplifier provides a gain of two to complete the
conversion to the Fahrenheit scale. Using the circuit in Figure 32,
a temperature of 0.0°F gives an output of 0.00 V. At room temp-
erature (70°F), the output voltage is 700 mV. A −40°C to +85°C
operating range translates into −40°F to +185°F. The errors are
essentially the same as for the circuit in Figure 31.
5
1
+15V
–15V
10p
F
VOUT = 0. 0V @ T = 0.0°F
(10mV/°F)
VPTAT
VREF
TMP01
1/2
OP297
7
4
3
6
2
100kΩ
100kΩ
90.9kΩ1.0kΩ
1/2
OP297
5
7
6
100kΩ
6.49kΩ121Ω
100kΩ
00333-032
Figure 32. Translating 5 mV/K to 10 mV/°F
TMP01
Rev. E | Page 17 of 20
OUTLINE DIMENSIONS
COMPLIANT TO JEDE C STANDARDS MS-001
CONTRO LLI NG DIMENSIONS ARE I N INCHES; MI LLI M E TER DIMENSIO NS
(IN PARENTHESES) ARE RO UNDED- O F F INCH EQ UIVALENTS F O R
REFERENCE ONLY AND ARE NOT APPRO PRIATE FOR USE I N DE SIGN.
CORNER LE ADS MAY BE CONF I G URE D AS WHOLE O R HALF LEADS .
070606-A
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
SEATING
PLANE
0.015
(0.38)
MIN
0.210 (5.33)
MAX
0.150 ( 3.81)
0.130 ( 3.30)
0.115 (2.92)
0.070 ( 1.78)
0.060 ( 1.52)
0.045 ( 1.14)
8
14
5
0.280 ( 7.11)
0.250 ( 6.35)
0.240 ( 6.10)
0.100 (2 . 5 4)
BSC
0.400 (10.16)
0.365 (9.27)
0.355 (9.02)
0.060 ( 1.52)
MAX
0.430 ( 10.92)
MAX
0.014 (0.36)
0.010 (0.25)
0.008 (0.20)
0.325 ( 8.26)
0.310 ( 7.87)
0.300 ( 7.62)
0.195 ( 4.95)
0.130 ( 3.30)
0.115 (2.9 2)
0.015 (0.38)
GAUGE
PLANE
0.005 ( 0.13)
MIN
Figure 33. .8-Lead Plastic Dual In-Line Package [PDIP]
Narrow Body
(N-8)
Dimensions shown in inches and (millimeters)
CONTROLLI N G DI M E NS IO NS ARE IN MIL LIMETERS ; IN CH DIME N SIO N S
(I N PARENTHE S E S ) ARE ROUNDE D- OF F MIL LI M E TE R EQUIVALE NTS FO R
REF E RENCE ON LY A ND ARE NOT APPRO PRI ATE FO R USE IN DES IG N.
COM P LIANT TO JEDEC S TANDARDS M S -012-AA
012407-A
0.25 (0.0098)
0.17 (0.0067)
1.27 (0. 0500)
0.40 (0. 0157)
0.50 (0. 0196)
0.25 (0. 0099) 45°
8°
0°
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.2 5 ( 0.009 8)
0.1 0 ( 0.004 0)
4
1
85
5.00 (0.1968)
4.80 (0.1890)
4.00 ( 0.157 4)
3.80 ( 0.149 7)
1.27 (0.0500)
BSC
6.20 (0. 2441)
5.80 (0. 2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
Figure 34. 8-Lead Standard Small Outline package [SOIC_N]
Narrow Body
(R-8)
Dimensions shown in millimeters and (inches)
TMP01
Rev. E | Page 18 of 20
CONT ROLLING DI M E NSIONS ARE I N INCHES; M IL L IMETER DIMENS IONS
(IN PARENT HESES) ARE ROUNDED-O FF INCH EQ UIVALENTS F OR
REFE RE NCE ONLY AND ARE NOT APPRO P RIATE FOR USE IN DESIGN.
COMP LIANT TO JEDEC STANDARDS MO-002- AK
0.2500 (6.35) MIN
0.5000 ( 1 2.70)
MIN
0.1850 (4.70)
0.1650 (4.19)
REFE RE NCE P LANE
0.0500 (1.27) MAX
0.0190 ( 0.48)
0.0160 ( 0.41)
0.0210 ( 0. 53 )
0.0160 ( 0. 41 )
0.0400 ( 1. 02 )
0.0100 ( 0. 25 )
0.0400 ( 1.02) M A X 0.0340 ( 0.86)
0.0280 ( 0.71)
0.0450 ( 1.14)
0.0270 ( 0.69)
0.1600 ( 4.06)
0.1400 ( 3.56)
0.1000 (2.54)
BSC
6
28
7
5
4
3
1
0.2000
(5.08)
BSC
0.1000
(2.54)
BSC
0.3700 ( 9. 40 )
0.3350 ( 8. 51 )
0.3350 (8.51)
0.3050 (7.75)
45° BSC
BASE & SEATING PLANE
022306-A
Figure 35. 8-Pin Metal Header [TO-99]
(H-08)
Dimensions shown in inches and (millimeters)
ORDERING GUIDE
Model/Grade Temperature Range Package Description Package Option
TMP01ES −40°C to +85°C 8-Lead SOIC_N R-8
TMP01ES-REEL −40°C to +85°C 8-Lead SOIC_N R-8
TMP01ESZ1−40°C to +85°C 8-Lead SOIC_N R-8
TMP01ESZ-REEL1 −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FP −40°C to +85°C 8-Lead PDIP N-8
TMP01FPZ1 −40°C to +85°C 8-Lead PDIP N-8
TMP01FS −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FS-REEL −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FS-REEL7 −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FSZ1 −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FSZ-REEL1 −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FSZ-REEL71 −40°C to +85°C 8-Lead SOIC_N R-8
TMP01FJ −40°C to +85°C 8-Pin Metal Header (TO-99) H-08
1 Z = RoHS Compliant Part.
TMP01
Rev. E | Page 19 of 20
NOTES
TMP01
Rev. E | Page 20 of 20
NOTES
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