Poly Phase Multifunction Energy Metering IC
with Per Phase Information
ADE7758
Rev. D
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responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
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FEATURES
Highly accurate; supports IEC 60687, IEC 61036, IEC 61268,
IEC 62053-21, IEC 62053-22, and IEC 62053-23
Compatible with 3-phase/3-wire, 3-phase/4-wire, and other
3-phase services
Less than 0.1% active energy error over a dynamic range of
1000 to 1 at 25°C
Supplies active/reactive/apparent energy, voltage rms,
current rms, and sampled waveform data
Two pulse outputs, one for active power and the other
selectable between reactive and apparent power with
programmable frequency
Digital power, phase, and rms offset calibration
On-chip, user-programmable thresholds for line voltage SAG
and overvoltage detections
An on-chip, digital integrator enables direct interface-to-
current sensors with di/dt output
A PGA in the current channel allows direct interface to
shunts and current transformers
An SPI®-compatible serial interface with IRQ
Proprietary ADCs and DSP provide high accuracy over large
variations in environmental conditions and time
Reference 2.4 V (drift 30 ppm/°C typical) with external
overdrive capability
Single 5 V supply, low power (70 mW typical)
GENERAL DESCRIPTION
The ADE7758 is a high accuracy, 3-phase electrical energy
measurement IC with a serial interface and two pulse outputs.
The ADE7758 incorporates second-order Σ-Δ ADCs, a digital
integrator, reference circuitry, a temperature sensor, and all the
signal processing required to perform active, reactive, and
apparent energy measurement and rms calculations.
The ADE7758 is suitable to measure active, reactive, and
apparent energy in various 3-phase configurations, such as
WYE or DELTA services, with both three and four wires. The
ADE7758 provides system calibration features for each phase,
that is, rms offset correction, phase calibration, and power
calibration. The APCF logic output gives active power
information, and the VARCF logic output provides instantaneous
reactive or apparent power information (continued on Page 4).
FUNCTIONAL BLOCK DIAGRAM
PHASE B
AND
PHASE C
DATA
4
AVDD
POWER
SUPPLY
MONITOR
12
REF
IN/OUT
11
AGND
ADC
–
+
9
ICP
10
ICN
PGA1
ADC
–
+
14
V
CP
13
VN
PGA2
ACTIVE/REACTIVE/APPARENT ENERGIES AND
VOLTAGE/CURRENT RMS CALCUL ATION FOR
PHASE C
(SEE PHASE A FOR DETAILED SIGNAL PATH)
ADC
–
+
7
IBP
8
IBN
PGA1
ADC
–
+
15
V
BP
PGA2
ACTIVE/REACTIVE/APPARENT ENERGIES AND
VOLTAGE/CURRENT RMS CALCUL ATION FOR
PHASE B
(SEE PHASE A FOR DETAILED SIGNAL PATH)
ADC
–
+
5
IAP
6
IAN
PGA1
ADC
–
+
16
VAP
PGA2
AIGAIN[11:0]
AVRMSGAIN[11:0]
AVAG[11:0]
|
X
|
APHCAL[6:0]
Φ
HPF INTEGRATOR
dt
AVAROS[11:0] AVARG[11:0]
LPF2
90° PHASE
SHIFTING FILTER
π
2
AWATTOS[11:0] AWG[11:0]
LPF2
22
DIN
24
DOUT
23
SCLK
21
CS
18
IRQ
ADE7758 REGISTERS AND
SERIAL INTERFACE
WDIV[7:0]
%
VARDIV[7:0]
%
VADIV[7:0]
%
AIRMSOS[11:0]
X
2
LPF
2.4V
REF
4kΩ
DFC
÷
APCFNUM[11:0]
APCFDEN[11:0]
ACTIVE POWER
1
APCF
3
DVDD
2
DGND
19
CLKIN
20
CLKOU
T
DFC
VARCFNUM[11:0]
VARCFDEN[11:0]
REACTIVE OR
APPARENT POWER
17
VARCF
ADE7758
AVRMSOS [11: 0]
04443-001
÷
Figure 1.
ADE7758
Rev. D | Page 2 of 72
TABLE OF CONTENTS
Features .............................................................................................. 1
General Description ......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 3
General Description ......................................................................... 4
Specifications ..................................................................................... 5
Timing Characteristics ................................................................ 6
Timing Diagrams .............................................................................. 7
Absolute Maximum Ratings ............................................................ 8
ESD Caution .................................................................................. 8
Pin Configuration and Function Descriptions ............................. 9
Terminology .................................................................................... 11
Typical Performance Characteristics ........................................... 12
Test Circuits ..................................................................................... 17
Theory of Operation ...................................................................... 18
Antialiasing Filter ....................................................................... 18
Analog Inputs .............................................................................. 18
Current Channel ADC............................................................... 19
di/dt Current Sensor and Digital Integrator ............................... 20
Peak Current Detection ............................................................. 21
Overcurrent Detection Interrupt ............................................. 21
Voltage Channel ADC ............................................................... 22
Zero-Crossing Detection ........................................................... 23
Phase Compensation .................................................................. 23
Period Measurement .................................................................. 25
Line Voltage SAG Detection ..................................................... 25
SAG Level Set .............................................................................. 26
Peak Voltage Detection .............................................................. 26
Phase Sequence Detection ......................................................... 27
Power-Supply Monitor ............................................................... 27
Reference Circuit ........................................................................ 27
Temperature Measurement ....................................................... 28
Root Mean Square Measurement ............................................. 28
Active Power Calculation .......................................................... 30
Reactive Power Calculation ...................................................... 35
Apparent Power Calculation ..................................................... 39
Energy Registers Scaling ........................................................... 41
Waveform Sampling Mode ....................................................... 41
Calibration ................................................................................... 42
Checksum Register..................................................................... 55
Interrupts ..................................................................................... 56
Using the Interrupts with an MCU .......................................... 56
Interrupt Timing ........................................................................ 56
Serial Interface ............................................................................ 56
Serial Write Operation ............................................................... 57
Serial Read Operation ................................................................ 59
Accessing the On-Chip Registers ............................................. 59
Registers ........................................................................................... 60
Communications Register ......................................................... 60
Operational Mode Register (0x13) .......................................... 64
Measurement Mode Register (0x14) ....................................... 64
Waveform Mode Register (0x15) ............................................. 65
Computational Mode Register (0x16) ..................................... 66
Line Cycle Accumulation Mode Register (0x17) ................... 67
Interrupt Mask Register (0x18) ................................................ 68
Interrupt Status Register (0x19)/Reset Interrupt Status
Register (0x1A) ........................................................................... 69
Outline Dimensions ....................................................................... 70
Ordering Guide .......................................................................... 70
ADE7758
Rev. D | Page 3 of 72
REVISION HISTORY
10/08—Rev. C to Rev. D
Changes to Figure 1 ........................................................................... 1
Changes to Phase Sequence Detection Section and Figure 60 .. 27
Changes to Current RMS Calculation Section ............................ 28
Changes to Voltage Channel RMS Calculation Section and
Figure 63 ........................................................................................... 29
Changes to Table 17 ........................................................................ 60
Changes to Ordering Guide ........................................................... 70
7/06—Rev. B to Rev. C
Updated Format .................................................................. Universal
Changes to Figure 1 ........................................................................... 1
Changes to Table 2 ............................................................................ 6
Changes to Table 4 ............................................................................ 9
Changes to Figure 34 and Figure 35 ............................................. 17
Changes to Current Waveform Gain Registers Section and
Current Channel Sampling Section .............................................. 19
Changes to Voltage Channel Sampling Section .......................... 22
Changes to Zero-Crossing Timeout Section ............................... 23
Changes to Figure 60 ...................................................................... 27
Changes to Current RMS Calculation Section ............................ 28
Changes to Current RMS Offset Compensation Section and
Voltage Channel RMS Calculation Section ................................. 29
Added Table 7 and Table 9; Renumbered Sequentially .............. 29
Changes to Figure 65 ...................................................................... 30
Changes to Active Power Offset Calibration Section ................. 31
Changes to Reactive Power Frequency Output Section ............. 38
Changes to Apparent Power Frequency Output Section and
Waveform Sampling Mode Section .............................................. 41
Changes to Gain Calibration Using Line
Accumulation Section .................................................................... 49
Changes to Example: Power Offset Calibration Using Line
Accumulation Section .................................................................... 53
Changes to Calibration of IRMS and VRMS Offset Section ..... 54
Changes to Table 18 ........................................................................ 64
Changes to Table 20 ........................................................................ 65
11/05—Rev. A to Rev. B
Changes to Table 1 ............................................................................ 5
Changes to Figure 23 Caption ....................................................... 14
Changes to Current Waveform Gain Registers Section ............. 19
Changes to di/dt Current Sensor and Digital
Integrator Section ............................................................................ 20
Changes to Phase Compensation Section .................................... 23
Changes to Figure 57 ...................................................................... 25
Changes to Figure 60 ...................................................................... 27
Changes to Temperature Measurement Section
and Root Mean Square Measurement Section ............................ 28
Inserted Table 6................................................................................ 28
Changes to Current RMS Offset Compensation Section .......... 29
Inserted Table 7................................................................................ 29
Added Equation 17 ......................................................................... 31
Changes to Energy Accumulation Mode Section ....................... 33
Changes to the Reactive Power Calculation Section .................. 35
Added Equation 32 ........................................................................... 36
Changes to Energy Accumulation Mode Section ....................... 38
Changes to the Reactive Power Frequency Output Section ...... 38
Changes to the Apparent Energy Calculation Section ............... 40
Changes to the Calibration Section .............................................. 42
Changes to Figure 76 through Figure 84 ............................... 43–54
Changes to Table 15 ........................................................................ 59
Changes to Table 16 ........................................................................ 63
Changes to Ordering Guide ........................................................... 69
9/04—Rev. 0 to Rev. A
Changed Hexadecimal Notation ...................................... Universal
Changes to Features List ................................................................... 1
Changes to Specifications Table ...................................................... 5
Change to Figure 25 ........................................................................ 16
Additions to the Analog Inputs Section ....................................... 19
Added Figures 36 and 37; Renumbered Subsequent Figures .... 19
Changes to Period Measurement Section .................................... 26
Change to Peak Voltage Detection Section ................................. 26
Added Figure 60 .............................................................................. 27
Change to the Current RMS Offset Compensation Section ..... 29
Edits to Active Power Frequency Output Section ...................... 33
Added Figure 68; Renumbered Subsequent Figures .................. 33
Changes to Reactive Power Frequency Output Section ............. 37
Added Figure 73; Renumbered Subsequent Figures .................. 38
Change to Gain Calibration Using Pulse Output Example ....... 44
Changes to Equation 37 ................................................................. 45
Changes to Example—Phase Calibration of Phase A
Using Pulse Output ......................................................................... 45
Changes to Equations 56 and 57 ................................................... 53
Addition to the ADE7758 Interrupts Section ............................. 54
Changes to Example-Calibration of RMS Offsets ...................... 54
Addition to Table 20 ....................................................................... 66
1/04—Revision 0: Initial Version
ADE7758
Rev. D | Page 4 of 72
GENERAL DESCRIPTION
(continued from Page 1) The ADE7758 has a waveform sample
register that allows access to the ADC outputs. The part also
incorporates a detection circuit for short duration low or high
voltage variations. The voltage threshold levels and the duration
(number of half-line cycles) of the variation are user
programmable. A zero-crossing detection is synchronized with
the zero-crossing point of the line voltage of any of the three
phases. This information can be used to measure the period of
any one of the three voltage inputs. The zero-crossing detection
is used inside the chip for the line cycle energy accumulation
mode. This mode permits faster and more accurate calibration
by synchronizing the energy accumulation with an integer
number of line cycles.
Data is read from the ADE7758 via the SPI serial interface. The
interrupt request output (IRQ) is an open-drain, active low logic
output. The IRQ output goes active low when one or more
interrupt events have occurred in the ADE7758. A status register
indicates the nature of the interrupt. The ADE7758 is available
in a 24-lead SOIC package.
ADE7758
Rev. D | Page 5 of 72
SPECIFICATIONS
AVDD = DVDD = 5 V ± 5%, AGND = DGND = 0 V, on-chip reference, CLKIN = 10 MHz XTAL, TMIN to TMAX = −40°C to +85°C.
Table 1.
Parameter1, 2Specification Unit Test Conditions/Comments
ACCURACY
Active Energy Measurement Error
(per Phase)
0.1 % typ Over a dynamic range of 1000 to 1
Phase Error Between Channels Line frequency = 45 Hz to 65 Hz, HPF on
PF = 0.8 Capacitive ±0.05 °max Phase lead 37°
PF = 0.5 Inductive ±0.05 °max Phase lag 60°
AC Power Supply Rejection AVDD = DVDD = 5 V + 175 mV rms/120 Hz
Output Frequency Variation 0.01 % typ V1P = V2P = V3P = 100 mV rms
DC Power Supply Rejection AVDD = DVDD = 5 V ± 250 mV dc
Output Frequency Variation 0.01 % typ V1P = V2P = V3P = 100 mV rms
Active Energy Measurement Bandwidth 14 kHz
IRMS Measurement Error 0.5 % typ Over a dynamic range of 500:1
IRMS Measurement Bandwidth 14 kHz
VRMS Measurement Error 0.5 % typ Over a dynamic range of 20:1
VRMS Measurement Bandwidth 260 Hz
ANALOG INPUTS See the Analog Inputs section
Maximum Signal Levels ±500 mV max Differential input
Input Impedance (DC) 380 kΩ min
ADC Offset Error3±30 mV max Uncalibrated error, see the Terminology section
Gain Error3
±6 % typ External 2.5 V reference
WAVEFORM SAMPLING Sampling CLKIN/128, 10 MHz/128 = 78.1 kSPS
Current Channels See the Current Channel ADC section
Signal-to-Noise Plus Distortion 62 dB typ
Bandwidth (−3 dB) 14 kHz
Voltage Channels See the Voltage Channel ADC section
Signal-to-Noise Plus Distortion 62 dB typ
Bandwidth (−3 dB) 260 Hz
REFERENCE INPUT
REFIN/OUT Input Voltage Range 2.6 V max 2.4 V + 8%
2.2 V min 2.4 V − 8%
Input Capacitance 10 pF max
ON-CHIP REFERENCE Nominal 2.4 V at REFIN/OUT pin
Reference Error ±200 mV max
Current Source 6 μA max
Output Impedance 4 kΩ min
Temperature Coefficient 30 ppm/°C typ
CLKIN All specifications CLKIN of 10 MHz
Input Clock Frequency 15 MHz max
5 MHz min
LOGIC INPUTS
DIN, SCLK, CLKIN, and CS
Input High Voltage, VINH 2.4 V min DVDD = 5 V ± 5%
Input Low Voltage, VINL 0.8 V max DVDD = 5 V ± 5%
Input Current, IIN ±3 μA max Typical 10 nA, VIN = 0 V to DVDD
Input Capacitance, CIN 10 pF max
ADE7758
Rev. D | Page 6 of 72
Parameter1, 2
Specification Unit Test Conditions/Comments
LOGIC OUTPUTS DVDD = 5 V ± 5%
IRQ, DOUT, and CLKOUT
IRQ is open-drain, 10 kΩ pull-up resistor
Output High Voltage, VOH 4 V min ISOURCE = 5 mA
Output Low Voltage, VOL 0.4 V max ISINK = 1 mA
APCF and VARCF
Output High Voltage, VOH 4 V min ISOURCE = 8 mA
Output Low Voltage, VOL 1 V max ISINK = 5 mA
POWER SUPPLY For specified performance
AVDD 4.75 V min 5 V − 5%
5.25 V max 5 V + 5%
DVDD 4.75 V min 5 V − 5%
5.25 V max 5 V + 5%
AIDD 8 mA max Typically 5 mA
DIDD 13 mA max Typically 9 mA
1 See the Typical Performance Characteristics.
2 See the Terminology section for a definition of the parameters.
3 See the Analog Inputs section.
TIMING CHARACTERISTICS
AVDD = DVDD = 5 V ± 5%, AGND = DGND = 0 V, on-chip reference, CLKIN = 10 MHz XTAL, TMIN to TMAX = −40°C to +85°C.
Table 2.
Parameter1, 2Specification Unit Test Conditions/Comments
WRITE TIMING
t1 50 ns (min) CS falling edge to first SCLK falling edge
t2 50 ns (min) SCLK logic high pulse width
t3 50 ns (min) SCLK logic low pulse width
t4 10 ns (min) Valid data setup time before falling edge of SCLK
t5 5 ns (min) Data hold time after SCLK falling edge
t6 900 ns (min) Minimum time between the end of data byte transfers
t7 50 ns (min) Minimum time between byte transfers during a serial write
t8 100 ns (min) CS hold time after SCLK falling edge
READ TIMING
t934 μs (min)
Minimum time between read command (that is, a write to communication register) and
data read
t10 50 ns (min) Minimum time between data byte transfers during a multibyte read
t114 30 ns (min) Data access time after SCLK rising edge following a write to the communications register
t125 100 ns (max) Bus relinquish time after falling edge of SCLK
10 ns (min)
t135
100 ns (max) Bus relinquish time after rising edge of CS
10 ns (min)
1 Sample tested during initial release and after any redesign or process change that may affect this parameter. All input signals are specified with tr = tf = 5 ns (10% to
90%) and timed from a voltage level of 1.6 V.
2 See the timing diagrams in Figure 3 and Figure 4 and the Serial Interface section.
3 Minimum time between read command and data read for all registers except waveform register, which is t9 = 500 ns min.
4 Measured with the load circuit in Figure 2 and defined as the time required for the output to cross 0.8 V or 2.4 V.
5 Derived from the measured time taken by the data outputs to change 0.5 V when loaded with the circuit in Figure 2. The measured number is then extrapolated back
to remove the effects of charging or discharging the 50 pF capacitor. This means that the time quoted here is the true bus relinquish time of the part and is
independent of the bus loading.
ADE7758
Rev. D | Page 7 of 72
TIMING DIAGRAMS
200µA I
OL
1.6mA I
OH
2.1V
TO OUTPUT
PIN C
L
50pF
04443-002
Figure 2. Load Circuit for Timing Specifications
DIN
S
CLK
CS
t2
t3
t1
t4t5
t7
t6
t8
COMMAND BYTE MOST SIGNIFICANT BYTE LEAST SIGNIFICANT BYTE
1A6 A4A5 A3 A2 A1 A0 DB7 DB0 DB7 DB0
t7
04443-003
Figure 3. Serial Write Timing
SCLK
CS
t
1
t
10
t
13
0A6 A4A5 A3 A2 A1 A0
DB0
DB7 DB0
DB7
DIN
DOUT
t
11
t
12
COMMAND BYTE MOST SIGNIFICANT BYTE LEAST SIGNIFICANT BYTE
t
9
04443-004
Figure 4. Serial Read Timing
ADE7758
Rev. D | Page 8 of 72
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
Table 3.
Parameter Rating
AVDD to AGND –0.3 V to +7 V
DVDD to DGND –0.3 V to +7 V
DVDD to AVDD –0.3 V to +0.3 V
Analog Input Voltage to AGND,
IAP, IAN, IBP, IBN, ICP, ICN, VAP,
VBP, VCP, VN
–6 V to +6 V
Reference Input Voltage to AGND –0.3 V to AVDD + 0.3 V
Digital Input Voltage to DGND –0.3 V to DVDD + 0.3 V
Digital Output Voltage to DGND –0.3 V to DVDD + 0.3 V
Operating Temperature
Industrial Range –40°C to +85°C
Storage Temperature Range –65°C to +150°C
Junction Temperature 150°C
24-Lead SOIC, Power Dissipation 88 mW
θJA Thermal Impedance 53°C/W
Lead Temperature, Soldering
Vapor Phase (60 sec) 215°C
Infrared (15 sec) 220°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
ADE7758
Rev. D | Page 9 of 72
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
APCF 1
DGND 2
DVDD 3
AVDD 4
DOUT24
SCLK23
DIN
22
CS21
IAP 5CLKOUT
20
IAN 6CLKIN19
IBP 7IRQ
18
IBN 8VARCF17
ICP 9VAP16
ICN 10 VBP
15
AGND 11 VCP14
REFIN/OUT 12 VN
13
ADE7758
TOP VIEW
(Not to Scale)
04443-005
Figure 5. Pin Configuration
Table 4. Pin Function Descriptions
Pin
No. Mnemonic Description
1 APCF Active Power Calibration Frequency (APCF) Logic Output. It provides active power information. This output
is used for operational and calibration purposes. The full-scale output frequency can be scaled by writing to
the APCFNUM and APCFDEN registers (see the Active Power Frequency Output section).
2 DGND This provides the ground reference for the digital circuitry in the ADE7758, that is, the multiplier, filters, and
digital-to-frequency converter. Because the digital return currents in the ADE7758 are small, it is acceptable to
connect this pin to the analog ground plane of the whole system. However, high bus capacitance on the DOUT
pin can result in noisy digital current that could affect performance.
3 DVDD Digital Power Supply. This pin provides the supply voltage for the digital circuitry in the ADE7758. The supply
voltage should be maintained at 5 V ± 5% for specified operation. This pin should be decoupled to DGND with
a 10 μF capacitor in parallel with a ceramic 100 nF capacitor.
4 AVDD Analog Power Supply. This pin provides the supply voltage for the analog circuitry in the ADE7758. The supply
should be maintained at 5 V ± 5% for specified operation. Every effort should be made to minimize power
supply ripple and noise at this pin by the use of proper decoupling. The Typical Performance Characteristics
show the power supply rejection performance. This pin should be decoupled to AGND with a 10 μF capacitor
in parallel with a ceramic 100 nF capacitor.
5, 6,
7, 8,
9, 10
IAP, IAN,
IBP, IBN,
ICP, ICN
Analog Inputs for Current Channel. This channel is used with the current transducer and is referenced in this
document as the current channel. These inputs are fully differential voltage inputs with maximum differential
input signal levels of ±0.5 V, ±0.25 V, and ±0.125 V, depending on the gain selections of the internal PGA (see
the Analog Inputs section). All inputs have internal ESD protection circuitry. In addition, an overvoltage
of ±6 V can be sustained on these inputs without risk of permanent damage.
11 AGND This pin provides the ground reference for the analog circuitry in the ADE7758, that is, ADCs, temperature sensor,
and reference. This pin should be tied to the analog ground plane or the quietest ground reference in the system.
This quiet ground reference should be used for all analog circuitry, for example, antialiasing filters, current, and
voltage transducers. To keep ground noise around the ADE7758 to a minimum, the quiet ground plane should be
connected to the digital ground plane at only one point. It is acceptable to place the entire device on the analog
ground plane.
12 REFIN/OUT This pin provides access to the on-chip voltage reference. The on-chip reference has a nominal value of
2.4 V ± 8% and a typical temperature coefficient of 30 ppm/°C. An external reference source can also be
connected at this pin. In either case, this pin should be decoupled to AGND with a 1 μF ceramic capacitor.
13, 14,
15, 16
VN, VCP,
VBP, VAP
Analog Inputs for the Voltage Channel. This channel is used with the voltage transducer and is referenced as
the voltage channels in this document. These inputs are single-ended voltage inputs with the maximum signal
level of ±0.5 V with respect to VN for specified operation. These inputs are voltage inputs with maximum input
signal levels of ±0.5 V, ±0.25 V, and ±0.125 V, depending on the gain selections of the internal PGA (see the
Analog Inputs section). All inputs have internal ESD protection circuitry, and in addition, an overvoltage of
±6 V can be sustained on these inputs without risk of permanent damage.
ADE7758
Rev. D | Page 10 of 72
Pin
No. Mnemonic Description
17 VARCF Reactive Power Calibration Frequency Logic Output. It gives reactive power or apparent power information
depending on the setting of the VACF bit of the WAVMODE register. This output is used for operational and
calibration purposes. The full-scale output frequency can be scaled by writing to the VARCFNUM and VARCFDEN
registers (see the Reactive Power Frequency Output section).
18 IRQ Interrupt Request Output. This is an active low open-drain logic output. Maskable interrupts include: an active
energy register at half level, an apparent energy register at half level, and waveform sampling up to 26 kSPS (see
the Interrupts section).
19 CLKIN Master Clock for ADCs and Digital Signal Processing. An external clock can be provided at this logic input.
Alternatively, a parallel resonant AT crystal can be connected across CLKIN and CLKOUT to provide a clock
source for the ADE7758. The clock frequency for specified operation is 10 MHz. Ceramic load capacitors of
a few tens of picofarad should be used with the gate oscillator circuit. Refer to the crystal manufacturer’s
data sheet for the load capacitance requirements
20 CLKOUT
A crystal can be connected across this pin and CLKIN as previously described to provide a clock source for
the ADE7758. The CLKOUT pin can drive one CMOS load when either an external clock is supplied at CLKIN or
a crystal is being used.
21 CS Chip Select. Part of the 4-wire serial interface. This active low logic input allows the ADE7758 to share the serial
bus with several other devices (see the Serial Interface section).
22 DIN Data Input for the Serial Interface. Data is shifted in at this pin on the falling edge of SCLK (see the Serial Interface
section).
23 SCLK Serial Clock Input for the Synchronous Serial Interface. All serial data transfers are synchronized to this clock
(see the Serial Interface section). The SCLK has a Schmidt-trigger input for use with a clock source that has a slow
edge transition time, for example, opto-isolator outputs.
24 DOUT Data Output for the Serial Interface. Data is shifted out at this pin on the rising edge of SCLK. This logic output
is normally in a high impedance state, unless it is driving data onto the serial data bus (see the Serial Interface
section).
ADE7758
Rev. D | Page 11 of 72
TERMINOLOGY
Measurement Error
The error associated with the energy measurement made by the
ADE7758 is defined by
%100
–×
=
EnergyTrue
EnergyTrueADE7758byRegisteredEnergy
Erro
r
t
Measuremen
(1)
Phase Error Between Channels
The high-pass filter (HPF) and digital integrator introduce a
slight phase mismatch between the current and the voltage
channel. The all-digital design ensures that the phase matching
between the current channels and voltage channels in all three
phases is within ±0.1° over a range of 45 Hz to 65 Hz and ±0.2°
over a range of 40 Hz to 1 kHz. This internal phase mismatch
can be combined with the external phase error (from current
sensor or component tolerance) and calibrated with the phase
calibration registers.
Power Supply Rejection (PSR)
This quantifies the ADE7758 measurement error as a
percentage of reading when the power supplies are varied. For
the ac PSR measurement, a reading at nominal supplies (5 V) is
taken. A second reading is obtained with the same input signal
levels when an ac signal (175 mV rms/100 Hz) is introduced
onto the supplies. Any error introduced by this ac signal is
expressed as a percentage of reading—see the Measurement
Error definition.
For the dc PSR measurement, a reading at nominal supplies
(5 V) is taken. A second reading is obtained with the same input
signal levels when the power supplies are varied ±5%. Any error
introduced is again expressed as a percentage of the reading.
ADC Offset Error
This refers to the dc offset associated with the analog inputs to
the ADCs. It means that with the analog inputs connected to
AGND that the ADCs still see a dc analog input signal. The
magnitude of the offset depends on the gain and input range
selection (see the Typical Performance Characteristics section).
However, when HPFs are switched on, the offset is removed
from the current channels and the power calculation is not
affected by this offset.
Gain Error
The gain error in the ADCs of the ADE7758 is defined as the
difference between the measured ADC output code (minus the
offset) and the ideal output code (see the Current Channel ADC
section and the Voltage Channel ADC section). The difference
is expressed as a percentage of the ideal code.
Gain Error Match
The gain error match is defined as the gain error (minus the
offset) obtained when switching between a gain of 1, 2, or 4. It is
expressed as a percentage of the output ADC code obtained
under a gain of 1.
ADE7758
Rev. D | Page 12 of 72
TYPICAL PERFORMANCE CHARACTERISTICS
0.5
–0.5
–0.4
–0.3
–0.2
–0.1
0
0.1
0.2
0.3
0.4
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
+25°C
PF = 1
+85°C
–40°C
04443-006
Figure 6. Active Energy Error as a Percentage of Reading (Gain = +1) over
Temperature with Internal Reference and Integrator Off
0.3
–0.3
–0.2
–0.1
0
0.1
0.2
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = +1, +25°C
PF = +0.5, +25°C
PF = –0.5, +25°C
PF = +0.5, +85°C
PF = +0.5, –40°C
04443-007
Figure 7. Active Energy Error as a Percentage of Reading (Gain = +1) over
Power Factor with Internal Reference and Integrator Off
0.3
–0.3
–0.2
–0.1
0
0.1
0.2
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
GAIN = +2
GAIN = +4
PF = 1
GAIN = +1
04443-008
Figure 8. Active Energy Error as a Percentage of Reading over
Gain with Internal Reference and Integrator Off
0.20
–0.20
–0.15
–0.10
–0.05
0
0.05
0.10
0.15
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = –0.5, +25°C
PF = +0.5, +25°C
PF = +0.5, –40°C
PF = +0.5, +85°C
04443-009
Figure 9. Active Energy Error as a Percentage of Reading (Gain = +1) over
Temperature with External Reference and Integrator Off
0.5
0.6
–0.2
–0.3
–0.4
–0.1
0
0.1
0.2
0.3
0.4
45 47 49 51 53 55 57 59 61 63 65
LINE FREQUENCY (Hz)
PERCENT ERROR (%)
WITH RESPECT TO 55Hz
PF = 1
PF = 0.5
04443-010
Figure 10. Active Energy Error as a Percentage of Reading (Gain = +1) over
Frequency with Internal Reference and Integrator Off
0.08
0.10
–0.06
–0.08
–0.10
–0.04
–0.02
0
0.02
0.04
0.06
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
WITH RESPECT TO 5V; 3A
V
DD
=5V
V
DD
=5.25V
V
DD
=4.75V
PF = 1
04443-011
Figure 11. Active Energy Error as a Percentage of Reading (Gain = +1) over
Power Supply with Internal Reference and Integrator Off
ADE7758
Rev. D | Page 13 of 72
0.20
0.25
–0.15
–0.20
–0.25
–0.10
–0.05
0
0.05
0.10
0.15
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PHASE A
PHASE B PHASE C
ALL PHASES
PF = 1
04443-012
Figure 12. APCF Error as a Percentage of Reading (Gain = +1)
with Internal Reference and Integrator Off
0.4
–0.4
–0.3
–0.2
–0.1
0
0.1
0.2
0.3
0.01 0.1 1 10 100
PF = 0, +25
°
C
PF = 0, +85
°
C
PF = 0, –40
°
C
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
04443-013
Figure 13. Reactive Energy Error as a Percentage of Reading (Gain = +1) over
Temperature with Internal Reference and Integrator Off
0.8
–0.8
–0.6
–0.4
–0.2
0
0.2
0.4
0.6
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = 0, +25
°
C
PF = –0.866, +25
°
C
PF = +0.866, +25
°
C
PF = +0.866, +85
°
C
PF = +0.866, –40
°
C
04443-014
Figure 14. Reactive Energy Error as a Percentage of Reading (Gain = +1) over
Power Factor with Internal Reference and Integrator Off
0.3
–0.3
–0.2
–0.1
0
0.1
0.2
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = 0, +25
°
C
PF = 0, +85
°
C
PF = 0, –40
°
C
04443-015
Figure 15. Reactive Energy Error as a Percentage of Reading (Gain = +1) over
Temperature with External Reference and Integrator Off
0.3
–0.3
–0.2
–0.1
0
0.1
0.2
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = 0, +25
°
C
PF = –0.866, +25
°
C
PF = +0.866, +25
°
C
PF = +0.866, +85
°
C
PF = +0.866, –40
°
C
04443-016
Figure 16. Reactive Energy Error as a Percentage of Reading (Gain = +1) over
Power Factor with External Reference and Integrator Off
0.8
–0.8
–0.6
–0.4
–0.2
0
0.2
0.4
0.6
45 47 49 51 53 55 57 59 61 63 65
LINE FREQUENCY (Hz)
PERCENT ERROR (%)
WITH RESPECT TO 55Hz
PF = 0
PF = 0.866
04443-017
Figure 17. Reactive Energy Error as a Percentage of Reading (Gain = +1) over
Frequency with Internal Reference and Integrator Off
ADE7758
Rev. D | Page 14 of 72
0.10
–0.10
–0.08
–0.06
–0.04
–0.02
0
0.02
0.04
0.06
0.08
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
WITH RESPECT TO 5V; 3A
5V
5.25V
4.75V
04443-018
Figure 18. Reactive Energy Error as a Percentage of Reading (Gain = +1) over
Supply with Internal Reference and Integrator Off
0.3
–0.3
–0.2
–0.1
0
0.1
0.2
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
GAIN = +1
GAIN = +2
GAIN = +4
PF = 0
04443-019
Figure 19. Reactive Energy Error as a Percentage of Reading over Gain with
Internal Reference and Integrator Off
0.4
–0.4
–0.2
–0.3
–0.1
0
0.1
0.2
0.3
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PHASE A
ALL PHASES
PHASE C
PHASE B
PF = 1
04443-020
Figure 20. VARCF Error as a Percentage of Reading (Gain = +1)
with Internal Reference and Integrator Off
0.3
–0.3
–0.2
–0.1
0
0.1
0.2
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
+25
°
C
+85
°
C
–40
°
C
04443-021
Figure 21. Active Energy Error as a Percentage of Reading (Gain = +4) over
Temperature with Internal Reference and Integrator On
0.5
0.4
–0.5
–0.4
–0.2
–0.3
–0.1
0
0.1
0.2
0.3
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = +1, +25
°
CPF = –0.5, +25
°
C
PF = +0.5, +25
°
C
PF = +0.5, +85
°
C
PF = +0.5, –40
°
C
04443-022
Figure 22. Active Energy Error as a Percentage of Reading (Gain = +4) over
Power Factor with Internal Reference and Integrator On
0.8
–0.8
–0.4
–0.6
–0.2
0
0.2
0.4
0.6
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = 0, +25
°
C
PF = +0.866, +25
°
CPF = –0.866, +25
°
C
PF = –0.866, +85
°
C
PF = –0.866, –40
°
C
04443-023
Figure 23. Reactive Energy Error as a Percentage of Reading (Gain = +4) over
Power Factor with Internal Reference and Integrator On
ADE7758
Rev. D | Page 15 of 72
0.4
–0.5
–0.4
–0.2
–0.3
–0.1
0
0.1
0.2
0.3
0.01 0.1 1 10 100
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
+25
°
C
+85
°
C
–40
°
C
PF = 0
04443-024
Figure 24. Reactive Energy Error as a Percentage of Reading (Gain = +4) over
Temperature with Internal Reference and Integrator On
0.5
0.4
–0.5
–0.4
–0.2
–0.3
–0.1
0
0.1
0.2
0.3
45 47 49 51 53 55 57 59 61 63 65
LINE FREQUENCY (Hz)
PERCENT ERROR (%)
PF = 1
PF = 0.5
04443-025
Figure 25. Active Energy Error as a Percentage of Reading (Gain = +4) over
Frequency with Internal Reference and Integrator On
1.2
1.0
–0.8
–0.6
–0.2
–0.4
0
0.2
0.4
0.6
0.8
45 47 49 51 53 55 57 59 61 63 65
LINE FREQUENCY (Hz)
PERCENT ERROR (%)
PF = 0.866
PF = 0
04443-026
Figure 26. Reactive Energy Error as a Percentage of Reading (Gain = +4) over
Frequency with Internal Reference and Integrator On
0.8
0.6
–1.2
–1.0
–0.6
–0.8
–0.4
–0.2
0
0.2
0.4
0.01 0.1 1 10 100
PF = 0.5
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
PF = 1
04443-027
Figure 27. IRMS Error as a Percentage of Reading (Gain = +1)
with Internal Reference and Integrator Off
0.8
0.6
–1.0
–0.6
–0.8
–0.4
–0.2
0
0.2
0.4
0.1 1 10 100
PF = +1
PF = –0.5
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
04443-028
Figure 28. IRMS Error as a Percentage of Reading (Gain = +4)
with Internal Reference and Integrator On
0.4
–0.4
–0.3
–0.2
–0.1
0
0.1
0.2
0.3
1 10 100
VOLTAGE (V)
PERCENT ERROR (%)
04443-029
Figure 29. VRMS Error as a Percentage of Reading (Gain = +1)
with Internal Reference
ADE7758
Rev. D | Page 16 of 72
1.5
–1.5
–1.0
–0.5
0
0.5
1.0
0.01 1 100.1 100
+25
°
C
+85
°
C
–40
°
C
PERCENT FULL-SCALE CURRENT (%)
PERCENT ERROR (%)
04443-030
–2024681012
18
21
15
12
9
6
3
0
CH 1 PhB OFFSET (mV)
HITS
MEAN: 6.5149
SD: 2.816
04443-032
Figure 30. Apparent Energy Error as a Percentage of Reading
(Gain = +1) over Temperature with Internal Reference and Integrator Off Figure 32. Phase B Channel 1 Offset Distribution
246810 1412
12
10
8
6
4
2
0
CH 1 PhC OFFSET (mV)
HITS
MEAN: 6.69333
SD: 2.70443
04443-033
–4–2024681012
18
15
12
9
6
3
0
CH 1 PhA OFFSET (mV)
HITS
MEAN: 5.55393
SD: 3.2985
04443-031
Figure 33. Phase C Channel 1 Offset Distribution
Figure 31. Phase A Channel 1 Offset Distribution
ADE7758
Rev. D | Page 17 of 72
TEST CIRCUITS
REF
IN/OUT
33nF
1kΩ
100nF
33nF
1kΩ
10µF
V
DD
VN
IAN
IBP
IBN
ICP
ICN
VAP
AVDD DVDD
VBP
VCP AGND DGND
DOUT
SCLK
APCF
CLKOUT
CLKIN
CS
DIN
IRQ
10MHz
22pF
22pF
PS2501-1
13 11 2
1
TO FRE Q.
COUNTER
14
23
20
IAP
RB
SAME AS
I
AP
, I
AN
9
8
7
10
16
15
14
100nF 10µF
33nF
1kΩ
1MΩ
220
V
33nF
1kΩ
825Ω
I
TO SPI BUS
34
19
5
6
24
23
21
22
18
12
SAME AS
I
AP
, I
AN
SAME AS V
AP
SAME AS V
AP
ADE7758
CURRENT
TRANSFORMER
17
VARCF
CT T URN RATIO 1800:1
CHANNEL 2 GAIN = + 1
CHANNEL 1 GAIN R
B
110Ω
25Ω
42.5Ω
81.25Ω
04443-034
Figure 34. Test Circuit for Integrator Off
REFIN/OUT
33nF
1kΩ
33nF
1kΩ
33nF
1kΩ
33nF
1kΩ
100nF
10µF
V
DD
VN
IAN
IBP
IBN
ICP
ICN
VAP
AVDD DVDD
VBP
VCP AGND DGND
DOUT
SCLK
APCF
CLKOUT
CLKIN
CS
DIN
IRQ
10MHz
22pF
22pF
PS2501-1
13 11 2
1
TO FRE Q.
COUNTER
14
23
20
IAP
SAME AS
IAP, IAN
9
8
7
10
16
15
14 100nF 10µF
33nF
1kΩ
1MΩ
220
V
33nF
1kΩ
825Ω
TO SPI BUS
34
19
5
6
24
23
21
22
18
12
SAME AS
IAP, IAN
SAME AS VAP
SAME AS VAP
ADE7758
I
di/dt SENSOR 17
VARCF
CHANNEL 1 GAIN = + 8
CHANNEL 2 GAIN = + 1
04443-035
Figure 35. Test Circuit for Integrator On
ADE7758
Rev. D | Page 18 of 72
THEORY OF OPERATION
ANTIALIASING FILTER
This filter prevents aliasing, which is an artifact of all sampled
systems. Input signals with frequency components higher than
half the ADC sampling rate distort the sampled signal at a fre-
quency below half the sampling rate. This happens with all ADCs,
regardless of the architecture. The combination of the high
sampling rate ∑-∆ ADC used in the ADE7758 with the relatively
low bandwidth of the energy meter allows a very simple low-
pass filter (LPF) to be used as an antialiasing filter. A simple RC
filter (single pole) with a corner frequency of 10 kHz produces
an attenuation of approximately 40 dB at 833 kHz. This is usually
sufficient to eliminate the effects of aliasing.
ANALOG INPUTS
The ADE7758 has six analog inputs divided into two channels:
current and voltage. The current channel consists of three pairs
of fully differential voltage inputs: IAP and IAN, IBP and IBN,
and ICP and ICN. These fully differential voltage input pairs
have a maximum differential signal of ±0.5 V. The current
channel has a programmable gain amplifier (PGA) with possible
gain selection of 1, 2, or 4. In addition to the PGA, the current
channels also have a full-scale input range selection for the ADC.
The ADC analog input range selection is also made using the
gain register (see Figure 38). As mentioned previously, the
maximum differential input voltage is ±0.5 V. However, by
using Bit 3 and Bit 4 in the gain register, the maximum ADC
input voltage can be set to ±0.5 V, ±0.25 V, or ±0.125 V on the
current channels. This is achieved by adjusting the ADC reference
(see the Reference Circuit section).
Figure 36 shows the maximum signal levels on the current
channel inputs. The maximum common-mode signal is
±25 mV, as shown in Figure 37.
DIFFERENTIAL INPUT
V
1
+ V
2
= 500mV MAX PEAK
+500mV
V
CM
V
1
IAP, IBP,
OR ICP
V
CM
–500mV
COMMON-MODE
±25mV MAX
V
1 +
V
2
V2IAN, I BN,
OR ICN
04443-036
Figure 36. Maximum Signal Levels, Current Channels, Gain = 1
The voltage channel has three single-ended voltage inputs: VAP,
VBP, and VCP. These single-ended voltage inputs have a
maximum input voltage of ±0.5 V with respect to VN. Both the
current and voltage channel have a PGA with possible gain
selections of 1, 2, or 4. The same gain is applied to all the inputs
of each channel.
Figure 37 shows the maximum signal levels on the voltage
channel inputs. The maximum common-mode signal is
±25 mV, as shown in Figure 36.
SINGLE - E NDE D INPUT
±500mV MAX PEAK
+500m
V
AGND
V
CM
V2
VAP, VB P,
OR VCP
V
CM
–500mV
COMMON-MODE
±25mV M AX
V
N
V
2
04443-037
Figure 37. Maximum Signal Levels, Voltage Channels, Gain = 1
The gain selections are made by writing to the gain register.
Bit 0 to Bit 1 select the gain for the PGA in the fully differential
current channel. The gain selection for the PGA in the single-
ended voltage channel is made via Bit 5 to Bit 6. Figure 38
shows how a gain selection for the current channel is made
using the gain register.
IAP, IBP, ICP
IAN, IBN, ICN
V
IN
K × V
IN
GAIN[7:0]
GAIN (K)
SELECTION
04443-038
Figure 38. PGA in Current Channel
Figure 39 shows how the gain settings in PGA 1 (current
channel) and PGA 2 (voltage channel) are selected by various
bits in the gain register.
GAIN REGISTER
1
CURRENT AND VOLTAGE CHANNE L PGA CONTRO L
7 6 5 4 3 2 1 0
0 0 0 0 0 0 0 0 ADDRESS: 0x23
RESERVED
1
REGI STE R CONTE NT S SHO W POWER- ON DEFAULTS
PGA 2 GAIN SELECT
00 = ×1
01 = ×2
10 = ×4
INT EG RATO R ENABLE
0 = DI S ABL E
1 = ENABL E
PGA 1 GAIN SELECT
00 = ×1
01 = ×2
10 = ×4
CURRENT I NP UT F UL L-S CALE SE LECT
00 = 0. 5V
01 = 0. 25V
10 = 0. 125 V
04443-039
Figure 39. Analog Gain Register
ADE7758
Rev. D | Page 19 of 72
Bit 7 of the gain register is used to enable the digital integrator
in the current signal path. Setting this bit activates the digital
integrator (see the DI/DT Current Sensor and Digital Integrator
section).
CURRENT CHANNEL ADC
Figure 41 shows the ADC and signal processing path for the
input IA of the current channels (same for IB and IC). In
waveform sampling mode, the ADC outputs are signed twos
complement 24-bit data-words at a maximum of 26.0 kSPS
(thousand samples per second). With the specified full-scale
analog input signal of ±0.5 V, the ADC produces its maximum
output code value (see Figure 41). This diagram shows a full-
scale voltage signal being applied to the differential inputs IAP
and IAN. The ADC output swings between 0xD7AE14
(−2,642,412) and 0x2851EC (+2,642,412).
Current Waveform Gain Registers
There is a multiplier in the signal path in the current channel
for each phase. The current waveform can be changed by ±50%
by writing a twos complement number to the 12-bit signed
current waveform gain registers (AIGAIN[11:0], BIGAIN[11:0],
and CIGAIN[11:0]). For example, if 0x7FF is written to those
registers, the ADC output is scaled up by +50%. To scale the
input by –50%, write 0x800 to the registers. Equation 2
describes mathematically the function of the current waveform
gain registers.
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+×
=
12
2
1gisterReGainCurrentofContent
OutputADC
WaveformCurrent
(2)
Changing the content of AIGAIN[11:0], BIGAIN[11:0], or
CIGAIN[11:0] affects all calculations based on its current; that
is, it affects the phase’s active/reactive/apparent energy as well as
its current rms calculation. In addition, waveform samples are
also scaled accordingly. IGAIN should not be used when using
Mode 0 of CONSEL, COMPMODE[1:0].
Current Channel Sampling
The waveform samples of the current channel can be routed to
the WFORM register at fixed sampling rates by setting the
WAVSEL[2:0] bit in the WAVMODE register to 000 (binary)
(see Table 20). The phase in which the samples are routed is set
by setting the PHSEL[1:0] bits in the WAVMODE register.
Energy calculation remains uninterrupted during waveform
sampling.
When in waveform sample mode, one of four output sample
rates can be chosen by using Bit 5 and Bit 6 of the WAVMODE
register (DTRT[1:0]). The output sample rate can be 26.04 kSPS,
13.02 kSPS, 6.51 kSPS, or 3.25 kSPS. By setting the WFSM bit in
the interrupt mask register to Logic 1, the interrupt request
output IRQ goes active low when a sample is available. The
timing is shown in . The 24-bit waveform samples are
transferred from the ADE7758 one byte (8-bits) at a time, with
the most significant byte shifted out first.
Figure 40
READ FROM WAVEFORM
0
SGN
CURRENT CHANNEL DATA –24 BI T S
0x12
SCLK
DIN
DOUT
IRQ
0
4443-040
Figure 40. Current Channel Waveform Sampling
The interrupt request output IRQ stays low until the interrupt
routine reads the reset status register (see the section). Interrupts
DIGITAL
INTEGRATOR1
GAIN[7]
ADC
REFERENCE
AIGAIN[11:0]
ACTIVE AND REACTIVE
POWER CALCULATION
WAVEFORM SAMPLE
REGISTER
CURRENT RMS (IRMS)
CALCULATION
IAP
IAN
PGA1
VIN
GAIN[4:3]
2.42V, 1.21V, 0.6V
GAIN[1:0]
×1, ×2, ×4
ANALOG
INPUT
RANGE
VIN
0V
0.5V/GAIN
0.25V/GAIN
0.125V/GAIN
ADC OUTPUT
WORD RANGE
CHANNEL 1
(CURRENT WAVEFORM)
DATA RANGE
0xD7AE14
0x000000
0x2851EC
50Hz CHANNEL 1 (CURRENT WAVEFORM)
DATA RANGE AF TER I NT E GRATOR
(50Hz AND AIGAIN[11:0] = 0x000)
0xCB2E48
0x000000
0x34D1B8
60Hz CHANNEL 1 (CURRENT WAVEFORM)
DATA RANGE AF TER I NT E GRATOR
(60Hz AND AIGAIN[11:0] = 0x000)
0xD4176D
0x000000
0x2BE893
HPF
04443-041
1WHEN DIGITAL I NTEGRATOR IS ENABLED, FULL-SCALE OUTP UT DATA IS
ATTENUATED DEP E NDI NG ON T HE SIGNAL FREQUENCY BECAUSE THE
INTEGRATOR HAS A –20dB/DECADE F REQUENCY RESPONSE. W HEN DISABLED,
THE O UTPUT WILL NO T BE FURTHER ATTENUAT E D.
Figure 41. Current Channel Signal Path
ADE7758
Rev. D | Page 20 of 72
di/dt CURRENT SENSOR AND DIGITAL INTEGRATOR
The di/dt sensor detects changes in the magnetic field caused by
the ac current. Figure 42 shows the principle of a di/dt current
sensor.
MAGNETIC FIELD CREATED BY CURRENT
(DIRECTLY PROPORTIONAL TO CURRENT)
+ EMF (ELECTROMOTIVE FORCE)
– INDUCED BY CHANGES IN
MAGNETIC FLUX DENSITY (di/dt)
04443-042
Figure 42. Principle of a di/dt Current Sensor
The flux density of a magnetic field induced by a current is
directly proportional to the magnitude of the current. The
changes in the magnetic flux density passing through a conductor
loop generate an electromotive force (EMF) between the two
ends of the loop. The EMF is a voltage signal that is propor-
tional to the di/dt of the current. The voltage output from the
di/dt current sensor is determined by the mutual inductance
between the current carrying conductor and the di/dt sensor.
The current signal needs to be recovered from the di/dt signal
before it can be used. An integrator is therefore necessary to
restore the signal to its original form. The ADE7758 has a built-
in digital integrator to recover the current signal from the di/dt
sensor. The digital integrator on Channel 1 is disabled by default
when the ADE7758 is powered up. Setting the MSB of the
GAIN[7:0] register turns on the integrator. Figure 43 to Figure 46
show the magnitude and phase response of the digital
integrator.
10 100 1k 10k
20
–50
–40
–30
–20
–10
0
10
FREQUENCY (Hz)
GAIN (dB)
04443-043
Figure 43. Combined Gain Response of the
Digital Integrator and Phase Compensator
10 100 1k 10k
80
91
90
89
88
87
86
85
84
83
82
81
FREQUENCY (Hz)
PHASE (Degrees)
04443-044
Figure 44. Combined Phase Response of the
Digital Integrator and Phase Compensator
40 706560555045
5
–1
0
1
2
3
4
FREQUENCY (Hz)
MAGNITUDE (dB)
04443-045
Figure 45. Combined Gain Response of the
Digital Integrator and Phase Compensator (40 Hz to 70 Hz)
40 706560555045
89.80
90.10
90.05
90.00
89.95
89.90
89.85
FREQUENCY (Hz)
PHASE (Degrees)
04443-046
Figure 46. Combined Phase Response of the
Digital Integrator and Phase Compensator (40 Hz to 70 Hz)
ADE7758
Rev. D | Page 21 of 72
Note that the integrator has a −20 dB/dec attenuation and
approximately −90° phase shift. When combined with a di/dt
sensor, the resulting magnitude and phase response should be a
flat gain over the frequency band of interest. However, the di/dt
sensor has a 20 dB/dec gain associated with it and generates
significant high frequency noise. A more effective antialiasing
filter is needed to avoid noise due to aliasing (see the Theory of
Operation section).
Note that the number of half-line cycles is based on counting
the zero crossing of the voltage channel. The ZXSEL[2:0] bits in
the LCYCMODE register determine which voltage channels are
used for the zero-crossing detection. The same signal is also
used for line cycle energy accumulation mode if activated (see
the Line Cycle Accumulation Mode Register (0x17) section).
OVERCURRENT DETECTION INTERRUPT
Figure 48 illustrates the behavior of the overcurrent detection.
When the digital integrator is switched off, the ADE7758 can be
used directly with a conventional current sensor, such as a
current transformer (CT) or a low resistance current shunt.
IPINTLVL[7:0]
READ RSTATUS
REGISTER
PKI INTERRUPT FLAG
(BIT 15 OF STATUS
REGISTER)
PKI RESET LOW
WHEN RSTATUS
REGISTER IS READ
CURRENT PEAK WAVEFORM BEING MONITORED
(SELECTED BY PKIRQSEL[2:0] IN MMODE REGISTER)
04443-048
PEAK CURRENT DETECTION
The ADE7758 can be programmed to record the peak of the
current waveform and produce an interrupt if the current
exceeds a preset limit.
Peak Current Detection Using the PEAK Register
The peak absolute value of the current waveform within a fixed
number of half-line cycles is stored in the IPEAK register.
Figure 47 illustrates the timing behavior of the peak current
detection.
L2
L1
CONTENT OF
IPEAK[7:0] 00 L1 L2 L1
NO. OF HALF
LINE CYCLES
SPECIFIED BY
LINECYC[15:0]
REGISTER
CURRENT WAVEFORM
(PHASE SELECTED BY
PEAKSEL[2:0] IN
MMODE REGISTER)
04443-047
Figure 48. ADE7758 Overcurrent Detection
Note that the content of the IPINTLVL[7:0] register is
equivalent to Bit 14 to Bit 21 of the current waveform sample.
Therefore, setting this register to 0xA1 represents putting peak
detection at full-scale analog input. Figure 48 shows a current
exceeding a threshold. The overcurrent event is recorded by
setting the PKI flag (Bit 15) in the interrupt status register. If the
PKI enable bit is set to Logic 1 in the interrupt mask register,
the IRQ logic output goes active low (see the
section).
Interrupts
Figure 47. Peak Current Detection Using the IPEAK Register Similar to peak level detection, multiple phases can be activated
for peak detection. If any of the active phases produce
waveform samples above the threshold, the PKI flag in the
interrupt status register is set. The phase of which overcurrent is
monitored is set by the PKIRQSEL[2:0] bits in the MMODE
register (see Table 19).
Note that the content of the IPEAK register is equivalent to
Bit 14 to Bit 21 of the current waveform sample. At full-scale
analog input, the current waveform sample is 0x2851EC. The
IPEAK at full-scale input is therefore expected to be 0xA1.
In addition, multiple phases can be activated for the peak
detection simultaneously by setting more than one of the
PEAKSEL[2:4] bits in the MMODE register to logic high. These
bits select the phase for both voltage and current peak
measurements. Note that if more than one bit is set, the VPEAK
and IPEAK registers can hold values from two different phases,
that is, the voltage and current peak are independently
processed (see the Peak Current Detection section).
ADE7758
Rev. D | Page 22 of 72
ADC
TO VOLTAGE RMS
CALCUL ATION AND
WAVEFORM SAMPLING
TO ACTI VE AND
REACTI V E ENE RG Y
CALCULATION
VAP +
–
VN
PGA
VA
GAIN[6:5]
×1, ×2, ×4
LPF OUTPUT
WO RD RANG E
0xD869
0x0
0x2797
50Hz
LPF OUTPUT
WO RD RANG E
0xD8B8
0x0
0x2748
60Hz
0xD7AE
0x0
0x2852
PHASE
CALIBRATION
PHCAL[6:0]
Φ
ANALO G INP UT
RANGE
VA
0V
0.5V
GAIN
LPF1
f3dB
= 260Hz
04443-049
Figure 49. ADC and Signal Processing in Voltage Channel
VOLTAGE CHANNEL ADC
Figure 49 shows the ADC and signal processing chain for the
input VA in the voltage channel. The VB and VC channels have
similar processing chains.
For active and reactive energy measurements, the output of the
ADC passes to the multipliers directly and is not filtered. This
solution avoids the much larger multibit multiplier and does not
affect the accuracy of the measurement. An HPF is not
implemented on the voltage channel to remove the dc offset
because the HPF on the current channel alone should be
sufficient to eliminate error due to ADC offsets in the power
calculation. However, ADC offset in the voltage channels
produces large errors in the voltage rms calculation and affects
the accuracy of the apparent energy calculation.
Voltage Channel Sampling
The waveform samples on the voltage channels can also be
routed to the WFORM register. However, before passing to the
WFORM register, the ADC outputs pass through a single-pole,
low-pass filter (LPF1) with a cutoff frequency at 260 Hz.
Figure 50 shows the magnitude and phase response of LPF1.
This filter attenuates the signal slightly. For example, if the line
frequency is 60 Hz, the signal at the output of LPF1 is
attenuated by 3.575%. The waveform samples are 16-bit, twos
complement data ranging between 0x2748 (+10,056d) and
0xD8B8 (−10,056d). The data is sign extended to 24-bit in the
WFORM register.
()
dB225.0974.0
Hz260
Hz60
1
1
2−==
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
+
=fH (3)
0
–20
–40
–60
–80
0
–10
–20
–30
–40
10 100 1k
FREQUENCY (Hz)
PHASE (Degrees)
GAIN (dB)
(60Hz; –0.2dB)
(60Hz; –13°)
04443-050
Figure 50. Magnitude and Phase Response of LPF1
Note that LPF1 does not affect the active and reactive energy
calculation because it is only used in the waveform sampling
signal path. However, waveform samples are used for the
voltage rms calculation and the subsequent apparent energy
accumulation.
The WAVSEL[2:0] bits in the WAVMODE register should be
set to 001 (binary) to start the voltage waveform sampling. The
PHSEL[1:0] bits control the phase from which the samples are
routed. In waveform sampling mode, one of four output sample
rates can be chosen by changing Bit 5 and Bit 6 of the
WAVMODE register (see Table 20). The available output sample
rates are 26.0 kSPS, 13.5 kSPS, 6.5 kSPS, or 3.3 kSPS. By setting
the WFSM bit in the interrupt mask register to Logic 1, the
interrupt request output IRQ goes active low when a sample is
available. The 24-bit waveform samples are transferred from the
ADE7758 one byte (8 bits) at a time, with the most significant
byte shifted out first.
ADE7758
Rev. D | Page 23 of 72
The sign of the register is extended in the upper 8 bits. The
timing is the same as for the current channels, as seen in Figure 40.
ZERO-CROSSING DETECTION
The ADE7758 has zero-crossing detection circuits for each of
the voltage channels (VAN, VBN, and VCN). Figure 51 shows
how the zero-cross signal is generated from the output of the
ADC of the voltage channel.
REFERENCE
ADC ZERO-
CROSSING
DETECTOR
PGA
V
AN,
V
BN,
V
CN
GAIN[6:5]
×1, ×2, ×4
LPF1
f
–3dB
=260Hz
24.8° @ 60Hz
ANALOG VOLTAGE
WAVEFORM
(VAN, VBN, OR VCN)
LPF1
OUTPUT
READ RSTATUS
IRQ
1.0
0.908
04443-051
Figure 51. Zero-Crossing Detection on Voltage Channels
The zero-crossing interrupt is generated from the output of
LPF1. LPF1 has a single pole at 260 Hz (CLKIN = 10 MHz). As
a result, there is a phase lag between the analog input signal of
the voltage channel and the output of LPF1. The phase response
of this filter is shown in the Voltage Channel Sampling section.
The phase lag response of LPF1 results in a time delay of
approximately 1.1 ms (at 60 Hz) between the zero crossing on
the voltage inputs and the resulting zero-crossing signal. Note
that the zero-crossing signal is used for the line cycle
accumulation mode, zero-crossing interrupt, and line
period/frequency measurement.
When one phase crosses from negative to positive, the
corresponding flag in the interrupt status register (Bit 9 to
Bit 11) is set to Logic 1. An active low in the IRQ output also
appears if the corresponding ZX bit in the interrupt mask
register is set to Logic 1. Note that only zero crossing from
negative to positive generates an interrupt.
The flag in the interrupt status register is reset to 0 when the
interrupt status register with reset (RSTATUS) is read. Each
phase has its own interrupt flag and mask bit in the interrupt
register.
Zero-Crossing Timeout
Each zero-crossing detection has an associated internal timeout
register (not accessible to the user). This unsigned, 16-bit
register is decreased by 1 every 384/CLKIN seconds. The
registers are reset to a common user-programmed value, that is,
the zero-crossing timeout register (ZXTOUT[15:0], Address 0x1B),
every time a zero crossing is detected on its associated input.
The default value of ZXTOUT is 0xFFFF. If the internal register
decrements to 0 before a zero crossing at the corresponding
input is detected, it indicates an absence of a zero crossing in
the time determined by the ZXTOUT[15:0]. The ZXTOx
detection bit of the corresponding phase in the interrupt status
register is then switched on (Bit 6 to Bit 8). An active low on the
IRQ output also appears if the ZXTOx mask bit for the
corresponding phase in the interrupt mask register is set to
Logic 1. shows the mechanism of the zero-crossing
timeout detection when the Line Voltage A stays at a fixed dc
level for more than 384/CLKIN × ZXTOUT[15:0] seconds.
Figure 52
ZXTOA
DETECTION BIT
READ
RSTATUS
VOLTAGE
CHANNEL A
ZXTOUT[15:0]
16-BIT INTERNAL
REGISTER VALUE
04443-052
Figure 52. Zero-Crossing Timeout Detection
PHASE COMPENSATION
When the HPF in the current channel is disabled, the phase
error between the current channel (IA, IB, or IC) and the
corresponding voltage channel (VA, VB, or VC) is negligible.
When the HPF is enabled, the current channels have phase
response (see Figure 53 through Figure 55). The phase response
is almost 0 from 45 Hz to 1 kHz. The frequency band is sufficient
for the requirements of typical energy measurement applications.
However, despite being internally phase compensated, the
ADE7758 must work with transducers that may have inherent
phase errors. For example, a current transformer (CT) with a
phase error of 0.1° to 0.3° is not uncommon. These phase errors
can vary from part to part, and they must be corrected to
perform accurate power calculations.
ADE7758
Rev. D | Page 24 of 72
40 706560555045
0.20
0.15
0.10
0.05
0
–0.05
–0.10
FREQUENCY (Hz)
PHASE (Degrees)
04443-054
The errors associated with phase mismatch are particularly
noticeable at low power factors. The ADE7758 provides a
means of digitally calibrating these small phase errors. The
ADE7758 allows a small time delay or time advance to be
introduced into the signal processing chain to compensate for
the small phase errors.
The phase calibration registers (APHCAL, BPHCAL, and
CPHCAL) are twos complement, 7-bit sign-extended registers
that can vary the time advance in the voltage channel signal
path from +153.6 µs to −75.6 µs (CLKIN = 10 MHz),
respectively. Negative values written to the PHCAL registers
represent a time advance, and positive values represent a time
delay. One LSB is equivalent to 1.2 µs of time delay or 2.4 µs of
time advance with a CLKIN of 10 MHz. With a line frequency
of 60 Hz, this gives a phase resolution of 0.026° (360° × 1.2 µs ×
60 Hz) at the fundamental in the positive direction (delay) and
0.052° in the negative direction (advance). This corresponds to
a total correction range of −3.32° to +1.63° at 60 Hz.
Figure 54. Phase Response of the HPF and Phase Compensation
(40 Hz to 70 Hz)
44 565452504846
0.10
0.08
0.06
0.04
0.02
0
–0.02
FREQUENCY (Hz)
PHASE (Degrees)
04443-055
Figure 56 illustrates how the phase compensation is used to
remove a 0.1° phase lead in IA of the current channel from the
external current transducer. To cancel the lead (0.1°) in the
current channel of Phase A, a phase lead must be introduced
into the corresponding voltage channel. The resolution of the
phase adjustment allows the introduction of a phase lead of
0.104°. The phase lead is achieved by introducing a time
advance into VA. A time advance of 4.8 µs is made by writing
−2 (0x7E) to the time delay block (APHCAL[6:0]), thus
reducing the amount of time delay by 4.8 µs or equivalently,
360° × 4.8 µs × 60 Hz = 0.104° at 60 Hz. Figure 55. Phase Response of HPF and Phase Compensation
(44 Hz to 56 Hz)
0 100 200 300 400 500 600 700 800 1k900
90
0
10
20
30
40
50
60
70
80
FREQUENCY (Hz)
PHASE (Degrees)
04443-053
Figure 53. Phase Response of the HPF and Phase Compensation
(10 Hz to 1 kHz)
ADE7758
Rev. D | Page 25 of 72
PGA1
IAP
IAN
IA ADC
HPF
PGA2
VAP
VN
VA ADC
60Hz
0.1°
IA
VA
RANGE OF PHAS E
CALIBRATION
1111100
60
APHCAL[6:0]
–153.6µs TO + 75. 6µ s
VA VA ADVANCED BY 4.8µs
(+0.104
°
@ 60Hz)
0x7E
IA
60Hz
DIGITAL
INTEGRATOR
ACTIVE AND
REACTIVE
ENERGY
CALCULATION
+1.36 °, –2.76° @ 50Hz ; 0.022 °, 0.043°
+1.63 °, –3.31° @ 60Hz ; 0.026 °, 0.052°
04443-056
Figure 56. Phase Calibration on Voltage Channels
PERIOD MEASUREMENT
The ADE7758 provides the period or frequency measurement
of the line voltage. The period is measured on the phase
specified by Bit 0 to Bit 1 of the MMODE register. The period
register is an unsigned 12-bit FREQ register and is updated
every four periods of the selected phase.
Bit 7 of the LCYCMODE selects whether the period register
displays the frequency or the period. Setting this bit causes the
register to display the period. The default setting is logic low,
which causes the register to display the frequency.
When set to measure the period, the resolution of this register is
96/CLKIN per LSB (9.6 µs/LSB when CLKIN is 10 MHz),
which represents 0.06% when the line frequency is 60 Hz. At
60 Hz, the value of the period register is 1737d. At 50 Hz, the
value of the period register is 2084d. When set to measure
frequency, the value of the period register is approximately 960d at
60 Hz and 800d at 50 Hz. This is equivalent to 0.0625 Hz/LSB.
LINE VOLTAGE SAG DETECTION
The ADE7758 can be programmed to detect when the absolute
value of the line voltage of any phase drops below a certain peak
value for a number of half cycles. Each phase of the voltage
channel is controlled simultaneously. This condition is
illustrated in Figure 57.
Figure 57 shows a line voltage fall below a threshold, which is
set in the SAG level register (SAGLVL[7:0]), for nine half cycles.
Because the SAG cycle register indicates a six half-cycle threshold
(SAGCYC[7:0] = 0x06), the SAG event is recorded at the end of
the sixth half cycle by setting the SAG flag of the corresponding
phase in the interrupt status register (Bit 1 to Bit 3 in the
interrupt status register).
If the SAG enable bit is set to Logic 1 for this phase (Bit 1 to
Bit 3 in the interrupt mask register), the IRQ logic output goes
active low (see the section). The phases are compared
to the same parameters defined in the SAGLVL and SAGCYC
registers.
Interrupts
SAGLVL[7:0]
FULL-SCALE
READ RSTATUS
REGISTER
SAGCYC[7:0] = 0x06
6HALFCYCLES
S
AG INTERRUPT FLAG
(BIT 3 TO BIT 5 OF
STATUS REGISTER)
VAP, VBP, OR VCP
SAG EVENT RESET LOW
WHEN VOLTAGE CHANNEL
EXCEEDS SAGLVL[7:0]
04443-057
Figure 57. ADE7758 SAG Detection
Figure 57 shows a line voltage fall below a threshold, which is
set in the SAG level register (SAGLVL[7:0]), for nine half cycles.
Because the SAG cycle register indicates a six half-cycle threshold
(SAGCYC[7:0] = 0x06), the SAG event is recorded at the end of
the sixth half cycle by setting the SAG flag of the corresponding
phase in the interrupt status register (Bit 1 to Bit 3 in the
interrupt status register). If the SAG enable bit is set to Logic 1
for this phase (Bit 1 to Bit 3 in the interrupt mask register), the
IRQ logic output goes active low (see the section).
The phases are compared to the same parameters defined in the
SAGLVL and SAGCYC registers.
Interrupts
ADE7758
Rev. D | Page 26 of 72
SAG LEVEL SET
The contents of the single-byte SAG level register, SAGLVL[0:7],
are compared to the absolute value of Bit 6 to Bit 13 from the
voltage waveform samples. For example, the nominal maximum
code of the voltage channel waveform samples with a full-scale
signal input at 60 Hz is 0x2748 (see the Voltage Channel Sampling
section). Bit 13 to Bit 6 are 0x9D. Therefore, writing 0x9D to the
SAG level register puts the SAG detection level at full scale and
sets the SAG detection to its most sensitive value.
The detection is made when the content of the SAGLVL[7:0]
register is greater than the incoming sample. Writing 0x00 puts
the SAG detection level at 0. The detection of a decrease of an
input voltage is disabled in this case.
PEAK VOLTAGE DETECTION
The ADE7758 can record the peak of the voltage waveform and
produce an interrupt if the current exceeds a preset limit.
Peak Voltage Detection Using the VPEAK Register
The peak absolute value of the voltage waveform within a fixed
number of half-line cycles is stored in the VPEAK register.
Figure 58 illustrates the timing behavior of the peak voltage
detection.
L2
L1
CONTENT OF
VPEAK[7:0] 00 L1 L2 L1
NO. OF HALF
LINE CYCLES
SPECIFIED BY
LINECYC[15:0]
REGISTER
VOLTAGE WAVEFORM
(PHASE SELECTED BY
PEAKSEL[2:4]
IN MMODE REGISTER)
04443-058
Figure 58. Peak Voltage Detection Using the VPEAK Register
Note that the content of the VPEAK register is equivalent to
Bit 6 to Bit 13 of the 16-bit voltage waveform sample. At full-
scale analog input, the voltage waveform sample at 60 Hz is
0x2748. The VPEAK at full-scale input is therefore expected to
be 0x9D.
In addition, multiple phases can be activated for the peak
detection simultaneously by setting multiple bits among the
PEAKSEL[2:4] bits in the MMODE register. These bits select
the phase for both voltage and current peak measurements.
Note that if more than one bit is set, the VPEAK and IPEAK
registers can hold values from two different phases, that is, the
voltage and current peak are independently processed (see the
Peak Current Detection section).
Note that the number of half-line cycles is based on counting
the zero crossing of the voltage channel. The ZXSEL[2:0] bits in
the LCYCMODE register determine which voltage channels are
used for the zero-crossing detection (see Table 22). The same
signal is also used for line cycle energy accumulation mode if
activated.
Overvoltage Detection Interrupt
Figure 59 illustrates the behavior of the overvoltage detection.
VPINTLVL[7:0]
READ RSTATUS
REGISTER
PKV INTERRUPT FLAG
(BIT 14 OF STATUS
REGISTER)
PKV RESET LOW
WHEN RSTATUS
REGISTER IS READ
VOLTAGE PEAK WAVEFORM BEING MONITORED
(SELECTED BY PKIRQSEL[5:7] IN MMODE REGISTER)
04443-059
Figure 59. ADE7758 Overvoltage Detection
Note that the content of the VPINTLVL[7:0] register is
equivalent to Bit 6 to Bit 13 of the 16-bit voltage waveform
samples; therefore, setting this register to 0x9D represents
putting the peak detection at full-scale analog input. Figure 59
shows a voltage exceeding a threshold. By setting the PKV flag
(Bit 14) in the interrupt status register, the overvoltage event is
recorded. If the PKV enable bit is set to Logic 1 in the interrupt
mask register, the IRQ logic output goes active low (see the
section). Interrupts
Multiple phases can be activated for peak detection. If any of the
active phases produce waveform samples above the threshold,
the PKV flag in the interrupt status register is set. The phase in
which overvoltage is monitored is set by the PKIRQSEL[5:7]
bits in the MMODE register (see Table 19).
ADE7758
Rev. D | Page 27 of 72
PHASE SEQUENCE DETECTION
The ADE7758 has an on-chip phase sequence error detection
interrupt. If the zero crossing of Phase A is not followed by
Phase C but by Phase B, the SEQERR bit (Bit 19) in the STATUS
register is set. If SEQERR is set in the mask register, the IRQ
logic output goes active low (see the section).
depicts how the interrupt is issued in two different
configurations.
Interrupts Figure 60
Note that if it is desired to have the interrupt occur when
Phase A is followed by Phase B and not Phase C, then the analog
inputs for Phase C and Phase B should be swapped. In this case,
the Phase B voltage input should be wired to the VCP pin and
the Phase C voltage input should be wired to the VBP pin.
04443-060
AB
SEQERR BIT OF STATUS REGISTER IS SET
A = 0° B = –120° C = +120°
C
AC
SEQERR BIT OF ST ATUS REGISTER IS NOT SET
A = 0° C = –120° B = +120°
B
VOLTAGE
W
AVEFORMS
ZERO
CROSSINGS
VOLTAGE
W
AVEFORMS
ZERO
CROSSINGS C A BC
B A CB A B AC
A C AB
Figure 60. Phase Sequence Detection
POWER-SUPPLY MONITOR
The ADE7758 also contains an on-chip power-supply monitor.
The analog supply (AVDD) is monitored continuously by the
ADE7758. If the supply is less than 4 V ± 5%, the ADE7758
goes into an inactive state, that is, no energy is accumulated
when the supply voltage is below 4 V. This is useful to ensure
correct device operation at power-up and during power-down.
The power-supply monitor has built-in hysteresis and filtering.
This gives a high degree of immunity to false triggering due to
noisy supplies. Figure 61 shows the behavior of the ADE7758
when the voltage of AVDD falls below the power-supply
monitor threshold. The power supply and decoupling for the
part should be designed such that the ripple at AVDD does not
exceed 5 V ± 5% as specified for normal operation.
AV
DD
5V
4V
0V
ADE7758
INTERNAL
C
ALCULATION
S
ACTIVE INACTIVEINACTIVE
TIME
04443-061
Figure 61. On-Chip, Power-Supply Monitoring
REFERENCE CIRCUIT
The nominal reference voltage at the REFIN/OUT pin is 2.42 V.
This is the reference voltage used for the ADCs in the
ADE7758. However, the current channels have three input
range selections (full scale is selectable among 0.5 V, 0.25 V, and
0.125 V). This is achieved by dividing the reference internally
by 1, ½, and ¼. The reference value is used for the ADC in the
current channels. Note that the full-scale selection is only
available for the current inputs.
The REFIN/OUT pin can be overdriven by an external source, for
example, an external 2.5 V reference. Note that the nominal
reference value supplied to the ADC is now 2.5 V and not
2.42 V. This has the effect of increasing the nominal analog
input signal range by 2.5/2.42 × 100% = 3% or from 0.5 V to
0.5165 V.
The voltage of the ADE7758 reference drifts slightly with
temperature; see the Specifications section for the temperature
coefficient specification (in ppm/°C). The value of the temperature
drift varies from part to part. Because the reference is used for
all ADCs, any x% drift in the reference results in a 2x%
deviation of the meter accuracy. The reference drift resulting
from temperature changes is usually very small and typically
much smaller than the drift of other components on a meter.
Alternatively, the meter can be calibrated at multiple temperatures.
ADE7758
Rev. D | Page 28 of 72
TEMPERATURE MEASUREMENT
The ADE7758 also includes an on-chip temperature sensor. A
temperature measurement is made every 4/CLKIN seconds.
The output from the temperature sensing circuit is connected to
an ADC for digitizing. The resultant code is processed and
placed in the temperature register (TEMP[7:0]). This register
can be read by the user and has an address of 0x11 (see the
Serial Interface section). The contents of the temperature
register are signed (twos complement) with a resolution of
3°C/LSB. The offset of this register may vary significantly from
part to part. To calibrate this register, the nominal value should
be measured, and the equation should be adjusted accordingly.
Temp (°C) =
[(TEMP[7:0] − Offset) × 3°C/LSB] + Ambient(°C) (4)
For example, if the temperature register produces a code of 0x46
at ambient temperature (25°C), and the temperature register
currently reads 0x50, then the temperature is 55°C :
Temp (°C) = [(0x50 – 0x46) × 3°C/LSB] + 25°C = 55°C
Depending on the nominal value of the register, some finite
temperature can cause the register to roll over. This should be
compensated for in the system master (MCU).
The ADE7758 temperature register varies with power supply. It
is recommended to use the temperature register only in
applications with a fixed, stable power supply. Typical error with
respect to power supply variation is show in Table 5.
Table 5. Temperature Register Error with Power Supply
Variation
4.5 V 4.75 V 5 V 5.25 V 5.5 V
Register Value 219 216 214 211 208
% Error +2.34 +0.93 0 −1.40 −2.80
ROOT MEAN SQUARE MEASUREMENT
Root mean square (rms) is a fundamental measurement of the
magnitude of an ac signal. Its definition can be both practical
and mathematical. Defined practically, the rms value assigned
to an ac signal is the amount of dc required to produce an
equivalent amount of power in the load. Mathematically, the
rms value of a continuous signal f(t) is defined as
()
dt
T
12
0
TtfFRMS ∫
= (5)
For time sampling signals, rms calculation involves squaring the
signal, taking the average, and obtaining the square root.
][
1
1
2nf
N
FRMS
N
n
∑
=
= (6)
The method used to calculate the rms value in the ADE7758 is
to low-pass filter the square of the input signal (LPF3) and take
the square root of the result (see Figure 62).
i(t) = √2 × IRMS × sin(ωt) (7)
then
i2(t) = IRMS2 − IRMS2 × cos(ωt) (8)
The rms calculation is simultaneously processed on the six
analog input channels. Each result is available in separate
registers.
While the ADE7758 measures nonsinusoidal signals, it should
be noted that the voltage rms measurement, and therefore the
apparent energy, are bandlimited to 260 Hz. The current rms as
well as the active power have a bandwidth of 14 kHz.
Current RMS Calculation
Figure 62 shows the detail of the signal processing chain for the
rms calculation on one of the phases of the current channel.
The current channel rms value is processed from the samples
used in the current channel waveform sampling mode. The
current rms values are stored in 24-bit registers (AIRMS,
BIRMS, and CIRMS). One LSB of the current rms register is
equivalent to one LSB of the current waveform sample. The
update rate of the current rms measurement is CLKIN/12.
SGN 225 224 223 217 216 215
CURRENT SIGNAL
FROM HPF OR
INTEGRATOR
(IF ENABLED)
0x1D3781
0x00
++
0x2851EC
0x0
0xD7AE14
X2
LPF3
AIRMS[24:0]
AIRMSOS[11:0]
04443-062
Figure 62. Current RMS Signal Processing
With the specified full-scale analog input signal of 0.5 V, the
ADC produces an output code that is approximately
±2,642,412d (see the Current Channel ADC section). The
equivalent rms value of a full-scale sinusoidal signal at 60 Hz is
1,914,753 (0x1D3781).
The accuracy of the current rms is typically 0.5% error from the
full-scale input down to 1/500 of the full-scale input. Additionally,
this measurement has a bandwidth of 14 kHz. It is recommended
to read the rms registers synchronous to the voltage zero
crossings to ensure stability. The IRQ can be used to indicate
when a zero crossing has occurred (see the Interrupts section).
Table 6 shows the settling time for the IRMS measurement,
which is the time it takes for the rms register to reflect the value
at the input to the current channel.
Table 6. Settling Time for IRMS Measurement
63% 100%
Integrator Off 80 ms 960 ms
Integrator On 40 ms 1.68 sec
ADE7758
Rev. D | Page 29 of 72
Current RMS Offset Compensation
The ADE7758 incorporates a current rms offset compensation
register for each phase (AIRMSOS, BIRMSOS, and CIRMSOS).
These are 12-bit signed registers that can be used to remove
offsets in the current rms calculations. An offset can exist in the
rms calculation due to input noises that are integrated in the dc
component of I2(t). Assuming that the maximum value from
the current rms calculation is 1,914,753d with full-scale ac
inputs (60 Hz), one LSB of the current rms offset represents
0.94% of the measurement error at 60 dB down from full scale.
The IRMS measurement is undefined at zero input. Calibration
of the offset should be done at low current and values at zero
input should be ignored. For details on how to calibrate the
current rms measurement, see the Calibration section.
IRMSOSIRMSIRMS 2
0×+= 16384 (9)
where IRMS0 is the rms measurement without offset correction.
Table 7. Approximate IRMS Register Values
Frequency (Hz) Integrator Off (d) Integrator On (d)
50 1,921,472 2,489,581
60 1,914,752 2,067,210
Voltage Channel RMS Calculation
Figure 63 shows the details of the signal path for the rms
estimation on Phase A of the voltage channel. This voltage rms
estimation is done in the ADE7758 using the mean absolute
value calculation, as shown in Figure 63.The voltage channel
rms value is processed from the waveform samples after the
low-pass filter LPF1. The output of the voltage channel ADC
can be scaled by ±50% by changing VRMSGAIN[11:0] registers
to perform an overall rms voltage calibration. The VRMSGAIN
registers scale the rms calculations as well as the apparent
energy calculation because apparent power is the product of the
voltage and current rms values. The voltage rms values are
stored in 24-bit registers (AVRMS, BVRMS, and CVRMS). One
LSB of a voltage waveform sample is approximately equivalent to
256 LSBs of the voltage rms register. The update rate of the
voltage rms measurement is CLKIN/12.
With the specified full-scale ac analog input signal of 0.5 V, the
LPF1 produces an output code that is approximately 63% of its
full-scale value, that is, ±9,372d, at 60 Hz (see the Vol tage
Channel ADC section). The equivalent rms value of a full-scale
ac signal is approximately 1,639,101 (0x1902BD) in the VRMS
register.
The accuracy of the VRMS measurement is typically 0.5% error
from the full-scale input down to 1/20 of the full-scale input.
Additionally, this measurement has a bandwidth of 260 Hz. It is
recommended to read the rms registers synchronous to the
voltage zero crossings to ensure stability. The IRQ can be used
to indicate when a zero crossing has occurred (see the
Interrupts section).
VAN
AVRMSGAIN[11:0]
0x2748
LPF OUTPUT
WORD RANGE
0x0
60Hz
0xD8B8
0x2797
LPF OUTPUT
WORD RANGE
0x0
50Hz
0xD869
LPF1
VOLT AG E SI G NAL –V(t)
0.5
GAIN
0x193504
50Hz
0x0
0x1902BD
60Hz
0x0
|
X
|
A
VRMS[23:0]
LPF3
SGN 216 215 214 282726
V
RMSOS[11:0]
+
+
04443-063
Figure 63. Voltage RMS Signal Processing
Table 8 shows the settling time for the VRMS measurement,
which is the time it takes for the rms register to reflect the value
at the input to the voltage channel.
Table 8. Settling Time for VRMS Measurement
63% 100%
100 ms 960 ms
Voltage RMS Offset Compensation
The ADE7758 incorporates a voltage rms offset compensation
for each phase (AVRMSOS, BVRMSOS, and CVRMSOS).
These are 12-bit signed registers that can be used to remove
offsets in the voltage rms calculations. An offset can exist in the
rms calculation due to input noises and offsets in the input
samples. It should be noted that the offset calibration does not
allow the contents of the VRMS registers to be maintained at 0
when no voltage is applied. This is caused by noise in the
voltage rms calculation, which limits the usable range between
full scale and 1/50th of full scale. One LSB of the voltage rms
offset is equivalent to 64 LSBs of the voltage rms register.
Assuming that the maximum value from the voltage rms
calculation is 1,639,101d with full-scale ac inputs, then 1 LSB of
the voltage rms offset represents 0.042% of the measurement
error at 1/10 of full scale.
VRMS = VRMS0 + VRMSOS × 64 (10)
where VRMS0 is the rms measurement without the offset
correction.
Table 9. Approximate VRMS Register Values
Frequency (Hz) Value (d)
50 1,678,210
60 1,665,118
ADE7758
Rev. D | Page 30 of 72
Voltage RMS Gain Adjust
The ADC gain in each phase of the voltage channel can be
adjusted for the rms calculation by using the voltage rms gain
registers (AVRMSGAIN, BVRMSGAIN, and CVRMSGAIN).
The gain of the voltage waveforms before LPF1 is adjusted by
writing twos complement, 12-bit words to the voltage rms gain
registers. Equation 11 shows how the gain adjustment is related
to the contents of the voltage gain register.
⎟
⎠
⎞
⎜
⎝
⎛+×
=
12
2
1VRMSGAIN
GainWithoutValuesRMSNominal
RegisterVRMSofContent
(11)
For example, when 0x7FF is written to the voltage gain register,
the RMS value is scaled up by 50%.
0x7FF = 2047d
2047/212 = 0.5
Similarly, when 0x800, which equals –2047d (signed twos
complement), is written the ADC output is scaled by –50%.
ACTIVE POWER CALCULATION
Electrical power is defined as the rate of energy flow from
source to load. It is given by the product of the voltage and
current waveforms. The resulting waveform is called the
instantaneous power signal and it is equal to the rate of energy
flow at every instant of time. The unit of power is the watt or
joules/sec. Equation 14 gives an expression for the instantaneous
power signal in an ac system.
v(t) = √2 × VRMS × sin(ωt) (12)
i(t) = √2 × IRMS × sin(ωt) (13)
where VRMS = rms voltage and IRMS = rms current.
p(t) = v(t) × i(t)
p(t) = IRMS × VRMS − IRMS × VRMS × cos(2ωt) (14)
The average power over an integral number of line cycles (n) is
given by the expression in Equation 15.
()
IRMSVRMSdttp
nT
p
nT ×== ∫
0
1 (15)
where:
t is the line cycle period.
P is referred to as the active or real power. Note that the active
power is equal to the dc component of the instantaneous power
signal p(t) in Equation 14, that is, VRMS × IRMS. This is the
relationship used to calculate the active power in the ADE7758
for each phase.
The instantaneous power signal p(t) is generated by multiplying
the current and voltage signals in each phase. The dc component
of the instantaneous power signal in each phase (A, B, and C) is
then extracted by LPF2 (the low-pass filter) to obtain the
average active power information on each phase. Figure 64
shows this process. The active power of each phase accumulates
in the corresponding 16-bit watt-hour register (AWATTHR,
BWATTHR, or CWATTHR). The input to each active energy
register can be changed depending on the accumulation mode
setting (see Table 22).
INSTANTANEOUS
POWER SIGNAL p(t) = VRMS × IRMS – VRMS × IRMS × cos(2ωt)
ACTIVE RE AL POWER
SIGNAL = VRMS × IRMS
0x19999A
V
RMS × IRMS
0xCCCCD
0x00000
CURRENT
i(t) = 2 × IRMS × sin(ωt)
VOLTAGE
v(t) = 2 × VRMS × sin(ωt)
04443-064
Figure 64. Active Power Calculation
Because LPF2 does not have an ideal brick wall frequency
response (see Figure 65), the active power signal has some
ripple due to the instantaneous power signal. This ripple is
sinusoidal and has a frequency equal to twice the line frequency.
Because the ripple is sinusoidal in nature, it is removed when
the active power signal is integrated over time to calculate the
energy.
0
–4
–8
–12
GAIN (dB)
–16
–20
–24 13108
FREQUENCY (Hz)
30 100
04443-065
Figure 65. Frequency Response of the LPF Used
to Filter Instantaneous Power in Each Phase
ADE7758
Rev. D | Page 31 of 72
Active Power Gain Calibration
Note that the average active power result from the LPF output
in each phase can be scaled by ±50% by writing to the phase’s
watt gain register (AWG, BWG, or CWG). The watt gain
registers are twos complement, signed registers and have a
resolution of 0.024%/LSB. Equation 16 describes
mathematically the function of the watt gain registers.
⎟
⎠
⎞
⎜
⎝
⎛+×
=
12
2
12 gisterReGainWatt
OutputLPF
DataPowerAverage
(16)
The REVPAP bit (Bit 17) in the interrupt status register is set if
the average power from any one of the phases changes sign. The
phases monitored are selected by TERMSEL bits in the
COMPMODE register (see Table 21). The TERMSEL bits are
also used to select which phases are included in the APCF and
VARCF pulse outputs. If the REVPAP bit is set in the mask
register, the IRQ logic output goes active low (see the
section). Note that this bit is set whenever there are sign
changes, that is, the REVPAP bit is set for both a positive-to-
negative change and a negative-to-positive change of the sign
bit. The response time of this bit is approximately 176 ms for a
full-scale signal, which has an average value of 0xCCCCD at the
low pass filter output. For smaller inputs, the time is longer.
Interrupts
The output is scaled by −50% by writing 0x800 to the watt gain
registers and increased by +50% by writing 0x7FF to them.
These registers can be used to calibrate the active power (or
energy) calculation in the ADE7758 for each phase. CLKINValueAverage
msTimesponseRe 4
25
2
601 ×
⎥
⎥
⎦
⎤
⎢
⎢
⎣
⎡
+≅ (17)
Active Power Offset Calibration The APCFNUM [15:13] indicate reverse power on each of the
individual phases. Bit 15 is set if the sign of the power on Phase A is
negative, Bit 14 for Phase B, and Bit 13 for Phase C.
The ADE7758 also incorporates a watt offset register on each
phase (AWATTOS, BWATTOS, and CWATTOS). These are
signed twos complement, 12-bit registers that are used to
remove offsets in the active power calculations. An offset can
exist in the power calculation due to crosstalk between channels
on the PCB or in the chip itself. The offset calibration allows the
contents of the active power register to be maintained at 0 when
no power is being consumed. One LSB in the active power
offset register is equivalent to 1/16 LSB in the active power
multiplier output. At full-scale input, if the output from the
multiplier is 0xCCCCD (838,861d), then 1 LSB in the LPF2
output is equivalent to 0.0075% of measurement error at 60 dB
down from full scale on the current channel. At −60 dB down
on full scale (the input signal level is 1/1000 of full-scale signal
inputs), the average word value from LPF2 is 838.861
(838,861/1000). One LSB is equivalent to 1/838.861/16 × 100%
= 0.0075% of the measured value. The active power offset register
has a correction resolution equal to 0.0075% at −60 dB.
No-Load Threshold
The ADE7758 has an internal no-load threshold on each phase.
The no-load threshold can be activated by setting the NOLOAD
bit (Bit 7) of the COMPMODE register. If the active power falls
below 0.005% of full-scale input, the energy is not accumulated
in that phase. As stated, the average multiplier output with full-
scale input is 0xCCCCD. Therefore, if the average multiplier
output falls below 0x2A, the power is not accumulated to avoid
creep in the meter. The no-load threshold is implemented only
on the active energy accumulation. The reactive and apparent
energies do not have the no-load threshold option.
Active Energy Calculation
As previously stated, power is defined as the rate of energy flow.
This relationship can be expressed mathematically as
dt
dEnergy
Power = (18)
Sign of Active Power Calculation
Note that the average active power is a signed calculation. If the
phase difference between the current and voltage waveform is
more than 90°, the average power becomes negative. Negative
power indicates that energy is being placed back on the grid.
The ADE7758 has a sign detection circuitry for active power
calculation.
Conversely, Energy is given as the integral of power.
()
dtp
∫
=tEnergy (19)
ADE7758
Rev. D | Page 32 of 72
AWG[11:0]
WDIV[7:0]
DIGITAL
INTEGRATOR
MULTIPLIER
I
V
HPF
CURRENT SIGNAL–i(t)
0x2851EC
0x00
0xD7AE14
VOLTAGE SIGNAL–v(t)
0x2852
000x
0xD7AE
+++
+
LPF2
%
SIGN 2
6
2
0
2
–1
2
–2
2
–3
2
–4
AWATTOS[11:0] AWATTHR[15:0]
15 0
40 0
TOTAL ACTIVE POWER IS
ACCUMULATED (INTEGRATED) IN
THE ACTIVE ENERGY REGISTER
TIME (nT)
T
AVERAGE POWER
SIGNAL–P
0xCCCCD
0x00000
PHCAL[6:0]
Φ
04443-066
Figure 66. ADE7758 Active Energy Accumulation
The ADE7758 achieves the integration of the active power
signal by continuously accumulating the active power signal in
the internal 41-bit energy registers. The watt-hr registers
(AWATTHR, BWATTHR, and CWATTHR) represent the upper
16 bits of these internal registers. This discrete time
accumulation or summation is equivalent to integration in
continuous time. Equation 20 expresses the relationship.
()
()
⎭
⎬
⎫
⎩
⎨
⎧×== ∑
∫∞
=
→0
0T
Lim
n
TnTpdttpEnergy (20)
where:
n is the discrete time sample number.
T is the sample period.
Figure 66 shows a signal path of this energy accumulation. The
average active power signal is continuously added to the internal
active energy register. This addition is a signed operation.
Negative energy is subtracted from the active energy register.
Note the values shown in Figure 66 are the nominal full-scale
values, that is, the voltage and current inputs at the corresponding
phase are at their full-scale input level. The average active power
is divided by the content of the watt divider register before it is
added to the corresponding watt-hr accumulation registers.
When the value in the WDIV[7:0] register is 0 or 1, active
power is accumulated without division. WDIV is an 8-bit
unsigned register that is useful to lengthen the time it takes
before the watt-hr accumulation registers overflow.
Figure 67 shows the energy accumulation for full-scale signals
(sinusoidal) on the analog inputs. The three displayed curves
show the minimum time it takes for the watt-hr accumulation
register to overflow when the watt gain register of the corre-
sponding phase equals to 0x7FF, 0x000, and 0x800. The watt
gain registers are used to carry out a power calibration in the
ADE7758. As shown, the fastest integration time occurs when
the watt gain registers are set to maximum full scale, that is, 0x7FF.
This is the time it takes before overflow can be scaled by writing
to the WDIV register and therefore can be increased by a
maximum factor of 255.
Note that the active energy register content can roll over to full-
scale negative (0x8000) and continue increasing in value when
the active power is positive (see Figure 66). Conversely, if the
active power is negative, the energy register would under flow
to full-scale positive (0x7FFF) and continue decreasing in value.
By setting the AEHF bit (Bit 0) of the interrupt mask register,
the ADE7758 can be configured to issue an interrupt (IRQ)
when Bit 14 of any one of the three watt-hr accumulation
registers has changed, indicating that the accumulation register
is half full (positive or negative).
Setting the RSTREAD bit (Bit 6) of the LCYMODE register
enables a read-with-reset for the watt-hr accumulation registers,
that is, the registers are reset to 0 after a read operation.
CONT ENTS OF WATT-HR
ACCUMULATION REGISTER
0x7FFF
0x3FFF
0x0000
0xC000
0x8000 TIME (Sec)
0.13 0.52 0.79 1.05 1.31 1.58
WATT
G
A
IN = 0x7FF
WATT G AI N = 0x00 0
WATT G AI N = 0x80 0
0
4443-067
Figure 67. Energy Register Roll-Over Time for Full-Scale Power
(Minimum and Maximum Power Gain)
ADE7758
Rev. D | Page 33 of 72
Integration Time Under Steady Load
The discrete time sample period (T) for the accumulation register
is 0.4 µs (4/CLKIN). With full-scale sinusoidal signals on the
analog inputs and the watt gain registers set to 0x000, the average
word value from each LPF2 is 0xCCCCD (see Figure 64 and
Figure 66). The maximum value that can be stored in the watt-
hr accumulation register before it overflows is 215 − 1 or 0x7FFF.
Because the average word value is added to the internal register,
which can store 240 − 1 or 0xFF, FFFF, FFFF before it overflows,
the integration time under these conditions with WDIV = 0 is
calculated as
sec0.524s0.4
0xCCCCD
FFFFFFFF,0xFF, =×=Time (21)
When WDIV is set to a value different from 0, the time before
overflow is scaled accordingly as shown in Equation 22.
Time = Time (WDIV = 0) × WDIV[7:0] (22)
Energy Accumulation Mode
The active power accumulated in each watt-hr accumulation
register (AWATTHR, BWATTHR, or CWATTHR) depends on
the configuration of the CONSEL bits in the COMPMODE
register (Bit 0 and Bit 1). The different configurations are
described in Table 10.
Table 10. Inputs to Watt-Hr Accumulation Registers
CONSEL[1, 0] AWATTHR BWAT THR CWATTHR
00 VA × IA VB × IB VC × IC
01 VA × (IA – IB) 0 VC × (IC – IB)
10 VA × (IA – IB) 0 VC × IC
11 Reserved Reserved Reserved
The contents of the watt-hr accumulation registers are affected
by both the current gain register (IGAIN) and the watt gain
register of the corresponding phase. IGAIN should not be used
when using Mode 0 of CONSEL, COMPMODE[1:0].
Depending on the poly phase meter service, the appropriate
formula should be chosen to calculate the active energy. The
American ANSI C12.10 Standard defines the different
configurations of the meter.
Table 11 describes which mode should be chosen in these
different configurations.
Table 11. Meter Form Configuration
ANSI Meter Form CONSEL (d) TERMSEL (d)
5S/13S 3-Wire Delta 0 3, 5, or 6
6S/14S 4-Wire Wye 1 7
8S/15S 4-Wire Delta 2 7
9S/16S 4-Wire Wye 0 7
Different gain calibration parameters are offered in the ADE7758
to cover the calibration of the meter in different configurations.
It should be noted that in CONSEL Mode 0d, the IGAIN and
WGAIN registers have the same effect on the end result. However,
changing IGAIN also changes all other calculations that use the
current waveform. In other words, changing IGAIN changes the
active, reactive, and apparent energy, as well as the rms current
calculation results. IGAIN should not be used when using Mode 0
of CONSEL COMPMODE[1:0].
Active Power Frequency Output
Pin 1 (APCF) of the ADE7758 provides frequency output for
the total active power. After initial calibration during manufac-
turing, the manufacturer or end customer often verifies the
energy meter calibration. One convenient way to verify the
meter calibration is for the manufacturer to provide an output
frequency that is proportional to the energy or active power
under steady load conditions. This output frequency can provide a
simple, single-wire, optically isolated interface to external
calibration equipment. Figure 68 illustrates the energy-to-
frequency conversion in the ADE7758.
INPUT TO BWATTHR
REGISTER
INPUT TO AWATTHR
REGISTER
INPUT TO CWATTHR
REGISTER
DFC APCF
APCFNUM[11:0]
APCFDEN[11:0]
÷
+
+
+÷4
04443-068
Figure 68. Active Power Frequency Output
A digital-to-frequency converter (DFC) is used to generate the
APCF pulse output from the total active power. The TERMSEL
bits (Bit 2 to Bit 4) of the COMPMODE register can be used to
select which phases to include in the total power calculation.
Setting Bit 2, Bit 3, and Bit 4 includes the input to the AWATTHR,
BWATTHR, and CWATTHR registers in the total active power
calculation. The total active power is signed addition. However,
setting the ABS bit (Bit 5) in the COMPMODE register enables
the absolute-only mode; that is, only the absolute value of the
active power is considered.
The output from the DFC is divided down by a pair of frequency
division registers before being sent to the APCF pulse output.
Namely, APCFDEN/APCFNUM pulses are needed at the DFC
output before the APCF pin outputs a pulse. Under steady load
conditions, the output frequency is directly proportional to the
total active power. The pulse width of APCF is 64/CLKIN if
APCFNUM and APCFDEN are both equal. If APCFDEN is
greater than APCFNUM, the pulse width depends on APCFDEN.
The pulse width in this case is T × (APCFDEN/2), where T is
the period of the APCF pulse and APCFDEN/2 is rounded to
the nearest whole number. An exception to this is when the
period is greater than 180 ms. In this case, the pulse width is
fixed at 90 ms.
ADE7758
Rev. D | Page 34 of 72
The maximum output frequency (APCFNUM = 0x00 and
APCFDEN = 0x00) with full-scale ac signals on one phase is
approximately 16 kHz.
From Equation 24, E(t) equals
()
)4cos(
8
2
14
–1
2
2
1
1
tf
f
tf
IRMSVRMS
tIRMSVRMS π×
⎥
⎥
⎥
⎥
⎦
⎤
⎢
⎢
⎢
⎢
⎣
⎡
+π
×
×× (25)
The ADE7758 incorporates two registers to set the frequency of
APCF (APCFNUM[11:0] and APCFDEN[11:0]). These are
unsigned 12-bit registers that can be used to adjust the frequency of
APCF by 1/212 to 1 with a step of 1/212. For example, if the
output frequency is 1.562 kHz while the contents of APCFDEN
are 0 (0x000), then the output frequency can be set to 6.103 Hz
by writing 0xFF to the APCFDEN register.
From Equation 25, it can be seen that there is a small ripple in
the energy calculation due to the sin(2ωt) component (see
Figure 69). The ripple gets larger with larger loads. Choosing a
lower output frequency for APCF during calibration by using a
large APCFDEN value and keeping APCFNUM relatively small
can significantly reduce the ripple. Averaging the output
frequency over a longer period achieves the same results.
If 0 were written to any of the frequency division registers, the
divider would use 1 in the frequency division. In addition, the
ratio APCFNUM/APCFDEN should be set not greater than 1 to
ensure proper operation. In other words, the APCF output
frequency cannot be higher than the frequency on the DFC output.
–
E(t)
t
Vlt
VI ×sin(4π×f
1
×t)
4π×f
1
1 +
2
2f
1
8
04443-069
The output frequency has a slight ripple at a frequency equal to
2× the line frequency. This is due to imperfect filtering of the
instantaneous power signal to generate the active power signal
(see the Active Power Calculation section). Equation 14 gives an
expression for the instantaneous power signal. This is filtered by
LPF2, which has a magnitude response given by Equation 23.
()
2
2
8
1
1
H
f
f
+
= (23)
Figure 69. Output Frequency Ripple
Line Cycle Active Energy Accumulation Mode
The ADE7758 is designed with a special energy accumulation
mode that simplifies the calibration process. By using the on-
chip, zero-crossing detection, the ADE7758 updates the watt-hr
accumulation registers after an integer number of zero crossings
(see Figure 70). The line-active energy accumulation mode for
watt-hr accumulation is activated by setting the LWATT bit
(Bit 0) of the LCYCMODE register. The total energy accumu-
lated over an integer number of half-line cycles is written to the
watt-hr accumulation registers after the LINECYC number of zero
crossings is detected. When using the line cycle accumulation
mode, the RSTREAD bit (Bit 6) of the LCYCMODE register
should be set to Logic 0.
The active power signal (output of the LPF2) can be rewritten as
()
() (
tf
f
IRMSVRMS
IRMSVRMStp 1
2
2
1
4cos
8
2
1
π×
⎥
⎥
⎥
⎥
⎦
⎤
⎢
⎢
⎢
⎢
⎣
⎡
+
×
−×=
)
(24)
where f1 is the line frequency, for example, 60 Hz.
ZXSEL0
1
ZERO-CROSSING
DETECTION
(PHASE A)
ZXSEL1
1
ZERO-CROSSING
DETECTION
(PHASE B)
ZXSEL2
1
ZERO-CROSSING
DETECTION
(PHASE C)
1
ZXSEL [ 0:2] ARE BITS 3 TO 5 IN T HE LCYCM ODE REGIS TER
CALIBRATION
CONTROL
LINECYC[15:0]
WATTOS[11:0] WG[11:0] WDIV[7:0]
+
+
%
+
+
WATTHR[15:0]
ACCUMULATE ACTIVE POWER FOR
LI NECYC NUMB E R OF Z E R O-CROSSING S;
WATT-HR ACCUMULATION REGISTERS
ARE UPDATED ONCE EV E RY LI NE C YC
NUMBER OF ZERO-CROSSINGS
ACTIVE POWER
15 0
40 0
04443-070
Figure 70. ADE7758 Line Cycle Active Energy Accumulation Mode
ADE7758
Rev. D | Page 35 of 72
Phase A, Phase B, and Phase C zero crossings are, respectively,
included when counting the number of half-line cycles by
setting ZXSEL[0:2] bits (Bit 3 to Bit 5) in the LCYCMODE
register. Any combination of the zero crossings from all three
phases can be used for counting the zero crossing. Only one
phase should be selected at a time for inclusion in the zero
crossings count during calibration (see the Calibration section).
The number of zero crossings is specified by the LINECYC
register. LINECYC is an unsigned 16-bit register. The ADE7758
can accumulate active power for up to 65535 combined zero
crossings. Note that the internal zero-crossing counter is always
active. By setting the LWATT bit, the first energy accumulation
result is, therefore, incorrect. Writing to the LINECYC register
when the LWATT bit is set resets the zero-crossing counter, thus
ensuring that the first energy accumulation result is accurate.
At the end of an energy calibration cycle, the LENERGY bit
(Bit 12) in the STATUS register is set. If the corresponding
mask bit in the interrupt mask register is enabled, the IRQ
output also goes active low; thus, the IRQ can also be used to
signal the end of a calibration.
Because active power is integrated on an integer number of half-
line cycles in this mode, the sinusoidal component is reduced to
0, eliminating any ripple in the energy calculation. Therefore, total
energy accumulated using the line-cycle accumulation mode is
E(t) = VRMS × IRMS × t (26)
where t is the accumulation time.
Note that line cycle active energy accumulation uses the same
signal path as the active energy accumulation. The LSB size of
these two methods is equivalent. Using the line cycle accumula-
tion to calculate the kWh/LSB constant results in a value that
can be applied to the WATTHR registers when the line
accumulation mode is not selected (see the Calibration section).
REACTIVE POWER CALCULATION
A load that contains a reactive element (inductor or capacitor)
produces a phase difference between the applied ac voltage and
the resulting current. The power associated with reactive elements
is called reactive power, and its unit is VAR. Reactive power is
defined as the product of the voltage and current waveforms when
one of these signals is phase shifted by 90°.
Equation 30 gives an expression for the instantaneous reactive
power signal in an ac system when the phase of the current
channel is shifted by +90°.
()
(
)
θ= –sin2 ωtVtv (27)
() ()
()
⎟
⎠
⎞
⎜
⎝
⎛π
+=
′
=
2
sin2i
sin2
ωtIt
ωtIti
(28)
where:
v = rms voltage.
i = rms current.
θ = total phase shift caused by the reactive elements in the load.
Then the instantaneous reactive power q(t) can be expressed as
(
)
(
)
(
)
()
⎟
⎠
⎞
⎜
⎝
⎛π
θ
⎟
⎠
⎞
⎜
⎝
⎛π
θ=
′
×
=
2
––2cos–
2
––cos ωtVIVItq
titvtq
(29)
where
(
)
ti
′
is the current waveform phase shifted by 90°.
Note that q(t) can be rewritten as
(
)
(
)(
θ
)
+
θ
=
–2sinsin ωtVIVItq (30)
The average reactive power over an integral number of line
cycles (n) is given by the expression in Equation 31.
()
()
∫××== nT
0
θsindt
nT
1IVtqQ (31)
where:
T is the period of the line cycle.
Q is referred to as the average reactive power. The instantaneous
reactive power signal q(t) is generated by multiplying the
voltage signals and the 90° phase-shifted current in each phase.
The dc component of the instantaneous reactive power signal in
each phase (A, B, and C) is then extracted by a low-pass filter to
obtain the average reactive power information on each phase.
This process is illustrated in Figure 71. The reactive power of
each phase is accumulated in the corresponding 16-bit VAR-
hour register (AVARHR, BVARHR, or CVARHR). The input to
each reactive energy register can be changed depending on the
accumulation mode setting (see Table 21).
ADE7758
Rev. D | Page 36 of 72
The frequency response of the LPF in the reactive power signal
path is identical to that of the LPF2 used in the average active
power calculation (see Figure 65).
VRMS × IRMS × sin(φ)
θ
0x00000
CURRENT
i(t) = 2 × IRMS × sin(ωt)
VOLTAGE
v(t) = 2 × VRMS × sin(ωt – θ)
INSTANTANEOUS
REACTIVE POWER SIG NAL
q(t) = VRMS × IRMS × sin(φ) + VRMS × IRMS × sin (2ωt + θ)
AVERAGE REACTIVE POWER SIGNAL
=
VRMS × IRMS × sin(θ)
04443-071
Figure 71. Reactive Power Calculation
The low-pass filter is nonideal, so the reactive power signal has
some ripple. This ripple is sinusoidal and has a frequency equal
to 2× the line frequency. Because the ripple is sinusoidal
in nature, it is removed when the reactive power signal is
integrated over time to calculate the reactive energy.
The phase-shift filter has –90° phase shift when the integrator is
enabled and +90° phase shift when the integrator is disabled. In
addition, the filter has a nonunity magnitude response. Because
the phase-shift filter has a large attenuation at high frequency,
the reactive power is primarily for the calculation at line
frequency. The effect of harmonics is largely ignored in the
reactive power calculation. Note that because of the magnitude
characteristic of the phase shifting filter, the LSB weight of the
reactive power calculation is slightly different from that of the
active power calculation (see the Energy Registers Scaling
section). The ADE7758 uses the line frequency of the phase
selected in the FREQSEL[1:0] bits of the MMODE[1:0] to
compensate for attenuation of the reactive energy phase shift
filter over frequency (see the Period Measurement section).
Reactive Power Gain Calibration
The average reactive power from the LPF output in each phase
can be scaled by ±50% by writing to the phase’s VAR gain register
(AVARG, BVARG, or CVARG). The VAR gain registers are twos
complement, signed registers and have a resolution of 0.024%/LSB.
The function of the VAR gain registers is expressed by
⎟
⎠
⎞
⎜
⎝
⎛+×
=
12
2
12 gisterReGainVAR
OutputLPF
Powe
r
ReactiveAverage
(32)
The output is scaled by –50% by writing 0x800 to the VAR gain
registers and increased by +50% by writing 0x7FF to them.
These registers can be used to calibrate the reactive power (or
energy) calculation in the ADE7758 for each phase.
Reactive Power Offset Calibration
The ADE7758 incorporates a VAR offset register on each phase
(AVAROS, BVAROS, and CVAROS). These are signed twos
complement, 12-bit registers that are used to remove offsets in
the reactive power calculations. An offset can exist in the power
calculation due to crosstalk between channels on the PCB or in
the chip itself. The offset calibration allows the contents of the
reactive power register to be maintained at 0 when no reactive
power is being consumed. The offset registers’ resolution is the
same as the active power offset registers (see the Apparent
Power Offset Calibration section).
Sign of Reactive Power Calculation
Note that the average reactive power is a signed calculation. As
stated previously, the phase shift filter has –90° phase shift when
the integrator is enabled and +90° phase shift when the
integrator is disabled.
Table 12 summarizes the relationship between the phase difference
between the voltage and the current and the sign of the resulting
VAR calc u lation.
The ADE7758 has a sign detection circuit for the reactive power
calculation. The REVPRP bit (Bit 18) in the interrupt status
register is set if the average reactive power from any one of the
phases changes. The phases monitored are selected by TERMSEL
bits in the COMPMODE register (see Table 21). If the REVPRP
bit is set in the mask register, the IRQ logic output goes active
low (see the section). Note that this bit is set whenever
there is a sign change; that is, the bit is set for either a positive-
to-negative change or a negative-to-positive change of the sign
bit. The response time of this bit is approximately 176 ms for a
full-scale signal, which has an average value of 0xCCCCD at the
low-pass filter output. For smaller inputs, the time is longer.
Interrupts
CLKINueAverageVal
mssponseTimeRe 42
601 25 ×
⎥
⎦
⎤
⎢
⎣
⎡
+≅ (33)
Table 12. Sign of Reactive Power Calculation
Φ1 Integrator Sign of Reactive Power
Between 0 to +90 Off Positive
Between −90 to 0 Off Negative
Between 0 to +90 On Positive
Between −90 to 0 On Negative
1 Φ is defined as the phase angle of the voltage signal minus the current
signal; that is, Φ is positive if the load is inductive and negative if the load is
capacitive.
ADE7758
Rev. D | Page 37 of 72
Reactive Energy Calculation
Reactive energy is defined as the integral of reactive power.
()
dttqEnergyReactive ∫
= (34)
VARDIV is an 8-bit unsigned register that is useful to lengthen
the time it takes before the VAR-hr accumulation registers
overflow.
Similar to reactive power, the fastest integration time occurs
when the VAR gain registers are set to maximum full scale,
that is, 0x7FF. The time it takes before overflow can be scaled
by writing to the VARDIV register; and, therefore, it can be
increased by a maximum factor of 255.
Similar to active power, the ADE7758 achieves the integration
of the reactive power signal by continuously accumulating the
reactive power signal in the internal 41-bit accumulation
registers. The VAR-hr registers (AVARHR, BVARHR, and
CVARHR) represent the upper 16 bits of these internal
registers. This discrete time accumulation or summation is
equivalent to integration in continuous time. Equation 35
expresses the relationship
When overflow occurs, the VAR-hr accumulation registers
content can rollover to full-scale negative (0x8000) and continue
increasing in value when the reactive power is positive. Con-
versely, if the reactive power is negative, the VAR-hr accumulation
registers content can roll over to full-scale positive (0x7FFF)
and continue decreasing in value.
()
()
⎭
⎬
⎫
⎩
⎨
⎧×== ∑
∫∞
=
→0n
0
Lim
dt TnTqtqEnergyReactive
T
(35)
By setting the REHF bit (Bit 1) of the interrupt mask register,
the ADE7758 can be configured to issue an interrupt (IRQ)
when Bit 14 of any one of the three VAR-hr accumulation
registers has changed, indicating that the accumulation register
is half full (positive or negative).
where:
n is the discrete time sample number.
T is the sample period.
Figure 72 shows the signal path of the reactive energy accumula-
tion. The average reactive power signal is continuously added
to the internal reactive energy register. This addition is a signed
operation. Negative energy is subtracted from the reactive energy
register. The average reactive power is divided by the content
of the VAR divider register before it is added to the corresponding
VAR-hr accumulation registers. When the value in the
VARDIV[7:0] register is 0 or 1, the reactive power is accumulated
without any division.
Setting the RSTREAD bit (Bit 6) of the LCYMODE register
enables a read-with-reset for the VAR-hr accumulation
registers; that is, the registers are reset to 0 after a read
operation.
VARG[11:0]
VARDIV[7:0]
90° PHASE
SHIFTING FILTER
MULTIPLIER
I
V
HPF
CURRENT SIGNAL–i(t)
0x2851EC
0x00
0xD7AE14
VOLTAGE SIGNAL–v(t)
0x2852
0x00
0xD7AE
+++
+
LPF2
%
SIGN 2
6
2
0
2
–1
2
–2
2
–3
2
–4
V
AR
OS[11:0] VARHR[15:0]
15 0
40 0
TOTAL REACTIVE POWER IS
ACCUMULAT E D (INT EG RATED) I N
THE VAR-HR ACCUMULATION REGISTERS
π
2
PHCAL[6:0]
Φ
04443-072
Figure 72. ADE7758 Reactive Energy Accumulation
ADE7758
Rev. D | Page 38 of 72
Integration Time Under Steady Load
The discrete time sample period (T) for the accumulation
register is 0.4 µs (4/CLKIN). With full-scale sinusoidal signals
on the analog inputs, a 90° phase difference between the voltage
and the current signal (the largest possible reactive power), and
the VAR gain registers set to 0x000, the average word value from
each LPF2 is 0xCCCCD.
The maximum value that can be stored in the reactive energy
register before it overflows is 215 − 1 or 0x7FFF. Because the
average word value is added to the internal register, which can
store 240 − 1 or 0xFF, FFFF, FFFF before it overflows, the
integration time under these conditions with VARDIV = 0 is
calculated as
sec0.5243s0.4
0xCCCCD
FFFFFFFF,0xFF, =×=Time (36)
When VARDIV is set to a value different from 0, the time
before overflow are scaled accordingly as shown in Equation 37.
Time = Time(VARDIV = 0) × VARDIV (37)
Energy Accumulation Mode
The reactive power accumulated in each VAR-hr accumulation
register (AVARHR, BVARHR, or CVARHR) depends on the
configuration of the CONSEL bits in the COMPMODE register
(Bit 0 and Bit 1). The different configurations are described in
Table 13. Note that IA’/IB’/IC’ are the current phase-shifted
current waveform.
Table 13. Inputs to VAR-Hr Accumulation Registers
CONSEL[1, 0] AVARHR BVARHR CVARHR
00 VA × IA’ VB × IB VC × IC’
01 VA (IA’ – IB’) 0 VC (IC’ – IB’)
10 VA (IA’ – IB’) 0 VC × IC’
11 Reserved Reserved Reserved
The contents of the VAR-hr accumulation registers are affected
by both the current gain register (IGAIN) and the VAR gain
register of the corresponding phase. IGAIN should not be used
when using Mode 0 of CONSEL COMPMODE[1:0].
Reactive Power Frequency Output
Pin 17 (VARCF) of the ADE7758 provides frequency output for
the total reactive power. Similar to APCF, this pin provides an
output frequency that is directly proportional to the total
reactive power. The pulse width of VARPCF is 64/CLKIN if
VARCFNUM and VARCFDEN are bot h equal. If VARCFDEN
is greater than VARCFNUM, the pulse width depends on
VARCFDEN. The pulse width in this case is T × (VARCFDEN/2),
where T is the period of the VARCF pulse and VARCFDEN/2
is rounded to the nearest whole number. An exception to this
is when the period is greater than 180 ms. In this case, the pulse
width is fixed at 90 ms.
A digital-to-frequency converter (DFC) is used to generate the
VARCF pulse output from the total reactive power. The TERMSEL
bits (Bit 2 to Bit 4) of the COMPMODE register can be used to
select which phases to include in the total reactive power calcu-
lation. Setting Bit 2, Bit 3, and Bit 4 includes the input to the
AVARHR, BVARHR, and CVARHR registers in the total
reactive power calculation. The total reactive power is signed
addition. However, setting the SAVAR bit (Bit 6) in the
COMPMODE register enables absolute value calculation. If the
active power of that phase is positive, no change is made to the
sign of the reactive power. However, if the sign of the active power
is negative in that phase, the sign of its reactive power is inverted
before summing and creating VARCF pulses. This mode should
be used in conjunction with the absolute value mode for active
power (Bit 5 in the COMPMODE register) for APCF pulses.
The effects of setting the ABS and SAVAR bits of the
COMPMODE register are as follows when ABS = 1 and
SAVAR = 1:
If watt > 0, APCF = Watts, VARCF = +VAR.
If watt < 0, APCF = |Watts|, VARCF = −VAR.
INPUT TO BVARHR
REGISTER
INPU
T
TO
A
V
A
RHR
REGISTER
INPUT TO CVARHR
REGISTER
+
+
+
INPUT TO BVAHR
REGISTER
INPUT TO AVAHR
REGISTER
INPUT TO CVAHR
REGISTER
+
+
+
0
1VARCF
VARCFNUM[11:0]
VARCFDEN[11:0]
÷
DFC
VACF BI T (BI T 7) OF
WAVMODE REGISTER
÷4
04443-073
Figure 73. Reactive Power Frequency Output
The output from the DFC is divided down by a pair of frequency
division registers before sending to the VARCF pulse output.
Namely, VARCFDEN/VARCFNUM pulses are needed at the
DFC output before the VARCF pin outputs a pulse. Under
steady load conditions, the output frequency is directly
proportional to the total reactive power.
Figure 73 illustrates the energy-to-frequency conversion in the
ADE7758. Note that the input to the DFC can be selected between
the total reactive power and total apparent power. Therefore,
the VARCF pin can output frequency that is proportional to the
total reactive power or total apparent power. The selection is
made by setting the VACF bit (Bit 7) in the WAVMODE register.
Setting this bit switches the input to the total apparent power.
The default value of this bit is logic low. Therefore, the default
output from the VARCF pin is the total reactive power.
All other operations of this frequency output are similar to that
of the active power frequency output (see the Active Power
Frequency Output section).
ADE7758
Rev. D | Page 39 of 72
Line Cycle Reactive Energy Accumulation Mode
The line cycle reactive energy accumulation mode is activated
by setting the LVAR bit (Bit 1) in the LCYCMODE register. The
total reactive energy accumulated over an integer number of
zero crossings is written to the VAR-hr accumulation registers
after the LINECYC number of zero crossings is detected. The
operation of this mode is similar to watt-hr accumulation (see
the Line Cycle Active Energy Accumulation Mode section).
When using the line cycle accumulation mode, the RSTREAD
bit (Bit 6) of the LCYCMODE register should be set to Logic 0.
APPARENT POWER CALCULATION
Apparent power is defined as the amplitude of the vector sum of
the active and reactive powers. Figure 74 shows what is typically
referred to as the power triangle.
REACTI V E POW E
R
ACTIVE POWER
APPARENT
POWER
θ
04443-074
Figure 74. Power Triangle
There are two ways to calculate apparent power: the arithmetical
approach or the vectorial method. The arithmetical approach
uses the product of the voltage rms value and current rms value
to calculate apparent power. Equation 38 describes the arithmetical
approach mathematically.
S = VRMS × IRMS (38)
where S is the apparent power, and VRMS and IRMS are the
rms voltage and current, respectively.
The vectorial method uses the square root of the sum of the
active and reactive power, after the two are individually squared.
Equation 39 shows the calculation used in the vectorial approach.
22 QPS += (39)
where:
S is the apparent power.
P is the active power.
Q is the reactive power.
For a pure sinusoidal system, the two approaches should yield
the same result. The apparent energy calculation in the ADE7758
uses the arithmetical approach. However, the line cycle energy
accumulation mode in the ADE7758 enables energy accumula-
tion between active and reactive energies over a synchronous
period, thus the vectorial method can be easily implemented in
the external MCU (see the Line Cycle Active Energy
Accumulation Mode section).
Note that apparent power is always positive regardless of the
direction of the active or reactive energy flows. The rms value of
the current and voltage in each phase is multiplied to produce
the apparent power of the corresponding phase.
The output from the multiplier is then low-pass filtered to obtain
the average apparent power. The frequency response of the LPF
in the apparent power signal path is identical to that of the LPF2
used in the average active power calculation (see Figure 65).
Apparent Power Gain Calibration
Note that the average active power result from the LPF output
in each phase can be scaled by ±50% by writing to the phase’s
VAGAIN register ( AVAG, BVAG, or CVAG). The VAGAIN
registers are twos complement, signed registers and have a
resolution of 0.024%/LSB. The function of the VAGAIN
registers is expressed mathematically as
⎟
⎠
⎞
⎜
⎝
⎛+×
=
12
2
12 RegisterVAGAIN
OutputLPF
PowerApparentAverage
(40)
The output is scaled by –50% by writing 0x800 to the VAR gain
registers and increased by +50% by writing 0x7FF to them.
These registers can be used to calibrate the apparent power (or
energy) calculation in the ADE7758 for each phase.
Apparent Power Offset Calibration
Each rms measurement includes an offset compensation register
to calibrate and eliminate the dc component in the rms value
(see the Current RMS Calculation section and the Vo ltage
Channel RMS Calculation section). The voltage and current
rms values are then multiplied together in the apparent power
signal processing. As no additional offsets are created in the
multiplication of the rms values, there is no specific offset
compensation in the apparent power signal processing. The offset
compensation of the apparent power measurement in each phase
should be done by calibrating each individual rms measurement
(see the Calibration section).
ADE7758
Rev. D | Page 40 of 72
Apparent Energy Calculation
Apparent energy is defined as the integral of apparent power.
Apparent Energy = ∫ S(t)dt (41)
Similar to active or reactive power accumulation, the fastest
integration time occurs when the VAGAIN registers are set to
maximum full scale, that is, 0x7FF. When overflow occurs, the
content of the VA-hr accumulation registers can roll over to 0
and continue increasing in value.
Similar to active and reactive energy, the ADE7758 achieves the
integration of the apparent power signal by continuously
accumulating the apparent power signal in the internal 41-bit,
unsigned accumulation registers. The VA-hr registers (AVAHR,
BVAHR, and CVAHR) represent the upper 16 bits of these
internal registers. This discrete time accumulation or
summation is equivalent to integration in continuous time.
Equation 42 expresses the relationship
By setting the VAEHF bit (Bit 2) of the mask register, the ADE7758
can be configured to issue an interrupt (IRQ) when the MSB of
any one of the three VA-hr accumulation registers has changed,
indicating that the accumulation register is half full.
Setting the RSTREAD bit (Bit 6) of the LCYMODE register
enables a read-with-reset for the VA-hr accumulation registers;
that is, the registers are reset to 0 after a read operation.
()
()
⎭
⎬
⎫
⎩
⎨
⎧×== ∑
∫∞
=
→0n
0T
Limdt TnTStSEnergyApparent (42)
Integration Time Under Steady Load
The discrete time sample period (T) for the accumulation register
is 0.4 µs (4/CLKIN). With full-scale, 60 Hz sinusoidal signals on
the analog inputs and the VAGAIN registers set to 0x000, the
average word value from each LPF2 is 0xB9954. The maximum
value that can be stored in the apparent energy register before it
overflows is 216 − 1 or 0xFFFF. As the average word value is first
added to the internal register, which can store 241 − 1 or 0x1FF,
FFFF, FFFF before it overflows, the integration time under these
conditions with VADIV = 0 is calculated as
where:
n is the discrete time sample number.
T is the sample period.
Figure 75 shows the signal path of the apparent energy accumu-
lation. The apparent power signal is continuously added to the
internal apparent energy register. The average apparent power is
divided by the content of the VA divider register before it is
added to the corresponding VA-hr accumulation register. When
the value in the VADIV[7:0] register is 0 or 1, apparent power is
accumulated without any division. VADIV is an 8-bit unsigned
register that is useful to lengthen the time it takes before the
VA-hr accumulation registers overflow.
sec1.157s0.4
0xB9954
FFFFFFFF,0x1FF, =×=Time (43)
When VADIV is set to a value different from 0, the time before
overflow is scaled accordingly, as shown in Equation 44.
Time = Time(VADIV = 0) × VADIV (44)
VOLTAGE RMS SIGNAL
0x174BAC
60Hz
0x0
0x17F263
50Hz
0x0
CURRENT RMS SIGNAL
0x1C82B
0x00
MULTIPLIER
IRMS
V
RMS
VAG[11:0]
VADIV[7:0]
+
+
LPF2
%
VARHR[15:0]
15 0
40 0
APPARENT POWER IS
ACCUMULATED (INTEGRATED) IN
THE VA-HR ACCUMULATION REGISTERS
04443-075
Figure 75. ADE7758 Apparent Energy Accumulation
ADE7758
Rev. D | Page 41 of 72
Table 14. Inputs to VA-Hr Accumulation Registers
CONSEL[1, 0] AVAHR1BVAHR CVAHR
00 AVRMS × AIRMS BVRMS × BIRMS CVRMS × CIRMS
01 AVRMS × AIRMS AVRMS + CVRMS/2 × BIRMS CVRMS × CIRMS
10 AVRMS × AIRMS BVRMS × BIRMS CVRMS × CIRMS
11 Reserved Reserved Reserved
1 AVRMS/BVRMS/CVRMS are the rms voltage waveform, and AIRMS/BIRMS/CIRMS are the rms values of the current waveform.
Energy Accumulation Mode
The apparent power accumulated in each VA-hr accumulation
register (AVAHR, BVAHR, or CVAHR) depends on the con-
figuration of the CONSEL bits in the COMPMODE register
(Bit 0 and Bit 1). The different configurations are described in
Table 14.
The contents of the VA-hr accumulation registers are affected
by both the registers for the current gain (IGAIN) and rms
voltage gain (VRMSGAIN), as well as the VAGAIN register of
the corresponding phase. IGAIN should not be used when
using CONSEL Mode 0, COMPMODE[1:0].
Apparent Power Frequency Output
Pin 17 (VARCF) of the ADE7758 provides frequency output for
the total apparent power. By setting the VACF bit (Bit 7) of the
WAVMODE register, this pin provides an output frequency that
is directly proportional to the total apparent power.
A digital-to-frequency converter (DFC) is used to generate the
pulse output from the total apparent power. The TERMSEL bits
(Bit 2 to Bit 4) of the COMPMODE register can be used to
select which phases to include in the total power calculation.
Setting Bit 2, Bit 3, and Bit 4 includes the input to the AVAHR,
BVAHR, and CVAHR registers in the total apparent power
calculation. A pair of frequency divider registers, namely
VARCFDEN and VARCFNUM, can be used to scale the output
frequency of this pin. Note that either VAR or apparent power
can be selected at one time for this frequency output (see the
Reactive Power Frequency Output section).
Line Cycle Apparent Energy Accumulation Mode
The line cycle apparent energy accumulation mode is activated
by setting the LVA bit (Bit 2) in the LCYCMODE register. The
total apparent energy accumulated over an integer number of
zero crossings is written to the VA-hr accumulation registers
after the LINECYC number of zero crossings is detected. The
operation of this mode is similar to watt-hr accumulation (see
the Line Cycle Active Energy Accumulation Mode section).
When using the line cycle accumulation mode, the RSTREAD
bit (Bit 6) of the LCYCMODE register should be set to Logic 0.
Note that this mode is especially useful when the user chooses
to perform the apparent energy calculation using the vectorial
method.
By setting LWATT and LVAR bits (Bit 0 and Bit 1) of the
LCYCMODE register, the active and reactive energies are
accumulated over the same period. Therefore, the MCU can
perform the squaring of the two terms and then take the square
root of their sum to determine the apparent energy over the
same period.
ENERGY REGISTERS SCALING
The ADE7758 provides measurements of active, reactive, and
apparent energies that use separate signal paths and filtering for
calculation. The differences in the datapaths can result in small
differences in LSB weight between the active, reactive, and
apparent energy registers. These measurements are internally
compensated so that the scaling is nearly one to one. The
relationship between the registers is shown in Table 15.
Table 15. Energy Registers Scaling
Frequency
60 Hz 50 Hz
Integrator Off
VAR 1.004 × WATT 1.0054 × WATT
VA 1.00058 × WATT 1.0085 × WATT
Integrator On
VAR 1.0059 × WATT 1.0064 × WATT
VA 1.00058 × WATT 1.00845 × WATT
WAVEFORM SAMPLING MODE
The waveform samples of the current and voltage waveform, as
well as the active, reactive, and apparent power multiplier out-
puts, can all be routed to the WAVEFORM register by setting
the WAVSEL[2:0] bits (Bit 2 to Bit 4) in the WAVMODE
register. The phase in which the samples are routed is set by
setting the PHSEL[1:0] bits (Bit 0 and Bit 1) in the WAVMODE
register. All energy calculation remains uninterrupted during
waveform sampling. Four output sample rates can be chosen by
using Bit 5 and Bit 6 of the WAVMODE register (DTRT[1:0]).
The output sample rate can be 26.04 kSPS, 13.02 kSPS,
6.51 kSPS, or 3.25 kSPS (see Table 20).
By setting the WFSM bit in the interrupt mask register to
Logic 1, the interrupt request output IRQ goes active low when
a sample is available. The 24-bit waveform samples are
transferred from the ADE7758 one byte (8 bits) at a time, with
the most significant byte shifted out first.
ADE7758
Rev. D | Page 42 of 72
The interrupt request output IRQ stays low until the interrupt
routine reads the reset status register (see the section). Interrupts
CALIBRATION
A reference meter or an accurate source is required to calibrate
the ADE7758 energy meter. When using a reference meter, the
ADE7758 calibration output frequencies APCF and VARCF are
adjusted to match the frequency output of the reference meter
under the same load conditions. Each phase must be calibrated
separately in this case. When using an accurate source for
calibration, one can take advantage of the line cycle accumulation
mode and calibrate the three phases simultaneously.
There are two objectives in calibrating the meter: to establish
the correct impulses/kW-hr constant on the pulse output and to
obtain a constant that relates the LSBs in the energy and rms
registers to Watt/VA/VAR hours, amps, or volts. Additionally,
calibration compensates for part-to-part variation in the meter
design as well as phase shifts and offsets due to the current
sensor and/or input networks.
Calibration Using Pulse Output
The ADE7758 provides a pulsed output proportional to the
active power accumulated by all three phases, called APCF.
Additionally, the VARCF output is proportional to either the
reactive energy or apparent energy accumulated by all three
phases. The following section describes how to calibrate the
gain, offset, and phase angle using the pulsed output information.
The equations are based on the pulse output from the ADE7758
(APCF or VARCF) and the pulse output of the reference meter
or CFEXPECTED.
Figure 76 shows a flowchart of how to calibrate the ADE7758
using the pulse output. Because the pulse outputs are
proportional to the total energy in all three phases, each phase
must be calibrated individually. Writing to the registers is fast to
reconfigure the part for calibrating a different phase; therefore,
Figure 76 shows a method that calibrates all phases at a given
test condition before changing the test condition.
ADE7758
Rev. D | Page 43 of 72
START CALIBRATE IRMS
OFFSET
CALIBRATE VRMS
OFFSET
MUST BE DONE
BEFORE VA GAIN
CALIBRATION
WATT AND VA
CAN BE CALIBRATED
SIMULTANEOUSLY @
PF = 1 BECAUSE THEY
HAVE SEPARATE PULSE OUTPUTS
ALL
PHASES
VA AND WATT
GAIN CAL?
YES NO
SET UP PULSE
OUTPUT FOR
A, B, OR C
CALIBRATE
WATT AND VA
GAIN @ I
TEST
,
PF = 1
ALL
PHASES
GAIN CAL
VAR?
YES NO
SET UP FOR
PHASE
A, B, OR C
CALIBRATE
VAR GAIN
@ I
TEST
, PF = 0,
INDUCTIVE
ALL
PHASES
PHASE ERROR
CAL?
YES NO
SET UP PULSE
OUTPUT FOR
A, B, OR C
CALIBRATE
PHASE @ I
TEST
,
PF = 0.5,
INDUCTIVE
ALL PHASES
VAR OFFSET
CAL?
YES NO
SET UP PULSE
OUTPUT FOR
A, B, OR C
CALIBRATE
VAR OFFSET
@ I
MIN
, PF = 0,
INDUCTIVE
ALL PHASES
WATT OFFSET
CAL?
YES NO
SET UP PULSE
OUTPUT FOR
A, B, OR C
CALIBRATE
WATT OFFSET
@ I
MIN
, PF = 1
END
04443-076
Figure 76. Calibration Using Pulse Output
Gain Calibration Using Pulse Output
Gain calibration is used for meter-to-meter gain adjustment,
APCF or VARCF output rate calibration, and determining the
Wh/LSB, VARh/LSB, and VAh/LSB constant. The registers used
for watt gain calibration are APCFNUM (0x45), APCFDEN
(0x46), and xWG (0x2A to 0x2C). Equation 50 through
Equation 52 show how these registers affect the Wh/LSB
constant and the APCF pulses.
For calibrating VAR gain, the registers in Equation 50 through
Equation 52 should be replaced by VARCFNUM (0x47),
VARCFDEN (0x48), and xVARG (0x2D to 0x2F). For VAGAIN,
they should be replaced by VARCFNUM (0x47), VARCFDEN
(0x48), and xVAG (0x30 to 0x32).
Figure 77 shows the steps for gain calibration of watts, VA, or
VAR using the pulse outputs.
ADE7758
Rev. D | Page 44 of 72
START
STEP 1
STEP 1A
ENABLE APCF AND
VARCF PULSE
OUTPUTS
STEP 2
CLEAR GAIN REGI ST ERS:
xWG, xVAG, xVARG
SELECT VA FOR
VARCF OUTPUT
CFNUM/VARCFNUM
SET TO CALCULATE
VALUES?
NO YES
ALL
PHASES VA
AND WATT
GAIN CAL?
YES NO
STEP 3
SET UP PULSE
OUTPUT FOR
PHASE A, B, OR C
STEP 5
SET UP SYSTEM
FOR I
TEST
,V
NOM
PF = 1
STEP 6
MEASURE %
ERROR FOR APCF
AND VARCF
STEP 7
CALCULATE AND
WRITE TO
xWG, xVAG
CALCULATE Wh/LSB
AND VAh/LSB
CONSTANTS
SET CFNUM/VARCFNUM
AND CFDEN/VARCFDEN
TO CALCULATED VALUES
STEP 4
END
ALL PHASES
VAR G AIN
CALIBRATED?
YES NO
SELECT VAR
FOR VARCF
OUTPUT
STEP 3
SET UP PULSE
OUTPUT FOR
PHASE A, B, OR C
VARCFNUM/
VARCFDEN
SET TO CALCULATED
VALUES?
NO YES
STEP 5
SET UP SYSTEM
FOR I
TEST
,V
NOM
PF = 0, I NDUCT I VE
STEP 6
MEASURE %
ERROR FOR
VARCF
STEP 7
CALCULATE AND
WRITE TO xVARG
CALCULATE
VARh/LSB
CONSTANT
SET
VARCFNUM/VARCFDEN TO
CALCULATED VALUES
STEP 4
04443-077
SELECT PHASE A,
B, OR C FOR L INE
PERIOD
MEASUREMENT
Figure 77. Gain Calibration Using Pulse Output
Step 1: Enable the pulse output by setting Bit 2 of the OPMODE
register (0x13) to Logic 0. This bit enables both the APCF and
VARCF pulses.
Step 1a: VAR and VA share the VARCF pulse output.
WAVMODE[7], Address (0x15), should be set to choose
between VAR or VA pulses on the output. Setting the bit to
Logic 1 selects VA. The default is Logic 0 or VARCF pulse
output.
Step 2: Ensure the xWG/xVARG/xVAG are zero.
Step 3: Disable the Phase B and Phase C contribution to the APCF
and VARCF pulses. This is done by the TERMSEL[2:4] bits of
the COMPMODE register (0x16). Setting Bit 2 to Logic 1 and
Bit 3 and Bit 4 to Logic 0 allows only Phase A to be included in
the pulse outputs. Select Phase A, Phase B, or Phase C for a line
period measurement with the FREQSEL[1:0] bits in the MMODE
register (0x14). For example, clearing Bit 1 and Bit 0 selects
Phase A for line period measurement.
ADE7758
Rev. D | Page 45 of 72
Step 4: Set APCFNUM (0x45) and APCFDEN (0x46) to the
calculated value to perform a coarse adjustment on the
imp/kWh ratio. For VAR/VA cal ibration, set VARCFNUM
(0x47) and VARCFDEN (0x48) to the calculated value.
The pulse output frequency with one phase at full-scale inputs
is approximately 16 kHz. A sample set of meters could be tested
to find a more exact value of the pulse output at full scale in the
user application.
To calculate the values for APCFNUM/APCFDEN and
VARCFNUM/VARCF DEN, us e the following for mulas:
FULLSCALE
TEST
FULLSCALE
NOM
NOMINAL I
I
V
V
APCF ××= kHz16 (45)
()
θ×
×
××
=cos
36001000
NOM
TEST
EXPECTED
VIMC
APCF (46)
⎟
⎠
⎞
⎜
⎝
⎛
=
EXPECTED
NOMINAL
APCF
APCF
INTAPCFDEN (47)
where:
MC is the meter constant.
ITEST is the test current.
VNOM is the nominal voltage at which the meter is tested.
VFULLSCALE and IFULLSCALE are the values of current and voltage,
which correspond to the full-scale ADC inputs of the ADE7758.
θ is the angle between the current and the voltage channel.
APCFEXPECTED is equivalent to the reference meter output under
the test conditions.
APCFNUM is written to 0 or 1.
The equations for calculating the VARCFNUM and
VARCFDEN during VAR cal ibration are similar :
()
θ×
×
××
=sin
36001000
NOM
TEST
EXPECTED
VIMC
VARCF (48)
Because the APCFDEN and VARCFDEN values can be
calculated from the meter design, these values can be written
to the part automatically during production calibration.
Step 5: Set the test system for ITEST, VNOM, and the unity power
factor. For VAR calibration, the power factor should be set to 0
inductive in this step. For watt and VA, the unity power factor
should be used. VAGAIN can be calibrated at the same time as
WGAIN because VAGAIN can be calibrated at the unity power
factor, and both pulse outputs can be measured simultaneously.
However, when calibrating VAGAIN at the same time as WGAIN,
the rms offsets should be calibrated first (see the Calibration of
IRMS and VRMS Offset section).
Step 6: Measure the percent error in the pulse output, APCF
and/or VARCF, from the reference meter:
%100
–
%×=
REF
REF
CF
CFAPCF
Error (49)
where CFREF = APCFEXPECTED = the pulse output of the reference
meter.
Step 7: Calculate xWG adjustment. One LSB change in xWG
(12 bits) changes the WATTHR register by 0.0244% and
therefore APCF by 0.0244%. The same relationship holds true
for VARCF.
[]
[]
[]
⎟
⎠
⎞
⎜
⎝
⎛+××
=
12
2
0:11
1
0:11
0:11 xWG
APCFDEN
APCFNUM
APCF
APC
F
NOMINAL
EXPECTED
(50)
%0244.0
%
–Error
xWG = (51)
When APCF is calibrated, the xWATTHR registers have the
same Wh/LSB from meter to meter if the meter constant and
the APCFNUM/APCFDEN ratio remain the same. The
Wh/LSB constant is
WDIVAPCFNU
M
APCFDENMC
LSB
Wh
1
1000
4
1
×××
= (52)
Return to Step 2 to calibrate Phase B and Phase C gain.
Example: Watt Gain Calibration of Phase A Using Pulse
Output
For this example, ITEST = 10 A, VNOM = 220 V, VFULLSCALE = 500 V,
IFULLSCALE = 130 A, MC = 3200 impulses/kWh, Power Factor = 1,
and Frequency = 50 Hz.
Clear APCFNUM (0x45) and write the calculated value to
APCFDEN (0x46) to perform a coarse adjustment on the
imp/kWh ratio, using Equation 45 through Equation 47.
kHz542.0
130
10
500
220
kHz16 =××=
NOMINAL
APCF
()
Hz9556.10cos
36001000
220103200 =×
×
××
=
EXPECTED
APCF
277
Hz9556.1
Hz542 =
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
=INTAPCFDEN
With Phase A contributing to CF, at ITEST, VNOM, and the unity
power factor, the example ADE7758 meter shows 2.058 Hz on
the pulse output. This is equivalent to a 5.26% error from the
reference meter value using Equation 49.
%26.5%100
Hz9556.1
Hz9556.1–Hz058.2 =×=%Error
The AWG value is calculated to be −216 d using Equation 51,
which means the value 0xF28 should be written to AWG.
ADE7758
Rev. D | Page 46 of 72
Step 5: Calculate xPHCAL.
2802165.215
%0244.0
%26.5
–xFAWG =−=−==
°
×××
=
360
1
)(
1
__
1
sPeriodLineWeightLSBPHCAL
ErrorPhase
xPHCAL
(54)
Phase Calibration Using Pulse Output
The ADE7758 includes a phase calibration register on each phase
to compensate for small phase errors. Large phase errors should
be compensated by adjusting the antialiasing filters. The ADE7758’s
phase calibration is a time delay with different weights in the
positive and negative direction (see the Phase Compensation
section). Because a current transformer is a source of phase error,
a fixed nominal value can be decided on to load into the xPHCAL
registers at power-up. During calibration, this value can be adjusted
for CT-to-CT error. Figure 78 shows the steps involved in
calibrating the phase using the pulse output.
where PHCAL_LSB_Weight is 1.2 s if the %Error is negative or
2.4 s if the %Error is positive (see the Phase Compensation
section).
If it is not known, the line period is available in the ADE7758’s
frequency register, FREQ (0x10). To configure line period
measurement, select the phase for period measurement in the
MMODE[1:0] and set LCYCMODE[7]. Equation 55 shows how
to determine the value that needs to be written to xPHCAL
using the period register measurement.
START
ALL
PHASES
PHASE ERROR
CALIBRATED?
END
YES NO
STEP 1
SET UP PULSE
OUTPUT FOR
PHASE A, B, OR C
AND ENABLE CF
OUTPUTS
STEP 2
SET UP SYSTEM
FOR I
TEST
,V
NOM
,
PF = 0.5, INDUCTIVE
STEP 3
MEASURE %
ERROR IN APCF
STEP 4
CALCULATE PHASE
ERROR (DEGREES)
STEP 5
PERIOD OF
SYSTEM
KNOWN?
MEASURE
PERIOD USING
FREQ[11:0]
REGISTER
NO YES
CALCULATE AND
WRITE TO
xPHCAL
04443-078
SELECT PHASE
FOR LINE PERIOD
MEASUREMENT
CONFIGURE
FREQ[11:0] FOR A
LINE PERIOD
MEASUREMENT
°
×
μ
×
=
360
]0:11[
__
6.9 FREQ
WeightLSBPHCAL
s
ErrorPhase
xPHCAL
(55)
Example: Phase Calibration of Phase A Using Pulse Output
For this example, ITEST = 10 A, VNOM = 220 V, VFULLSCALE = 500 V,
IFULLSCALE = 130 A, MC = 3200 impulses/kWh, Power Factor =
0.5 inductive, and Frequency = 50 Hz.
With Phase A contributing to CF, at ITEST, VNOM, and 0.5
inductive power factor, the example ADE7758 meter shows
0.9668 Hz on the pulse output. This is equivalent to −1.122%
error from the reference meter value using Equation 49.
The Phase Error in degrees using Equation 53 is 0.3713°.
()
°=
⎟
⎠
⎞
⎜
⎝
⎛
×
=° 3713.0
3%100
1.122%–
–ArcsinErrorPhase
If at 50 Hz the FREQ register = 2083d, the value that should be
written to APHCAL is 17d, or 0x11 using Equation 55. Note
that a PHCAL_LSB_Weight of 1.2 s is used because the
%Error is negative.
1101719.17
360
2083
s2.1
s6.9
3713.0 xAPHCAL ===
°
××°=
Figure 78. Phase Calibration Using Pulse Output
Power Offset Calibration Using Pulse Output
Step 1: Step 1 and Step 3 from the gain calibration should be
repeated to configure the ADE7758 pulse output. Ensure the
xPHCAL registers are zero.
Power offset calibration should be used for outstanding
performance over a wide dynamic range (1000:1). Calibration
of the power offset is done at or close to the minimum current
where the desired accuracy is required.
Step 2: Set the test system for ITEST, VNOM, and 0.5 power factor
inductive.
Step 3: Measure the percent error in the pulse output, APCF,
from the reference meter using Equation 49.
Step 4: Calculate the Phase Error in degrees by
()
⎟
⎠
⎞
⎜
⎝
⎛
×
=° 3%100
–%Error
ArcsinErrorPhase (53)
The ADE7758 has power offset registers for watts and VAR
(xWATTOS and xVAROS). Offsets in the VA measurement are
compensated by adjusting the rms offset registers (see the
Calibration of IRMS and VRMS Offset section). Figure 79
shows the steps to calibrate the power offsets using the pulse
outputs.
ADE7758
Rev. D | Page 47 of 72
START
STEP 1
ENABLE CF
OUTPUTS
STEP 2
CLEAR OFFSET
REGISTERS
xWATTOS, xVAROS
ALL PHASES
WAT T OFFSET
CALIBRATED?
YES NO
ALL PHASES
VAR OF FSE T
CALIBRATED?
YES NO
SET UP APCF
PULSE O UT P UT
FOR PHASE A, B,
OR C
STEP 4
STEP 3
SET UP SYSTEM
FOR IMIN,V
NOM,
PF = 1
STEP 5
MEASURE %
ERROR FOR
APCF
STEP 6
CALCULATE AND
WRITE TO
xWATTOS
END SET UP VARCF
PULSE OUTPUT
FOR PHASE A, B,
OR C
STEP 4
STEP 3
SET UP SYSTEM
FOR IMIN,V
NOM,
PF = 0, INDUCTIVE
STEP 5
MEASURE %
ERROR FOR VARCF
MEASURE PERIOD
USING FREQ[11:0]
REGISTER
STEP 6
CALCULATE AND
WRITE TO
xVAROS
STEP 7.
REPEAT STEP
3TOSTEP6
FOR xVAROS
SELECT PHASE
FOR LINE PERIOD
MEASUREMENT
CONFIGURE
FREQ[11:0] FOR A
LINE PERIOD
MEASUREMENT
04443-079
Figure 79. Offset Calibration Using Pulse Output
Step 1: Repeat Step 1 and Step 3 from the gain calibration to
configure the ADE7758 pulse output.
Step 2: Clear the xWATTOS and xVAROS registers.
Step3: Disable the Phase B and Phase C contribution to the APCF
and VARCF pulses. This is done by the TERMSEL[2:4] bits of
the COMPMODE register (0x16). Setting Bit 2 to Logic 1 and
Bit 3 and Bit 4 to Logic 0 allows only Phase A to be included in
the pulse outputs. Select Phase A, Phase B, or Phase C for a line
period measurement with the FREQSEL[1:0] bits in the MMODE
register (0x14). For example, clearing Bit 1 and Bit 0 selects
Phase A for line period measurement.
Step 4: Set the test system for IMIN, VNOM, and unity power factor.
For Step 6, set the test system for IMIN, VNOM, and zero-power
factor inductive.
Step 5: Measure the percent error in the pulse output, APCF or
VARCF, from the reference meter using Equation 49.
Step 6: Calculate xWATTOS using Equation 56 (for xVAROS
use Equation 57).
APCFNUM
APCFDEN
Q
APCF
%APCF
xWATTOS
EXPECTED
ERROR ××
⎟
⎠
⎞
⎜
⎝
⎛×
=
4
2
%100
– (56)
ADE7758
Rev. D | Page 48 of 72
VARCFNU
M
VARCFDEN
Q
VARCF
%VARCF
xVAROS
EXPECTED
ERROR ××
⎟
⎠
⎞
⎜
⎝
⎛×
=
4
2
%100
– (57)
where Q is defined in Equation 58 and Equation 59.
For xWATTOS,
4
1
2
1
425 ××= CLKIN
Q (58)
For xVAROS,
4
1
4
0]:[11
202
2
1
424 ×
⎟
⎠
⎞
⎜
⎝
⎛
××= FREQ
CLKIN
Q (59)
where the FREQ (0x10) register is configured for line period
measurements.
Step 7: Repeat Step 3 to Step 6 for xVAROS calibration.
Example: Offset Calibration of Phase A Using Pulse Output
For this example, IMIN = 50 mA, VNOM = 220 V, VFULLSCALE =
500 V, IFULLSCALE = 130 A, MC = 3200 impulses/kWh, Power
Factor = 1, Frequency = 50 Hz, and CLKIN = 10 MHz.
With IMIN, VNOM, and unity power factor, the example ADE7758
meter shows 0.009789 Hz on the APCF pulse output. When the
power factor is changed to 0.5 inductive, the VARCF output is
0.009769 Hz.
This is equivalent to 0.1198% for the watt measurement and
−0.0860% for the VAR measurement. Using Equation 56
through Equation 59, the values 0xFFD and 0x3 should be
written to AWATTOS (0x39) and AVAROS (0x3C), respectively.
0xFFD3– –2.8
1
277
0.01863
2
0.009778
%100
0.1198%
–4===××
⎟
⎠
⎞
⎜
⎝
⎛×
=AWATTOS
32.6
1
277
0.01444
2
0.009778
%100
0.0860%–
–4==××
⎟
⎠
⎞
⎜
⎝
⎛×=AVAROS
For AWATTOS,
01863.0
4
1
2
1
4
610
25 =××= E
Q
For AVAROS,
0.01444
4
1
4
2083
202
2
1
4
610
24 =×××= E
Q
Calibration Using Line Accumulation
Line cycle accumulation mode configures the nine energy
registers such that the amount of energy accumulated over an
integer number of half line cycles appears in the registers after
the LENERGY interrupt. The benefit of using this mode is that
the sinusoidal component of the active energy is eliminated.
Figure 80 shows a flowchart of how to calibrate the ADE7758
using the line accumulation mode. Calibration of all phases and
energies can be done simultaneously using this mode to save
time during calibration.
START
CAL IRMS OFFSET
CAL VRMS OFFSET
CAL WATT AND VA
GAIN ALL PHASES
@ PF = 1
CAL VAR GAIN ALL
PHASES @ PF = 0,
INDUCTIVE
CALIBRATE PHASE
ALL PHASES
@ PF = 0.5,
INDUCTIVE
CALIBRATE ALL
PHASES WATT
OFFSET @ I
MIN
AND
PF = 1
CALIBRATE ALL
PHASES VAR
OFFSETS @ I
MIN
AND PF = 0,
INDUCTIVE
END
04443-080
Figure 80. Calibration Using Line Accumulation
ADE7758
Rev. D | Page 49 of 72
Gain Calibration Using Line Accumulation Step 2: Select Phase A, Phase B, or Phase C for a line period
measurement with the FREQSEL[1:0] bits in the MMODE
register (0x14). For example, clearing Bit 1 and Bit 0 selects
Phase A for line period measurement.
Gain calibration is used for meter-to-meter gain adjustment,
APCF or VARCF output rate calibration, and determining the
Wh/LSB, VARh/LSB, and VAh/LSB constant.
Step 0: Before performing the gain calibration, the APCFNUM/
APCFDEN (0x45/0x46) and VARCFNUM/ VARCFDEN
(0x47/0x48) values can be set to achieve the correct impulses/kWh,
impulses/kVAh, or impulses/kVARh using the same method
outlined in Step 4 in the Gain Calibration Using Pulse Output
section. The calibration of xWG/xVARG/xVAG (0x2A through
0x32) is done with the line accumulation mode. Figure 81 shows
the steps involved in calibrating the gain registers using the line
accumulation mode.
Step 3: Set up ADE7758 for line accumulation by writing 0xBF
to LCYCMODE. This enables the line accumulation mode on
the xWATTHR, xVARHR, and xVAHR (0x01 to 0x09) registers
by setting the LWATT, LVAR, and LVA bits, LCYCMODE[0:2]
(0x17), to Logic 1. It also sets the ZXSEL bits, LCYCMODE[3:5],
to Logic 1 to enable the zero-crossing detection on all phases for
line accumulation. Additionally, the FREQSEL bit,
LCYCMODE[7], is set so that FREQ (0x10) stores the line
period. When using the line accumulation mode, the
RSTREAD bit of LCYCMODE should be set to 0 to disable the
read with reset mode. Select the phase for line period
measurement in MMODE[1:0].
Step 1: Clear xWG, xVARG, and xVAG.
Step 4: Set the number of half-line cycles for line accumulation
by writing to LINECYC (0x1C).
FREQUENCY
KNOWN?
NO YES
STEP 0 SET
APCFNUM/APCFDEN
AND
VARCFNUM/VARCFDEN
STEP 1
STEP 2
CLEAR
xWG/xVAR/xVAG
STEP 3
SET LYCMODE
REGISTER
STEP 4
SET ACCUMULATION
TIME (LINECYC)
STEP 5
SET MASK FOR
LENE RGY INTERRUPT
STEP 6
SET UP SYSTEM FOR
I
TEST
,V
NOM
,PF=1
STEP 7
READ FREQ[11:0]
REGISTER
STEP 8
RESET STATUS
REGISTER
STEP 9
READ ALL xWATTHR
AND xVAHR AFTER
LENERGY
INTERRUPT
STEP 9A
CALCULATE xWG
STEP 9B
CALCULATE xVAG
STEP 10
WRITE TO xWG AND
xVAG
CALIBRATE WATT
AND VA @ PF = 1
STEP 11
SET UP TEST
SYSTEM FOR
I
TEST
,V
NOM
,
PF = 0, INDUCTIV E
STEP 12
RESET STATUS
REGISTER
STEP 13
READ ALL xVARHR
AFTER LENERGY
INTERRUPT
STEP 14
CALCULATE xVARG
STEP 15
WRITE TO xVARG
STEP 16
CALCULATE
Wh/LSB, VAh/LSB,
VARh/LSB
END
04443-081
SELECT PHASE
FOR LINE PERIOD
MEASUREMENT
CONFIGURE
FREQ[11:0] FOR A
LINE PERIOD
MEASUREMENT
Figure 81. Gain Calibration Using Line Accumulation
ADE7758
Rev. D | Page 50 of 72
Step 5: Set the LENERGY bit, MASK[12] (0x18), to Logic 1
to enable the interrupt signaling the end of the line cycle
accumulation.
Step 6: Set the test system for ITEST, VNOM, and unity power factor
(calibrate watt and VA simultaneously and first).
Step 7: Read the FREQ (0x10) register if the line frequency is
unknown.
Step 8: Reset the interrupt status register by reading
RSTATUS (0x1A).
Step 9: Read all six xWATTHR (0x01 to 0x03) and xVAHR
(0x07 to 0x09) energy registers after the LENERGY interrupt
and store the values.
Step 9a: Calculate the values to be written to xWG registers
according to the following equations:
()
WDI
V
APCFNU
M
APCFDEN
AccumTimeθcosVIMC
WATTHR
NOM
TEST
EXPECTED
1
36001000
4
×
×
×
×××××
=
(60)
where AccumTime is
[]
SelectedPhasesofNo.FrequencyLine
:LINECYC
××2
015 (61)
where:
MC is the meter constant.
θ is the angle between the current and voltage.
Line Frequency is known or calculated from the FREQ[11:0]
register. With the FREQ[11:0] register configured for line period
measurements, the line frequency is calculated with Equation 62.
6-
109.60]:[11
1
××
=FREQ
FrequencyLine (62)
No. of Phases Selected is the number of ZXSEL bits set to Logic 1
in LCYCMODE (0x17).
Then, xWG is calculated as
12
21 ×
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−=
MEASURED
EXPECTED
WATTHR
WATTHR
xWG (63)
Step 9b: Calculate the values to be written to the xVAG registers
according to the following equation:
VADIVVARCFNUM
VARCFDEN
AccumTimeVIMC
VAHR
NOM
TEST
EXPECTED
1
36001000
4××
×
××××
=
(64)
12
21 ×
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−=
MEASURED
EXPECTED
VAHR
VAHR
xVAG
Step 10: Write to xWG and xVAG.
Step 11: Set the test system for ITEST, VNOM, and zero power
factor inductive to calibrate VAR gain.
Step 12: Repeat Step 7.
Step 13: Read the xVARHR (0x04 to 0x06) after the LENERGY
interrupt and store the values.
Step 14: Calculate the values to be written to the xVARG
registers (to adjust VARCF to the expected value).
()
VARDIVVARCFNU
M
VARCFDEN
AccumTimeθsinVIMC
VARHR
NOM
TEST
EXPECTED
1
36001000
4
×
×
×
×××××
=
(65)
12
21 ×
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−=
MEASURED
EXPECTED
VARHR
VARHR
xVARG
Step 15: Write to xVARG.
Step 16: Calculate the Wh/LSB, VARh/LSB, and VAh/LSB
constants.
()
xWATTHR
AccumTimeθcosVI
LSB
Wh NOM
TEST
×
×
×
×
=3600 (66)
xVAHR
AccumTimeVI
LSB
VAh NOM
TEST
×
×
×
=3600 (67)
()
xVARHR
AccumTimeθsinVI
LSB
VARh NOM
TEST
×
×
×
×
=3600 (68)
Example: Watt Gain Calibration Using Line Accumulation
This example shows only Phase A watt calibration. The steps
outlined in the Gain Calibration Using Line Accumulation
section show how to calibrate watt, VA, and VAR. All three
phases can be calibrated simultaneously because there are nine
energy registers.
For this example, ITEST = 10 A, VNOM = 220 V, Power Factor = 1,
Frequency = 50 Hz, LINECYC (0x1C) is set to 0x800, and MC =
3200 imp/kWhr.
ADE7758
Rev. D | Page 51 of 72
To set APCFNUM (0x45) and APCFDEN (0x46) to the
calculated value to perform a coarse adjustment on the
imp/kW-hr ratio, use Equation 45 to Equation 47.
kHz5415.0
130
10
500
220
kH16 =××= zAPCFNOMINAL
()
Hz1.956cos
36001000
220103200 =θ×
×
××
=
EXPECTED
APCF
277
Hz956.1
Hz5.541
INT =
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
=
APCFDEN
Under the test conditions above, the AWATTHR register value
is 15559d after the LENERGY interrupt. Using Equation 60 and
Equation 61, the value to be written to AWG is −199d, 0xF39.
[]
SelectedPhasesofNo.
FREQ
:LINECYC
AccumTime ×
××
×
=
−6
106.9]0:11[
1
2
015
6.832128s
3
106.92085
1
2
8000
6
=
×
××
×
=
−
x
AccumTime
148041
1
277
36001000
832.612201032004 =××
×
×××××
=
EXPECTED
WATTHR
0xF39–199–198.8764021
15559
14804 12 ===×
⎟
⎠
⎞
⎜
⎝
⎛−=xWG
Using Equation 66, the Wh/LSB constant is
00.000282
148043600
832.622010 =
×
××
=
LSB
Wh
Phase Calibration Using Line Accumulation
The ADE7758 includes a phase calibration register on each
phase to compensate for small phase errors. Large phase errors
should be compensated by adjusting the antialiasing filters. The
ADE7758’s phase calibration is a time delay with different weights
in the positive and negative direction (see the Phase Compensation
section). Because a current transformer is a source of phase
error, a fixed nominal value can be decided on to load into the
xPHCAL (0x3F to 0x41) registers at power-up. During
calibration, this value can be adjusted for CT-to-CT error.
Figure 82 shows the steps involved in calibrating the phase
using the line accumulation mode.
STEP 1
SET LCYCMODE,
LINECYC AND MASK
REGISTERS
STEP 2
SET UP SYSTEM FOR
ITEST,V
NOM, PF = 0.5,
INDUCTIVE
STEP 3
RESET STATUS
REGISTER
STEP 4
READ ALL xWATTHR
REGISTERS AFTER
LENERGY
INTERRUPT
STEP 5
CALCULATE PHASE
ERROR IN DE GREES
FOR ALL PHASES
STEP 6
CALCULATE AND
WRITE TO ALL
xPHCAL REGISTERS
04443-082
Figure 82. Phase Calibration Using Line Accumulation
Step 1: If the values were changed after gain calibration, Step 1,
Step 3, and Step 4 from the gain calibration should be repeated
to configure the LCYCMODE and LINECYC registers.
Step 2: Set the test system for ITEST, VNOM, and 0.5 power factor
inductive.
Step 3: Reset the interrupt status register by reading RSTATUS
(0x1A).
Step 4: The xWATTHR registers should be read after the
LENERGY interrupt. Measure the percent error in the energy
register readings (AWATTHR, BWATTHR, and CWATTHR)
compared to the energy register readings at unity power factor
(after gain calibration) using Equation 69. The readings at unity
power factor should have been repeated after the gain calibration
and stored for use in the phase calibration routine.
2
2
–
1PF
1PF
5PF
=
=
=
=xWATTHR
xWATTHR
xWATTHR
Error (69)
Step 5: Calculate the Phase Error in degrees using the equation
()
⎟
⎠
⎞
⎜
⎝
⎛
=° 3
–Error
ArcsinErrorPhase (70)
Step 6: Calculate xPHCAL and write to the xPHCAL registers
(0x3F to 0x41).
°
×××
=
360
1
)(
1
__
1
sPeriodLineWeightLSBPHCAL
ErrorPhase
xPHCAL
(71)
ADE7758
Rev. D | Page 52 of 72
With ITEST, VNOM, and 0.5 inductive power factor, the example
ADE7758 meter shows 7318d in the AWATTHR (0x01) register.
For unity power factor (after gain calibration), the meter shows
14804d in the AWATTHR register. This is equivalent to
−1.132% error.
where PHCAL_LSB_Weight is 1.2 s if the %Error is negative
or 2.4 s if the %Error is positive (see the Phase Compensation
section).
If it is not known, the line period is available in the ADE7758’s
frequency register, FREQ (0x10). To configure line period
measurement, select the phase for period measurement in the
MMODE[1:0] and set LCYCMODE[7]. Equation 72 shows how
to determine the value that needs to be written to xPHCAL
using the period register measurement.
%132.101132.0
2
148042
14804
–7318 −=−==Error
The Phase Error in degrees using Equation 66 is 0.374°.
°
××
=
360
]0:11[
__
s6.9 FREQ
WeightLSBPHCAL
ErrorPhase
xPHCAL
(72)
()
°=
⎟
⎠
⎞
⎜
⎝
⎛−
=° 374.0
3
01132.0
sin–ArcErrorPhase
Example: Phase Calibration Using Line Accumulation Using Equation 72, the value written to APHCAL (0x3F), if at
50 Hz, the FREQ (0x10) register = 2085d, is 17d. Note that a
PHCAL_LSB_Weight of 1.2 µs is used because the %Error is
negative.
This example shows only Phase A phase calibration. All three
PHCAL registers can be calibrated simultaneously using the
same method.
11x017
360
2085
2.1
6.9
374.0 ==××°=APHCAL
For this example, ITEST = 10 A, VNOM = 220 V, Power Factor = 0.5
inductive, and Frequency = 50 Hz. Also, LINECYC = 0x800.
STEP 1
SET MMODE,
LCYCMODE,
LINECYC AND
MASK REGISTERS
STEP 2
SET UP SYSTEM
FOR I
MIN
,V
NOM
@PF=1
STEP 3
RESET STATUS
REGISTER
STEP 4
READ ALL
xWATTHR
REGISTERS AFTER
LENERGY
INTERRUPT
END
FOR STEP 8
READ ALL xVARHR
AFTER LENERGY
INTERRUPT
FOR STEP 8, CALCULATE
xVAROS FOR ALL PHASES
STEP 5
CALCULATE
xWATTOS FOR ALL
PHASES
FOR STEP 8, WRITE TO
ALL xVAROS REGISTERS
STEP 6
WRITE TO ALL
xWATTOS
REGISTERS
STEP 7
SET UP SYSTEM
FOR I
TEST
,V
NOM
@
PF = 0, I NDUCTIVE
STEP 8
REPEAT STEP 3 TO
STEP 8 FOR
xVARHR, xVAROS
CALIBRATION
04443-083
Figure 83. Power Offset Calibration Using Line Accumulation
ADE7758
Rev. D | Page 53 of 72
Power Offset Calibration Using Line Accumulation
Power offset calibration should be used for outstanding
performance over a wide dynamic range (1000:1). Calibration
of the power offset is done at or close to the minimum current.
The ADE7758 has power offset registers for watts and VAR,
xWATTOS (0x39 to 0x3B) and xVAROS (0x3C to 0x3E). Offsets in
the VA measurement are compensated by adjusting the rms offset
registers (see the Calibration of IRMS and VRMS Offset section).
More line cycles could be required at the minimum current to
minimize the effect of quantization error on the offset
calibration. For example, if a current of 40 mA results in an
active energy accumulation of 113 after 2000 half line cycles,
one LSB variation in this reading represents an 0.8% error. This
measurement does not provide enough resolution to calibrate
out a <1% offset error. However, if the active energy is
accumulated over 37,500 half line cycles, one LSB variation
results in 0.05% error, reducing the quantization error.
Figure 83 shows the steps to calibrate the power offsets using
the line accumulation mode.
Step 1: If the values change after gain calibration, Step 1, Step 3,
and Step 4 from the gain calibration should be repeated to
configure the LCYCMODE, LINECYC, and MASK registers.
Select Phase A, Phase B, or Phase C for a line period measure-
ment with the FREQSEL[1:0] bits in the MMODE register (0x14).
For example, clearing Bit 1 and Bit 0 selects Phase A for line
period measurement.
Step 2: Set the test system for IMIN, VNOM, and unity power factor.
Step 3: Reset the interrupt status register by reading RSTATUS
(0x1A).
Step 4: Read all xWATTHR energy registers (0x01 to 0x03) after
the LENERGY interrupt and store the values.
Step 4a: If it is not known, the line period is available in the
ADE7758’s frequency register, FREQ (0x10). To configure line
period measurement, select the phase for period measurement
in the MMODE[1:0] and set LCYCMODE[7].
Step 5: Calculate the value to be written to the xWATTOS
registers according to the following equations:
TEST
MIN
MIN
ITEST
IMIN
I
TEST
I
II
I
LINECYC
LINECYC
xWATTHRIxWATTHR
Offset
TEST
MIN
–
–×
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛××
=
(73)
[]
29
2
4
0:11 ×
×
×
=CLKINAccumTime
Offset
xWATTOS (74)
where:
AccumTime is defined in Equation 58.
is the value in the energy register at ITEST.
is the value in the energy register at IMIN.
LINECYCIMIN is the number of line cycles accumulated at IMIN.
LINECYCIMAX is the number of line cycles accumulated at IMAX.
TEST
I
xWATTHR
MIN
I
xWATTHR
Step 6: Write to all xWATTOS registers (0x39 to 0x3B).
Step 7: Set the test system for IMIN, VNOM, and zero power factor
inductive to calibrate VAR gain.
Step 8: Repeat Steps 3, 4, and 5.
Step 9: Calculate the value written to the xVAROS registers
according to the following equations:
TEST
MIN
MIN
ITEST
IMIN
I
TEST
I
II
I
LINECYC
LINECYC
xVARHRIxVARHR
Offset
TEST
MIN
–
–×
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛××
=
(75)
26
2
202
]0:11[
4×FREQ
Offset
0]:[11 ××
×
=CLKINAccumTime
xVAROS (76)
where the FREQ[11:0] register is configured for line period
Power Offset Calibration Using Line Accumulation
MIN ITEST = 10 A, VNOM = 220 V,
Wh,
ing r ga ion at
d in the
readings.
Example:
This example only shows Phase A of the phase active power
offset calibration. Both active and reactive power offset for
all phases can be calibrated simultaneously using the method
explained in the Power Offset Calibration Using Line
Accumulation section.
For this example, I = 50 mA,
VFULLSCALE = 500 V, IFULLSCALE = 130 A, MC = 3200 impulses/k
Power Factor = 1, Frequency = 50 Hz, and CLKIN = 10 MHz.
Also, LINECYCITEST = 0x800 and LINECYCIMIN = 0x4000.
After accumulat over 0x800 line cycles fo in calibrat
ITEST, the example ADE7758 meter shows 14804d in the
AWATTHR (0x01) register. At IMIN, the meter shows 592
AWATTHR register. By using Equation 73, this is equivalent to
0.161 LSBs of offset; therefore, using Equation 61 and Equation 74,
the value written to AWATTOS is 0d.
0.16
10–0.05
0.05
0x800
0x4000
14804–10592
=
×
⎟
⎠
⎞
⎜
⎝
⎛××
=
Offse
t
ADE7758
Rev. D | Page 54 of 72
s64.45
3
106.92085
1
2
40000
6
=
×
××
×
×
=
−
AccumTime The low-pass filter used to obtain the rms measurements is not
ideal; therefore, it is recommended to synchronize the readings
with the zero crossings of the voltage waveform and to average a
few measurements when reading the rms registers.
00.0882
MHz1054.64
40.161 29 =−=×
×
×
=AWATTOS The ADE7758 IRMS measurement is linear over a 500:1 range,
and the VRMS measurement is linear over a 20:1 range. To
measure the voltage VRMS offset (xVRMSOS), measure rms
values at two different nonzero current levels, for example,
VNOM and VFULLSCALE/20.
Calibration of IRMS and VRMS Offset
noise and offset
e
ffset
IRMSOS and VRMSOS are used to cancel
contributions from the inputs. The calibration method is th
same whether calibrating using the pulse outputs or line
accumulation. Reading the registers is required for this
calibration because there is no rms pulse output. The rms o
calibration should be performed before VAGAIN calibration.
The rms offset calibration also removes offset from the VA
calculation. For this reason, no VA offset register exists in the
ADE7758.
To measure the current rms offset (IRMSOS), measure rms
values at two different nonzero current levels, for example, ITEST
and IFULLSCALE/500. This translates to two test conditions: ITEST
and VNOM, and IFULLSCALE/500 and VFULLSCALE/20. Figure 84 shows
a flowchart for calibrating the rms measurements.
STEP 1
SET
CONFIGURATION
REGISTERS FOR
ZERO CROSSING
ON ALL PHASES
STEP 2
SET I NT E RRUPT
MASK FOR
ZERO CROSSING
ON ALL PHASES
STEP 3
STEP 4
READ RMS
REGISTERS
STEP 5
WRITE TO
xVRMSOS
xIRMSOS
SET UP
SYSTEM FOR
ITEST,V
NOM
SET UP SYSTEM
FOR IFULLSCALE/500,
VFULLSCALE/20
START
TESTED
ALL
PHASES?
YES NO
TESTED
ALL
CONDITIONS?
1
2
STEP 4A
CHOOSE N
n=0
STEP 4D
READ
xIRMS
xVRMS
STEP 4E
CALCULATE THE
AVERAGE OF N
SAMPLES
STEP 4B
RESET INTERRUPT
STATUS REGISTER END
n=n+1
n=N?
NO YES
YES NO
STEP 4C
INTERRUPT?
04443-084
Figure 84. RMS Calibration Routine
ADE7758
Rev. D | Page 55 of 72
Step 1: Set configuration registers for zero crossings on all
phases by writing the value 0x38 to the LCYCMODE register
(0x17). This sets all of the ZXSEL bits to Logic 1.
Step 2: Set the interrupt mask register for zero-crossing
detection on all phases by writing 0xE00 to the MASK[0:24]
register (0x18). This sets all of the ZX bits to Logic 1.
Step 3: Set up the calibration system for one of the two test
conditions: ITEST and VNOM, and IFULLSCALE/500 and VFULLSCALE/20.
Step 4: Read the rms registers after the zero-crossing interrupt
and take an average of N samples. This is recommended to get
the most stable rms readings. This procedure is detailed in
Figure 84: Steps 4a through 4e.
Step 4a. Choose the number of samples, N, to be averaged.
Step 4b. Reset the interrupt status register by reading RSTATUS
(0x1A).
Step 4c. Wait for the zero-crossing interrupt. When the zero-
crossing interrupt occurs, move to Step 4d.
Step 4d. Read the xIRMS and xVRMS registers. These values
will be averaged in Step 4e.
Step 4e: Average the N samples of xIRMS and xVRMS. The
averaged values will be used in Step 5.
Step 5: Write to the xVRMSOS (0x33 to 0x35) and xIRMSOS
(0x36 to 0x38) registers according to the following equations:
()( )
22
2222
16384
1
TEST
MIN
ITEST
MINIMIN
TEST
I–I
IRMSI–IRMSI
xIRMSOS
××
×
=
(77)
where:
IMIN is the full scale current/500.
ITEST is the test current.
IRMSIMIN and IRMSITEST are the current rms register values
without offset correction for the inputs IMIN and ITEST,
respectively.
NOM
MIN
VNOM
MINVMIN
NOM
V–V
VRMSV–VRMSV
x
VRMSOS
××
×
=
64
1
(78)
where:
VMIN is the full scale voltage/20
VNOM is the nominal line voltage.
VRMSVMIN and VRMSVNOM are the voltage rms register values
without offset correction for the input VMIN and VNOM,
respectively.
Example: Calibration of RMS Offsets
For this example, ITEST = 10 A, IMAX = 100 A, VNOM = 220 V,
VFULLSCALE = 500 V, Power Factor = 1, and Frequency = 50 Hz.
Twenty readings are taken synchronous to the zero crossings of
all three phases at each current and voltage to determine the
average xIRMS and xVRMS readings. At ITEST and VNOM, the
example ADE7758 meter gets an average AIRMS (0x0A)
reading of 148242.2 and 744570.8 in the AVRMS (0x0D)
register. Then the current is set to IMIN = IFULLSCALE/500 or
260 mA. At IMIN, the average AIRMS reading is 3885.68. At
VMIN = VFULLSCALE/20 or 25 V, the example meter gets an average
AVRMS of 86362.36. Using this data, −15d is written to
AIRMSOS (0x36) and −31d is written to AVRMSOS (0x33)
registers according to the Equation 77 and Equation 78.
()(
() )
0xFF2158.14
10–260.0
148242.2260.0–3885.6810
16384
1
2
2222
=−=−
=
××
×
=
AIRMSOS
()()
()
0xFE1319.30
220–25
744570.825–86362.36220
64
1=−=−=
××
×
=
AVRMSOS
This example shows the calculations and measurements for
Phase A only. However, all three xIRMS and xVRMS registers
can be read simultaneously to compute the values for each
xIRMSOS and xVRMSOS register.
CHECKSUM REGISTER
The ADE7758 has a checksum register CHKSUM[7:0] (0x7E)
to ensure the data bits received in the last serial read operation
are not corrupted. The 8-bit checksum register is reset before
the first bit (MSB of the register to be read) is put on the DOUT
pin. During a serial read operation, when each data bit becomes
available on the rising edge of SCLK, the bit is added to the
checksum register. In the end of the serial read operation, the
contents of the checksum register are equal to the sum of all the
1s in the register previously read. Using the checksum register, the
user can determine if an error has occurred during the last read
operation. Note that a read to the checksum register also
generates a checksum of the checksum register itself.
DOUT
ADDR: 0x7E
CHECKSUM
REGISTER
CONTENT OF REGISTERS
(N-BYTES)
04443-085
Figure 85. Checksum Register for Serial Interface Read
ADE7758
Rev. D | Page 56 of 72
INTERRUPTS
The ADE7758 interrupts are managed through the interrupt
status register (STATUS[23:0], Address 0x19) and the interrupt
mask register (MASK[23:0], Address 0x18). When an interrupt
event occurs in the ADE7758, the corresponding flag in the
interrupt status register is set to a Logic 1 (see Table 24). If the
mask bit for this interrupt in the interrupt mask register is
Logic 1, then the IRQ logic output goes active low. The flag bits
in the interrupt status register are set irrespective of the state of
the mask bits. To determine the source of the interrupt, the
MCU should perform a read from the reset interrupt status
register with reset. This is achieved by carrying out a read from
RSTATUS, Address 0x1A. The IRQ output goes logic high on
completion of the interrupt status register read command (see
the section). When carrying out a read with
reset, the ADE7758 is designed to ensure that no interrupt
events are missed. If an interrupt event occurs just as the
interrupt status register is being read, the event is not lost, and
the
Interrupt Timing
IRQ logic output is guaranteed to go logic high for the
duration of the interrupt status register data transfer before
going logic low again to indicate the pending interrupt. Note
that the reset interrupt bit in the status register is high for only
one clock cycle, and it then goes back to 0.
USING THE INTERRUPTS WITH AN MCU
Figure 86 shows a timing diagram that illustrates a suggested
implementation of ADE7758 interrupt management using an
MCU. At time t1, the IRQ line goes active low indicating that
one or more interrupt events have occurred in the ADE7758.
The IRQ logic output should be tied to a negative-edge-
triggered external interrupt on the MCU. On detection of the
negative edge, the MCU should be configured to start executing
its interrupt service routine (ISR). On entering the ISR, all
interrupts should be disabled using the global interrupt mask
bit. At this point, the MCU external interrupt flag can be
cleared to capture interrupt events that occur during the current
ISR. When the MCU interrupt flag is cleared, a read from the
reset interrupt status register with reset is carried out. (This
causes the IRQ line to be reset logic high (t2); see the
section.) The reset interrupt status register contents are
used to determine the source of the interrupt(s) and hence the
appropriate action to be taken. If a subsequent interrupt event
occurs during the ISR (t3) that event is recorded by the MCU
external interrupt flag being set again.
Interrupt
Timing
On returning from the ISR, the global interrupt mask bit is
cleared (same instruction cycle) and the external interrupt flag
uses the MCU to jump to its ISR once again. This ensures that
the MCU does not miss any external interrupts. The reset bit in
the status register is an exception to this and is only high for one
clock cycle after a reset event.
INTERRUPT TIMING
The Serial Interface section should be reviewed before
reviewing this section. As previously described, when the IRQ
output goes low, the MCU ISR must read the interrupt status
register to determine the source of the interrupt. When reading
the interrupt status register contents, the IRQ output is set high
on the last falling edge of SCLK of the first byte transfer (read
interrupt status register command). The IRQ output is held
high until the last bit of the next 8-bit transfer is shifted out
(interrupt status register contents), as shown in . If an
interrupt is pending at this time, the
Figure 87
IRQ output goes low again.
If no interrupt is pending, the IRQ output remains high.
SERIAL INTERFACE
The ADE7758 has a built-in SPI interface. The serial interface
of the ADE7758 is made of four signals: SCLK, DIN, DOUT,
and CS. The serial clock for a data transfer is applied at the
SCLK logic input. This logic input has a Schmitt trigger input
structure that allows slow rising (and falling) clock edges to be
used. All data transfer operations are synchronized to the serial
clock. Data is shifted into the ADE7758 at the DIN logic input
on the falling edge of SCLK. Data is shifted out of the ADE7758
at the DOUT logic output on a rising edge of SCLK.
The CS logic input is the chip select input. This input is used
when multiple devices share the serial bus. A falling edge on CS
also resets the serial interface and places the ADE7758 in
communications mode.
The CS input should be driven low for the entire data transfer
operation. Bringing CS high during a data transfer operation
aborts the transfer and places the serial bus in a high impedance
state. The CS logic input can be tied low if the ADE7758 is the
only device on the serial bus.
However, with CS tied low, all initiated data transfer operations
must be fully completed. The LSB of each register must be
transferred because there is no other way of bringing the
ADE7758 back into communications mode without resetting
the entire device, that is, performing a software reset using Bit 6
of the OPMODE[7:0] register, Address 0x13.
The functionality of the ADE7758 is accessible via several on-
chip registers (see Figure 88). The contents of these registers can
be updated or read using the on-chip serial interface. After a
falling edge on CS, the ADE7758 is placed in communications
mode. In communications mode, the ADE7758 expects the first
communication to be a write to the internal communications
register. The data written to the communications register
contains the address and specifies the next data transfer to be a
read or a write command. Therefore, all data transfer operations
with the ADE7758, whether a read or a write, must begin with a
write to the communications register.
ADE7758
Rev. D | Page 57 of 72
GLOBAL
INTERRUPT
MASK
ISR RETURN
GLOBAL INTERRUPT
MASK RESET
CLEAR MCU
INTERRUPT
FLAG
READ
STATUS WITH
RESET (0x1A)
ISR ACTION
(BASED ON STATUS CONTENTS)
MCU
INTERRUPT
FLAG SET
PROGRAM
SEQUENCE
t
1
t
2
t
3
JUMP
TO
ISR
JUMP
TO
ISR
IRQ
04443-086
Figure 86. ADE7758 Interrupt Management
STATUS REGISTER CONTENTS
SCLK
DIN
DOUT
READ STATUS REGISTER COMMAND
t
1
CS
0001 000
DB15 DB8 DB7 DB0
1
t
9
t
11
t
12
IRQ
04443-087
Figure 87. ADE7758 Interrupt Timing
COMMUNICATIONS
REGISTER
IN
OUT
IN
OUT
IN
OUT
IN
OUT
IN
OUT
REGISTER NO. 1
REGISTER NO. 2
REGISTER NO. 3
REGISTER NO. n–1
REGISTER NO. n
REGISTER
ADDRESS
DECODE
DIN
DOUT
04443-088
Figure 88. Addressing ADE7758 Registers via the Communications Register
The communications register is an 8-bit, write-only register.
The MSB determines whether the next data transfer operation
is a read or a write. The seven LSBs contain the address of the
register to be accessed (see Table 16).
Figure 89 and Figure 90 show the data transfer sequences for a
read and write operation, respectively.
MULTIBYTE
COMMUNICATIONS REGISTER WRITE
DIN
SCLK
DOUT READ DATA
ADDRESS0
CS
04443-089
Figure 89. Reading Data from the ADE7758 via the Serial Interface
COMMUNICATIONS REGISTER WRITE
DIN
S
CLK
ADDRESS1
CS
MULTIBYTE READ DATA
04443-090
Figure 90. Writing Data to the ADE7758 via the Serial Interface
On completion of a data transfer (read or write), the ADE7758
once again enters into communications mode, that is, the next
instruction followed must be a write to the communications
register.
A data transfer is completed when the LSB of the ADE7758
register being addressed (for a write or a read) is transferred to
or from the ADE7758.
SERIAL WRITE OPERATION
The serial write sequence takes place as follows. With the
ADE7758 in communications mode and the CS input logic low,
a write to the communications register takes place first. The
MSB of this byte transfer must be set to 1, indicating that the
next data transfer operation is a write to the register. The seven
LSBs of this byte contain the address of the register to be written
to. The ADE7758 starts shifting in the register data on the next
falling edge of SCLK. All remaining bits of register data are
shifted in on the falling edge of the subsequent SCLK pulses
(see ). Figure 91
ADE7758
Rev. D | Page 58 of 72
As explained earlier, the data write is initiated by a write to the
communications register followed by the data. During a data
write operation to the ADE7758, data is transferred to all on-
chip registers one byte at a time. After a byte is transferred into
the serial port, there is a finite time duration before the content
in the serial port buffer is transferred to one of the ADE7758
on-chip registers. Although another byte transfer to the serial
port can start while the previous byte is being transferred to the
destination register, this second-byte transfer should not finish
until at least 900 ns after the end of the previous byte transfer.
This functionality is expressed in the timing specification t6 (see
Figure 91). If a write operation is aborted during a byte transfer
(CS brought high), then that byte is not written to the
destination register.
Destination registers can be up to 3 bytes wide (see the
Accessing the On-Chip Registers section). Therefore, the first
byte shifted into the serial port at DIN is transferred to the most
significant byte (MSB) of the destination register. If the
destination register is 12 bits wide, for example, a two-byte data
transfer must take place. The data is always assumed to be right
justified; therefore, in this case, the four MSBs of the first byte
would be ignored, and the four LSBs of the first byte written to
the ADE7758 would be the four MSBs of the 12-bit word.
Figure 92 illustrates this example.
DIN
SCLK
CS
t
2
t
3
t
1
t
4
t
5
t
7
t
6
t
8
COMMAND BYTE MOST SIGNIFICANT BYTE LEAST SIGNIFICANT BYTE
1A6 A4A5 A3 A2 A1 A0 DB7 DB0 DB7 DB0
t
7
04443-091
Figure 91. Serial Interface Write Timing Diagram
SCLK
DIN X X X X DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
MOST SIGNIFICANT BYTE LEAST SIGNIFICANT BYTE
04443-092
Figure 92. 12-Bit Serial Write Operation
SCLK
CS
t
1
t
10
t
13
0A6 A4A5 A3 A2 A1 A0
DB0
DB7 DB0DB7
DIN
DOUT
t
11
t
12
COMMAND BYTE MOST SIGNIFICANT BYTE LEAST SIGNIFICANT BYTE
t
9
04443-093
Figure 93. Serial Interface Read Timing Diagram
ADE7758
Rev. D | Page 59 of 72
SERIAL READ OPERATION
During a data read operation from the ADE7758, data is shifted
out at the DOUT logic output on the rising edge of SCLK. As
was the case with the data write operation, a data read must be
preceded with a write to the communications register.
With the ADE7758 in communications mode and CS logic low,
an 8-bit write to the communications register takes place first.
The MSB of this byte transfer must be a 0, indicating that the
next data transfer operation is a read. The seven LSBs of this
byte contain the address of the register that is to be read. The
ADE7758 starts shifting out of the register data on the next
rising edge of SCLK (see ). At this point, the DOUT
logic output switches from a high impedance state and starts
driving the data bus. All remaining bits of register data are
shifted out on subsequent SCLK rising edges. The serial
interface enters communications mode again as soon as the
read is completed. The DOUT logic output enters a high
impedance state on the falling edge of the last SCLK pulse.
Figure 93
The read operation can be aborted by bringing the CS logic
input high before the data transfer is completed. The DOUT
output enters a high impedance state on the rising edge of CS.
When an ADE7758 register is addressed for a read operation,
the entire contents of that register are transferred to the serial
port. This allows the ADE7758 to modify its on-chip registers
without the risk of corrupting data during a multibyte transfer.
Note that when a read operation follows a write operation, the
read command (that is, write to communications register)
should not happen for at least 1.1 µs after the end of the write
operation. If the read command is sent within 1.1 µs of the write
operation, the last byte of the write operation can be lost.
ACCESSING THE ON-CHIP REGISTERS
All ADE7758 functionality is accessed via the on-chip registers.
Each register is accessed by first writing to the communications
register and then transferring the register data. For a full
description of the serial interface protocol, see the Serial
Interface section.
ADE7758
Rev. D | Page 60 of 72
REGISTERS
COMMUNICATIONS REGISTER
The communications register is an 8-bit, write-only register that
controls the serial data transfer between the ADE7758 and the
host processor. All data transfer operations must begin with a
write to the communications register.
The data written to the communications register determines
whether the next operation is a read or a write and which
register is being accessed.
Table 16 outlines the bit designations for the communications
register.
Table 16. Communications Register
Bit Location Bit Mnemonic Description
0 to 6 A0 to A6 The seven LSBs of the communications register specify the register for the data transfer operation.
Table 17 lists the address of each ADE7758 on-chip register.
7 W/R When this bit is a Logic 1, the data transfer operation immediately following the write to the
communications register is interpreted as a write to the ADE7758. When this bit is a Logic 0, the data
transfer operation immediately following the write to the communications register is interpreted as a
read operation.
DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
W/R A6 A5 A4 A3 A2 A1 A0
Table 17. ADE7758 Register List
Address
[A6:A0] Name R/W1Length Type2
Default
Value Description
0x00 Reserved – Reserved.
0x01 AWATTHR R 16 S 0 Watt-Hour Accumulation Register for Phase A. Active power is
accumulated over time in this read-only register. The AWATTHR register
can hold a maximum of 0.52 seconds of active energy information with
full-scale analog inputs before it overflows (see the Active Energy
Calculation section). Bit 0 and Bit 1 of the COMPMODE register determine
how the active energy is processed from the six analog inputs.
0x02 BWATTHR R 16 S 0 Watt-Hour Accumulation Register for Phase B.
0x03 CWATTHR R 16 S 0 Watt-Hour Accumulation Register for Phase C.
0x04 AVARHR R 16 S 0 VAR-Hour Accumulation Register for Phase A. Reactive power is
accumulated over time in this read-only register. The AVARHR register
can hold a maximum of 0.52 seconds of reactive energy information
with full-scale analog inputs before it overflows (see the Reactive Energy
Calculation section). Bit 0 and Bit 1 of the COMPMODE register
determine how the reactive energy is processed from the six analog
inputs.
0x05 BVARHR R 16 S 0 VAR-Hour Accumulation Register for Phase B.
0x06 CVARHR R 16 S 0 VAR-Hour Accumulation Register for Phase C.
0x07 AVAHR R 16 S 0 VA-Hour Accumulation Register for Phase A. Apparent power is
accumulated over time in this read-only register. The AVAHR register can
hold a maximum of 1.15 seconds of apparent energy information with
full-scale analog inputs before it overflows (see the Apparent Energy
Calculation section). Bit 0 and Bit 1 of the COMPMODE register determine
how the apparent energy is processed from the six analog inputs.
0x08 BVAHR R 16 S 0 VA-Hour Accumulation Register for Phase B.
0x09 CVAHR R 16 S 0 VA-Hour Accumulation Register for Phase C.
0x0A AIRMS R 24 S 0 Phase A Current Channel RMS Register. The register contains the rms
component of the Phase A input of the current channel. The source is
selected by data bits in the mode register.
0x0B BIRMS R 24 S 0 Phase B Current Channel RMS Register.
0x0C CIRMS R 24 S 0 Phase C Current Channel RMS Register.
0x0D AVRMS R 24 S 0 Phase A Voltage Channel RMS Register.
ADE7758
Rev. D | Page 61 of 72
Address
[A6:A0] Name R/W1Length Type2
Default
Value Description
0x0E BVRMS R 24 S 0 Phase B Voltage Channel RMS Register.
0x0F CVRMS R 24 S 0 Phase C Voltage Channel RMS Register.
0x10 FREQ R 12 U 0 Frequency of the Line Input Estimated by the Zero-Crossing Processing.
It can also display the period of the line input. Bit 7 of the LCYCMODE
register determines if the reading is frequency or period. Default is
frequency. Data Bit 0 and Bit 1 of the MMODE register determine the
voltage channel used for the frequency or period calculation.
0x11 TEMP R 8 S 0 Temperature Register. This register contains the result of the latest
temperature conversion. Refer to the Temperature Measurement
section for details on how to interpret the content of this register.
0x12 WFORM R 24 S 0 Waveform Register. This register contains the digitized waveform of one
of the six analog inputs or the digitized power waveform. The source is
selected by Data Bit 0 to Bit 4 in the WAVMODE register.
0x13 OPMODE R/W 8 U 4 Operational Mode Register. This register defines the general
configuration of the ADE7758 (see Table 18).
0x14 MMODE R/W 8 U 0xFC
Measurement Mode Register. This register defines the channel used for
period and peak detection measurements (see Table 19).
0x15 WAVMODE R/W 8 U 0 Waveform Mode Register. This register defines the channel and sampling
frequency used in the waveform sampling mode (see Table 20).
0x16 COMPMODE R/W 8 U 0x1C
Computation Mode Register. This register configures the formula
applied for the energy and line active energy measurements (see Table 22).
0x17 LCYCMODE R/W 8 U 0x78
Line Cycle Mode Register. This register configures the line cycle
accumulation mode for WATT-HR, VAR-HR, and VA-Hr (see Table 23).
0x18 MASK R/W 24 U 0 IRQ Mask Register. It determines if an interrupt event generates an
active-low output at the IRQ pin (see the section). Interrupts
0x19 STATUS R 24 U 0 IRQ Status Register. This register contains information regarding the
source of the ADE7758 interrupts (see the section). Interrupts
0x1A RSTATUS R 24 U 0 IRQ Reset Status Register. Same as the STATUS register, except that its
contents are reset to 0 (all flags cleared) after a read operation.
0x1B ZXTOUT R/W 16 U 0xFFFF
Zero-Cross Timeout Register. If no zero crossing is detected within the
time period specified by this register, the interrupt request line (IRQ)
goes active low for the corresponding line voltage. The maximum
timeout period is 2.3 seconds (see the section). Zero-Crossing Detection
0x1C LINECYC R/W 16 U 0xFFFF
Line Cycle Register. The content of this register sets the number of
half-line cycles that the active, reactive, and apparent energies are
accumulated for in the line accumulation mode.
0x1D SAGCYC R/W 8 U 0xFF SAG Line Cycle Register. This register specifies the number of consecutive
half-line cycles where voltage channel input may fall below a threshold
level. This register is common to the three line voltage SAG detection.
The detection threshold is specified by the SAGLVL register (see the Line
Voltage SAG Detection section).
0x1E SAGLVL R/W 8 U 0 SAG Voltage Level. This register specifies the detection threshold for the
SAG event. This register is common to all three phases’ line voltage SAG
detections. See the description of the SAGCYC register for details.
0x1F VPINTLVL R/W 8 U 0xFF
Voltage Peak Level Interrupt Threshold Register. This register sets the
level of the voltage peak detection. Bit 5 to Bit 7 of the MMODE register
determine which phases are to be monitored. If the selected voltage
phase exceeds this level, the PKV flag in the IRQ status register is set.
0x20 IPINTLVL R/W 8 U 0xFF
Current Peak Level Interrupt Threshold Register. This register sets the
level of the current peak detection. Bit 5 to Bit 7 of the MMODE register
determine which phases are to be monitored. If the selected current
phase exceeds this level, the PKI flag in the IRQ status register is set.
0x21 VPEAK R 8 U 0 Voltage Peak Register. This register contains the value of the peak
voltage waveform that has occurred within a fixed number of half-line
cycles. The number of half-line cycles is set by the LINECYC register.
ADE7758
Rev. D | Page 62 of 72
Address
[A6:A0] Name R/W1Length Type2
Default
Value Description
0x22 IPEAK R 8 U 0 Current Peak Register. This register holds the value of the peak current
waveform that has occurred within a fixed number of half-line cycles.
The number of half-line cycles is set by the LINECYC register.
0x23 GAIN R/W 8 U 0 PGA Gain Register. This register is used to adjust the gain selection for
the PGA in the current and voltage channels (see the Analog Inputs section).
0x24 AVRMSGAIN R/W 12 S 0 Phase A VRMS Gain Register. The range of the voltage rms calculation
can be adjusted by writing to this register. It has an adjustment range of
±50% with a resolution of 0.0244%/LSB.
0x25 BVRMSGAIN R/W 12 S 0 Phase B VRMS Gain Register.
0x26 CVRMSGAIN R/W 12 S 0 Phase C VRMS Gain Register.
0x27 AIGAIN R/W 12 S 0 Phase A Current Gain Register. The range of the current rms calculation
can be adjusted by writing to this register. It has an adjustment range of
±50% with a resolution of 0.0244%/LSB. Adjusting this register also
scales the watt and VAR calculation. Not for use with Mode 0 of CONSEL,
COMPMODE[1:0].
0x28 BIGAIN R/W 12 S 0 Phase B Current Gain Register. Not for use with Mode 0 of CONSEL
COMPMODE[1:0].
0x29 CIGAIN R/W 12 S 0 Phase C Current Gain Register. Not for use with Mode 0 of CONSEL
COMPMODE[1:0].
0x2A AWG R/W 12 S 0 Phase A Watt Gain Register. The range of the watt calculation can be
adjusted by writing to this register. It has an adjustment range of ±50%
with a resolution of 0.0244%/LSB.
0x2B BWG R/W 12 S 0 Phase B Watt Gain Register.
0x2C CWG R/W 12 S 0 Phase C Watt Gain Register.
0x2D AVARG R/W 12 S 0 Phase A VAR Gain Register. The range of the VAR calculation can be
adjusted by writing to this register. It has an adjustment range of ±50%
with a resolution of 0.0244%/LSB.
0x2E BVARG R/W 12 S 0 Phase B VAR Gain Register.
0x2F CVARG R/W 12 S 0 Phase C VAR Gain Register.
0x30 AVAG R/W 12 S 0 Phase A VA Gain Register. The range of the VA calculation can be
adjusted by writing to this register. It has an adjustment range of ±50%
with a resolution of 0.0244%/LSB.
0x31 BVAG R/W 12 S 0 Phase B VA Gain Register.
0x32 CVAG R/W 12 S 0 Phase C VA Gain Register.
0x33 AVRMSOS R/W 12 S 0 Phase A Voltage RMS Offset Correction Register.
0x34 BVRMSOS R/W 12 S 0 Phase B Voltage RMS Offset Correction Register.
0x35 CVRMSOS R/W 12 S 0 Phase C Voltage RMS Offset Correction Register.
0x36 AIRMSOS R/W 12 S 0 Phase A Current RMS Offset Correction Register.
0x37 BIRMSOS R/W 12 S 0 Phase B Current RMS Offset Correction Register.
0x38 CIRMSOS R/W 12 S 0 Phase C Current RMS Offset Correction Register.
0x39 AWATTOS R/W 12 S 0 Phase A Watt Offset Calibration Register.
0x3A BWATTOS R/W 12 S 0 Phase B Watt Offset Calibration Register.
0x3B CWATTOS R/W 12 S 0 Phase C Watt Offset Calibration Register.
0x3C AVAROS R/W 12 S 0 Phase A VAR Offset Calibration Register.
0x3D BVAROS R/W 12 S 0 Phase B VAR Offset Calibration Register.
0x3E CVAROS R/W 12 S 0 Phase C VAR Offset Calibration Register.
0x3F APHCAL R/W 7 S 0 Phase A Phase Calibration Register. The phase relationship between
the current and voltage channel can be adjusted by writing to this
signed 7-bit register (see the Phase Compensation section).
0x40 BPHCAL R/W 7 S 0 Phase B Phase Calibration Register.
0x41 CPHCAL R/W 7 S 0 Phase C Phase Calibration Register.
0x42 WDIV R/W 8 U 0 Active Energy Register Divider.
0x43 VARDIV R/W 8 U 0 Reactive Energy Register Divider.
0x44 VADIV R/W 8 U 0 Apparent Energy Register Divider.
ADE7758
Rev. D | Page 63 of 72
Address
[A6:A0] Name R/W1Length Type2
Default
Value Description
0x45 APCFNUM R/W 16 U 0 Active Power CF Scaling Numerator Register. The content of this
register is used in the numerator of the APCF output scaling calculation.
Bits [15:13] indicate reverse polarity active power measurement for
Phase A, Phase B, and Phase C in order; that is, Bit 15 is Phase A, Bit 14
is Phase B, and so on.
0x46 APCFDEN R/W 12 U 0x3F
Active Power CF Scaling Denominator Register. The content of this
register is used in the denominator of the APCF output scaling.
0x47 VARCFNUM R/W 16 U 0 Reactive Power CF Scaling Numerator Register. The content of
this register is used in the numerator of the VARCF output scaling.
Bits [15:13] indicate reverse polarity reactive power measurement for
Phase A, Phase B, and Phase C in order; that is, Bit 15 is Phase A, Bit 14
is Phase B, and so on.
0x48 VARCFDEN R/W 12 U 0x3F
Reactive Power CF Scaling Denominator Register. The content of this
register is used in the denominator of the VARCF output scaling.
0x49 to
0x7D
RESERVED − − – − Reserved.
0x7E CHKSUM R 8 U − Checksum Register. The content of this register represents the sum of all
the ones in the last register read from the SPI port.
0x7F VERSION R 8 U − Version of the Die.
1 This column specifies the read/write capability of the register. R = Read only register. R/W = Register that can be both read and written.
2 Type decoder: U = unsigned; S = signed.
ADE7758
Rev. D | Page 64 of 72
OPERATIONAL MODE REGISTER (0x13)
The general configuration of the ADE7758 is defined by writing to the OPMODE register. Table 18 summarizes the functionality of each
bit in the OPMODE register.
Table 18. OPMODE Register
Bit
Location
Bit
Mnemonic
Default
Value Description
0 DISHPF 0 The HPFs in all current channel inputs are disabled when this bit is set.
1 DISLPF 0 The LPFs after the watt and VAR multipliers are disabled when this bit is set.
2 DISCF 1 The frequency outputs APCF and VARCF are disabled when this bit is set.
3 to 5 DISMOD 0 By setting these bits, ADE7758’s ADCs can be turned off. In normal operation, these bits should
be left at Logic 0.
DISMOD[2:0] Description
0 0 0 Normal operation.
1 0 0
Redirect the voltage inputs to the signal paths for the current channels and
the current inputs to the signal paths for the voltage channels.
0 0 1 Switch off only the current channel ADCs.
1 0 1
Switch off current channel ADCs and redirect the current input signals to the
voltage channel signal paths.
0 1 0 Switch off only the voltage channel ADCs.
1 1 0
Switch off voltage channel ADCs and redirect the voltage input signals to the
current channel signal paths.
0 1 1 Put the ADE7758 in sleep mode.
1 1 1 Put the ADE7758 in power-down mode (reduces AIDD to 1 mA typ).
6 SWRST 0 Software Chip Reset. A data transfer to the ADE7758 should not take place for at least 18 μs after
a software reset.
7 RESERVED 0 This should be left at 0.
MEASUREMENT MODE REGISTER (0x14)
The configuration of the PERIOD and peak measurements made by the ADE7758 is defined by writing to the MMODE register. Table 19
summarizes the functionality of each bit in the MMODE register.
Table 19. MMODE Register
Bit
Location
Bit
Mnemonic
Default
Value Description
0 to 1 FREQSEL 0 These bits are used to select the source of the measurement of the voltage line frequency.
FREQSEL1 FREQSEL0 Source
0 0 Phase A
0 1 Phase B
1 0 Phase C
1 1 Reserved
2 to 4 PEAKSEL 7 These bits select the phases used for the voltage and current peak registers. Setting Bit 2 switches
the IPEAK and VPEAK registers to hold the absolute values of the largest current and voltage
waveform (over a fixed number of half-line cycles) from Phase A. The number of half-line cycles is
determined by the content of the LINECYC register. At the end of the LINECYC number of half-line
cycles, the content of the registers is replaced with the new peak values. Similarly, setting Bit 3 turns
on the peak detection for Phase B, and Bit 4 for Phase C. Note that if more than one bit is set, the
VPEAK and IPEAK registers can hold values from two different phases, that is, the voltage and
current peak are independently processed (see the Peak Current Detection section).
5 to 7 PKIRQSEL 7 These bits select the phases used for the peak interrupt detection. Setting Bit 5 switches on the
monitoring of the absolute current and voltage waveform to Phase A. Similarly, setting Bit 6 turns on
the waveform detection for Phase B, and Bit 7 for Phase C. Note that more than one bit can be set for
detection on multiple phases. If the absolute values of the voltage or current waveform samples in
the selected phases exceeds the preset level specified in the VPINTLVL or IPINTLVL registers the
corresponding bit(s) in the STATUS registers are set (see the Peak Current Detection section).
ADE7758
Rev. D | Page 65 of 72
WAVEFORM MODE REGISTER (0x15)
The waveform sampling mode of the ADE7758 is defined by writing to the WAVMODE register. Table 20 summarizes the functionality of
each bit in the WAVMODE register.
Table 20. WAVMODE Register
Bit
Location
Bit
Mnemonic
Default
Value Description
0 to 1 PHSEL 0 These bits are used to select the phase of the waveform sample.
PHSEL[1:0] Source
0 0 Phase A
0 1 Phase B
1 0 Phase C
1 1 Reserved
2 to 4 WAVSEL 0 These bits are used to select the type of waveform.
WAVSEL[2:0] Source
0 0 0 Current
0 0 1 Voltage
0 1 0 Active Power Multiplier Output
0 1 1 Reactive Power Multiplier Output
1 0 0 VA Multiplier Output
-Others- Reserved
5 to 6 DTRT 0 These bits are used to select the data rate.
DTRT[1:0] Update Rate
0 0 26.04 kSPS (CLKIN/3/128)
0 1 13.02 kSPS (CLKIN/3/256)
1 0 6.51 kSPS (CLKIN/3/512)
1 1 3.25 kSPS (CLKIN/3/1024)
7 VACF 0 Setting this bit to Logic 1 switches the VARCF output pin to an output frequency that is
proportional to the total apparent power (VA). In the default state, Logic 0, the VARCF pin outputs
a frequency proportional to the total reactive power (VAR).
ADE7758
Rev. D | Page 66 of 72
COMPUTATIONAL MODE REGISTER (0x16)
The computational method of the ADE7758 is defined by writing to the COMPMODE register. Table 21 summarizes the functionality of
each bit in the COMPMODE register.
Table 21. COMPMODE Register
Bit
Location
Bit
Mnemonic
Default
Value Description
0 to 1 CONSEL 0 These bits are used to select the input to the energy accumulation registers. CONSEL[1:0] = 11 is
reserved. IA, IB, and IC are IA, IB, and IC phase shifted by –90°, respectively.
Registers CONSEL[1, 0] = 00 CONSEL[1, 0] = 01 CONSEL[1, 0] = 10
AWATTHR VA × IA VA × (IA – IB) VA × (IA–IB)
BWATTHR VB × IB 0 0
CWATTHR VC × IC VC × (IC – IB) VC × IC
AVARHR
VA × IA VA × (IA – IB) VA × (IA–IB)
BVARHR
VB × IB 0 0
CVARHR
VC × IC VC × (IC – IB) VC × IC
AVAHR VARMS × IARMS VARMS × IARMS VARMS × ARMS
BVAHR VBRMS × IBRMS (VARMS + VCRMS)/2 × IBRMS VARMS × IBRMS
CVAHR VCRMS × ICRMS VCRMS × ICRMS VCRMS × ICRMS
2 to 4 TERMSEL 7 These bits are used to select the phases to be included in the APCF and VARCF pulse outputs. Setting
Bit 2 selects Phase A (the inputs to AWATTHR and AVARHR registers) to be included. Bit 3 and Bit 4
are for Phase B and Phase C, respectively. Setting all three bits enables the sum of all three phases to
be included in the frequency outputs (see the Active Power Frequency Output and the Reactive
Power Frequency Output sections).
5 ABS 0 Setting this bit places the APCF output pin in absolute only mode. Namely, the APCF output
frequency is proportional to the sum of the absolute values of the watt-hour accumulation registers
(AWATTHR, BWATTHR, and CWATTHR). Note that this bit only affects the APCF pin and has no effect
on the content of the corresponding registers.
6 SAVAR 0 Setting this bit places the VARCF output pin in the signed adjusted mode. Namely, the VARCF output
frequency is proportional to the sign-adjusted sum of the VAR-hour accumulation registers (AVARHR,
BVARHR, and CVARHR). The sign of the VAR is determined from the sign of the watt calculation from
the corresponding phase, that is, the sign of the VAR is flipped if the sign of the watt is negative, and
if the watt is positive, there is no change to the sign of the VAR. Note that this bit only affects the
VARCF pin and has no effect on the content of the corresponding registers.
7 NOLOAD 0 Setting this bit activates the no-load threshold in the ADE7758.
ADE7758
Rev. D | Page 67 of 72
LINE CYCLE ACCUMULATION MODE REGISTER (0x17)
The functionalities involved the line-cycle accumulation mode in the ADE7758 are defined by writing to the LCYCMODE register.
Table 22 summarizes the functionality of each bit in the LCYCMODE register.
Table 22. LCYCMODE Register
Bit
Location
Bit
Mnemonic
Default
Value Description
0 LWATT 0 Setting this bit places the watt-hour accumulation registers (AWATTHR, BWATTHR, and CWATTHR
registers) into line-cycle accumulation mode.
1 LVAR 0 Setting this bit places the VAR-hour accumulation registers (AVARHR, BVARHR, and CVARHR registers)
into line-cycle accumulation mode.
2 LVA 0 Setting this bit places the VA-hour accumulation registers (AVAHR, BVAHR, and CVAHR registers) into
line-cycle accumulation mode.
3 to 5 ZXSEL 7 These bits select the phases used for counting the number of zero crossings in the line-cycle
accumulation mode. Bit 3, Bit 4, and Bit 5 select Phase A, Phase B, and Phase C, respectively. More than
one phase can be selected for the zero-crossing detection, and the accumulation time is shortened
accordingly.
6 RSTREAD 1 Setting this bit enables the read-with-reset for all the WATTHR, VARHR, and VAHR registers for all three
phases, that is, a read to those registers resets the registers to 0 after the content of the registers have
been read. This bit should be set to Logic 0 when the LWATT, LVAR, or LVA bits are set to Logic 1.
7 FREQSEL 0 Setting this bit causes the FREQ (0x10) register to display the period, instead of the frequency of the
line input.
ADE7758
Rev. D | Page 68 of 72
INTERRUPT MASK REGISTER (0x18)
When an interrupt event occurs in the ADE7758, the IRQ logic output goes active low if the mask bit for this event is Logic 1 in the
MASK register. The IRQ logic output is reset to its default collector open state when the RSTATUS register is read. describes the
function of each bit in the interrupt mask register.
Table 23
Table 23. Function of Each Bit in the Interrupt Mask Register
Bit
Location
Interrupt
Flag
Default
Value Description
0 AEHF 0 Enables an interrupt when there is a change in Bit 14 of any one of the three WATTHR registers,
that is, the WATTHR register is half full.
1 REHF 0 Enables an interrupt when there is a change in Bit 14 of any one of the three VARHR registers,
that is, the VARHR register is half full.
2 VAEHF 0 Enables an interrupt when there is a 0 to 1 transition in the MSB of any one of the three VAHR
registers, that is, the VAHR register is half full.
3 SAGA 0 Enables an interrupt when there is a SAG on the line voltage of the Phase A.
4 SAGB 0 Enables an interrupt when there is a SAG on the line voltage of the Phase B.
5 SAGC 0 Enables an interrupt when there is a SAG on the line voltage of the Phase C.
6 ZXTOA 0 Enables an interrupt when there is a zero-crossing timeout detection on Phase A.
7 ZXTOB 0 Enables an interrupt when there is a zero-crossing timeout detection on Phase B.
8 ZXTOC 0 Enables an interrupt when there is a zero-crossing timeout detection on Phase C.
9 ZXA 0 Enables an interrupt when there is a zero crossing in the voltage channel of Phase A (see the
Zero-Crossing Detection section).
10 ZXB 0 Enables an interrupt when there is a zero crossing in the voltage channel of Phase B (see the
Zero-Crossing Detection section).
11 ZXC 0 Enables an interrupt when there is a zero crossing in the voltage channel of Phase C (see the
Zero-Crossing Detection section).
12 LENERGY 0 Enables an interrupt when the energy accumulations over LINECYC are finished.
13 RESERVED 0 Reserved.
14 PKV 0 Enables an interrupt when the voltage input selected in the MMODE register is above the value
in the VPINTLVL register.
15 PKI 0 Enables an interrupt when the current input selected in the MMODE register is above the value
in the IPINTLVL register.
16 WFSM 0 Enables an interrupt when data is present in the WAVEMODE register.
17 REVPAP 0 Enables an interrupt when there is a sign change in the watt calculation among any one of the
phases specified by the TERMSEL bits in the COMPMODE register.
18 REVPRP 0 Enables an interrupt when there is a sign change in the VAR calculation among any one of the
phases specified by the TERMSEL bits in the COMPMODE register.
19 SEQERR 0 Enables an interrupt when the zero crossing from Phase A is followed not by the zero crossing
of Phase C but with that of Phase B.
ADE7758
Rev. D | Page 69 of 72
INTERRUPT STATUS REGISTER (0x19)/RESET INTERRUPT STATUS REGISTER (0x1A)
The interrupt status register is used to determine the source of an interrupt event. When an interrupt event occurs in the ADE7758, the
corresponding flag in the interrupt status register is set. The IRQ pin goes active low if the corresponding bit in the interrupt mask
register is set. When the MCU services the interrupt, it must first carry out a read from the interrupt status register to determine the
source of the interrupt. All the interrupts in the interrupt status register stay at their logic high state after an event occurs. The state of the
interrupt bit in the interrupt status register is reset to its default value once the reset interrupt status register is read.
Table 24. Interrupt Status Register
Bit
Location
Interrupt
Flag
Default
Value Event Description
0 AEHF 0 Indicates that an interrupt was caused by a change in Bit 14 among any one of the three
WATTHR registers, that is, the WATTHR register is half full.
1 REHF 0 Indicates that an interrupt was caused by a change in Bit 14 among any one of the three
VARHR registers, that is, the VARHR register is half full.
2 VAEHF 0 Indicates that an interrupt was caused by a 0 to 1 transition in Bit 15 among any one of
the three VAHR registers, that is, the VAHR register is half full.
3 SAGA 0 Indicates that an interrupt was caused by a SAG on the line voltage of the Phase A.
4 SAGB 0 Indicates that an interrupt was caused by a SAG on the line voltage of the Phase B.
5 SAGC 0 Indicates that an interrupt was caused by a SAG on the line voltage of the Phase C.
6 ZXTOA 0 Indicates that an interrupt was caused by a missing zero crossing on the line voltage of the Phase A.
7 ZXTOB 0 Indicates that an interrupt was caused by a missing zero crossing on the line voltage of the Phase B.
8 ZXTOC 0 Indicates that an interrupt was caused by a missing zero crossing on the line voltage of the Phase C.
9 ZXA 0 Indicates a detection of a rising edge zero crossing in the voltage channel of Phase A.
10 ZXB 0 Indicates a detection of a rising edge zero crossing in the voltage channel of Phase B.
11 ZXC 0 Indicates a detection of a rising edge zero crossing in the voltage channel of Phase C.
12 LENERGY 0 In line energy accumulation, indicates the end of an integration over an integer number of half-
line cycles (LINECYC). See the Calibration section.
13 RESET 1 Indicates that the 5 V power supply is below 4 V. Enables a software reset of the ADE7758 and sets
the registers back to their default values. This bit in the STATUS or RSTATUS register is logic high
for only one clock cycle after a reset event.
14 PKV 0 Indicates that an interrupt was caused when the selected voltage input is above the value
in the VPINTLVL register.
15 PKI 0 Indicates that an interrupt was caused when the selected current input is above the value
in the IPINTLVL register.
16 WFSM 0 Indicates that new data is present in the waveform register.
17 REVPAP 0 Indicates that an interrupt was caused by a sign change in the watt calculation among any one
of the phases specified by the TERMSEL bits in the COMPMODE register.
18 REVPRP 0 Indicates that an interrupt was caused by a sign change in the VAR calculation among any one
of the phases specified by the TERMSEL bits in the COMPMODE register.
19 SEQERR 0 Indicates that an interrupt was caused by a zero crossing from Phase A followed not by the zero
crossing of Phase C but by that of Phase B.
ADE7758
Rev. D | Page 70 of 72
OUTLINE DIMENSIONS
COMP LIANT TO JEDEC STANDARDS MS -013-AD
CONT ROLLING DIM E NS IONS ARE IN MI L LI M E TERS; INCH DIM E NSIO NS
(IN PARENTHESES) ARE ROUNDED-OFF M IL L IMETER E QUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
15.60 (0.6142 )
15.20 (0.5984 )
0.30 ( 0 .0 118)
0.10 ( 0 .0039)
2.65 (0.1043)
2.35 (0.0925)
10.65 (0.4193)
10.00 (0.3937)
7.60 ( 0.2992)
7.40 ( 0.2913)
0.75 (0.0295)
0.25 (0.0098)
45°
1.27 ( 0.0500)
0.40 ( 0.0157)
COPLANARITY
0.10 0.33 (0 .0130)
0.20 ( 0 .0079)
0.51 ( 0.0201)
0.31 ( 0.0122)
SEATING
PLANE
8°
0°
24 13
12
1
1.27 ( 0.0500)
BSC
060706-A
Figure 94. 24-Lead Standard Small Outline Package [SOIC_W]
Wide Body (RW-24)
Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model Temperature Range Package Description Package Option
ADE7758ARW −40°C to + 85°C 24-Lead Wide Body SOIC_W RW-24
ADE7758ARWRL −40°C to + 85°C 24-Lead Wide Body SOIC_W RW-24
ADE7758ARWZ1−40°C to + 85°C 24-Lead Wide Body SOIC_W RW-24
ADE7758ARWZRL1
−40°C to + 85°C 24-Lead Wide Body SOIC_W RW-24
EVAL-ADE7758EBZ1
Evaluation Board
1 Z = RoHS Compliant Part.
ADE7758
Rev. D | Page 71 of 72
NOTES
ADE7758
Rev. D | Page 72 of 72
NOTES
©2004–2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04443-0-10/08(D)
