DC237A - Linear Technology

LT1506

4.5A, 500kHz Step-Down

Switching Regulator

■

Constant 500kHz Switching Frequency

■

Easily Synchronizable

■

Operates with Input as Low as 4V

■

Uses All Surface Mount Components

■

Inductor Size Reduced to 1.8

■

Saturating Switch Design: 0.07Ω

■

Shutdown Current: 20µA

■

Cycle-by-Cycle Current Limiting

The LT

1506 is a 500kHz monolithic buck mode switching

regulator functionally identical to the LT1374 but optimized

for lower input voltage applications. It will operate over a

4V to 15V input range compared with 5.5V to 25V for the

LT1374. A 4.5A switch is included on the die along with all

the necessary oscillator, control and logic circuitry. High

switching frequency allows a considerable reduction in the

size of external components. The topology is current mode

for fast transient response and good loop stability. Both

fixed output voltage and adjustable parts are available.

A special high speed bipolar process and new design tech-

niques achieve high efficiency at high switching frequency.

Efficiency is maintained over a wide output current range

by keeping quiescent supply current to 4mA

and by utiliz-

ing a supply boost

capacitor to saturate the power switch.

The LT1506 fits into standard 7-pin DD and fused lead

SO-8 packages. Full cycle-by-cycle short-circuit protection

and thermal shutdown are provided. Standard surface

mount external parts are used, including the inductor and

capacitors. There is the optional function of shutdown or

synchronization. A shutdown signal reduces supply current

to 20µA. Synchronization allows an external logic level sig-

nal to increase the internal oscillator from 580kHz to 1MHz.

■

Portable Computers

■

Battery-Powered Systems

■

Battery Charger

■

Distributed Power

, LTC and LT are registered trademarks of Linear Technology Corporation.

LOAD CURRENT (A)

EFFICIENCY (%)

70 2.0 2.5 3.0 3.5

1506 TA02

0.5 1.0 1.5 4.0

OUT

= 3.3V

= 5V

L = 10µH

Efficiency vs Load Current

5V to 3.3V Down Converter

BOOST

LT1506-3.3

OUTPUT

3.3V

INPUT

1506 TA01

0.68µF

1.5nF D1

MBRS330T3

100µF, 10V

SOLID

TANTALUM

10µF TO

50µF

CERAMIC

1N914

5µH

SENSESHDN

OPEN

HIGH

= ON GND V

FEATURES

DESCRIPTIO

APPLICATIO S

TYPICAL APPLICATIO

LT1506

ABSOLUTE MAXIMUM RATINGS

Input Voltage .......................................................... 16V

BOOST Pin Above Input Voltage ............................. 15V

SHDN Pin Voltage..................................................... 7V

FB Pin Voltage (Adjustable Part)............................ 3.5V

FB Pin Current (Adjustable Part)............................ 1mA

Sense Voltage (Fixed 3.3V Part) ............................... 5V

SYNC Pin Voltage ..................................................... 7V

Operating Junction Temperature Range

LT1506C............................................... 0°C to 125° C

LT1506I ........................................... –40°C to 125°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

ELECTRICAL CHARACTERISTICS

PARAMETER CONDITIONS MIN TYP MAX UNITS

Feedback Voltage (Adjustable) 2.39 2.42 2.45 V

All Conditions ●2.36 2.48 V

Sense Voltage (Fixed 3.3V) 3.25 3.3 3.35 V

All Conditions ●3.23 3.37 V

SENSE Pin Resistance 4 6.6 9.5 kΩ

Reference Voltage Line Regulation 4.3V ≤ V

≤ 15V 0.01 0.03 %/V

Feedback Input Bias Current ●0.5 2 µA

Error Amplifier Voltage Gain (Notes 2, 8) 200 400

Error Amplifier Transconductance ∆I (V

) = ±10µA (Note 8) 1500 2000 2700 µMho

●1000 3100 µMho

Pin to Switch Current Transconductance 5.3 A/V

Error Amplifier Source Current V

= 2.1V or V

SENSE

= 2.9V ●140 225 320 µA

Error Amplifier Sink Current V

= 2.7V or V

SENSE

= 3.7V ●140 225 320 µA

Pin Switching Threshold Duty Cycle = 0 0.9 V

Pin High Clamp 2.1 V

Switch Current Limit V

Open, V

= 2.1V or V

SENSE

= 2.9V, DC ≤ 50% ●4.5 6 8.5 A

Slope Compensation DC = 80% 0.8 A

TJ = 25°C, VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.

PACKAGE/ORDER INFORMATION

ORDER PART

NUMBER

ORDER PART

NUMBER

LT1506CR

LT1506CR-3.3

LT1506CR-SYNC

LT1506CR-3.3 SYNC

LT1506IR

LT1506IR-3.3

LT1506IR-SYNC

LT1506IR-3.3 SYNC

JMAX

= 125°C, θ

= 30°C/W

WITH PACKAGE SOLDERED TO 0.5 SQUARE INCH

COPPER AREA OVER BACKSIDE GROUND PLANE OR

INTERNAL POWER PLANE. θ

CAN VARY FROM 20°C/W

TO >40°C/W DEPENDING ON MOUNTING TECHNIQUES 1506I

506I33

1506

150633

S8 PART MARKING

LT1506CS8

LT1506CS8-3.3

LT1506IS8

LT1506IS8-3.3

*Default is the adjustable output voltage device with FB pin and shutdown function. Option -3.3 replaces FB with SENSE pin for fixed 3.3V output

applications. -SYNC replaces SHDN with SYNC pin for applications requiring synchronization. Consult factory for Military grade parts.

FB OR SENSE*

BOOST

GND

SYNC OR SHDN*

R PACKAGE

7-LEAD PLASTIC DD PAK

FRONT VIEW

TAB

GND

TOP VIEW

S8 PACKAGE

8-LEAD PLASTIC SO

BOOST

GND**

SYNC

SHDN

FB OR

SENSE*

= 80°C/W

**WITH FUSED (GND) GROUND PIN

CONNECTED TO GROUND PLANE OR

LARGE LANDS

(Note 1)

LT1506

ELECTRICAL CHARACTERISTICS

PARAMETER CONDITIONS MIN TYP MAX UNITS

Switch On Resistance (Note 7) I

= 4.5A 0.07 0.1 Ω

●0.13 Ω

Maximum Switch Duty Cycle V

= 2.1V or V

SENSE

= 2.9V 90 93 %

●86 93 %

Switch Frequency V

Set to Give 50% Duty Cycle 460 500 540 kHz

●440 560 kHz

Switch Frequency Line Regulation 4.3V

≤ V

≤ 15V ●0 0.15 %/V

Frequency Shifting Threshold on FB Pin ∆f = 10kHz ●0.8 1.0 1.3 V

Minimum Input Voltage (Note 3) ●4.0 4.3 V

Minimum Boost Voltage (Note 4) I

≤ 4.5A ●2.3 3.0 V

Boost Current (Note 5) I

= 1A ●20 35 mA

= 4.5A ●90 140 mA

Input Supply Current (Note 6) ●3.8 5.4 mA

Shutdown Supply Current V

SHDN

= 0V, V

Open 15 50 µA

●75 µA

Lockout Threshold V

Open ●2.3 2.38 2.46 V

Shutdown Thresholds V

Open Device Shutting Down ●0.13 0.37 0.60 V

Device Starting Up ●0.25 0.45 0.7 V

Synchronization Threshold ●1.5 2.2 V

Synchronizing Range 580 1000 kHz

SYNC Pin Input Resistance 40 kΩ

TJ = 25°C, VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.

The ● denotes specifications which apply over the full operating

temperature range.

Note 1: Absolute Maximum Ratings are those values beyond which the life

of a device may be impaired.

Note 2: Gain is measured with a V

swing equal to 200mV above the

switching threshold level to 200mV below the upper clamp level.

Note 3: Minimum input voltage is not measured directly, but is guaranteed

by other tests. It is defined as the voltage where internal bias lines are still

regulated so that the reference voltage and oscillator frequency remain

constant. Actual minimum input voltage to maintain a regulated output will

depend on output voltage and load current. See Applications Information.

Note 4: This is the minimum voltage across the boost capacitor needed to

guarantee full saturation of the internal power switch.

Note 5: Boost current is the current flowing into the boost pin with the pin

held 5V above input voltage. It flows only during switch on time.

Note 6: Input supply current is the bias current drawn by the input pin

with switching disabled.

Note 7: Switch on resistance is calculated by dividing V

to V

voltage

by the forced current (4.5A). See Typical Performance Characteristics for

the graph of switch voltage at other currents.

Note 8: Transconductance and voltage gain refer to the internal amplifier

exclusive of the voltage divider. To calculate gain and transconductance

refer to SENSE pin on fixed voltage parts. Divide values shown by the ratio

OUT

/2.42.

LT1506

TYPICAL PERFORMANCE CHARACTERISTICS

Feedback Pin Voltage

TEMPERATURE (°C)

–50

2.430

2.425

2.420

2.415

2.410 100

1506 G03

–25 0 25 50 75 125

FEEDBACK VOLTAGE (V)

Switch Peak Current Limit

DUTY CYCLE (%)

SWITCH PEAK CURRENT (A)

6.5

6.0

5.5

5.0

4.5

4.0

3.5

3.0 80

1506 G02

20 40 60 100

TYPICAL

MINIMUM

Lockout and Shutdown

ThresholdsShutdown Pin Bias Current Shutdown Supply Current

JUNCTION TEMPERATURE (°C)

–50

2.40

2.36

2.32

0.8

0.4

025 75

1506 G05

–25 0 50 100 125

SHUTDOWN PIN VOLTAGE (V)

LOCKOUT

START-UP

SHUTDOWN

INPUT VOLTAGE (V)

INPUT SUPPLY CURRENT (µA)

051015

1506 G06

VSHDN = 0V

Shutdown Supply Current

SHUTDOWN VOLTAGE (V)

INPUT SUPPLY CURRENT (µA)

00.1 0.2 0.3 0.4

1506 G07

= 10V

FREQUENCY (Hz)

GAIN (µMho)

PHASE (DEG)

3000

2500

2000

1500

1000

500

200

150

100

–50

100 10k 100k 10M

1506 G09

1k 1M

GAIN

PHASE

ERROR AMPLIFIER EQUIVALENT CIRCUIT

OUT

200k

OUT

12pF

LOAD

= 50Ω

2 × 10

–3

)(

Error Amplifier Transconductance

JUNCTION TEMPERATURE (°C)

–50

TRANSCONDUCTANCE (µMho)

2500

2000

1500

1000

500

0050 75

1506 G08

–25 25 100 125

Minimum Input Voltage

with 3.3V Output

LOAD CURRENT (mA)

4.1

INPUT VOLTAGE (V)

4.3

4.5

4.7

10 100 1000

1506 G12

3.9

3.7

3.5

3.3

TEMPERATURE (°C)

–50

–500

–400

–300

–200

–8

–4

025 75

1506 G04

–25 0 50 100 125

CURRENT (µA)

AT 0.37V SHUTDOWN THRESHOLD.

AFTER SHUTDOWN, CURRENT

DROPS TO A FEW µA

AT 2.38V LOCKOUT THRESHOLD

LT1506

TYPICAL PERFORMANCE CHARACTERISTICS

Frequency Foldback

FEEDBACK PIN VOLTAGE (V)

SWITCHING FREQUENCY (kHz) OR CURRENT (µA)

500

400

300

200

100

02.0

1506 G10

0.5 1.0 1.5 2.5

SWITCHING

FREQUENCY

FEEDBACK PIN

CURRENT

Inductor Core Loss for 3.3V Output

INDUCTANCE (µH)

CORE LOSS (W)

1.0

0.1

0.01

0.001 46810

1506 G01

TYPE 52

Metglas

Kool Mµ

PERMALLOY

µ = 125

Switching Frequency

TEMPERATURE (°C)

–50

550

540

530

520

510

500

490

480

470

460

450 100

1506 G11

–25 0 25 50 75 125

FREQUENCY (kHz)

BOOST Pin Current

SWITCH CURRENT (A)

BOOST PIN CURRENT (mA)

12345

1506 G14

100

90 DUTY CYCLE = 100%

Maximum Load Current

at VOUT = 3.3V

INPUT VOLTAGE (V)

4.0

4.2

4.4

1506 G13

3.8

3.6

610814

3.4

3.2

3.0

LOAD CURRENT (A)

L= 10µH

L= 5µH

L= 3µH

L= 1.8µH

Maximum Load Current

at VOUT = 5V

INPUT VOLTAGE (V)

2.6

LOAD CURRENT (A)

2.8

3.2

3.4

3.6

913 15

4.4

1506 G17

3.0

711

3.8

4.0

4.2 L= 10µH

L= 5µH

L= 3µH

L= 1.8µH

Switch Voltage Drop

SWITCH CURRENT (A)

SWITCH VOLTAGE (mV)

250

200

400 125°C

25°C

350

300

1506 G16

150

100

0123

500

450

–40°C

VC Pin Shutdown Threshold

JUNCTION TEMPERATURE (°C)

–50

1.4

1.2

1.0

0.8

0.6

0.4 100

1506 G15

–25 0 25 50 75 125

THRESHOLD VOLTAGE (V)

SHUTDOWN

OUTPUT VOLTAGE (%)

1506 G18

20 40 60 100

OUTPUT CURRENT (A)

FOLDBACK

CHARACTERISTICS

CURRENT

SOURCE

LOAD

RESISTOR

LOAD

MOS LOAD

POSSIBLE UNDESIRED

STABLE POINT FOR

CURRENT SOURCE

LOAD*

Current Limit Foldback

Kool Mµ is a registered trademark of Magnetics, Inc.

Metglas is a registered trademark of AlliedSignal Inc.

*See “More Than Just Voltage Feedback” in the Applications Information section.

LT1506

PIN FUNCTIONS

UUU

FB/SENSE: The feedback pin is used to set output voltage

using an external voltage divider that generates 2.42V at

the pin with the desired output voltage. The fixed voltage

(-3.3) parts have the divider included on the chip and the

FB pin is used as a SENSE pin, connected directly to the

3.3V output. Three additional functions are performed by

the FB pin. When the pin voltage drops below 1.7V, switch

current limit is reduced. Below 1.5V the external sync

function is disabled. Below 1V, switching frequency is also

reduced. See Feedback Pin Function section in Applica-

tions Information for details.

BOOST: The BOOST pin is used to provide a drive voltage,

higher than the input voltage, to the internal bipolar NPN

power switch. Without this added voltage, the typical

switch voltage loss would be about 1.5V. The additional

boost voltage allows the switch to saturate and voltage

loss approximates that of a 0.07Ω FET structure, but with

much smaller die area. Efficiency improves from 75% for

conventional bipolar designs to > 89% for these new parts.

: This is the collector of the on-chip power NPN switch.

This pin powers the internal circuitry and internal regula-

tor. At NPN switch on and off, high dI/dt edges occur on

this pin. Keep the external bypass and catch diode close to

this pin. All trace inductance on this path will create a

voltage spike at switch off, adding to the V

voltage

across the internal NPN.

GND: The GND pin connection needs consideration for

two reasons. First, it acts as the reference for the regulated

output, so load regulation will suffer if the “ground” end of

the load is not at the same voltage as the GND pin of the

IC. This condition will occur when load current or other

currents flow through metal paths between the GND pin

and the load ground point. Keep the ground path short

between the GND pin and the load and use a ground plane

when possible. The second consideration is EMI caused

by GND pin current spikes. Internal capacitance between

the V

pin and the GND pin creates very narrow (<10ns)

current spikes in the GND pin. If the GND pin is connected

to system ground with a long metal trace, this trace may

radiate excess EMI. Keep the path between the input

bypass and the GND pin short. The GND pin of the SO-8

package is directly attached to the internal tab. This pin

should be attached to a large copper area to improve

thermal resistance.

: The switch pin is the emitter of the on-chip power

NPN switch. This pin is driven up to the input pin voltage

during switch on time. Inductor current drives the switch

pin negative during switch off time. Negative voltage is

clamped with the external catch diode. Maximum negative

switch voltage allowed is –0.8V.

SYNC: The sync pin is used to synchronize the internal

oscillator to an external signal. It is directly logic compat-

ible and can be driven with any signal between 10% and

90% duty cycle. The synchronizing range is equal to

initial

operating frequency, up to 1MHz. This pin replaces SHDN

on -SYNC option parts. See Synchronizing section in

Applications Information for details. When not in use, this

pin should be grounded.

SHDN: The shutdown pin is used to turn off the regulator

and to reduce input drain current to a few microamperes.

Actually, this pin has two separate thresholds, one at

2.38V to disable switching, and a second at 0.4V to force

complete micropower shutdown. The 2.38V threshold

functions as an accurate undervoltage lockout (UVLO).

This is sometimes used to prevent the regulator from

operating until the input votlage has reached a predeter-

mined level.

: The V

pin is the output of the error amplifier and the

input of the peak switch current comparator. It is normally

used for frequency compensation, but can do double duty

as a current clamp or control loop override. This pin sits

at about 1V for very light loads and 2V at maximum load.

It can be driven to ground to shut off the regulator, but if

driven high, current must be limited to 4mA.

LT1506

BLOCK DIAGRAM

and output capacitor, then an abrupt 180° shift will occur.

The current fed system will have 90° phase shift at a much

lower frequency, but will not have the additional 90° shift

until well beyond the LC resonant frequency. This makes

it much easier to frequency compensate the feedback loop

and also gives much quicker transient response.

High switch efficiency is attained by using the BOOST pin

to provide a voltage to the switch driver which is higher

than the input voltage, allowing switch to be saturated.

This boosted voltage is generated with an external capaci-

tor and diode. Two comparators are connected to the

shutdown pin. One has a 2.38V threshold for undervoltage

lockout and the second has a 0.4V threshold for complete

shutdown.

The LT1506 is a constant frequency, current mode buck

converter. This means that there is an internal clock and

two feedback loops that control the duty cycle of the power

switch. In addition to the normal error amplifier, there is a

current sense amplifier that monitors switch current on a

cycle-by-cycle basis. A switch cycle starts with an oscilla-

tor pulse which sets the R

flip-flop to turn the switch on.

When switch current reaches a level set by the inverting

input of the comparator, the flip-flop is reset and the

switch turns off. Output voltage control is obtained by

using the output of the error amplifier to set the switch

current trip point. This technique means that the error

amplifier commands current to be delivered to the output

rather than voltage. A voltage fed system will have low

phase shift up to the resonant frequency of the inductor

Figure 1. Block Diagram

–

INPUT

2.9V BIAS

REGULATOR

500kHz

OSCILLATOR

FREQUENCY

SHIFT CIRCUIT

GND

1506 BD

SLOPE COMP

0.01Ω

INTERNAL

CURRENT

SENSE

AMPLIFIER

VOLTAGE GAIN = 20

SYNC

SHDN

SHUTDOWN

COMPARATOR

LOCKOUT

COMPARATOR

CURRENT

COMPARATOR

ERROR

AMPLIFIER

= 2000µMho

FOLDBACK

CURRENT

LIMIT

CLAMP

BOOST

FLIP-FLOP DRIVER

CIRCUITRY

0.9V

POWER

SWITCH

PARASITIC DIODES

DO NOT FORWARD BIAS

2.42V

–

0.4V

3.5µA

2.38V

LT1506

APPLICATIONS INFORMATION

WUUU

FEEDBACK PIN FUNCTIONS

The feedback (FB) pin on the LT1506 is used to set output

voltage and provide several overload protection features.

The first part of this section deals with selecting resistors

to set output voltage and the remaining part talks about

foldback frequency and current limiting created by the FB

pin. Please read both parts before committing to a final

design. The fixed 3.3V LT1506-3.3 has internal divider

resistors and the FB pin is renamed SENSE, connected

directly to the output.

The suggested value for the output divider resistor (see

Figure 2) from FB to ground (R2) is 5k or less, and a

formula for R1 is shown below. The output voltage error

caused by ignoring the input bias current on the FB pin is

less than 0.25% with R2 = 5k. Please read the following

if divider resistors are increased above the suggested

values.

RRV

OUT

12242

242

=−

()

More Than Just Voltage Feedback

The feedback pin is used for more than just output voltage

sensing. It also reduces switching frequency and current

limit when output voltage is very low (see the Frequency

Foldback graph in Typical Performance Characteristics).

This is done to control power dissipation in both the IC and

in the external diode and inductor during short-circuit

conditions. A shorted output requires the switching regu-

lator to operate at very low duty cycles, and the average

current through the diode and inductor is equal to the

short-circuit current limit of the switch (typically 6A for the

LT1506, folding back to less than 3A). Minimum switch on

time limitations would prevent the switcher from attaining

a sufficiently low duty cycle if switching frequency were

maintained at 500kHz, so frequency is reduced by about

5:1 when the feedback pin voltage drops below 1V (see

Frequency Foldback graph). This does not affect operation

with normal load conditions; one simply sees a gear shift

in switching frequency during start-up as the output

voltage rises.

In addition to lower switching frequency, the LT1506 also

operates at lower switch current limit when the feedback

pin voltage drops below 1.7V. Q2 in Figure 2 performs this

function by clamping the V

pin to a voltage less than its

normal 2.1V upper clamp level. This

foldback current limit

greatly reduces power dissipation in the IC, diode and

inductor during short-circuit conditions. External synchro-

nization is also disabled to prevent interference with

foldback operation. Again, it is nearly transparent to the

user under normal load conditions. The only loads that may

be affected are current source loads which maintain full

load current with output voltage less than 50% of final value.

In these rare situations the feedback pin can be clamped

above 1.5V with an external diode to defeat foldback cur-

rent limit.

Caution:

clamping the feedback pin means that

frequency shifting will also be defeated, so a combination

of high input voltage and dead shorted output may cause

the LT1506 to lose control of current limit.

Figure 2. Frequency and Current Limit Foldback

–

2.4V

VSW

VCGND

TO SYNC CIRCUIT

1506 F02

TO FREQUENCY

SHIFTING

1k R4

OUTPUT

ERROR

AMPLIFIER

1.6V Q1

LT1506

APPLICATIONS INFORMATION

WUUU

The internal circuitry which forces reduced switching

frequency also causes current to flow out of the feedback

pin when output voltage is low. The equivalent circuitry is

shown in Figure 2. Q1 is completely off during normal

operation. If the FB pin falls below 1V, Q1 begins to

conduct current and reduces frequency at the rate of

approximately 5kHz/µA. To ensure adequate frequency

foldback (under worst-case short-circuit conditions), the

external divider Thevinin resistance must be low enough

to pull 150µA out of the FB pin with 0.6V on the pin (R

DIV

≤ 4k).

The net result is that reductions in frequency and

current limit are affected by output voltage divider imped-

ance. Although divider impedance is not critical, caution

should be used if resistors are increased beyond the

suggested values and short-circuit conditions will occur

with high input voltage

. High frequency pickup will

increase and the protection accorded by frequency and

current foldback will decrease.

MAXIMUM OUTPUT LOAD CURRENT

Maximum load current for a buck converter is limited by

the maximum switch current rating (I

) of the LT1506.

This current rating is 4.5A up to 50% duty cycle (DC),

decreasing to 3.7A at 80% duty cycle. This is shown

graphically in Typical Performance Characteristics and as

shown in the formula below:

= 4.5A for DC ≤ 50%

= 3.21 + 5.95(DC) – 6.75(DC)

for 50% < DC < 90%

DC = Duty cycle = V

OUT

Example: with V

OUT

= 5V, V

= 8V; DC = 5/8 = 0.625, and;

SW(MAX)

= 3.21 + 5.95(0.625) – 6.75(0.625)

= 4.3A

Current rating decreases with duty cycle because the

LT1506 has internal slope compensation to prevent cur-

rent mode subharmonic switching. For more details, read

Application Note 19. The LT1506 is a little unusual in this

regard because it has nonlinear slope compensation which

gives better compensation with less reduction in current

limit.

Maximum load current would be equal to maximum

switch current

for an infinitely large inductor

, but with

finite inductor size, maximum load current is reduced by

one-half peak-to-peak inductor current. The following

formula assumes continuous mode operation, implying

that the term on the right is less than one-half of I

OUT(MAX)

Continuous Mode

For the conditions above and L = 3.3µH,

OUT MAX

(

)

−

=−

()

−

()











()

=− =

43 58 5

2 3 3 10 500 10 8

43 057 373

..• •

.. .

At V

= 15V, duty cycle is 33%, so I

is just equal to a fixed

4.5A, and I

OUT(MAX)

is equal to:

45 515 5

2 3 3 10 500 10 15

45 101 349

..• •

.. .

−

()

−

()











()

=− =

−

Note that there is less load current available at the higher

input voltage because inductor ripple current increases.

This is not always the case. Certain combinations of

inductor value and input voltage range may yield lower

available load current at the lowest input voltage due to

reduced peak switch current at high duty cycles. If load

current is close to the maximum available, please check

maximum available current at both input voltage ex-

tremes. To calculate actual peak switch current with a

given set of conditions, use:

VVV

LfV

SW PEAK OUT OUT IN OUT

(

)

=+ −

()

()()( )

P−

()

−

()

()()( )

VVV

LfV

OUT IN OUT

LT1506

APPLICATIONS INFORMATION

WUUU

CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR

For most applications the output inductor will fall in the

range of 3µH to 20µH. Lower values are chosen to reduce

physical size of the inductor. Higher values allow more

output current because they reduce peak current seen by

the LT1506 switch, which has a 4.5A limit. Higher values

also reduce output ripple voltage, and reduce core loss.

Graphs in the Typical Performance Characteristics section

show maximum output load current versus inductor size

and input voltage. A second graph shows core loss versus

inductor size for various core materials.

When choosing an inductor you might have to consider

maximum load current, core and copper losses, allowable

component height, output voltage ripple, EMI, fault cur-

rent in the inductor, saturation, and of course, cost. The

following procedure is suggested as a way of handling

these somewhat complicated and conflicting requirements.

1. Choose a value in microhenries from the graphs of

maximum load current and core loss. Choosing a small

inductor with lighter loads may result in discontinuous

mode of operation, but the LT1506 is designed to work

well in either mode. Keep in mind that lower core loss

means higher cost, at least for closed core geometries

like toroids. The core loss graphs show absolute loss

for a 3.3V output, so actual percent losses must be

calculated for each situation.

Assume that the average inductor current is equal to

load current and decide whether or not the inductor

must withstand continuous fault conditions. If maxi-

mum load current is 0.5A, for instance, a 0.5A inductor

may not survive a continuous 4.5A overload condition.

Dead shorts will actually be more gentle on the induc-

tor because the LT1506 has foldback current limiting.

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current

can be significantly higher than output current, espe-

cially with smaller inductors and lighter loads, so don’t

omit this step. Powdered iron cores are forgiving

because they saturate softly, whereas ferrite cores

saturate abruptly. Other core materials fall in between

somewhere. The following formula assumes continu-

ous mode of operation, but it errs only slightly on the

high side for discontinuous mode, so it can be used for

all conditions.

VVV

fLV

PEAK OUT OUT IN OUT

=+ −

()

()()( )

= Maximum input voltage

f = Switching frequency, 500kHz

3. Decide if the design can tolerate an “open” core geom-

etry like a rod or barrel, which have high magnetic field

radiation, or whether it needs a closed core like a toroid

to prevent EMI problems. One would not want an open

core next to a magnetic storage media, for instance!

This is a tough decision because the rods or barrels are

temptingly cheap and small and there are no helpful

guidelines to calculate when the magnetic field radia-

tion will be a problem.

4. Start shopping for an inductor (see representative

surface mount units in Table 2) which meets the

requirements of core shape, peak current (to avoid

saturation), average current (to limit heating), and fault

current (if the inductor gets too hot, wire insulation will

melt and cause turn-to-turn shorts). Keep in mind that

all good things like high efficiency, low profile, and high

temperature operation will increase cost, sometimes

dramatically. Get a quote on the cheapest unit first to

calibrate yourself on price, then ask for what you really

want.

5. After making an initial choice, consider the secondary

things like output voltage ripple, second sourcing, etc.

Use the experts in the Linear Technology’s applica-

tions department if you feel uncertain about the final

choice. They have experience with a wide range of

inductor types and can tell you about the latest devel-

opments in low profile, surface mounting, etc.

LT1506

APPLICATIONS INFORMATION

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Table 2

SERIES CORE

VENDOR/ VALUE DC CORE RESIS- MATER- HEIGHT

PART NO. (

H) (Amps) TYPE TANCE(

Ω

) IAL (mm)

Coiltronics

CTX2-1 2 4.1 Tor 0.011 KMµ4.2

CTX5-4 5 4.4 Tor 0.019 KMµ6.4

CTX8-4 8 3.5 Tor 0.020 KMµ6.4

CTX2-1P 2 3.4 Tor 0.014 52 4.2

CTX2-3P 2 4.6 Tor 0.012 52 4.8

CTX5-4P 5 3.3 Tor 0.027 52 6.4

Sumida

CDRH125 10 4.0 SC 0.025 Fer 6

CDRH125 12 3.5 SC 0.027 Fer 6

CDRH125 15 3.3 SC 0.030 Fer 6

CDRH125 18 3.0 SC 0.034 Fer 6

Coilcraft

DT3316-222 2.2 5 SC 0.035 Fer 5.1

DT3316-332 3.3 5 SC 0.040 Fer 5.1

DT3316-472 4.7 3 SC 0.045 Fer 5.1

Pulse

PE-53650 4 4.8 Tor 0.017 Fer 9.1

PE-53651 5 5.4 Tor 0.018 Fer 9.1

PE-53652 9 5.5 Tor 0.022 Fer 10

PE-53653 16 5.1 Tor 0.032 Fer 10

Dale

IHSM-4825 2.7 5.1 Open 0.034 Fer 5.6

IHSM-4825 4.7 4.0 Open 0.047 Fer 5.6

IHSM-5832 10 4.3 Open 0.053 Fer 7.1

IHSM-5832 15 3.5 Open 0.078 Fer 7.1

IHSM-7832 22 3.8 Open 0.054 Fer 7.1

Tor = Toroid

SC = Semiclosed geometry

Fer = Ferrite core material

52 = Type 52 powdered iron core material

KMµ = Kool Mµ

Output Capacitor

The output capacitor is normally chosen by its Effective

Series Resistance (ESR), because this is what determines

output ripple voltage. At 500kHz, any polarized capacitor

is essentially resistive. To get low ESR takes

volume

, so

physically smaller capacitors have high ESR. The ESR

range for typical LT1506 applications is 0.05Ω to 0.2Ω. A

typical output capacitor is an AVX type TPS, 100µF at 10V,

with a guaranteed ESR less than 0.1Ω. This is a “D” size

surface mount solid tantalum capacitor. TPS capacitors

are specially constructed and tested for low ESR, so they

give the lowest ESR for a given volume. The value in

microfarads is not particularly critical, and values from

22µF to greater than 500µF work well, but you cannot

cheat mother nature on ESR. If you find a tiny 22µF solid

tantalum capacitor, it will have high ESR, and output ripple

voltage will be terrible. Table 3 shows some typical solid

tantalum surface mount capacitors.

Table 3. Surface Mount Solid Tantalum Capacitor ESR

and Ripple Current

E Case Size ESR (Max.,

Ω

) Ripple Current (A)

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

AVX TAJ 0.7 to 0.9 0.4

D Case Size

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

C Case Size

AVX TPS 0.2 (typ) 0.5 (typ)

Many engineers have heard that solid tantalum capacitors

are prone to failure if they undergo high surge currents.

This is historically true, and type TPS capacitors are

specially tested for surge capability, but surge ruggedness

is not a critical issue with the

output

capacitor. Solid

tantalum capacitors fail during very high

turn-on

surges,

which do not occur at the output of regulators. High

discharge

surges, such as when the regulator output is

dead shorted, do not harm the capacitors.

Unlike the input capacitor, RMS ripple current in the

output capacitor is normally low enough that ripple cur-

rent rating is not an issue. The current waveform is

triangular with a typical value of 200mA

RMS

. The formula

to calculate this is:

Output Capacitor Ripple Current (RMS):

IVVV

LfV

RIPPLE RMS OUT IN OUT

(

)

()

−

()

()()( )

029.

LT1506

APPLICATIONS INFORMATION

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Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now

available in smaller case sizes. These are ideal for input

bypassing because of their high ripple rating and tolerance

to turn-on surges. As output capacitors, caution must be

used. Solid tantalum capacitor’s ESR generates a loop

“zero” at 5kHz to 50kHz that is beneficial in giving accept-

able loop phase margin. Ceramic capacitors remain ca-

pacitive to beyond 300kHz and usually resonate with their

ESL before ESR becomes effective. When using ceramic

output capacitors, the loop compensation pole frequency

must be reduced by a typical factor of 10.

OUTPUT RIPPLE VOLTAGE

Figure 3 shows a typical output ripple voltage waveform

for the LT1506. Ripple voltage is determined by the high

frequency impedance of the output capacitor, and ripple

current through the inductor. Peak-to-peak ripple current

through the inductor into the output capacitor is:

IVVV

VLf

POUT IN OUT

-P =

()

−

()

()()()

For high frequency switchers, the sum of ripple current

slew rates may also be relevant and can be calculated

from:

ΣdI

Peak-to-peak output ripple voltage is the sum of a

triwave

created by peak-to-peak ripple current times ESR, and a

square

wave created by parasitic inductance (ESL) and

ripple current slew rate. Capacitive reactance is assumed

to be small compared to ESR or ESL.

V I ESR ESL dI

RIPPLE

()( )

()

P-P

Example: with V

=10V, V

OUT

= 5V, L = 10µH, ESR = 0.1Ω,

ESL = 10nH:

RIPPLE

P-P

()

−

()









=

()()

+









=+=

−

510 5

10 10 10 500 10 05

10 10 10

05 01 10 10 10

0 05 0 01 60

••

•

.. •

OUT

AT I

OUT

= 1A

OUT

AT I

OUT

= 50mA

INDUCTOR CURRENT

AT I

OUT

= 1A

0.5µs/DIV 1374 F03

INDUCTOR CURRENT

AT I

OUT

= 50mA

20mV/DIV

0.5A/DIV

Figure 3. LT1506 Ripple Voltage Waveform

CATCH DIODE

The suggested catch diode (D1) is a 1N5821 Schottky, or

its Motorola equivalent, MBR330. It is rated at 3A average

forward current and 30V reverse voltage. Typical forward

voltage is 0.5V at 3A. The diode conducts current only

during switch off time. Peak reverse voltage is equal to

regulator input voltage. Average forward current in normal

operation can be calculated from:

IIVV

D AVG OUT IN OUT

(

)

=−

()

This formula will not yield values higher than 3A with

maximum load current of 4.25A unless the ratio of input to

output voltage exceeds 3.4:1. The only reason to consider

a larger diode is the worst-case condition of a high input

voltage and

overloaded

(not shorted) output. Under short-

circuit conditions, foldback current limit will reduce diode

current to less than 2.6A, but if the output is overloaded

LT1506

APPLICATIONS INFORMATION

WUUU

and does not fall to less than 1/3 of nominal output voltage,

foldback will not take effect. With the overloaded condi-

tion, output current will increase to a typical value of 5.7A,

determined by peak switch current limit of 6A. With

= 15V, V

OUT

= 4V (5V overloaded) and I

OUT

= 5.7A:

D AVG

()

=−

()

5 7 15 4

15 418

This is safe for short periods of time, but it would be

prudent to check with the diode manufacturer if continu-

ous operation under these conditions must be tolerated.

BOOST␣ PIN␣ CONSIDERATIONS

For most applications, the boost components are a 0.27µF

capacitor and a 1N914 or 1N4148 diode. The anode is

connected to the regulated output voltage and this gener-

ates a voltage across the boost capacitor nearly identical

to the regulated output. In certain applications, the anode

may instead be connected to the unregulated input volt-

age. This could be necessary if the regulated output

voltage is very low (< 3V) or if the input voltage is less than

5V. Efficiency is not affected by the capacitor value, but the

capacitor should have an ESR of less than 1Ω to ensure

that it can be recharged fully under the worst-case condi-

tion of minimum input voltage. Almost any type of film or

ceramic capacitor will work fine.

For nearly all applications, a 0.27µF boost capacitor works

just fine, but for the curious, more details are provided

here. The size of the boost capacitor is determined by

switch drive current requirements. During switch on time,

drain current on the capacitor is approximately I

OUT

/ 50. At

peak load current of 4.25A, this gives a total drain of 85mA.

Capacitor ripple voltage is equal to the product of on time

and drain current divided by capacitor value;

∆V = (t

)(85mA/C). To keep capacitor ripple voltage to

less than 0.6V (a slightly arbitrary number) at the worst-

case condition of t

= 1.8µs, the capacitor needs to be

0.27µF. Boost capacitor ripple voltage is not a critical

parameter, but if the minimum voltage across the capaci-

tor drops to less than 3V, the power switch may not

saturate fully and efficiency will drop. An

approximate

formula for absolute minimum capacitor value is:

CIVV

fV V

MIN OUT OUT IN

OUT

()( )

()

−

()

f = Switching frequency

OUT

= Regulated output voltage

= Minimum input voltage

This formula can yield capacitor values substantially less

than 0.27µF, but it should be used with caution since it

does not take into account secondary factors such as

capacitor series resistance, capacitance shift with tem-

perature and output overload.

SHUTDOWN FUNCTION AND UNDERVOLTAGE

LOCKOUT

Figure 4 shows how to add undervoltage lockout (UVLO)

to the LT1506. Typically, ULVO is used in situations where

the input supply is

current limited

, or has a relatively high

source resistance. A switching regulator draws constant

power from the source, so source current increases as

source voltage drops. This looks like a negative resistance

load to the source and can cause the source to current limit

or latch low under low source voltage conditions. ULVO

prevents the regulator from operating at source voltages

where these problems might occur.

Threshold voltage for lockout is about 2.38V, slightly less

than the internal 2.42V reference voltage. A 3.5µA bias

current flows

out

of the pin at threshold. This internally

generated current is used to force a default high state on

the shutdown pin if the pin is left open. When low shut-

down current is not an issue, the error due to this current

can be minimized by making R

10k or less. If shutdown

current is an issue, R

can be raised to 100k, but the error

due to initial bias current and changes with temperature

should be considered.

RRV V

VR A

HI LO IN

()

=−

()

−

()

238

238 35

to 100k 25k suggested

..µ

= Minimum input voltage

LT1506

APPLICATIONS INFORMATION

WUUU

2.38V

0.4V

GND

VSW

LT1506

INPUT

RFB

RHI

1506 F04

OUTPUT

SHDN

LOCKOUT

TOTAL

SHUTDOWN

3.5µA

RLO

Figure 4. Undervoltage Lockout

Keep the connections from the resistors to the shutdown

pin short and make sure that interplane or surface capaci-

tance to the switching nodes are minimized. If high

resistor values are used, the shutdown pin should be

bypassed with a 1000pF capacitor to prevent coupling

problems from the switch node. If hysteresis is desired in

the undervoltage lockout point, a resistor R

can be

added to the output node. Resistor values can be calcu-

lated from:

RRV VV V

RRV V

LO IN OUT

FB HI OUT

=−+

()

[]

−

()

()( )

238 1

238 235

∆∆

∆

25k suggested for R

= Input voltage at which switching stops as input

voltage descends to trip level

∆V = Hysteresis in input voltage level

Example: output voltage is 5V, switching is to stop if input

voltage drops below 6V and should not restart unless

input rises back to 7.5V. ∆V is therefore 1.5V and V

= 6V.

Let R

= 25k.

Rk k

=−+

()

[]

−

()

25 623815 5 1 15

238 25 35

25 5 2

229 48

48 5 1 5 160

../ .

SWITCH NODE CONSIDERATIONS

For maximum efficiency, switch rise and fall times are

made as short as possible. To prevent radiation and high

frequency resonance problems, proper layout of the com-

ponents connected to the switch node is essential. B field

(magnetic) radiation is minimized by keeping catch diode,

switch pin, and input bypass capacitor leads as short as

possible. E field radiation is kept low by minimizing the

length and area of all traces connected to the switch pin

and BOOST pin. A ground plane should always be used

under the switcher circuitry to prevent interplane cou-

pling. A suggested layout for the critical components is

shown in Figure 5. Note that the feedback resistors and

compensation components are kept as far as possible

LT1506

APPLICATIONS INFORMATION

WUUU

from the switch node. Also note that the high current

ground path of the catch diode and input capacitor are kept

very short and separate from the analog ground line.

The high speed switching current path is shown schemati-

cally in Figure 6. Minimum lead length in this path is

essential to ensure clean switching and low EMI. The path

including the switch, catch diode, and input capacitor is

the only one containing nanosecond rise and fall times. If

you follow this path on the PC layout, you will see that it is

irreducibly short. If you move the diode or input capacitor

away from the LT1506, get your resumé in order. The

other paths contain only some combination of DC and

500kHz triwave, so are much less critical.

Figure 5. Suggested Layout (Topside Only Shown)

Figure 6. High Speed Switching Path

CONNECT TO

GROUND PLANE

KEEP FB AND V

COMPONENTS

AWAY FROM HIGH FREQUENCY,

HIGH CURRENT COMPONENTS

PLACE FEEDTHROUGHS

AROUND GND PIN FOR GOOD

THERMAL CONDUCTIVITY

R3 D2

1506 F05

C5 C6GND

OUT

D1C3V

GND

CONNECT TO

GROUND PLANE

MINIMIZE LT1506 C3, D1 LOOP

KELVIN SENSE

OUT

TAKE OUTPUT

DIRECTLY FROM

END OF OUTPUT

CAPACITOR

1506 F06

HIGH

FREQUENCY

CIRCULATING

PATH

LOAD

SWITCH NODE

LT1506

APPLICATIONS INFORMATION

WUUU

PARASITIC RESONANCE

Resonance or “ringing” may sometimes be seen on the

switch node (see Figure 7). Very high frequency ringing

following switch rise time is caused by switch/diode/input

capacitor lead inductance and diode capacitance. Schot-

tky diodes have very high “Q” junction capacitance that

can ring for many cycles when excited at high frequency.

If total lead length for the input capacitor, diode and switch

path is 1 inch, the inductance will be approximately 25nH.

At switch off, this will produce a spike across the NPN

output device in addition to the input voltage. At higher

currents this spike can be in the order of 10V to 20V or

higher with a poor layout, potentially exceeding the abso-

lute max switch voltage. The path around switch, catch

diode and input capacitor must be kept as short as

possible to ensure reliable operation. When looking at this,

a >100MHz oscilloscope must be used, and waveforms

should be observed on the leads of the package. This

switch off spike will also cause the SW node to go below

ground. The LT1506 has special circuitry inside which

RISE AND FALL

WAVEFORMS ARE

SUPERIMPOSED

(PULSE WIDTH IS

NOT

120ns)

5V/DIV

Figure 7. Switch Node Resonance

20ns/DIV 1375/76 F07

INDUCTOR

CURRENT

20ns/DIV 1375/76 F11

0.5µs/DIV 1375/76 F08

Figure 8. Discontinuous Mode Ringing

5V/DIV

100mA/DIV

SWITCH NODE

VOLTAGE

mitigates this problem, but negative voltages over 1V

lasting longer than 10ns should be avoided. Note that

100MHz oscilloscopes are barely fast enough to see the

details of the falling edge overshoot in Figure 7.

A second, much lower frequency ringing is seen during

switch off time if load current is low enough to allow the

inductor current to fall to zero during part of the switch off

time (see Figure 8). Switch and diode capacitance reso-

nate with the inductor to form damped ringing at 1MHz to

10 MHz. This ringing is not harmful to the regulator and it

has not been shown to contribute significantly to EMI. Any

attempt to damp it with a resistive snubber will degrade

efficiency.

INPUT BYPASSING AND VOLTAGE RANGE

Input Bypass Capacitor

Step-down converters draw current from the input supply

in pulses. The average height of these pulses is equal to

load current, and the duty cycle is equal to V

OUT

. Rise

and fall time of the current is very fast. A local bypass

capacitor across the input supply is necessary to ensure

proper operation of the regulator and minimize the ripple

current fed back into the input supply.

The capacitor also

forces switching current to flow in a tight local loop,

minimizing EMI

Do not cheat on the ripple current rating of the Input

bypass capacitor, but also don’t get hung up on the value

in microfarads

. The input capacitor is intended to absorb

all the switching current ripple, which can have an RMS

value as high as one half of load current. Ripple current

ratings on the capacitor must be observed to ensure

reliable operation. In many cases it is necessary to parallel

two capacitors to obtain the required ripple rating. Both

capacitors must be of the same value and manufacturer to

guarantee power sharing. The actual value of the capacitor

in microfarads is not particularly important because at

500kHz, any value above 5µF is essentially resistive. RMS

ripple current rating is the critical parameter. Actual RMS

current can be calculated from:

IIVVVV

RIPPLE RMS OUT OUT IN OUT IN

(

)

=−

()

LT1506

APPLICATIONS INFORMATION

WUUU

The term inside the radical has a maximum value of 0.5

when input voltage is twice output, and stays near 0.5 for

a relatively wide range of input voltages. It is common

practice therefore to simply use the worst-case value and

assume that RMS ripple current is one half of load current.

At maximum output current of 4.5A for the LT1506, the

input bypass capacitor should be rated at 2.25A ripple

current. Note however, that there are many secondary

considerations in choosing the final ripple current rating.

These include ambient temperature, average versus peak

load current, equipment operating schedule, and required

product lifetime. For more details, see Application Notes

19 and 46, and Design Note 95.

Input Capacitor Type

Some caution must be used when selecting the type of

capacitor used at the input to regulators. Aluminum

electrolytics are lowest cost, but are physically large to

achieve adequate ripple current rating, and size con-

straints (especially height), may preclude their use.

Ceramic capacitors are now available in larger values, and

their high ripple current and voltage rating make them

ideal for input bypassing. Cost is fairly high and footprint

may also be somewhat large. Solid tantalum capacitors

would be a good choice, except that they have a history of

occasional spectacular failures when they are subjected to

large current surges during power-up. The capacitors can

short and then burn with a brilliant white light and lots of

nasty smoke. This phenomenon occurs in only a small

percentage of units, but it has led some OEM companies

to forbid their use in high surge applications. The input

bypass capacitor of regulators can see these high surges

when a battery or high capacitance source is connected.

Several manufacturers have developed a line of solid

tantalum capacitors specially tested for surge capability

(AVX TPS series for instance, see Table 3), but even these

units may fail if the input voltage surge approaches the

maximum voltage rating of the capacitor. AVX recom-

mends derating capacitor voltage by 2:1 for high surge

applications.

Larger capacitors may be necessary when the input volt-

age is very close to the minimum specified on the data

sheet. Small voltage dips during switch on time are not

normally a problem, but at very low input voltage they may

cause erratic operation because the input voltage drops

below the minimum specification. Problems can also

occur if the input-to-output voltage differential is near

minimum. The amplitude of these dips is normally a

function of capacitor ESR and ESL because the capacitive

reactance is small compared to these terms. ESR tends to

be the dominate term and is inversely related to physical

capacitor size within a given capacitor type.

SYNCHRONIZING (-SYNC Option for DD Package)

The SYNC pin, is used to synchronize the internal oscilla-

tor to an external signal. The SYNC input must pass from

a logic level low, through the maximum synchronization

threshold with a duty cycle between 10% and 90%. The

input can be driven directly from a logic level output. The

synchronizing range is equal to

initial

operating frequency

up to 1MHz. This means that

minimum

practical sync

frequency is equal to the worst-case

high

self-oscillating

frequency (560kHz), not the typical operating frequency of

500kHz. Caution should be used when synchronizing

above 700kHz because at higher sync frequencies the

amplitude of the internal slope compensation used to

prevent subharmonic switching is reduced. This type of

subharmonic switching only occurs at input voltages less

than twice output voltage. Higher inductor values will tend

to eliminate this problem. See Frequency Compensation

section for a discussion of an entirely different cause of

subharmonic switching before assuming that the cause is

insufficient slope compensation. Application Note 19 has

more details on the theory of slope compensation.

At power-up, when V

is being clamped by the FB pin (see

Figure 2, Q2), the sync function is disabled. This allows the

frequency foldback to operate in the shorted output con-

dition. During normal operation, switching frequency is

controlled by the internal oscillator until the FB pin reaches

1.5V, after which the SYNC pin becomes operational.

THERMAL CALCULATIONS

Power dissipation in the LT1506 chip comes from four

sources: switch DC loss, switch AC loss, boost circuit

current, and input quiescent current. The following

LT1506

APPLICATIONS INFORMATION

WUUU

formulas show how to calculate each of these losses.

These formulas assume continuous mode operation, so

they should not be used for calculating efficiency at light

load currents.

Switch loss:

PRI V

Vns I V f

SW SW OUT OUT

IN OUT IN

()( )

()()()

Boost current loss:

PVI

BOOST OUT OUT

()

50/

Quiescent current loss:

PV V

Q IN OUT

OUT

()











()

0 001 0 005

0 002

.. .

= Switch resistance (≈0.07)

24ns = Equivalent switch current/voltage overlap time

f = Switch frequency

Example: with V

= 10V, V

OUT

= 5V and I

OUT

= 3A:

BOOST

( )()()

+





()( )







=+=

()( )

()

()( )

−

007 3 5

10 24 10 3 10 500 10

0 32 0 36 0 68

5350

10 015

10 0 001 5 0 005 5 0 002

10 004

.••

...

.. ..

Total power dissipation is 0.68 + 0.15 + 0.04 = 0.87W.

Thermal resistance for LT1506 package is influenced by

the presence of internal or backside planes. With a full

plane under the SO package, thermal resistance will be

about 80°C/W. No plane will increase resistance to about

120°C/W. To calculate die temperature, use the proper

thermal resistance number for the desired package and

add in worst-case ambient temperature:

= T

+ θ

TOT

)

With the SO-8 package (θ

= 80°C/W), at an ambient

temperature of 50°C,

= 50 + 80 (0.87) = 120°C

Die temperature is highest at low input voltage, so use

lowest continuous input operating voltage for thermal

calculations.

FREQUENCY COMPENSATION

Loop frequency compensation of switching regulators

can be a rather complicated problem because the reactive

components used to achieve high efficiency also

introduce multiple poles into the feedback loop. The

inductor and output capacitor on a conventional step-

down converter actually form a resonant tank circuit that

can exhibit peaking and a rapid 180° phase shift at the

resonant frequency. By contrast, the LT1506 uses a “cur-

rent mode” architecture to help alleviate phase shift cre-

ated by the inductor. The basic connections are shown in

Figure 9. Figure 10 shows a Bode plot of the phase and gain

of the power section of the LT1506, measured from the V

pin to the output. Gain is set by the 5.3A/V transconduc-

tance of the LT1506 power section and the effective

complex impedance from output to ground. Gain rolls off

smoothly above the 600Hz pole frequency set by the

100µF output capacitor. Phase drop is limited to about

70°. Phase recovers and gain levels off at the zero fre-

quency (≈16kHz) set by capacitor ESR (0.1Ω).

Figure 9. Model for Loop Response

–

2.42V

LT1506

GND

1506 F09

OUTPUT

ESR

ERROR

AMPLIFIER

CURRENT MODE

POWER STAGE

= 5.3A/V

LT1506

APPLICATIONS INFORMATION

WUUU

Figure 10. Response from VC Pin to Output

FREQUENCY (Hz)

GAIN (µMho)

PHASE (DEG)

3000

2500

2000

1500

1000

500

200

150

100

–50

100 10k 100k 10M

1506 F11

1k 1M

GAIN

PHASE

OUT

200k

OUT

12pF

ERROR AMPLIFIER EQUIVALENT CIRCUIT

LOAD

= 50Ω

2 × 10

–3

)(

Figure 11. Error Amplifier Gain and Phase

Figure 12. Overall Loop Characteristics

What About a Resistor in the Compensation Network?

It is common practice in switching regulator design to add

a “zero” to the error amplifier compensation to increase

loop phase margin. This zero is created in the external

network in the form of a resistor (R

) in series with the

compensation capacitor. Increasing the size of this resis-

tor generally creates better and better loop stability, but

there are two limitations on its value. First, the combina-

tion of output capacitor ESR and a large value for R

may

cause loop gain to stop rolling off altogether, creating a

gain margin problem. An approximate formula for R

where gain margin falls to zero is:

R Loop V

G G ESR

COUT

MP MA

Gain =1

()

()()()()

242.

FREQUENCY (Hz)

GAIN: V

PIN TO OUTPUT (dB)

PHASE: V

PIN TO OUTPUT (DEG)

–20

–40

–80

–120

10 1k 10k 1M

1505 F10

100 100k

GAIN

PHASE

= 10V

OUT

= 5V

OUT

= 2A

FREQUENCY (Hz)

LOOP GAIN (dB)

LOOP PHASE (DEG)

–20

200

150

100

–50

10 1k 10k 1M

1505 F12

100 100k

GAIN

PHASE

= 10V

OUT

= 5V, I

OUT

= 2A

OUT

= 100µF, 10V, AVX TPS

= 1.5nF, R

= 0, L = 10µH

Error amplifier transconductance phase and gain are shown

in Figure 11. The error amplifier can be modeled as a

transconductance of 2000µMho, with an output imped-

ance of 200kΩ in parallel with 12pF. In all practical

applications, the compensation network from V

pin to

ground has a much lower impedance than the output

impedance of the amplifier at frequencies above 500Hz.

This means that the error amplifier characteristics them-

selves do not contribute excess phase shift to the loop, and

the phase/gain characteristics of the error amplifier sec-

tion are completely controlled by the external compensa-

tion network.

In Figure 12, full loop phase/gain characteristics are

shown with a compensation capacitor of 1.5nF, giving the

error amplifier a pole at 530Hz, with phase rolling off to 90°

and staying there. The overall loop has a gain of 74dB at

low frequency, rolling off to unity-gain at 100kHz. Phase

shows a two-pole characteristic until the ESR of the output

capacitor brings it back above 10kHz. Phase margin is

about 60° at unity-gain.

Analog experts will note that around 4.4kHz, phase dips

very close to the zero phase margin line. This is typical of

switching regulators, especially those that operate over a

wide range of loads. This region of low phase is not a

problem as long as it does not occur near unity-gain. In

practice, the variability of output capacitor ESR tends to

dominate all other effects with respect to loop response.

Variations in ESR

will

cause unity-gain to move around,

but at the same time phase moves with it so that adequate

phase margin is maintained over a very wide range of ESR

(≥ ±3:1).

LT1506

APPLICATIONS INFORMATION

WUUU

cases, the resistor may have to be larger to get acceptable

phase response, and some means must be used to control

ripple voltage at the V

pin. The suggested way to do this

is to add a capacitor (C

) in parallel with the R

network

on the V

pin. Pole frequency for this capacitor is typically

set at one-fifth of switching frequency so that it provides

significant attenuation of switching ripple, but does not

add unacceptable phase shift at loop unity-gain frequency.

With R

= 3k,

CfR kpF

()()()

=





()

2 500 10 3 531

ππ•

How Do I Test Loop Stability?

The “standard” compensation for LT1506 is a 1.5nF

capacitor for C

, with R

= 0. While this compensation will

work for most applications, the “optimum” value for loop

compensation components depends, to various extent, on

parameters which are not well controlled. These include

inductor value

(±30% due to production tolerance, load

current and ripple current variations),

output capacitance

(±20% to ±50% due to production tolerance, tempera-

ture, aging and changes at the load),

output capacitor ESR

(±200% due to production tolerance, temperature and

aging), and finally,

DC input voltage and output load

current

. This makes it important for the designer to check

out the final design to ensure that it is “robust” and tolerant

of all these variations.

I check switching regulator loop stability by pulse loading

the regulator output while observing transient response at

the output, using the circuit shown in Figure 13. The

regulator loop is “hit” with a small transient AC load

current at a relatively low frequency, 50Hz to 1kHz. This

causes the output to jump a few millivolts, then settle back

to the original value, as shown in Figure 14. A well behaved

loop will settle back cleanly, whereas a loop with poor

phase or gain margin will “ring” as it settles. The

number

of rings indicates the degree of stability, and the

frequency

of the ringing shows the approximate unity-gain fre-

quency of the loop.

Amplitude

of the signal is not particu-

larly important, as long as the amplitude is not so high that

the loop behaves nonlinearly.

= Transconductance of power stage = 5.3A/V

= Error amplifier transconductance = 2(10

–3

)

ESR = Output capacitor ESR

2.42 = Reference voltage

With V

OUT

= 5V and ESR = 0.03Ω, a value of 6.5k for R

would yield zero gain margin, so this represents an upper

limit. There is a second limitation however which has

nothing to do with theoretical small signal dynamics. This

resistor sets high frequency gain of the error amplifier,

including the gain at the switching frequency. If switching

frequency gain is high enough, output ripple voltage will

appear at the V

pin with enough amplitude to muck up

proper operation of the regulator. In the marginal case,

subharmonic

switching occurs, as evidenced by alternat-

ing pulse widths seen at the switch node. In more severe

cases, the regulator squeals or hisses audibly even though

the output voltage is still roughly correct. None of this will

show on a theoretical Bode plot because Bode is an

amplitude insensitive analysis.

Tests have shown that if

ripple voltage on the V

is held to less than 100mV

P-P

, the

LT1506 will be well behaved

. The formula below will give

an estimate of V

ripple voltage when R

is added to the

loop, assuming that R

is large compared to the reactance

of C

at 500kHz.

VR G V V ESR

VLf

C RIPPLE C MA IN OUT

(

)

()( )

−

()()()

24.

= Error amplifier transconductance (2000µMho)

If a computer simulation of the LT1506 showed that a

series compensation resistor of 3k gave best overall loop

response, with adequate gain margin, the resulting V

ripple voltage with V

= 10V, V

OUT

= 5V, ESR = 0.1Ω,

L = 10µH, would be:

VkV

C RIPPLE

(

)

−

()





−

()()()

()









=

3 2 10 10 5 0 1 2 4

10 10 10 500 10 0 144

•..

••

This ripple voltage is high enough to possibly create

subharmonic switching. In most situations a compromise

value (<2k in this case) for the resistor gives acceptable

phase margin and no subharmonic problems. In other

LT1506

APPLICATIONS INFORMATION

WUUU

Figure 13. Loop Stability Test Circuit

OSCILLOSCOPE

SYNC

ADJUSTABLE

DC LOAD

ADJUSTABLE

INPUT SUPPLY

100Hz TO 1kHz

100mV TO 1V

P-P

100µF TO

1000µF

RIPPLE FILTER

1506 F13

TO X1

OSCILLOSCOPE

PROBE

3300pF 330pF

50Ω

470Ω4.7k

SWITCHING

REGULATOR

0.2ms/DIV 1375/76 F14

10mV/DIV

OUT

AT I

OUT

500mA

BEFORE FILTER

OUT

AT I

OUT

500mA

AFTER FILTER

OUT

AT I

OUT

= 50mA

AFTER FILTER

LOAD PULSE

THROUGH 50Ω

f ≈ 780Hz

5A/DIV

Figure 14. Loop Stability Check

The output of the regulator contains both the desired low

frequency transient information and a reasonable amount

of high frequency (500kHz) ripple. The ripple makes it

difficult to observe the small transient, so a two-pole,

100kHz filter has been added. This filter is not particularly

critical; even if it attenuated the transient signal slightly,

this wouldn’t matter because amplitude is not critical.

After verifying that the setup is working correctly, I start

varying load current and input voltage to see if I can find

any combination that makes the transient response look

suspiciously “ringy.” This procedure may lead to an

adjustment for best loop stability or faster loop transient

response. Nearly always you will find that loop response

looks better if you add in several kΩ for R

. Do this only

if necessary, because as explained before, R

above 1k

may require the addition of C

to control V

pin ripple. If

everything looks OK, I use a heat gun and cold spray on the

circuit (especially the output capacitor) to bring out any

temperature-dependent characteristics.

Keep in mind that this procedure does not take initial

component tolerance into account. You should see fairly

clean response under all load and line conditions to ensure

that component variations will not cause problems. One

note here: according to Murphy, the component most

likely to be changed in production is the output capacitor,

because that is the component most likely to have manu-

facturer variations (in ESR) large enough to cause prob-

lems. It would be a wise move to lock down the sources of

the output capacitor in production.

A possible exception to the “clean response” rule is at very

light loads, as evidenced in Figure 14 with I

LOAD

= 50mA.

Switching regulators tend to have dramatic shifts in loop

response at very light loads, mostly because the inductor

current becomes discontinuous. One common result is very

slow but stable characteristics. A second possibility is low

phase margin, as evidenced by ringing at the output with

transients. The good news is that the low phase margin at

light loads is not particularly sensitive to component varia-

tion, so if it looks reasonable under a transient test, it will

probably not be a problem in production. Note that

fre-

quency

of the light load ringing may vary with component

tolerance but phase margin generally hangs in there.

CURRENT SHARING MULTIPHASE SUPPLY

The circuit in Figure 15 uses multiple LT1506s to produce

a 5V, 12A power supply. There are several advantages to

using a multiple switcher approach compared to a single

larger switcher. The inductor size is considerably reduced.

Three 4A inductors store less energy (LI

/2) than one 12A

coil so are far smaller. In addition, synchronizing three

LT1506

APPLICATIONS INFORMATION

WUUU

converters 120° out of phase with each other reduces

input and output ripple currents. This reduces the ripple

rating, size and cost of filter capacitors.

Current Sharing/Split Input Supplies

Current sharing is accomplished by joining the V

pins to

a common compensation capacitor. The output of the

error amplifier is a gm stage, so any number of devices can

be connected together. The effective gm of the composite

error amplifier is the multiple of the individual devices. In

Figure 15, the compensation capacitor C4 has been

increased by ×3. Tolerances in the reference voltages

result in small offset currents to flow between the V

pins.

The overall effect is that the loop regulates the output at a

voltage between the minimum and maximum reference of

the devices used. Switch current matching between

devices will be typically better than 300mA. The negative

temperature coefficient of the V

to switch current transcon-

ductance prevents current hogging.

A common V

voltage forces each LT1506 to operate at the

same switch current, not duty cycle. Each device operates

at the duty cycle defined by its respective input voltage. In

Figure 15, the input could be split and each device oper-

ated at a different voltage. The common V

ensures

loading is shared between inputs.

Figure 15. Current Sharing 12A Supply

68nF

25V

C1, C3: MARCON THCS50E1E106Z

D1: ROHM RB051L-40

D2: 1N914

L1: DO3316P-682

C3C

10µF

25V

C2C

330nF

10V

D1C

D2C

1506 F15

L1C

6.8µH

10µF

25V

12A

5.36k

4.99k

1.8MHz

3-BIT RING

COUNTER

C3B

10µF

25V

C2B

330nF

10V

D1B

D2B

L1B

6.8µH

C3A

10µF

25V

INPUT

6V TO 15V

C2A

330nF

10V

D1A

D2A

L1A

6.8µH

SYNC SW GND

LT1506-SYNC

BOOST FBV

SYNC SW GND

LT1506-SYNC

BOOST FBV

SYNC SW GND

LT1506-SYNC

BOOST FB

Synchronized Ripple Currents

A ring counter generates three synchronization signals at

600kHz, 33% duty cycle phased 120° apart. The sync

input will operate over a wide range of duty cycles, so no

further pulse conditioning is needed. Each device’s maxi-

mum input ripple current is a 4A square wave at 600kHz.

When synchronously added together, the ripple remains

at 4A but frequency increases to 1.8MHz. Likewise, the

output ripple current is a 1.8MHz triangular waveform,

with maximum amplitude of 350mA at 10V V

. Interest-

ingly, at 7.6V and 15V V

, the theoretical summed output

ripple current cancels completely. To reduce board space

and ripple voltage, C1 and C3 are ceramic capacitors. Loop

compensation C4 must be adjusted when using ceramic

output capacitors due to the lack of effective series resis-

tance. The typical tantalum compensation of 1.5nF is

increased to 22nF (×3) for the ceramic output capacitor.

If synchronization is not used and the internal oscillators

free run, the circuit will operate correctly, but ripple

cancellation will not occur. Input and output capacitors

must be ripple rated for the total output current.

LT1506

APPLICATIONS INFORMATION

WUUU

Redundant Operation

The circuit shown in Figure 15 is fault tolerant when

operating at less than 8A of output current. If one device

fails, the output will remain in regulation. The feedback

loop will compensate by raising the voltage on the V

pin,

increasing switch current of the two remaining devices.

BUCK CONVERTER WITH ADJUSTABLE SOFT START

Large capacitive loads can cause high input currents at

start-up. Figure 16 shows a circuit that limits the dv/dt of

the output at start-up, controlling the capacitor charge

rate. The buck converter is a typical configuration with the

addition of R3, R4, C

and Q1. As the output starts to rise,

Q1 turns on, regulating switch current via the V

pin to

maintain a constant dv/dt at the output. Output rise time is

controlled by the current through C

defined by R4 and

Q1’s V

. Once the output is in regulation, Q1 turns off and

the circuit operates normally. R3 is transient protection for

the base of Q1.

RiseTime RC V

SS OUT

=()( )( )

()

Using the values shown in Figure 16,

RiseTime ms==

(• )(• )()

–

47 10 15 10 5

07 5

The ramp is linear and rise times in the order of 100ms are

possible. Since the circuit is voltage controlled, the ramp

rate is unaffected by load characteristics and maximum

output current is unchanged. Variants of this circuit can be

used for sequencing multiple regulator outputs.

Dual Output SEPIC Converter

The circuit in Figure 17 generates both positive and

negative 5V outputs with a single piece of magnetics. The

two inductors shown are actually just two windings on a

standard B H Electronics inductor. The topology for the 5V

output is a standard buck converter. The –5V topology

would be a simple flyback winding coupled to the buck

converter if C4 were not present. C4 creates a SEPIC

(Single-Ended Primary Inductance Converter) topology

whicn improves regulation and reduces ripple current in

L1. Without C4, the voltage swing on L1B compared to

L1A would vary due to relative loading and coupling

losses. C4 provides a low impedance path to maintain an

equal voltage swing in L1B, improving regulation. In a

flyback converter, during switch on time, all the converter’s

energy is stroed in L1A only, since no current flows in L1B.

At switch off, energy is transferred by magnetic coupling

into L1B, powering the –5V rail. C4 pulls L1B positive

during switch on time, causing current to flow, and energy

to build in L1B and C4. At switch off, the energy stored in

both L1B and C4 supply the –5V rail. This reduces the

current in L1A and changes L1B current waveform from

square to triangular. For details on this circuit see Design

Note 100.

Figure 16. Buck Converter with Adjustable Soft Start

OUTPUT

–5V

†

* L1 IS A SINGLE CORE WITH TWO WINDINGS

BH ELECTRONICS #501-0726

** TOKIN IE475ZY5U-C304

†

IF LOAD CAN GO TO ZERO, AN OPTIONAL

PRELOAD OF 1k TO 5k MAY BE USED TO

IMPROVE LOAD REGULATION

D1, D3: MBRD340

INPUT

TO 15V

GND

1506 F17

0.27µF

1.5nF D1

C1**

100µF

10V TANT

C5**

100µF

10V TANT

10µF

25V

CERAMIC

C4**

4.7µF

1N914

L1*

6.8µH

L1*

5.36k

4.99k

BOOST

LT1506

FBSHDN

GND V

BOOST

LT1506

OUTPUT

INPUT

12V

1506 F16

0.33µF

100µF

15nF

1.5nF

10µF

1N914

5µH

5.36k

SHDN

GND V

4.99k

47k

Figure 17. Dual Output SEPIC Converter

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

LT1506

1506f LT/TP 1198 4K • PRINTED IN USA

 LINEAR T ECHNOLOGY CORPORATION 1998

Dimensions in inches (millimeters) unless otherwise noted.

PACKAGE DESCRIPTION

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R Package

7-Lead Plastic DD Pak

(LTC DWG # 05-08-1462)

R (DD7) 0396

0.026 – 0.036

(0.660 – 0.914)

0.143+0.012

–0.020

()

3.632+0.305

–0.508

0.040 – 0.060

(1.016 – 1.524) 0.013 – 0.023

(0.330 – 0.584)

0.095 – 0.115

(2.413 – 2.921)

0.004+0.008

–0.004

()

0.102+0.203

–0.102

0.050 ± 0.012

(1.270 ± 0.305)

0.059

(1.499)

TYP

0.045 – 0.055

(1.143 – 1.397)

0.165 – 0.180

(4.191 – 4.572)

0.330 – 0.370

(8.382 – 9.398)

0.060

(1.524)

TYP

0.390 – 0.415

(9.906 – 10.541)

15° TYP

0.300

(7.620)

0.075

(1.905)

0.183

(4.648)

0.060

(1.524)

0.060

(1.524)

0.256

(6.502)

BOTTOM VIEW OF DD PAK

HATCHED AREA IS SOLDER PLATED

COPPER HEAT SINK

1234

0.150 – 0.157**

(3.810 – 3.988)

8765

0.189 – 0.197*

(4.801 – 5.004)

0.228 – 0.244

(5.791 – 6.197)

0.016 – 0.050

0.406 – 1.270

0.010 – 0.020

(0.254 – 0.508)× 45°

0°– 8° TYP

0.008 – 0.010

(0.203 – 0.254)

SO8 0996

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

TYP

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

●

FAX: (408) 434-0507

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