y TECHNOLOGY LIN } \D LT1513/LT1513-2 SEPIC Constant- or Programmable-Current/ Constant-Voltage Battery Charger FEATURES Charger Input Voltage May Be Higher, Equal to or Lower Than Battery Voltage Charges Any Number of Cells Up to 20V 1% Voltage Accuracy for Rechargeable Lithium Batteries 100mV Current Sense Voltage for High Efficiency (LT1513) OmV Current Sense Voltage for Easy Current Programming (LT1513-2) Battery Can Be Directly Grounded 500kHz Switching Frequency Minimizes Inductor Size Charging Current Easily Programmable or Shut Down APPLICATIONS = Charging of NiCd, NiMH, Lead-Acid or Lithium Rechargeable Cells Precision Current Limited Power Supply Constant-Voltage/Constant-Current Supply Transducer Excitation Universal Input CCFL Driver DESCRIPTION The LT1513 is a 500kHz current mode switching regula- tor specially configured to create a constant- or program- mable-current/constant-voltage battery charger. In addition to the usual voltage feedback node, it has a current sense feedback circuit for accurately controlling output current of a flyback or SEPIC (Single-Ended Primary Inductance Converter) topology charger. These topologies allow the current sense circuit to be ground referred and completely separated from the battery itself, simplifying battery switch- ing and system grounding problems. In addition, these topologies allow charging even when the input voltage is lower than the battery voltage. The LT1513 can also drive a CCFL Royer converter with high efficiency in floating or grounded mode. Maximum switch current on the LT1513 is 3A. This allows battery charging currents up to 2A for a single lithium-ion cell. Accuracy of 1% in constant-voltage mode is perfect for lithium battery applications. Charging current can be easily programmed for all battery types. 4Y, LTC and LT are registered trademarks of Linear Technology Corporation. TYPICAL APPLICATION WALL ADAPTER LIA* Maximum Charging Current 2.4 +] 63 INPUT uF 47uF ont 2.2 SINGLE Li-lon CELL Tosv 1.254 2.0 (4.1V) CHARGE ~ L1B* 1.8 UBLE Li-lon SYNC sf Ez 46 CELL ( AND/OR = SHUTDOWN SHUTDOWN ct a 14 ee SR2 | e 22uF C4 12V, TS 5V ot LL x2 1.0 C4 S$R3 > 65 0.22F $0.08a 08 TOU 0.6 BATTERY . + VOLTAGE * LIA, L1B ARE TWO 10H WINDINGS ON A COMMON CORE: COILTRONICS CTX10-4 ** CERAMIC MARCON THCR40EIE475Z OR TOKIN 1475ZY5U-C304 + MBRD340 OR MBRS340T3. MBRD340 HAS 5yA TYPICAL LEAKAGE, MBRS340T3 SOpA TYPICAL Figure 1. SEPIC Charger with 1.25A Output Current 0.4 111543 * TA01 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 INDUCTOR = 10H ACTUAL PROGRAMMED CHARGING CURRENT WILL BE INDEPENDENT OF INPUT VOLTAGE IF IT DOES NOT EXCEED VALUES SHOWN 111513 + TAI2 LY WhineLT1513/LT1513-2 ABSOLUTE MAXIMUM RATINGS SUPPly VOIAQE oo... cece ccs esecececsssetesetecssseseseases 30V Operating Junction Temperature Range SWITCH VOITAQGE oe ecccececcccecsesetetecetsesesetecseeeseseeanes 40V Ds ko) ko | Ce 0C to 125C S/S Pin VOUHAGE oe cccseccscsestecsssssteessscseeesecseeesevaes 30V LTVD VSD oo ceeeessssseteesesseeeseneeeees -40C to 125C FB Pin Voltage (Transient, 10MS) oes +10V Short Circuit... eects 0C to 150C Veep PIN CUITENE 0... ccccecceeeteteeseeeteeseseteeseeeees 10mA Storage Temperature Range................ -65C to 150C lrg Pin Voltage (Transient, 10MS) oes +10V _ Lead Temperature (Soldering, 10 sec)............0. 300C FRONT VIEW ORDER PART ORDER PART NUMBER NUMBER sy FRONT VIEW TAB Vew LT15130R Vy LT1513CT7-2 GND Ira LT1513CR-2 O Vgw LT15131T/-2 ve LT15131R en 7-LEAD PLASTIC DD T7 PACKAGE Tymax = 125C, Oya = 80C/ W 7-LEAD TO-220 WITH PACKAGE SOLDERED TO 0.5INCH? COPPER Tymax = 125C, Oyq = 50C/W, Oj = 4C/W AREA OVER BACKSIDE GROUND PLANE OR INTERNAL POWER PLANE, 6), CAN VARY FROM 20C/W TO > 40C/W DEPENDING ON MOUNTING TECHNIQUE Consult factory for Military grade parts. Vin = 5V, Ve = 0.6V, Vep = Veer, Ipg = OV, Vw and S/S pins open, unless otherwise noted. SYMBOL | PARAMETER CONDITIONS MIN TYP MAX UNITS Veer FB Reference Voltage Measured at FB Pin 1.233 1.245 1.257 V Vo = 0.8V @| 1.228 1.245 1.262 V FB Input Current Vee = Veer 300 550 nA e 600 nA FB Reference Voltage Line Regulation 2.7V < Vin $ 25V, Vo = 0.8V e 0.01 0.03 olV Viper lrg Reference Voltage (LT1513) Measured at Irp Pin -107 -100 93 mV Veg = OV, Vo = 0.8V @| -110 -100 90 mV lrg Input Current Vieg = Viner (Note 2) e 10 25 35 pA Irg Reference Voltage Line Regulation 2.7V < Vin $ 25V, Vo = 0.8V e 0.01 0.05 oN leBvos Irg Voltage Offset (LT1513-2) (Note 3) lyrg = 6OpA (Note 4) @| -75 2.5 12.5 mV lrg Input Current Viep = Virer @; -200 -10 0 nA Vep Source Current Viper = 10mV, Veg = 1.2V @| -700 -300 -100 pA Om Error Amplifier Transconductance Alc = +25yA 1100 1500 1900 pmho @| 700 2300 uumho Error Amplifier Source Current Vep = Vaer 150mV, Vo = 1.5V @; 120 200 350 pA Error Amplifier Sink Current Vep = Veer + 150mV, Ve = 1.5V e 1400 2400 pALT1513/LT1513-2 ELECTRICAL CHARACTERISTICS Vin = 5V, Ve = 0.6V, Veg = Vrer, Ig = OV, Vow and S/S pins open, unless otherwise noted. SYMBOL | PARAMETER CONDITIONS MIN TYP MAX UNITS Error Amplifier Clamp Voltage High Clamp, Vrg = 1V 1.70 1.95 2.30 V Low Clamp, Vrg = 1.5V 0.25 0.40 0.52 V Ay Error Amplifier Voltage Gain 500 VV Vc Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V f Switching Frequency 2.7V < Vin < 25V 450 500 550 kHz 0C <T, < 125C 430 500 580 kHz Tj < 0C 400 580 kHz Maximum Switch Duty Cycle e 85 95 % Switch Current Limit Blanking Time 130 260 ns BV Output Switch Breakdown Voltage 0C <Ty < 125C 40 47 V Tj < 0C 35 V Vsat Output Switch ON Resistance Isw = 2A e 0.25 0.45 Q lim Switch Current Limit Duty Cycle = 50% @; 3.0 3.8 5.4 A Duty Cycle = 80% (Note 1) @| 26 3.4 5.0 A Aljy/Algy | Supply Current Increase During Switch ON Time 15 25 mA/A Control Voltage to Switch Current 4 AV Transconductance Minimum Input Voltage e 2.4 27 V lg Supply Current 2.7V <Viy < 25V e 4 5.5 mA Shutdown Supply Current 2.7V < Vin < 25V, Vsj5 < 0.6V, Tj > 0C | @ 12 30 pA Ty < 0C 50 pA Shutdown Threshold 2.7V <Viy < 25V @| 06 1.3 2 V Shutdown Delay e 5 12 25 bs S/S Pin Input Current OV < Vs/g < 5V @; -10 15 pA Synchronization Frequency Range @| 600 800 kHz The @ denotes specifications which apply over the full operating temperature range. Note 1: For duty cycles (DC) between 50% and 85%, minimum guaranteed switch current is given by I.) = 1.33 (2.75 DC). Note 2: The I-p pin is servoed to its regulating state with Vp = 0.8V. Note 3: Consult factory for grade selected parts. Note 4: The Irp pin is sevoed to regulate FB to 1.245V LY WhineLT1513/LT1513-2 TYPICAL PERFORMANCE CHARACTERISTICS Switch Saturation Voltage Switch Current Limit Minimum Input Voltage vs Switch Current vs Duty Cycle vs Temperature 6 3.0 = _ 8 2.8 << = _ 25C AND _ 5 = 4 1125G = = 26 oO = => | lu : - Bos | 3 E a 3 a 24 << oc S oc 5 > oO KE & 2 5 wa o 2 S 22 x E ~ S B = 1 2.0 wn 0 0 1.8 0 04 0.8 1.2 1.6 2.0 2.4 28 3.2 36 4.0 0 10 20 30 40 50 60 70 80 90 100 -50 -25 0 25 50 75 100 125 150 SWITCH CURRENT (A) DUTY CYCLE (%) TEMPERATURE (C) LT1513* G01 Negative Feedback Input Current vs Temperature 0 = = kK i -10 cc oc Da oO 5 -20 a = Ms oO =< -30 a lu HH lu => -40 & a = 50 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C) 111513 * GN6 Minimum Peak-to-Peak Synchronization Voltage vs Temperature = 3.0 fgyne = 700kHz 6 25 <= 5 Oo = 2.0 S & Lae N 15 S co oc S 1.0 = > wn = 05 = = = 0 -5[0 -25 0 25 50 75 100 125 150 TEMPERATURE (C) L11513 + G04 BATTERY VOLTAGE (V) FEEDBACK INPUT CURRENT (nA) T1513 * G2 LT1543 + G03 Output Charging Characteristics Showing Constant-Current and Constant-Voltage Operation MAXIMUM AVAILABLE | * CHARGING CURRENT WITH 12V INPUT Vin = 12V (A) 8.4V BATTERY IcHRG = 0.5A (B) 8.4V BATTERY IcHRa = 1A (C) 4.2V BATTERY IcHRG = 1.5A (A) (B) 0 0 02 04 06 08 1.0 1.2 14 16 1.8 2.0 CHARGING CURRENT (A) 1513 GN? Feedback Input Current vs Temperature 800 700 Veg = VREF 600 500 400 300 200 100 0 -50 -25 0 25 50 75 TEMPERATURE (C) 100 125 150 (11513 + G5LT1513/LT1513-2 PIN FUNCTIONS Vc (Pin 1): The compensation pin is primarily used for frequency compensation, but it can also be used for soft Starting and current limiting. It is the output of the error amplifier and the input of the current comparator. Peak switch current increases from OA to 3.6A as the Vc voltage varies from 1V to 1.9V. Current out of the Vo pin is about 200uA when the pin is externally clamped below the internal 1.9V clamp level. Loop frequency compensation is performed with a capacitor or series RC network from the Vc pin directly to the ground pin (avoid ground loops). FB (Pin 2): The feedback pin is used for positive output voltage sensing. The R1/R2 voltage divider connected to FB defines Li-lon float voltage at full charge, or acts as a voltage limiter for NiCd or NiMH applications. FB is the inverting input to the voltage error amplifier. Input bias current is typically 300nA, so divider current is normally set to 100uA to swamp out any output voltage errors due to bias current. The noninverting input of this amplifier is tied internally to a 1.245V reference. The grounded end of the output voltage divider should be connected directly to the LT1513 ground pin (avoid ground loops). Irp (Pin 3): The current feedback pin is used to sense charging current. Itis the input to acurrent sense amplifier that controls charging current when the battery voltage is below a programmed limit. During constant-current operation, the LT1513 Ip pin regulates at-100mV. Input resistance of this pin is 5kQ, so filter resistance (R4, Figure 1) should be less than 50Q. The 39Q, 0.22uF filter shown in Figure 1 is used to convert the pulsating current in the sense resistor to a smooth DC current feedback signal. The LT1513-2 lrg pin regulates at OmV to provide programmable current limit. The current through Rd, Figure 5, is balanced by the current through R4, program- ming the maximum voltage across R3. GND (Pin 4): The ground pin is common to both control circuitry and switch current. Vo, FB and S/S signals must be Kelvin and connected as close as possible to this pin. The TAB of the R package should also be connected to the power ground. Vsw (Pin 5): The switch pin is the collector of the power switch, carrying up to 3A of current with fast rise and fall times. Keep the traces on this pin as short as possible to minimize radiation and voltage spikes. In particular, the path in Figure 1 which includes SW to C2, D1, C1 and around to the LT1513 ground pin should be as short as possible to minimize voltage spikes at switch turn-off. S/S (Pin 6): This pin can be used for shutdown and/or synchronization. It is logic level compatible, but can be tied to Vix if desired. It defaults to a high ON state when floated. A logic low state will shut down the charger to a micropower state. Driving the S/S pin with a continuous logic signal of 6(00kHz to 800kHz will synchronize switch- ing frequency to the external signal. Shutdown is avoided in this mode with an internal timer. Vin (Pin 7): The input supply pin should be bypassed with a low ESR capacitor located right next to the IC chip. The grounded end of the capacitor must be connected directly to the ground plane to which the TAB is connected. TAB: The TAB on the surface mount R package is electri- cally connected to the ground pin, but a low inductance connection must be made to both the TAB and the pin for proper circuit operation. See suggested PC layout in Figure 4. LY WhineLT1513/LT1513-2 BLOCK DIAGRAM VIN SW | 7s cH SYNC sone >| Logic >| DRIVER {i swirei I * ko IFBA lFB - 50k* COMP PS a 4 L Vep FA IA S0.040 Ct - > 1.245V Ve vee REF + + *REMOVE ON LT1513-2 vrisis 60 Figure 2 OPERATION The LT1513 is a current mode switcher. This means that switch duty cycle is directly controlled by switch current rather than by output voltage or current. Referring to the Block Diagram, the switch isturned on atthe start of each oscillator cycle. It is turned off? when switch current reaches a predetermined level. Control of output voltage and current is obtained by using the output of a dual feedback voltage sensing error amplifier to set switch current trip level. This technique has the advantage of simplified loop frequency compensation. A low dropout internal regulator provides a 2.3V supply for all internal circuitry on the LT1513. This low dropout design allows input voltage to vary from 2.7V to 25V. A 500kHz oscillator is the basic clock for all internal timing. It turns on the Output switch via the logic and driver circuitry. Special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. This minimizes driver dissipation and provides very rapid turn-off of the switch. A unique error amplifier design has two inverting inputs which allow for sensing both output voltage and current. A 1.245V bandgap reference biases the noninverting input. The first inverting input of the error amplifier is brought out for positive output voltage sensing. The second inverting input is driven by a current amplifier which is sensing Output current via an external current sense resistor. The current amplifier is set to a fixed gain of -12.5 which provides a100mvV current limit sense voltage. The LT1513-2 option removes the feedback resistors around the lrg amplifier and connects its output to the FB signal. This provides a ground referenced current sense voltage suitable for external current programming and makes amplifier input and output available for external loop compensation. The error signal developed at the amplifier output is brought out externally and is used for frequency compen- sation. During normal regulator operation this pin sits at a voltage between 1V (low output current) and 1.9V (high output current). Switch duty cycle goes to zero if the Vo pin is pulled below the Vc pin threshold, placing the LT1513 in an idle mode. 6 LY WeeLT1513/LT1513-2 APPLICATIONS INFORMATION The LT1513 is an IC battery charger chip specifically opti- mized to use the SEPIC converter topology. A complete charger schematic is shown in Figure 1. The SEPIC topology has unique advantages for battery charging. It will operate with input voltages above, equal to or below the battery voltage, has no path for battery discharge when turned off, and eliminates the snubber losses of flyback designs. It also has a current sense point that is ground referred and need not be connected directly to the battery. The two inductors shown are actually just two identical windings on one inductor core, although two separate inductors can be used. Acurrent sense voltage is generated with respect to ground across R3 in Figure 1. The average current through R3 is always identical to the current delivered to the battery. The LT1513 current limit loop will servo the voltage across R3 to -100mV when the battery voltage is below the voltage limit set by the output divider R1/R2. Constant-current charging is therefore set at 100mV/R3. R4 and C4 filter the current signal to deliver a smooth feedback voltage to the Iep pin. R1 and R2 forma dividerfor battery voltage sensing and set the battery float voltage. The suggested value for R2 is 12.4k. R1 is calculated from: {= R2(Vpat 1.245) 1.245 +R2(0.3uA) Vpat = battery float voltage 0.3uA = typical FB pin bias current A value of 12.4k for R2 sets divider current at 100uA. This is a constant drain on the battery when power to the charger is off. If this drain is too high, R2 can be increased to 41.2k, reducing divider current to 30uA. This introduces an addi- tional uncorrectable error to the constant voltage float mode of about +0.5% as calculated by: +0.15A(R1)(R2) 1.245(R1+R2) +0.15y/A = expected variation in FB bias current around the nominal 0.3pA typical value. With R2 = 41.2k and R1 = 228k, (Vpat = 8.2V), the error due to variations in bias current would be+0.42%. A second option is to disconnect the divider when charger poweris off. This can be done with a small NFET as shown in Vpat Error = Figure 3. D2, C6 and R6 form apeak detector to drivethe gate of the FET to about the same as the battery voltage. If power is turned off, the gate will drop to OV and the only drain onthe battery will be the reverse leakage of the catch diode D1. See Diode Selection for a discussion of diode leakage. ADAPTER INPUT j Rt ------- {ie + re <= SN = > bis 4 R$ $5 == dropF $470k S* I ! a. s. Lt SCHEMATIC SIMPLIFIED FOR CLARITY me D2 = 1N914, 1N4148 OR EQUIVALENT Figure 3. Eliminating Divider Current Maximum Input Voltage Maximum input voltage for the LT1513 is partly determined by battery voltage. A SEPIC converter has a maximum switch voltage equal to input voltage plus output voltage. The LT1513 has a maximum input voltage of 30V and a maximum switch voltage of 40V, so this limits maximum input voltage to 30V, or 40V Vpar, whichever is less. Shutdown and Synchronization The dual function S/S pin provides easy shutdown and synchronization. It is logic level compatible and can be pulled high or left floating for normal operation. A logic low on the S/S pin activates shutdown, reducing input supply currentto 12uA. To synchronize switching, drive the S/S pin between 600kHz and 800kHz. Inductor Selection L1Aand L1B are normally justtwo identical windings on one core, although two separate inductors can be used. Atypical value is 10uH, which gives about 0.5A peak-to-peak induc- tor current. Lower values will give higher ripple current, which reduces maximum charging current. 5uH can be used if charging currents are at least 20% lower than the values LY WhineLT1513/LT1513-2 APPLICATIONS INFORMATION shown in the maximum charging current graph. Higher inductance values give slightly higher maximum charging current, but are larger and more expensive. A low loss toroid core such as Kool Mu, Molypermalloy or Metglas is recommended. Series resistance should be less than 0.04Q for each winding. Open core inductors, such as rods or barrels are not recommended because they generate large magnetic fields which may interfere with other electronics close to the charger. Input Capacitor The SEPIC topology has relatively low input ripple current compared to other topologies and higher harmonics are especially low. RMS ripple current in the input capacitor is less than 0.25A with L = 10uH and less than 0.5A with L=5uH. AlowESR 22uF, 25V solid tantalum capacitor (AVX type TPS or Sprague type 593D) is adequate for most applications with the following caveat. Solid tantalum capacitors can be destroyed with a very high turn-on surge current such as would be generated if alow impedance input source were hot switched to the charger input. If this condition can occur, the input capacitor should have the highest possible voltage rating, at least twice the surge input voltage if possible. Consult with the capacitor manufacturer beforeafinal choiceis made. A4.7uF ceramic capacitor such as the one used for the coupling capacitor can also be used. These capacitors do not have a turn-on surge limitation. The input capacitor must be connected directly to the Vj, pin and the ground plane close to the LT1513. Output Capacitor It is assumed as a worst case that all the switching output ripple current from the battery charger could flow in the output capacitor. This is a desirable situation if it is neces- sary to have very low switching ripple current in the battery itself. Ferrite beads or line chokes are often inserted in series with the battery leads to eliminate high frequency currents that could create EMI problems. This forces all the ripple current into the output capacitor. Total RMS current into the capacitor has a maximum value of about 1A, and this is handled with the two paralleled 22uF, 25V capacitors shown Kool Muis a registered trademark of Magnetics, Inc. Metglas is a registered trademark of AlliedSignal Inc. in Figure 1. These are AVX type TPS or Sprague type 593D Surface mount solid tantalum units intended for switching applications. Do not substitute other types without ensuring that they have adequate ripple current ratings. See Input Capacitor section for details of surge limitation on solid tantalum capacitors if the battery may be hot switched to the output of the charger. Coupling Capacitor C2 in Figure 1 is the coupling capacitor that allows a SEPIC converter topology to work with input voltages either higher or lower than the battery voltage. DC bias on the capacitor is equal to input voltage. RMS ripple current in the coupling capacitor has a maximum value of about 1A at full charging current. A conservative formula to calculate this is: | +V 1.1 Icour(aMs) = oe (1.1 is a fudge factor to account for inductor ripple current and other losses) With IcHRG =1.2A, Vin=15V and Vaart =8.2V, Icgup=1.02A. The recommended capacitor is a 4.7uF ceramic type from Marcon or Tokin. These capacitors have extremely low ESR and high ripple current ratings in a small package. Solid tantalum units can be substituted iftheir ripple current rating is adequate, but typical values will increase to 22uF or more to meet the ripple current requirements. Diode Selection The switching diode should be a Schottky type to minimize both forward and reverse recovery losses. Average diode currentis the same as output charging current, so this will be under 2A. A 3Adiode is recommended for most applications, although smaller devices could be used at reduced charging current. Maximum diode reverse voltage will be equal to input voltage plus battery voltage. Diode reverse leakage current will be of some concern during charger shutdown. This leakage current is a direct drain on the battery when the charger is not powered. High 8 L) TECHNOLOGYLT1513/LT1513-2 APPLICATIONS INFORMATION current Schottky diodes have relatively high leakage cur- rents (5A to 500uA) even at room temperature. The latest very-low-forward devices have especially high leakage cur- rents. Ithas been noted that surface mountversions of some Schottky diodes have as much as ten times the leakage of their through-hole counterparts. This may be because alow forward voltage process is used to reduce power dissipation in the surface mount package. In any case, check leakage Specifications carefully before making a final choice for the switching diode. Be aware that diode manufacturers want to specify a maximum leakage current that is ten times higher than the typical leakage. It is very difficult to get them to specify alow leakage current in high volume production. This is an on going problem for all battery charger circuits and most customers have to settle for a diode whose typical leakage is adequate, but theoretically has a worst-case condition of higher than desired battery drain. Thermal Considerations Care should be taken to ensure that worst-case conditions do not cause excessive die temperatures. Typical thermal resistance is 30C/W forthe R package but this number will GROUND PLANE C1,03,C5 AND R3 TIED DIRECTLY TO GROUND PLANE vary depending on the mounting technique (copper area, airflow, etc.). Average supply current (including driver current) is: iy =4mA + (VBat)(IcHRG) 0.024) Vin Switch power dissipation is given by: Pay = loHRG) sw) Vaart + Vin) Vaan (Vin) Rew = Output switch ON resistance Total power dissipation of the die is equal to supply current times supply voltage, plus switch power: PpToTaLy = (lin)(Vin) + Pow For Viy = 10V, VpaT =8.2V, Icurg = 1.2A, Rew = 0.3Q, ly =4mA +24mA =28mA Pow =0.64W Pp = (10)(0.028) + 0.64 =0.92W VBAT 2 WINDING INDUCTOR 171513 *F04 VIN Figure 4. LT1513 Suggested Partial Layout for Critical Thermal and Electrical Paths LY WhineLT1513/LT1513-2 APPLICATIONS INFORMATION Programmed Charging Current LT1513-2 charging current can be programmed with a DC voltage source or equivalent PWM signal, as shown in Figure 5. In constant-current mode, lrg acts as a virtual ground. The Ise voltage across R65 is balanced by the voltage across R4 in the ratio R4/R5. Charging current is given by: (Viget)(R4/R5)leBvos R3 leg input current is small and can normally be ignored, but lcp Offset voltage must be considered if operating over a wide range of program currents. The voltage across R3 at maximum charge current can be increased to reduce offset errors at lower charge currents. In Figure 5, Isey from OV to 5V corresponds to an Icyarge of OA to 1A +37/-62mA. C4 and R4 smooth the switch current wave- form. During constant-current operation, the voltage feed- back network loads the FB pin, whichis held at Vper by the leg amplifier. It is recommended that this load does not ICHARGE = exceed 60uA to maintain a sharp constant voltage to constant current crossover characteristic. IcuarRge can also be controlled by a PWM input. Assuming the signal is aCMO$S rail-to-rail output with a source impedance of less than a few hundred ohms, effective Isez is Veg multiplied by the PWM ratio. Icuarge has good linearity over the entire 0% to 100% range. Voltage Mode Loop Stability The LT1513 operates in constant-voltage mode during the final phase of charging lithium-ion and lead-acid batteries. This feedback loop is stabilized with a series resistor and capacitor on the Vc pin of the chip. Figure 6 shows the simplified model for the voltage loop. The error amplifier is modeled as a transconductance stage with gp =1500umho LiB ISeT C4 Ss R38 O1pF S020 cet 1513 FOS Figure 5 MODULATOR SECTION V1 |p ___4(Vin) ; 9m = V1 ~ Vin + Vpat bo, Vin = DC INPUT VOLTAGE L p Vpar = DC BATTERY VOLTAGE RBAT 0.12 mt I I I | BATTERY _2 t ; 1513 FO6 * FOR 8.4V BATTERY. ADJUST VALUE OF R1 FOR ACTUAL BATTERY VOLTAGE * Rp AND Cp MODEL PHASE DELAY IN THE MODULATOR THIS IS A SIMPLIFIED AC MODEL FOR THE LT1513 IN CONSTANT- VOLTAGE MODE. RESISTOR AND CAPACITOR NUMBERS CORRESPOND TO THOSE USED IN FIGURE 1. Rp AND Cp MODEL THE PHASE DELAY IN THE MODULATOR. C3 1S 3pF FOR A 10uH INDUCTOR. IT SHOULD BE SCALED PROPORTIONALLY FOR OTHER INDUCTOR VALUES (6pF FOR 20,H). THE MODULATOR ISA AS SHOWN, THIS LOOP HAS A UNITY-GAIN FREQUENCY OF ABOUT 250Hz. UNITY-GAIN WILL MOVE OUT TO SEVERAL KILOHERTZ IF BATTERY RESISTANCE INCREASES TO SEVERAL OHMS. R5 IS NOT USED IN ALL APPLICATIONS, BUT IT GIVES BETTER PHASE MARGIN IN CONSTANT-VOLTAGE MODE WITH HIGH BATTERY RESISTANCE. TRANSCONDUCTANCE WHOSE GAIN IS A FUNCTION OF INPUT AND BATTERY VOLTAGE AS SHOWN. Figure 6. Constant-Voltage Small-Signal Model 10LT1513/LT1513-2 APPLICATIONS INFORMATION (from the Electrical Characteristics). Amplifier output resis- tance is modeled with a 330k resistor. The power stage (modulator section) ofthe LT1513is modeled as atranscon- ductance whose value is 4(Viy)/(Viy + Vgar). This is a very simplified model of the actual power stage, butitis sufficient when the unity-gain frequency of the loop is low compared to the switching frequency. The outputfilter capacitor model includes its ESR (Reap). A series resistance (Rgat) is also assigned to the battery model. Analysis of this loop normally shows an extremely stable system for all conditions, even with 0Q for R5. The one condition which can cause reduced phase margin is with a very large battery resistance (>5Q), or with the battery replaced with a resistive load. The addition of R5 gives good phase margin even under these unusual conditions. R5 should not be increased above 330Q without checking for two possible problems. The first is instability in the constant current region (see Constant-Current Mode Loop Stability), andthe second is subharmonic switching where switch duty cycle varies from cycle to cycle. This duty cycle instability is caused by excess switching frequency ripple voltage on the Vc pin. Normally this ripple is very low because of the filtering effect of C5, but large values of Rd can allow high ripple onthe Vc pin. Normal loop analysis does not show this problem, and indeed small signal loop stability can be excellent even in the presence of subharmonic switching. The primary issue with subharmonics is the presence of EMI at frequencies below 500kHz. Constant-Current Mode Loop Stability The LT1513 is normally very stable when operating in con- stant-current mode (see Figure 7), but there are certain con- ditions which may create instabilities. The combination of highervalue current sense resistors (low programmed charg- ing current), higher input voltages, and the addition ofaloop compensation resistor (R5) on the Vc pin may create an un- stable current mode loop. (A resistor is sometimes added in series with C5 to improve loop phase margin when the loop is operating in voltage mode.) Instability results because loop gain is too high in the 50kHz to 150kHz region where excess phase occurs in the current sensing amplifier and the modulator. The Irga amplifier (gain of -12.5) has a pole at approximately 150kHz. The modulator section con- sisting of the current comparator, the power switch and the magnetics, has a pole at approximately 50kHz when the coupled inductorvalueis 1 0uH. Higher inductance will reduce the pole frequency proportionally. The design procedure pre- sented here is to roll off the loop to unity-gain at a frequency of 25kHz or lower to avoid these excess phase regions. MODULATOR SECTION 4(V1) Vin) Vin + VBaT La 10pF VOLTAGE GAIN = 12 rT A THE CURRENT AMPLIFIER HAS A FIXED VOLTAGE GAIN OF 12. ITS PHASE DELAY IS MODELED WITH Ra AND Ca. THE ERROR AMPLIFIER HAS A TRANSCONDUCTANCE OF 1500,mho AND AN INTERNAL OUTPUT SHUNT RESISTANCE OF 330k. AS SHOWN, THIS LOOP HAS A UNITY-GAIN FREQUENCY OF ABOUT 27kHz. RS 1S NOT USED IN ALL APPLICATIONS, BUT IT GIVES BETTER PHASE MARGIN IN CONSTANT VOLTAGE MODE. THIS IS A SIMPLIFIED AC MODEL FOR THE LT1513 IN CONSTANT-CURRENT MODE. RESISTOR AND CAPACITOR NUMBERS CORRESPOND TO THOSE USED IN FIGURE 1. Rp AND Cp MODEL THE PHASE DELAY IN THE PowerPath. C3 IS 3pF FOR A 10uH INDUCTOR. IT SHOULD BE SCALED PROPORTIONALLY FOR OTHER INDUCTOR VALUES (6pF FOR 20H). THE PowerPath IS A TRANSCONDUCTANCE WHOSE GAIN IS A FUNCTION OF INPUT AND BATTERY VOLTAGE AS SHOWN. 1543 FO? Figure 7. Constant-Current Small-Signal Model LY Whine 11LT1513/LT1513-2 APPLICATIONS INFORMATION The suggested way to control unity loop frequency is to increase the filter time constant on the leg pin (R4/C4 in Figures 1 and 7). The filter resistor cannot be arbitrarily increased because high values will affect charging current accuracy. Charging current will increase by 1% for each 40Q increase in R4. There is no inherent limitation on the value of C4, but if this capacitor is ceramic, it should be an X7R type to maintain its value over temperature. X7R dielectric requires a larger footprint. The formula for calculating the minimum value for the filter capacitor C4 is: og PNAC yye12)(1500n)(R5) 2n(f)(R4)(Viy + Vat) Vin = Highest input voltage 1500u= Transconductance of error amplifier (EA)f = Desired unity-gain frequency Vpat = Battery voltage For example, assume Vin max) = 15V, R3 = 0.4Q (charging current set to 0.25A), R4 = 240, R5 = 330Q and Vaart = 8V, og 2-4(4V15)112)(0.0016)(330) _ 6.3(25000)(39)(15+ 8) The value for C4 could be reduced toa more manageable size by increasing R4 to 75Q and reducing Rd to 300Q, yielding 0.47uF for C4. The 2% increase in charging current can be ignored or factored into the value for R3. More Help Linear Technology Field Application Engineers have a CAD Spreadsheet program for detailed calculations of circuit operating conditions. In addition, our Applications Depart- ment is always ready to lend a helping hand. The LT1371 data sheet may also be helpful. The LT1513 is identical except for the current amplifier circuitry. 12LT1513/LT1513-2 TYPICAL APPLICATIONS Lithium-lon Battery Charger with Switchable Charge Current Many battery chemistries require several constant-current settings during the charging cycle. The circuit shown in Figure 8 uses the LT 1513-2 to provide switchable 1.35A and 0.13A constant-current modes. The circuit is based on a Standard SEPIC battery charger circuit set to a single lithium-ion cell charge voltage of 4.1V. The LT1513-2 has Ire referenced to ground allowing a simple resistor network to set the charging current values. In constant-current mode, the lrg error amplifier drives the FB pin, increasing charging current, until Ipq is balanced by Ips. (Ik5 )(R4) IFavos R3 There are several ways to control Ips including DAC, PWM or resistor network as shown here. If the lithium cell requires precharging, Q1 is turned on, setting a constant current of 0.13A. When charge voltage is reached, Q1 is turned off, programming the full charge current of 1.35A. As the cell voltage approaches 4.1V, the voltage sensing network (R1, R2) starts driving the Veg pin, changing the LT1513-2 to constant-voltage mode. As charging currentfalls, the output remains in constant-voltage mode for the remainder of the charging cycle. When charging is complete, the LT1513-2 can be shut down with the S/S pin. ICHARGE = HA C2 D1 CTX10-4 4.7uF MBRS330T3 Liclon VIN | tc RECHARGABLE I J CELL en LiB 787k CHARGE F 0.5% e SHUTDOWN 4 ay +] Ct 22uF x6 7 ai 10k 47k x2 AAA WVV t0 Q 2 R3 eRe PRECHARGE 2 D 34k \ | 0 O.2QuF F0.252 SMe CHARGE GND 1543 FO8 Figure 8. Lithium-lon Battery Charger LY Whine 13LT1513/LT1513-2 TYPICAL APPLICATIONS This Cold Cathode Fluorescent Lamp driver uses a Royer class self-oscillating sine wave converter to driver a high voltage lamp with an AC waveform. CCFL Royer converters have significantly degraded efficiency if they must operate at low input voltages, and this circuit was designed to handle input voltages as low as 2.7V. Therefore, the LT1513 is connected to generate a negative current through L2 that allows the Royer to operate as if it were connected to a constant higher voltage input. C1 2.7V 47uF TO 20V The Royer output winding and the bulb are allowed to float in this circuit. This can yield significantly higher efficiency in Situations where the stray bulb capacitance to surrounding enclosure is high. To regulate bulb current in Figure 9, Royer input current is sensed with R2 and filtered with R3 and C6. This negative feedback signal is applied to the Irp pin of the LT1513. Formore information on this circuit contactthe LTC Applications Department and see Design Note 133. Consid- erable written application literature on Royer CCFL circuits is also available from other LTC Application and Design Notes. 4.7 uF C2 CERAMIC 2 Q2 20uH m 4 5 LYON LAMP CURRENT 10nF R7 2k C5 To DI 8A 5.6mA NEGATIVE VOLTAGE IS GENERATED HERE PWM DIMMING (=1kHz) C2: TOKIN MULTILAYER CERAMIC C3: MUST BE A LOW LOSS CAPACITOR, WIMA MKP-20 OR EQUIVALENT L1, L2: COILTRONICS CTX20-4 (MUST BE SEPARATE INDUCTORS) Qi, Q2: ZETEX ZTX849 OR FZT849 T1: COILTRONICS CTX110605 (67:1) - 1513 FOS Figure 9. CCFL Power Supply for Floaing Lamp Configuration Operates on 2.7V 14LT1513/LT1513-2 PACKAGE DESCRIPTION 0.256 0.080 (1.524) R Package 7-Lead Plastic DD Pak (LTC DWG # 05-08-1462) 0.060 (1.524) TYP ! 0.390 - 0.415 (9.906 10.541) \ | tf 0.330 0.370 (8.382 9.398) __ EBEEEH 0.040 0.060 7800 0.143 +0 OTe | | maa! le pier 09 (1.016 1.524) ee 0.026 0. 036 BOTTOM VIEW OF DD PAK . 632+? ad (0.660 0.914) HATCHED AREA IS SOLDER PLATED COPPER HEAT SINK T7 Package 7-Lead Plastic TO-220 (Standard) (LTC DWG # 05-08-1422) 0.147 - 0.155 0.390-0.415 | (3.734 3.937) (9.906 10.541) DIA + 0.230 - 0.270 (5.842 6.858) 0.460 - 0.500 (11.684 12.700) 0.570 - 0.620 (14.478 15.748) 0.330 - 0.370 (8.382 9.398) 0.040 0.060 J |< | le (1.016 1.524) 0.026 0.036 (0.660 0.914) 0.260 0.320 (3.860 5.130) 1 (6.604 8.128) = ' 0.165 0.180 0.165 0.180 0.059 (1.499) TYP t 0.013 - 0,023 (0.330 0.584) (4.191 = 4572) ; | 0.620 TYP 0.700 0.728 (17.780 18.491) Y 0.152 0.202 j 15.75) 0.135 0.165 Dimensions in inches (millimeters) unless otherwise noted. | (4191-4572) [Se 0.045 - 0.055 =" Typ (1.143 1.397) +0.008 0.004" 004 +0.203 [o-io2" Oi] 0,095 - 0.115 peas 2.921) Lt | 0.050 + 0.012 (1.270 + 0.305) R(DD7} 0396 0.045 0.055 (1.143 1.397) 0.095 0.115 | (2.413-2.921) 0.013 0.023 (0.330 0.584) 0.155 - 0.195 (3.429 4.191) (3.937 4.953) T? (10-221) (FORMED) 1197, LS. LINGAR Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen- tation that the interconnection ofits circuits as described herein will not infringe on existing patent rights. 15LT1513/LT1513-2 PART NUMBER} DESCRIPTION COMMENTS LT1239 Backup Battery Management System Charges Backup Battery and Regulates Backup Battery Output when Main Battery Removed LTC1325 Microprocessor Controlled Battery Management System | Can Charge, Discharge and Gas Gauge NiCd, NiMH and Pb-Acid Batteries with Software Charging Profiles LT1510 1.5A Constant-Current/Constant-Voltage Battery Charger | Step-Down Charger for Li-lon, NiCd and NiMH LT1511 3.0A Constant-Current/Constant-Voltage Battery Charger | Step-Down Charger that Allows Charging During Computer Operation and with Input Current Limiting Prevents Wall-Adapter Overload LT1512 SEPIC Constant-Current/Constant-Voltage Battery Charger) Step-Up/Step-Down Charger for Up to 1A Current Linear Technology Corporation 1 6 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977 www.linear-tech.com 1513fa LT/TP 0198 REV A 4K PRINTED IN THE USA LY Wie LINEAR TECHNOLOGY CORPORATION 1996