Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw1
ANPEC reserves the right to make changes to improve reliability or manufacturability without notice, and
advise customers to obtain the latest version of relevant information to verify before placing orders.
5V to 12V Synchronous Buck Controller
Features
• Wide Operation Supply Voltage from 5V to 12V
• Power-On-Reset Monitoring on VCC
• Excellent Reference Voltage Regulations
- 0.8V Internal Reference
- ±1% Over Temperature Range
• Integrated Soft-Start
• Automatic PSM/PWM Modes
• Voltage Mode PWM Operation with 90% (Max.)
Duty Cycle
• Under-Voltage Protection
• Adjustable Over-Current Protection Threshold
- Sensing the RDS(ON) of Low-Side MOSFET
• Over-Voltage Protection
• Under-Voltage Protection
• Simple SOP-8 Package
• Lead Free and Green Devices Available
(RoHS Compliant)
Applications
General Description
• Graphic Cards
• DSL, Switch HUB
• Wireless Lan
• Notebook Computer
• Mother Board
• LCD Monitor/TV
The APW7165 is a voltage mode, fixed 300kHz-switching
frequency, and synchronous buck controller. The
APW7165 allows wide input voltage that is either a single
5~12V or two supply voltages for various applications. A
power-on-reset (POR) circuit monitors the VCC supply
voltage to prevent wrong logic controls. A built-in digital
soft-start circuit prevents the output voltages from over-
shoot as well as limits the input current. An internal 0.8V
temperature-compensated reference voltage with high
accuracy is designed to meet the requirement of low out-
put voltage applications. The APW7165 provides excel-
lent output voltage regulations against load current
variation.
The controller’s over-current protection monitors the out-
put current by using the voltage drop across the RDS(ON) of
low-side MOSFET, eliminating the need for a current sens-
ing resistor that features high efficiency and low cost.
The APW7165 also integrates over-voltage protection
(OVP) and under-voltage protection circuit which moni-
tors the FB voltage to prevent the PWM output from over
and under voltage.
The APW7165 is available in a simple SOP-8 package.
Pin Configuration
Simplified Application Circuit
SOP-8
(Top View)
BOOT 1
LGATE 4
GND 3
UGATE 2 6 FB
7 COMP
8 PHASE
5 VCC
PHASE
FB GND
VCC
LGATE
COMP
APW7165 VIN
VOUT
UGATE
BOOT
5
7
6
3
4
8
2
1
ONOFF
VVCC
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw2
Ordering and Marking Information
Note: ANPEC lead-free products contain molding compounds/die attach materials and 100% matte tin plate termination finish; which
are fully compliant with RoHS. ANPEC lead-free products meet or exceed the lead-free requirements of IPC/JEDEC J-STD-020D for
MSL classification at lead-free peak reflow temperature. ANPEC defines “Green” to mean lead-free (RoHS compliant) and halogen
free (Br or Cl does not exceed 900ppm by weight in homogeneous material and total of Br and Cl does not exceed 1500ppm by
weight).
Absolute Maximum Ratings (Note 1)
Symbol
Parameter Rating Unit
VVCC VCC Supply Voltage (VCC to GND) -0.3 ~ 16 V
VBOOT BOOT to PHASE Voltage -0.3 ~ 16 V
> 400ns -0.3 ~ VBOOT+0.3 V
VUGATE UGATE to PHASE Voltage < 400ns -5 ~ VBOOT+5 V
> 400ns -0.3 ~ VVCC+0.3 V
VLGATE LGATE to GND Voltage < 400ns -5 ~ VVCC+5 V
> 200ns -0.3 ~ 16 V
VPHASE PHASE to GND Voltage < 200ns -10 ~ 30 V
FB and COMP to GND (< VVCC + 0.3V) -0.3 ~ 7 V
TJ Maximum Junction Temperature 150 °C
TSTG Storage Temperature -65 ~ 150 °C
TSDR Maximum Lead Soldering Temperature, 10 Seconds 260 °C
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
Symbol
Parameter Typical Value Unit
θJA Thermal Resistance - Junction to Ambient (Note 2) SOP-8
150 °C/W
θJC Thermal Resistance - Junction to Case SOP-8
28 °C/W
Thermal Characteristics
Note 2 : θJA is measured with the component mounted on a high effective thermal conductivity test board in free air.
APW7165
Handling Code
Temperature Range
Package Code
Assembly Material
APW7165 K: XXXXX - Date Code
Package Code
K : SOP-8
Operating Ambient Temperature Range
E : -20 to 70oC
Handling Code
TR : Tape & Reel
Assembly Material
G : Halogen and Lead Free Device
APW7165
XXXXX
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw3
Recommended Operating Conditions (Note 3)
Symbol
Parameter Range Unit
VVCC VCC Supply Voltage (VCC to GND) 4.5 ~ 13.2 V
VOUT Converter Output Voltage 0.9 ~ 5 V
VIN Converter Input Voltage 2.9 ~ VVCC V
IOUT Converter Output Current 0 ~ 20 A
TA Ambient Temperature -20 ~ 70 °C
TJ Junction Temperature -20 ~ 125 °C
Electrical Characteristics
APW7165
Symbol
Parameter Test Conditions Min.
Typ. Max.
Unit
INPUT SUPPLY VOLTAGE AND CURRENT
VCC Supply Current (Shutdown Mode)
UGATE and LGATE open;
COMP=GND - 4 6
IVCC VCC Supply Current UGATE and LGATE open - 16 24 mA
POWER-ON-RESET(POR)
Rising VCC POR Threshold 3.7 4.1 4.4 V
VCC POR Hysteresis 0.3 0.45
0.6 V
OSCILLATOR
FOSC Oscillator Frequency 270 300 330 kHz
∆VOSC Oscillator Sawtooth Amplitude (Note 4) - 1.5 - V
DMAX Maximum Duty Cycle 85 - 90 %
ERROR AMPLIFIER
VREF Reference Voltage TA = -20 ~ 70°C 0.792
0.8 0.808
V
Converter Load Regulation (Note 4) IOUT = 2 ~ 12A - - 0.2 %
gm Transconductance - 667 - µA/V
FB Input Leakage Current VFB = 0.8V - 0.1 1 µA
COMP High Voltage RL = 10kΩ to GND - 2.5 -
COMP Low Voltage RL = 10kΩ to GND - 1 - V
Maximum COMP Source Current VCOMP = 2V - 200 -
Maximum COMP Sink Current VCOMP = 2V - 200 - µA
GATE DRIVERS
High-Side Gate Driver Source Current
VBOOT= 12V, VUGATE-PHASE = 2V - 1.8 - A
High-Side Gate Driver Sink Impedance
BOOT=12V, IUGATE = 0.1A - 2.3 - Ω
Low-Side Gate Driver Source Current VVCC = 12V, VLGATE = 2V - 1.8 - A
Low-Side Gate Driver Sink Impedance
VVCC =12V, IUGATE = 0.1A - 1.3 - Ω
TD Dead-Time (Note 4) - 30 - ns
Refer to the typical application circuit. These specifications apply over VVCC = 12V, TA = -20°C to 70°C, unless otherwise
noted. Typical values are at TA = 25°C.
Note 3 : Refer to the application circuit for further information.
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw4
Electrical Characteristics (Cont.)
Refer to the typical application circuit. These specifications apply over VVCC = 12V, TA = -20°C to 70°C, unless otherwise
noted. Typical values are at TA = 25°C.
APW7165
Symbol
Parameter Test Conditions Min.
Typ. Max.
Unit
PROTECTIONS
VFB_UV FB Under-Voltage Protection Trip Point
Percentage of VREF 45 50 55 %
VFB_OV FB Over-Voltage Protection Trip Point
VFB rising 115 120 125 %
FB Over-Voltage Protection Hysteresis
- 5 - %
VOCP_MAX
Built-in Maximum OCP Voltage 300 - - mV
IOCSET OCSET Current Source 19.5 21.5
23.5 µA
SOFT-START
VDISABLE Shutdown Threshold of VCOMP - - 0.6 V
TSS Internal Soft-Start Interval (Note 4) - 1.7 - ms
Note 4 : Guaranteed by design, not production tested.
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw5
Operating Waveforms
Refer to the typical application circuit. The test condition is VIN=12V, TA= 25oC unless otherwise specified.
Enable
CH1: VCOMP, 1V/Div
CH2: VOUT, 500mV/Div
CH3: VPHASE, 10V/Div
Time: 500µs/Div
VPHASE
1
3
2 VOUT
VCOMP
Shutdown
CH1: VCOMP, 1V/Div
CH2: VOUT, 500mV/Div
CH3: VPHASE, 10V/Div
Time: 1ms/Div
1
3
2
RLOAD
=10Ω
VCOMP
VOUT
VPHASE
Power OnPower Off
CH1: VIN, 5V/Div
CH2: VOUT, 500mV/Div
CH3: VUGATE, 10V/Div
Time: 1ms/Div
1
3
2
VIN
VOUT
VUGATE
CH1: VIN, 5V/Div
CH2: VOUT, 500mV/Div
CH3: VUGATE, 10V/Div
Time: 200ms/Div
1
3
2
VIN
VOUT
VUGATE
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw6
Operating Waveforms (Cont.)
Refer to the typical application circuit. The test condition is VIN=12V, TA= 25oC unless otherwise specified.
UGATE Falling UGATE Rising
1,2
3
VUGATE2
VLGATE2
VPHASE2
CH1: VUGATE, 20V/Div
CH2: VLGATE, 10V/Div
CH3: VPHASE, 10V/Div
Time: 20ns/Div
CH1: VUGATE, 20V/Div
CH2: VLGATE, 10V/Div
CH3: VPHASE, 10V/Div
Time: 20ns/Div
2
3
VLGATE
1VUGATE
VPHASE
PSM to PWMPWM to PSM
CH1: VOUT, 500mV/Div
Time: 500µs/Div
CH2: VPHASE, 10V/Div
CH3: IL, 2A/Div
CH1: VOUT, 1V/Div
Time: 500µs/Div
CH2: VPHASE, 10V/Div
CH3: IL, 2A/Div
2
3
1
VOUT
IOUT=10mA to 2A
VPHASE
IL
2
3
1
IOUT= 2A to 10mA
VOUT
VPHASE
IL
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw7
Operating Waveforms (Cont.)
Refer to the typical application circuit. The test condition is VIN=12V, TA= 25oC unless otherwise specified.
Load Transient Response
CH1: VOUT, 50mV/Div
Time: 200µs/Div
CH2: IOUT, 5A/Div
Over Current Protection
3
1
2
VOUT
VPHASE
IL
ROCSET
=5.1kΩ,RDS(low-side)=10mΩ
CH1: VOUT, 500mV/Div
Time: 5µs/Div
CH2: VPHASE, 20A/Div
CH3: IL, 10A/Div
CH1: VOUT, 500mV/Div
Time: 50µs/Div
CH2: VPHASE, 20A/Div
CH3: IL, 10A/Div
3
1
2
VOUT
VPHASE
IL
Over Current Protection
VOUT
IOUT Slew rate=10A /µs
IOUT=10mA->10A->10mA
IOUT
1
2
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw8
Pin Description
PIN
NO. NAME FUNCTION
1 BOOT This pin provides the bootstrap voltage to the high-side gate driver for driving the N-channel MOSFET.
An external capacitor from PHASE to BOOT, an internal diode, and the power supply voltage VCC,
generates the bootstrap voltage for the high-side gate driver (UGATE).
2 UGATE High-side Gate Driver Output. This pin is the gate driver for high-side MOSFET.
3 GND Signal and Power ground. Connecting this pin to system ground.
4 LGATE Low-side Gate Driver Output and Over-Current Setting Input. This pin is the gate driver for low-side
MOSFET. It also used to set the maximum inductor current. Refer to the section in “Function
Description” for detail.
5 VCC Power Supply Input for Control Circuitry. Connect a nominal 5V to 12V power supply voltage to this
pin. A power-on-reset function monitors the input voltage at this pin. It is recommended that a
decoupling capacitor (1 to 10µF) be connected to GND for noise decoupling.
6 FB Feedback Input of Converter. The converter senses feedback voltage via FB and regulates the FB
voltage at 0.8V. Connecting FB with a resistor-divider from the output sets the output voltage of the
converter.
7 COMP
This is a multiplexed pin. During the soft-start and normal converter operation, this pin represents the
output of the error amplifier. It is used to compensate the regulation control loop in combination with
the FB pin.
Pulling COMP low (VDISABLE = 0.6V typical) will shut down the controller. When the pull-down device is
released, the COMP pin will start to rise. When the COMP pin rises above the VDISABLE trip point, the
APW7165 will begin a new initialization and soft-start cycle.
8 PHASE This pin is the return path for the high-side gate driver. Connecting this pin to the high-side MOSFET
source and connecting a capacitor to BOOT for the bootstrap voltage. This pin is also used to monitor
the voltage drop across the low-side MOSFET for over-current protection.
Typical Application Circuit
PHASE
FB GND
VCC
LGATE
COMP
APW7165 CIN2
470µF x 2
CIN1
1µF
L1
1µH
VIN
VOUT
UGATE
COUT
470µF x 2
VCC Supply
(5~12V)
BOOT
R1
1kΩ
R2
2kΩ
5
7
6
3
4
8
2
1
R5
2R2
C5
1µF
C1
15nF
R3
15kΩ
C2
15pF
Q1
APM2510
Q2
APM2556
ON OFF
C4
0.1µF
ROCSET
Q3
2N7002
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw9
Block Diagram
Power-On-Reset
Sample
and
Hold
To
LGATE
VROCSET
Soft-Start
and
Fault Logic
Gate
Control
Sense Low Side
Error Amplifier
VREF Oscillator
PWM
Comparator
UVP Comparator
Regulator
3V
VREF
(0.8V typical)
IZCMP
PHASE
UGATE
BOOT
COMP GNDFB
VCC
Soft-Start
Inhibit
IOCSET
(21.5µA typical)
LGATE
OVP Comparator
1.2
1/2
0.6V Disable
VCC
0.8V
2xVROCSET
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw10
Function Description
Power-On-Reset (POR)
Soft-Start
( )
512x
F1
ttTOSC
23SS =−=
Over-Current Protection
The Power-On-Reset (POR) function of APW7165 con-
tinually monitors the input supply voltage (VCC) and en-
sures that the IC has sufficient supply voltage and can
work well. The POR function initiates a soft-start process
while the VCC voltage exceeds the POR threshold; the
POR function also inhibits the operations of the IC while
the VCC voltage falls below the POR threshold.
The APW7165 builds in a 40-steps digital soft-start to
control the output voltage rise as well as limit the current
surge at the start-up. During the soft-start, the internal
step voltage connected to the one of the positive inputs of
the error amplifier replaces the reference voltage (0.8V
typical) until the step voltage reaches the reference
voltage. The digital soft-start circuit interval (shown as
figure 1) depends on the switching frequency.
The over-current function protects the switching converter
against over-current or short-circuit conditions. The con-
troller senses the inductor current by detecting the drain-
to-source voltage which is the product of the inductor’s
current and the on-resistance of the low-side MOSFET
during it’s on-state.
A resistor (ROCSET), connected from the LGATE to the GND,
programs the over-current trip level. Before the IC ini-
tiates a soft-start process, an internal current source, IOCSET
(21.5µA typical), flowing through the ROCSET develops a
voltage (VROCSET) across the ROCSET. During the normal
operation, the device holds VROCSET and stops the current
source, IOCSET. When the voltage across the low-side
MOSFET exceeds the double VROCSET (2 x VROCSET), the IC
shuts off the converter and then initiates a new soft-start
process. After 2 over-current events are counted, the de-
vice is shut down and all the gate drivers (UGATE, LGATE,
and DRIVE) are off. Both the output of the PWM converter
and linear controller are latched to be floating.
The APW7165 has an internal OCP voltage, VOCP_MAX, and
the value is 0.3V minimum. When the ROCSET x IOCSET ex-
ceeds 0.3V or the ROCSET is floating or not connected, the
VROCSET will be the default value 0.3V. The over current
threshold would be 0.7V across low-side MOSFET. The
threshold of the valley inductor current-limit is therefore
given by:
)sidelow(RRI2
I)ON(DS
OCSETOCSET
LIMIT −
××
=
For the over-current is never occurred in the normal oper-
ating load range; the variation of all parameters in the
above equation should be considered:
- The RDS(ON) of low-side MOSFET is varied by tempera-
ture and gates to source voltage. Users should deter-
mine the maximum RDS(ON) by using the manufacturer’s
datasheet.
- The minimum IOCSET (19.5µA) and minimum ROCSET
should be used in the above equation.
- Note that the ILIMIT is the current flow through the low-
side MOSFET; ILIMIT must be greater than valley inductor
current which is output current minus the half of induc-
tor ripple current.
2I
II )MAX(OUTLIMIT ∆
−>
Where ΔI = output inductor ripple current
- The overshoot and transient peak current also should
be considered.
Voltage(V)
Time
VVCC
VOUT
POR
t2
t0t3
t1
OCSET count completed
OCSET count start
(OCSET duration, t2 - t1, less than 0.9ms)
Figure 1. Soft-Start Interval
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw11
Under-Voltage Protection
Over-Voltage Protection (OVP)
Shutdown and Enable
Pulse Skipping Mode (PSM)
Adaptive Shoot-Through Protection
The under-voltage function monitors the voltage on FB
(VFB) by Under-Voltage (UV) comparator to protect the PWM
converter against short-circuit conditions. When the VFB
falls below the falling UVP threshold (50% VREF), a fault
signal is internally generated and the device turns off
high-side and low-side MOSFETs. The converter is shut-
down and the output is latched to be floating.
The over-voltage protection monitors the FB voltage to
prevent the output from over-voltage condition. When the
output voltage rises above 120% of the nominal output
voltage, the APW7165 turns off the high-side MOSFET
and turns on the low-side MOSFET until the output volt-
age falls below the falling OVP threshold, regulating the
output voltage around the OVP threshold.
The APW7165 can be shut down or enabled by pulling
low the voltage on COMP. The COMP is a dual-function
pin. During normal operation, this pin represents the
output of the error amplifier. It is used to compensate the
regulation control loop in combination with the FB pin.
Pulling the COMP low (VDISABLE = 0.6V typical) places the
controller into shutdown mode which UGATE and LGATE
are pulled to PHASE and GND respectively.
When the pull-down device is released, the COMP volt-
age will start to rise. When the COMP voltage rises above
the VDISABLE threshold, the APW7165 will begin a new ini-
tialization and soft-start process.
At light loads, the inductor current may reach zero or re-
verse on each pulse. The low-side MOSFET is turned off
by the current reversal comparator, IZCMP, to block the
negative inductor current. In this condition, the converter
enters discontinuous current mode operation.
At very light loads, the APW7165 will automatically skip
pulses in pulse skipping mode operation to reduce
switching losses as well as maintain output regulation
for efficient applications.
The gate drivers incorporate an adaptive shoot-through
protection to prevent high-side and low-side MOSFETs
from conducting simultaneously and shorting the input
supply. This is accomplished by ensuring the falling gate
has turned off one MOSFET before the other is allowed to
rise.
During turn-off of the low-side MOSFET, the LGATE volt-
age is monitored until it is below 1.5V threshold, at which
time the UGATE is released to rise after a constant delay.
During turn-off of the high-side OCSFET, the UGATE-to-
PHASE voltage is also monitored until it is below 1.5V
threshold, at which time the LGATE is released to rise
after a constant delay.
Function Description (Cont.)
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw12
Application Information
Output Capacitor Selection
The selection of COUT is determined by the required effec-
tive series resistance (ESR) and voltage rating rather than
the actual capacitance requirement. Therefore, selecting
high performance low ESR capacitors is intended for
switching regulator applications. In some applications,
multiple capacitors have to be paralleled to achieve the
desired ESR value. If tantalum capacitors are used, make
sure they are surge tested by the manufactures. If in doubt,
consult the capacitors manufacturer.
Input Capacitor Selection
The input capacitor is chosen based on the voltage rating
and the RMS current rating. For reliable operation, select
the capacitor voltage rating to be at least 1.3 times higher
than the maximum input voltage. The maximum RMS
current rating requirement is approximately IOUT/2 where
IOUT is the load current. During power up, the input capaci-
tors have to handle large amount of surge current. If tanta-
lum capacitors are used, make sure they are surge tested
by the manufactures. If in doubt, consult the capacitors
manufacturer.
For high frequency decoupling, a ceramic capacitor be-
tween 0.1µF to 1µF can connect between VCC and ground
pin.
Inductor Selection
The inductance of the inductor is determined by the out-
put voltage requirement. The larger the inductance, the
lower the inductor’s current ripple. This will translate into
lower output ripple voltage. The ripple current and ripple
voltage can be approximated by:
where Fs is the switching frequency of the regulator.
∆VOUT = IRIPPLE x ESR
A tradeoff exists between the inductor’s ripple current and
the regulator load transient response time. A smaller in-
ductor will give the regulator a faster load transient re-
sponse at the expense of higher ripple current and vice
versa. The maximum ripple current occurs at the maxi-
mum input voltage. A good starting point is to choose the
ripple current to be approximately 30% of the maximum
output current.
Once the inductance value has been chosen, selecting
an inductor is capable of carrying the required peak cur-
rent without going into saturation. In some types of
inductors, especially core that is make of ferrite, the ripple
current will increase abruptly when it saturates. This will
result in a larger output ripple voltage.
Compensation
The output LC filter of a step down converter introduces a
double pole, which contributes with –40dB/decade gain
slope and 180 degrees phase shift in the control loop. A
compensation network between COMP pin and ground
should be added. The simplest loop compensation net-
work is shown in Figure 5.
The output LC filter consists of the output inductor and
output capacitors. The transfer function of the LC filter is
given by:
The poles and zero of this transfer function are:
The FLC is the double poles of the LC filter, and FESR is
the zero introduced by the ESR of the output capacitor.
Output Voltage Selection
The output voltage can be programmed with a resistive
divider. Use 1% or better resistors for the resistive divider
is recommended. The FB pin is the inverter input of the
error amplifier, and the reference voltage is 0.8V. The
output voltage is determined by:
+×= 2
1
OUT R
R
10.8V
Where R1 is the resistor connected from VOUT to FB and
R2 is the resistor connected from FB to the GND.
IN
OUT
SW
OUTIN
RIPPLE V
V
LFVV
I×
×
−
=
1CESRsCLsCESRs1
= GAIN OUTOUT
2OUT
LC +××+××
OUTCL××π×21
= FLC
OUTCESR××π×21
= FESR
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw13
Application Information (Cont.)
Figure 3. The LC Filter Gain & Frequency
The PWM modulator is shown in Figure 4. The input is
the output of the error amplifier and the output is the PHASE
node. The transfer function of the PWM modulator is given
by:
Figure 4. The PWM Modulator
The compensation circuit is shown in Figure 5. R3 and
C1 introduce a zero and C2 introduces a pole to reduce
the switching noise. The transfer function of error ampli-
fier is given by:
The pole and zero of the compensation network are:
Compensation (Cont.)
Figure 5. Compensation Network
The closed loop gain of the converter can be written as:
Figure 6 shows the converter gain and the following guide-
lines will help to design the compensation network.
1.Select the desired zero crossover frequency FO:
(1/5 ~ 1/10) x FSW >FO>FZ
Use the following equation to calculate R3:
FESR
FLC
Frequency
-40dB/dec
-20dB/dec
Gain
VOSC
PWM
Comparator
Driver
Driver
Output of
Error
Amplifier
VIN
PHASE
gm
F
R2
R2R1
2
F
F
V
V
R3 O
LC
ESR
IN
OSC ×
××
∆
=
Figure 2. The Output LC Filter
L
COUT
ESR
Output
PHASE
OSC
PWM V
= GAIN ∆IN
V
+×=× sC2
1
//
sC1
1
R3gmgm OAMP Z = GAIN
C2
C2C1R3 C2C1
ss
C1R31
s
gm ×
×× +
+×
×
+
×=
C2C1 C2C1
R32
1
+
×
××π×
=
P
F
C1R321××π×
=
Z
F
C2
VOUT
R2
R1
R3
Error
Amplifier
VREF
C1
COMP
FB -
+
AMPPWMLG GAINGAINGAIN ×
+
×× R2R1
R2
Where:
gm = 667µA/V
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw14
Application Information (Cont.)
Compensation (Cont.)
Figure 6. Converter Gain & Frequency
MOSFET Selection
The selection of the N-channel power MOSFETs is deter-
mined by the RDS(ON), reverse transfer capacitance (CRSS),
and maximum output current requirement.The losses in
the MOSFETs have two components: conduction loss and
transition loss. For the upper and lower MOSFET, the
losses are approximately given by the following equations:
PUPPER = IOUT2 (1+ TC)(RDS(ON))D + (0.5)(Iout)(VIN)(tsw)FSW
PLOWER = IOUT2 (1+ TC)(RDS(ON))(1-D)
where IOUT is the load current
TC is the temperature dependency of RDS(ON)
FSW is the switching frequency
tsw is the switching interval
D is the duty cycle
Note that both MOSFETs have conduction losses while
the upper MOSFET include an additional transition loss.
The switching internal, tsw, is the function of the reverse
transfer capacitance CRSS. Figure 7 illustrates the switch-
ing waveform internal of the MOSFET.
The (1+TC) term factors in the temperature dependency
of the RDS(ON) and can be extracted from the “RDS(ON) vs.
Temperature” curve of the power MOSFET.
1FC1R3 C1
C2 SW −×××π
=
Layout Consideration
In any high switching frequency converter, a correct lay-
out is important to ensure proper operation of the
regulator. With power devices switching at 300kHz, the
resulting current transient will cause voltage spike across
the interconnecting impedance and parasitic circuit
elements. As an example, consider the turn-off transition
of the PWM MOSFET. Before turn-off, the MOSFET is car-
rying the full load current. During turn-off, current stops
flowing in the MOSFET and is free-wheeling by the lower
MOSFET and parasitic diode. Any parasitic inductance of
the circuit generates a large voltage spike during the
switching interval. In general, using short and wide printed
circuit traces should minimize interconnecting imped-
ances and the magnitude of voltage spike. And signal
and power grounds are to be kept separating till com-
bined using the ground plane construction or single point
grounding. Figure 8. illustrates the layout, with bold lines
indicating high current paths; these traces must be short
Figure 7. Switching Waveform Across MOSFET
2. Place the zero FZ before the LC filter double poles FLC:
FZ = 0.75 x FLC
Calculate the C1 by the equation:
3. Set the pole at the half the switching frequency:
FP = 0.5xFSW
Calculate the C2 by the equation:
LCF0.75R121
C1 ×××π×
=
FLC
FESR
FP=0.5FSW
FZ=0.75FLC
FO
Frequency
PWM &
Filter Gain
Compensation
Gain
Converter
Gain
Gain
20 . log(gm . R3)
∆VOSC
VIN
20.log
Voltage across
drain and source of MOSFET
Time
VDS
tsw
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw15
Application Information (Cont.)
Layout Consideration (Cont.)
Figure 8. Layout Guidelines
- Keep the switching nodes (UGATE, LGATE, and PHASE)
away from sensitive small signal nodes since these
nodes are fast moving signals. Therefore, keep traces
to these nodes as short as possible.
- The traces from the gate drivers to the MOSFETs (UG
and LG) should be short and wide.
- Place the source of the high-side MOSFET and the drain
of the low-side MOSFET as close as possible. Minimiz-
ing the impedance with wide layout plane between the
two pads reduces the voltage bounce of the node.
- Decoupling capacitor, compensation component, the
resistor dividers, and boot capacitors should be close
their pins. (For example, place the decoupling ceramic
capacitor near the drain of the high-side MOSFET as
close as possible. The bulk capacitors are also placed
near the drain).
- The input capacitor should be near the drain of the up-
per MOSFET; the output capacitor should be near the
loads. The input capacitor GND should be close to the
output capacitor GND and the lower MOSFET GND.
- The drain of the MOSFETs (VIN and PHASE nodes) should
be a large plane for heat sinking.
- The ROCSET resistance should be placed near the IC as
close as possible.
and wide. Components along the bold lines should be
placed lose together. Below is a checklist for your layout:
VCC
BOOT
PHASE
UGATE
LGATE
VIN
VOUT
L
O
A
D
APW7165
ROCSET
Close to IC
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw16
Package Information
SOP-8
L
VIEW A
0.25
SEATING PLANE
GAUGE PLANE
Note: 1. Followed from JEDEC MS-012 AA.
2. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion or gate burrs shall not exceed 6 mil per side.
3. Dimension “E” does not include inter-lead flash or protrusions.
Inter-lead flash and protrusions shall not exceed 10 mil per side.
SEE VIEW A
c
h X 45o
E1
E
D
e b
A2
A
A1
NX aaa c
S
Y
M
B
O
LMIN. MAX.
1.75
0.10
0.17 0.25
0.25
A
A1
c
D
E
E1
e
h
L
MILLIMETERS
b0.31 0.51
SOP-8
0.25 0.50
0.40 1.27
MIN. MAX.
INCHES
0.069
0.004
0.012 0.020
0.007 0.010
0.010 0.020
0.016 0.050
0o8o0o8o
0.010
1.27 BSC 0.050 BSC
A2 1.25 0.049
4.80 5.00 0.189 0.197
3.80 4.00 0.150 0.157
5.80 6.20 0.228 0.244
0.10 0.004
aaa
θ
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw17
Application
A H T1 C d D W E1 F
330.0±2.00
50 MIN.
12.4+2.00
-0.00
13.0+0.50
-0.20
1.5 MIN.
20.2 MIN.
12.0±0.30
1.75±0.10
5.5±0.05
P0 P1 P2 D0 D1 T A0 B0 K0
SOP-8
4.0±0.10
8.0±0.10
2.0±0.05
1.5+0.10
-0.00
1.5 MIN.
0.6+0.00
-0.40
6.40±0.20
5.20±0.20
2.10±0.20
(mm)
Carrier Tape & Reel Dimensions
Devices Per Unit
Package Type Unit Quantity
SOP-8 Tape & Reel 2500
H
T1
A
d
A
E1
A
B
W
F
T
P0
OD0
BA0
P2
K0
B0
SECTION B-B
SECTION A-A
OD1
P1
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw18
Taping Direction Information
SOP-8
Classification Profile
USER DIRECTION OF FEED
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw19
Classification Reflow Profiles
Profile Feature Sn-Pb Eutectic Assembly Pb-Free Assembly
Preheat & Soak
Temperature min (Tsmin)
Temperature max (Tsmax)
Time (Tsmin to Tsmax) (ts)
100 °C
150 °C
60-120 seconds
150 °C
200 °C
60-120 seconds
Average ramp-up rate
(Tsmax to TP) 3 °C/second max. 3 °C/second max.
Liquidous temperature (TL)
Time at liquidous (tL) 183 °C
60-150 seconds 217 °C
60-150 seconds
Peak package body Temperature
(Tp)* See Classification Temp in table 1 See Classification Temp in table 2
Time (tP)** within 5°C of the specified
classification temperature (Tc) 20** seconds 30** seconds
Average ramp-down rate (Tp to Tsmax)
6 °C/second max. 6 °C/second max.
Time 25°C to peak temperature 6 minutes max. 8 minutes max.
* Tolerance for peak profile Temperature (Tp) is defined as a supplier minimum and a user maximum.
** Tolerance for time at peak profile temperature (tp) is defined as a supplier minimum and a user maximum.
Table 2. Pb-free Process – Classification Temperatures (Tc)
Package
Thickness Volume mm3
<350 Volume mm3
350-2000 Volume mm3
>2000
<1.6 mm 260 °C 260 °C 260 °C
1.6 mm – 2.5 mm 260 °C 250 °C 245 °C
≥2.5 mm 250 °C 245 °C 245 °C
Table 1. SnPb Eutectic Process – Classification Temperatures (Tc)
Package
Thickness Volume mm3
<350 Volume mm3
≥350
<2.5 mm 235 °C 220 °C
≥2.5 mm 220 °C 220 °C
Reliability Test Program
Test item Method Description
SOLDERABILITY JESD-22, B102 5 Sec, 245°C
HOLT JESD-22, A108 1000 Hrs, Bias @ Tj=125°C
PCT JESD-22, A102 168 Hrs, 100%RH, 2atm, 121°C
TCT JESD-22, A104 500 Cycles, -65°C~150°C
HBM MIL-STD-883-3015.7 VHBM≧2KV
MM JESD-22, A115 VMM≧200V
Latch-Up JESD 78 10ms, 1tr≧100mA
Copyright ANPEC Electronics Corp.
Rev. A.3 - Jan., 2011
APW7165
www.anpec.com.tw20
Customer Service
Anpec Electronics Corp.
Head Office :
No.6, Dusing 1st Road, SBIP,
Hsin-Chu, Taiwan, R.O.C.
Tel : 886-3-5642000
Fax : 886-3-5642050
Taipei Branch :
2F, No. 11, Lane 218, Sec 2 Jhongsing Rd.,
Sindian City, Taipei County 23146, Taiwan
Tel : 886-2-2910-3838
Fax : 886-2-2917-3838