104PW02F - NEC - Datasheet.Live

LM48511

LM48511 3W, Ultra-Low EMI, Filterless, Mono, Class D Audio Power

Amplifier with Spread Spectrum

Literature Number: SNAS416C

February 25, 2008

LM48511

3W, Ultra-Low EMI, Filterless, Mono, Class D Audio Power

Amplifier with Spread Spectrum

General Description

The LM48511 integrates a boost converter with a high effi-

ciency Class D audio power amplifier to provide 3W continu-

ous power into an 8Ω speaker when operating from a 5V

power supply. When operating from a 3V to 4V power supply,

the LM48511 can be configured to drive 1 to 2.5W into an

8Ω load with less than 1% distortion (THD+N). The Class D

amplifier features a low noise PWM architecture that elimi-

nates the output filter, reducing external component count,

board area consumption, system cost, and simplifying design.

A selectable spread spectrum modulation scheme suppress-

es RF emissions, further reducing the need for output filters.

The LM48511’s switching regulator is a current-mode boost

converter operating at a fixed frequency of 1MHz. Two se-

lectable feedback networks allow the LM48511 regulator to

dynamically switch between two different output voltages, im-

proving efficiency by optimizing the amplifier’s supply voltage

based on battery voltage and output power requirements.

The LM48511 is designed for use in portable devices, such

as GPS, mobile phones, and MP3 players. The high, 80%

efficiency at 5V, extends battery life when compared to Boost-

ed Class AB amplifiers. Independent regulator and amplifier

shutdown controls optimize power savings by disabling the

regulator when high output power is not required.

The gain of the LM48511 is set by external resistors, which

allows independent gain control from multiple sources by

summing the signals. Output short circuit and thermal over-

load protection prevent the device from damage during fault

conditions. Superior click and pop suppression eliminates au-

dible transients during power-up and shutdown.

Key Specifications

■ Quiescent Power Supply Current

VDD = 3V

VDD = 5V

9mA (typ)

13.5mA (typ)

■ PO at VDD = 5V, PV1 = 7.8V

RL = 8Ω, THD+N = 1% 3.0W (typ)

■ PO at VDD = 3V, PV1 = 4.8V

RL = 8Ω, THD+N = 1% 1W (typ)

■ PO at VDD = 5V, PV1 = 7.8V

RL = 4Ω, THD+N = 1% 5.4W (typ)

■ Shutdown Current at VDD = 3V 0.01μA (typ)

Features

■3W Output into 8Ω at 5V with THD+N = 1%

■Selectable spread spectrum mode reduces EMI

■80% Efficiency

■Independent Regulator and Amplifier Shutdown Controls

■Dynamically Selectable Regulator Output Voltages

■Filterless Class D

■3.0V – 5.5V operation

■Low Shutdown Current

■Click and Pop Suppression

Applications

■GPS

■Portable media

■Cameras

■Mobile Phones

■Handheld games

EMI Graph

300222h5

FIGURE 1. LM48511 RF Emissions — 3 inch cable

Boomer® is a registered trademark of National Semiconductor Corporation.

LM48511 3W, Ultra-Low EMI, Filterless, Mono, Class D Audio Power Amplifier

with Spread Spectrum

Typical Application

300222i3

FIGURE 2. Typical LM48511 Audio Amplifier Application Circuit

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LM48511

Connection Diagrams

SQ Package

300222d4

Top View

Order Number LM48511SQ

See NS Package Number SQA24B

SQ Package Marking

300222d5

Top View

U — Wafer fab code

Z — Assembly plant

XY — 2 Digit date code

TT — Lot traceability

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LM48511

Pin Descriptions

LLP-24 Pin Name Function

1 FB_SEL

Regulator Feedback Select. Connect to VDD to select feedback

network connected to FB_GND1. Connect to GND to select

feedback network connected to FB_GND0.

2,3 SW Drain of the Internal FET Switch

4 SOFTSTART Soft Start Capacitor

5 SD_AMP Amplifier Active Low Shutdown. Connect to VDD for normal

operation. Connect to GND to disable amplifier.

6 SS/FF Modulation Mode Select. Connect to VDD for spread spectrum

mode (SS). Connect to GND for fixed frequency mode (FF).

7 GND Signal Ground

8 LS+ Amplifier Non-Inverting Output

9, 11 LSGND Amplifier H-Bridge Ground

10 PV1 Amplifier H-Bridge Power Supply. Connect to V1.

12 LS- Amplifier Inverting Output

13 V1 Amplifier Supply Voltage. Connect to PV1

14 VG0+ Amplifier Non-Inverting Gain Output

15 IN- Amplifier Inverting Input

16 IN+ Amplifier Non-Inverting Input

17 VG0– Amplifier Inverting Gain Output

18 VDD Power Supply

19 FB Regulator Feedback Input. Connect FB to an external resistive

voltage divider to set the boost output voltage.

20 FB_GND1 Ground return for R3, R1 resistor divider

21 FB_GND0 Ground return for R3, R2 resistor divider

22,23 REGGND Power Ground (Booster)

24 SD_BOOST Regulator Active Low Shutdown. Connect to VDD for normal

operation. Connect to GND to disable regulator.

DAP To be soldered to board for enhanced thermal dissipation. Connect

to GND plane.

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LM48511

Absolute Maximum Ratings (Notes 2, 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage (VDD, PV1, V1)9V

Storage Temperature −65°C to +150°C

Input Voltage −0.3V to VDD + 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150°C

Thermal Resistance

θJC (SQ) 3.8°C/W

θJA (SQ) 32.8°C/W

Operating Ratings

Temperature Range

TMIN ≤ TA ≤ TMAX −40°C ≤ TA ≤ +85°C

Supply Voltage (VDD) 3.0V ≤ VDD ≤ 5.5V

Amplifier Voltage (PV1, V1) 4.8V ≤ PV1 ≤ 8.0V

Electrical Characteristics VDD = 5.0V (Notes 1, 2, 10)

The following specifications apply for VDD = 5.0V, PV1 = 7.8V (continuos mode), AV = 2V/V, R3 = 25.5kΩ, RLS = 4.87kΩ, RL =

8Ω, f = 1kHz, SS/FF = GND, unless otherwise specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions

LM48511 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7)

IDD Quiescent Power Supply Current

VIN = 0, RLOAD = ∞

Fixed Frequency Mode (FF) 13.5 mA (max)

Spread Spectrum Mode (SS) 14.5 22 mA (max)

ISD Shutdown Current VSD_BOOST = VSD_AMP = SS =

FB_SEL = GND 0.11 1 μA (max)

VIH Logic Voltage Input High 1.03 1.4 V (min)

VIL Logic Voltage Input Low 0.92 0.4 V (min)

TWU Wake-up Time CSS = 0.1μF49 ms

VOS Output Offset Voltage Note 12 0.01 3 mV

POOutput Power

RL = 8Ω

f = 1kHz, BW = 22kHz

THD+N = 1%

3.0

2.6 W (min)

THD+N = 10%

3.8

RL = 4Ω

f = 1kHz, BW = 22kHz

THD+N = 1%

5.4

THD+N = 10%

6.7

THD+N Total Harmonic Distortion + Noise

PO = 2W, f = 1kHz, RL = 8Ω

0.03

PO = 3W, f = 1kHz, RL = 4Ω

0.04

0.05

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LM48511

Symbol Parameter Conditions

LM48511 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7)

εOS Output Noise

f = 20Hz to 20kHz

Inputs to AC GND, No weighting

µVRMS

f = 20Hz to 20kHz

Inputs to AC GND, A weighted

µVRMS

PSRR Power Supply Rejection Ratio

(Input Referred)

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 217Hz,

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 1kHz,

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 10kHz,

CMRR Common Mode Rejection Ratio

(Input Referred)

VRIPPLE = 1VP-P, fRIPPLE = 217Hz 73 dB

ηEfficiency f = 1kHz, RL = 8Ω, PO = 1W 80 %

VFB Feedback Pin Reference Voltage 1.23 V

Electrical Characteristics VDD = 3.6V (Notes 1, 2, 10)

The following specifications apply for VDD = 3.6V, PV1 = 7V (continuous mode), AV = 2V/V, R3 = 25.5kΩ, RLS = 5.36kΩ, RL = 8Ω,

f = 1kHz, SS/FF = GND, unless otherwise specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions

LM48511 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7)

IDD Quiescent Power Supply Current

VIN = 0, RLOAD = ∞

Fixed Frequency Mode (FF) 16 mA (max)

Spread Spectrum Mode (SS) 17.5 26.6 mA (max)

ISD Shutdown Current VSD_BOOST = VSD_AMP = SS =

FB_SEL = GND 0.03 1 μA (max)

VIH Logic Voltage Input High 0.96 1.4 V (min)

VIL Logic Voltage Input Low 0.84 0.4 V (min)

TWU Wake-up Time CSS = 0.1μF50 ms

VOS Output Offset Voltage Note 12 0.04 mV

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LM48511

Symbol Parameter Conditions

LM48511 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7)

POOutput Power

RL = 8Ω, f = 1kHz, BW = 22kHz

THD+N = 1%

2.5

THD+N = 10%

3.0

RL = 4Ω, f = 1kHz, BW = 22kHz

THD+N = 1%

4.3

4.2

THD+N = 10%

5.4

5.3

THD+N Total Harmonic Distortion + Noise

PO = 1.5W, f = 1kHz, RL = 8Ω

0.03

0.03 %

PO = 3W, f = 1kHz, RL = 4Ω

0.04

0.05 %

εOS Output Noise

f = 20Hz to 20kHz

Inputs to AC GND, No weighting

µVRMS

f = 20Hz to 20kHz

Inputs to AC GND, A weighted

µVRMS

PSRR Power Supply Rejection Ratio

(Input Referred)

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 217Hz

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 1kHz

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 10kHz

CMRR Common Mode Rejection Ratio

(Input Referred)

VRIPPLE = 1VP-P, fRIPPLE = 217Hz 73 dB

ηEfficiency f = 1kHz, RL = 8Ω, PO = 1W 77 %

VFB Feedback Pin Reference Voltage 1.23 V

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LM48511

Electrical Characteristics VDD = 3.0V (Notes 1, 2, 10)

The following specifications apply for VDD = 3.0V, PV1 = 4.8V (continuos mode), AV = 2V/V, R3 = 25.5kΩ, RLS = 9.31kΩ, RL =

8Ω, f = 1kHz, SS/FF = GND, unless otherwise specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions

LM48511 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7)

IDD Quiescent Power Supply Current

VIN = 0, RLOAD = ∞

Fixed Frequency Mode (FF) 9 mA (max)

Spread Spectrum Mode (SS) 9.5 mA (max)

ISD Shutdown Current VSD_BOOST = VSD_AMP = SS =

FB_SEL = GND 0.01 1 μA

VIH Logic Voltage Input High 0.91 V (min)

VIL Logic Voltage Input Low 0.79 V

TWU Wake-up Time CSS = 0.1μF49 ms

VOS Output Offset Voltage Note 12 0.04 mV

POOutput Power

RL = 8Ω, f = 1kHz, BW = 22kHz

THD+N = 1%

0.84 W (min)

THD+N = 10%

1.3

RL = 4Ω, f = 1kHz, BW = 22kHz

THD+N = 1%

1.8

THD+N = 10%

2.2

THD+N Total Harmonic Distortion + Noise

PO = 500mW, f = 1kHz, RL = 8Ω

0.02

0.03

PO = 500mW, f = 1kHz, RL = 4Ω

0.04

0.06

εOS Output Noise

f = 20Hz to 20kHz

Inputs to AC GND, No weighting

µVRMS

f = 20Hz to 20kHz

Inputs to AC GND, A weighted

µVRMS

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LM48511

Symbol Parameter Conditions

LM48511 Units

(Limits)

Typical

(Note 6)

Limit

(Note 7)

PSRR Power Supply Rejection Ratio

(Input Referred)

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 217Hz

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 1kHz

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 10kHz

CMRR Common Mode Rejection Ratio

(Input Referred)

VRIPPLE = 1VP-P, fRIPPLE = 217Hz 71 dB

ηEfficiency f = 1kHz, RL = 8Ω, PO = 1W 75 %

VFB Feedback Pin Reference Voltage 1.23 V

Note 1: “Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability

and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in

the Recommended Operating Conditions is not implied. TheRecommended Operating Conditions indicate conditions at which the device is functional and the

device should not be operated beyond such conditions. All voltages are measured with respect to the ground pin, unless otherwise specified.

Note 2: The Electrical Characteristics tables list guaranteed specifications under the listed Recommended Operating Conditions except as otherwise modified

or specified by the Electrical Characteristics Conditions and/or Notes. Typical specifications are estimations only and are not guaranteed.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX = (TJMAX - TA) / θJA or the number given in Absolute Maximum Ratings, whichever is lower. For the LM48511, see power

derating curves for additional information.

Note 4: Human body model, applicable std. JESD22-A114C.

Note 5: Machine model, applicable std. JESD22-A115-A.

Note 6: Typical values represent most likely parametric norms at TA = +25ºC, and at the Recommended Operation Conditions at the time of product

characterization and are not guaranteed.

Note 7: Datasheet min/max specification limits are guaranteed by test or statistical analysis.

Note 8: Shutdown current is measured with components R1 and R2 removed.

Note 9: Feedback pin reference voltage is measured with the Audio Amplifier disconnected from the Boost converter (the Boost converter is unloaded).

Note 10: RL is a resistive load in series with two inductors to simulate an actual speaker load for RL = 8Ω, the load is 15μH+8Ω+15μH. For RL = 4Ω, the load is

15μH+4Ω+15μH.

Note 11: Offset voltage is determined by: (IDD (with load) — IDD (no load)) x RL.

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LM48511

Typical Performance Characteristics

THD+N vs Frequency

VDD = 5V, RL = 8Ω

PO = 2W, filter = 22kHz, PV1 = 7.8V

300222g9

THD+N vs Frequency

VDD = 3.6V, RL = 8Ω

PO = 500mW, filter = 22kHz, PV1 = 4.8V

300222g7

THD+N vs Frequency

VDD = 3V, RL = 8Ω

PO = 1.5W, filter = 22kHz, PV1 = 7V

300222g8

THD+N vs Output Power

VDD = 5V, RL = 8Ω

PO = 1.5W, f = 1kHz, filter = 22kHz, PV1 = 7.8V

300222h4

THD+N vs Output Power

VDD = 3.6V, RL = 8Ω

f = 1kHz, filter = 22kHz, PV1 = 7V

300222h1

THD+N vs Output Power

VDD = 3V, RL = 8Ω

f = 1kHz, filter = 22kHz, PV1 = 4.8V

300222h3

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LM48511

THD+N vs Output Power

VDD = 3V, 3.6V, 5V, RL = 8Ω

f = 1kHz, filter = 22kHz, R1 = 4.87kΩ, FF

300222i1

THD+N vs Output Power

VDD = 3.6V, RL = 8Ω

filter = 22kHz, PV1 = 7.8V, PV1 = 7V, PV1 = 4.8V, FF

300222i0

Boost Amplifier vs Output Power

VDD = 5V, RL = 8Ω

f = 1kHz, PV1 = 7.8V

300222f7

Boost Amplifier vs Output Power

VDD = 3.6V, RL = 8Ω

f = 1kHz, PV1 = 7V

300222f5

Boost Amplifier vs Output Power

VDD = 3V, RL = 8Ω

f = 1kHz, PV1 = 4.8V

300222f6

PSRR vs Frequency

VDD = 5V, RL = 8Ω

VRIPPLE = 200mVPP, PV1 = 7.8V

300222g3

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LM48511

PSRR vs Frequency

VDD = 3.6V, RL = 8Ω

VRIPPLE = 200mVPP, PV1 = 7V

300222g1

PSRR vs Frequency

VDD = 3V, RL = 8Ω

VRIPPLE = 200mVPP, PV1 = 4.8V

300222g2

Supply Current vs Supply Voltage

PV1 = 7.8V

300222g6

Supply Current vs Supply Voltage

PV1 = 7V

300222g5

Supply Current vs Supply Voltage

PV1 = 4.8V

300222g4

Power Dissipation vs Output Power

VDD = 5V, RL = 8Ω

PV1 = 7.8V, FF

300222g0

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LM48511

Power Dissipation vs Output Power

VDD = 3.6V, RL = 8Ω

PV1 = 7V, FF

300222f8

Power Dissipation vs Output Power

VDD = 3V, RL = 8Ω

PV1 = 4.8V, FF

300222f9

Boost Converter Efficiency vs ILOAD(DC)

VDD = 5V, PV1 = 7.8V

300222h8

Boost Converter Efficiency vs ILOAD(DC)

VDD = 3.6V, PV1 =7V

300222h6

Boost Converter Efficiency vs ILOAD(DC)

VDD = 3V, PV1 = 4.8V

300222h7

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LM48511

Application Information

GENERAL AMPLIFIER FUNCTION

The LM48511 features a Class D audio power amplifier that

utilizes a filterless modulation scheme, reducing external

component count, conserving board space and reducing sys-

tem cost. The outputs of the device transition from PV1 to

GND with a 300kHz switching frequency. With no signal ap-

plied, the outputs (VLS+ and VLS-) switch with a 50% duty

cycle, in phase, causing the two outputs to cancel. This can-

cellation results in no net voltage across the speaker, thus

there is no current to the load in the idle state.

With the input signal applied, the duty cycle (pulse width) of

the LM48511 outputs changes. For increasing output voltage,

the duty cycle of VLS+ increases, while the duty cycle of VLS-

decreases. For decreasing output voltages, the converse

occurs. The difference between the two pulse widths yields

the differential output voltage.

FIXED FREQUENCY

The LM48511 features two modulations schemes, a fixed fre-

quency mode (FF) and a spread spectrum mode (SS). Select

the fixed frequency mode by setting SS/FF = GND. In fixed

frequency mode, the amplifier outputs switch at a constant

300kHz. In fixed frequency mode, the output spectrum con-

sists of the fundamental and its associated harmonics (see

Typical Performance Characteristics).

SPREAD SPECTRUM MODE

The logic selectable spread spectrum mode eliminates the

need for output filters, ferrite beads or chokes. In spread

spectrum mode, the switching frequency varies randomly by

10% about a 330kHz center frequency, reducing the wide-

band spectral contend, improving EMI emissions radiated by

the speaker and associated cables and traces. Where a fixed

frequency class D exhibits large amounts of spectral energy

at multiples of the switching frequency, the spread spectrum

architecture of the LM48511 spreads that energy over a larger

bandwidth (See Typical Performance Characteristics). The

cycle-to-cycle variation of the switching period does not affect

the audio reproduction, efficiency, or PSRR. Set SS/FF =

VDD for spread spectrum mode.

DIFFERENTIAL AMPLIFIER EXPLANATION

The LM48511 includes fully differential amplifier that features

differential input and output stages. A differential amplifier

amplifies the difference between the two input signals. Tradi-

tional audio power amplifiers have typically offered only sin-

gle-ended inputs resulting in a 6dB reduction in signal to noise

ratio relative to differential inputs. The LM48511 also offers

the possibility of DC input coupling which eliminates the two

external AC coupling, DC blocking capacitors. The LM48511

can be used, however, as a single ended input amplifier while

still retaining it's fully differential benefits. In fact, completely

unrelated signals may be placed on the input pins. The

LM48511 simply amplifies the difference between the signals.

A major benefit of a differential amplifier is the improved com-

mon mode rejection ratio (CMRR) over single input amplifiers.

The common-mode rejection characteristic of the differential

amplifier reduces sensitivity to ground offset related noise in-

jection, especially important in high noise applications.

AUDIO AMPLIFIER POWER DISSIPATION AND

EFFICIENCY

The major benefit of a Class D amplifier is increased efficiency

versus a Class AB. The efficiency of the LM48511 is attributed

to the region of operation of the transistors in the output stage.

The Class D output stage acts as current steering switches,

consuming negligible amounts of power compared to their

Class AB counterparts. Most of the power loss associated

with the output stage is due to the IR loss of the MOSFET on-

resistance, along with switching losses due to gate charge.

REGULATOR POWER DISSIPATION

At higher duty cycles, the increased ON-time of the switch

FET means the maximum output current will be determined

by power dissipation within the LM48511 FET switch. The

switch power dissipation from ON-time conduction is calcu-

lated by:

PD(SWITCH) = DC x (IINDUCTOR(AVE))2 x RDS(ON) (W) (1)

where DC is the duty cycle.

SHUTDOWN FUNCTION

The LM48511 features independent amplifier and regulator

shutdown controls, allowing each portion of the device to be

disabled or enabled independently. SD_AMP controls the

Class D amplifiers, while SD_BOOST controls the regulator.

Driving either inputs low disables the corresponding portion

of the device, and reducing supply current.

When the regulator is disabled, both FB_GND switches open,

further reducing shutdown current by eliminating the current

path to GND through the regulator feedback network. Without

the GND switches, the feedback resistors as shown in Figure

1 would consume an additional 165μA from a 5V supply. With

the regulator disabled, there is still a current path from VDD,

through the inductor and diode, to the amplifier power supply.

This allows the amplifier to operate even when the regulator

is disabled. The voltage at PV1 and V1 will be:

(VDD - [VD + (IL x DCR)] (2)

Where VD is the forward voltage of the Schottky diode, IL is

the current through the inductor, and DCR is the DC resis-

tance of the inductor. Additionally, when the regulator is dis-

abled, an external voltage between 5V and 8V can be applied

directly to PV1 and V1 to power the amplifier.

It is best to switch between ground and VDD for minimum cur-

rent consumption while in shutdown. The LM48511 may be

disabled with shutdown voltages in between GND and VDD,

the idle current will be greater than the typical 0.1µA value.

Increased THD+N may also be observed when a voltage of

less than VDD is applied to SD_AMP .

REGULATOR FEEDBACK SELECT

The LM45811 regulator features two feedback paths as

shown in Figure 1, which allow the regulator to easily switch

between two different output voltages. The voltage divider

consists of the high side resistor, R3, and the low side resis-

tors (RLS), R1 and R2. R3 is connected to the output of the

boost regulator, the mid-point of each divider is connected to

FB, and the low side resistors are connected to either

FB_GND1 or FB_GND0. FB_SEL determines which

FB_GND switch is closed, which in turn determines which

feedback path is used. For example if FB_SEL = VDD, the

FB_GND1 switch is closed, while the FB_GND0 switch re-

mains open, creating a current path through the resistors

connected to FB_GND1. Conversely, if FB_SEL = GND, the

FB_GND0 switch is closed, while the FB_GND1 switch re-

mains open, creating a current path through the resistors

connected to FB_GND0.

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LM48511

FB_SEL can be susceptible to noise interference. To prevent

an accidental state change, either bypass FB_SEL with a

0.1µF capacitor to GND, or connect the higher voltage feed-

back network to FB_GND0, and the lower voltage feedback

network to FB_GND1. Because the higher output voltage

configuration typically generates more noise on VDD, this con-

figuration minimizes the VDD noise exposure of FB_SEL, as

FB_SEL = GND for FB_GND0 (high voltage output) and

FB_SEL = VDD for FB_GND1 (low voltage output).

The selectable feedback networks maximize efficiency in two

ways. In applications where the system power supply voltage

changes, such as a mobile GPS receiver, that transitions from

battery power, to AC line, to a car power adapter, the

LM48511 can be configured to generate a lower voltage when

the system power supply voltages is lower, and conversely,

generate a higher voltage when the system power supply is

higher. See the Setting the Regulator Output Voltage (PV1)

section.

In applications where the same speaker/amplifier combina-

tion is used for different purposes with different audio power

requirements, such as a cell phone ear piece/speaker phone

speaker, the ability to quickly switch between two different

voltages allows for optimization of the amplifier power supply,

increasing overall system efficiency. When audio power de-

mands are low (ear piece mode) the regulator output voltage

can be set lower, reducing quiescent current consumption.

When audio power demands increase (speaker phone

mode), a higher voltage increases the amplifier headroom,

increasing the audio power delivered to the speaker.

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications using

integrated power amplifiers, and switching DC-DC convert-

ers, is critical for optimizing device and system performance.

Consideration to component values must be used to maxi-

mize overall system quality. The best capacitors for use with

the switching converter portion of the LM48511 are multi-layer

ceramic capacitors. They have the lowest ESR (equivalent

series resistance) and highest resonance frequency, which

makes them optimum for high frequency switching convert-

ers. When selecting a ceramic capacitor, only X5R and X7R

dielectric types should be used. Other types such as Z5U and

Y5F have such severe loss of capacitance due to effects of

temperature variation and applied voltage, they may provide

as little as 20% of rated capacitance in many typical applica-

tions. Always consult capacitor manufacturer’s data curves

before selecting a capacitor. High-quality ceramic capacitors

can be obtained from Taiyo-Yuden and Murata.

POWER SUPPLY BYPASSING

As with any amplifier, proper supply bypassing is critical for

low noise performance and high power supply rejection. The

capacitor location on both PV1, V1 and VDD pins should be

as close to the device as possible.

AUDIO AMPLIFIER GAIN SETTING RESISTOR

SELECTION

The amplifier gain of the LM48511 is set by four external re-

sistors, the input resistors, R5 and R7, and the feed back

resistors R6 and R8.. The amplifier gain is given by:

Where RIN is the input resistor and RF is the feedback resistor.

AVD = 2 X RF / RIN (3)

Careful matching of the resistor pairs, R6 and R8, and R5 and

R7, is required for optimum performance. Any mismatch be-

tween the resistors results in a differential gain error that leads

to an increase in THD+N, decrease in PSRR and CMRR, as

well as an increase in output offset voltage. Resistors with a

tolerance of 1% or better are recommended.

The gain setting resistors should be placed as close to the

device as possible. Keeping the input traces close together

and of the same length increases noise rejection in noisy en-

vironments. Noise coupled onto the input traces which are

physically close to each other will be common mode and eas-

ily rejected.

AUDIO AMPLIFIER INPUT CAPACITOR SELECTION

Input capacitors may be required for some applications, or

when the audio source is single-ended. Input capacitors block

the DC component of the audio signal, eliminating any conflict

between the DC component of the audio source and the bias

voltage of the LM48511. The input capacitors create a high-

pass filter with the input resistors RIN. The -3dB point of the

high pass filter is found by:

f = 1 / 2πRINCIN (4)

In single-ended configurations, the input capacitor value af-

fects click-and-pop performance. The LM48511 features a

50mg turn-on delaly. Choose the input capacitor / input re-

sistor values such that the capacitor is charged before the

50ms turn-on delay expires. A capacitor value of 0.18μF and

a 20kΩ input resistor are recommended. In differential appli-

cations, the charging of the input capacitor does not affect

click-and-pop significantly.

The input capacitors can also be used to remove low fre-

quency content from the audio signal. High pass filtering the

audio signal helps protect speakers that can not reproduce or

may be damaged by low frequencies. When the LM48511 is

using a single-ended source, power supply noise on the

ground is seen as an input signal. Setting the high-pass filter

point above the power supply noise frequencies, 217Hz in a

GSM phone, for example, filters out the noise such that it is

not amplified and heard on the output. Capacitors with a tol-

erance of 10% or better are recommended for impedance

matching and improved CMRR and PSRR.

SELECTING REGULATOR OUTPUT CAPACITOR

A single 100µF low ESR tantalum capacitor provides suffi-

cient output capacitance for most applications. Higher capac-

itor values improve line regulation and transient response.

Typical electrolytic capacitors are not suitable for switching

converters that operate above 500kHz because of significant

ringing and temperature rise due to self-heating from ripple

current. An output capacitor with excessive ESR reduces

phase margin and causes instability.

SELECTING REGULATING BYPASS CAPACITOR

A supply bypass capacitor is required to serve as an energy

reservoir for the current which must flow into the coil each time

the switch turns on. This capacitor must have extremely low

ESR, so ceramic capacitors are the best choice. A nominal

value of 10μF is recommended, but larger values can be

used. Since this capacitor reduces the amount of voltage rip-

ple seen at the input pin, it also reduces the amount of EMI

passed back along that line to other circuitry.

SELECTING THE SOFTSTART (CSS) CAPACITOR

The soft-start function charges the boost converter reference

voltage slowly. This allows the output of the boost converter

to ramp up slowly thus limiting the transient current at startup.

15 www.national.com

LM48511

Selecting a soft-start capacitor (CSS) value presents a trade

off between the wake-up time and the startup transient cur-

rent. Using a larger capacitor value will increase wake-up time

and decrease startup transient current while the apposite ef-

fect happens with a smaller capacitor value. A general guide-

line is to use a capacitor value 1000 times smaller than the

output capacitance of the boost converter (C2). A 0.1uF soft-

start capacitor is recommended for a typical application.

The following table shows the relationship between CSS start-

up time and surge current.

CSS

(μF)

Boost Set-up Time

(ms)

Input Surge Current

(mA)

0.1 5.1 330

0.22 10.5 255

0.47 21.7 220

VDD = 5V, PV1 = 7.8V (continuous mode)

SELECTING DIODE (D1)

Use a Schottkey diode, as shown in Figure 1. A 30V diode

such as the DFLS230LH from Diodes Incorporated is recom-

mended. The DFLS230LH diodes are designed to handle a

maximum average current of 2A.

DUTY CYCLE

The maximum duty cycle of the boost converter determines

the maximum boost ratio of output-to-input voltage that the

converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined by:

Duty Cycle = (PV1+VD-VDD)/(PV1+VD-VSW) (5)

This applies for continuous mode operation.

SELECTING INDUCTOR VALUE

Inductor value involves trade-offs in performance. Larger in-

ductors reduce inductor ripple current, which typically means

less output voltage ripple (for a given size of output capacitor).

Larger inductors also mean more load power can be delivered

because the energy stored during each switching cycle is:

E = L/2 x IP2(6)

Where “lp” is the peak inductor current. The LM48511 will limit

its switch current based on peak current. With IP fixed, in-

creasing L will increase the maximum amount of power avail-

able to the load. Conversely, using too little inductance may

limit the amount of load current which can be drawn from the

output. Best performance is usually obtained when the con-

verter is operated in “continuous” mode at the load current

range of interest, typically giving better load regulation and

less output ripple. Continuous operation is defined as not al-

lowing the inductor current to drop to zero during the cycle.

Boost converters shift over to discontinuous operation if the

load is reduced far enough, but a larger inductor stays con-

tinuous over a wider load current range.

INDUCTOR SUPPLIES

The recommended inductor for the LM48511 is the

IHLP-2525CZ-01 from Vishay Dale. When selecting an in-

ductor, the continuous current rating must be high enough to

avoid saturation at peak currents. A suitable core type must

be used to minimize switching losses, and DCR losses must

be considered when selecting the current rating. Use shielded

inductors in systems that are susceptible to RF interference.

SETTING THE REGULATOR OUTPUT VOLTAGE (PV1)

The output voltage of the regulator is set through one of two

external resistive voltage-dividers (R3 in combination with ei-

ther R1 or R2) connected to FB (Figure 1). The resistor, R4

is only for compensation purposes and does not affect the

regulator output voltage. The regulator output voltage is set

by the following equation:

PV1 = VFB [1+R3/RLS] (7)

Where VFB is 1.23V, and RLS is the low side resistor (R1 or

R2). To simplify resistor selection:

RLS = (R3VFB) / (PV1–VFB) (8)

A value of approximately 25.5kΩ is recommended for R3.

The quiescent current of the boost regulator is directly related

to the difference between its input and output voltages, the

larger the difference, the higher the quiescent current. For

improved power consumption the following regulator input/

output voltage combinations are recommended:

VDD (V) PV1 (V) R3(kΩ) RLS (kΩ) POUT into 8Ω (W)

3.0 4.8 25.5 9.31 1

3.6 7.1 25.5 5.35 2.5

5 7.8 25.5 4.87 3

The values of PV1 are for continuous mode operation.

For feedback path selection, see Regulator Feedback Select

section.

DISCONTINUOUS/CONTINUOUS OPERATION

The LM48511 regulator features two different switching

modes. Under light load conditions, the regulator operates in

a variable frequency, discontinuous, pulse skipping mode,

that improves light load efficiency by minimizing losses due

to MOSFET gate charge. Under heavy loads, the LM48511

regulator automatically switches to a continuous, fixed fre-

quency PWM mode, improving load regulation. In discontin-

uous mode, the regulator output voltage is typically 400mV

higher than the expected (calculated) voltage in continuous

mode.

ISWFEED-FORWARD COMPENSATION FOR BOOST

CONVERTER

Although the LM48511 regulator is internally compensated,

an external feed-forward capacitor, (C1) may be required for

stability (Figure 1). The compensation capacitor places a zero

in regulator loop response. The recommended frequency of

the zero (fZ) is 22.2kHz. The value of C1 is given by:

C1 =1 / 2πR3fZ(9)

In addition to C1, a compensation resistor, R4 is required to

cancel the zero contributed by the ESR of the regulator output

capacitor. Calculate the zero frequency of the output capaci-

tor by:

fCO =1 / 2πRCOCO(10)

www.national.com 16

LM48511

Where R CO is the ESR of the output capacitor. The value of

RFB3 is given by:

R4 =1 / 2πfCOC1 (11)

CALCULATING REGULATOR OUTPUT CURRENT

The load current of the boost converter is related to the av-

erage inductor current by the relation:

IAMP = IINDUCTOR(AVE) x (1 - DC) (A) (12)

Where "DC" is the duty cycle of the application.

The switch current can be found by:

ISW = IINDUCTOR(AVE) + 1/2 (IRIPPLE) (A) (13)

Inductor ripple current is dependent on inductance, duty cy-

cle, supply voltage and frequency:

IRIPPLE = DC x (VDD-VSW) / (f x L) (A) (14)

where f = switching frequency = 1MHz

combining all terms, we can develop an expression which al-

lows the maximum available load current to be calculated:

IAMP(max) = (1–DC)x[ISW(max)–DC(V-VSW)]/2fL (A) (15)

The equation shown to calculate maximum load current takes

into account the losses in the inductor or turn-off switching

losses of the FET and diode.

DESIGN PARAMETERS VSW AND ISW

The value of the FET "ON" voltage (referred to as VSW in

equations 9 thru 12) is dependent on load current. A good

approximation can be obtained by multiplying the on resis-

tance (RDS(ON) of the FET times the average inductor current.

The maximum peak switch current the device can deliver is

dependent on duty cycle.

17 www.national.com

LM48511

Build Of Materials

Designator Description Footprint Quantity Value

Cf1 CHIP CAPACITOR GENERIC CAP 0805 1 470pF

CINA CHIP CAPACITOR GENERIC CAP 1210 1 1μF

CINB CHIP CAPACITOR GENERIC CAP 1210 1 1μF

Co CHIP CAPACITOR GENERIC CAP 1210 1 10μF

Cs1 CHIP CAPACITOR GENERIC CAP 1210 1 2.2μF

Cs2 CHIP CAPACITOR GENERIC CAP 1210 1 4.7μF

D1 SCHOTTKY DIO DIODE MBR0520 IR 1

L1 IND_COILCRAFT-DO1813P 1 4.7μH

R1 CHIP RESISTOR GENERIC RES 0805 1 41.2K

R2 CHIP RESISTOR GENERIC RES 0805 1 13.3K

RINA CHIP RESISTOR GENERIC RES 0805 1 150K

RINB CHIP RESISTOR GENERIC RES 0805 1 150K

www.national.com 18

LM48511

Revision History

Rev Date Description

1.0 07/24/07 Initial release.

1.1 07/25/07 Input some text edits.

1.2 09/25/07 Changed the Amplifier Voltage (Operating Ratings section) from 5.0V to

4.8V.

1.3 11/06/07 Added another Po (@Vdd = 5V, Rl = 4Ω) section in the Key Specification

division.

1.4 02/25/08 Edited the Notes section.

19 www.national.com

LM48511

Physical Dimensions inches (millimeters) unless otherwise noted

SQ Package

Order Number LM48511SQ

NS Package Number SQA24B

www.national.com 20

LM48511

Notes

21 www.national.com

LM48511

Notes

LM48511 3W, Ultra-Low EMI, Filterless, Mono, Class D Audio Power Amplifier

with Spread Spectrum

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