LTC3121 15V, 1.5A Synchronous Step-Up DC/DC Converter with Output Disconnect Description Features n n n n n n n n n n n n VIN Range: 1.8V to 5.5V, 500mV After Start-Up Output Voltage Range: 2.2V to 15V 400mA Output Current for VIN = 5V and VOUT = 12V Output Disconnects from Input When Shut Down Synchronous Rectification: Up to 95% Efficiency Inrush Current Limit Up to 3MHz Adjustable Switching Frequency Synchronizable to External Clock Selectable Burst Mode(R) Operation: 25A IQ Output Overvoltage Protection Soft-Start <1A IQ in Shutdown 12-Lead, 3mm x 4mm Thermally Enhanced DFN Package The LTC(R)3121 is a synchronous step-up DC/DC converter with true output disconnect and inrush current limiting. The 1.5A current limit along with the ability to program output voltages up to 15V makes the LTC3121 well suited for a variety of demanding applications. Once started, operation will continue with inputs down to 500mV, extending run time in many applications. Applications n n n n PCI Express Cards/Systems Piezo Actuators Small DC Motors 12V Analog Rail From Battery, 5V, or Backup Capacitor L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. The LTC3121 features output disconnect in shutdown, dramatically reducing input power drain and enabling VOUT to completely discharge. Adjustable PWM switching from 100kHz to 3MHz optimizes applications for highest efficiency or smallest solution footprint. The oscillator can also be synchronized to an external clock for noise sensitive applications. Selectable Burst Mode operation reduces quiescent current to 25A, ensuring high efficiency across the entire load range. An internal soft-start limits inrush current during start-up. Other features include a <1A shutdown current and robust protection under short-circuit, thermal overload, and output overvoltage conditions. The LTC3121 is offered in a low profile 12-lead (3mm x 4mm x 0.75mm) DFN package. Typical Application 5V to 12V Synchronous Boost Converter with Output Disconnect 100 6.8H 90 SW 4.7F OFF ON BURST PWM SD LTC3121 PWM/SYNC 100nF CAP RT FB VCC VC SGND 57.6k 4.7F VOUT 12V 400mA VOUT PGND 1.02M 22F 113k 210k 80 10 Burst Mode OPERATION 70 1 PWM 60 50 40 30 0.1 PWM POWER LOSS POWER LOSS (W) VIN EFFICIENCY (%) VIN 5V Efficiency Curve 20 10 10pF 0 0.01 390pF 0.1 10 1 LOAD CURRENT (mA) 100 0.01 600 3121 TA01b 3121 TA01a 3121fa For more information www.linear.com/LTC3121 1 LTC3121 Absolute Maximum Ratings Pin Configuration (Note 1) VIN Voltage ................................................... -0.3V to 6V VOUT Voltage ............................................. -0.3V to 18V SW Voltage (Note 2)................................... -0.3V to 18V SW Voltage (Pulsed < 100ns) (Note 2)........ -0.3V to 19V VC, RT Voltage ........................................... -0.3V to VCC CAP Voltage VOUT < 5.7V.............................-0.3V to (VOUT + 0.3V) 5.7V VOUT 11.7V...... (VOUT - 6V) to (VOUT + 0.3V) VOUT > 11.7V..................................(VOUT - 6V) to 12V All Other Pins................................................ -0.3V to 6V Operating Junction Temperature Range (Notes 3, 4)............................................. -40C to 125C Storage Temperature Range................... -65C to 150C Order Information TOP VIEW SW 1 12 CAP PGND 2 11 VOUT VIN 3 PWM/SYNC 4 VCC RT 13 PGND 10 SGND 9 SD 5 8 FB 6 7 VC DE PACKAGE 12-LEAD (4mm x 3mm) PLASTIC DFN TJMAX = 125C, JA = 43C/W (NOTE 5), JC = 5C/W EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB FOR RATED THERMAL PERFORMANCE http://www.linear.com/product/LTC3121#orderinfo LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3121EDE#PBF LTC3121EDE#TRPBF 3121 12-Lead (4mm x 3mm) Plastic DFN -40C to 125C LTC3121IDE#PBF LTC3121IDE#TRPBF 3121 12-Lead (4mm x 3mm) Plastic DFN -40C to 125C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 2 3121fa For more information www.linear.com/LTC3121 LTC3121 Electrical Characteristics The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25C (Note 3). VIN = 3.6V, VOUT = 12V, RT = 57.6k unless otherwise noted. PARAMETER CONDITIONS Minimum Start-Up Voltage VOUT = 0V l MIN. Input Voltage Range After VOUT 2.2V l 1.7 0.5 Output Voltage Adjust Range l 2.2 Feedback Voltage l 1.178 Feedback Input Current TYP VFB = 1.4V MAX UNITS 1.8 V 5.5 V 15 V 1.202 1.225 V 1 50 nA Quiescent Current, Shutdown VSD = 0V, VOUT = 0V, Not Including Switch Leakage 0.01 1 A Quiescent Current, Active VC = 0V, Measured On VIN, Non-Switching 500 700 A Quiescent Current, Burst Measured on VIN, VFB > 1.4V Measured on VOUT, VFB > 1.4V 25 10 40 20 A A N-channel MOSFET Switch Leakage Current VSW = 15V, VOUT = 15V, VC = 0V (Note 6) l 0.1 30 A P-channel MOSFET Switch Leakage Current VSW = VIN = 0V, VOUT = 15V (Note 6) l 0.1 70 A N-channel MOSFET Switch On-Resistance 0.121 P-channel MOSFET Switch On-Resistance 0.188 N-channel MOSFET Current Limit VIN = 3.3V l 1.5 1.8 Maximum Duty Cycle VFB = 1.0V l 90 94 Minimum Duty Cycle VFB = 1.4V l Switching Frequency 0.85 SYNC Frequency Range l 0.1 PWM/SYNC Input High l 0.9 *VCC PWM/SYNC Input Low l VPWM/SYNC = 5.5V CAP Clamp Voltage VOUT > 6.1V, Referenced to VOUT Error Amplifier Transconductance l SD Input Low SD Input Current Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Voltage transients on the SW pin beyond the DC limit specified in the Absolute Maximum Ratings are non-disruptive to normal operations when using good layout practices, as shown on the demo board or described in the data sheet or application notes. Note 3: The LTC3121 is tested under pulsed load conditions such that TA TJ. The LTC3121E is guaranteed to meet performance specifications from 0C to 85C. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3121I is guaranteed to meet specifications over the full -40C to 125C operating junction MHz 3 MHz V 0.01 1 A -5.2 -5.6 -6.0 V 70 100 130 S 25 A 10 ms 1.6 V l VSD = 5.5V 1.15 V Soft-Start Time l % 0.1*VCC Error Amplifier Output Current SD Input High 1 A % 0 l PWM/SYNC Input Current 2.7 1 0.25 V 2 A temperature range. The junction temperature (TJ in C) is calculated from the ambient temperature (TA in C) and power dissipation (PD in watts) according to the formula: TJ = TA + (PD * JA) where JA is the thermal impedance of the package. Note 4: The LTC3121 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature shutdown is active. Continuous operation above the specified maximum operating junction temperature may result in device degradation or failure. Note 5: Failure to solder the exposed backside of the package to the PC board ground plane will result in a thermal impedance much higher than the rated package specifications. Note 6: Measured using a propietary test mode to ensure anti-ringing switch between VIN ans SW is not active. 3121fa For more information www.linear.com/LTC3121 3 LTC3121 Typical Performance Characteristics Configured as front page application unless otherwise specified. Efficiency vs Load Current, VOUT = 7.5V Efficiency vs Load Current, VOUT = 5V 100 100 90 90 80 80 EFFICIENCY (%) EFFICIENCY (%) 70 60 50 PWM 40 30 20 0 0.01 0.1 1 100 10 LOAD CURRENT (mA) 70 60 50 PWM 40 30 0 0.01 0.1 1 100 10 LOAD CURRENT (mA) PWM Mode Operation ILOAD = 200mA OUTPUT CURRENT 250mA/DIV 40mA 40mA INDUCTOR CURRENT 1A/DIV 3121 G05 2ms/DIV CHANGE IN RDS(ON) FROM 25C (%) -0.3 -0.4 -0.5 140 1.0 3121 G07 60 40 20 0 -20 -40 -50 3121 G06 Oscillator Frequency vs Temperature 80 -0.2 1000 VOUT 5V/DIV CHANGE IN FREQUENCY FROM 25C (%) 0.2 -0.1 1 100 10 LOAD CURRENT (mA) 3121 G03 RDS(ON) vs Temperature, Both NMOS and PMOS 0 0.1 Inrush Current Control 500s/DIV Feedback vs Temperature 0.1 VIN = 5.4V VIN = 4.2V VIN = 2.6V SD 5V/DIV 3121 G04 1s/DIV CHANGE IN VFB FROM 25C (%) 30 0 0.01 1000 400mA INDUCTOR CURRENT 1A/DIV 4 PWM 40 10 VOUT 500mV/DIV AC-COUPLED 40 90 TEMPERATURE (C) 50 Load Transient Response VOUT 20mV/DIV AC-COUPLED -10 60 3121 G02 3121 G01 -0.6 -60 70 20 VIN = 5.4V VIN = 3.8V VIN = 2.3V 10 1000 BURST 80 20 VIN = 4.2V VIN = 3.3V VIN = 0.6V 10 90 BURST EFFICIENCY (%) BURST 100 Efficiency vs Load Current, VOUT = 12V -10 70 110 30 TEMPERATURE (C) 150 3121 G08 0.5 0 -0.5 -1.0 -1.5 -2.0 -60 -10 90 40 TEMPERATURE (C) 140 3121 G09 3121fa For more information www.linear.com/LTC3121 LTC3121 Typical Performance Characteristics Peak Current Limit Change vs Temperature 1.0 0.8 0.6 0.4 0.2 0 0.5 1.5 2.5 3.5 VIN (V) 4.5 70 1 60 0 -1 -2 70 110 30 TEMPERATURE (C) 0 150 Burst Mode No Load Input Current vs VIN 10000 200 150 100 2.5 3.5 VIN , FALLING (V) 4.5 100 10 0.5 5.5 1.5 3121 G13 SD Pin Threshold 2.5 3.5 VIN , FALLING (V) FREQUENCY (MHz) 400mV 1s/DIV 3121 G16 15 0 4 0.5 2 500 0 600 3121 G17 4 -10 70 110 30 TEMPERATURE (C) 150 VOUT = 15V VOUT = 3.6V VOUT = 2.2V 3 10 1.0 400 300 RT (k) 3121 G12 Frequency Accuracy 6 200 6 30 12 1.5 100 5 3121 G15 8 0 4 3121 G14 2.0 0 3 VIN (V) 45 -15 -50 5.5 PERIOD (s) VSD 500mV/DIV 4.5 FREQUENCY PERIOD 2.5 900mV 2 60 Frequency vs RT 3.0 VOUT 5V/DIV 1 75 VOUT = 5V VOUT = 7.5V VOUT = 12V 50 1.5 0 Burst Mode Quiescent Current Change vs Temperature 1000 0 0.5 20 3121 G11 INPUT CURRENT (A) OUTPUT CURRENT (mA) 250 -10 3121 G10 VOUT = 2.2V VOUT = 5V VOUT = 7.5V VOUT = 12V 300 30 10 -4 -50 Burst Mode Maximum Output Current vs VIN 350 40 -3 5.5 VOUT = 5V VOUT = 7.5V VOUT = 12V 50 CHANGE IN CURRENT FROM 25C (%) OUTPUT CURRENT (A) 1.2 2 INPUT CURRENT (mA) VOUT = 5V VOUT = 7.5V VOUT = 12V PWM Operation No Load Input Current vs VIN CHANGE IN FREQUENCY (%) 1.4 PEAK CURRENT LIMIT CHANGE FROM 25C (%) PWM Mode Maximum Output Current vs VIN 2 1 0 -1 -2 -3 -4 0 1 3 2 4 VIN FALLING (V) 5 6 3121 G18 3121fa For more information www.linear.com/LTC3121 5 LTC3121 Typical Performance Characteristics Efficiency vs Frequency 0 100 90 60 50 40 30 20 fOSC = 200kHz fOSC = 1MHz fOSC = 3MHz 10 10 100 OUTPUT CURRENT (mA) 4.0 -2 -3 VCC (V) VCAP, REFERRED TO VOUT (V) 70 EFFICIENCY (%) 4.5 -1 80 0 VCC vs VIN CAP Pin Voltage vs VOUT -4 -5 3.5 3.0 -6 1000 -7 0 2 4 8 6 10 VOUT (V) 3121 G19 12 14 2.5 VIN FALLING VIN RISING 0 3121 G20 Burst Mode Operation to PWM Mode Burst Mode Operation VOUT 100mV/DIV AC-COUPLED 1 2 4 3 VIN (V) 3121 G21 VOUT 100mV/DIV AC-COUPLED VOUT 100mV/DIV AC-COUPLED VPWM/SYNC 2V/DIV INDUCTOR CURRENT 0.5A/DIV VPWM/SYNC 2V/DIV OUTPUT CURRENT = 70mA OUTPUT CURRENT = 70mA OUTPUT CURRENT = 50mA 3121 G22 5s/DIV 20s/DIV Burst Mode Transient 3121 G23 3121 G24 20s/DIV Synchronized Operation Short-Circuit Response SHORT-CIRCUIT APPLIED ILOAD = 100mA VOUT 5V/DIV VOUT 200mV/DIV AC-COUPLED 10mA SHORT-CIRCUIT REMOVED VSW 5V/DIV 100mA VPWM/SYNC 5V/DIV 10mA 200s/DIV 6 6 PWM Mode to Burst Mode Operation VSW 10V/DIV OUTPUT CURRENT 100mA/DIV 5 3121 G25 SYNCHRONIZED TO 1.3MHz 1s/DIV 3121 G26 INDUCTOR CURRENT 1A/DIV 200s/DIV 3121 G27 3121fa For more information www.linear.com/LTC3121 LTC3121 Pin Functions SW (Pin 1): Switch Pin. Connect an inductor from this pin to VIN. Keep PCB trace lengths as short and wide as possible to reduce EMI and voltage overshoot. When VOUT VIN + 2V, an internal anti-ringing resistor is connected between SW and VIN after the inductor current has dropped to near zero, to minimize EMI. The anti-ringing resistor is also activated in shutdown and during the sleep periods of Burst Mode operation. VCC (Pin 5): VCC Regulator Output. Connect a low-ESR filter capacitor of at least 4.7F from this pin to GND to provide an internal regulated rail approximately equal to the lower of VIN and 4.25V. When VOUT is higher than VIN, and VIN falls below 3V, VCC will regulate to the lower of approximately VOUT and 4.25V. A UVLO event occurs if VCC drops below 1.6V. Switching is inhibited, and a soft-start is initiated when VCC returns above 1.7V. PGND (Pin 2, Exposed Pad Pin 13): Power Ground. When laying out your PCB, provide a short, direct path between PGND and the output capacitor and tie directly to the ground plane. The exposed pad is ground and must be soldered to the PCB ground plane for rated thermal performance. RT (Pin 6): Frequency Adjust Pin. Connect an external resistor (RT) from this pin to SGND to program the oscillator frequency according to the formula: VIN (Pin 3): Input Supply Pin. The device is powered from VIN unless VOUT exceeds VIN and VIN is less than 3V. Place a low ESR ceramic bypass capacitor of at least 4.7F from VIN to PGND. X5R and X7R dielectrics are preferred for their superior voltage and temperature characteristics. PWM/SYNC (Pin 4): Burst Mode Operation Select and Oscillator Synchronization. Do not leave this pin floating. * PWM/SYNC = High. Disable Burst Mode Operation and maintain low noise, constant frequency operation. * PWM/SYNC = Low. The converter operates in Burst Mode operation, independent of load current. * PWM/SYNC = External CLK. The internal oscillator is synchronized to the external CLK signal. Burst Mode operation is disabled. A clock pulse width between 100ns and 2s is required to synchronize the oscillator. An external resistor must be connected between RT and GND to program the oscillator slightly below the desired synchronization frequency. In non-synchronized applications, repeated clocking of the PWM/SYNC pin to affect an operating mode change is supported with these restrictions: * Boost Mode (VOUT > VIN): IOUT <500A: PWM/SYNC 100Hz, IOUT 500A: PWM/SYNC 5kHz * Buck Mode (VOUT < VIN): IOUT <5mA: PWM/SYNC 5Hz, IOUT 5mA: PWM/SYNC 5kHz RT = 57.6/OSC where OSC is in MHz and RT is in k. VC (Pin 7): Error Amplifier Output. A frequency compensation network is connected to this pin to compensate the control loop. See Compensating the Feedback Loop section for guidelines. FB (Pin 8): Feedback Input to the Error Amplifier. Connect the resistor divider tap to this pin. Connect the top of the divider to VOUT and the bottom of the divider to SGND. The output voltage can be adjusted from 2.2V to 15V according to this formula: VOUT = 1.202V * (1 + R1/R2) SD (Pin 9): Logic Controlled Shutdown Input. Bringing this pin above 1.6V enables normal, free-running operation, forcing this pin below 0.25V shuts the LTC3121 down, with quiescent current below 1A. Do not leave this pin floating. SGND (Pin 10): Signal Ground. When laying out a PC board, provide a short, direct path between SGND and the (-) side of the output capacitor. VOUT (Pin 11): Output Voltage Sense and the Source of the Internal Synchronous Rectifier MOSFET. Driver bias is derived from VOUT. Connect the output filter capacitor from VOUT to PGND, as close to the IC as possible. A minimum value of 10F ceramic is recommended. VOUT is disconnected from VIN when SD is low. CAP (Pin 12): Serves as the Low Reference for the Synchronous Rectifier Gate Drive. Connect a low ESR filter capacitor (typically 100nF) from this pin to VOUT to provide an elevated ground rail, approximately 5.6V below VOUT, used to drive the synchronous rectifier. 3121fa For more information www.linear.com/LTC3121 7 LTC3121 Block Diagram 1 BULK CONTROL SIGNALS SW VIN ANTI-RING L1 VIN 1.8V TO 5.5V 3 COUT TSD VREF_UP OSC SD OVLO SD SHUTDOWN PWM LOGIC AND DRIVERS + - CURRENT SENSE SD PWM/SYNC PWM BURST SYNC CONTROL + - IZERO COMP OVLO -+ - - + VC 5 LDO 6 FB CPL 8 1.202V VC R2 OSC REFERENCE UVLO SOFT-START VC CLAMP RC CC VREF_UP 1.202V RT THERMAL SD 7 CF SD TSD OVLO CVCC 4.7F OSCILLATOR RPL 12 gm ERROR AMPLIFIER ILIM REF ADAPTIVE SLOPE COMPENSATION VCC CAP R1 VIN VOUT VBEST C1 100nF 16.2V PGND 4 VOUT 2.2V TO 15V 11 VIN CIN 9 VOUT TSD RT SGND 10 PGND 2 EXPOSED PAD 13 LTC3121 3121 BD THE VALUES OF RC, CC, AND CF ARE BASED UPON OPERATING CONDITIONS. PLEASE REFER TO COMPENSATING THE FEEDBACK LOOP SECTION FOR GUIDELINES TO DETERMINE OPTIMAL VALUES OF THESE COMPONENTS. 8 3121fa For more information www.linear.com/LTC3121 LTC3121 Operation The LTC3121 is an adjustable frequency, 100kHz to 3MHz synchronous boost converter housed in a 12-lead 4mm x 3mm DFN. The LTC3121 offers the unique ability to startup and regulate the output from inputs as low as 1.8V and continue to operate from inputs as low as 0.5V. Output voltages can be programmed between 2.2V and 15V. The device also features fixed frequency, current mode PWM control for exceptional line and load regulation. The current mode architecture with adaptive slope compensation provides excellent transient load response and requires minimal output filtering. An internal 10ms closed loop soft-start simplifies the design process while minimizing the number of external components. With its low RDS(ON) and low gate charge internal N-channel MOSFET switch and P-channel MOSFET synchronous rectifier, the LTC3121 achieves high efficiency over a wide range of load current. High efficiency is achieved at light loads when Burst Mode operation is commanded. Operation can be best understood by referring to the Block Diagram. Low Voltage Operation The LTC3121 is designed to allow start-up from input voltages as low as 1.8V. When VOUT exceeds 2.2V, the LTC3121 continues to regulate its output, even when VIN falls to as low as 0.5V. The limiting factors for the application become the availability of the input source to supply sufficient power to the output at the low voltages, and the maximum duty cycle. Note that at low input voltages, small voltage drops due to series resistance become critical and greatly limit the power delivery capability of the converter. This feature extends operating times by maximizing the amount of energy that can be extracted from the input source. zero to its final programmed value. This limits the inrush current drawn from the input source. As a result, the duration of the soft-start is largely unaffected by the size of the output capacitor or the output regulation voltage. The closed loop nature of the soft-start allows the converter to respond to load transients that might occur during the soft-start interval. The soft-start period is reset by a shutdown command on SD, a UVLO event on VCC (VCC < 1.6V), an overvoltage event on VOUT (VOUT 16.2V), or an overtemperature event (thermal shutdown is invoked when the die temperature exceeds 170C). Upon removal of these fault conditions, the LTC3121 will soft-start the output voltage. Error Amplifier The non-inverting input of the transconductance error amplifier is internally connected to the 1.202V reference and the inverting input is connected to FB. An external resistive voltage divider from VOUT to ground programs the output voltage from 2.2V to 15V via FB as shown in Figure 1. R1 VOUT = 1.202V 1+ R2 Selecting an R2 value of 121k to have approximately 10A of bias current in the VOUT resistor divider yields the formula: R1 = 100.67*(VOUT - 1.202V) where R1 is in k. Power converter control loop compensation is set by a simple RC network between VC and ground. VOUT LTC3121 Low Noise Fixed Frequency Operation R1 + - Soft-Start The LTC3121 contains internal circuitry to provide closedloop soft-start operation. The soft-start utilizes a linearly increasing ramp of the error amplifier reference voltage from zero to its nominal value of 1.202V in approximately 10ms, with the internal control loop driving VOUT from FB 1.202V R2 3121 F01 Figure 1. Programming the Output Voltage 3121fa For more information www.linear.com/LTC3121 9 LTC3121 Operation Internal Current Limit The current limit comparator shuts off the N-channel MOSFET switch once its threshold is reached. Peak switch current is limited to 1.8A, independent of input or output voltage, except when VOUT is below 1.5V, resulting in the current limit being approximately half of the nominal peak. Lossless current sensing converts the peak current signal of the N-channel MOSFET switch into a voltage that is summed with the internal slope compensation. The summed signal is compared to the error amplifier output to provide a peak current control command for the PWM. Zero Current Comparator The zero current comparator monitors the inductor current being delivered to the output and shuts off the synchronous rectifier when this current reduces to approximately 50mA. This prevents the inductor current from reversing in polarity, improving efficiency at light loads. Oscillator The internal oscillator is programmed to the desired switching frequency with an external resistor from the RT pin to GND according to the following formula: 57.6 OSC (MHz) = RT (k) The oscillator also can be synchronized to an external frequency by applying a pulse train to the PWM/SYNC pin. An external resistor must be connected between RT and GND to program the oscillator to a frequency approximately 25% below that of the externally applied pulse train used for synchronization. RT is selected in this case according to this formula: 73.2 RT (k) = SYNC (MHz) Output Disconnect The LTC3121's output disconnect feature eliminates body diode conduction of the internal P-channel MOSFET rectifier. This allows for VOUT to discharge to 0V during 10 shutdown, and draw no current from the input source. It also allows for inrush current limiting at turn-on, minimizing surge currents seen by the input supply. Note that to obtain the advantages of output disconnect, there must not be an external Schottky diode connected between SW and VOUT. The output disconnect feature also allows VOUT to be pulled high, without reverse current being backfed into the power source connected to VIN. Shutdown The boost converter is disabled by pulling SD below 0.25V and enabled by pulling SD above 1.6V. Note that SD can be driven above VIN or VOUT, as long as it is limited to less than the absolute maximum rating. Thermal Shutdown If the die temperature exceeds 170C typical, the LTC3121 will go into thermal shutdown (TSD). All switches will be turned off until the die temperature drops by approximately 7C, when the device re-initiates a soft-start and switching can resume. Boost Anti-Ringing Control When VOUT VIN + 2V, the anti-ringing control connects a resistor across the inductor to damp high frequency ringing on the SW pin during discontinuous current mode operation when the inductor current has dropped to near zero. Although the ringing of the resonant circuit formed by L and CSW (capacitance on SW pin) is low energy, it can cause EMI radiation. VCC Regulator An internal low dropout regulator generates the 4.25V (nominal) VCC rail from VIN or VOUT, depending upon operating conditions. VCC is supplied from VIN when VIN is greater than 3.5V, otherwise the greater of VIN and VOUT is used. The VCC rail powers the internal control circuitry and power MOSFET gate drivers of the LTC3121. The VCC regulator is disabled in shutdown to reduce quiescent current and is enabled by forcing the SD pin above its threshold. A 4.7F or larger capacitor must be connected between VCC and SGND. 3121fa For more information www.linear.com/LTC3121 LTC3121 Operation An overvoltage condition occurs when VOUT exceeds approximately 16.2V. Switching is disabled and the internal soft-start ramp is reset. Once VOUT drops below approximately 15.6V, a soft-start cycle is initiated and switching is enabled. If the boost converter output is lightly loaded so that the time constant product of the output capacitance, COUT, and the output load resistance, ROUT is near or greater than the soft-start time of approximately 10ms, the soft-start ramp may end before or soon after switching resumes, defeating the inrush current limiting of the closed loop soft-start following an overvoltage event. Short-Circuit Protection The LTC3121 output disconnect feature allows output short-circuit protection. To reduce power dissipation under overload and short-circuit conditions, the peak switch current limit is reduced to 1A. Once VOUT > 1.5V, the current limit is set to its nominal value of 1.8A. VIN > VOUT Operation The LTC3121 step-up converter will maintain voltage regulation even when the input voltage is above the desired output voltage. Note that operating in this mode will exhibit lower efficiency and a reduced output current capability. Refer to the Typical Performance Characteristics section for details. Burst Mode Operation When the PWM/SYNC pin is held low, the boost converter operates in Burst Mode operation to improve efficiency at light loads and reduce standby current at no load. The input thresholds for this pin are determined relative to VCC with a low being less than 10% of VCC and a high being greater than 90% of VCC. The LTC3121 will operate in fixed frequency PWM mode even if Burst Mode operation is commanded during soft-start. In Burst Mode operation, the LTC3121 switches asynchronously. The inductor current is first charged to 600mA by turning on the N-channel MOSFET switch. Once this current threshold is reached, the N-channel is turned off and the P-channel synchronous switch is turned on, delivering current to the output. When the inductor current discharges to approximately zero, the cycle repeats. In Burst Mode operation, energy is delivered to the output until the nominal regulation value is reached, at which point the LTC3121 transitions to sleep mode. In sleep, the output switches are turned off and the LTC3121 consumes only 25A of quiescent current. When the output voltage droops approximately 1%, switching resumes. This maximizes efficiency at very light loads by minimizing switching and quiescent losses. Output voltage ripple in Burst Mode operation is typically 1% to 2% peak-to-peak. Additional output capacitance (10F or greater), or the addition of a small feed-forward capacitor (10pF to 50pF) connected between VOUT and FB can help further reduce the output ripple. The maximum output current (IOUT) capability in Burst Mode operation varies with VIN and VOUT, as shown in Figure 2. 350 300 OUTPUT CURRENT (mA) Overvoltage Lockout 250 VOUT = 2.2V VOUT = 5V VOUT = 7.5V VOUT = 12V 200 150 100 50 0 0.5 1.5 2.5 3.5 VIN, FALLING (V) 4.5 5.5 3121 F02 Figure 2. Burst Mode Maximum Output Current vs VIN 3121fa For more information www.linear.com/LTC3121 11 LTC3121 Applications Information PCB Layout Guidelines Component Selection The high switching frequency of the LTC3121 demands careful attention to board layout. A careless layout will result in reduced performance. Maximizing the copper area for ground will help to minimize die temperature rise. A multilayer board with a separate ground plane is ideal, but not absolutely necessary. See Figure 3 for an example of a two-layer board layout. PGND PGND CAP SW VIN 1 12 2 11 3 4 VCC RT 13 PGND VOUT 9 8 6 7 The LTC3121 can utilize small surface mount inductors due to its capability of setting a fast (up to 3MHz) switching frequency. Larger values of inductance will allow slightly greater output current capability by reducing the inductor ripple current. To design a stable converter the range of inductance values is bounded by the targeted magnitude of the internal slope compensation and is inversely proportional to the switching frequency. The inductor selection for the LTC3121 has the following bounds: 10 3 H > L > H f f The inductor peak-to-peak ripple current is given by the following equation: 10 SGND 5 Inductor Selection FB V * (VOUT - VIN) RIPPLE(A) = IN f * L * VOUT VC where: L = Inductor Value in H f = Switching Frequency in MHz 3121 F02 Figure 3. Traces Carrying High Current Are Direct (PGND, SW, VIN and VOUT). Trace Area at FB and VC Are Kept Low. Trace Length to Input Supply Should Be Kept Short. VIN and VOUT Ceramic Capacitors Should Be Placed as Close to the LTC3121 Pins as Possible Schottky Diode Although it is not required, adding a Schottky diode from SW to VOUT can improve the converter efficiency by about 4%. Note that this defeats the output disconnect and shortcircuit protection features of the LTC3121. 12 The inductor ripple current is a maximum at the minimum inductor value. Substituting 3/f for the inductor value in the above equation yields the following: V * (VOUT - VIN) RIPPLEMAX(A) = IN 3 * VOUT To realize greater output current capability at the guaranteed minimum (over temperature) 1.5A current limit, it is recommended that the inductor ripple current be limited to one-third of this minimum value, or to approximately 0.5A. Choosing a minimum inductor value of 6/f H (where f = switching frequency in MHz) or greater typically results in an inductor ripple current of 0.5A or less for the majority of step-up ratios. High frequency ferrite core inductor materials reduce frequency dependent power losses compared to cheaper powdered iron types, improving efficiency. 3121fa For more information www.linear.com/LTC3121 LTC3121 Applications Information The inductor should have low DCR (series resistance of the windings) to reduce the I2R power losses, and must be able to support the peak inductor current without saturating. Molded chokes and most chip inductors usually do not have enough core area to support the peak inductor currents of 2A to 3A seen on the LTC3121. To minimize radiated noise, use a shielded inductor. See Table 1 for suggested components and suppliers Table 1. Recommended Inductors PART NUMBER VALUE DCR ISAT (H) (m) (A) SIZE (mm) WxLxH Coilcraft XAL4020-222ME Coilcraft XAL4030-332ME Coilcraft XAL4030-472ME Coilcraft XAL5050-682ME Coilcraft XAL6060-223ME Coilcraft MSS1260T-333ML 2.2 3.3 4.7 6.8 22 33 39 29 44 29 61 57 5.6 4.3 x 4.3 x 2.1 5.5 4.3 x 4.3 x 3.1 4.5 4.3 x 4.3 x 3.1 6.0 5.3 x 5.3 x 5.1 5.6 6.3 x 6.3 x 6.1 4.34 12.3 x 12.3 x 6.2 Coiltronics DR73-2R2-R Coiltronics DR74-4R7-R Coiltronics DR125-330-R Coiltronics DR127-470-R 2.2 4.7 33 47 17 25 51 72 5.52 4.37 3.84 5.28 7.6 x 7.6 x 3.55 7.6 x 7.6 x 4.35 12.5 x 12.5 x 6 12.5 x 12.5 x 8 Sumida CDR7D28MNNP-2R2NC Sumida CDR7D28MNNP-6R8NC 2.2 6.8 18 46 4.9 3.5 7.6 x 7.6 x 3 7.6 x 7.6 x 3 Taiyo-Yuden NR5040T3R3N 3.3 35 3.8 5x5x4 TDK LTF5022T-2R2N3R2-LC TDK SPM6530T-3R3M TDK VLP8040T-4R7M 2.2 3.3 4.7 40 30 25 3.2 6.8 4.4 5 x 5.2 x 2.2 7.1 x 6.5 x 3 8 x 7.7 x 4 Wurth WE-PD7447789002 Wurth WE-PD7447789003 Wurth WE-PD7447789003 Wurth WE-PD7447779006 Wurth WE-HCI7443251000 Wurth WE-PD744770122 Wurth WE-PD744770133 Wurth WE-PD7447709470 2.2 3.3 4.7 6.8 10 22 33 47 23 30 35 35 16 43 64 60 4.8 4.2 4.2 3.3 8.5 5 3.6 4.5 7.3 x 7.3 x 3.2 7.3 x 7.3 x 3.2 7.3 x 7.3 x 3.2 7.3 x 7.3 x 4.5 10 x 10 x 5 12 x 12 x 8 12 x 12 x 8 12 x 12 x 10 Output and Input Capacitor Selection Low ESR (equivalent series resistance) capacitors should be used to minimize the output voltage ripple. Multilayer ceramic capacitors are an excellent choice as they have extremely low ESR and are available in small footprints. X5R and X7R dielectric materials are preferred for their ability to maintain capacitance over wide voltage and temperature ranges. Y5V types should not be used. Although ceramic capacitors are recommended, low ESR tantalum capacitors may be used as well. When selecting output capacitors, the magnitude of the peak inductor current, together with the ripple voltage specification, determine the choice of the capacitor. Both the ESR (equivalent series resistance) of the capacitor and the charge stored in the capacitor each cycle contribute to the output voltage ripple. The ripple due to the charge is approximately: VRIPPLE(CHARGE) IP * VIN COUT * VOUT * where IP is the peak inductor current. The ESR of COUT is usually the most dominant factor for ripple in most power converters. The ripple due to the capacitor ESR is: V VRIPPLE(ESR) = ILOAD * RESR * OUT VIN where RESR = capacitor equivalent series resistance. The input filter capacitor reduces peak currents drawn from the input source and reduces input switching noise. A low ESR bypass capacitor with a value of at least 4.7F should be located as close to the VIN pin as possible. Low ESR and high capacitance are critical to maintain low output voltage ripple. Capacitors can be used in parallel for even larger capacitance values and lower effective ESR. Ceramic capacitors are often utilized in switching converter applications due to their small size, low ESR and low leakage currents. However, many ceramic capacitors experience significant loss in capacitance from their rated value with increased DC bias voltage. It is not uncommon for a small surface mount capacitor to lose more than 50% of its rated capacitance when operated near its rated voltage. As a result it is sometimes necessary to use a larger capacitor value or a capacitor with a larger value and case size, such as 1812 rather than 1206, in order to actually realize the intended capacitance at the full operating voltage. Be sure to consult the vendor's curve of capacitance vs DC bias voltage. Table 2 shows a sampling of capacitors suited for LTC3121 applications. 3121fa For more information www.linear.com/LTC3121 13 LTC3121 Applications Information Operating Frequency Selection Table 2. Representative Output Capacitors MANUFACTURER, PART NUMBER VALUE (F) VOLTAGE (V) SIZE L x W x H (mm) TYPE, ESR (m) AVX, 12103D226MAT2A 22 25 3.2 x 2.5 x 2.79, X5R Ceramic Kemet, C2220X226K3RACTU 22 25 5.7 x 5.0 x 2.4, X7R Ceramic Kemet, A700D226M016ATE030 22 16 7.3 x 4.3 x 2.8, Alum. Polymer, 30m Murata, GRM32ER71E226KE15L 22 25 3.2 x 2.5 x 2.5, X7R Ceramic Nichicon, PLV1E121MDL1 82 25 8 x 8 x 12, Alum. Polymer, 25m Panasonic, ECJ-4YB1E226M 22 25 3.2 x 2.5 x 2.5, X5R Ceramic Sanyo, 25TQC22MV 22 25 7.3 x 4.3 x 3.1, POSCAP, 50m Sanyo, 16TQC100M 100 16 7.3 x 4.3 x 1.9, POSCAP, 45m Sanyo, 25SVPF47M 47 25 6.6 x 6.6 x 5.9, OS-CON, 30m Taiyo Yuden, TMK325BJ226MM-T 22 25 3.2 x 2.5 x 2.5, X5R Ceramic TDK, CKG57NX5R1E476M 47 25 6.5 x 5.5 x 5.5, X5R Ceramic Cap-XX GS230F 1.2Farads 4.5 39 x 17 x 3.8 28m Cooper A1030-2R5155 1.5Farads 2.5 O = 10, L = 30 60m Maxwell BCAP0050-P270 50Farads 2.5 O = 18, L = 40 20m For applications requiring a very low profile and very large capacitance, the GS, GS2 and GW series from Cap-XX and PowerStor Aerogel Capacitors from Cooper all offer very high capacitance and low ESR in various low profile packages. A method for improving the converter's transient response uses a small feed-forward series network of a capacitor and a resistor across the top resistor of the feedback divider (from VOUT to FB). This adds a phase-lead zero and pole to the transfer function of the converter as calculated in the Compensating the Feedback Loop section. 14 There are several considerations in selecting the operating frequency of the converter. Typically the first consideration is to stay clear of sensitive frequency bands, which cannot tolerate any spectral noise. For example, in products incorporating RF communications, the 455kHz IF frequency is sensitive to any noise, therefore switching above 600kHz is desired. Some communications have sensitivity to 1.1MHz and in that case a 1.5MHz switching converter frequency may be employed. A second consideration is the physical size of the converter. As the operating frequency is increased, the inductor and filter capacitors typically can be reduced in value, leading to smaller sized external components. The smaller solution size is typically traded for efficiency, since the switching losses due to gate charge increase with frequency. Another consideration is whether the application can allow pulse-skipping. When the boost converter pulse-skips, the minimum on-time of the converter is unable to support the duty cycle. This results in a low frequency component to the output ripple. In many applications where physical size is the main criterion, running the converter in this mode is acceptable. In applications where it is preferred not to enter this mode, the maximum operating frequency is given by: VOUT - VIN MAX _ NOSKIP Hz VOUT * tON(MIN) where tON(MIN) = minimum on-time = 100ns. Thermal Considerations For the LTC3121 to deliver its full power, it is imperative that a good thermal path be provided to dissipate the heat generated within the package. This can be accomplished by taking advantage of the large thermal pad on the underside of the IC. It is recommended that multiple vias in the printed circuit board be used to conduct heat away from the IC and into a copper plane with as much area as possible. If the junction temperature rises above ~170C, the part will go into thermal shutdown, and all switching will stop until the temperature drops approximately 7C. 3121fa For more information www.linear.com/LTC3121 LTC3121 Applications Information Compensating the Feedback Loop The LTC3121 uses current mode control, with internal adaptive slope compensation. Current mode control eliminates the second order filter due to the inductor and output capacitor exhibited in voltage mode control, and simplifies the power loop to a single pole filter response. Because of this fast current control loop, the power stage of the IC combined with the external inductor can be modeled by a transconductance amplifier gmp and a current controlled current source. Figure 4 shows the key equivalent small signal elements of a boost converter. The DC small-signal loop gain of the system shown in Figure 4 is given by the following equation: GBOOST = GEA * GMP * GPOWER * R2 R1+ R2 where GEA is the DC gain of the error amplifier, GMP is the modulator gain, and GPOWER is the inductor current to VOUT gain. GEA = g ma * RO 950V/V (Not Adjustable; g ma = 95S, RO 10M) GMP = g mp = GPOWER = IL VC 3.4S (Not Adjustable) VOUT * VIN = IL 2 *IOUT Combining the two equations above yields: GDC = GMP * GPOWER 1.7 * * VIN IOUT Converter efficiency will vary with IOUT and switching frequency OSC as shown in the typical performance characteristics curves. Output Pole: P1 = 2 Hz 2 * * RL * COUT Error Amplifier Pole: P2 = 1 Hz 2 * * RO * (CC + CF ) Error Amplifier Zero: Z1 = 1 Hz 2 * * RC * CC - + gmp IL VOUT * VIN *I 2 * VOUT L COUT MODULATOR RPL 1.202V REFERENCE CF gma RC CC RO ERROR AMPLIFIER - RL ESR Zero: Z2 = CPL + VC RESR 1 2 * * RESR * COUT VIN2 * RL R1 RHP Zero: Z3 = R2 High Frequency Pole: P3 > FB 2 * * VOUT 2 * L 3121 F04 CC: COMPENSATION CAPACITOR COUT: OUTPUT CAPACITOR CPL: PHASE LEAD CAPACITOR CF : HIGH FREQUENCY FILTER CAPACITOR gma: TRANSCONDUCTANCE AMPLIFIER INSIDE IC gmp: POWER STAGE TRANSCONDUCTANCE AMPLIFIER RC: COMPENSATION RESISTOR RL: OUTPUT RESISTANCE DEFINED AS VOUT/ILOADMAX RO: OUTPUT RESISTANCE OF gma RPL: PHASE LEAD RESISTOR R1, R2: FEEDBACK RESISTOR DIVIDER NETWORK RESR: OUTPUT CAPACITOR ESR : CONVERTER EFFICIENCY (~90% AT HIGHER CURRENTS) Figure 4. Boost Converter Equivalent Model V/V Phase Lead Zero: Z4 = Hz Hz OSC 3 1 Hz 2 * * (R1+RPL ) * CPL 1 Hz R1* R2 +RPL * CPL 2* * R1+R2 Error Amplifier Filter Pole: Phase Lead Pole: P4 = P5 = 1 Hz C *C F C 2 * * RC * CC + CF 3121fa For more information www.linear.com/LTC3121 15 LTC3121 Applications Information The current mode zero (Z3) is a right half plane zero which can be an issue in feedback control design, but is manageable with proper external component selection. As a general rule, the frequency at which the open-loop gain of the converter is reduced to unity, known as the crossover frequency C, should be set to less than one third of the right half plane zero (Z3), and under one eighth of the switching frequency OSC. Once C is selected, the values for the compensation components can be calculated using a bode plot of the power stage or two generally valid assumptions: P1 dominates the gain of the power stage for frequencies lower than C and C is much higher than P2. First calculate the power stage gain at C, GC in V/V. Assuming the output pole P1 dominates GC for this range, it is expressed by: GC GDC 2 1+ C P1 V/V Decide how much phase margin (m) is desired. Greater phase margin can offer more stability while lower phase margin can yield faster transient response. Typically, m 60 is optimal for minimizing transient response time while allowing sufficient margin to account for component variability. 1 is the phase boost of Z1, P2, and P5 while 2 is the phase boost of Z4 and P4. Select 1 and 2 such that V 1 74 ; 2 2 * tan-1 OUT - 90 and 1.2V 1 + 2 = m + tan-1 C Z3 where VOUT is in V and C and Z3 are in kHz. Setting Z1, P5, Z4, and P4 such that Z1= C , P5 = C a1, Z4 = C , P4 = C a2 a1 a2 The compensation will force the converter gain GBOOST to unity at C by using the following expression for CC: CC = 103 * g ma * R2 * GC ( a1 - 1) a2 2 * C * (R1+ R2) a1 pF (gma in S, C in kHz, GC in V/V) Once CC is calculated, RC and CF are determined by: 106 * a1 RC = k (C in kHz, C C in pF) 2 * C * CC CF = CC a1 - 1 The values of the phase lead components are given by the expressions: RPL R1* R2 R1- a2 * R1+R2 = k and a2 - 1 CPL = 106 ( a2 - 1) (R1+R2) 2 * C * R12 a2 pF where R1, R2, and RPL are in k and C is in kHz. Note that selecting 2 = 0 forces a2 = 1, and so the converter will have Type II compensation and therefore no feedforward: RPL is open (infinite impedance) and CPL = 0pF. If a2 = 0.833 * VOUT (its maximum), feedforward is maximized; RPL = 0 and CPL is maximized for this compensation method. Once the compensation values have been calculated, obtaining a converter bode plot is strongly recommended to verify calculations and adjust values as required. Using the circuit in Figure 5 as an example, Table 3 shows the parameters used to generate the bode plot shown in Figure 6. allows a1 and a2 to be determined using 1 and 2 + 90 2 +90 a1 = tan2 1 , a2 = tan 2 2 2 16 3121fa For more information www.linear.com/LTC3121 LTC3121 Applications Information VIN 5V L1 6.8H SW VIN CIN 4.7F SD OFF ON LTC3121 PWM/SYNC BURST PWM C1 100nF FB VCC VC SGND R1 1.02M CAP RT RT 57.6k VOUT 12V 400mA VOUT COUT 22F R2 113k RC 210k PGND CF 10pF CC 390pF CVCC 4.7F 3121 F05a Transient Response with 200mA to 400mA Load Step Switching Waveforms with 400mA Load VOUT 100mV/DIV AC-COUPLED VOUT 100mV/DIV AC-COUPLED SW 10V/DIV OUTPUT CURRENT 200mA/DIV INDUCTOR CURRENT 1A/DIV 3121 F05b 100s/DIV 3121 F05c Figure 5. 1MHz, 5V to 12V, 400mA Boost Converter 170 180 150 140 PHASE 130 100 110 60 90 20 70 -20 50 -60 30 -100 GAIN 10 -140 -180 -10 -30 0.01 PHASE (deg) GAIN (dB) 200ns/DIV 0.1 10 1 FREQUENCY (kHz) 100 -220 1000 3121 F06 Figure 6. Bode Plot for Example Converter 3121fa For more information www.linear.com/LTC3121 17 LTC3121 Applications Information From Figure 6, the phase is 60 when the gain reaches 0dB, so the phase margin of the converter is 60. The crossover frequency is 15kHz, which is more than three times lower than the 121.3kHz frequency of the RHP zero to achieve adequate phase margin. Table 3. Bode Plot Parameters for Type II Compensation PARAMETER VALUE UNITS COMMENT VIN 5 V App Specific VOUT 12 V App Specific RL 30 App Specific COUT 22 F App Specific RESR L 5 m App Specific 6.8 H App Specific 1 MHz Adjustable R1 1020 k Adjustable R2 113 k Adjustable gma 95 S Fixed RO 10 M Fixed gmp 3.4 S Fixed 92 % App Specific RC 210 k Adjustable CC 390 pF Adjustable CF 10 pF Adjustable RPL Open k Optional CPL 0 pF Optional OSC VIN 5V L1 6.8H SW VIN CIN 4.7F The circuit in Figure 7 shows the same application as that in Figure 5 with Type III compensation. This is accomplished by adding CPL and RPL and adjusting CC, CF, and RC accordingly. Table 4 shows the parameters used to generate the bode plot shown in Figure 8. SD OFF ON BURST PWM C1 100nF LTC3121 PWM/SYNC VOUT 12V 400mA VOUT CAP RPL 604k CPL 10pF VCC CVCC 4.7F COUT 22F FB RT RT 57.6k R1 1.02M SGND PGND VC RC 127k CC 220pF CF 33pF R2 113k 3121 F06 Figure 7. Boost Converter with Phase Lead 18 3121fa For more information www.linear.com/LTC3121 LTC3121 Applications Information Table 4. Bode Plot Parameters for Type III Compensation VALUE UNITS COMMENT VIN 5 V App Specific VOUT 12 V App Specific RL 30 App Specific COUT 22 F App Specific 170 RESR L OSC 5 m App Specific 150 H App Specific 130 1 MHz Adjustable 110 60 1020 k Adjustable 90 20 70 -20 R2 113 k Adjustable gma 95 S Fixed RO 10 M Fixed gmp 3.4 S Fixed 92 % App Specific RC 127 k Adjustable CC 220 pF Adjustable CF 33 pF Adjustable RPL 604 k Adjustable CPL 10 pF Adjustable 140 100 PHASE 50 -60 GAIN 30 -100 10 -140 -10 -180 -30 0.01 0.1 10 1 FREQUENCY (kHz) PHASE (deg) R1 180 6.8 GAIN (dB) PARAMETER From Figure 8, the phase margin is still optimized at 60 and the crossover frequency remains 15kHz. Adding CPL and RPL provides some feedforward signal in Burst Mode operation, leading to lower output voltage ripple. 100 -220 1000 3121 F08 Figure 8. Bode Plot Showing Phase Lead 3121fa For more information www.linear.com/LTC3121 19 LTC3121 Typical Applications 3.3V to 12V, 2MHz Synchronous Boost Converter with Output Disconnect, 250mA L1 3.3H VIN 3.3V 100 SW C1 100nF RT FB VCC VC RT 28k SGND R1 1.02M CAP PWM/SYNC BURST PWM 80 COUT 22F R2 113k RC 280k PGND CVCC 4.7F Burst Mode OPERATION 70 60 50 1 PWM 40 30 20 CF 10pF CC 220pF 10 POWER LOSS (W) CIN 4.7F LTC3121 SD OFF ON VOUT 12V 250mA VOUT EFFICIENCY (%) VIN 90 10 0 0.01 CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 22F, 16V, X7R, 1812 L1: COILCRAFT XAL5050-332ME 3121 TA02a PWM POWER LOSS 0.1 1 10 LOAD CURRENT (mA) 0.1 100 3121 TA02b Single Li-Cell to 6V, 2.5W, 3MHz Synchronous Boost Converter for RF Transmitter VIN 2.5V TO 4.2V L1 2.2H SW VIN CIN 4.7F OFF ON SD LTC3121 PWM/SYNC C1 100nF FB VCC VC SGND CVCC 4.7F PGND VIN = 3.6V VOUT 200mV/DIV AC-COUPLED R1 487k CAP RT RT 17.4k VOUT 6V 425mA VOUT COUT 47F RC 137k CC 150pF 420mA R2 121k CF 12pF OUTPUT CURRENT 200mA/DIV 40mA 100s/DIV CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 47F, 10V, X7R, 1812 L1: COILCRAFT XAL5030-222ME 20 3121 TA03b 3121 TA03a 3121fa For more information www.linear.com/LTC3121 LTC3121 Typical Applications 2 AA Cell to 12V Synchronous Boost Converter, 100mA L1 6.8H 1.2 VIN CIN 4.7F VOUT C1 100nF RT FB VCC VC SGND 1.0 R1 1.02M CAP PWM/SYNC RT 57.6k VOUT 12V 100mA COUT 22F R2 113k RC 200k PGND CVCC 4.7F CC 560pF 70 60 0.8 50 40 0.6 30 20 0.4 CF 10pF 0.2 1.6 CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 22F, 16V, X7R, 1812 L1: COILCRAFT XAL5050-682ME 80 EFFICIENCY (%) LTC3121 SD OFF ON 100 90 SW INPUT CURRENT (A) VIN 1.8V TO 3V EFFICIENCY INPUT CURRENT 1.8 2 2.2 3121 TA04a 2.4 2.6 VIN (V) 10 2.8 0 3.2 3 3121 TA04b 3.3V to 12V, 300kHz Synchronous Boost Converter with Output Disconnect, 250mA L1 22H 100 90 SW OFF ON CIN 4.7F BURST PWM SD LTC3121 PWM/SYNC C1 100nF R1 1.02M CAP COUT 68F FB RT SGND CVCC 4.7F CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 68F, 16V, X7R, 1812 L1: COILCRAFT XAL6060-223ME PGND RC 154k CC 1.2nF 80 R2 113k VC VCC RT 196k VOUT 12V 250mA VOUT 10 Burst Mode OPERATION 70 1 POWER LOSS (W) VIN EFFICIENCY (%) VIN 3.3V 60 50 PWM 40 0.1 30 20 CF 56pF PWM POWER LOSS 10 0 0.01 3121 TA05a 0.1 10 1 LOAD CURRENT (mA) 0.01 100 3121 TA05b 3121fa For more information www.linear.com/LTC3121 21 LTC3121 Typical Applications USB/Battery Powered Synchronous Boost Converter, 4.3V to 5V, 500mA L1 3.3H VIN 4.3V TO 5.5V SW VIN CIN 4.7F C2 4.7F LTC3121 SD OFF ON PWM/SYNC C1 100nF FB VCC VC SGND VIN 2V/DIV R1 383k CAP RT RT 57.6k VOUT 5V 500mA VOUT VOUT 2V/DIV COUT 47F R2 121k RC 43.2k PGND CVCC 4.7F INPUT CURRENT 0.5A/DIV CF 68pF CC 1nF CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 47F, 6.3V, X7R, 1812 C2: LELON VE-4R7M1ATR-0305 L1: TDK SPM6530T-3R3M 2ms/DIV RLOAD = 20 VIN = USB 2.0 PORT HOTPLUGGED 3121 TA06a 3121 TA06b 5V to Dual Output Synchronous Boost Converter, 15V C2 470nF L1 6.8H -15.1 SW VIN OFF ON LTC3121 VOUT RT FB VCC VC SGND PGND CVCC 4.7F CIN, CVCC: 4.7F, 6.3V, X7R, 1206 COUT2: 47F, 16V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT1: 22F, 16V, X7R, 1812 C2: 470nF, 25V, X7R, 1206 L1: COILCRAFT XAL5050-682ME U1: CENTRAL SEMICONDUCTOR CBAT54S Z1: DIODES, INC. DDZ16ASF-7 22 C1 100nF CAP PWM/SYNC RT 57.6k VOUT1 15V RC 365k CC 150pF -14.9 U1 R1 1.3M COUT1 22F R2 113k CF 10pF COUT2 47F VOUT2 -15V Z1 14.9 -14.8 14.8 -14.7 14.7 -14.6 14.6 -14.5 14.5 VOUT2 -14.4 14.4 -14.3 14.3 -14.2 14.2 -14.1 3121 TA07a 15.0 VOUT1 0 105 35 70 OUTPUT CURRENT (mA) VOUT1 (V) CIN 4.7F SD 15.1 -15.0 VOUT2 (V) VIN 5V 14.1 140 3121 TA07b 3121fa For more information www.linear.com/LTC3121 LTC3121 Typical Applications Single Li-Cell 3-LED Driver, 2.5V/4.2V to 175mA L1 3.3H 100 1.0 EFFICIENCY SW VIN CIN 4.7F SD LTC3121 PWM/SYNC FB LT1006 VC SGND PGND CVCC 4.7F CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 22F, 16V, X7R, 1812 L1: TDK SPM6530T-3R3M D1, D2, D3: CREE XPGWHT-L1-0000-00G51 D2 VCC CAP VCC RT 57.6k D1 C1 100nF RT 0.6 0.5 RC 2k CC 3.9nF + - R1 1.02M COUT1 22F 40 0.4 D3 30 0.3 RS 0.2 0.2 20 POWER LOSS (W) OFF ON 50 60 VOUT EFFICIENCY (%) VIN 2.5V TO 4.2V POWER LOSS R2 30.9k 3121 TA08a 10 0.1 2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3 VIN (V) 3121 TA08b 3121fa For more information www.linear.com/LTC3121 23 LTC3121 Package Description Please refer to http://www.linear.com/product/LTC3121#packaging for the most recent package drawings. DE/UE Package 12-Lead Plastic DFN (4mm x 3mm) (Reference LTC DWG # 05-08-1695 Rev D) 0.70 0.05 3.60 0.05 2.20 0.05 3.30 0.05 1.70 0.05 PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.50 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 0.10 (2 SIDES) 7 R = 0.115 TYP 0.40 0.10 12 R = 0.05 TYP PIN 1 TOP MARK (NOTE 6) 0.200 REF 3.30 0.10 3.00 0.10 (2 SIDES) 1.70 0.10 0.75 0.05 6 0.25 0.05 1 PIN 1 NOTCH R = 0.20 OR 0.35 x 45 CHAMFER (UE12/DE12) DFN 0806 REV D 0.50 BSC 2.50 REF 0.00 - 0.05 BOTTOM VIEW--EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE A VARIATION OF VERSION (WGED) IN JEDEC PACKAGE OUTLINE M0-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 24 3121fa For more information www.linear.com/LTC3121 LTC3121 Revision History REV DATE DESCRIPTION PAGE NUMBER A 04/16 Added Note 6. 3 Added R1 label to schematic. 18 Modified R1 and R2 values in Table 4. 19 3121fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of itsinformation circuits as described herein will not infringe on existing patent rights. For more www.linear.com/LTC3121 25 LTC3121 Typical Application Dual Supercapacitor Backup Power Supply, 0.5V to 5V L1 3.3H VIN 0.5V TO 5V CIN 4.7F OFF ON VIN SW SD LTC3121 SC2 50F RT 57.6k C1 100nF RT FB VCC VC SGND R1 383k CAP PWM/SYNC SC1 50F VOUT 5V VOUT PGND CVCC 4.7F SD 2V/DIV COUT 47F RC 43.2k CC 1nF R2 121k VOUT 5V/DIV OUTPUT CURRENT 50mA/DIV CF 68pF CIN, CVCC: 4.7F, 6.3V, X7R, 1206 C1: 100nF, 6.3V, X7R, 1206 COUT: 47F, 6.3V, X7R, 1812 L1: TDK SPM6530T-3R3M SC1, SC2: MAXWELL BCAP0050-P270 VIN 2V/DIV 200s/DIV 3121 TA09b 3121 TA09a Related Parts PART NUMBER DESCRIPTION COMMENTS LTC3421 3A ISW, 3MHz, Synchronous Step-Up DC/DC Converter with Output Disconnect 95% Efficiency, VIN = 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 12A, ISD < 1A, QFN24 Package LTC3422 1.5A ISW, 3MHz Synchronous Step-Up DC/DC Converter with Output Disconnect 95% Efficiency, VIN = 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 25A, ISD < 1A, 3mm x 3mm DFN Package LTC3112 2.5A ISW, 750kHz, Synchronous Buck-Boost DC/DC Converter with Output Disconnect, Burst Mode Operation 95% Efficiency, VIN = 2.7V to 15V, VOUT(MAX) = 14V, IQ = 50A, ISD < 1A, 4mm x 5mm DFN and TSSOP Packages LTC3458 1.4A ISW, 1.5MHz, Synchronous Step-Up DC/DC Converter/ Output Disconnect/Burst Mode Operation 93% Efficiency, VIN = 1.5V to 6V, VOUT(MAX) = 7.5V, IQ = 15A, ISD < 1A, DFN12 Package LTC3528 1A ISW, 1MHz, Synchronous Step-Up DC/DC Converter with Output Disconnect/Burst Mode Operation 94% Efficiency, VIN = 700mV to 5.25V, VOUT(MAX) = 5.25V, IQ = 12A, ISD < 1A, 3mm x 2mm DFN Package LTC3539 2A ISW, 1MHz/2MHz, Synchronous Step-Up DC/DC Converters with Output Disconnect/Burst Mode Operation 94% Efficiency, VIN = 700mV to 5.25V, VOUT(MAX) = 5.25V, IQ = 10A, ISD < 1A, 3mm x 2mm DFN Package LTC3459 70mA ISW, 10V Micropower Synchronous Boost Converter/ Output Disconnect/Burst Mode Operation VIN = 1.5V to 5.5V, VOUT(MAX) = 10V, IQ = 10A, ISD < 1A, ThinSOTTM Package LTC3499 750mA Synchronous Step-Up DC/DC Converters with Reverse-Battery Protection 94% Efficiency, VIN = 1.8V to 5.5V, VOUT(MAX) = 6V, IQ = 20A, ISD < 1A, 3mm x 3mm DFN and MSOP Packages LTC3115-1 40V, 2A Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.7V to 40V, VOUT(MAX) = 40V, IQ = 50A, ISD < 3A, 4mm x 5mm DFN and TSSOP Packages LTC3122 2.5A ISW, 3MHz, Synchronous Step-Up DC/DC Converter with Output Disconnect, Burst Mode Operation 95% Efficiency, VIN = 1.8V to 5.5V [500mV After Start-Up], VOUT(MAX) = 15V, IQ = 25A, ISD < 1A, 3mm x 4mm DFN and MSOP Packages LTC3124 5A ISW, 6MHz, Dual Phase, Synchronous Step-Up DC/DC Converter with Output Disconnect, Burst Mode Operation 95% Efficiency, VIN = 1.8V to 5.5V [500mV After Start-Up], VOUT(MAX) = 15V, IQ = 25A, ISD < 1A, 3mm x 5mm DFN and TSSOP Packages 26 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LTC3121 (408) 432-1900 FAX: (408) 434-0507 www.linear.com/LTC3121 3121fa LT 0416 REV A * PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2015