LTC3121
1
3121fa
For more information www.linear.com/LTC3121
Typical applicaTion
FeaTures DescripTion
15V, 1.5A Synchronous
Step-Up DC/DC Converter
with Output Disconnect
The LT C
®
3121 is a synchronous step-up DC/DC converter
with true output disconnect and inrush current limiting. The
1.5A current limit along with the ability to program output
voltages up to 15V makes the LTC3121 well suited for a
variety of demanding applications. Once started, operation
will continue with inputs down to 500mV, extending run
time in many applications.
The LTC3121 features output disconnect in shutdown,
dramatically reducing input power drain and enabling
VOUT to completely discharge. Adjustable PWM switching
from 100kHz to 3MHz optimizes applications for highest
efficiency or smallest solution footprint. The oscillator
can also be synchronized to an external clock for noise
sensitive applications. Selectable Burst Mode operation
reduces quiescent current to 25µA, ensuring high efficiency
across the entire load range. An internal soft-start limits
inrush current during start-up.
Other features include a <1µA shutdown current and robust
protection under short-circuit, thermal overload, and output
overvoltage conditions. The LTC3121 is offered in a low
profile 12-lead (3mm × 4mm × 0.75mm) DFN package.
5V to 12V Synchronous Boost Converter with Output Disconnect
applicaTions
n VIN Range: 1.8V to 5.5V, 500mV After Start-Up
n Output Voltage Range: 2.2V to 15V
n 400mA Output Current for VIN = 5V and VOUT = 12V
n Output Disconnects from Input When Shut Down
n Synchronous Rectification: Up to 95% Efficiency
n Inrush Current Limit
n Up to 3MHz Adjustable Switching Frequency
Synchronizable to External Clock
n Selectable Burst Mode
®
Operation: 25µA IQ
n Output Overvoltage Protection
n Soft-Start
n <1µA IQ in Shutdown
n 12-Lead, 3mm × 4mm Thermally Enhanced DFN
Package
n PCI Express Cards/Systems
n Piezo Actuators
n Small DC Motors
n 12V Analog Rail From Battery, 5V, or Backup Capacitor
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Efficiency Curve
3121 TA01a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND 210k
390pF
SW
6.8µH
113k
1.02M
10pF
4.7µF
100nF
22µF
VOUT
12V
400mA
57.6k
VIN
5V
4.7µF ONOFF
PWMBURST
LOAD CURRENT (mA)
0.01
EFFICIENCY (%)
POWER LOSS (W)
100
90
70
50
40
80
60
30
20
10
0
10
1
0.1
0.01
100.1
3121 TA01b
6001 100
PWM
Burst Mode
OPERATION
PWM POWER LOSS
LTC3121
2
3121fa
For more information www.linear.com/LTC3121
absoluTe MaxiMuM raTings
VIN Voltage .................................................. –0.3V to 6V
VOUT Voltage ............................................ –0.3V to 18V
SW Voltage (Note 2) .................................. –0.3V to 18V
SW Voltage (Pulsed < 100ns) (Note 2) ....... –0.3V to 19V
VC, RT Voltage .......................................... –0.3V to VCC
CAP Voltage
VOUT < 5.7V ............................–0.3V to (VOUT + 0.3V)
5.7V ≤ VOUT ≤ 11.7V...... (VOUT – 6V) to (VOUT + 0.3V)
VOUT > 11.7V .................................(VOUT – 6V) to 12V
All Other Pins ............................................... –0.3V to 6V
Operating Junction Temperature Range
(Notes 3, 4) ............................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
(Note 1)
12
11
10
9
8
7
13
PGND
1
2
3
4
5
6
CAP
VOUT
SGND
SD
FB
VC
SW
PGND
VIN
PWM/SYNC
VCC
RT
TOP VIEW
DE PACKAGE
12-LEAD (4mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W (NOTE 5), θJC = 5°C/W
EXPOSED PAD (PIN 13) IS PGND,
MUST BE SOLDERED TO PCB FOR RATED THERMAL PERFORMANCE
pin conFiguraTion
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3121EDE#PBF LTC3121EDE#TRPBF 3121 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C
LTC3121IDE#PBF LTC3121IDE#TRPBF 3121 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
http://www.linear.com/product/LTC3121#orderinfo
LTC3121
3
3121fa
For more information www.linear.com/LTC3121
elecTrical characTerisTics
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Voltage transients on the SW pin beyond the DC limit specified in
the Absolute Maximum Ratings are non-disruptive to normal operations
when using good layout practices, as shown on the demo board or
described in the data sheet or application notes.
Note 3: The LTC3121 is tested under pulsed load conditions such that
TA ≈ TJ. The LTC3121E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3121I is guaranteed
to meet specifications over the full –40°C to 125°C operating junction
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C (Note 3). VIN = 3.6V, VOUT = 12V, RT = 57.6k unless otherwise noted.
PARAMETER CONDITIONS MIN. TYP MAX UNITS
Minimum Start-Up Voltage VOUT = 0V l1.7 1.8 V
Input Voltage Range After VOUT ≥ 2.2V l0.5 5.5 V
Output Voltage Adjust Range l2.2 15 V
Feedback Voltage l1.178 1.202 1.225 V
Feedback Input Current VFB = 1.4V 1 50 nA
Quiescent Current, Shutdown VSD = 0V, VOUT = 0V, Not Including Switch Leakage 0.01 1 µA
Quiescent Current, Active VC = 0V, Measured On VIN, Non-Switching 500 700 µA
Quiescent Current, Burst Measured on VIN, VFB > 1.4V
Measured on VOUT, VFB > 1.4V
25
10
40
20
µA
µA
N-channel MOSFET Switch Leakage Current VSW = 15V, VOUT = 15V, VC = 0V (Note 6) l0.1 30 µA
P-channel MOSFET Switch Leakage Current VSW = VIN = 0V, VOUT = 15V (Note 6) l0.1 70 µA
N-channel MOSFET Switch On-Resistance 0.121 Ω
P-channel MOSFET Switch On-Resistance 0.188 Ω
N-channel MOSFET Current Limit VIN = 3.3V l1.5 1.8 2.7 A
Maximum Duty Cycle VFB = 1.0V l90 94 %
Minimum Duty Cycle VFB = 1.4V l0 %
Switching Frequency l0.85 1 1.15 MHz
SYNC Frequency Range l0.1 3 MHz
PWM/SYNC Input High l0.9 •VCC V
PWM/SYNC Input Low l0.1•VCC V
PWM/SYNC Input Current VPWM/SYNC = 5.5V 0.01 1 µA
CAP Clamp Voltage VOUT > 6.1V, Referenced to VOUT –5.2 –5.6 –6.0 V
Error Amplifier Transconductance l70 100 130 µS
Error Amplifier Output Current ±25 µA
Soft-Start Time 10 ms
SD Input High l1.6 V
SD Input Low l0.25 V
SD Input Current VSD = 5.5V 1 2 µA
temperature range. The junction temperature (TJ in °C) is calculated from
the ambient temperature (TA in °C) and power dissipation (PD in watts)
according to the formula: TJ = TA + (PD • θJA) where θJA is the thermal
impedance of the package.
Note 4: The LTC3121 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature shutdown is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
Note 5: Failure to solder the exposed backside of the package to the PC
board ground plane will result in a thermal impedance much higher than
the rated package specifications.
Note 6: Measured using a propietary test mode to ensure anti-ringing
switch between VIN ans SW is not active.
LTC3121
4
3121fa
For more information www.linear.com/LTC3121
Typical perForMance characTerisTics
PWM Mode Operation Load Transient Response Inrush Current Control
Feedback vs Temperature
RDS(ON) vs Temperature,
Both NMOS and PMOS
Oscillator Frequency
vs Temperature
Efficiency vs Load Current,
VOUT = 5V
Efficiency vs Load Current,
VOUT = 7.5V
Efficiency vs Load Current,
VOUT = 12V
Configured as front page application unless otherwise specified.
1µs/DIV 3121 G04
VOUT
20mV/DIV
AC-COUPLED
INDUCTOR
CURRENT
1A/DIV
ILOAD = 200mA
500µs/DIV 3121 G05
VOUT
500mV/DIV
AC-COUPLED
OUTPUT
CURRENT
250mA/DIV
40mA 40mA
400mA
TEMPERATURE (°C)
–50
CHANGE IN RDS(ON) FROM 25°C (%)
80
60
40
20
0
–20
–40 70 110–10
3121 G08
30 150
TEMPERATURE (°C)
–60
CHANGE IN FREQUENCY FROM 25°C (%)
1.0
0.5
–0.5
–1.0
0
–1.5
–2.0 90–10
3121 G09
40 140
TEMPERATURE (°C)
–60
CHANGE IN VFB FROM 25°C (%)
0.2
0.1
–0.2
–0.1
0
–0.3
–0.4
–0.5
–0.6 40 90–10
3121 G07
140
LOAD CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3121 G01
1 100 1000
BURST
PWM
VIN = 4.2V
VIN = 3.3V
VIN = 0.6V
LOAD CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3121 G03
10001 100
PWM
VIN = 5.4V
VIN = 4.2V
VIN = 2.6V
BURST
LOAD CURRENT (mA)
0.01
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0100.1
3121 G02
1 100 1000
PWM
VIN = 5.4V
VIN = 3.8V
VIN = 2.3V
BURST
2ms/DIV 3121 G06
VOUT
5V/DIV
SD
5V/DIV
INDUCTOR
CURRENT
1A/DIV
LTC3121
5
3121fa
For more information www.linear.com/LTC3121
Typical perForMance characTerisTics
Burst Mode Maximum Output
Current vs VIN
Burst Mode No Load Input Current
vs VIN
Burst Mode Quiescent Current
Change vs Temperature
SD Pin Threshold Frequency vs RT Frequency Accuracy
PWM Mode Maximum Output
Current vs VIN
Peak Current Limit Change
vs Temperature
PWM Operation No Load Input
Current vs VIN
VIN (V)
0.5
OUTPUT CURRENT (A)
0.8
0.6
0.4
1.4
1.2
1.0
0.2
03.5 4.51.5
3121 G10
2.5 5.5
VOUT = 5V
VOUT = 7.5V
VOUT = 12V
VIN (V)
0
INPUT CURRENT (mA)
70
40
30
20
60
50
10
04 521
3121 G12
3 6
VOUT = 5V
VOUT = 7.5V
VOUT = 12V
VIN, FALLING (V)
0.5
OUTPUT CURRENT (mA)
350
200
150
100
300
250
50
04.52.51.5
3121 G13
3.5 5.5
VOUT = 2.2V
VOUT = 5V
VOUT = 7.5V
VOUT = 12V
VIN, FALLING (V)
0.5
INPUT CURRENT (µA)
10000
1000
100
10 4.52.51.5
3121 G14
3.5 5.5
VOUT = 5V
VOUT = 7.5V
VOUT = 12V
TEMPERATURE (°C)
–50
CHANGE IN CURRENT FROM 25°C (%)
75
60
45
30
15
0
–15 70 110–10
3121 G15
30 150
TEMPERATURE (°C)
–50
PEAK CURRENT LIMIT CHANGE FROM 25°C (%)
2
1
0
–1
–2
–3
–4 70 110–10
3121 G11
30 150
RT (kΩ)
0
FREQUENCY (MHz)
PERIOD (µs)
3.0
2.0
1.5
1.0
2.5
0.5
0
12
8
6
4
10
2
0
400200100
3121 G17
300 600500
FREQUENCY
PERIOD
VIN FALLING (V)
0
CHANGE IN FREQUENCY (%)
4
2
1
0
–1
3
–2
–3
–4 421
3121 G18
3 65
VOUT = 15V
VOUT = 3.6V
VOUT = 2.2V
1s/DIV 3121 G16
VOUT
5V/DIV
VSD
500mV/DIV
900mV
400mV
LTC3121
6
3121fa
For more information www.linear.com/LTC3121
Typical perForMance characTerisTics
Burst Mode Operation
Burst Mode Operation
to PWM Mode
PWM Mode to Burst Mode
Operation
Burst Mode Transient Synchronized Operation Short-Circuit Response
Efficiency vs Frequency CAP Pin Voltage vs VOUT VCC vs VIN
OUTPUT CURRENT (mA)
10
EFFICIENCY (%)
100
40
30
20
60
50
70
90
80
10
0100
3121 G19
1000
fOSC = 200kHz
fOSC = 1MHz
fOSC = 3MHz
VOUT (V)
0
VCAP, REFERRED TO VOUT (V)
0
–3
–4
–5
–2
–1
–6
–7 10642
3121 G20
8 1412
1µs/DIV 3121 G26
VSW
5V/DIV
VPWM/SYNC
5V/DIV
SYNCHRONIZED TO 1.3MHz
20µs/DIV 3121 G23
VOUT
100mV/DIV
AC-COUPLED
VPWM/SYNC
2V/DIV
OUTPUT CURRENT = 70mA
20µs/DIV 3121 G24
VOUT
100mV/DIV
AC-COUPLED
VPWM/SYNC
2V/DIV
OUTPUT CURRENT = 70mA
5µs/DIV 3121 G22
VOUT
100mV/DIV
AC-COUPLED
VSW
10V/DIV
INDUCTOR
CURRENT
0.5A/DIV OUTPUT CURRENT = 50mA
200µs/DIV 3121 G25
VOUT
200mV/DIV
AC-COUPLED
OUTPUT
CURRENT
100mA/DIV 10mA 10mA
100mA
VIN (V)
0
VCC (V)
4.5
4.0
3.5
3.0
2.5 421
3121 G21
3 65
VIN FALLING
VIN RISING
200µs/DIV 3121 G27
VOUT
5V/DIV
INDUCTOR
CURRENT
1A/DIV
SHORT-CIRCUIT APPLIED
SHORT-CIRCUIT
REMOVED
ILOAD = 100mA
LTC3121
7
3121fa
For more information www.linear.com/LTC3121
pin FuncTions
SW (Pin 1): Switch Pin. Connect an inductor from this
pin to VIN. Keep PCB trace lengths as short and wide as
possible to reduce EMI and voltage overshoot. When VOUT
≥ VIN + 2V, an internal anti-ringing resistor is connected
between SW and VIN after the inductor current has dropped
to near zero, to minimize EMI. The anti-ringing resistor is
also activated in shutdown and during the sleep periods
of Burst Mode operation.
PGND (Pin 2, Exposed Pad Pin 13): Power Ground. When
laying out your PCB, provide a short, direct path between
PGND and the output capacitor and tie directly to the ground
plane. The exposed pad is ground and must be soldered
to the PCB ground plane for rated thermal performance.
VIN (Pin 3): Input Supply Pin. The device is powered from
VIN unless VOUT exceeds VIN and VIN is less than 3V. Place
a low ESR ceramic bypass capacitor of at least 4.7µF from
VIN to PGND. X5R and X7R dielectrics are preferred for
their superior voltage and temperature characteristics.
PWM/SYNC (Pin 4): Burst Mode Operation Select and
Oscillator Synchronization. Do not leave this pin floating.
• PWM/SYNC = High. Disable Burst Mode Operation and
maintain low noise, constant frequency operation.
• PWM/SYNC = Low. The converter operates in Burst
Mode operation, independent of load current.
• PWM/SYNC = External CLK. The internal oscillator is
synchronized to the external CLK signal. Burst Mode
operation is disabled. A clock pulse width between
100ns and 2µs is required to synchronize the oscillator.
An external resistor must be connected between RT
and GND to program the oscillator slightly below the
desired synchronization frequency.
In non-synchronized applications, repeated clocking of
the PWM/SYNC pin to affect an operating mode change
is supported with these restrictions:
• Boost Mode (VOUT > VIN): IOUT <500µA: ƒPWM/SYNC ≤
100Hz, IOUT ≥ 500µA: ƒPWM/SYNC ≤ 5kHz
• Buck Mode (VOUT < VIN): IOUT <5mA: ƒPWM/SYNC ≤ 5Hz,
IOUT ≥ 5mA: ƒPWM/SYNC ≤ 5kHz
VCC (Pin 5): VCC Regulator Output. Connect a low-ESR
filter capacitor of at least 4.7µF from this pin to GND to
provide an internal regulated rail approximately equal to
the lower of VIN and 4.25V. When VOUT is higher than VIN,
and VIN falls below 3V, VCC will regulate to the lower of
approximately VOUT and 4.25V. A UVLO event occurs if VCC
drops below 1.6V. Switching is inhibited, and a soft-start
is initiated when VCC returns above 1.7V.
RT (Pin 6): Frequency Adjust Pin. Connect an external
resistor (RT) from this pin to SGND to program the oscil-
lator frequency according to the formula:
RT = 57.6/ƒOSC
where ƒOSC is in MHz and RT is in kΩ.
VC (Pin 7): Error Amplifier Output. A frequency compen-
sation network is connected to this pin to compensate
the control loop. See Compensating the Feedback Loop
section for guidelines.
FB (Pin 8): Feedback Input to the Error Amplifier. Connect
the resistor divider tap to this pin. Connect the top of the
divider to VOUT and the bottom of the divider to SGND.
The output voltage can be adjusted from 2.2V to 15V ac-
cording to this formula:
VOUT = 1.202V • (1 + R1/R2)
SD (Pin 9): Logic Controlled Shutdown Input. Bringing this
pin above 1.6V enables normal, free-running operation,
forcing this pin below 0.25V shuts the LTC3121 down, with
quiescent current below 1μA. Do not leave this pin floating.
SGND (Pin 10): Signal Ground. When laying out a PC
board, provide a short, direct path between SGND and
the (–) side of the output capacitor.
VOUT (Pin 11): Output Voltage Sense and the Source of
the Internal Synchronous Rectifier MOSFET. Driver bias
is derived from VOUT. Connect the output filter capacitor
from VOUT to PGND, as close to the IC as possible. A
minimum value of 10µF ceramic is recommended. VOUT
is disconnected from VIN when SD is low.
CAP (Pin 12): Serves as the Low Reference for the Syn-
chronous Rectifier Gate Drive. Connect a low ESR filter
capacitor (typically 100nF) from this pin to VOUT to provide
an elevated ground rail, approximately 5.6V below VOUT,
used to drive the synchronous rectifier.
LTC3121
8
3121fa
For more information www.linear.com/LTC3121
block DiagraM
3121 BD
LTC3121
PWM
LOGIC
AND
DRIVERS
SHUTDOWN SD CURRENT
SENSE
TSD
VREF_UP
OSC
SD
OVLO
ANTI-RING
PWM
BURST
SYNC
CONTROL
3
1
L1 11
12
7
8
10
2
13
+––
+ –
ADAPTIVE SLOPE COMPENSATION
ILIM
REF
4
5
9
LDO
VBEST
VIN VOUT
6
OSCILLATOR OSC
+ –
BULK CONTROL
SIGNALS
SOFT-START
VC CLAMP
SD
TSD
OVLO
REFERENCE
UVLO
THERMAL SD
VREF_UP
1.202V
TSD SGND
PGND
EXPOSED PAD
RT
VCC
VIN
CIN
RT
SW
PWM/SYNC
SD
VIN
1.8V TO 5.5V
CVCC
4.7µF
COUT
IZERO
COMP OVLO
PGND
16.2V
1.202V
gm ERROR
AMPLIFIER
VOUT
CAP
FB
R1
R2
VC
CPL
VC
RPL
C1
100nF
CC
RC
CF
VOUT
2.2V TO 15V
THE VALUES OF RC, CC, AND CF ARE BASED UPON OPERATING CONDITIONS.
PLEASE REFER TO COMPENSATING THE FEEDBACK LOOP SECTION FOR
GUIDELINES TO DETERMINE OPTIMAL VALUES OF THESE COMPONENTS.
VIN
–
+
LTC3121
9
3121fa
For more information www.linear.com/LTC3121
operaTion
The LTC3121 is an adjustable frequency, 100kHz to 3MHz
synchronous boost converter housed in a 12-lead 4mm ×
3mm DFN. The LTC3121 offers the unique ability to start-
up and regulate the output from inputs as low as 1.8V and
continue to operate from inputs as low as 0.5V. Output
voltages can be programmed between 2.2V and 15V. The
device also features fixed frequency, current mode PWM
control for exceptional line and load regulation. The cur-
rent mode architecture with adaptive slope compensation
provides excellent transient load response and requires
minimal output filtering. An internal 10ms closed loop
soft-start simplifies the design process while minimizing
the number of external components.
With its low RDS(ON) and low gate charge internal N-channel
MOSFET switch and P-channel MOSFET synchronous
rectifier, the LTC3121 achieves high efficiency over a wide
range of load current. High efficiency is achieved at light
loads when Burst Mode operation is commanded. Operation
can be best understood by referring to the Block Diagram.
LOW VOLTAGE OPERATION
The LTC3121 is designed to allow start-up from input
voltages as low as 1.8V. When VOUT exceeds 2.2V, the
LTC3121 continues to regulate its output, even when VIN
falls to as low as 0.5V. The limiting factors for the applica-
tion become the availability of the input source to supply
sufficient power to the output at the low voltages, and
the maximum duty cycle. Note that at low input voltages,
small voltage drops due to series resistance become
critical and greatly limit the power delivery capability of
the converter. This feature extends operating times by
maximizing the amount of energy that can be extracted
from the input source.
LOW NOISE FIXED FREQUENCY OPERATION
Soft-Start
The LTC3121 contains internal circuitry to provide closed-
loop soft-start operation. The soft-start utilizes a linearly
increasing ramp of the error amplifier reference voltage
from zero to its nominal value of 1.202V in approximately
10ms, with the internal control loop driving VOUT from
zero to its final programmed value. This limits the inrush
current drawn from the input source. As a result, the du-
ration of the soft-start is largely unaffected by the size of
the output capacitor or the output regulation voltage. The
closed loop nature of the soft-start allows the converter
to respond to load transients that might occur during
the soft-start interval. The soft-start period is reset by a
shutdown command on SD, a UVLO event on VCC (VCC <
1.6V), an overvoltage event on VOUT (VOUT ≥ 16.2V), or
an overtemperature event (thermal shutdown is invoked
when the die temperature exceeds 170°C). Upon removal
of these fault conditions, the LTC3121 will soft-start the
output voltage.
Error Amplifier
The non-inverting input of the transconductance error
amplifier is internally connected to the 1.202V reference
and the inverting input is connected to FB. An external
resistive voltage divider from VOUT to ground programs
the output voltage from 2.2V to 15V via FB as shown in
Figure 1.
VOUT =1.202V 1+R1
R2
⎛
⎝
⎜⎞
⎠
⎟
Selecting an R2 value of 121kΩ to have approximately
10µA of bias current in the VOUT resistor divider yields
the formula:
R1 = 100.67•(VOUT – 1.202V)
where R1 is in kΩ.
Power converter control loop compensation is set by a
simple RC network between VC and ground.
Figure 1. Programming the Output Voltage
3121 F01
FB
LTC3121
1.202V R2
R1
VOUT
+
–
LTC3121
10
3121fa
For more information www.linear.com/LTC3121
operaTion
Internal Current Limit
The current limit comparator shuts off the N-channel
MOSFET switch once its threshold is reached. Peak switch
current is limited to 1.8A, independent of input or output
voltage, except when VOUT is below 1.5V, resulting in the
current limit being approximately half of the nominal peak.
Lossless current sensing converts the peak current sig-
nal of the N-channel MOSFET switch into a voltage that
is summed with the internal slope compensation. The
summed signal is compared to the error amplifier output
to provide a peak current control command for the PWM.
Zero Current Comparator
The zero current comparator monitors the inductor current
being delivered to the output and shuts off the synchro-
nous rectifier when this current reduces to approximately
50mA. This prevents the inductor current from reversing
in polarity, improving efficiency at light loads.
Oscillator
The internal oscillator is programmed to the desired switch-
ing frequency with an external resistor from the RT pin to
GND according to the following formula:
ƒOSC (MHz) =57.6
R
T
(kΩ)
⎛
⎝
⎜⎞
⎠
⎟
The oscillator also can be synchronized to an external
frequency by applying a pulse train to the PWM/SYNC pin.
An external resistor must be connected between RT and
GND to program the oscillator to a frequency approximately
25% below that of the externally applied pulse train used
for synchronization. RT is selected in this case according
to this formula:
RT(kΩ)=73.2
ƒ
SYNC
(MHz)
⎛
⎝
⎜⎞
⎠
⎟
Output Disconnect
The LTC3121’s output disconnect feature eliminates body
diode conduction of the internal P-channel MOSFET
rectifier. This allows for VOUT to discharge to 0V during
shutdown, and draw no current from the input source. It
also allows for inrush current limiting at turn-on, minimiz-
ing surge currents seen by the input supply. Note that to
obtain the advantages of output disconnect, there must
not be an external Schottky diode connected between SW
and VOUT. The output disconnect feature also allows VOUT
to be pulled high, without reverse current being backfed
into the power source connected to VIN.
Shutdown
The boost converter is disabled by pulling SD below 0.25V
and enabled by pulling SD above 1.6V. Note that SD can
be driven above VIN or VOUT, as long as it is limited to less
than the absolute maximum rating.
Thermal Shutdown
If the die temperature exceeds 170°C typical, the LTC3121
will go into thermal shutdown (TSD). All switches will be
turned off until the die temperature drops by approximately
7°C, when the device re-initiates a soft-start and switching
can resume.
Boost Anti-Ringing Control
When VOUT ≥ VIN + 2V, the anti-ringing control connects
a resistor across the inductor to damp high frequency
ringing on the SW pin during discontinuous current mode
operation when the inductor current has dropped to near
zero. Although the ringing of the resonant circuit formed
by L and CSW (capacitance on SW pin) is low energy, it
can cause EMI radiation.
VCC Regulator
An internal low dropout regulator generates the 4.25V
(nominal) VCC rail from VIN or VOUT, depending upon
operating conditions. VCC is supplied from VIN when VIN
is greater than 3.5V, otherwise the greater of VIN and VOUT
is used. The VCC rail powers the internal control circuitry
and power MOSFET gate drivers of the LTC3121. The VCC
regulator is disabled in shutdown to reduce quiescent
current and is enabled by forcing the SD pin above its
threshold. A 4.7µF or larger capacitor must be connected
between VCC and SGND.
LTC3121
11
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Overvoltage Lockout
An overvoltage condition occurs when VOUT exceeds ap-
proximately 16.2V. Switching is disabled and the internal
soft-start ramp is reset. Once VOUT drops below approxi-
mately 15.6V, a soft-start cycle is initiated and switching
is enabled. If the boost converter output is lightly loaded
so that the time constant product of the output capaci-
tance, COUT, and the output load resistance, ROUT is near
or greater than the soft-start time of approximately 10ms,
the soft-start ramp may end before or soon after switching
resumes, defeating the inrush current limiting of the closed
loop soft-start following an overvoltage event.
Short-Circuit Protection
The LTC3121 output disconnect feature allows output
short-circuit protection. To reduce power dissipation under
overload and short-circuit conditions, the peak switch cur-
rent limit is reduced to 1A. Once VOUT > 1.5V, the current
limit is set to its nominal value of 1.8A.
VIN > VOUT Operation
The LTC3121 step-up converter will maintain voltage regu-
lation even when the input voltage is above the desired
output voltage. Note that operating in this mode will exhibit
lower efficiency and a reduced output current capability.
Refer to the Typical Performance Characteristics section
for details.
Burst Mode OPERATION
When the PWM/SYNC pin is held low, the boost converter
operates in Burst Mode operation to improve efficiency
at light loads and reduce standby current at no load. The
input thresholds for this pin are determined relative to VCC
with a low being less than 10% of VCC and a high being
greater than 90% of VCC. The LTC3121 will operate in
fixed frequency PWM mode even if Burst Mode operation
is commanded during soft-start.
In Burst Mode operation, the LTC3121 switches asynchro-
nously. The inductor current is first charged to 600mA
by turning on the N-channel MOSFET switch. Once this
current threshold is reached, the N-channel is turned off
and the P-channel synchronous switch is turned on, de-
livering current to the output. When the inductor current
discharges to approximately zero, the cycle repeats. In
Burst Mode operation, energy is delivered to the output
until the nominal regulation value is reached, at which
point the LTC3121 transitions to sleep mode. In sleep, the
output switches are turned off and the L
TC3121 consumes
only 25μA of quiescent current. When the output volt-
age droops approximately 1%, switching resumes. This
maximizes efficiency at very light loads by minimizing
switching and quiescent losses. Output voltage ripple in
Burst Mode operation is typically 1% to 2% peak-to-peak.
Additional output capacitance (10μF or greater), or the
addition of a small feed-forward capacitor (10pF to 50pF)
connected between VOUT and FB can help further reduce
the output ripple.
The maximum output current (IOUT) capability in Burst
Mode operation varies with VIN and VOUT, as shown in
Figure 2.
Figure 2. Burst Mode Maximum Output Current vs VIN
VIN, FALLING (V)
0.5
OUTPUT CURRENT (mA)
350
300
200
100
50
250
150
03.51.5
3121 F02
5.52.5 4.5
VOUT = 2.2V
VOUT = 5V
VOUT = 7.5V
VOUT = 12V
operaTion
LTC3121
12
3121fa
For more information www.linear.com/LTC3121
applicaTions inForMaTion
PCB LAYOUT GUIDELINES
The high switching frequency of the LTC3121 demands
careful attention to board layout. A careless layout will
result in reduced performance. Maximizing the copper
area for ground will help to minimize die temperature rise.
A multilayer board with a separate ground plane is ideal,
but not absolutely necessary. See Figure 3 for an example
of a two-layer board layout.
COMPONENT SELECTION
Inductor Selection
The LTC3121 can utilize small surface mount inductors due
to its capability of setting a fast (up to 3MHz) switching
frequency. Larger values of inductance will allow slightly
greater output current capability by reducing the inductor
ripple current. To design a stable converter the range of
inductance values is bounded by the targeted magnitude
of the internal slope compensation and is inversely propor-
tional to the switching frequency. The inductor selection
for the LTC3121 has the following bounds:
µH >L>
µH
The inductor peak-to-peak ripple current is given by the
following equation:
RIPPLE(A) =
V
IN
•(V
OUT
– V
IN
)
f•L•VOUT
where:
L = Inductor Value in µH
f = Switching Frequency in MHz
The inductor ripple current is a maximum at the minimum
inductor value. Substituting 3/f for the inductor value in
the above equation yields the following:
RIPPLEMAX(A) =
V
IN
•(V
OUT
– V
IN
)
3•VOUT
To realize greater output current capability at the guaran-
teed minimum (over temperature) 1.5A current limit, it is
recommended that the inductor ripple current be limited to
one-third of this minimum value, or to approximately 0.5A.
Choosing a minimum inductor value of 6/f μH (where f =
switching frequency in MHz) or greater typically results in
an inductor ripple current of 0.5A or less for the majority of
step-up ratios. High frequency ferrite core inductor materi-
als reduce frequency dependent power losses compared
to cheaper powdered iron types, improving efficiency.
Figure 3. Traces Carrying High Current Are Direct (PGND, SW, VIN
and VOUT). Trace Area at FB and VC Are Kept Low. Trace Length to
Input Supply Should Be Kept Short. VIN and VOUT Ceramic Capacitors
Should Be Placed as Close to the LTC3121 Pins as Possible
12
11
10
9
8
7
13
PGND
3121 F02
VIN
PGND
1
2
3
4
5
6
SGND
FB
SW
VCC
VOUT
PGND
CAP
VC
RT
SCHOTTKY DIODE
Although it is not required, adding a Schottky diode from
SW to VOUT can improve the converter efficiency by about
4%. Note that this defeats the output disconnect and short-
circuit protection features of the LTC3121.
LTC3121
13
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For more information www.linear.com/LTC3121
applicaTions inForMaTion
The inductor should have low DCR (series resistance of
the windings) to reduce the I2R power losses, and must be
able to support the peak inductor current without saturat-
ing. Molded chokes and most chip inductors usually do
not have enough core area to support the peak inductor
currents of 2A to 3A seen on the L
TC3121. To minimize
radiated noise, use a shielded inductor.
See Table 1 for suggested components and suppliers
Table 1. Recommended Inductors
PART NUMBER
VALUE
(µH)
DCR
(mΩ)
ISAT
(A)
SIZE (mm)
W × L × H
Coilcraft XAL4020-222ME
Coilcraft XAL4030-332ME
Coilcraft XAL4030-472ME
Coilcraft XAL5050-682ME
Coilcraft XAL6060-223ME
Coilcraft MSS1260T-333ML
2.2
3.3
4.7
6.8
22
33
39
29
44
29
61
57
5.6
5.5
4.5
6.0
5.6
4.34
4.3 × 4.3 × 2.1
4.3 × 4.3 × 3.1
4.3 × 4.3 × 3.1
5.3 × 5.3 × 5.1
6.3 × 6.3 × 6.1
12.3 × 12.3 × 6.2
Coiltronics DR73-2R2-R
Coiltronics DR74-4R7-R
Coiltronics DR125-330-R
Coiltronics DR127-470-R
2.2
4.7
33
47
17
25
51
72
5.52
4.37
3.84
5.28
7.6 × 7.6 × 3.55
7.6 × 7.6 × 4.35
12.5 × 12.5 × 6
12.5 × 12.5 × 8
Sumida CDR7D28MNNP-2R2NC
Sumida CDR7D28MNNP-6R8NC
2.2
6.8
18
46
4.9
3.5
7.6 × 7.6 × 3
7.6 × 7.6 × 3
Taiyo-Yuden NR5040T3R3N 3.3 35 3.8 5 × 5 × 4
TDK LTF5022T-2R2N3R2-LC
TDK SPM6530T-3R3M
TDK VLP8040T-4R7M
2.2
3.3
4.7
40
30
25
3.2
6.8
4.4
5 × 5.2 × 2.2
7.1 × 6.5 × 3
8 × 7.7 × 4
Würth WE-PD7447789002
Würth WE-PD7447789003
Würth WE-PD7447789003
Würth WE-PD7447779006
Würth WE-HCI7443251000
Würth WE-PD744770122
Würth WE-PD744770133
Würth WE-PD7447709470
2.2
3.3
4.7
6.8
10
22
33
47
23
30
35
35
16
43
64
60
4.8
4.2
4.2
3.3
8.5
5
3.6
4.5
7.3 × 7.3 × 3.2
7.3 × 7.3 × 3.2
7.3 × 7.3 × 3.2
7.3 × 7.3 × 4.5
10 × 10 × 5
12 × 12 × 8
12 × 12 × 8
12 × 12 × 10
Output and Input Capacitor Selection
Low ESR (equivalent series resistance) capacitors should
be used to minimize the output voltage ripple. Multilayer
ceramic capacitors are an excellent choice as they have
extremely low ESR and are available in small footprints.
X5R and X7R dielectric materials are preferred for their
ability to maintain capacitance over wide voltage and tem-
perature ranges. Y5V types should not be used. Although
ceramic capacitors are recommended, low ESR tantalum
capacitors may be used as well.
When selecting output capacitors, the magnitude of the
peak inductor current, together with the ripple voltage
specification, determine the choice of the capacitor. Both
the ESR (equivalent series resistance) of the capacitor and
the charge stored in the capacitor each cycle contribute
to the output voltage ripple.
The ripple due to the charge is approximately:
VRIPPLE(CHARGE) ≈
I
P
•V
IN
COUT •VOUT •ƒ
where IP is the peak inductor current.
The ESR of COUT is usually the most dominant factor for
ripple in most power converters. The ripple due to the
capacitor ESR is:
VRIPPLE(ESR) =ILOAD •RESR •
V
OUT
V
IN
where RESR = capacitor equivalent series resistance.
The input filter capacitor reduces peak currents drawn from
the input source and reduces input switching noise. A low
ESR bypass capacitor with a value of at least 4.7µF should
be located as close to the VIN pin as possible.
Low ESR and high capacitance are critical to maintain low
output voltage ripple. Capacitors can be used in parallel
for even larger capacitance values and lower effective
ESR. Ceramic capacitors are often utilized in switching
converter applications due to their small size, low ESR and
low leakage currents. However, many ceramic capacitors
experience significant loss in capacitance from their rated
value with increased DC bias voltage. It is not uncommon
for a small surface mount capacitor to lose more than 50%
of its rated capacitance when operated near its rated volt-
age. As a result it is sometimes necessary to use a larger
capacitor value or a capacitor with a larger value and case
size, such as 1812 rather than 1206, in order to actually
realize the intended capacitance at the full operating volt-
age. Be sure to consult the vendor’s curve of capacitance
vs DC bias voltage. Table 2 shows a sampling of capacitors
suited for L
TC3121 applications.
LTC3121
14
3121fa
For more information www.linear.com/LTC3121
Table 2. Representative Output Capacitors
MANUFACTURER,
PART NUMBER
VALUE
(µF)
VOL
TAGE
(V)
SIZE L × W × H (mm)
TYPE, ESR (mΩ)
AVX,
12103D226MAT2A
22 25 3.2 × 2.5 × 2.79,
X5R Ceramic
Kemet,
C2220X226K3RACTU
22 25 5.7 × 5.0 × 2.4,
X7R Ceramic
Kemet,
A700D226M016ATE030
22 16 7.3 × 4.3 × 2.8,
Alum. Polymer, 30mΩ
Murata,
GRM32ER71E226KE15L
22 25 3.2 × 2.5 × 2.5,
X7R Ceramic
Nichicon,
PLV1E121MDL1
82 25 8 × 8 × 12,
Alum. Polymer, 25mΩ
Panasonic,
ECJ-4YB1E226M
22 25 3.2 × 2.5 × 2.5,
X5R Ceramic
Sanyo,
25TQC22MV
22 25 7.3 × 4.3 × 3.1,
POSCAP, 50mΩ
Sanyo,
16TQC100M
100 16 7.3 × 4.3 × 1.9,
POSCAP, 45mΩ
Sanyo,
25SVPF47M
47 25 6.6 × 6.6 × 5.9,
OS-CON, 30mΩ
Taiyo Yuden,
TMK325BJ226MM-T
22 25 3.2 × 2.5 × 2.5,
X5R Ceramic
TDK,
CKG57NX5R1E476M
47 25 6.5 × 5.5 × 5.5,
X5R Ceramic
Cap-XX
GS230F
1.2Farads 4.5 39 × 17 × 3.8
28mΩ
Cooper
A1030-2R5155
1.5Farads 2.5 Ø = 10, L = 30
60mΩ
Maxwell
BCAP0050-P270
50Farads 2.5 Ø = 18, L = 40
20mΩ
For applications requiring a very low profile and very large
capacitance, the GS, GS2 and GW series from Cap-XX
and PowerStor Aerogel Capacitors from Cooper all offer
very high capacitance and low ESR in various low profile
packages.
A method for improving the converter’s transient response
uses a small feed-forward series network of a capacitor and
a resistor across the top resistor of the feedback divider
(from VOUT to FB). This adds a phase-lead zero and pole
to the transfer function of the converter as calculated in
the Compensating the Feedback Loop section.
OPERATING FREQUENCY SELECTION
There are several considerations in selecting the operating
frequency of the converter. Typically the first consideration
is to stay clear of sensitive frequency bands, which cannot
tolerate any spectral noise. For example, in products incor-
porating RF communications, the 455kHz IF frequency is
sensitive to any noise, therefore switching above 600kHz
is desired. Some communications have sensitivity to
1.1MHz and in that case a 1.5MHz switching converter
frequency may be employed. A second consideration is the
physical size of the converter. As the operating frequency
is increased, the inductor and filter capacitors typically
can be reduced in value, leading to smaller sized external
components. The smaller solution size is typically traded
for efficiency, since the switching losses due to gate charge
increase with frequency.
Another consideration is whether the application can allow
pulse-skipping. When the boost converter pulse-skips, the
minimum on-time of the converter is unable to support
the duty cycle. This results in a low frequency component
to the output ripple. In many applications where physical
size is the main criterion, running the converter in this
mode is acceptable. In applications where it is preferred
not to enter this mode, the maximum operating frequency
is given by:
ƒMAX _NOSKIP ≤
V
OUT −
V
IN
VOUT •tON(MIN)
Hz
where tON(MIN) = minimum on-time = 100ns.
Thermal Considerations
For the LTC3121 to deliver its full power, it is imperative
that a good thermal path be provided to dissipate the heat
generated within the package. This can be accomplished
by taking advantage of the large thermal pad on the un-
derside of the IC. It is recommended that multiple vias in
the printed circuit board be used to conduct heat away
from the IC and into a copper plane with as much area as
possible. If the junction temperature rises above ~170°C,
the part will go into thermal shutdown, and all switching
will stop until the temperature drops approximately 7°C.
applicaTions inForMaTion
LTC3121
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Compensating the Feedback Loop
The LTC3121 uses current mode control, with internal
adaptive slope compensation. Current mode control elimi-
nates the second order filter due to the inductor and output
capacitor exhibited in voltage mode control, and simplifies
the power loop to a single pole filter response. Because
of this fast current control loop, the power stage of the IC
combined with the external inductor can be modeled by a
transconductance amplifier gmp and a current controlled
current source. Figure 4 shows the key equivalent small
signal elements of a boost converter.
The DC small-signal loop gain of the system shown in
Figure 4 is given by the following equation:
GBOOST =GEA •GMP •GPOWER •
R2
R1+R2
where GEA is the DC gain of the error amplifier, GMP is
the modulator gain, and GPOWER is the inductor current
to VOUT gain.
GEA =gma •RO≈950V/V
(Not Adjustable; gma = 95µS, RO≈10MΩ)
GMP = gmp = ΔIL
ΔVC
≈3.4S (Not Adjustable)
GPOWER = ΔVOUT
ΔIL
=η•VIN
2•IOUT
Combining the two equations above yields:
GDC =GMP •GPOWER ≈
1.7 •
η
•V
IN
IOUT
V/V
Converter efficiency η will vary with IOUT and switching
frequency ƒOSC as shown in the typical performance
characteristics curves.
Output Pole: P1 =
2
2•π•RL•COUT
Hz
Error Amplifier Pole: P2 = 1
2•π•RO•(CC+CF)Hz
Error Amplifier Zero: Z1 = 1
2•π•RC•CC
Hz
ESR Zero: Z2 = 1
2•π•RESR •COUT
Hz
RHP Zero: Z3 = V
IN2•RL
2•π•VOUT2•LHz
High Frequency Pole: P3 >ƒOSC
3
Phase Lead Zero: Z4 =1
2•π•(R1+RPL )•CPL
Hz
Phase Lead Pole: P4 =1
2•π•R1•R2
R1+R2 +RPL
⎛
⎝
⎜⎞
⎠
⎟•CPL
Hz
Error Amplifier Filter Pole:
P5 =1
2•π•RC•CC•CF
CC+CF
⎛
⎝
⎜⎞
⎠
⎟
Hz
applicaTions inForMaTion
Figure 4. Boost Converter Equivalent Model
3121 F04
VOUT
–
+
–
+
RC
VC
RO
gma
gmp
CC
CF
IL
MODULATOR
ERROR
AMPLIFIER
1.202V
REFERENCE
RPL
R1
FB
R2
RESR RL
CPL
COUT
• IL
η • VIN
2 • VOUT
CC: COMPENSATION CAPACITOR
COUT: OUTPUT CAPACITOR
CPL: PHASE LEAD CAPACITOR
CF: HIGH FREQUENCY FILTER CAPACITOR
gma: TRANSCONDUCTANCE AMPLIFIER INSIDE IC
gmp: POWER STAGE TRANSCONDUCTANCE AMPLIFIER
RC: COMPENSATION RESISTOR
RL: OUTPUT RESISTANCE DEFINED AS VOUT/ILOADMAX
RO: OUTPUT RESISTANCE OF gma
RPL: PHASE LEAD RESISTOR
R1, R2: FEEDBACK RESISTOR DIVIDER NETWORK
RESR: OUTPUT CAPACITOR ESR
η : CONVERTER EFFICIENCY (~90% AT HIGHER CURRENTS)
LTC3121
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The current mode zero (Z3) is a right half plane zero
which can be an issue in feedback control design, but is
manageable with proper external component selection.
As a general rule, the frequency at which the open-loop
gain of the converter is reduced to unity, known as the
crossover frequency ƒC, should be set to less than one
third of the right half plane zero (Z3), and under one eighth
of the switching frequency ƒOSC. Once ƒC is selected, the
values for the compensation components can be calculated
using a bode plot of the power stage or two generally valid
assumptions: P1 dominates the gain of the power stage
for frequencies lower than ƒC and ƒC is much higher than
P2. First calculate the power stage gain at ƒC, GƒC in V/V.
Assuming the output pole P1 dominates GƒC for this range,
it is expressed by:
GƒC ≈
G
DC
1+ƒC
P1
⎛
⎝
⎜⎞
⎠
⎟
2V/V
Decide how much phase margin (Φm) is desired. Greater
phase margin can offer more stability while lower phase mar-
gin can yield faster transient response. Typically, Φm ≈ 60°
is optimal for minimizing transient response time while
allowing sufficient margin to account for component vari-
ability. Φ1 is the phase boost of Z1, P2, and P5 while Φ2 is
the phase boost of Z4 and P4. Select Φ1 and Φ2 such that
Φ1≤74°;Φ2≤2•tan−1VOUT
1.2V
⎛
⎝
⎜⎞
⎠
⎟−90° and
Φ1+ Φ2= Φm+tan−1ƒC
Z3
⎛
⎝
⎜⎞
⎠
⎟
where VOUT is in V and ƒC and Z3 are in kHz.
Setting Z1, P5, Z4, and P4 such that
Z1=
ƒ
C
a
1
, P5 =ƒCa1, Z4 =
ƒ
C
a
2
, P4 =ƒCa2
allows a1 and a2 to be determined using Φ1 and Φ2
a1=tan2Φ1+
90
°
2
⎛
⎝
⎜⎞
⎠
⎟, a2 = tan2Φ2+
90
°
2
⎛
⎝
⎜⎞
⎠
⎟
The compensation will force the converter gain GBOOST
to unity at ƒC by using the following expression for CC:
CC=103•gma •R2 •GƒC a1−1
( )
a2
2π•ƒC•R1+R2
( )
a1
pF
(gma in µS, ƒC in kHz, GƒC in V/V)
Once CC is calculated, RC and CF are determined by:
RC=106•a1
2π•ƒC•CC
kΩ (ƒC in kHz, CC in pF)
CF=CC
a
1
−1
The values of the phase lead components are given by
the expressions:
RPL =
R1−a2•
R1•R2
R1+R2
⎛
⎝
⎜⎞
⎠
⎟
a2−1kΩ and
CPL = 106a2−1
( )
R1+R2
( )
2
π
•ƒ
C
•R12a
2
pF
where R1, R2, and RPL are in kΩ and ƒC is in kHz.
Note that selecting Φ2 = 0° forces a2 = 1, and so the
converter will have Type II compensation and therefore
no feedforward: RPL is open (infinite impedance) and CPL
= 0pF. If a2 = 0.833 • VOUT (its maximum), feedforward is
maximized; RPL = 0 and CPL is maximized for this com-
pensation method.
Once the compensation values have been calculated, ob-
taining a converter bode plot is strongly recommended to
verify calculations and adjust values as required.
Using the circuit in Figure 5 as an example, Table 3 shows
the parameters used to generate the bode plot shown in
Figure 6.
applicaTions inForMaTion
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applicaTions inForMaTion
Figure 5. 1MHz, 5V to 12V, 400mA Boost Converter
Figure 6. Bode Plot for Example Converter
Transient Response with
200mA to 400mA Load Step
Switching Waveforms with 400mA Load
3121 F05a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
210k
CC
390pF
SW
L1
6.8µH
R2
113k
R1
1.02M
CF
10pF
CVCC
4.7µF
C1
100nF
COUT
22µF
VOUT
12V
400mA
RT
57.6k
VIN
5V
CIN
4.7µF
ONOFF
PWMBURST
FREQUENCY (kHz)
0.01
GAIN (dB)
PHASE (deg)
170
150
110
70
50
130
90
30
10
–10
–30
180
140
100
60
20
–20
–60
–100
–140
–180
–220
100.1
3121 F06
10001 100
GAIN
PHASE
200ns/DIV 3121 F05b
VOUT
100mV/DIV
AC-COUPLED
SW
10V/DIV
INDUCTOR
CURRENT
1A/DIV
100µs/DIV 3121 F05c
VOUT
100mV/DIV
AC-COUPLED
OUTPUT
CURRENT
200mA/DIV
LTC3121
18
3121fa
For more information www.linear.com/LTC3121
Figure 7. Boost Converter with Phase Lead
3121 F06
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
127k
CC
220pF
SW
L1
6.8µH
R2
113k
R1
1.02M
CF
33pF
CVCC
4.7µF
C1
100nF
COUT
22µF
VOUT
12V
400mA
RT
57.6k
VIN
5V
CIN
4.7µF
ONOFF
PWMBURST
RPL
604k
CPL
10pF
applicaTions inForMaTion
Table 3. Bode Plot Parameters for Type II Compensation
PARAMETER VALUE UNITS COMMENT
VIN 5 V App Specific
VOUT 12 V App Specific
RL30 Ω App Specific
COUT 22 µF App Specific
RESR 5 mΩ App Specific
L 6.8 µH App Specific
ƒOSC 1 MHz Adjustable
R1 1020 kΩ Adjustable
R2 113 kΩ Adjustable
gma 95 µS Fixed
RO10 MΩ Fixed
gmp 3.4 S Fixed
η92 % App Specific
RC210 kΩ Adjustable
CC390 pF Adjustable
CF10 pF Adjustable
RPL Open kΩ Optional
CPL 0 pF Optional
From Figure 6, the phase is 60° when the gain reaches
0dB, so the phase margin of the converter is 60°. The
crossover frequency is 15kHz, which is more than three
times lower than the 121.3kHz frequency of the RHP zero
to achieve adequate phase margin.
The circuit in Figure 7 shows the same application as
that in Figure 5 with Type III compensation. This is ac-
complished by adding CPL and RPL and adjusting CC, CF,
and RC accordingly. Table 4 shows the parameters used
to generate the bode plot shown in Figure 8.
LTC3121
19
3121fa
For more information www.linear.com/LTC3121
Figure 8. Bode Plot Showing Phase Lead
applicaTions inForMaTion
Table 4. Bode Plot Parameters for Type III Compensation
PARAMETER VALUE UNITS COMMENT
VIN 5 V App Specific
VOUT 12 V App Specific
RL30 Ω App Specific
COUT 22 µF App Specific
RESR 5 mΩ App Specific
L 6.8 µH App Specific
ƒOSC 1 MHz Adjustable
R1 1020 kΩ Adjustable
R2 113 kΩ Adjustable
gma 95 µS Fixed
RO10 MΩ Fixed
gmp 3.4 S Fixed
η92 % App Specific
RC127 kΩ Adjustable
CC220 pF Adjustable
CF33 pF Adjustable
RPL 604 kΩ Adjustable
CPL 10 pF Adjustable
FREQUENCY (kHz)
0.01
GAIN (dB)
PHASE (deg)
170
150
110
70
50
130
90
30
10
–10
–30
180
140
100
60
20
–20
–60
–100
–140
–180
–220
100.1
3121 F08
10001 100
GAIN
PHASE
From Figure 8, the phase margin is still optimized at 60°
and the crossover frequency remains 15kHz. Adding CPL
and RPL provides some feedforward signal in Burst Mode
operation, leading to lower output voltage ripple.
LTC3121
20
3121fa
For more information www.linear.com/LTC3121
Single Li-Cell to 6V, 2.5W, 3MHz Synchronous Boost Converter for RF Transmitter
3121 TA03a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
137k
CC
150pF
SW
L1
2.2µH
R2
121k
R1
487k
CF
12pF
CVCC
4.7µF
C1
100nF
COUT
47µF
VOUT
6V
425mA
RT
17.4k
VIN
2.5V TO 4.2V
CIN
4.7µF ONOFF
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 47µF, 10V, X7R, 1812
L1: COILCRAFT XAL5030-222ME
100µs/DIV
40mA
420mA
3121 TA03b
OUTPUT
CURRENT
200mA/DIV
VOUT
200mV/DIV
AC-COUPLED
VIN = 3.6V
Typical applicaTions
3121 TA02a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
280k
CC
220pF
SW
L1
3.3µH
R2
113k
R1
1.02M
CF
10pF
CVCC
4.7µF
C1
100nF
COUT
22µF
VOUT
12V
250mA
RT
28k
VIN
3.3V
CIN
4.7µF
ONOFF
PWMBURST
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 22µF, 16V, X7R, 1812
L1: COILCRAFT XAL5050-332ME
3.3V to 12V, 2MHz Synchronous Boost Converter with Output Disconnect, 250mA
LOAD CURRENT (mA)
0.01
EFFICIENCY (%)
100
80
90
60
40
30
70
50
20
10
0
POWER LOSS (W)
10
1
0.1
10 1000.1
3121 TA02b
1
PWM
Burst Mode OPERATION
PWM POWER LOSS
LTC3121
21
3121fa
For more information www.linear.com/LTC3121
Typical applicaTions
2 AA Cell to 12V Synchronous Boost Converter, 100mA
3121 TA04a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
200k
CC
560pF
SW
L1
6.8µH
R2
113k
R1
1.02M
CF
10pF
CVCC
4.7µF
C1
100nF
COUT
22µF
VOUT
12V
100mA
RT
57.6k
VIN
1.8V TO 3V
CIN
4.7µF
ONOFF
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 22µF, 16V, X7R, 1812
L1: COILCRAFT XAL5050-682ME
VIN (V)
1.6
INPUT CURRENT (A)
1.2
0.6
1.0
0.8
0.4
0.2
EFFICIENCY (%)
100
90
70
50
40
80
60
30
20
10
0
2.2 2.4 2.6 2.81.8
3121 TA04b
2 3.23
EFFICIENCY
INPUT CURRENT
3.3V to 12V, 300kHz Synchronous Boost Converter with Output Disconnect, 250mA
3121 TA05a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
154k
CC
1.2nF
SW
L1
22µH
R2
113k
R1
1.02M
CF
56pF
CVCC
4.7µF
C1
100nF
COUT
68µF
VOUT
12V
250mA
RT
196k
VIN
3.3V
CIN
4.7µF
ONOFF
PWMBURST
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 68µF, 16V, X7R, 1812
L1: COILCRAFT XAL6060-223ME
LOAD CURRENT (mA)
0.01
EFFICIENCY (%)
100
80
90
60
40
30
70
50
20
10
0
POWER LOSS (W)
10
1
0.1
0.01
10 1000.1
3121 TA05b
1
PWM
Burst Mode OPERATION
PWM POWER LOSS
LTC3121
22
3121fa
For more information www.linear.com/LTC3121
Typical applicaTions
USB/Battery Powered Synchronous Boost Converter, 4.3V to 5V, 500mA
3121 TA06a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
43.2k
CC
1nF
SW
L1
3.3µH
R2
121k
R1
383k
CF
68pF
CVCC
4.7µF
C1
100nF
COUT
47µF
VOUT
5V
500mA
RT
57.6k
VIN
4.3V TO 5.5V
CIN
4.7µF
C2
4.7µF
ONOFF
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 47µF, 6.3V, X7R, 1812
C2: LELON VE-4R7M1ATR-0305
L1: TDK SPM6530T-3R3M
2ms/DIV
3121 TA06b
INPUT
CURRENT
0.5A/DIV
VOUT
2V/DIV
VIN
2V/DIV
RLOAD = 20Ω
VIN = USB 2.0
PORT HOTPLUGGED
5V to Dual Output Synchronous Boost Converter, ±15V
OUTPUT CURRENT (mA)
0
V
OUT2
(V)
–15.1
–15.0
–14.8
–14.6
–14.5
–14.9
–14.7
–14.4
–14.3
–14.2
–14.1
V
OUT1
(V)
15.1
15.0
14.8
14.6
14.5
14.9
14.7
14.4
14.3
14.2
14.1
14035 70
3121 TA07b
105
VOUT1
VOUT2
3121 TA07a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
365k
CC
150pF
SW
L1
6.8µH
R2
113k
Z1
R1
1.3M
CF
10pF
CVCC
4.7µF
C1
100nF U1
C2
470nF
COUT1
22µF
COUT2
47µF
V
OUT1
15V
V
OUT2
–15V
RT
57.6k
V
IN
5V
CIN
4.7µF
ONOFF
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
COUT2: 47µF, 16V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT1: 22µF, 16V, X7R, 1812
C2: 470nF, 25V, X7R, 1206
L1: COILCRAFT XAL5050-682ME
U1: CENTRAL SEMICONDUCTOR CBAT54S
Z1: DIODES, INC. DDZ16ASF-7
LTC3121
23
3121fa
For more information www.linear.com/LTC3121
Typical applicaTions
Single Li-Cell 3-LED Driver, 2.5V/4.2V to 175mA
3121 TA08a
VIN
SD
PWM/SYNC
RT
VCC
VOUT
CAP
FB
VC
LTC3121
SGND PGND RC
2k
CC
3.9nF
SW
L1
3.3µH
CVCC
4.7µF
VCC
C1
100nF
LT1006
C
OUT1
22µF
RT
57.6k
VIN
2.5V TO
4.2V
CIN
4.7µF
ONOFF
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 22µF, 16V, X7R, 1812
L1: TDK SPM6530T-3R3M
D1, D2, D3: CREE XPGWHT-L1-0000-00G51
+
–
R1
1.02M
R2
30.9k
RS
0.2Ω
D1
D2
D3
VIN (V)
2.5
EFFICIENCY (%)
100
10
20
30
40
50
60
POWER LOSS (W)
1.0
0.3
0.4
0.5
0.6
0.2
0.1
4.1 4.32.7 2.9 3.1
3121 TA08b
3.93.5 3.73.3
EFFICIENCY
POWER LOSS
LTC3121
24
3121fa
For more information www.linear.com/LTC3121
package DescripTion
DE/UE Package
12-Lead Plastic DFN (4mm × 3mm)
(Reference LTC DWG # 05-08-1695 Rev D)
4.00 ±0.10
(2 SIDES)
3.00 ±0.10
(2 SIDES)
NOTE:
1. DRAWING PROPOSED TO BE A VARIATION OF VERSION
(WGED) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
0.40 ± 0.10
BOTTOM VIEW—EXPOSED PAD
1.70 ± 0.10
0.75 ±0.05
R = 0.115
TYP
R = 0.05
TYP
2.50 REF
16
127
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
TOP MARK
(NOTE 6)
0.200 REF
0.00 – 0.05
(UE12/DE12) DFN 0806 REV D
2.50 REF
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
2.20 ±0.05
0.70 ±0.05
3.60
±0.05
PACKAGE OUTLINE
3.30 ±0.10
0.25 ± 0.05
0.50 BSC
1.70 ± 0.05
3.30 ±0.05
0.50 BSC
0.25 ± 0.05
Please refer to http://www.linear.com/product/LTC3121#packaging for the most recent package drawings.
LTC3121
25
3121fa
For more information www.linear.com/LTC3121
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision hisTory
REV DATE DESCRIPTION PAGE NUMBER
A 04/16 Added Note 6.
Added R1 label to schematic.
Modified R1 and R2 values in Table 4.
3
18
19
LTC3121
26
3121fa
For more information www.linear.com/LTC3121
LINEAR TECHNOLOGY CORPORATION 2015
LT 0416 REV A • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LTC3121
relaTeD parTs
Typical applicaTion
Dual Supercapacitor Backup Power Supply, 0.5V to 5V
PART NUMBER DESCRIPTION COMMENTS
LTC3421 3A ISW, 3MHz, Synchronous Step-Up DC/DC Converter
with Output Disconnect
95% Efficiency, VIN = 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 12μA,
ISD < 1μA, QFN24 Package
LTC3422 1.5A ISW, 3MHz Synchronous Step-Up DC/DC Converter
with Output Disconnect
95% Efficiency, VIN = 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 25μA,
ISD < 1μA, 3mm × 3mm DFN Package
LTC3112 2.5A ISW, 750kHz, Synchronous Buck-Boost DC/DC Converter
with Output Disconnect, Burst Mode Operation
95% Efficiency, VIN = 2.7V to 15V, VOUT(MAX) = 14V, IQ = 50μA,
ISD < 1μA, 4mm × 5mm DFN and TSSOP Packages
LTC3458 1.4A ISW, 1.5MHz, Synchronous Step-Up DC/DC Converter/
Output Disconnect/Burst Mode Operation
93% Efficiency, VIN = 1.5V to 6V, VOUT(MAX) = 7.5V, IQ = 15μA,
ISD < 1μA, DFN12 Package
LTC3528 1A ISW, 1MHz, Synchronous Step-Up DC/DC Converter
with Output Disconnect/Burst Mode Operation
94% Efficiency, VIN = 700mV to 5.25V, VOUT(MAX) = 5.25V, IQ = 12µA,
ISD < 1µA, 3mm × 2mm DFN Package
LTC3539 2A ISW, 1MHz/2MHz, Synchronous Step-Up DC/DC Converters
with Output Disconnect/Burst Mode Operation
94% Efficiency, VIN = 700mV to 5.25V, VOUT(MAX) = 5.25V, IQ = 10µA,
ISD < 1µA, 3mm × 2mm DFN Package
LTC3459 70mA ISW, 10V Micropower Synchronous Boost Converter/
Output Disconnect/Burst Mode Operation
VIN = 1.5V to 5.5V, VOUT(MAX) = 10V, IQ = 10μA, ISD < 1μA,
ThinSOT™ Package
LTC3499 750mA Synchronous Step-Up DC/DC Converters with
Reverse-Battery Protection
94% Efficiency, VIN = 1.8V to 5.5V, VOUT(MAX) = 6V, IQ = 20µA,
ISD < 1µA, 3mm × 3mm DFN and MSOP Packages
LTC3115-1 40V, 2A Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.7V to 40V, VOUT(MAX) = 40V, IQ = 50µA,
ISD < 3µA, 4mm × 5mm DFN and TSSOP Packages
LTC3122 2.5A ISW, 3MHz, Synchronous Step-Up DC/DC Converter with
Output Disconnect, Burst Mode Operation
95% Efficiency, VIN = 1.8V to 5.5V [500mV After Start-Up],
VOUT(MAX) = 15V, IQ = 25µA, ISD < 1µA, 3mm × 4mm DFN and MSOP
Packages
LTC3124 5A ISW, 6MHz, Dual Phase, Synchronous Step-Up DC/DC
Converter with Output Disconnect, Burst Mode Operation
95% Efficiency, VIN = 1.8V to 5.5V [500mV After Start-Up],
VOUT(MAX) = 15V, IQ = 25µA, ISD < 1µA, 3mm × 5mm DFN and TSSOP
Packages
3121 TA09a
VIN VOUT
LTC3121
SGND PGND RC
43.2k
CC
1nF
SW
L1
3.3µH
R2
121k
R1
383k
CF
68pF
CVCC
4.7µF
C1
100nF
COUT
47µF
V
OUT
5V
RT
57.6k
VIN
0.5V TO 5V
CIN, CVCC: 4.7µF, 6.3V, X7R, 1206
C1: 100nF, 6.3V, X7R, 1206
COUT: 47µF, 6.3V, X7R, 1812
L1: TDK SPM6530T-3R3M
SC1, SC2: MAXWELL BCAP0050-P270
SD
SC1
50F
SC2
50F
PWM/SYNC
RT
VCC
CAP
FB
VC
ON
OFF
CIN
4.7µF
200s/DIV 3121 TA09b
SD
2V/DIV
OUTPUT
CURRENT
50mA/DIV
VOUT
5V/DIV
VIN
2V/DIV