19-3690; Rev 1; 2/09 KIT ATION EVALU E L B A AVAIL High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator Features The MAX8566 high-efficiency switching regulator delivers up to 10A load current at output voltages from 0.6V to (0.87 x VIN). The IC operates from 2.3V to 3.6V input supplies, making it ideal for point-of-load applications. The total output-voltage set error is less than 1% over load, line, and temperature. The MAX8566 operates in pulse-width-modulation (PWM) mode with a 250kHz to 2.4MHz switching frequency range that is programmable by an external resistor. The IC can be synchronized to an external clock in the same frequency range using the SYNC input. The high operating frequency minimizes the size of external components. Using low-RDS(ON) n-channel MOSFETs for both high- and low-side switches maintains high efficiency at both heavy-load and highswitching frequencies. o Internal 8m On-Resistance MOSFETs o 10A Output PWM Step-Down Regulator o 1% Output Accuracy over Load, Line, and Temperature o Operates from 2.3V to 3.6V Input Supply o Adjustable Output from 0.6V to (0.87 x VIN) o 250kHz to 2.4MHz Adjustable Frequency or SYNC Input o Allows All-Ceramic-Capacitor Design o SYNCOUT Drives 2nd Regulator 180 Out-of-Phase o Prebiased or Monotonic Soft-Start o Programmable Soft-Start Time o Output Tracking or Sequencing o Sourcing and Sinking Output Current o Power-Good Output o 32-Lead Thin QFN Package o REFIN for DDR-Termination Application PART MAX8566ETJ+ Typical Operating Circuit IN IN IN IN REFIN PGND REFIN FOR TRACKING LSS IN VDD INPUT 2.25V TO 3.6V PGND SS PGND EN PGND SYNC INPUT SYNC MAX8566ETJ+ PROGRAMMABLE FREQUENCY FREQ STEP-DOWN REGULATOR TQFN 5mm x 5mm LX SYNCOUT LX GND LX L1 330nH/10A LX LX BST LX LX FB COMP SYNC OUTPUT 180 PGND LX SYSTEM ENABLE ASIC/CPU/DSP Core Voltages Base-Station Power Supplies Fiber Power Supplies Telecom Power Supplies Network Power Supplies PIN-PACKAGE 32 Thin QFN-EP* +Denotes lead(Pb)/RoHS-compliant free package. *EP = Exposed Pad. Applications POL Power Supplies DDR Power Supplies TEMP RANGE -40C to +85C PWRGD The MAX8566 is available in a 32-pin, 5mm x 5mm thin QFN package. The MAX8566 and all the required external components fit into a footprint of less than 0.80in2. Ordering Information MODE The MAX8566 employs a voltage-mode control architecture with a high-bandwidth (> 10MHz) error amplifier. The voltage-mode control architecture makes switching frequencies greater than 1MHz possible, achieving all-ceramic-capacitor designs to minimize PC board space. The error amplifier works with Type 3 compensation to fully utilize the bandwidth of the highfrequency switching to obtain fast transient response. Adjustable soft-start time provides flexibility to minimize input startup inrush current. An open-drain, powergood (PWRGD) signal goes high when the output reaches 90% of its regulation point. The MAX8566 provides a SYNCOUT output to synchronize a second MAX8566 or a second regulator switching 180 out-of-phase with the first to reduce the input ripple current, which consequently reduces the inputcapacitance requirements. The MAX8566 also provides an external reference input (REFIN) for output-tracking applications. OUTPUT UP TO 10A C5 2 x 22F 6.3V MONOTONIC SS SELECTION POWER-GOOD OUTPUT COMPENSATION Pin Configuration appears at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com. 1 MAX8566 General Description MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator ABSOLUTE MAXIMUM RATINGS EN/SS, EN, IN, SYNC, SKIP, VDD, LSS, PWRGD to GND ..........-0.3V to +4V (4.5V nonswitching) SYNCOUT, SS, COMP, FB, REFIN, FREQ to GND .........................................-0.3V to (VDD + 0.3V) LX Current (Note 1) .................................................-12A to +12A BST to LX .................................-0.3V to +4V (4.5V nonswitching) PGND to GND .......................................................-0.3V to +0.3V Continuous Power Dissipation (TA = +85C) 32-Pin Thin QFN (derate 33.3mW/C above +70C) .....2666.7W Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed the IC's package power-dissipation limits. Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = VDD = VEN = 3.3V, VFB = 0.5V, VSYNC = 0V, TA = 0C to +85C, typical values are at TA = +25C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS IN/VDD IN and VDD Voltage Range 2.3 3.6 V LSS Voltage Range 2.3 3.6 V IN Supply Current VDD Supply Current Quiescent current, VFB = 0.7V 0.7 fS = 1MHz, no load 14 Quiescent current, VFB = 0.7V 1.8 fS = 1MHz, VLSS = VDD 16 Total Shutdown Current into IN and VDD VIN = VDD = VLSS = (VBST - VLX) = TA = +25C 3.6V, VEN = 0V TA = 0C to +85C VDD Undervoltage-Lockout Threshold LX starts/stops switching, 2s deglitch VDD falling VIN = VDD = VBST = 3.6V, VLX = 3.6V or 0V, VEN = 0V TA = 0C to +85C 2.2 4 50 3 VDD rising 2.0 1.72 2.2 1.90 mA mA A V BST Shutdown Supply Current TA = +25C 10 0.05 A PWM COMPARATOR Comparator Propagation Delay 10mV overdrive 20 ns COMP Clamp Voltage, High VIN = 2.3V to 3.6V, VFB = 0.7V Slew Rate Shutdown Resistance 1.80 2.0 0.75 1.4 From COMP to GND, VEN = 0V 2.15 V V/s 30 100 0.6 0.606 V ERROR AMPLIFIER FB Regulation Voltage Error-Amplifier Common-Mode Input Range Error-Amplifier Maximum Output Current 2 VCOMP = 1V to 2V, VDD = 2.5V and 3.3V 0.594 VDD = 2.3V to 2.6V 0 VDD 1.65 VDD = 2.6V to 3.6V 0 VDD 1.7 V 0.8 _______________________________________________________________________________________ mA High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator (VIN = VDD = VEN = 3.3V, VFB = 0.5V, VSYNC = 0V, TA = 0C to +85C, typical values are at TA = +25C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS FB Input Bias Current VFB = 0.7V, TA = +25C 40 200 nA REFIN Input Bias Current VREFIN = 0.6V, TA = +25C 70 250 nA VDD = 2.3V to 2.6V 0 VDD 1.65 VDD = 2.6V to 3.6V 0 VDD 1.7 REFIN Common-Mode Range V LX (ALL PINS COMBINED) VIN = VBST - VLX = 3.3V 8 16 VIN = VBST - VLX = 2.5V 12 20 VIN = VLSS = 3.3V 8 16 VIN = VLSS = 2.5V 12 20 On-Resistance, High Side ILX = -2A On-Resistance, Low Side ILX = 2A Current-Limit Threshold VIN = 2.5V or 3.3V, high side 12 VLX = 3.6V Leakage Current VIN = 3.6V, VEN = 0V, TA = +25C Switching Frequency VIN = 2.5V or 3.3V Minimum Off-Time VIN = 2.5V or 3.3V Maximum Duty Cycle RFREQ = 50k, VIN = 2.5V or 3.3V Minimum Duty Cycle RFREQ = 50k, VIN = 2.5V or 3.3V VLX = 0V 15 20 5 200 m A A -200 +5 RFREQ = 50k 0.8 1 1.2 RFREQ = 23.3k 1.7 2 2.3 50 75 87 m MHz ns 95 % 10 % RMS LX Output Current 10 A 0.7 V ENABLE/SOFT-START EN Input Logic-Low Threshold 0.4 EN Input Logic-High Threshold 1.65 Monotonic start MODE Input Threshold VDD = 2.3V to 3.6V EN, MODE Input Current VEN = VMODE = 0V or 3.6V, VDD = 3.6V, TA = +25C Soft-Start Charging Current VSS = 0.3V SS Discharge Resistance 1.90 30 45 20 % of VDD 0.01 1 A 8 11 A No monotonic start 5 V 8 k _______________________________________________________________________________________ 3 MAX8566 ELECTRICAL CHARACTERISTICS (continued) MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator ELECTRICAL CHARACTERISTICS (continued) (VIN = VDD = VEN = 3.3V, VFB = 0.5V, VSYNC = 0V, TA = 0C to +85C, typical values are at TA = +25C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS 2.40 MHz SYNC Capture Range VDD = 2.3V to 3.6V Pulse Width VDD = 2.3V to 3.6V Input Threshold VDD = 2.3V to 3.6V Input Current VSYNC = 0V or 3.6V, VDD = 3.6V 0.25 tLO 100 tHI 100 VIH 0.4 VIL IIH IIL, TA = +25C ns 0.95 1 -1 -1 1.6 +10 +0.01 +1 V A SYNCOUT Frequency Range VDD = 2.3V to 3.6V 0.25 Phase Shift from SYNC or Internal Oscillator Frequency = 1MHz 160 180 Output Voltage ISYNCOUT = 1mA, VDD = 2.3V to 3.6V VDD 0.4 VDD 0.05 VOH VOL 0.05 2.40 MHz 230 Degrees V 0.4 THERMAL SHUTDOWN Thermal-Shutdown Threshold When LX stops switching Thermal-Shutdown Hysteresis +165 C 20 C POWER GOOD Threshold Voltage VFB falling, 3mV hysteresis Falling-Edge Deglitch 86 90 93 % of VREFIN or 0.6V 30 50 80 s Output Low Voltage IPWRGD = 4mA 0.15 0.3 V Leakage Current VPWRGD = 3.6V, VFB = 0.9V, TA = +25C 0.01 1 A 4 _______________________________________________________________________________________ High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator (VIN = VDD = VEN = 3.3V, VFB = 0.5V, VSYNC = 0V, TA = -40C to +85C, unless otherwise noted. Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS 2.325 3.600 V 2.325 3.600 V 2.2 mA 4 mA IN/VDD IN and VDD Voltage Range LSS Voltage Range IN Supply Current Quiescent current, VFB = 0.7V VDD Supply Current Quiescent current, VFB = 0.7V VDD Undervoltage-Lockout Threshold LX starts/stops switching, 2s rising/falling-edge delay VDD rising VDD falling 2.2 1.72 V COMP Clamp Voltage, High VIN = 2.3V to 3.6V, VFB = 0.7V 1.80 Slew Rate 2.18 0.75 Shutdown Resistance V V/s 100 0.591 0.609 V VDD = 2.325V to 2.6V 0 VDD 1.65 VDD = 2.6V to 3.6V 0 VDD 1.7 From COMP to GND, VEN = 0V ERROR AMPLIFIER FB Regulation Voltage Error-Amplifier Common-Mode Input Range VCOMP = 1V to 2V, VIN = 2.3V or 3.6V Error-Amplifier Maximum Output Current 0.8 VDD = 2.325V to 2.5V VDD = 2.6V to 3.6V mA 0 VDD 1.65 0 VDD 1.7 REFIN Common-Mode Range V V LX (ALL PINS COMBINED) On-Resistance, High Side ILX = -2A On-Resistance, Low Side ILX = 2A Current-Limit Threshold VIN = 2.5V or 3.3V VIN = VBST - VLX = 3.3V 16 VIN = VBST - VLX = 2.5V 20 VIN = VLSS = 3.3V 15 VIN = VLSS = 2.5V 20 12 20 m m A _______________________________________________________________________________________ 5 MAX8566 ELECTRICAL CHARACTERISTICS MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator ELECTRICAL CHARACTERISTICS (continued) (VIN = VDD = VEN = 3.3V, VFB = 0.5V, VSYNC = 0V, TA = -40C to +85C, unless otherwise noted. Note 2) PARAMETER Switching Frequency CONDITIONS VIN = 2.5V or 3.3V MIN TYP MAX RFREQ = 50k 0.8 1.2 RFREQ = 23.3k 1.7 2.3 Minimum Off-Time VIN = 2.5V or 3.3V Maximum Duty Cycle RFREQ = 50k, VIN = 2.5V or 3.3V 90 87 RMS Output Current UNITS MHz ns % 10 A 0.7 V 45 20 % of VDD ENABLE/SOFT-START EN Input Logic-Low Threshold EN Input Logic-High Threshold MODE Input Threshold 1.65 VIN = 2.3V to 3.6V Monotonic start 30 No monotonic start EN, MODE Input Current VEN or VMODE = 0V or 3.6V, VDD = 3.6V Soft-Start Charging Current VSS = 0.3V V 1 A 5 12 A 0.25 2.40 MHz SYNC Capture Range VIN = 2.3V to 3.6V Pulse Width VIN = 2.3V to 3.6V Input Threshold VIN = 2.3V to 3.6V 6 tLO 100 tHI 100 VIH 0.4 VIL _______________________________________________________________________________________ ns 1.6 V High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator (VIN = VDD = VEN = 3.3V, VFB = 0.5V, VSYNC = 0V, TA = -40C to +85C, unless otherwise noted. Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS SYNCOUT Frequency Range VDD = 2.3V to 3.6V 0.25 2.40 MHz Phase Shift from SYNC or Internal Oscillator Frequency = 1MHz 160 230 Degrees Output Voltage ISYNCOUT = 1mA, VDD = 2.3V to 3.6V VDD 0.4 VOH V VOL 0.4 POWER-GOOD Threshold Voltage VFB falling, 3mV hysteresis 85 Falling-Edge Deglitch 30 PWRGD Output Voltage IPWRGD = 4mA 93 % of VREF 80 s 0.3 V Note 2: Specifications to -40C are guaranteed by design and not production tested. Typical Operating Characteristics (Typical values are at VIN = VDD = 3.3V, VOUT = 1.8V, RFREQ = 50k, IOUT = 10A, and TA = +25C.) 90 85 EFFICIENCY (%) EFFICIENCY (%) 90 VOUT = 1.8V 80 75 95 90 VOUT = 1.8V 85 VOUT = 1.5V 80 75 75 65 65 65 60 60 10 100 VOUT = 0.8V 70 60 1 VOUT = 1.5V 80 70 LOAD CURRENT (A) VOUT = 1.8V 85 70 0.1 MAX8566 toc03 95 EFFICIENCY (%) VOUT = 2.5V 100 MAX8566 toc02 95 100 MAX8566 toc01 100 EFFICIENCY vs. LOAD CURRENT VIN = 2.5V, VLSS = 3.3V EFFICIENCY vs. LOAD CURRENT VIN = VLSS = 2.5V EFFICIENCY vs. LOAD CURRENT VIN = VLSS = 3.3V 0.1 1 10 LOAD CURRENT (A) 100 0.1 1 10 100 LOAD CURRENT (A) _______________________________________________________________________________________ 7 MAX8566 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (continued) (Typical values are at VIN = VDD = 3.3V, VOUT = 1.8V, RFREQ = 50k, IOUT = 10A, and TA = +25C.) 0.61 0.60 0.59 0.58 1.5 RFREQ = 50k 1.0 RFREQ = 100k 0.57 0.5 0.05 0 VOUT = 2.5V VOUT = 1.8V -0.05 -0.10 -0.15 -0.20 VOUT = 0.8V -0.25 -0.30 -0.35 0.56 -0.40 0 -40 0 40 120 80 -15 10 35 60 1 2 3 4 5 6 7 8 9 LOAD CURRENT (A) SHUTDOWN SUPPLY CURRENT vs. INPUT VOLTAGE MAXIMUM OUTPUT CURRENT vs. OUTPUT VOLTAGE EXPOSED PADDLE TEMPERATURE vs. LOAD CURRENT 14.5 14.0 OUTPUT CURRENT (A) 8 MAX8566 toc08 MAX8566 toc07 15.0 7 6 5 4 3 13.5 13.0 12.5 12.0 11.5 2 11.0 1 10.5 0 10.0 1.0 1.5 2.0 2.5 3.0 3.5 4.0 70 TA = +25C 20 TA = -40C -30 MAX8566 EV KIT PCB 200LFM 0 OUTPUT VOLTAGE (V) 0.4 0.3 0.2 ILOAD = 0A 0 -0.1 -0.2 ILOAD = 4.5A ILOAD = 10A -0.3 -0.4 2.50 2.75 3.00 3.25 3.50 INPUT VOLTAGE (V) 4 6 LOAD CURRENT (A) 3.75 4.00 10.0 OUTPUT SHORT-CIRCUIT CURRENT (A) MAX8566 toc10 0.5 0.1 2 OUTPUT SHORT-CIRCUIT CURRENT vs. INPUT VOLTAGE LINE REGULATION OUTPUT VOLTAGE CHANGE (%) TA = +85C 10 -80 0.5 0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 INPUT VOLTAGE (V) -0.5 2.25 120 MAX8566 toc11 0.5 EXPOSED PADDLE TEMPERATURE (C) TEMPERATURE (C) VEN = 0V 0 0 85 TEMPERATURE (C) 10 9 -40 MAX8566 toc09 0.55 8 MAX8566 toc06 2.0 0.62 0.10 OUTPUT VOLTAGE CHANGE (%) 0.63 RFREQ = 23.3k MAX8566 toc05 MAX8566 toc04 2.5 FREQUENCY (MHz) REFERENCE VOLTAGE (V) 0.64 LOAD REGULATION FREQUENCY vs. TEMPERATURE REFERENCE VOLTAGE vs. TEMPERATURE 0.65 SHUTDOWN SUPPLY CURRENT (A) MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator 9.5 9.0 8.5 8.0 7.5 7.0 6.5 6.0 2.25 2.45 2.65 2.85 3.05 3.25 3.45 3.65 3.85 INPUT VOLTAGE (V) _______________________________________________________________________________________ 8 10 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator GAIN/PHASE OF THE VOLTAGE LOOP LOAD TRANSIENT (0 TO 5A) MAX8566 toc13 MAX8566 toc12 147 kHz VOUT AC-COUPLED (50mV/div) 0dB GAIN (10dB/div) 5A 56 0 IOUT (2A/div) PHASE (45/div) 1 10 100 1000 0 t = 10s/div FREQUENCY (kHz) STARTUP INTO 0.18 LOAD (RLOAD = 0.18) FULL-LOAD SWITCHING WAVEFORMS MAX8566 toc14 MAX8566 toc15 7A (PEAK) IIN (5A/div) 0A IL 12A (2A/div) 10A 3.3V VOUT (10mV/div) 3V VLX (2V/div) 0V 0A t = 400ns/div VEN (2V/div) 0V 1.8V VOUT (1V/div) 3V 0V VPWRGD 0V (2V/div) t = 400s _______________________________________________________________________________________ 9 MAX8566 Typical Operating Characteristics (continued) (Typical values are at VIN = VDD = 3.3V, VOUT = 1.8V, RFREQ = 50k, IOUT = 10A, and TA = +25C.) Typical Operating Characteristics (continued) (Typical values are at VIN = VDD = 3.3V, VOUT = 1.8V, RFREQ = 50k, IOUT = 10A, and TA = +25C.) SOFT-START WITH REFIN SYNCHRONIZED OPERATION (NO LOAD) MAX8566 toc16 MAX8566 toc17 6.5A IIN (5A/div) IIN (AC-COUPLED) (20mA/div) 0A IL1 (2A/div) 0A VREFIN 0.6V (500mV/div) 0V 1.8V VOUT (1V/div) 0V IL2 (2A/div) 0A 3V VPWRGD (2V/div) 0V VLX1 (5V/div) VLX2 (5V/div) 3.3V 0V 3.3V 0V t = 400s/div t = 400ns/div SOFT-START TIME vs. SOFT-START CAPACITANCE STARTUP INTO PREBIASED OUTPUT (RLOAD = 0.18) MAX8566 toc19 MAX8566 toc18 800 700 SOFT-START TIME (ms) MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator 600 500 400 300 200 7.5A IIN (PEAK) (5A/div) 0A 3.3V VEN (12V/div) 0V 1.8V VOUT 0.9V (1V/div) 0V 3V VPWRGD (2V/div) 0V 100 0 0 1 2 3 4 5 6 7 8 9 10 t = 400s CSS (F) 10 ______________________________________________________________________________________ High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator PIN NAME FUNCTION 1 MODE Monotonic Startup Enable/Disable. Connect MODE to GND or to the center tap of an external resistordivider to enable/disable monotonic startup mode. 2 COMP Error-Amplifier Output. Connect the necessary compensation network from COMP to FB. COMP is internally pulled to GND when the IC is in shutdown mode. 3 PWRGD 4 BST 5-12 LX 13-17 PGND 18-22 IN 23 LSS Low-Side MOSFET-Driver Supply Voltage. Connect LSS to a 2.3V to 3.6V supply voltage. 24 VDD IC Supply Voltage Input. Connect VDD to IN through an external 2 resistor. Bypass VDD to GND with a 4.7F capacitor. 25 REFIN External Reference Input. Connect to an external reference. FB regulates to the voltage at REFIN. Connect REFIN to SS to use the internal reference. 26 SS Soft-Start Input. Connect a capacitor from SS to GND to set the soft-start time. See the Soft-Start section. 27 EN Enable Input. Active-high logic input to enable/disable the MAX8566. Connect EN to IN to enable the IC; connect to GND to disable the IC. 28 SYNC Synchronization Input. Synchronize to an external clock with a frequency of 250kHz to 2.4MHz. Leave SYNC unconnected to disable the synchronization function. 29 FREQ Oscillator Frequency Selection. Connect a resistor from FREQ to GND to select the switching frequency. See the Frequency Select section. 30 SYNCOUT 31 GND 32 FB Feedback Input. Connect FB to the center tap of an external resistor-divider from the output to GND to set the output voltage. -- EP Exposed Pad. Internally connected to GND. Connect to a large ground plane to maximize thermal performance. Not indented as an electrical connection point. Power-Good Output. Open-drain output that is high impedance when VFB 90% of 0.6V. Otherwise, PWRGD is internally pulled low. PWRGD is internally pulled low when the IC is in shutdown mode, VDD is below the UVLO threshold, or the IC is in thermal shutdown. High-Side MOSFET Driver Supply. Bypass BST to LX with a 0.1F capacitor. BST is connected to LSS through an internal pMOS switch. Inductor Connection. All LX pins are internally connected together. Connect all LX pins to the switched side of the inductor. LX is high impedance when the IC is in shutdown mode. Power Ground. All PGND pins are internally connected. Connect all PGND pins externally to the power ground plane. Input Power Supply. All IN pins are internally connected. Connect all IN pins externally to an input supply from 2.3V to 3.6V. Bypass IN to PGND with 20F of ceramic capacitance. Oscillator Output. The SYNCOUT output is 180 out-of-phase from the internal oscillator or the SYNC signal to facilitate running a second regulator 180 out-of-phase with the first to reduce input ripple current. Analog Circuit Ground ______________________________________________________________________________________ 11 MAX8566 Pin Description MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator VDD SHUTDOWN CONTROL EN UVLO CIRCUITRY CURRENT-LIMIT COMPARATOR BST ILIM THRESHOLD LX BIAS GENERATOR IN P VOLTAGE REFERENCE LSS N LX CONTROL LOGIC SS SOFT-START REFIN THERMAL SHUTDOWN + ERROR AMPLIFIER - FB N PWM COMPARATOR - PGND LSS MODE + COMP FREQ SYNC SYNCOUT OSCILLATOR COMP LOW DETECTOR SHDN PWRGD FB MAX8566 N 0.54V GND Figure 1. Functional Diagram 12 ______________________________________________________________________________________ High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator The MAX8566 high-efficiency, voltage-mode switching regulator is capable of delivering up to 10A of output current. The MAX8566 provides output voltages from 0.6V to (0.87 x VIN) from 2.3V to 3.6V input supplies, making it ideal for on-board point-of-load applications. The output voltage accuracy is better than 1% over load, line, and temperature. The MAX8566 features a wide switching frequency range, allowing the user to achieve all-ceramic-capacitor designs and faster transient responses. The high operating frequency minimizes the size of external components. The MAX8566 also features a wide 2.3V to 3.6V input voltage range, making it ideal for point-ofload applications with both 3.3V and 2.5V input voltages. The MAX8566 is available in a small (5mm x 5mm), 32-pin thin QFN package. The SYNCOUT function allows end users to operate two MAX8566s at the same switching frequency with 180 out-of-phase operation to minimize the input ripple current, consequently reducing the input capacitance requirements. The REFIN function makes the MAX8566 an ideal candidate for DDR and tracking power supplies. Using internal low-RDS(ON) (8m) n-channel MOSFETs for both highand low-side switches maintains high efficiency at both heavy-load and high-switching frequencies. In addition, the MAX8566 features a low-side-driver supply input (LSS) to boost the efficiency with a higher driver voltage (3.3V) for 2.5V input applications. The MAX8566 employs the voltage-mode control architecture with a high bandwith (> 10MHz) error amplifier. The voltage-mode control architecture allows above 2MHz switching, reducing board area. The op-amp voltage error amplifier works with Type 3 compensation to fully utilize the bandwidth of the high-frequency switching to obtain fast transient response. Adjustable soft-start time provides flexibilities to minimize input startup inrush current. An open-drain power-good (PWRGD) output goes high when VFB reaches 0.54V. Principle of Operation The controller logic block is the central processor that determines the duty cycle of the high-side MOSFET under different line, load, and temperature conditions. Under normal operation, where the current limit and temperature protection are not triggered, the controller logic block takes the output from the PWM comparator and generates the driver signals for both high-side and low-side MOSFETs. The break-before-make logic and the timing for charging the bootstrap capacitors are calculated by the controller logic block. The error signal from the voltage error amplifier is compared with the ramp signal generated by the oscillator at the PWM comparator and thus the required PWM signal is produced. The high-side switch is turned on at the beginning of the oscillator cycle and turns off when the ramp voltage exceeds the VCOMP signal or the current-limit threshold is exceeded. The low-side switch is then turned on for the remainder of the oscillator cycle. Current Limit The internal, high-side MOSFET has a typical 15A peak current-limit threshold. When current flowing out of LX exceeds this limit, the high-side MOSFET turns off and the synchronous rectifier turns on. The synchronous rectifier remains on until the inductor current falls below the low-side current limit. This lowers the duty cycle and causes the output voltage to droop until the current limit is no longer exceeded. The MAX8566 uses a hiccup mode to prevent overheating during short-circuit output conditions. The device enters hiccup mode when V FB drops below 420mV and the current limit is reached. The IC turns off for 3.4ms and then enters soft-start. If the short-circuit condition remains after the soft-start time, the IC shuts down for another 3.4ms. The IC repeats this behavior until the short-circuit condition is removed. Soft-Start and REFIN The MAX8566 utilizes an adjustable soft-start function to limit inrush current during startup. An 8A (typ) current source charges an external capacitor connected to SS to increase the capacitor voltage in a controlled manner. The soft-start time is adjusted by the value of the external capacitor from SS to GND. The required capacitance value is determined as: C= 8A x t SS 0.6V where tSS is the required soft-start time in seconds. The MAX8566 also features an external reference input (REFIN). The IC regulates FB to the voltage applied to REFIN. The internal soft-start is not available when using an external reference. A method of soft-start when using an external reference is shown in Figure 2. Connect REFIN to SS to use the internal 0.6V reference. ______________________________________________________________________________________ 13 MAX8566 Detailed Description High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator MAX8566 High-Side MOSFET Driver Supply (BST) The gate-drive voltage for the high-side, n-channel switch is generated by a flying-capacitor boost circuit. The capacitor between BST and LX is charged from the VLSS supply while the low-side MOSFET is on. When the low-side MOSFET is switched off, the stored voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage for the high-side internal MOSFET. R1 REFIN C R2 MAX8566 Frequency Select (FREQ) The switching frequency is resistor programmable from 250kHz to 2.4MHz. Set the switching frequency of the IC with a resistor from FREQ to GND (RFREQ). RFREQ is calculated as: Figure 2. Soft-Start Implementation with External Reference Undervoltage Lockout (UVLO) The UVLO circuitry inhibits switching when V DD is below 2V. Once VDD rises above 2V, UVLO clears and the soft-start function activates. A 100mV hysteresis is built in for glitch immunity. RFREQ = Monotonic Startup Modes (MODE) where fS is the desired switching frequency in Hz. When starting up into a precharged output, the MAX8566 does not discharge the output prior to entering soft-start (known as monotonic startup). Drive MODE to 1/3 of VDD to enable monotonic startup mode. Connect MODE to GND to disable monotonic startup mode. 23 18 VIN 2.3V TO 3.6V C1 10F SYNC Function (SYNC, SYNCOUT) The MAX8566 features a SYNC function that allows the switching frequency to be synchronized to any frequency between 250kHz to 2.4MHz. Drive SYNC with a C5 0.047F C24 OPEN 19 20 21 C3 0.22F C2 10F 22 BST LSS IN LX IN LX IN IN IN LX LX LX R1 10 24 POWER-GOOD OUTPUT R2 20k VDD MAX8566 C4 1F 3 27 LX LX LX PGND PGND PGND PWRGD PGND PGND EN R17 20k FB 1 25 R18 10k 28 29 R3 50k 50k 1 x - 0.05s 0.95s fs MODE COMP REFIN SYNC SYNCOUT SS FREQ GND 31 L1 0.47H 4 5 6 7 8 C6 22F 2.4k VOUT 1.8V AT 10A C7 22F R4 100 9 10 11 12 17 16 3300pF R5 24.9k 15 14 13 R6 12.4k 32 2 R7 16.9k C9 330pF 30 26 C11 0.022F C10 22pF Figure 3. Typical Application Circuit. 14 ______________________________________________________________________________________ C8 120pF High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator The MAX8566 has a SYNCOUT output that generates a clock signal that is 180 out-of-phase with its internal oscillator, or the signal applied to SYNC. This allows for another regulator to be synchronized 180 out-of-phase to reduce the input ripple current. Power-Good Output (PWRGD) PWRGD is an open-drain output that goes high impedance once the soft-start ramp has concluded, provided V FB is above 0.54V. PWRGD pulls low when V FB is below 0.54V for at least 50s. PWRGD is low during shutdown. Low-Side MOSFET Driver Supply (LSS) The MAX8566 provides an external input for the lowside MOSFET driver supply (LSS). This allows for higher gate-drive voltages to maximize converter efficiency at low input voltages. Shutdown Mode Drive EN to GND to shut down the IC and reduce quiescent current to 4A. During shutdown, the output is high impedance. Drive EN high to enable the MAX8566. Thermal Protection Thermal-overload protection limits total power dissipation in the device. When the junction temperature exceeds T J = +165C a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction temperature cools by 20C, causing a pulsed output during continuous overload conditions. The soft-start sequence begins after a thermal-shutdown condition. Applications Information VDD Decoupling To decrease the noise effects due to the high switching frequency and maximize the output accuracy of the MAX8566, decouple VDD with a 4.7F capacitor from VDD to GND and a 2 resistor from VDD to VIN. Place the capacitor as close to VDD as possible. Inductor Design Choose an inductor with the following equation: L= VOUT x (VIN - VOUT ) fs x VIN x LIR x IOUT(MAX) where LIR is the ratio of the inductor ripple current to average continuous current at the minimum duty cycle. Choose the LIR between 20% to 40% for best performance and stability. Use a low-loss inductor with the lowest possible DC resistance that fits in the allotted dimensions. Powered iron ferrite core types are often the best choice for performance. With any core material the core must be large enough not to saturate at the peak inductor current (IPEAK). Calculate IPEAK as follows: LIR IPEAK = 1 + x IOUT(MAX) 2 Output Capacitor Selection The key selection parameters for the output capacitor are capacitance, ESR, ESL, and voltage rating requirements. These affect the overall stability, output ripple voltage, and transient response of the DC-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor's ESR, and the voltage drop due to the capacitor's ESL. Calculate the output voltage ripple due to the output capacitance, ESR, and ESL as: VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL) where the output ripple due to output capacitance, ESR, and ESL are: IP-P 8 x COUT x fs VRIPPLE(ESR) = IP-P x ESR VRIPPLE(C) = I VRIPPLE(ESL) = P-P x ESL t ON I or VRIPPLE(ESL) = P-P x ESL, whichever is greater. t OFF ______________________________________________________________________________________ 15 MAX8566 square wave at the desired synchronization frequency. A rising edge on SYNC triggers the internal SYNC circuitry. The frequency of the input into SYNC must be higher than the internal oscillator frequency set by RFREQ. Leave SYNC disconnected to disable the function and operate on the internal oscillator. MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator The peak inductor current (IP-P) is: V -V V IP-P = IN OUT x OUT fs x L VIN Use these equations for initial capacitor selection. Determine final values by testing a prototype or an evaluation circuit. A smaller ripple current results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output voltage ripple decreases with larger inductance. Use ceramic capacitors for low ESR and low ESL at the switching frequency of the converter. The low ESL of ceramic capacitors makes ripple voltages negligible. Load-transient response depends on the selected output capacitance. During a load transient, the output instantly changes by ESR x ILOAD. Before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. After a short time (see the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its predetermined value. The controller response time depends on the closed-loop bandwidth. A higher bandwidth yields a faster response time, preventing the output from deviating further from its regulating value. See the Compensation Design section for more details. Input Capacitor Selection The input capacitor reduces the current peaks drawn from the input power supply and reduces switching noise in the IC. The impedance of the input capacitor at the switching frequency should be less than that of the input source so high-frequency switching currents do not pass through the input source but are instead shunted through the input capacitor. High source impedance requires high input capacitance. The input capacitor must meet the ripple-current requirement imposed by the switching currents. The RMS input ripple current is given by: IRIPPLE = ILOAD x VOUT x (VIN - VOUT ) VIN where IRIPPLE is the input RMS ripple current. Compensation Design The power transfer function consists of one double pole and one zero. The double pole is introduced by the output filtering inductor, L, and the output filtering capacitor, C O . The ESR of the output filtering capacitor 16 determines the zero. The double pole and zero frequencies are given as follows: fP1_ LC = fP2 _ LC = fZ _ ESR = 1 R + ESR 2 x L x C O x O RO + RL 1 2 x ESR x CO where RL is equal to the sum of the output inductor's DCR and the internal switch resistance, RDSON. A typical value for R DSON is 8m. R O is the output load resistance, which is equal to the rated output voltage divided by the rated output current. ESR is the total equivalent series resistance of the output filtering capacitor. If there is more than one output capacitor of the same type in parallel, the value of the ESR in the above equation is equal to that of the ESR of a single output capacitor divided by the total number of output capacitors. The high switching frequency range of the MAX8566 allows the use of ceramic output capacitors. Since the ESR of ceramic capacitors is typically very low, the frequency of the associated transfer-function zero is higher than the unity-gain crossover frequency, fC, and the zero cannot be used to compensate for the double pole created by the output filtering inductor and capacitor. The double pole produces a gain drop of 40dB and a phase shift of 90 degrees per decade. The error amplifier must compensate for this gain drop and phase shift to achieve a stable high-bandwidth closed-loop system. Therefore, use Type 3 compensation as shown in Figure 4. Type 3 compensation possesses three poles and two zeros with the first pole, fP1_EA, located at 0 frequency (DC). Locations of other poles and zeros of the Type 3 compensation are given by: 1 2 x R1 x C1 1 fZ2 _ EA = 2 x R3 x C3 1 fP2 _ EA = 2 x R1 x C2 fZ1_ EA = fP3 _ EA = 1 2 x R2 x C3 The above equations are based on the assumptions that C1>>C2, and R3>>R2, which are true in most applications. Placement of these poles and zeros is ______________________________________________________________________________________ High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator L LX R1 = 1 x 0.8 x C1 C3 = 1 x 0.8 x R3 R2 MAX8566 R3 C3 FB R1 C1 COMP R4 C2 Figure 4. Type 3 Compensation Network determined by the frequencies of the double pole and ESR zero of the power transfer function. It is also a function of the desired closed-loop bandwidth. The following section outlines the step-by-step design procedure to calculate the required compensation components. Begin by setting the desired output voltage. The output voltage is set using a resistor-divider from the output to GND with FB at the center tap (R3 and R4 in Figure 4). Use 20k for R4 and calculate R3 as: V R3 = R4 x OUT - 1 0.6V The zero-cross frequency of the closed-loop, fC, should be less than 20% of the switching frequency, fS. Higher zero-cross frequency results in faster transient response. It is recommended that the zero-cross frequency of the closed loop should be chosen between 10% and 20% of the switching frequency. Once fC is chosen, C1 is calculated from the following equation: C1 = 1.56 x VIN R fC x 2 x x R3 x 1 + L RO Due to the underdamped nature of the output LC double pole, set the two zero frequencies of the Type 3 compensation less than the LC double-pole frequency L x CO x (RO + ESR) RL + RO L x CO x (RO + ESR) RL + RO Set the second compensation pole, fP2_EA, at fZ_ESR yields: C2 = CO x C1 x ESR R1 x C1- CO x ESR Set the third compensation pole at 1/2 of the switching frequency to gain some phase margin. Calculate R2 as follows: R2 = 1 x C3 x fS The above equations provide accurate compensation when the zero-cross frequency is significantly higher than the double-pole frequency. When the zero-cross frequency is near the double-pole frequency, the actual zero-cross frequency is higher than the calculated frequency. In this case, lowering the value of R1 reduces the zero-cross frequency. Also, set the third pole of the Type 3 compensation close to the switching frequency if the zero-cross frequency is above 200kHz to boost the phase margin. Please note that the value of R4 can be altered to make the values of the compensation components practical. The recommended range for R4 is 10k to 50k. PCB Layout Considerations and Thermal Performance The MAX8566EVKIT provides an optimal layout and should be followed closely. For custom design, follow these guidelines: 1) Place decoupling capacitors (VDD and SS) as close to the IC as possible. Keep the power ground plane (connected to PGND) and signal ground plane (connected to GND) separate. ______________________________________________________________________________________ 17 MAX8566 to provide adequate phase boost. Set the two zero frequencies to 80% of the LC double-pole frequency. Hence: MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator OPEN-LOOP GAIN COMPENSATION TRANSFER FUNCTION THE THIRD POLE DOUBLE POLE GAIN (dB) THE SECOND POLE POWER-STAGE TRANSFER FUNCTION THE FIRST AND SECOND ZEROS f Figure 5. Transfer Function for Type 3 Compensation 2) Connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 3) Keep the high-current paths as short and wide as possible. Keep the path of switching current short and minimize the loop area formed by LX, the output capacitors, and the input capacitors. 18 4) Connect IN, LX, and PGND separately to a large copper area to help cool the IC to further improve efficiency and long-term reliability. 5) Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close to the IC as possible. 6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP). ______________________________________________________________________________________ High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator VDD LSS IN IN IN IN IN PGND PROCESS: BiCMOS TOP VIEW 24 23 22 21 20 19 18 17 REFIN 25 16 PGND Package Information SS 26 15 PGND EN 27 14 PGND For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. SYNC 28 13 PGND MAX8566 LX FB 32 9 LX + 1 2 3 4 5 6 7 8 LX 10 LX 31 LX GND LX LX BST 11 PWRGD 30 SYNCOUT COMP LX 29 MODE 12 FREQ PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 32 TQFN-EP T3255-4 21-0140 THIN QFN *CONNECT EP TO GND. ______________________________________________________________________________________ 19 MAX8566 Chip Information Pin Configuration MAX8566 High-Efficiency, 10A, PWM Internal-Switch Step-Down Regulator Revision History REVISION NUMBER REVISION DATE 0 8/05 Initial release 1 2/09 Made corrections to Ordering Information, Pin Description, Compensation Design section, Pin Configuration, and Package Information. DESCRIPTION PAGES CHANGED -- 1, 11, 17, 19-21 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2009 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.