LM48510SDBD - National Semiconductor

October 2007

LM48510

Boosted Class D Audio Power Amplifier

General Description

The LM48510 integrates a boost converter with a high effi-

ciency mono, Class D audio power amplifier to provide 1.2W

continuous power into an 8Ω speaker when operating on a

3.3V power supply with boost voltage (PV1) of 5.0V. When

operating on a 3.3V power supply, the LM48510 is capable of

driving a 4Ω speaker load at a continuous average output of

1.7W with less than 1% THD+N. The Class D amplifier is a

low noise, filterless PWM architecture that eliminates the out-

put filter, reducing external component count, board area

consumption, system cost, and simplifying design.

The LM48510's switching regulator is a current-mode boost

converter operating at a fixed frequency of 0.6MHz.

The LM48510 is designed for use in mobile phones and other

portable communication devices. The high (76%) efficiency

extends battery life when compared to Boosted Class AB am-

plifiers. The LM48510 features a low-power consumption

shutdown mode. Shutdown may be enabled by driving the

Shutdown pin to a logic low (GND).

The gain of the Class D is externally configurable which allows

independent gain control from multiple sources by summing

the signals. Output short circuit and Thermal shutdown pro-

tection prevent the device from damage during fault condi-

tions. Superior click and pop suppression eliminates audible

transients during power-up and shutdown.

Key Specifications

■ Quiescent Power Supply Current 6mA (typ)

■ Output Power

(RL = 8Ω, THD+N ≤ 1%,

VDD = 3.3V, PV1 = 5.0V) 1.2W (typ)

■ Shutdown Current 0.01μA (typ)

Features

■Click and Pop Suppression

■Low 0.01μA Shutdown Current

■76% Efficiency

■Filterless Class D

■2.7V - 5.0V operation (VDD)

■Externally configurable gain on Class D

■Very fast turn on time: 17μs

■Independent Boost and Amplifier shutdown pins

Applications

■Mobile Phones

■PDAs

■Portable media

■Cameras

■Handheld games

Typical Application

20123266

FIGURE 1. Typical LM48510 Audio Amplifier Application Circuit

Boomer® is a registered trademark of National Semiconductor Corporation.

LM48510 Boosted Class D Audio Power Amplifier

Connection Diagram

LM48510SD

20123267

Top View

Order Number LM48510SD

See NS Package Number SDA16B

LLP-14 Pin Name Function

1 VO1 Amplifier Output

2 GND1 Ground

3 PV1 Amplifier Power Input

4 VO2 Amplifier Output

5 NC1 No Connect

6 SDBOOST Boost Regulator Active Low

Shutdown

7 GND2 Signal Ground (Booster)

8 FB Feedback point that connects to

external resistive divider

9 SW Drain of the Internal FET Switch

10 GND3 Power Ground (Booster)

11 VDD Power Supply

12 NC2 No Connect

13 IN+ Amplifier Non-Inverting Input

14 IN- Amplifier Inverting Input

15 V1 Amplifier Power Input

16 SDAMP Amplifier Active Low Shutdown

DAP To be soldered to board for

enhanced thermal dissipation.

www.national.com 2

LM48510

Absolute Maximum Ratings (Notes 2, 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage (VDD, V1)6V

Storage Temperature −65°C to +150°C

Input Voltage −0.3V to VDD + 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150°C

Thermal Resistance

θJA (SD) 37°C/W

See AN-1187 'Leadless Leadframe Packaging (LLP)'

Operating Ratings

Temperature Range

TMIN ≤ TA ≤ TMAX −40°C ≤ TA ≤ +85°C

Supply Voltage (VDD) 2.7V ≤ VDD ≤ 5.0V

Supply Voltage (V1) 4.5V ≤ V1 ≤ 5.5V

Electrical Characteristics VDD = 3.3V (Notes 1, 2)

The following specifications apply for VDD = 3.3V, PV1 = V1 = 5.0V, AV = 6dB (Ri = 150kΩ), RL = 15µH + 8Ω +15µH, fIN = 1kHz,

unless otherwise specified. Limits apply for TA = 25°C.

Symbol Parameter Conditions LM48510 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

IDD Quiescent Power Supply Current VIN = 0, RLOAD = ∞6.06 8.75 mA (max)

ISD Shutdown Current SDAMP = SDBOOST = GND

(Note 9) 0.01 1 μA (max)

VSDIH Shutdown Voltage Input High SD1 Boost

SD2 Amplifier

1.5

1.4

V (min)

VSDIL Shutdown Voltage Input Low SD1 Boost

SD2 Amplifier

0.5

0.4

V (max)

TWU Wake-up Time 17 μs

VOS Output Offset Voltage 10 mV

POOutput Power

RL = 15μH + 4Ω + 15μH

THD+N = 1% (max),

f = 1kHz, 22kHz, BW

VDD = 3.3V 1.7

RL = 15μH + 8Ω + 15μH

THD+N = 1% (max),

f = 1kHz, 22kHz, BW

VDD = 3.3V 1.2 0.9 W (min)

RL = 15μH + 4Ω + 15μH

THD+N = 10% (max),

f = 1kHz, 22kHz, BW

VDD = 2.7V

VDD = 3.3V

1.11

1.9

RL = 15μH + 8Ω + 15μH

THD+N = 10% (max),

f = 1kHz, 22kHz, BW

VDD = 2.7V

VDD = 3.3V

0.98

1.55

THD+N Total Harmonic Distortion + Noise

PO = 500mW, f = 1kHz,

RL = 15μH + 8Ω + 15μH,

VDD = 2.7V

0.06 %

PO = 500mW, f = 1kHz,

RL = 15μH + 8Ω + 15μH,

VDD = 3.3V

0.07

3 www.national.com

LM48510

Symbol Parameter Conditions LM48510 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

εOS Output Noise

VDD = 3.3V, f = 20Hz – 20kHz

Inputs to AC GND, No weighting

input referred

µVRMS

VDD = 3.3V, f = 20Hz – 20kHz

Inputs to AC GND, A weighted

input referred

µVRMS

AVGain 300kΩ/Ri V/V

PSRR Power Supply Rejection Ratio

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 217Hz 89 dB

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 1kHz 83 dB

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 10kHz 55 dB

CMRR Common Mode Rejection Ratio VRIPPLE = 1VP-P, fRIPPLE = 217Hz 70 dB

ηEfficiency PO = 1W, f = 1kHz,

RL = 15μH + 8Ω + 15μH, VDD = 3.3V 76 %

VFB Feedback Pin Reference Voltage (Note 10) 1.23 V

Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions

which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters

where no limit is given, however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower.

Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.

Note 5: Machine Model, 220pF–240pF discharged through all pins.

Note 6: Typicals are measured at 25°C and represent the parametric norm.

Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: Shutdown current is measured with components R1 and R2 removed.

Note 10: Feedback pin reference voltage is measured with the Audio Amplifier disconnected from the Boost converter (the Boost converter is unloaded).

www.national.com 4

LM48510

Typical Performance Characteristics

THD+N vs Frequency

VDD = 2.7V, RL = 15μH + 4Ω + 15μH

20123237

THD+N vs Frequency

VDD = 2.7V, RL = 15μH + 8Ω + 15μH

20123250

THD+N vs Frequency

VDD = 3.3V, RL = 15μH + 4Ω + 15μH

20123239

THD+N vs Frequency

VDD = 3.3V, RL = 15μH + 8Ω + 15μH

20123240

THD+N vs Output Power

VDD = 2.7V, RL = 15μH + 4Ω + 15μH

20123241

THD+N vs Output Power

VDD = 2.7V, RL = 15μH + 8Ω + 15μH

20123251

5 www.national.com

LM48510

THD+N vs Output Power

VDD = 3.3V, RL = 15μH + 4Ω + 15μH

20123243

THD+N vs Output Power

VDD = 3.3V, RL = 15μH + 8Ω + 15μH

20123244

Power Dissipation vs Output Power

VDD = 2.7V

20123234

Power Dissipation vs Output Power

VDD = 3.3V

20123235

Power Dissipation vs Output Power

VDD = 4.2V

20123236

Power Supply Current vs Output Power

VDD = 2.7V

20123229

www.national.com 6

LM48510

Power Supply Current vs Output Power

VDD = 3.3V

20123230

Power Supply Current vs Output Power

VDD = 4.2V

20123231

PSRR vs. Frequency

VDD = 3.3V, RL = 15μH + 8Ω + 15μH

20123232

CMRR vs Frequency

VDD = 3.3V, RL = 15μH + 8Ω + 15μH

20123212

Supply Current vs. Supply Voltage

RL = no load

20123233

SW Current vs. Duty Cycle

20123245

7 www.national.com

LM48510

Feedback Voltage vs. Temperature

20123224

Feedback Bias Current vs. Temperature

20123225

Max Duty Cycle vs. Temperature

20123246

RDS(ON) vs. Temperature

20123227

RDS(ON) vs. VIN

20123228

Output Power vs. Efficiency

RL = 4Ω

201232c2

www.national.com 8

LM48510

Output Power vs. Efficiency

RL = 8Ω

201232c3

Boost Converter Max. Load Current

vs. VDD

20123265

9 www.national.com

LM48510

Application Information

GENERAL AMPLIFIER FUNCTION

The audio amplifier portion of LM48510 is a Class D featuring

a filterless modulation scheme. The differential outputs of the

device switch at 300kHz from PV1 to GND. When there is no

input signal applied, the two outputs (VO1 and VO2) switch

with a 50% duty cycle, with both outputs in phase. Because

the outputs of the Class D are differential, the two signals

cancel each other. This results in no net voltage across the

speaker, thus there is no load current during an idle state,

conserving power.

With an input signal applied, the duty cycle (pulse width) of

the Class D outputs changes. For increasing output voltages,

the duty cycle of VO1 increases, while the duty cycle of VO2

decreases. For decreasing output voltages, the converse oc-

curs, the duty cycle of VO2 increases while the duty cycle of

VO1 decreases. The difference between the two pulse widths

yields the differential output voltage.

OPERATING RATINGS

The LM48510 has independent power supplies for the Class

D audio power amplifier (PV1, V1) and the Boost Converter

(VDD). The Class D amplifier operating rating is 2.4V≤(PV1,

V1)≤5.5V when being used without the Boost.

Note the output voltage (PV1, V1) has to be more than VDD.

DIFFERENTIAL AMPLIFIER EXPLANATION

As logic supply voltages continue to shrink, designers are in-

creasingly turning to differential analog signal handling to

preserve signal to noise ratios with restricted voltage swing.

The amplifier portion of the LM48510 is a fully differential am-

plifier that features differential input and output stages. A

differential amplifier amplifies the difference between the two

input signals. Traditional audio power amplifiers have typical-

ly offered only single-ended inputs resulting in a 6dB reduc-

tion in signal to noise ratio relative to differential inputs. The

amplifier also offers the possibility of DC input coupling which

eliminates the two external AC coupling, DC blocking capac-

itors. The amplifier can be used, however, as a single ended

input amplifier while still retaining it's fully differential benefits.

In fact, completely unrelated signals may be placed on the

input pins. The amplifier portion of the LM48510 simply am-

plifies the difference between the signals. A major benefit of

a differential amplifier is the improved common mode rejec-

tion ratio (CMRR) over single input amplifiers. The common-

mode rejection characteristic of the differential amplifier

reduces sensitivity to ground offset related noise injection,

especially important in high noise applications.

AMPLIFIER DISSIPATION

In general terms, efficiency is considered to be the ratio of

useful work output divided by the total energy required to pro-

duce it with the difference being the power dissipated, typi-

cally, in the IC. The key here is “useful” work. For audio

systems, the energy delivered in the audible bands is con-

sidered useful including the distortion products of the input

signal. Sub-sonic (DC) and super-sonic components

(>22kHz) are not useful. The difference between the power

flowing from the power supply and the audio band power be-

ing transduced is dissipated in the LM48510 and in the trans-

ducer load. The amount of power dissipation in the LM48510

is very low. This is because the ON resistance of the switches

used to form the output waveforms is typically less than

0.25Ω. This leaves only the transducer load as a potential

"sink" for the small excess of input power over audio band

output power. The amplifier dissipates only a fraction of the

excess power requiring no additional PCB area or copper

plane to act as a heat sink.

BOOST CONVERTER POWER DISSIPATION

At higher duty cycles, the increased ON time of the FET

means the maximum output current will be determined by

power dissipation within the boost converter FET switch. The

switch power dissipation from ON-state conduction is calcu-

lated by Equation 1.

PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON) (1)

Where DC is the duty cycle.

There will be some switching losses as well, so some derating

needs to be applied when calculating IC power dissipation.

SHUTDOWN FUNCTION

To reduce power consumption while not in use, the amplifier

of LM48510 contains shutdown circuitry that reduces current

draw to less than 0.01µA. It is best to switch between ground

and supply (PV1, V1) for minimum current usage while in the

shutdown state. While the LM48510 may be disabled with

shutdown voltages in between ground and supply, the idle

current will be greater than the typical 0.01µA value. In-

creased THD may also be observed with voltages less than

VDD on the SDAMP pin when in PLAY mode.

The amplifier has an internal resistor connected between

GND and SDAMP pins. The purpose of this resistor is to elim-

inate any unwanted state changes when the SDAMP pin is

floating. The amplifier will enter the shutdown state when the

SDAMP pin is left floating or if not floating, when the shutdown

voltage has crossed the threshold. To minimize the supply

current while in the shutdown state, the SDAMP pin should be

driven to GND or left floating. If the SDAMP pin is not driven to

GND, the amount of additional resistor current due to the in-

ternal shutdown resistor can be found by Equation (2) below.

(VSD - GND) / 300kΩ(2)

With only a 0.5V difference, an additional 1.7µA of current will

be drawn while in the shutdown state.

In many applications, a microcontroller or microprocessor

output is used to control the shutdown circuitry to provide a

quick, smooth transition into shutdown. Another solution is to

use a single-pole, single-throw switch, and a pull-up resistor.

One terminal of the switch is connected to GND. The other

side is connected to the two shutdown pins and the terminal

of the pull-up resistor. The remaining resistance terminal is

connected to VDD. If the switch is open, then the external pull-

up resistor connected to VDD will enable the LM48510. This

scheme guarantees that the shutdown pins will not float thus

preventing unwanted state changes.

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications using

integrated power amplifiers, and switching DC-DC convert-

ers, is critical for optimizing device and system performance.

Consideration to component values must be used to maxi-

mize overall system quality.

The best capacitors for use with the switching converter por-

tion of the LM48510 are multi-layer ceramic capacitors. They

have the lowest ESR (equivalent series resistance) and high-

www.national.com 10

LM48510

est resonance frequency, which makes them optimum for

high frequency switching converters.

When selecting a ceramic capacitor, only X5R and X7R di-

electric types should be used. Other types such as Z5U and

Y5F have such severe loss of capacitance due to effects of

temperature variation and applied voltage, they may provide

as little as 20% of rated capacitance in many typical applica-

tions. Always consult capacitor manufacturer’s data curves

before selecting a capacitor. High-quality ceramic capacitors

can be obtained from Taiyo-Yuden, AVX, and Murata.

The gain of the amplifier is set by the external resistors, Ri in

Figure 1. The gain is given by Equation (3) below. Best THD

+N performance is achieved with a gain of 2V/V (6dB).

AV = 2 * 150kΩ / Ri (V/V) (3)

It is recommended that resistors with 1% tolerance or better

be used to set the gain of the amplifier. The Ri resistors should

be placed close to the input pins of the amplifier. Keeping the

input traces close to each other and of the same length in a

high noise environment will aid in noise rejection due to the

good CMRR of the Class D. Noise coupled onto input traces

which are physically close to each other will be common mode

and easily rejected by the amplifier.

Input capacitors may be needed for some applications or

when the source is single-ended (see Figure1). Input capac-

itors are needed to block any DC voltage at the source so that

the DC voltage seen between the input terminals of the Class

D is 0V. Input capacitors create a high-pass filter with the input

resistors, Ri. The –3dB point of the high-pass filter is found

using Equation (4) below.

fC = 1 / (2πRi Ci ) (Hz) (4)

The input capacitors may also be used to remove low audio

frequencies. Small speakers cannot reproduce low bass fre-

quencies so filtering may be desired . When the Class D is

using a single-ended source, power supply noise on the

ground is seen as an input signal by the +IN input pin that is

capacitor coupled to ground. Setting the high-pass filter point

above the power supply noise frequencies, 217Hz in a GSM

phone, for example, will filter out this noise so it is not ampli-

fied and heard on the output. Capacitors with a tolerance of

10% or better are recommended for impedance matching.

POWER SUPPLY BYPASSING FOR AMPLIFIER

As with any amplifier, proper supply bypassing is critical for

low noise performance and high power supply rejection. The

capacitor (Cs2, see Figure 1) location on both PV1 and V1 pin

should be as close to the device as possible.

SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER

One of the major considerations is the closedloop bandwidth

of the amplifier. To a large extent, the bandwidth is dictated

by the choice of external components shown in Figure 1. The

input coupling capacitor, Ci, forms a first order high pass filter

which limits low frequency response. This value should be

chosen based on needed frequency response for a few dis-

tinct reasons.

High value input capacitors are both expensive and space

hungry in portable designs. Clearly, a certain value capacitor

is needed to couple in low frequencies without severe atten-

uation. But ceramic speakers used in portable systems,

whether internal or external, have little ability to reproduce

signals below 100Hz to 150Hz. Thus, using a high value input

capacitor may not increase actual system performance.

In addition to system cost and size, click and pop performance

is affected by the value of the input coupling capacitor, Ci. A

high value input coupling capacitor requires more charge to

reach its quiescent DC voltage (nominally 1/2 VDD). This

charge comes from the output via the feedback and is apt to

create pops upon device enable. Thus, by minimizing the ca-

pacitor value based on desired low frequency response, turn-

on pops can be minimized.

SELECTING OUTPUT CAPACITOR (CO) FOR BOOST

CONVERTER

A single 4.7µF to 10µF ceramic capacitor will provide suffi-

cient output capacitance for most applications. If larger

amounts of capacitance are desired for improved line support

and transient response, tantalum capacitors can be used.

Aluminum electrolytics with ultra low ESR such as Sanyo Os-

con can be used, but are usually prohibitively expensive.

Typical electrolytic capacitors are not suitable for switching

frequencies above 500 kHz because of significant ringing and

temperature rise due to self-heating from ripple current. An

output capacitor with excessive ESR can also reduce phase

margin and cause instability.

In general, if electrolytics are used, it is recommended that

they be paralleled with ceramic capacitors to reduce ringing,

switching losses, and output voltage ripple.

SELECTING INPUT CAPACITOR (Cs1) FOR BOOST

CONVERTER

An input capacitor is required to serve as an energy reservoir

for the current which must flow into the coil each time the

switch turns ON. This capacitor must have extremely low

ESR, so ceramic is the best choice. A nominal value of 4.7µF

is recommended, but larger values can be used. Since this

capacitor reduces the amount of voltage ripple seen at the

input pin, it also reduces the amount of EMI passed back

along that line to other circuitry.

SETTING THE OUTPUT VOLTAGE (V1) OF BOOST

CONVERTER

The output voltage is set using the external resistors R1 and

R2 (see Figure 1). A value of approximately 13.3kΩ is rec-

ommended for R2 to establish a divider current of approxi-

mately 92µA. R1 is calculated using the formula:

R1 = R2 X (V1/1.23 − 1) (5)

FEED-FORWARD COMPENSATION FOR BOOST

CONVERTER

Although the LM48510's internal Boost converter is internally

compensated, the external feed-forward capacitor Cf1 is re-

quired for stability (see Figure 1). Adding this capacitor puts

a zero in the loop response of the converter. The recom-

mended frequency for the zero fz should be approximately

6kHz. Cf1 can be calculated using the formula:

Cf1 = 1 / (2π X R1 X fz) (6)

SELECTING DIODES FOR BOOST

The external diode used in Figure 1 should be a Schottky

diode. A 20V diode such as the MBR0520 is recommended.

11 www.national.com

LM48510

The MBR05XX series of diodes are designed to handle a

maximum average current of 0.5A. For applications exceed-

ing 0.5A average but less than 1A, a Microsemi UPS5817 can

be used.

DUTY CYCLE

The maximum duty cycle of the boost converter determines

the maximum boost ratio of output-to-input voltage that the

converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined as:

Duty Cycle = V1 + VDIODE - VDD / V1 + VDIODE - VSW

This applies for continuous mode operation.

INDUCTANCE VALUE

The inductor is the largest sized component and usually the

most costly. “How small can the inductor be?” The answer is

not simple and involves trade-offs in performance. Larger in-

ductors mean less inductor ripple current, which typically

means less output voltage ripple (for a given size of output

capacitor). Larger inductors also mean more load power can

be delivered because the energy stored during each switch-

ing cycle is:

E = L/2 X (lp)2

Where lp is the peak inductor current. An important point to

observe is that the LM48510 will limit its switch current based

on peak current. This means that since lp(max) is fixed, in-

creasing L will increase the maximum amount of power avail-

able to the load. Conversely, using too little inductance may

limit the amount of load current which can be drawn from the

output.

Best performance is usually obtained when the converter is

operated in “continuous” mode at the load current range of

interest, typically giving better load regulation and less output

ripple. Continuous operation is defined as not allowing the in-

ductor current to drop to zero during the cycle. It should be

noted that all boost converters shift over to discontinuous op-

eration as the output load is reduced far enough, but a larger

inductor stays “continuous” over a wider load current range.

To better understand these trade-offs, a typical application

circuit (5V to 12V boost with a 10µH inductor) will be analyzed.

We will assume:

VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V

Since the frequency is 0.6MHz (nominal), the period is ap-

proximately 1.66µs. The duty cycle will be 62.5%, which

means the ON-time of the switch is 1.04µs. It should be noted

that when the switch is ON, the voltage across the inductor is

approximately 4.5V. Using the equation:

V = L (di/dt)

We can then calculate the di/dt rate of the inductor which is

found to be 0.17 A/µs during the ON-time. Using these facts,

we can then show what the inductor current will look like dur-

ing operation:

20123248

FIGURE 2. 10μH Inductor Current

5V - 12V Boost (LM48510)

During the 1.04µs ON-time, the inductor current ramps up

0.176A and ramps down an equal amount during the OFF-

time. This is defined as the inductor “ripple current”. A similar

analysis can be performed on any boost converter, to make

sure the ripple current is reasonable and continuous opera-

tion will be maintained at the typical load current values.

MAXIMUM SWITCH CURRENT

The maximum FET switch current available before the current

limiter cuts in is dependent on duty cycle of the application.

This is illustrated in a graph in the typical performance char-

acterization section which shows typical values of switch

current as a function of effective (actual) duty cycle.

CALCULATING OUTPUT CURRENT OF BOOST

CONVERTER (IAMP)

As shown in Figure 2 which depicts inductor current, the load

current is related to the average inductor current by the rela-

tion:

ILOAD = IIND(AVG) x (1 - DC) (7)

Where "DC" is the duty cycle of the application. The switch

current can be found by:

ISW = IIND(AVG) + 1/2 (IRIPPLE) (8)

Inductor ripple current is dependent on inductance, duty cy-

cle, input voltage and frequency:

IRIPPLE = DC x (VIN-VSW) / (f x L) (9)

combining all terms, we can develop an expression which al-

lows the maximum available load current to be calculated:

ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/2fL (10)

The equation shown to calculate maximum load current takes

into account the losses in the inductor or turn-OFF switching

losses of the FET and diode.

DESIGN PARAMETERS VSW AND ISW

The value of the FET ON voltage (referred to as VSW in equa-

tions 4 thru 7) is dependent on load current. A good approxi-

mation can be obtained by multiplying the RDS(ON) of the FET

times the average inductor current.

www.national.com 12

LM48510

FET on resistance increases at VIN values below 5V, since

the internal N-FET has less gate voltage in this input voltage

range (see Typical Performance Characteristics curves).

Above VIN = 5V, the FET gate voltage is internally clamped to

5V.

The maximum peak switch current the device can deliver is

dependent on duty cycle. For higher duty cycles, see Typical

Performance Characteristics curves.

INDUCTOR SUPPLIERS

Recommended suppliers of inductors for the LM48510 in-

clude, but are not limited to Taiyo-Yuden, Sumida, Coilcraft,

Panasonic, TDK and Murata. When selecting an inductor,

make certain that the continuous current rating is high enough

to avoid saturation at peak currents. A suitable core type must

be used to minimize core (switching) losses, and wire power

losses must be considered when selecting the current rating.

PCB LAYOUT GUIDELINES

High frequency boost converters require very careful layout

of components in order to get stable operation and low noise.

All components must be as close as possible to the LM48510

device. It is recommended that a four layer PCB be used so

that internal ground planes are available.

Some additional guidelines to be observed:

1. Keep the path between L1, D1, and Co extremely short.

Parasitic trace inductance in series with D1 and Co will in-

crease noise and ringing.

2. The feedback components R1, R2 and Cf1 must be kept

close to the FB pin to prevent noise injection on the FB pin

trace.

3. If internal ground planes are available (recommended) use

vias to connect directly to ground at pin 2 of U1, as well as the

negative sides of capacitors Cs1 and Co.

GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION

This section provides practical guidelines for mixed signal

PCB layout that involves various digital/analog power and

ground traces. Designers should note that these are only

"rule-of-thumb" recommendations and the actual results will

depend heavily on the final layout.

Power and Ground Circuits

For two layer mixed signal design, it is important to isolate the

digital power and ground trace paths from the analog power

and ground trace paths. Star trace routing techniques (bring-

ing individual traces back to a central point rather than daisy

chaining traces together in a serial manner) can have a major

impact on low level signal performance. Star trace routing

refers to using individual traces to feed power and ground to

each circuit or even device. This technique will take require a

greater amount of design time but will not increase the final

price of the board. The only extra parts required may be some

jumpers.

Single-Point Power / Ground Connection

The analog power traces should be connected to the digital

traces through a single point (link). A "Pi-filter" can be helpful

in minimizing high frequency noise coupling between the ana-

log and digital sections. It is further recommended to place

digital and analog power traces over the corresponding digital

and analog ground traces to minimize noise coupling.

Placement of Digital and Analog Components

All digital components and high-speed digital signals traces

should be located as far away as possible from analog com-

ponents and circuit traces.

Avoiding Typical Design / Layout Problems

Avoid ground loops or running digital and analog traces par-

allel to each other (side-by-side) on the same PCB layer.

When traces must cross over each other do it at 90 degrees.

Running digital and analog traces at 90 degrees to each other

from the top to the bottom side as much as possible will min-

imize capacitive noise coupling and crosstalk.

13 www.national.com

LM48510

20123268

FIGURE 3. Demo Board Schematic Reference

www.national.com 14

LM48510

Demonstration Board Layout

20123298

FIGURE 4. Top Layer Silkscreen

20123264

FIGURE 5. Top Trace Layer

20123262

FIGURE 6. GND Layer (middle 1)

20123263

FIGURE 7. Power Trace Layer (middle 2)

20123261

FIGURE 8. Bottom Layer

20123260

FIGURE 9. Bottom Silkscreen

15 www.national.com

LM48510

Build Of Material

Designator Description Footprint Quantity Value

Cf1 CHIP CAPACITOR GENERIC CAP 0805 1 470pF

CINA CHIP CAPACITOR GENERIC CAP 1210 1 1μF

CINB CHIP CAPACITOR GENERIC CAP 1210 1 1μF

Co CHIP CAPACITOR GENERIC CAP 1210 1 10μF

Cs1 CHIP CAPACITOR GENERIC CAP 1210 1 2.2μF

Cs2 CHIP CAPACITOR GENERIC CAP 1210 1 4.7μF

D1 SCHOTTKY DIO DIODE MBR0520 IR 1

L1 IND_COILCRAFT-DO1813P 1 4.7μH

R1 CHIP RESISTOR GENERIC RES 0805 1 41.2K

R2 CHIP RESISTOR GENERIC RES 0805 1 13.3K

RINA CHIP RESISTOR GENERIC RES 0805 1 150K

RINB CHIP RESISTOR GENERIC RES 0805 1 150K

www.national.com 16

LM48510

Revision History

Rev Date Description

1.0 11/16/06 Initial release.

1.1 03/07/07 Changed the Limit value on the VSDIH and VSDIL to 1.5 and 0.5 respectively.

1.2 10/15/07 Changed the typical value of Vfb = 1.24 to 1.23

17 www.national.com

LM48510

Physical Dimensions inches (millimeters) unless otherwise noted

LLP Package

Order Number LM48510SD

NS Package Number SDA16B

www.national.com 18

LM48510

Notes

19 www.national.com

LM48510

Notes

LM48510 Boosted Class D Audio Power Amplifier

THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION

(“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY

OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO

SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS,

IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS

DOCUMENT.

TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT

NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL

PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR

APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND

APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE

NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS.

EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO

LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE

AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR

PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY

RIGHT.

LIFE SUPPORT POLICY

NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR

SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL

COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and

whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected

to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform

can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness.

National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other

brand or product names may be trademarks or registered trademarks of their respective holders.

For the most current product information visit us at www.national.com

National Semiconductor

Americas Customer

Support Center

Email:

new.feedback@nsc.com

Tel: 1-800-272-9959

National Semiconductor Europe

Customer Support Center

Fax: +49 (0) 180-530-85-86

Email: europe.support@nsc.com

Deutsch Tel: +49 (0) 69 9508 6208

English Tel: +49 (0) 870 24 0 2171

Français Tel: +33 (0) 1 41 91 8790

National Semiconductor Asia

Pacific Customer Support Center

Email: ap.support@nsc.com

National Semiconductor Japan

Customer Support Center

Fax: 81-3-5639-7507

Email: jpn.feedback@nsc.com

Tel: 81-3-5639-7560