DCDC Converter
Digital SupIRBuck
IR38063
1
Rev 3.13
Dec 15, 2017
25A Single-input Voltage, Synchronous
Buck Regulator with PMBus Interface
FEATURES
DESCRIPTION
Internal LDO allows single 21V operation
Output Voltage Range: 0.5V to 0.875*PVin
0.5% accurate Reference Voltage
Programmable Switching Frequency up to 1.5
MHz using Rt/Sync pin or PMBus
Internal Soft-Start with Pre-Bias Start-up
Enable input with Voltage Monitoring Capability
Remote Sense Amplifier with True Differential
Voltage Sensing
Fast mode I2C and 400 kHz PMBus interface
Sequencing and tracking capable
Selectable analog mode or digital mode
66 PMBus commands for configuration, control,
fault protection and telemetry.
Thermally compensated current limit with
configurable overcurrent responses
Optional light load efficiency mode
External synchronization with Smooth Clocking
Dedicated output voltage sensing protection
which remains active even when Enable is low.
Integrated MOSFETs and Bootstrap diode
Operating junction temp: -40oC<Tj<125oC
Small Size 5mmx7mm PQFN
Pb-Free (RoHS Compliant)
The IR38063 PMBus SupIRBuck™ is an easy-to-use,
fully integrated and highly efficient DC/DC regulator
with I2C/PMBus interface. The onboard PWM controller
and MOSFETs make IR38063 a space-efficient
solution, providing accurate power delivery for low
output voltage and high current applications.
The IR38063 can be comprehensively configured via
PMBus and the configuration stored in internal
memory. In addition, PMBus commands allow run-time
control, fault status and telemetry.
The IR38063 can also operate as a standard analog
regulator without any programming and can provide
current and temperature telemetry in an analog format.
APPLICATIONS
Server Applications
Netcomm applications
Embedded telecom Systems
Distributed Point Of Load Architectures
ORDERING INFORMATION
Base
Part
Package
Type
Standard Pack
Orderable Part
Number
Application Description
Form
Quantity
IR38063
QFN
5x7 mm
Tape
&
Reel
4000
IR38063MTRPBF
Standard part, 0.5 Vout
IR38063MBC01TRP
Broadcom SAS34xx, SAS35xx, SAS36xx, SAS37xx
250
111-4190PBF
Standard part, 0.5Vout for Distributors
IR38063
2
Rev 3.14
Dec 15, 2017
BASIC APPLICATION
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
5.5V <Vin<21V
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Vp
P1V8 Track_EN
Figure 1: Typical Application Circuit
Figure 2: Performance Curve
PINOUT DIAGRAM
Figure 3: IR38063 package (Top View) 5mm x 7mm PQFN
Note: Pins 23 and 26 are connected internally
but appear separated externally (refer to
assembly drawing)
IR38063
3
Rev 3.14
Dec 15, 2017
BLOCK DIAGRAM
Figure 4: IR38063 Simplified Block Diagram
IR38063
4
Rev 3.14
Dec 15, 2017
PIN DESCRIPTIONS
PIN #
PIN NAME
PIN DESCRIPTION
1
PVIN
Input voltage for power stage. Bypass capacitors between PVin and PGND should
be connected very close to this pin and PGND. Typical applications use 4 X22 uF
input capacitors and a low ESR, low ESL 0.1uF decoupling capacitor in a
0603/0402 case size. A 3.3nF capacitor may also be used in parallel with these
input capacitors to reduce ringing on the Sw node.
2
Boot
Supply voltage for high side driver. A 0.1uF capacitor should be connected from this
pin to the Sw pin. For PVin > 16V, it is recommended to use a 1 ohm to 4.02 ohm
resistor in series with the boot capacitor.
3
Track_En¯¯¯¯¯¯¯¯¯ *
Pull low to enable tracking function. For normal, non-tracking operation, connect a
100 kOhm resistor from this pin to P1V8. An alternative to using 100kohm to P1V8
is to connect a 750 kohm resistor from Track_En# to LGND when the Track_En#
pin is not used for a tracking function. One of these two options must be used to
disable tracking functionality. The 100kOhm is the preferred method.
4
Vp
Used for sequencing and tracking applications. Leave open if not used.
5
Vsns
Sense pin for OVP and PGood
6
FB
Inverting input to the error amplifier. This pin is connected directly to the output of
the regulator or to the output of the remote sense amplifier, via resistor divider to
set the output voltage and provide feedback to the error amplifier.
7
COMP
Output of error amplifier. An external resistor and capacitor network is typically
connected from this pin to FB to provide loop compensation.
8
RSo
Remote Sense Amplifier Output
9
RS-
Remote Sense Amplifier input. Connect to ground at the load.
10
RS+
Remote Sense Amplifier input. Connect to output at the load.
11
PGood
Power Good status pin. Output is open drain. Connect a pull up resistor from this
pin to VCC. If the power good voltage before VCC UVLO needs to be limited to <
500 mV, use a 49.9K pullup, otherwise a 4.99K pullup will suffice.
12,25
PGND
Power ground. This pin should be connected to the system’s power ground plane.
Bypass capacitors between PVin and PGND should be connected very close to the
PVIN pin (pin 1) and this pin.
13
LGND
Signal ground for internal reference and control circuitry.
14
RT/Sync
In analog mode, use an external resistor from this pin to GND to set the switching
frequency. The resistor should be placed very close to the pin. This pin can also be
used for external synchronization. In digital mode this pin is typically left floating
however a 15K resistor from this pin to GND may be used instead of floating the
pin.
15
EN/FCCM
Enable pin to turn on and off the IC. In analog mode, also serves as a mode pin,
forcing the converter to operate in CCM when pulled to<3.1V.
16
ADDR
A resistor should be connected from this pin to LGnd to set the PMBus address
offset for the device. It is recommended to provide a placement for a 10 nF
capacitor in parallel with the offset resistor. If communication is not needed, as in
analog mode, this pin should be left floating.
IR38063
5
Rev 3.14
Dec 15, 2017
PIN #
PIN NAME
PIN DESCRIPTION
17
SALERT¯¯¯¯¯¯¯
/TMON
SMBus Alert¯¯¯¯ line; open drain SMBALERT# pin. This should be pulled up to 3.3V-
5V with a 1K-5K resistor; this pin provides a voltage proportional to the junction
temperature if digital communication is not needed, as in analog mode.
18
SDA/IMON
SMBus data serial input/output line; This should be pulled up to 3.3V-5V with a 1K-
5K resistor; this pin provides a voltage proportional to the output current if digital
communication is not needed, as in analog mode.
19
SCL/OCSet
SMBus clock line; This should be pulled up to 3.3V-5V with a 1K-5K resistor. This
pin is used to set OC thresholds if digital communication is not needed, as in analog
mode. In analog mode recommend 4.7KΩ for the pull-up to VCC or pull down to
GND when setting the OCP value.
20
P1V8
This is the supply for the digital circuits; bypass with a minimum 2.2uF capacitor to
PGnd. A 10uF capacitor is recommended.
21
Vin
Input Voltage for LDO.
22
VCC
Bias Voltage for IC and driver section, output of LDO. Add 10 uF bypass cap from
this pin to PGnd.
23,26
NC
NC
24
SW
Switch node. This pin is connected to the output inductor.
*Design has simulated the Track_En# input threshold test for a 750K over:
the temperature range of -40 to 150degC,
Vcc of 4.5V to 5.5V
Over all corners of silicon
IR38063
6
Rev 3.14
Dec 15, 2017
ABSOLUTE MAXIMUM RATINGS
Stresses beyond these listed under “Absolute Maximum Ratings” may cause permanent damage to the device.
These are stress ratings only and functional operation of the device at these or any other conditions beyond those
indicated in the operational sections of the specifications are not implied.
PVin, Vin
-0.3V to 25V
VCC
-0.3V to 6V
P1V8
-0.3V to 2 V
SW
-0.3V to 25V (DC), -4V to 25V (AC, 100ns)
BOOT
-0.3V to 31V
PGD, other Input/output pins
-0.3V to 6V (Note 1)
BOOT to SW
-0.3V to 6V (DC), -0.3V to 6.5V (AC, 100ns)
PGND to GND, RS- to GND
-0.3V to + 0.3V
THERMAL INFORMATION
Junction to Case Thermal Resistance ƟJC-TOP
30oC/W
Junction to Ambient Thermal Resistance ƟJA
13.8oC/W
Junction to PCB Thermal Resistance ƟJ-PCB
2.05oC/W
Storage Temperature Range
-55°C to 150°C
Junction Temperature Range
-40°C to 150°C
(Voltages referenced to GND unless otherwise specified)
Note 1: Must not exceed 6V.
IR38063
7
Rev 3.14
Dec 15, 2017
ELECTRICAL SPECIFICATIONS
RECOMMENDED OPERATING CONDITIONS
SYMBOL
DEFINITION
MIN
MAX
UNITS
PVin
Input Bus Voltage
1.2
21*
V
Vin
LDO supply voltage
5.5
21
VCC
LDO output/Bias supply voltage
4.5
5.5
Boot to SW
High Side driver gate voltage
4.5
5.5
VO
Output Voltage
0.5
0.875*PVin
IO
Output Current
0
25
A
Fs
Switching Frequency
225
1650
kHz
TJ
Junction Temperature
-40
125
°C
* SW Node must not exceed 25V
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, these specification apply over, 1.5V < PVin < 21V, 4.5V < Vcc < 5.5, 0C < TJ <
125C.
Typical values are specified at TA = 25C.
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
MOSFET Rds(on)
Top Switch
Rds(on)_Top
VBoot – VSW = 5V, ID = 25A, Tj
= 25°C
2.7
4
5.6
mΩ
Bottom Switch
Rds(on)_Bot
Vcc =5V, ID = 25A, Tj = 25°C
1.11
1.58
2.05
Reference Voltage
Accuracy
00C<Tj<850C
1.25V<VFB<2.555V
VOUT_SCALE_LOOP=1;
-1
+1
%
0.75V<VFB<1.25V
VOUT_SCALE_LOOP=1;
-0.75
+0.75
0.45V<VFB<0.75V
VOUT_SCALE_LOOP=1;
-0.5
+0.5
%
Accuracy
-400C<Tj<1250C
1.25V<VFB<2.555V
VOUT_SCALE_LOOP=1;
-1.6
+1.6
%
0.75V<VFB<1.25V
VOUT_SCALE_LOOP=1;
-1.0
+1.0
%
0.45V<VFB<0.75V
VOUT_SCALE_LOOP=1;
-2.0
+2.0
%
IR38063
8
Rev 3.14
Dec 15, 2017
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
Supply Current
PVin range (using
external Vcc=5.1V)
1.2-
21
V
Vin range (using
internal LDO)
Fsw=600kHz
5.3-
21
V
Fsw=1.5MHz
5.5-
21
Vin range (when
Vin=Vcc)
4.5
5.1
5.5
V
Vin Supply Current
(Standby) (internal Vcc)
Iin(Standby)
Enable low, No Switching,
Vin=21V, low power mode
enabled
2.7
4
mA
Vin Supply Current
(Dyn)(internal Vcc)
Iin(Dyn)
Enable high, Fs = 600kHz,
Vin=21V
39
50
mA
VCC Supply Current
(Standby)(external Vcc)
Icc(Standby)
Enable low, No Switching,
Vcc=5.5V, low power mode
enabled
2.7
5
mA
VCC Supply Current
(Dyn)(external Vcc)
Icc(Dyn)
Enable high, Fs = 600kHz,
Vcc=5.5V
39
50
mA
Under Voltage Lockout
VCC – Start –
Threshold
VCC_UVLO_Start
VCC Rising Trip Level
4.0
4.2
4.4
V
VCC – Stop –
Threshold
VCC_UVLO_Stop
VCC Falling Trip Level
3.7
3.9
4.1
PVin-Start-Threshold
PVin_UVLO_Start
PVin Rising Trip Level
0.85
0.95
1.05
V
PVin-Stop-Threshold
PVin_UVLO_Stop
PVin Falling Trip Level
0.35
0.45
0.55
Enable – Start –
Threshold
Enable_UVLO_Start
Supply ramping up
1.14
1.2
1.36
V
Enable – Stop –
Threshold
Enable_UVLO_Stop
Supply ramping down
0.9
1.0
1.06
Enable leakage current
Ien
Enable=5.5V
1
uA
Oscillator
Rt current (analog mode
only)
Rt pin voltage < 1.1V
98
100
102
uA
Frequency Range
FS
Rt=1.54K
360
400
440
kHz
Rt=3.83K
540
600
660
Rt=11.8K
1350
1500
1650
Min Pulse Width
Dmin (ctrl)
Note 2
35
50
ns
Fixed Off Time
Note 2 Fs=1.5MHz
100
150
ns
Max Duty Cycle
Dmax
Fs=400kHz
86.5
87.5
88.5
%
Sync Frequency Range
Note 2
225
1650
kHz
Sync Pulse Duration
100
200
ns
IR38063
9
Rev 3.14
Dec 15, 2017
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
Sync Level Threshold
High
2.1
V
Low
1
Error Amplifier
Input Offset Voltage
Vos_Vp
VFb – Vp, Vp = 0.5V
-1.5
+1.5
%
Input Bias Current
IFb(E/A)
-0.5
+0.5
µA
Input Bias Current
IVp(E/A)
0
7
µA
Sink Current
Isink(E/A)
0.6
1.1
1.8
mA
Source Current
Isource(E/A)
8
13
25
mA
Slew Rate
SR
Note 2
7
12
20
V/µs
Maximum Voltage
Vmax(E/A)
2.8
3.9
4.3
V
Minimum Voltage
Vmin(E/A)
100
mV
Common Mode Voltage
Vcm_Vp
Note 2
0
2.555
V
Remote Sense Differential Amplifier
Unity Gain Bandwidth
BW_RS
Note 2
3
6.4
MHz
DC Gain
Gain_RS
Note 2
110
dB
Offset Voltage
Offset_RS
0.5V<RS+<2.555V, 4kOhm
load
270C<Tj<850C
-1.6
0
1.6
mV
0.5V<RS+<2.555V, 4kOhm
load
-400C<Tj<1250C
-3
3
Source Current
Isource_RS
V_RSO=1.5V, V_RSP=4V
11
16
mA
Sink Current
Isink_RS
0.4
1
2
mA
Slew Rate
Slew_RS
Note 2, Cload = 100pF
2
4
8
V/µs
RS+ input impedance
Rin_RS+
36
55
74
Kohm
RS- input impedance
Rin_RS-
Note 2
36
55
74
Kohm
Maximum Voltage
Vmax_RS
V(VCC) – V(RS+)
0.5
1
1.5
V
Minimum Voltage
Min_RS
4
20
mV
Bootstrap Diode
Forward Voltage
I(Boot) = 40mA
150
300
450
mV
Switch Node
SW Leakage Current
lsw
SW = 0V, Enable = 0V
1
µA
Isw_En
SW=0; Enable= 2V
18
Internal Regulator (VCC/LDO)
Output Voltage
VCC
Vin(min) = 5.5V, Io=0mA,
Cload = 10uF
4.8
5.15
5.4
V
Vin(min) = 5.5V, Io=70mA,
Cload = 10uF
4.5
4.99
5.2
IR38063
10
Rev 3.14
Dec 15, 2017
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
VCC dropout
VCC_drop
Io=0-70mA, Cload = 10uF,
Vin=5.1V
0.7
V
Short Circuit Current
Ishort
110
mA
Internal Regulator (P1V8)
Output Voltage
P1V8
Vin(min) = 4.5V, Io = 0‐
10mA, Cload = 2.2uF
1.795
1.83
1.905
V
Adaptive On time Mode
AOT Threshold
High
En/Fccm
3.8
3.9
4.1
V
Low
3.1
3.6
3.8
Zero-crossing
comparator threshold
ZC_Vth
-4
-1
2
mV
Zero-crossing
comparator delay
ZC_Tdly
8/Fs
s
FAULTS
Power Good
Power Good High
threshold
Power_Good_High
Vsns rising,
VOUT_SCALE_LOOP=1,
Track_EN floating,
VDAC1=0.5V
91
%VDAC1
Vsns rising,
VOUT_SCALE_LOOP=1,
Track_EN low, Vp=0.5V
90
%Vp
Power Good Low
Threshold
Power_Good_Low
Vsns falling,
VOUT_SCALE_LOOP=1,
Track_EN floating,
VDAC1=0.5V
86
%VDAC1
Vsns falling,
VOUT_SCALE_LOOP=1,
Track_EN low, Vp=0.5V
84.5
%Vp
Power Good High
Threshold Rising Delay
TPDLY
Vsns rising, Vsns >
Power_Good_High
0
ms
Power Good Low
Threshold Falling delay
VPG_low_Dly
Vsns falling, Vsns <
Power_Good_Low
150
175
200
us
Tracker Comparator
Upper Threshold
VPG(tracker_
upper)
Vp Rising,
VOUT_SCALE_LOOP=1,
Track_EN low, Vsns=Vp
0.38
0.4
0.42
V
Tracker Comparator
Lower Threshold
VPG(tracker_
lower)
Vp Falling,
VOUT_SCALE_LOOP=1,
Track_EN low, Vsns=Vp
0.28
0.3
0.32
V
PGood Voltage Low
PG (voltage)
IPGood = -5mA
0.5
V
Over Voltage Protection (OVP)
OVP Trip Threshold
OVP (trip)
Vsns rising,
VOUT_SCALE_LOOP=1,
Track_EN floating,
VDAC1=0.5V
115
121
125
%VDAC1
IR38063
11
Rev 3.14
Dec 15, 2017
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
Vsns rising,
VOUT_SCALE_LOOP=1,
Track_EN low, Vp=0.5V
115
120
125
%Vp
OVP comparator
Hysteresis
OVP (hyst)
Vsns falling,
VOUT_SCALE_LOOP=1,
Track_EN floating,
VDAC1=0.5V
2.5
4.5
5.8
%OVP
(trip)
Vsns Falling,
VOUT_SCALE_LOOP=1,
Track_EN low, Vp=0.5V
2.5
4.5
5.8
%OVP
(trip)
OVP Fault Prop Delay
OVP (delay)
Vsns rising, Vsns-
OVP(trip)>200 mV
200
ns
Over-Current Protection
OC Trip Current
ITRIP
Analog mode: OCSet pulled
high to VCC via resistor.
VCC = 5.05V, Tj=250C
30
33
36
A
Analog mode: OCSet left
floating.
VCC = 5.05V, Tj=250C
24.5
27
29.5
A
Analog mode: OCSet pulled
low to GND via resistor.
VCC = 5.05V, Tj=250C
18.3
21
23.7
A
OCset Current
Temperature coefficient
OCSET(temp)
-400C to 1250C, VCC=5.05V,
Note 2
5900
ppm/°C
Hiccup blanking time
Tblk_Hiccup
Note 2
20
ms
Thermal Shutdown
Thermal Shutdown
Note 2
145
°C
Hysteresis
Note 2
25
°C
Input Over-Voltage Protection
PVin overvoltage
threshold
PVinOV
22
23.7
25
V
PVin overvoltage
Hysteresis
PVin ov hyst
2.4
V
MONITORING AND REPORTING
Bus Speed1
100
400
kHz
Iout & Vout filter
78
Hz
Iout & Vout Update rate
31.2
5
kHz
Vin & Temperature filter
78
Hz
Vin & Temperature
update rate
31.2
5
kHz
Output Voltage Reporting
Resolution
NVout
Note 2
1/256
V
IR38063
12
Rev 3.14
Dec 15, 2017
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
Lowest reported Vout
Vomon_low
Vsns=0V
0
V
Highest reported Vout
Vomon_high
VOUT_SCALE_LOOP=1,
Vsns=3.3V
3.3
V
VOUT_SCALE_LOOP=0.5,
Vsns=3.3V
6.6
V
VOUT_SCALE_LOOP=0.25,
Vsns=3.3V
13.2
V
VOUT_SCALE_LOOP=0.125
, Vsns=3.3V
26.4
V
Vout reporting accuracy
00C to 850C, 4.5V<Vcc<5.5V,
1V<Vsns≤ 1.5V
VOUT_SCALE_LOOP=1
+/-
0.6
%
00C to 850C, 4.5V<Vcc<5.5V,
Vsns> 1.5V
VOUT_SCALE_LOOP=1
+/-1
00C to 1250C,
4.5V<Vcc<5.5V, Vsns>0.9V
VOUT_SCALE_LOOP=1
+/-
1.5
00C to 1250C,
4.5V<Vcc<5.5V,
0.5V<Vsns<0.9V
VOUT_SCALE_LOOP=1
+/-3
Iout Reporting
Resolution
NIout
Note 2
62.5
mA
Iout (digital) monitoring
Range
Iout_dig
0
37.5
A
Iout_dig Accuracy
00C to 1250C,
4.5V<Vcc<5.5V, 5A < Iout <2
5A
+/-5
%
Imon (analog) voltage
Imon
0.3
1.1
V
Imon ( analog) accuracy
00C to 1250C,
4.5V<Vcc<5.5V, 5A < Iout <2
5A, -30uA< I_IMON<30uA
+/-
1.5
A
Temperature Reporting
Resolution
NTmon
Note 2
1
°C
Temperature Monitoring
(digital) Range
Tmon_dig
-40
150
°C
Temperature Monitoring
(digital) accuracy
-400C to 1250C,
4.5V<Vcc<5.5V, -30uA<
I_TMON<30uA; Guaranteed
by char
-5
5
°C
Analog monitoring
range
Tmon
-400C to 1500C
500
1100
mV
Analog Monitoring
-400C to 1250C,
-9
9
°C
IR38063
13
Rev 3.14
Dec 15, 2017
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNIT
Accuracy
4.5V<Vcc<5.5V, -30uA<
I_TMON<30uA, Note 2
Temperature coefficient
2.27
mV/°C
Thermal shutdown
hysteresis
Note 2
25
°C
Input Voltage Reporting
Resolution
NPVin
Note 2
1/32
V
Monitoring Range
PMBVinmon
0
21
V
Monitoring accuracy
00C to 850C, 4.5V<Vcc<5.5V,
PVin>10V
-1.5
1.5
%
-400C to 1250C,
4.5V<Vcc<5.5V, PVin>14V
-1.5
1.5
-400C to 1250C,
4.5V<Vcc<5.5V,
6V<PVin<14V
-3
3
PMBus Interface Timing Specifications
SMBus Operating
frequency
FSMB
400
kHz
Bus Free time between
Start and Stop condition
TBUF
1.3
us
Hold time after
(Repeated) Start
Condition. After this
period, the first clock is
generated.
THD:STA
0.6
us
Repeated start
condition setup time
TSU:STA
0.6
us
Stop condition setup
time
TSU:STO
0.6
us
Data Rising Threshold
1.339
1.766
V
Data Falling Threshold
1.048
1.495
V
Clock Rising Threshold
1.339
1.766
V
Clock Falling Threshold
1.048
1.499
V
Data Hold Time
THD:DAT
300
900
ns
Data Setup Time
TSU:DAT
100
ns
Clock low time out
TTIMEOUT
25
35
ms
Clock low period
TLOW
1.3
us
Clock High Period
THIGH
0.6
50
us
Notes
2. Guaranteed by design but not tested in production
3. Guaranteed by statistical correlation, but not tested in production
IR38063
14
Rev 3.14
Dec 15, 2017
TYPICAL APPLICATION DIAGRAMS
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
5.5V <PVin<21V
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
P1V8
Vp
Track_EN
Figure 5: Using the internal LDO, digital mode, Vo < 2.555V
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Vp
5.5V <PVin<21V
P1V8 Track_EN
Recommend R2 =499Ω
R2
Figure 6: Using the internal LDO, digital mode, Vo > 2.555V
IR38063
15
Rev 3.14
Dec 15, 2017
TYPICAL APPLICATION DIAGRAMS
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
5.5V < PVin< 21V
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Vp
P1V8 Track_EN
Figure 7: Using the internal LDO, analog mode, Vo<2.555V
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
1.2V <PVin<21V
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Vp
Vcc=5V P1V8 Track_EN
Figure 8: Using external Vcc, digital mode, Vo<2.555V
IR38063
16
Rev 3.14
Dec 15, 2017
TYPICAL APPLICATION DIAGRAMS
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
PVin=Vin=Vcc= 5V
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Vp
P1V8 Track_EN
Figure 9: Single 5V application, digital mode, Vo<2.555V
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
Rt/SYNC
5.5V <PVin<21V
PVin
Vin
En/FCCM
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Track_EN
P1V8 Vp
Figure 10: Using the internal LDO, digital mode, tracking mode
IR38063
17
Rev 3.14
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TYPICAL OPERATING CHARACTERISTICS (-40°C TO +125°C)
IR38063
18
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING CHARACTERISTICS (-40°C TO +125°C)
IR38063
19
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING CHARACTERISTICS (-40°C TO +125°C)
IR38063
20
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING CHARACTERISTICS (-40°C TO +125°C)
IR38063
21
Rev 3.14
Dec 15, 2017
TYPICAL EFFICIENCY AND POWER LOSS CURVES
PVin = Vin = 12V, VCC = Internal LDO, Io=0-25A, Fs= 600kHz, Room Temperature, No Air Flow. Note that the
losses of the inductor, input and output capacitors are also considered in the efficiency and power loss curves.
The table below shows the indicator used for each of the output voltages in the efficiency measurement.
VOUT (V)
LOUT (uH)
P/N
DCR (mΩ)
0.6
0.15
HCB178380D-151 (Delta)
0.15
0.8
0.15
HCB178380D-151 (Delta)
0.15
1
0.15
HCB178380D-151 (Delta)
0.15
1.2
0.215
PCDC1008-R215EMO (Cyntec)
0.29
1.5
0.215
PCDC1008-R215EMO (Cyntec)
0.29
1.8
0.215
PCDC1008-R215EMO (Cyntec)
0.29
2.5
0.3
59PR987N (Vitec)
0.29
3.3
0.47
WE-HCM 744 309 047 (Wurth)
0.15
5
0.47
WE-HCM 744 309 047 (Wurth)
0.15
IR38063
22
Rev 3.14
Dec 15, 2017
TYPICAL EFFICIENCY AND POWER LOSS CURVES
PVin = Vin = VCC = 5V, Io=0-25A, Fs= 600kHz, Room Temperature, No Air Flow. Note that the losses of the
inductor, input and output capacitors are also considered in the efficiency and power loss curves. The table
below shows the indicator used for each of the output voltages in the efficiency measurement.
VOUT (V)
LOUT (uH)
P/N
DCR (mΩ)
0.6
0.15
HCB178380D-151 (Delta)
0.15
0.8
0.15
HCB178380D-151 (Delta)
0.15
1
0.15
HCB178380D-151 (Delta)
0.15
1.2
0.15
HCB178380D-151 (Delta)
0.15
1.5
0.15
HCB178380D-151 (Delta)
0.15
1.8
0.215
PCDC1008-R215EMO (Cyntec)
0.29
2.5
0.215
PCDC1008-R215EMO (Cyntec)
0.29
3.3
0.215
PCDC1008-R215EMO (Cyntec)
0.29
IR38063
23
Rev 3.14
Dec 15, 2017
THEORY OF OPERATION
DESCRIPTION
The IR38063 is a 25A synchronous buck regulator
with a selectable digital interface and an externally
compensated fast, analog, PWM voltage mode
control scheme to provide good noise immunity as
well as fast dynamic response in a wide variety of
applications. At the same time, enabling the digital
PMBus interface allows complete configurability of
output setting and fault functions, as well as
telemetry.
The switching frequency is programmable to 1.5MHz
and provides the capability of optimizing the design
in terms of size and performance.
IR38063 provides precisely regulated output voltage
from 0.5V to 0.875*PVin programmed via two
external resistors or digitally through PMBus
commands. The IR38063 operates with an internal
bias supply (LDO), typically 5.2V. This allows
operation with a single supply. The output of this
LDO is brought out at the Vcc pin and may be
bypassed to the system power ground with a 10 uF
decoupling capacitor. The Vcc pin may also be
connected to the Vin pin, and an external Vcc supply
between 4.5V and 5.5V may be used, allowing an
extended operating bus voltage (PVin) range from
1.2V to 21V.
The device utilizes the on-resistance of the low side
MOSFET (synchronous MOSFET) as current sense
element. This method enhances the converter’s
efficiency and reduces cost by eliminating the need
for external current sense resistor.
IR38063 includes two low Rds(on) MOSFETs using
IR’s HEXFET technology. These are specifically
designed for high efficiency applications.
DEVICE POWER-UP AND INITIALIZATION
During the power-up sequence, when Vin is brought
up, the internal LDO converts it to a regulated 5.2V
at Vcc. There is another LDO which further converts
this down to 1.8V to supply the internal digital
circuitry. An under-voltage lockout circuit monitors
the voltage of VCC pin and the P1V8 pin, and holds
the Power-on-reset (POR) low until these voltages
exceed their thresholds and the internal 48 MHz
oscillator is stable. When the device comes out of
reset, it initializes a multiple times programmable
memory (MTP) load cycle, where the contents of the
MTP are loaded into the working registers. Once the
registers are loaded from MTP, the designer can use
PMBus commands to re-configure the various
parameters to suit the specific VR design
requirements if desired, irrespective of the status of
Enable.
In the default configuration, power conversion is
enabled only when the En/FCCM pin voltage
exceeds its undervoltage threshold, the PVin bus
voltage exceeds its undervoltage threshold, the
contents of the MTP have been fully loaded into the
working registers and the device address has been
read. The initialization sequence is shown in Figure
Figure 11.
IR38063 provides additional options to enable the
device power conversion through software and
these options may be configured to override the
default by using the I2C interface or PMBus, if used
in digital mode. For further details see the UN0060
IR3806x PMBus commandset user note.
PVIN=VIN
VCC
P1V8
UVOK
clkrdy
POR
Initialization
done
Enable
Vout
Figure 11: IR38063 Initialization sequence
ANALOG AND DIGITAL MODE OPERATION
The IR38063 has 2 7-bit registers that are used to
set the base I2C address and base PMBus address
of the device, as shown below in Table 1.
IR38063
24
Rev 3.14
Dec 15, 2017
Table 1: Registers used to set device base
address
Register
Description
I2c_address[6:0]
The chip I2C address. An
address of 0 will disable
communication
Pmbus_address[6:0]
The chip PMBus address.
An address of 0 will disable
communication.
In addition, a resistor may be connected between
the ADDR and LGND pins to set an offset from the
default preconfigured I2C address (0x10) /PMBus
address (0x40) in the MTP. Up to 16 different offsets
can be set, allowing 16 IR38063 devices with unique
addresses in a single system. This offset, and
hence, the device address, is read by the internal 10
bit ADC during the initialization sequence. It is
recommended that a layout placement be provided
for a 10nF capacitor in parallel with this offset
resistor. On systems that have more noise, this
capacitor will help to prevent the 10 bit ADC from
incorrectly reading the offset and calculating the
wrong address offset.
Table 2 below provides the resistor values needed
to set the 16 offsets from the base address.
Table 2 : Address offset vs. External
Resistor(RADDR)
ADDR Resistor
(Ohm)
Address Offset
499
+0
1050
+1
1540
+2
2050
+3
2610
+4
3240
+5
3830
+6
4530
+7
5230
+8
6040
+9
6980
+10
7870
+11
8870
+12
9760
+13
10700
+14
11800
+15
The device will then respond to I2C/PMbus
commands sent to this address. This mode in which
digital communication to and from the device is
allowed following the MTP load sequence is referred
to as the digital mode of operation. However, if the
ADDR pin is left floating, the IR38063 disables
digital communication and will not respond to
commands sent over the bus. In fact, the 3 pins
used for digital communication are dual purpose
pins which get reconfigured for analog applications if
ADDR is left floating. Hence, in the analog mode,
the default configuration parameters loaded in to the
working registers from the MTP during the
initialization sequence cannot be modified on the fly,
and the device can be operated similar to an analog
only SupIRBuck such as IR3847.
BUS VOLTAGE UVLO
In the analog mode of operation or with the default
configuration, if the input to the Enable pin is derived
from the bus voltage by a suitably programmed
resistive divider, it can be ensured that the IR38063
does not turn on until the bus voltage reaches the
desired level as shown in Figure 12. Only after the
bus voltage reaches or exceeds this level and
voltage at the Enable pin exceeds its threshold
(typically 1.2V) IR38063 will be enabled. Therefore,
in addition to being a logic input pin to enable the
IR38063, the Enable feature, with its precise
threshold, also allows the user to override the
default 1 V Under-Voltage Lockout for the bus
voltage (PVin). This is desirable particularly for high
output voltage applications, where we might want
the IR38063 to be disabled at least until PVin
exceeds the desired output voltage level.
Alternatively, the default 1 V PVin UVLO threshold
may be reconfigured/overridden using the VIN_ON
and VIN_OFF PMBus commands. It should be noted
that while the input voltage is also fed to an ADC
through a 21:1 internal resistive divider, the digitized
IR38063
25
Rev 3.14
Dec 15, 2017
input voltage is used only for the purposes of
reporting the input voltage through the READ_VIN
PMBUs command and has no impact on the bus
voltage UVLO, input overvoltage faults and input
undervoltage warnings, all of which are implemented
by using analog comparators to compare the input
voltage to the corresponding thresholds
programmed by the PMBus commands VIN_ON,
VIN_OFF, VIN_OV_FAULT_LIMIT and
VIN_UV_WARN_LIMIT respectively. The bus
voltage reading as reported by READ_VIN has no
effect on the input feedforward function either.
Vcc
PVin
DAC2 (Reference DAC)
EN
> 1.2V
1.2V
EN_UVLO_START
10.2V
12V
1 V
Figure 12: Normal Start up, device turns on when
the bus voltage reaches 10.2V
A resistor divider is used at EN pin from PVin to turn
on the device at 10.2V.
Vcc
PVin=Vin
EN > 1.2V
Vp
DAC2 (Reference DAC)
Figure 13: Recommended startup for Normal
operation
Vcc
PVin=Vin
EN > 1.2V
Vp
DAC2 (Reference DAC)
Figure 14: Recommended startup for sequencing
operation (ratiometric or simultaneous)
Vcc
PVin=Vin
EN > 1.2V
Vp
Track_En 0V
DAC2 (Reference DAC)
Figure 15: Recommended startup for memory
tracking operation (DDR-VTT)
Figure 13 shows the recommended startup
sequence for the normal (non-tracking, non-
sequencing) operation of IR38063, when Enable is
used as logic input. In this operating mode, a 100
kOhm resistor is connected from Track_En¯¯¯¯¯¯¯¯¯ to
P1V8. Figure 14 shows the recommended startup
sequence for sequenced operation of IR38063 with
Enable used as logic input. For this mode
of operation also, a 100 kOhm resistor is connected
from Track_En¯¯¯¯¯¯¯¯¯ to P1V8. Figure 15 shows the
recommended startup sequence for tracking
operation of IR38063 with Enable used as logic
input. For this mode of operation, Track_En¯¯¯¯¯¯¯¯¯ should
be connected to LGND.
PRE-BIAS STARTUP
IR38063 is able to start up into pre-charged output,
which prevents oscillation and disturbances of the
output voltage.
IR38063
26
Rev 3.14
Dec 15, 2017
The output starts in asynchronous fashion and
keeps the synchronous MOSFET (Sync FET) off
until the first gate signal for control MOSFET (Ctrl
FET) is generated. Figure 16 shows a typical Pre-
Bias condition at start up. The sync FET always
starts with a narrow pulse width (12.5% of a
switching period) and gradually increases its duty
cycle with a step of 12.5%, with 16 cycles at each
step, until it reaches the steady state value. Figure
17 shows the series of 16x8 startup pulses.
Vo
[V]
[Time]
Pre-Bias
Voltage
Figure 16: Pre-Bias startup
... ... ...
HDRv
... ... ...
16 End of
PB
LDRv
12.5% 25%87.5%
16 ...
...
...
...
Figure 17: Pre-Bias startup pulses
SOFT-START (REFERENCE DAC RAMP)
IR38063 has an internal soft starting DAC to control
the output voltage rise and to limit the current surge
at the start-up. In the default configuration and in
analog mode, to ensure correct start-up, the DAC
sequence initiates only after power conversion is
enabled when the En/FCCM pin voltage exceeds its
undervoltage threshold, the PVin bus voltage
exceeds its undervoltage threshold and the contents
of the MTP have been fully loaded into the working
registers. In analog mode and in the default
configuration, the reference DAC signal linearly rises
to 0.5V in 2 ms. Figure 18 shows the waveforms
during soft start In digital mode, the reference DAC
soft-start may be delayed from time power
conversion is enabled. The range for this
programmable delay is 0ms to 127 ms, and the
resolution is 1 ms. Further, the soft start time may be
configured from 1ms to 127 ms with 1 ms resolution.
For more details on the PMBus commands
TON_DELAY and TON_RISE used to program the
startup sequence, please see the UN0060 IR3806x
PMBus commandset user note.
Note however, that a shorter Ton_Rise can lead to a
slight overshoot on the output voltage during startup.
Infineon recommends using a rise time that would
limit the soft start rate to <0.4mV/us. Also, it is
recommended that the system designer should
verify in the actual design that the selected rise time
keeps the overshoot within limits acceptable to the
system.
Reference
DAC
Vout
t1
0.5V
Internal Enable
t3
t2
Ton_delay Ton_rise
Figure 18: DAC2 (VREF) Soft start
During the startup sequence the over-current
protection (OCP) and over-voltage protection (OVP)
are active to protect the device for any short circuit
or over voltage condition.
OPERATING FREQUENCY
In analog mode, the switching frequency can be
programmed between 306kHz – 1500kHz by
connecting an external resistor from Rt pin to LGnd.
This frequency is set during the initialization
sequence, when the 10 bit ADC reads the voltage at
the RT pin. It should be noted that after the
initialization sequence is complete, the ADC no
longer reads the voltage at the ADC pin, so
changing the resistor on the fly after initialization will
IR38063
27
Rev 3.14
Dec 15, 2017
not affect the switching frequency. Table 3 tabulates
the oscillator frequency versus Rt.
Table 3: Switching Frequency (Fs) vs. External
Resistor(Rt)
Rt
Resistor
(Ohm)
F s(kHz)
499
306
1050
356
1540
400
2050
444
2610
500
3240
550
3830
600
4530
706
5230
750
6040
800
6980
923
7870
1000
8870
1091
9760
1200
10700
1333
11800
1500
In the digital mode, the default switching frequency
is configured to be 607 kHz, and is programmable
from 250 kHz to 1500 kHz. The user can override
this using the FREQUENCY_SWITCH PMBus
command. In the digital mode of operation no
resistor is used or needed on the Rt/Sync pin. For
best telemetry accuracy, it is recommended that the
following switching frequencies be avoided: 250
kHz, 300 kHz, 400 kHz, 500 kHz, 600 kHz, 750 kHz,
800 kHz, 1 MHz, 1.2 MHz and 1.5 MHz. Instead,
Infineon suggests using the following values 251
kHz, 302 kHz, 403 kHz, 505 kHz, 607 kHz, 762 kHz,
813 kHz, 978 kHz, 1171 kHz and 1454 kHz
respectively.
EXTERNAL SYNCHRONIZATION
IR38063 incorporates an internal phase lock loop
(PLL) circuit which enables synchronization of the
internal oscillator to an external clock. This function
is important to avoid sub-harmonic oscillations due
to beat frequency for embedded systems when
multiple point-of-load (POL) regulators are used. A
multi-function pin, Rt/Sync, is used to connect the
external clock. In the analog mode, if the external
clock is applied before the initialization sequence is
done, the internal ADC cannot read the value of the
RT resistor and hence, for proper operation, it is
mandatory that the external clock remains applied. If
the synchronization clock is then lost after
initialization, the IR38063 will treat this as a
symptom of a failure in the system and disable
power conversion. Therefore, for such applications,
where the switching frequency is always determined
by an external synchronization clock, the Rt/Sync
pin can be connected to the external clock signal
solely and no other resistor is needed. If the external
clock is applied after the initialization sequence, the
IR38063 treats this as an application where the
converter switching frequency needs to toggle
between the external clock frequency and the
internal free-running frequency, and in the analog
mode, an external resistor from Rt/Sync pin to LGnd
is required to set the free-running frequency. In the
digital mode, the resistor is not needed because the
free running frequency is set in an internal register.
When an external clock is applied to Rt/Sync pin
after the converter runs in steady state with its free-
running frequency, a transition from the free-running
frequency to the external clock frequency will
happen. This transition is to gradually make the
actual switching frequency equal to the external
clock frequency, no matter which one is higher.
When the external clock signal is removed from
Rt/Sync pin, the switching frequency is also changed
to free-running gradually.
IR38063
28
Rev 3.14
Dec 15, 2017
SW
SYNC
...
...
Gradually change
Fs1
Fs2
Fs1
Free Running
Frequency Synchronize to the
external clock Return to free-
running freq
Gradually change
Figure 19: Timing Diagram for Synchronization
to the external clock (Fs1>Fs2 or Fs1<Fs2)
An internal circuit is used to change the PWM ramp
slope according to the clock frequency applied on
Rt/Sync pin. Even though the frequency of the
external synchronization clock can vary in a wide
range, the PLL circuit will make sure that the ramp
amplitude is kept constant, requiring no adjustment
of the loop compensation. PVin variation also affects
the ramp amplitude, which will be discussed
separately in Feed-Forward section.
It must be noted here that in analog mode, since the
voltage at the Rt/Sync pin is read by the ADC at
startup special care must be taken if a low
impedance system clock is used for synchronization
and is applied before the initialization sequence is
done. The circuit shown in Figure 20 below shows
how this may be done using a diode-capacitor
combination. This couples the clock edges to the
Rt/Sync pin while not loading the Rt/SYNC pin with
the impedance of the synchronization clock, and
thus not affecting the Rt voltage read by the ADC at
startup.
Figure 20: Synchronizing a low impedance clock
in analog mode
It must be re-iterated that this is not a concern in
digital mode and the clock may be directly applied to
the Rt/Sync pin.
SHUTDOWN
In the default configuration, IR38063 can be
shutdown by pulling the Enable pin below its 1.0V
threshold. During shutdown the high side and the
low side drivers are turned off. By default, the device
exhibits an immediate shutdown with no delay and
no soft stop.
Alternatively, in digital mode, the part may be
configured to allow shutdown using the
OPERATION PMBus command as well.
CURRENT SENSING, TELEMETRY AND
OVER CURRENT PROTECTION
Current sensing for both, telemetry as well as
overcurrent protection is done by sensing the
voltage across the sync FET RDson. This method
enhances the converter’s efficiency, reduces cost by
eliminating a current sense resistor and any
minimizes sensitivity to layout related noise issues.
A novel, patented scheme allows reconstruction of
the average inductor current from the voltage
sensed across the Sync FET Rdson. It should be
noted here that it is this reconstructed average
inductor current that is digitized by the ADC and
used for output current reporting as well as for
overcurrent warning, the threshold for which may be
set using the IOUT_OC_WARN_LIMIT command.
IR38063
29
Rev 3.14
Dec 15, 2017
The current is reported in 1/16A resolution using the
READ_IOUT PMBus command.
The Over current (OC) fault protection circuit also
uses the voltage sensed across the RDS(on) of the
Synchronous MOSFET; however, the protection
mechanism relies on a fast comparator to compare
the sensed signal to the overcurrent threshold and
does not depend on the ADC or reported current. In
the analog mode of operation, the current limit can
be set to one of three possible settings by floating
the OCSet pin, or pulling it up to Vcc or pulling it
down to PGnd. The current limit scheme in the
IR38063 uses an internal temperature compensated
current source that has the same temperature
coefficient as the RDS(on) of the Synchronous
MOSFET. As a result, the over-current trip threshold
remains almost constant over temperature.
Over Current Protection circuitry senses the inductor
current flowing through the Synchronous FET closer
to the valley point. The OCP circuit samples this
current for 75 ns typically after the rising edge of the
PWM set pulse which is an internal signal that has a
width of 12.5% of the switching period. The PWM
pulse that turns on the high side FET starts at the
falling edge of the PWM set pulse. This makes valley
current sense more robust as current is sensed
close to the bottom of the inductor downward slope
where transient and switching noise is low. This
helps to prevent false tripping due to noise and
transients.
The actual DC output current limit point will be
greater than the valley point by an amount equal to
approximately half of the peak to peak inductor
ripple current. The current limit point will be a
function of the inductor value, input voltage, output
voltage and the frequency of operation. On equation
1, ILIMIT is the value set when configuring the OCP
value. The user should account for the inductor
ripple to obtain the actual DC output current limit.
2i
II LIMITOCP
(1)
IOCP = DC current limit hiccup point
ILIMIT = Current Limit Valley Point
Δi = Inductor ripple current
0
IL
0
HDrv
Current Limit
0
LDrv
...
...
0
PGood
Hiccup
Tblk_Hiccup
20 ms
Figure 21: Timing Diagram for Current Limit
Hiccup
In the default configuration and in analog mode, if
the overcurrent detection trips the OCP comparator,
the IR38063 goes into a hiccup mode after 8 current
limit cycles. In order to put the part into hiccup, a
total of 8 current limited cycles in needed. If the
device sees 6 consecutive cycles without an
overcurrent condition it resets the 8 cycle counter.
The hiccup is performed by de-asserting the internal
Enable signal to the analog and power conversion
circuitry and holding it low for 20 ms.
Following this, the OCP signal resets and the
converter recovers. After every hiccup cycle, the
converter stays in this mode until the overload or
short circuit is removed. This behavior is shown in
Figure 21.
It should be noted that on some units, a false OCP
maybe experienced during IR38063 device start-up
due to noise. The part will ride through this false
OCP due to the pulse by pulse current limiting
feature of the IR38063 and successfully ramp to the
correct output voltage. However, Infineon
recommends sending a PMBUS Clear_Faults
command after start-up to reset the PMBUS SAlert#
to a high and to clear the PMBUS status register for
faults.
IR38063
30
Rev 3.14
Dec 15, 2017
Note that the IR38063 allows the user to override
the default overcurrent threshold using the PMBus
command IOUT_OC_FAULT_LIMIT.
Also, using the PMBus command
IOUT_OC_FAULT_RESPONSE, the part may be
configured to respond to an overcurrent fault in one
of two ways
1) Pulse by pulse current limiting for a programmed
number of switching cycles (8 to 64 cycles, in 8 cycle
resolution) followed by a latched shutdown.
2) Pulse by pulse current limiting for a programmed
number (8 to 64 cycles, in 8 cycle resolution) of
switching cycles followed by hiccup.
The pulse-by-pulse or constant current limiting
mechanism is briefly explained below.
Figure 22: Pulse by pulse current limiting for 8
cycles, followed by hiccup.
In Figure 22 above, with the overcurrent response
set to pulse-by-pulse current limiting for 8 cycles
followed by hiccup, the converter is operating at
D<0.125 when the overcurrent condition occurs. In
such a case, no duty cycle limiting is applied.
Figure 23: Constant current limiting.
Figure 23 depicts a case where the
overcurrent condition happens when the converter is
operating at D>0.5 and the overcurrent response
has been set to Constant current operation through
pulse by pulse current limiting. In such a case, after
3 consecutive overcurrent cycles are recognized, the
pulse width is dropped such that D=0.5 and then
after 3 more consecutive OCP cycles, to 0.25 and
then finally to 0.125 at which it keeps running until
the total OCP count reaches the programmed
maximum following which the part enters hiccup
mode. Conversely, when the overcurrent condition
disappears, the pulse width is restored to its nominal
value gradually, by a similar mechanism in reverse;
every sequence of 4 consecutive cycles in which the
current is below the overcurrent threshold doubles
the duty cycle, so that D goes from 0.125 to 0.25,
then to 0.5 and finally to its nominal value.
DIE TEMPERATURE SENSING, TELEMETRY
AND THERMAL SHUTDOWN
IR38063 uses on die temperature sensing for
accurate temperature reporting and over
temperature detection. The READ_TEMEPRATURE
PMBus command reports this temperature in 10C
resolution. The trip threshold is set by default to
145oC. The default over temperature response of the
IR38063 (also the response in analog mode) is to
inhibit power conversion while the fault is present,
followed by automatic restart after the fault condition
is cleared. Hence, in the default configuration, when
trip threshold is exceeded, the internal Enable signal
to the power conversion circuitry is de-asserted,
turning off both MOSFETs.
0
IL
0
HDrv
Fs
IOUT_OC_FAULT_LIMIT
0
0
CLK
LDrv
20 ms
Internal
Enable
OCP High
1 2 34 5 6 7 8
0
IL
0
HDrv
Fs
IO UT_OC_FAULT_LIMIT
0
0
CLK
LDrv
Internal
Enable
OCP High
1 2 3456 7 8 910 ...
11
IR38063
31
Rev 3.14
Dec 15, 2017
Automatic restart is initiated when the sensed
temperature drops within the operating range. There
is a 25oC hysteresis in the thermal shutdown
threshold.
The default overtemperature threshold as well as
overtemperature response may be re-configured or
overridden using the OT_FAULT_LIMIT and
OT_FAULT_RESPONSE PMBus commands
respectively. The devices support three types of
responses to an over-temperature fault:
1) Ignore
2) Inhibit when over temperature condition exists
and auto-restart when over temperature condition
disappears
3) Latched shutdown.
REMOTE VOLTAGE SENSING
True differential remote sensing in the feedback loop
is critical to high current applications where the
output voltage across the load may differ from the
output voltage measured locally across an output
capacitor at the output inductor, and to applications
that require die voltage sensing.
The RS+ and RS- pins of the IR38063 form the
inputs to a remote sense differential amplifier with
high speed, low input offset and low input bias
current which ensure accurate voltage sensing and
fast transient response in such applications.
The input range for the differential amplifier is limited
to 1.5V below the VCC rail. Therefore, for
applications in which the output voltage is more than
3V, it is recommended to use local sensing, or if
remote sensing is a must, then the output voltage
between the RS+ and RS-pins must be divided
down to less than 3V using a resistive voltage
divider. Practically, since designs for output voltage
greater than 2.555V require the use of a resistive
divider anyway, it is recommended that this divider
be placed at the input of the remote sense amplifier.
Please note, however, that this modifies the open
loop transfer function and requires a change in the
compensation network to optimally stabilize the loop.
FEED-FORWARD
Feed-Forward (F.F.) is an important feature,
because it can keep the converter stable and
preserve its load transient performance when PVin
varies over a wide range. The PWM ramp
amplitude (Vramp) is proportionally changed with
PVin to maintain PVin/Vramp almost constant
throughout PVin variation range (as shown in Figure
24). Thus, the control loop bandwidth and phase
margin can be maintained constant. Feed-forward
function can also minimize impact on output voltage
from fast PVin change. The feedforward is disabled
for PVin<4.7V. Hence, for PVin<4.7V, a re-
calculation of control loop parameters is needed for
re-compensation.
0
0
PVin
PWM Ramp
12V
Ramp Offset
21V
5V 12V
Figure 24: Timing Diagram for Feed-Forward
(F.F.) Function
LIGHT LOAD EFFICIENCY ENHANCEMENT
(AOT)
The IR38063 implements an Adaptive On Time
control or AOT scheme to improve light load
efficiency. It is based on a COT (Constant On Time)
control scheme with some novel advancements that
make the on-time during diode emulation adaptive
and dependent upon the pulse width in constant
frequency operation. This allows the scheme to be
combined with a PWM scheme, while providing
relatively smooth transition between the two modes
of operation. In other words, the switching regulator
can operate in AOT mode at light loads and
automatically switch to PWM at medium and heavy
loads and vice versa. Therefore, the regulator will
benefit from the high efficiency of the AOT mode at
light loads, and from the constant frequency and fast
transient response of the PWM at medium to heavy
loads.
IR38063
32
Rev 3.14
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In order to enable this light load efficiency
enhancement mode in analog operation, the voltage
at the En/FCCM pin needs to be kept above 4V. In
digital mode, a MFR_SPECIFIC PMBus command
(MFR_FCCM) can be used to enable AOT operation
at light load.
Shortly after the reference voltage has finished
ramping up, an internal circuit which is called the
“calibration circuit” starts operation. It samples the
Comp voltage (output of the error amplifier), digitizes
it and stores it in a register. There is a DAC which
converts the value of this register to an analog
voltage which is equal to the sampled Comp voltage.
At this time, the regulator is ready to enter AOT
mode if the load condition is appropriate. If the load
is so low that the inductor current becomes negative
before the next SW pulse, the operation can be
switched to AOT mode. The condition to enter AOT
is the occurrence of 8 consecutive inductor current
zero crossings in eight consecutive switching cycles.
If this happens, operation is switched to AOT mode
as shown in Figure 25. The inductor current is
sensed using the RDS_ON of the Sync-FET and no
direct inductor current measuring is required. In AOT
mode, just like COT operation, pulses with constant
width are generated and diode emulation is utilized.
This means that a pulse is generated and LDrv is
held on until the inductor current becomes zero.
Then both HDrv and LDrv remain off until the
voltage of the sense pin comes down and reaches
the reference voltage. At this moment the next pulse
is generated. The sense pin is connected to the
output voltage by a resistor divider which has the
same ratio as the voltage divider which is connected
to the feedback pin (Fb).
HDrv
0
0
LDrv
0
SW
0
IL
Ton
0
Vout ...
...
... ...
...
...
...
...
1/Fs Reduced Switching
Frequency
8/Fs delay
Diode
Emulation
Figure 25: Timing Diagram for Reduced
Switching Frequency and Diode Emulation in
Light Load Condition (AOT mode)
When the load increases beyond a certain value, the
control is switched back to PWM through either of
the following two mechanisms:
- If due to the increase in load, the output voltage
drops to 95% of the reference voltage.
-If Vsense remains below the reference voltage for
3 consecutive inductor current zero-cross events
It is worth mentioning that in AOT mode, when
Vsense comes down to reference voltage level, a
new pulse in generated only if the inductor current is
already zero. If at this time the inductor current
(sensed on the Sync-FET) is still positive, the new
pulse generation is postponed till the current decays
to zero. The second condition mentioned above
usually happens when the load is gradually
increased.
It should be noted that in tracking mode, AOT
operation is disabled and the IR38063 can only
operate in continuous conduction mode even at light
loads.
In digital mode, if the output voltage and hence the
reference voltage is commanded to a different
voltage, AOT is disabled during the transition. It is
enabled only after reference voltage finishes its
ramp (up or down) and the calibration circuit has
sampled and held the new Comp voltage.
In general, AOT operation is more jittery and noisier
than FCCM operation, where the switching
frequency may vary from cycle to cycle, giving
increased Vout ripple. Therefore, it is recommended
to use FCCM mode of operation as far as possible.
OUTPUT VOLTAGE TRACKING AND
SEQUENCING
IR38063 can accommodate user programmable
tracking and/or sequencing options using Vp,
Track_En¯¯¯¯¯¯¯¯¯ , Enable, and Power Good pins. The
error-amplifier (E/A) has two non-inverting inputs.
Ideally, the input with the lowest voltage is used for
IR38063
33
Rev 3.14
Dec 15, 2017
regulating the output voltage and the other input is
ignored. In practice the voltage of the other input
should be about 200mV greater than the
low-voltage input so that its effects can completely
be ignored. Vp and Track_En¯¯¯¯¯¯¯¯¯ are internally biased
to 5V via a high impedance path. For normal
operation, Vp is left floating and a 100 kOhm resistor
is connected from Track_En¯¯¯¯¯¯¯¯¯ to P1V8. Therefore, in
normal operating condition, after Enable goes high,
DAC2 ramps up the output voltage until Vfb (voltage
of feedback/Fb pin) reaches about 0.5V.
Tracking-mode operation is achieved by connecting
Track_En¯¯¯¯¯¯¯¯¯ to LGND. In tracking mode, Vfb always
follows Vp which means Vout is always proportional
to Vp voltage (typical for DDR/Vtt rail applications).
The effective Vp variation range is 0V~2.555V.
In sequencing mode of operation (simultaneous or
ratiometric), a 100 kOhm resistor is connected from
Track_En¯¯¯¯¯¯¯¯¯ to P1V8 and Vp is kept to ground level
until DAC2 signal reaches the final value. Then Vp is
ramped up and Vfb follows Vp. When Vp>DAC2
(0.5V in analog mode or default configuration) the
error-amplifier switches to DAC2 and the output
voltage is regulated with DAC2. The final Vp voltage
after sequencing startup should between 0.7V ~ 5V.
Boot
Vcc
Fb
Comp
PGnd
SW
Vo1
(master)
PGood
PGood
Rt/SYNC
5.5V <Vin<21V
PVin
Vin
Vsns
SCL/OCSet
RS+
RSo
RS-
Track_En
Vp En/FCCM
RA
RB
SDA/IMON
SALERT/TMON
ADDR
IR38063
master
LGnd
P1V8
Boot
Vcc
Fb
Comp
PGnd
SW
Vo2
(slave)
PGood
PGood
Rt/SYNC
5.5V <Vin<21V
PVin
Vin
Vsns
SCL/OCSet
RS+
RSo
RS-
Vp En/FCCM
Vo1(master)
RE
RF
RD
RC
Track_En
SDA/IMON
SALERT/TMON
ADDR
IR38063
slave
LGnd
P1V8
Figure 26: Application Circuit for Simultaneous
and Ratiometric Sequencing
Tracking and sequencing operations can be
implemented to be simultaneous or ratiometric (refer
to Figure 27 and Figure 28). Figure 26 shows typical
circuit configuration for sequencing operation. With
this power-up configuration, the voltage at the Vp pin
of the slave reaches 0.5V before the Fb pin of the
master. If RE/RF =RC/RD, simultaneous startup is
achieved. That is, the output voltage of the slave
follows that of the master until the voltage at the Vp
pin of the slave reaches 0.5 V. After the voltage at
the Vp pin of the slave exceeds 0.5V, the internal
0.5V reference of the slave dictates its output
voltage. In reality the regulation gradually shifts from
Vp to internal DAC2. The circuit shown in Figure 26
can also be used for simultaneous or ratiometric
tracking operation if the Track_En¯¯¯¯¯¯¯¯¯ pin of the slave
is connected to LGND. Table 4 summarizes the
required conditions to achieve simultaneous /
ratiometric tracking or sequencing operations.
IR38063
34
Rev 3.14
Dec 15, 2017
Vcc
Reference DAC=0.5V
1.2V Soft Start (slave)
Enable (slave)
Vo1 (master)
Vo2 (slave)
(a)
Vo1 (master)
Vo2 (slave)
(b)
Figure 27: Typical waveforms for sequencing
mode of operation: (a) simultaneous, (b)
ratiometric
Vcc
Track_En=0V (slave)
1.2V Soft Start (slave)
Enable (slave)
Vo1 (master)
Vo2 (slave)
(a)
Vo1 (master)
Vo2 (slave)
(b)
Figure 28: Typical waveforms in tracking mode
of operation: (a) simultaneous, (b) ratiometric
Table 4: Required Conditions for Simultaneous /
Ratiometric Tracking and Sequencing (Figure 26)
Operating
Mode
Track_E
nable
(Slave)
Vp
Required
Condition
Normal
(Non-
sequencing,
Non-tracking)
100
kOhm to
P1V8
Floating
―
Simultaneous
Sequencing
100
kOhm to
P1V8
Ramp
up from
0V
RA/RB>RE/
RF=RC/RD
Ratiometric
100
Ramp
RA/RB>RE/
Sequencing
kOhm to
P1V8
up from
0V
RF>RC/RD
Simultaneous
Tracking
0V
Ramp
up from
0V
RE/RF
=RC/RD
Ratiometric
Tracking
0V
Ramp
up from
0V
RE/RF
>RC/RD
TRACK_EN¯¯¯¯¯¯¯¯¯¯¯
This pin is used to choose between tracking or non-
tracking mode of operation. To enable operation in
tracking mode, this pin must be tied to LGnd. For
non-tracking or sequencing mode, a 100 kOhm
resistor is connected from this pin to P1V8.
OUTPUT VOLTAGE SENSING, TELEMETRY
AND FAULTS
In the IR38063, the voltage sense and regulation
circuits are decoupled, enabling ease of testing as
well as redundancy. In order to do this, IR38063
uses the sense voltage at the dedicated Vsns pin for
output voltage reporting (in 1/256 V resolution, using
the READ_VOUT PMBus command) as well as for
power good detection and output overvoltage
protection.
Power good detection and output overvoltage
detection rely on fast analog comparator circuits,
whereas overvoltage warnings as well as
undervoltage faults and warnings rely on comparing
the digitized Vsns to the corresponding thresholds
programmed using PMBus commands
VOUT_OV_WARN_LIMIT,VOUT_UV_FAULT_LIMIT
and VOUT_UV_WARN_LIMIT respectively.
Power Good Output
The Vsns voltage is an input to the window
comparator with default upper and lower thresholds
of 0.45V and 0.42V respectively. PGood signal is
high whenever Vsns voltage is within the PGood
comparator window thresholds. The PGood pin is
open drain and it needs to be externally pulled high.
High state indicates that output is in regulation. It
should be noted, that in digital mode, the Power
Good thresholds may be changed through the
POWER_GOOD_ON and POWER_GOOD_OFF
commands, which set the rising and falling PGood
IR38063
35
Rev 3.14
Dec 15, 2017
thresholds respectively. However, when no resistive
divider is used, such as for output voltages lower
than 2.555V, the Power Good thresholds must be
programmed to within 630 mV of the output voltage,
otherwise, the effective power good threshold
changes from an absolute threshold to one that
tracks the output voltage with a 630 mV offset.
The threshold is set differently in different operating
modes and the result of the comparison sets the
PGood signal. Figure 29, Figure 30 and Figure 31
show the timing diagram of the PGood signal in
different operating modes. The Vsns signal is also
used by OVP comparator to detect an output over
voltage condition. By default, the PGood signal will
assert as soon as the Vsns signal enters the
regulation window. In digital mode, this delay is
programmable from 0 to 10ms with a 1 ms
resolution, using the MFR_TPGDLY command.
0
0
0
Fault DAC
PGD
Vsns
0.5 V
0.45V
0.42V
160us
0
Reference DAC 0.5 V
Figure 29: Non-sequenced, Non-tracking Startup
0.3V
0
0
0
Vp
Vsns
0.4V
PGD
0.9*Vp
1.2*Vp
Figure 30: Vp Tracking ( Track_En¯¯¯¯¯¯¯¯¯ = 0V)
0
0
0
Reference DAC
PGD
Vsns
(1V<Vp<5V)
0.5V
0.605V
0.45V
0
Vp 0.5V
Figure 31: Vp Sequencing (100 kOhm from
Track_En¯¯¯¯¯¯¯¯¯ to P1V8)
Over-Voltage Protection (OVP)
Over-voltage protection in IR38063 is achieved by
comparing sense pin voltage Vsns to a configurable
overvoltage threshold.
For non-tracking operation, in analog mode, or in
digital mode using the default configuration, the OVP
threshold is set to 0.605V; for tracking operation, it is
set at 1.2*Vp.
For non-tracking operation, in digital mode, the OVP
threshold may be reprogrammed to within 655 mV of
the output voltage (for output voltages lower than
2.555mV, without any resistive divider on the Fb
pin), using the VOUT_OV_FAULT_LIMIT PMBus
command. For an OVP threshold programmed to be
more than 655 mV greater than the output voltage,
the effective OV threshold ceases to be an absolute
IR38063
36
Rev 3.14
Dec 15, 2017
value and instead tracks the output voltage with a
655 mV offset.
When Vsns exceeds the over voltage threshold, an
over voltage trip signal asserts after 200ns (typ.)
delay. The default response is that the high side
drive signal HDrv is latched off immediately and
PGood flags are set low. The low side drive signal is
kept on until the Vsns voltage drops below the
threshold. HDrv remains latched off until a reset is
performed by cycling either Vcc or Enable, or in the
digital mode, using the OPERATION command.
IR38063 allows the user to reconfigure this response
by the use of the VOUT_OV_FAULT_RESPONSE
PMBus command. In addition to the default
response described above, this command can be
used to configure the device such that Vout
overvoltage faults are ignored and the converter
remains enabled. (however, they will still be flagged
in the STATUS_REGISTERS and by SAlert¯¯¯¯¯ ). For
further details on the corresponding PMBus
commands related to OVP, please refer to the
UN0060 IR3806x PMBus commandset user note.
Vsns voltage is set by an external resistive voltage
divider connected to the output.
Figure 32: Timing Diagram for OVP in non-
tracking mode
MINIMUM ON TIME CONSIDERATIONS
The minimum ON time is the shortest amount of time
for Ctrl FET to be reliably turned on. This is a very
critical parameter for low duty cycle, high frequency
applications. In the conventional approach, when the
error amplifier output is near the bottom of the ramp
waveform with which it is compared to generate the
PWM output, propagation delays can be high
enough to cause pulse skipping, and hence limit the
minimum pulse width that can be realized. Moreover,
in the conventional approach, the bottom of the
ramp often presents a high gain region to the error
amplifier output, making the modulator more
susceptible to noise and requiring the use of lower
control loop bandwidth to prevent noise, jitter and
pulse skipping.
IR has developed a proprietary scheme to improve
and enhance the minimum pulse width which
minimizes these delays and hence, allows stable
operation with pulse-widths as small as 35ns. At the
same time, this scheme also has greater noise
immunity, thus allowing stable, jitter free operation
down to very low pulse widths even with a high
control loop bandwidth, thus reducing the required
output capacitance.
Any design or application using IR38063 must
ensure operation with a pulse width that is higher
than the minimum on-time and at least 50 ns of on-
time is recommended in the application. This is
necessary for the circuit to operate without jitter and
pulse-skipping, which can cause high inductor
current ripple and high output voltage ripple.
s
out
s
on FPVin
V
F
D
t
(2)
In any application that uses IR38063, the following
condition must be satisfied:
onon tt
(min)
(3)
sin
out
on FPV
V
t
(min)
(4)
(min)on
out
sin tV
FPV
(5)
The minimum output voltage is limited by the
reference voltage and hence Vout(min) = 0.5V.
Therefore, for Vout(min) = 0.5V,
HDrv
0
0
0
LDrv
Vout DAC1+OV_OFFSET_DAC
Comp
0
0
PGood
DAC1
200 ns
hysteresis
200 ns
IR38063
37
Rev 3.14
Dec 15, 2017
(min)on
out
sin tV
FPV
(6)
s
ns
V
FPV sin V/μ 10
50
5.0
Therefore, at the maximum recommended input
voltage 21V and minimum output voltage, the
converter should be designed at a switching
frequency that does not exceed 476 kHz.
Conversely, for operation at the maximum
recommended operating frequency (1.5 MHz) and
minimum output voltage (0.5V), the input voltage
(PVin) should not exceed 6.7V, otherwise pulse
skipping may happen.
VOLTAGE REFERNCE
The default reference voltage of the error amplifier is
0.5V for both digital and analog mode. The default
voltage scale loop setting is 1.
MAXIMUM DUTY RATIO
A certain off-time is specified for IR38063. This
provides an upper limit on the operating duty ratio at
any given switching frequency. The off-time remains
at a relatively fixed ratio to switching period in the
low and mid frequency range, while at higher
frequencies, the maximum duty ratio at which
IR38063 can operate shows a corresponding
decrease. Figure 33 shows a plot of the maximum
duty ratio vs. the switching frequency with built in
input voltage feed forward mechanism.
Figure 33: Maximum duty cycle vs. switching
frequency
IR38063
38
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DESIGN EXAMPLE
The following example is a typical application for the
IR38063.
PVin = Vin12V
Fs = 607kHz
Vo = 1.2V
Io = 25A
Ripple Voltage = ± 1% * Vo
ΔVo = ± 5% * Vo (for 30% load transient)
Digital mode operation
Enabling the IR38063
As explained earlier, in analog mode, the precise
threshold of the Enable lends itself well to
implementation of a UVLO for the Bus Voltage as
shown in Figure 34.
R1
R2
Enable
IR38063
Vin
Figure 34: Using Enable pin for UVLO
implementation
For a typical Enable threshold of VEN = 1.2 V
2
(min) 12
1.2
in EN
R
PV V
RR
(7)
21 (min)
EN
in EN
V
RR
PV V
(8)
For PVin (min)=9.2V, R1=49.9K and R2=7.5K ohm is a
good choice.
Alternatively, if used in digital mode, the PVin UVLO
thresholds may be programmed to suitable values
such as 9V and 8V, through the VIN_ON and
VIN_OFF PMBus commands or through the
appropriate configuration registers respectively.
Programming the frequency
The device is programmed with a default switching
frequency=607kHz. This value may be read using the
FREQUENCY_SWITCH PMBus command.
If operating in analog mode, the timing resistor Rt
should be chosen to be 3.83K
Output Voltage Programming
The IR38063 offers flexibility for programming the
output voltage. Two distinct methods of programming
the Output voltage are available and the appropriate
one should be chosen depending upon if the mode of
operation is analog or digital.
In the analog mode of operation, the output voltage is
programmed by the reference voltage and an external
resistive divider. The FB pin is the inverting input of
the error amplifier, which is internally referenced to
VREF.
The divider ratio is set such that the voltage at the
VREF pin equals that at the FB pin when the output is
at its desired value. When an external resistor divider
is connected to the output as shown in Figure 35, the
output voltage is defined by using the following
equation:
6
5
1R
R
VV refo
(9)
refo
ref
VV
V
RR 56
(10)
R5
R6
FB
IR38063
Vo
Figure 35: Typical application of the IR38063 for
programming the output voltage
However, in the digital mode of operation, the Vout
related PMBus commands and the Vout related
registers allow the user to program the output voltage
directly, by changing the reference voltage (up to a
maximum of 2.555V) in response to the commanded
IR38063
39
Rev 3.14
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voltage. Therefore, no resistive divider is necessary
for this design since Vo=1.2V.
Bootstrap Capacitor Selection
To drive the Control FET, it is necessary to supply a
gate voltage at least 4V greater than the voltage at the
SW pin, which is connected to the source of the
Control FET. This is achieved by using a bootstrap
configuration, which comprises the internal bootstrap
diode and an external bootstrap capacitor (C1). The
operation of the circuit is as follows: When the sync
FET is turned on, the capacitor node connected to SW
is pulled down to ground. The capacitor charges
towards Vcc through the internal bootstrap diode
(Figure 36), which has a forward voltage drop VD. The
voltage Vc across the bootstrap capacitor C1 is
approximately given as:
DcccVVV
(11)
When the control FET turns on in the next cycle, the
capacitor node connected to SW rises to the bus
voltage PVin. However, if the value of C1 is
appropriately chosen, the voltage Vc across C1
remains approximately unchanged and the voltage at
the Boot pin becomes:
Boot in cc D
V PV V V
(12)
L
Vc
C1
PVIN
Vcc
SW
+
-
Boot
PGnd
+ VD -
IR38063
Cvin
Figure 36: Bootstrap circuit to generate Vc voltage
A bootstrap capacitor of value 0.1uF is suitable for
most applications.
Input Capacitor Selection
The ripple currents generated during the on time of
the control FETs should be provided by the input
capacitor. The RMS value of this ripple for each
channel is expressed by:
DDII oRMS 1
(13)
o
in
V
DPV
(14)
Where:
D is the Duty Cycle
IRMS is the RMS value of the input capacitor
current.
Io is the output current.
Io=25A and D = 0.1, the IRMS = 7.5A.
Ceramic capacitors are recommended due to their
peak current capabilities. They also feature low ESR
and ESL at higher frequency which enables better
efficiency. For this application, it is advisable to have
4x22uF, 25V ceramic capacitors,
C3216X5R1E226M160AB from TDK. In addition to
these, although not mandatory, a 1x330uF, 25V SMD
capacitor EEV-FK1E331P from Panasonic may also
be used as a bulk capacitor and is recommended if
the input power supply is not located close to the
converter.
Inductor Selection
Inductors are selected based on output power,
operating frequency and efficiency requirements. A
low inductor value causes large ripple current,
resulting in the smaller size, faster response to a load
transient but poor efficiency and high output noise.
Generally, the selection of the inductor value can be
reduced to the desired maximum ripple current in the
inductor (Δi). The optimum point is usually found
between 20% and 50% ripple of the output current.
For the buck converter, the inductor value for the
desired operating ripple current can be determined
using the following relation:
1
;
in o s
i
PV V L t D
tF
o
in o in s
V
L PV V PV i F
(15)
Where:
PVin = Maximum input voltage
IR38063
40
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V0 = Output Voltage
Δi = Inductor Ripple Current
Fs = Switching Frequency
Δt = On time for Control FET
D = Duty Cycle
If Δi ≈ 34%*Io, then the inductor is calculated to be
0.212μH. Select L=0.215μH, PCDC1008-R215EMO,
from Cyntec which provides a compact, low profile
inductor suitable for this application. The selected
inductor value give a peak-to-peak inductor ripple
current=8.3A.
Output Capacitor Selection
The voltage ripple and transient requirements
determine the output capacitors type and values. The
criterion is normally based on the value of the
Effective Series Resistance (ESR). However the
actual capacitance value and the Equivalent Series
Inductance (ESL) are other contributing components.
These components can be described as:
)(CoESLoESRoo VVVV
ESRIV LESR )(0
0( ) in o
ESL PV V
V ESL
L
so
L
CFCI
V
8
)(0
(16)
Where:
ΔV0 = Output Voltage Ripple
ΔIL = Inductor Ripple Current
Since the output capacitor has a major role in the
overall performance of the converter and determines
the result of transient response, selection of the
capacitor is critical. The IR38063 can perform well
with all types of capacitors.
As a rule, the capacitor must have low enough ESR to
meet output ripple and load transient requirements.
The goal for this design is to meet the voltage ripple
requirement in the smallest possible capacitor size.
Therefore it is advisable to select ceramic capacitors
due to their low ESR and ESL and small size. Seven
of TDK C2012X5R0J476M (47uF/0805/X5R/6.3V)
capacitors is a good choice.
It is also recommended to use a 0.1µF ceramic
capacitor at the output for high frequency filtering.
Feedback Compensation
The IR38063, while allowing flexibility and
configurability through the digital wrapper of the
PMBus interface, still employs a high performance
voltage mode control engine. The control loop
is a single voltage feedback path including error
amplifier and a PWM comparator. To achieve fast
transient response and accurate output regulation, a
compensation circuit is necessary. The goal of the
compensation network is to provide a closed-loop
transfer function with the highest
0 dB crossing frequency and adequate phase margin
(greater than 45o).
The output LC filter introduces a double pole, -
40dB/decade gain slope above its corner resonant
frequency, and a total phase lag of 180o. The resonant
frequency of the LC filter is expressed as follows:
oo
LC CL
F
2
1
(17)
Figure 37 shows gain and phase of the LC filter. Since
we already have 180o phase shift from the output filter
alone,
the system runs the risk of being unstable.
Phase
00
FLC
0Frequency
FLC Frequency
00
-180
0
0dB
-40dB/Decade
-90
Gain
Figure 37: Gain and Phase of LC filter
IR38063
41
Rev 3.14
Dec 15, 2017
The IR38063 uses a voltage-type error amplifier with
high-gain (90dB) and high-bandwidth (30MHz). The
output of the amplifier is available for DC gain control
and AC phase compensation.
The error amplifier can be compensated either in type
II or type III compensation.
Local feedback with Type II compensation is shown in
Figure 38.
This method requires that the output capacitor have
enough ESR to satisfy stability requirements. If the
output capacitor’s ESR generates a zero at 5kHz to
50kHz, the zero generates acceptable phase margin
and the Type II compensator can be used.
The ESR zero of the output capacitor is expressed as
follows:
o
ESR CESR
F
21
(18)
VOUT
VREF
R6
R5
CPOLE
C3
R3
Ve
FZFPOLE
E/A
Zf
Frequency
Gain(dB)
H(s) dB
Fb Comp
ZIN
Figure 38: Type II compensation network and its
asymptotic gain plot
The transfer function (Ve/Vout) is given by:
35
33
1
)( CsR CsR
Z
Z
sH
V
V
IN
f
out
e
(19)
The (s) indicates that the transfer function varies as a
function of frequency. This configuration introduces a
gain and zero, expressed by:
5
3
)( R
R
sH
(20)
33
21CR
Fz
(21)
First select the desired zero-crossover frequency (Fo):
ESRo FF
and
so FF )10/1~5/1(
(22)
Use the following equation to calculate R3:
5
32
osc o ESR
in LC
V F F R
RPV F
(23)
Where:
PVin = Maximum Input Voltage
Vosc =Effective amplitude of the oscillator ramp
Fo = Crossover Frequency
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
R5 = Feedback Resistor
To cancel one of the LC filter poles, place the zero
before the LC filter resonant frequency pole:
LCZFF %75
oo
ZCL
F
2
1
75.0
(24)
Use equation (22), (23) and (24) to calculate C3.
One more capacitor is sometimes added in parallel
with C3 and R3. This introduces one more pole which
is mainly used to suppress the switching noise.
The additional pole is given by:
POLE
POLE
p
CC CC
R
F
3
3
3
2
1
(25)
The pole sets to one half of the switching frequency
which results in the capacitor CPOLE:
IR38063
42
Rev 3.14
Dec 15, 2017
S
S
POLE FR
C
FR
C
3
3
3
1
1
1
(26)
For a general unconditional stable solution for any
type of output capacitors with a wide range of ESR
values, we use a local feedback with a type III
compensation network. The typically used
compensation network for voltage-mode controller is
shown in Figure 39.
VOUT
VREF
R6
R5
R4
C4
C2
C3
R3
Ve
FZ1FZ2FP2FP3
E/A
Zf
ZIN
Frequency
Gain (dB)
|H(s)| dB
Fb Comp
Figure 39: Type III Compensation network and its
asymptotic gain plot
Again, the transfer function is given by:
IN
f
out
eZ
Z
sH
V
V )(
By replacing Zin and Zf, according to Figure 39, the
transfer function can be expressed as:
44
32
32
3325
54433
11
11
)(
CsR
CC CC
sRCCsR
RRsCCsR
sH
(27)
The compensation network has three poles and two
zeros and they are expressed as follows:
0
1
P
F
(28)
44
221CR
FP
(29)
23
32
32
3
321
2
1CR
CC CC
R
FP
(30)
33
121CR
FZ
(31)
54534
221
21RCRRC
FZ
(32)
Cross over frequency is expressed as:
34 1
2
in
oosc o o
PV
F R C V L C
(33)
Based on the frequency of the zero generated by the
output capacitor and its ESR, relative to the crossover
frequency, the compensation type can be different.
Table 5 shows the compensation types for relative
locations of the crossover frequency.
Table 5: Different types of compensators
Compensator
Type
FESR vs FO
Typical Output
Capacitor
Type II
FLC < FESR < FO <
FS/2
Electrolytic
Type III
FLC < FO < FESR
SP Cap,
Ceramic
The higher the crossover frequency is, the potentially
faster the load transient response will be. However,
the crossover frequency should be low enough to
allow attenuation of switching noise. Typically, the
control loop bandwidth or crossover frequency (Fo) is
selected such that:
so F F * 1/10~1/5
The DC gain should be large enough to provide high
DC-regulation accuracy. The phase margin should be
greater than 45o for overall stability.
In this design, we target Fo= 75 kHz.
IR38063
43
Rev 3.14
Dec 15, 2017
The specifications
PVin = 12V
Vo = 1.2V
Vosc = 1.357 (This is a function of PVin, duty cycle
and switching frequency. Infineon’s
SupIRBuck online design tool can help the
user in accounting for this operating point
dependency of the effective oscillator ramp
amplitude)
Vref = 1.2V
Lo = 0.215 µH
Co = 7 x 47µF, ESR≈3mΩ each
It must be noted here that the value of the
capacitance used in the compensator design must be
the small signal value. For instance, the small signal
capacitance of the 47µF capacitor used in this design
is 34µF at 1.2 V DC bias and 607 kHz frequency. It is
this value that must be used for all computations
related to the compensation. The small signal value
may be obtained from the manufacturer’s datasheets,
design tools or SPICE models. Alternatively, they may
also be inferred from measuring the power stage
transfer function of the converter and measuring the
double pole frequency FLC and using equation
(22) to compute the small signal Co.
These result to:
FLC = 22.2 kHz
FESR = 1220 kHz
Fs/2 = 300 kHz
Select crossover frequency F0=75 kHz
Since FLC<F0<Fs/2<FESR, Type III is selected to place
the pole and zeros.
Detailed calculation of compensation Type III:
Desired Phase Margin Θ = 70°
sin1 sin1
2oZ FF
13.22 kHz
sin1 sin1
2oP FF
425.35 kHz
Select:
21 5.0 ZZ FF
6.61 kHz and
sP FF 5.0
3
300 kHz
Select C4 = 2.2nF.
Calculate R3, C3 and C2:
34
2o o o osc
in
F L C V
RC PV
; R3 = 1.19 kΩ,
Select: R3 = 1.21 kΩ
31
321RF
C
Z
; C3 = 20.03 nF, Select: C3 = 22
nF
33
221RF
C
P
; C2 = 450 pF, Select: C2 = 390
pF
Calculate R4, R5 and R6:
24
421
P
FC
R
; R4 = 170 Ω, Select R4 = 182
Ω
24
521
Z
FC
R
; R5 = 5.29kΩ, Select R5 = 5.62
kΩ
In digital mode, R6 is not necessary.
Setting the Power Good Threshold
In digital mode, the PMBus commands
POWER_GOOD_ON and POWER_GOOD_OFF, or
the corresponding registers may be used to adjust the
power good thresholds to within 630 mV of the output
voltage (for output voltages <2.555V). In this design,
the power good thresholds have been set such that
the Power Good is asserted when the output voltage
rises above 1.074V, and is de-asserted when the
output voltage falls below 1V, giving 74mV of
hysteresis.
IR38063
44
Rev 3.14
Dec 15, 2017
In this design, a power good assertion delay of 0 ms
was programmed. Therefore, the PGood signal
asserts as soon as the output voltage rises above the
power good assertion threshold, and remains
asserted until the output voltage drops below the
power good de-assertion threshold. There is a fixed
160us delay for power good de-assertion. It should be
noted, however, that an overvoltage condition or any
fault condition that causes a shutdown will lead to
PGood de-assertion without any delay.
Selecting Power Good Pull-Up Resistor
The PGood is an open drain output and require pull
up resistors to VCC. The value of the pull-up resistors
should limit the current flowing into the PGood pin to
less than 5mA. A typical value used is 4.99kΩ.
Setting the Overvoltage Threshold
In digital mode, the overvoltage protection threshold
may be programmed using the PMBus command
VOUT_OV_FAULT_LIMIT, or the corresponding
configuration registers, to within 655 mV of the output
voltage (for output voltages <2.555V). In this design,
the threshold has been set to 1.5V. The fault response
has been set to shutdown, so that an overvoltage
condition will cause the part to shutdown with the sync
FET remaining on until the voltage drops 5% below
the overvoltage threshold. In analog
Setting the Overcurrent Threshold
For this 25A design, the overcurrent protection
threshold has been programmed such that the part
goes into a hiccup current limiting mode when the
inductor valley current exceeds 33A, or when the load
current exceeds ~37A.
Communicating on the I2C/PMBus
In order to enable digital mode, as explained earlier, a
resistor needs to be connected from the ADDR pin to
LGnd. In this design, RADDR was chosen to be 499
ohm, to have no offset from the base i2c/PMBus
address.
Further, Infineon’s PowIRCenter USB-to-I2C dongles
have their SCL and SDA lines internally pulled up to
3.3V. Therefore, although this design provides
placeholders for the bus pullups, they may be left
unpopulated if the PowIRCenter dongle is used. The
SAlert¯¯¯¯¯ line is pulled up to Vcc with a 4.99K resistor.
IR38063
45
Rev 3.14
Dec 15, 2017
Boot
Vcc/
LDO_out
Fb
Comp
SW Vo
PGood
PGood
PVin
Vin
Vsns
RS+
RSo
RS-
ADDR
SCL/OCSet
SDA/IMON
SAlert/TMON
PGnd LGnd
Vp En/
FCCM
R1
49.9 KCin=1X330uF/25V
+ 4X22uF/1206/
X5R/16V
R2
7.5 K
Vin=12V
R4
182 C4
2.2nF
R5
5.62 K
R3
1.21 K
C3
22 nF
C2
390 pF
Co=7X47uF/0805/
X5R/6.3V
Lo
0.215uH
RPG
4.99 K
RADDR
499
CVCC=1X10uF/
0805/X5R/10V
Cboot=0.1uF/0603/
X7R/50V
CADDR
(optional)
10 nF
Rt/SYNC
P1V8
Track_EN
RTRACK
100 K
CP1V8=1X2.2uF/
0603/X5R/10V
Figure 40: Application circuit for a single supply, 12V to 1.2V, 25A Point of Load Converter
IR38063
46
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING WAVEFORMS
Vin = PVin = 12V, Vout = 1.2V, Iout = 0-25A, Room Temperature, No Air Flow
Figure 44: Operation 00, Immediate OFF, 25A
load
Ch1:PVin, Ch2:Vout, Ch3:PGood, Ch4:Enable
Figure 45: Inductor node at 25A load
Ch3:SW node
Figure 42:PVin Start up at 25A Load
Ch1:PVin, Ch2:Vout, Ch3:PGood,Ch4:Vcc
Figure 43:Operation 80,Turn ON without margining, 25A
load
Ch1:PVin, Ch2:Vout, Ch3:PGood, Ch4:Enable
Figure 46: Output voltage ripple at 25A load
Ch2:Vout
Figure 41: PVin Start up at 25A Load
Ch1:PVin, Ch2:Vout, Ch3:PGood, Ch4:Enable
IR38063
47
Rev 3.14
Dec 15, 2017
Figure 48: Short-circuit recovery (Hiccup) at 25A
load
Ch2:Vout, Ch3:PGood
Figure 47: 0.35V Prebias voltage startup at 0A
load
Ch2:Vout, Ch3:PGood
TYPICAL OPERATING WAVEFORMS
Vin = PVin = 12V, Vout = 1.2V, Iout = 0-25A, Room Temperature, No Air Flow
IR38063
48
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING WAVEFORMS
Vin = PVin = 12V, Vout = 1.2V, Iout = 0-25A, Room Temperature, No Air Flow
Figure 49: Transient Response, 2.5A to 10A step (2.5A/us)
Ch2:Vout, Ch4:Iout
Figure 50: Transient Response, 17.5A to 25A step (2.5A/us)
Ch2:Vout, Ch4:Iout
IR38063
49
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING WAVEFORMS
Vin = PVin = 12V, Vout = 1.2V, Iout = 0-25A, Room Temperature, No Air Flow
Figure 51: Bode Plot at 0A load
Bandwidth = 83.3kHz, Phase Margin = 56.3o , Gain Margin = 10.8dB
Figure 52: Bode Plot at 25A load
Bandwidth = 78.7kHz, Phase Margin = 50.8o, Gain Margin = 11.3dB
12
-60
-40
-20
0
20
40
60
-200
-150
-100
-50
0
50
100
150
200
103104105
TR1/dB
TR2/°
f/Hz
TR1: Mag(Gain) TR2: Unwrapped Phase(Gain)
12
-60
-40
-20
0
20
40
60
-200
-150
-100
-50
0
50
100
150
200
103104105
TR1/dB
TR2/°
f/Hz
TR1: Mag(Gain) TR2: Unwrapped Phase(Gain)
IR38063
50
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING WAVEFORMS
Vin = PVin = 12V, Vout = 1.2V, Iout = 0-25A, Room Temperature, No Air Flow
Figure 54: Power loss versus load current
Figure 53: Efficiency versus load current
IR38063
51
Rev 3.14
Dec 15, 2017
TYPICAL OPERATING WAVEFORMS
Vin = PVin = 12V, Vout = 1.2V, Iout = 0-25A, Room Temperature, No Air Flow
Figure 55: Thermal Image of the board at 25A load
IR38063: 72.9oC, inductor: 58.4oC, Ambient:26.5oC
IR38063
52
Rev 3.14
Dec 15, 2017
Figure 56: PMBus Command Summary for the 1.2V design
IR38063
53
Rev 3.14
Dec 15, 2017
I2C PROTOCOLS
All registers may be accessed using either I2C or PMBus protocols. I2C allows the use of a simple format
whereas PMBus provides error checking capability. Figure 57 shows the I2C format employed by Manhattan
A A
S Slave
Address
W Register
Address
Data Byte
1 7 1 8
S
1
7
W
1
A P
8 1 1
1
1
8
WRITE
READ
P
1 1
…
S
1
7
R
1 8
N P
1 1
S: Start Condition
A: Acknowledge (0')
N: Not Acknowledge (1')
Sr: Repeated Start Condition
P: Stop Condition
R: Read (1')
W: Write (0')
PEC: Packet Error Checking
*: Present if PEC is enabled
: Master to Slave
: Slave to Master
AA Register
Address
A
Slave
Address
Slave
Address Data Byte
Figure 57: I2C Format
SMBUS/PMBUS PROTOCOLS
To access IR’s configuration and monitoring registers, 4 different protocols are required:
the SMBus Read/Write Byte/Word protocol with/without PEC (for status and monitoring)
the SMBus Send Byte protocol with/without PEC (for CLEAR_FAULTS only)
the SMBus Block Read protocol for accessing Model and Revision information
the SMBus Process call (for accessing Configuration Registers)
In addition, Manhattan supports:
Alert Response Address (ARA)
Bus timeout (10ms)
Group Command for writing to many VRs within one command
SSlave
Address W Command
Code Data Byte
1 7 1 8
S
1
Slave
Address
7
W
1
A
A
A A P
8 1 1
1
1
Command
Code
Data Byte
High
8
A
8
1
Data Byte
Low A
8 1
PEC*
8
1
A*
A P
1 1
PEC*
8
1
A*
BYTE
WORD
…
S: Start Condition
A: Acknowledge (0')
N: Not Acknowledge (1')
Sr: Repeated Start Condition
P: Stop Condition
R: Read (1')
W: Write (0')
PEC: Packet Error Checking
*: Present if PEC is enabled
: Master to Slave
: Slave to Master
Figure 58: SMBus Write Byte/Word
IR38063
54
Rev 3.14
Dec 15, 2017
P
Sr
7
R
8
Data Byte
S Slave
Address
W Command
Code
1 7 1 8
A N
A A
1 1 1 1 1 1 1
P
N
Data Byte
Low
1 1
Data Byte
High
8
Sr
7
R
S
W
1 7 1 8
A A A
1 1 1 1 1 8
PEC*
8
A*
1
1
…
PEC*
8
A*
1
BYTE
WORD
Slave
Address
Slave
Address
Command
Code
Slave
Address A
Figure 59: SMBus Read Byte/Word
P
PEC*
8
SSlave
Address
W Command
Code
1 7 1 8
A A*A
1 1 1 1
Figure 60: SMBus Send Byte
A
1
S Slave
Address
WCommand
Code
1 7 1 8
A A
1 1
Byte Count =1
Data Byte
8
Sr
R
1 7 1
A
1 8
…
P N
1 1
A*
1
PEC*
8
Slave
Address
Figure 61: SMBus Block Read with Byte Count=1
Figure 62: MFR specific command to Write an internal Register
SPMBus
Address WCommand
D1h Register
Address Data Byte
A A A A P
PEC* A
IR38063
55
Rev 3.14
Dec 15, 2017
Figure 63: SMBus Custom Process Call to Read an internal Register
A
1
A
High
Data Byte
A
AA
W
ALow
Data Byte AA
W
A A A A
8 8
S W
1 7 1 8
1 1 1 1
…
…
8
PEC1*
1
High
Data Byte
1 or more bytes
8 8
Sr Command
Code 2
1 7 1 8
1 1 1 1
A …
…
8
PEC2*
1
A*
1 or more bytes
8 8
Sr
1 7 1 8
1 1 1 1
…
8
PECn*
1 or more bytes
P
1
A*
Low
Data Byte
Slave
Address 1
Slave
Address 2
Command
Code 1
High
Data Byte
Slave
Address n
Command
Code n
Low
Data Byte
Figure 64: Group Command
IR38063
56
Rev 3.14
Dec 15, 2017
LAYOUT RECOMMENDATIONS
The layout is very important when designing high
frequency switching converters. Layout will affect
noise pickup and can cause a good design to
perform with less than expected results.
Make the connections for the power components in
the top layer with wide, copper filled areas or
polygons. In general, it is desirable to make proper
use of power planes and polygons for power
distribution and heat dissipation.
The input capacitors, inductor, output capacitors and
the IR38063 should be as close to each other as
possible. This helps to reduce the EMI radiated by
the power traces due to the high switching currents
through them. Place the input capacitor directly at
the PVin pin of IR38063.
The feedback part of the system should be kept
away from the inductor and other noise sources.
The critical bypass components such as capacitors
for Vin, VCC and 1.8V should be close to their
respective pins. It is important to place the feedback
components including feedback resistors and
compensation components close to Fb and Comp
pins.
In a multilayer PCB use one layer as a power
ground plane and have a control circuit ground
(analog ground), to which all signals are referenced.
The goal is to localize the high current path to a
separate loop that does not interfere with the more
sensitive analog control functions. These two
grounds must be connected together on the PC
board layout at a single point. It is recommended to
place all the compensation parts over the analog
ground plane in top layer.
The Power QFN is a thermally enhanced package.
Based on thermal performance it is recommended to
use at least a 6-layers PCB. To effectively remove
heat from the device the exposed pad should be
connected to the ground plane using vias.
IR38063 has 3 pins, SCL, SDA and SALERT¯¯¯¯¯¯¯ that
are used for I2C/PMBus communication. It is
recommended that the traces used for these
communication lines be at least 10 mils wide with a
spacing between the SCL and SDA traces that is at
least 2-3 times the trace width.
IR38063
57
Rev 3.14
Dec 15, 2017
SUPPORTED PMBUS COMMANDS
Command
Code
Command Name
SMBus
transaction
No. of
bytes
Range
Resolution
Default
Value
Description
01h
OPERATION
R/W Byte
1
Enables or disables the device and controls
margining
02h
ON_OFF_CONFIG
R/W Byte
1
Configures the combination of Enable pin input and
serial bus commands needed to turn the unit on and
off.
03h
CLEAR_FAULTS
Send Byte
0
Clear contents of Fault registers
10h
WRITE_PROTECT
R/W Byte
1
Used to control writing to the PMBus device. The
intent of this command is to provide protection
against accidental changes.
15h
STORE_USER_ALL
Send Byte
0
Burns the User section registers into OTP memory
16h
RESTORE_USER_ALL
Send Byte
0
Copies the OTP registers into User memory
19h
CAPABILITY
Read Byte
1
Returns 1011xxxx to indicate Packet Error Checking
is supported, maximum bus speed is 400kHz and
SMBAlert# is supported.
1Bh
SMBALERT_MASK
Write
word/Block
read
Process call
2
May be used to prevent a warning or fault condition
from asserting the SMBALERT# signal.
21h
VOUT_COMMAND16
R/W Word
2
0-
2.555V/VS
5mV/VS
0.5V
Causes the device to set its output voltage to the
commanded value.
VS= VOUT_SCALE_LOOP
22h
VOUT_TRIM16
R/W Word
2
-128-
+128V
0V
Available to the device user to trim the output voltage
24h
VOUT_MAX16
R/W Word
2
6V
Sets an upper limit on the output voltage the unit can
command regardless of any other commands or
combinations.
25h
VOUT_MARGIN_HIGH16
R/W Word
2
0-
2.555V/VS
5mV/VS
0.55V
Sets the MARGIN high voltage when commanded by
OPERATION
VS= VOUT_SCALE_LOOP
26h
VOUT_MARGIN_LOW16
R/W Word
2
0-
2.555V/VS
5mV/VS
0.45V
Sets the MARGIN low voltage when commanded by
OPERATION
VL= VOUT_SCALE_LOOP
27h
VOUT_TRANSITION_RATE11
R/W Word
2
0-
127ms/us
0.125mV
/us
Sets the rate in mV/μs at which the output should
change voltage. Exponent 0 to -4 allowed.
29h
VOUT_SCALE_LOOP11
R/W Word
2
0.125-1
1
Compensates for external resistor divider in feedback
path and in the sense path. Values 1, 0.5, 0.25,
0.125 allowed. Exponent -3 allowed.
33h
FREQUENCY_SWITCH11
R/W Word
2
166-
1500kHz
607kHz
Sets the switching frequency, in kHz. Exponent 0 to
1 allowed.
35h
VIN_ON11
R/W Word
2
0-16.5V
0.5V
1V
Sets the value of the input voltage, in volts, at which
the unit should start power conversion. Exponent -1
allowed.
36h
VIN_OFF11
R/W Word
2
0-16V
0.5V
0.5V
Sets the value of the input voltage, in volts, at which
the unit, once operation has started, should stop
power conversion. Exponent -1 allowed.
39h
IOUT_CAL_OFFSET11
R/W Word
2
-128A-
+127.5A
0.5A
0A
Used to null out any offsets in the output current
sensing circuit. Exponent -1 allowed.
40h
VOUT_OV_FAULT_LIMIT16
R/W Word
2
(25-
655mV)/VS
10mV/VS
0.605V
Sets the value of the output voltage measured at the
sense pin that causes an output overvoltage fault.
VS= VOUT_SCALE_LOOP
41h
VOUT_OV_FAULT_RESPONS
E
R/W Byte
1
Ignore/Shut
down
Shutdow
n
Instructs the device on what action to take in
response to an output overvoltage fault.
42h
VOUT_OV_WARN_LIMIT16
R/W Word
2
3.9mV
0.56V
Sets the value of the output voltage at the
sense pin that causes an output voltage high
warning.
43h
VOUT_UV_WARN_LIMIT16
R/W Word
2
3.9mV
0.44V
Sets the value of the output voltage at the
Sense pin that causes an output voltage low warning.
IR38063
58
Rev 3.14
Dec 15, 2017
44h
VOUT_UV_FAULT_LIMIT16
R/W Word
2
3.9mV
0.395V
Sets the value of the output voltage at the
sense pin that causes an output undervoltage fault.
45h
VOUT_UV_FAULT_RESPONS
E
R/W Byte
1
Ignore/Shut
down
Ignore
Instructs the device on what action to
take in response to an output undervoltage fault.
46h
IOUT_OC_FAULT_LIMIT11
R/W Word
2
3-37.5A
0.5A
33A
Sets the value of the output current, in
amperes, that causes the overcurrent detector to
indicate an overcurrent fault. Exponent -1 allowed.
47h
IOUT_OC_FAULT_RESPONSE
R/W Byte
1
Pulse by
pulse for
8 cycles,
then
hiccup or
latch off
Instructs the device on what action to
take in response to an output overcurrent fault.
4Ah
IOUT_OC_WARN_LIMIT11
R/W Word
2
0-63.5A
0.5A
28A
Sets the value of the output current, in
amperes, that causes the overcurrent detector to
indicate an overcurrent warning. Exponent -1
allowed.
4Fh
OT_FAULT_LIMIT11
R/W Word
2
0-150°C
1°C
145°C
Set the temperature, in degrees Celsius, of the unit at
which it should indicate an Overtemperature Fault.
Exponent 0 allowed.
50h
OT_FAULT_RESPONSE
R/W Byte
1
Ignore/Shut
down/Inhibi
it
Inhibit
Instructs the device on what action to take in
response to an overtemperature fault.
51h
OT_WARN_LIMIT11
R/W Word
2
0-150°C
1°C
125°C
Set the temperature, in degrees Celsius, of the unit at
which it should indicate an Overtemperature Warning
alarm. Exponent 0 allowed.
55h
VIN_OV_FAULT_LIMIT11
R/W Word
2
6.25V-24V
0.25V
24V
Sets the value of the input voltage that causes an
input overvoltage fault. Exponent -2 allowed.
56h
VIN_OV_FAULT_RESPONSE
R/W Byte
1
Ignore/Shut
down
Shutdow
n
Instructs the device on what action to take
in response to an input overvoltage fault.
58h
VIN_UV_WARN_LIMIT11
R/W Word
2
0-16V
0.5V
0.5V
Sets the value of the input voltage PVin, in volts,
that causes an input overvoltage fault. Exponent -1
allowed.
5Eh
POWER_GOOD_ON16
R/W Word
2
(0-
0.63V)/VS
10mV/VS
0.45V
Sets the output voltage at which an optional
POWER_GOOD signal should be asserted.
VS=VOUT_SCALE_LOOP
5Fh
POWER_GOOD_OFF16
R/W Word
2
(0-
0.63V)/VS
10mV/VS
0.42V
Sets the output voltage at which an optional
POWER_GOOD signal should be negated.
VS=VOUT_SCALE_LOOP
60h
TON_DELAY11
R/W Word
2
0-127ms
1ms
0ms
Sets the time, in milliseconds, from when a start
condition is received (as programmed by the
ON_OFF_CONFIG command) until the output
voltage starts to rise. Exponent 0 allowed.
61h
TON_RISE11
R/W Word
2
1-127ms
1ms
2ms
Sets the time, in milliseconds, from when the output
starts to rise until the voltage has entered the
regulation band. Exponent 0 allowed.
62h
TON_MAX_FAULT_LIMIT11
R/W Word
2
0-127ms
1ms
0 (No
limit)
Sets an upper limit, in milliseconds, on how long the
unit can attempt to power up the output without
reaching the output undervoltage fault limit.
Exponent 0 allowed.
63h
TON_MAX_FAULT_RESPONS
E
R/W Byte
1
Ignore/Shut
down
Ignore
Instructs the device on what action to
take in response to a TON_MAX fault.
64h
TOFF_DELAY (not supported)
R/W Word
2
0-127ms
1ms
0ms
Sets the time, in milliseconds, from a stop condition
is received (as programmed by the
ON_OFF_CONFIG command) until the unit stops
transferring energy to the output. Exponent 0
allowed.
65h
TOFF_FALL (not supported)
R/W Word
2
0-127ms
1ms
1ms
Device will acknowledge this command but ignore it.
78h
STATUS BYTE
Read Byte
1
Returns 1 byte where the bit meanings are:
Bit <7> device busy fault
Bit <6> output off (due to fault or enable)
IR38063
59
Rev 3.14
Dec 15, 2017
Bit <5> Output over-voltage fault
Bit <4> Output over-current fault
Bit <3> Input Under-voltage fault
Bit <2> Temperature fault
Bit <1> Communication/Memory/Logic fault
Bit <0>: None of the above
79h
STATUS WORD
Read Word
2
Returns 2 bytes where the Low byte is the same as
the STATUS_BYTE data. The High byte has bit
meanings are:
Bit <7> Output high or low fault
Bit <6> Output over-current fault
Bit <5> Input under-voltage fault
Bit <4> Reserved; hardcoded to 0
Bit <3> Output power not good
Bit <2:0> Hardcoded to 0
7Ah
STATUS_VOUT
Read Byte
1
Reports types of VOUT related faults.
7Bh
STATUS_IOUT
Read Byte
1
Reports types of IOUT related faults.
7Ch
STATUS_INPUT
Read Byte
1
Reports types of INPUT related faults.
7Dh
STATUS_TEMPERATURE
Read Byte
1
Returns Over Temperature warning and Over
Temperature fault (OTP level). Does not report under
temperature warning/fault. The bit meanings are:
Bit <7> Over Temperature Fault
Bit <6> Over Temperature Warning
Bit <5> Under Temperature Warning
Bit <4> Under Temperature Fault
Bit <3:0> Reserved
7Eh
STATUS_CML
Read Byte
1
Returns 1 byte where the bit meanings are:
Bit <7> Command not Supported
Bit <6> Invalid data
Bit <5> PEC fault
Bit <4> OTP fault
Bit <3:2> Reserved
Bit<1> Other communication fault
Bit<0> Other memory or logic fault; hardcoded to 0
88h
READ_VIN11
Read Word
2
Returns the input voltage in Volts
8Bh
READ_VOUT16
Read Word
2
Returns the output voltage in Volts
8Ch
READ_IOUT11
Read Word
2
Returns the output current in Amperes
8Dh
READ_TEMPERATURE11
Read Word
2
Returns the device temperature in degrees Celsius
96h
READ_POUT11
Read Word
2
Returns the output power in Watts
98h
PMBUS_REVISION
Read Byte
1
Reports PMBus Part I rev 1.1 & PMBus
Part II rev 1.2(draft)
99h
MFR_ID
Block
Read/Write
3
IR
Returns 2 bytes used to read the manufacturer’s ID.
User can overwrite with any value.
9Ah
MFR_MODEL
Block
Read/Write
2
Set 00
If set to 00h, returns a 1 byte code corresponding to
IC_DEVICE_ID.
Alternatively, user can set to any non-zero value
9Bh
MFR_REVISION
Block
Read/Write
2
Set 00
If set to 00h, returns a 1 byte code corresponding to
IC_DEVICE_REV.
Alternatively, user can set to any non-zero value
ADh
IC_DEVICE_ID
Block Read
2
Used to read the type or part number of an IC.
IR38060: 30h
IR38063
60
Rev 3.14
Dec 15, 2017
IR38062: 32h
IR38063: 33h
IR38064:34h
AEh
IC_DEVICE_REV
Block Read
2
Used to read the revision of the IC
D0h
MFR_READ_REG
Custom
2
Manufacturer Specific: Read from configuration
registers
D1h
MFR_WRITE_REG
Custom
2
Manufacturer Specific: Write to configuration & status
registers
D8h
MFR_TPGDLY
R/W Word
2
0-10ms
1ms
0ms
Sets the delay in ms, between the output voltage
entering the regulation window and the assertion of
the PGood signal. Exponent 0 allowed.
D9h
MFR_FCCM
R/W Byte
1
0-1
1 (CCM)
Allows the user to choose between forced continuous
conduction mode and adaptive on-time operation at
light load.
D6h
MFR_I2C_address
R/W Word
1
0-7Fh
10h
Sets and returns the device I2C base address
DBh
MFR_VOUT_PEAK16
Read Word
2
Continuously records and reports the highest value of
Read Vout.
DCh
MFR_IOUT_PEAK11
Read Word
2
Continuously records and reports the highest value of
Read Iout.
DDh
MFR_TEMPERATURE_PEAK11
Read Word
2
Continuously records and reports the highest value of
Read_Temperature
IR38063
61
Rev 3.14
Dec 15, 2017
PCB PADS AND COMPONENT
IR38063
62
Rev 3.14
Dec 15, 2017
PCB COPPER AND SOLDER RESIST (PAD SIZES)
IR38063
63
Rev 3.14
Dec 15, 2017
PCB COPPER AND SOLDER RESIST (PAD SPACING)
IR38063
64
Rev 3.14
Dec 15, 2017
SOLDER PASTE STENCIL (PAD SIZES)
IR38063
65
Rev 3.14
Dec 15, 2017
SOLDER PASTE STENCIL (PAD SPACING)
IR38063
66
Rev 3.14
Dec 15, 2017
MARKING INFORMATION
TAPE AND REEL INFORMATION
Refer to Application Note AN-1132 for more information.
IRXXXX
IRXXXX
IR38063
67
Rev 3.14
Dec 15, 2017
PACKAGE INFORMATION
34 32 31
30
29
28
27
26
25
24
23
22
21
20
19
1817161514131210
9
8
7
6
5
4
3
2
1
11
SIDE VIEW (Back)
33
SIDE VIEW (Front)
TOP VIEW
SIDE VIEW (Left) SIDE VIEW (Right)
PIN
1
AB
C
IR38063
68
Rev 3.14
Dec 15, 2017
DIMENSION TABLE
SYMBOL MINIMUM NOMINAL MAXIMUM
A 0.80 0.90 1.00
A1 0.00 0.02 0.05
A3 0.203 Ref
b1 0.45 0.50 0.55
b2 0.30 0.35 0.40
b3 0.20 0.25 0.30
b4 0.325 0.375 0.425
D 5.00 BSC
E 7.00 BSC
D1 3.450 3.500 3.550
E1 1.725 1.775 1.825
D2 0.725 0.775 0.825
E2 1.292 1.342 1.392
D3 1.823 1.873 1.923
E3 1.932 1.982 2.032
L1 0.35 0.40 0.45
L2 0.822 0.872 0.922
e1 0.500 BSC
e2 0.625 BSC
e3 1.125 BSC
e4 1.250 BSC
e5 0.925 BSC
e6 1.373 BSC
e7 0.825 BSC
e8 0.750 BSC
aaa 0.05
bbb 0.10
ccc 0.10
ddd 0.05
eee 0.08
N34
10 18
19
30
3134
1
9
1
9
1810
19
30
3134
PIN 1
(R0.20)
BOTTOM VIEW
IR38063
69
Rev 3.14
Dec 15, 2017
ENVIRONMENTAL QUALIFICATIONS
Qualification Level
Industrial
Moisture Sensitivity Level
5mm x 7mm PQFN
MSL 2 260C
ESD
Machine Model
(JESD22-A115A)
JEDEC Class B
Human Body Model
(JESD22-A114F)
JEDEC Class 2 (2KV)
Charged Device Model
(JESD22-C101D)
JEDEC Class 3
RoHS Compliant
Yes (with Exemption 7a)
† Qualification standards can be found at International Rectifier web site: http://www.irf.com
IR38063
70
Rev 3.14
Dec 15, 2017
REVISION HISTORY
Rev.
Date
Description
3.0
9/9/2015
Initial DR3 Release
3.1
10/5/2015
Corrected OC test condition
3.2
10/17/2015
Recommended Vcc range, corrected typos
Updated Frequency_Switch default to 607kHz (was 600kHz)
3.3
10/21/2015
Added reference to UN0060 PMBus commandset
Corrected default Ton_Rise to 2ms (was 1ms)
3.4
11/06/2015
Added Reference accuracy over -40C+125C
Updated package drawing to include L-shaped pin
3.5
1/15/2016
Improved assembly drawing quality
Removed unnecessary info from Marking diagram
Added Linear telemetry formats to PMBus command table
Corrected Mfr_ID/Model/Rev and other descriptions in PMBus command
table
Clearly specified Vin/Vcc operating ranges
Added Tape & Reel information
Converted to Infineon format
3.6
1/25/2016
Added AC specification for Boot to SW, explicitly stated that no Rt resistor
needed in digital mode, corrected a typo in Vin operating range to PVin
operating range
3.7
3/4/2016
Corrected IOUT_OC_FAULT_LIMIT, IOUT_OC_FAULT_RESPONSE and
IOUT_OC_WARN_LIMIT range and defaults in PMBus commandset table
3.8
5/26/2016
Changed Vcc recommended limits, changed Iout reporting resolution
display format from 0.0625A to 62.5 mA
3.9
8/24/2016
Changed OC response types; also changed PMBus default. Changed pad,
stencil and solder drawings, added info about decoupling caps, added
placement for 10nF cap on addr resistor in typical apps diagrams,
removed gain and bandwidth specs of RSA and EA. Added note about
preferring to use FCCM because AOT is noisier. Changed ADDR resistor for
0 offset to 499 ohm, min. RT resistor also changed from 0 ohm to 499
ohm. Added a list of recommended Fsw. Added note about SS rate.
Updated Ton_Rise range in PMBus table. Corrected IC_DEVICE_ID in
PMBus table by removing IR8061 and IR38065
3.10
12/2/2016
Updates related to 750 k on track_en pin, updated LDO test condition in
spec table
3.11
1/11/2017
Updates related to 100K from track_en to P1V8
3.12
3/1/2017
Update to Vp bias current limit, note on the 750 K option from track_en#
to LGnd
Added 250pc reel, Broadcom, IBM part numbers into Ordering info
3.13
5/17/2017
Updated Ordering Information
3.14
12/15/2017
Update to PMBus commands. TOFF_DELAY and TOFF_FALL not supported
in Manhattan. Specified how to set OCP settings in analog mode. Added
note, Rt pin can be grounded to LGND through a series resistor of 15kohm
IR38063
71
Rev 3.14
Dec 15, 2017
Rev.
Date
Description
instead of floating. Added recommendation to use 10uF bypass capacitor
at P1V8 pin. Updated the schematic diagrams and OCP timing diagram.
Updated Figure 11, to show VCC goes high correctly with PVIN. Updated
Figure 40 to show 499 ohms on the ADDR pin.
IR38063
72
Rev 3.14
Dec 15, 2017
Published by
Infineon Technologies AG
81726 München, Germany
© Infineon Technologies AG 2015
All Rights Reserved.
IMPORTANT NOTICE
The information given in this document shall in no event be regarded as a guarantee of conditions or
characteristics (“Beschaffenheitsgarantie”). With respect to any examples, hints or any typical values stated
herein and/or any information regarding the application of the product, Infineon Technologies hereby disclaims
any and all warranties and liabilities of any kind, including without limitation warranties of non-infringement of
intellectual property rights of any third party.
In addition, any information given in this document is subject to customer’s compliance with its obligations stated
in this document and any applicable legal requirements, norms and standards concerning customer’s products
and any use of the product of Infineon Technologies in customer’s applications.
The data contained in this document is exclusively intended for technically trained staff. It is the responsibility of
customer’s technical departments to evaluate the suitability of the product for the intended application and the
completeness of the product information given in this document with respect to such application.
For further information on the product, technology, delivery terms and conditions and prices please contact your
nearest Infineon Technologies office (www.infineon.com).
WARNINGS
Due to technical requirements products may contain dangerous substances. For information on the types in
question please contact your nearest Infineon Technologies office.
Except as otherwise explicitly approved by Infineon Technologies in a written document signed by authorized
representatives of Infineon Technologies, Infineon Technologies’ products may not be used in any applications
where a failure of the product or any consequences of the use thereof can reasonably be expected to result in
personal injury.
