LTC3601
1
3601fc
For more information www.linear.com/LTC3601
TYPICAL APPLICATION
FEATURES
APPLICATIONS
DESCRIPTION
1.5A, 15V Monolithic
Synchronous Step-Down
Regulator
n Distributed Power Systems
n Lithium-Ion Battery-Powered Instruments
n Point-of-Load Power Supply
L, LT , LT C , LT M , Burst Mode, Linear Technology and the Linear logo are registered trademarks
and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 5481178, 5847554,
6580258, 6304066, 6476589, 6774611.
Efficiency and Power Loss vs Load Current
n 4V to 15V Operating Input Voltage Range
n 1.5A Output Current
n Up to 96% Efficiency
n Very Low Duty Cycle Operation: 5% at 2.25MHz
n Adjustable Switching Frequency: 800kHz to 4MHz
n External Frequency Synchronization
n Current Mode Operation for Excellent Line and Load
Transient Response
n User Selectable Burst Mode
®
(No Load IQ = 300µA) or
Forced Continuous Operation
n 0.6V Reference Allows Low Output Voltages
n Short-Circuit Protected
n Output Voltage Tracking Capability
n Programmable Soft-Start
n Power Good Status Output
n Available in Small, Thermally Enhanced 16-Pin QFN
(3mm × 3mm) and MSOP Packages
The LT C
®
3601 is a high efficiency, monolithic synchronous
buck regulator using a phase-lockable controlled on-time,
current mode architecture capable of supplying up to 1.5A
of output current. The operating supply voltage range is
from 4V to 15V, making it suitable for a wide range of
power supply applications.
The operating frequency is programmable from 800kHz to
4MHz with an external resistor enabling the use of small
surface mount inductors. For switching noise sensitive
applications, the LTC3601 can be externally synchronized
over the same frequency range. An internal phase-locked
loop aligns the on-time of the top power MOSFET to the
internal or external clock. This unique constant frequency/
controlled on-time architecture is ideal for high step-down
ratio applications that demand high switching frequencies
and fast transient response.
The LTC3601 offers two operational modes: Burst Mode
operation and forced continuous mode to allow the user
to optimize output voltage ripple, noise, and light load
efficiency for a given application. Maximum light load
efficiency is achieved with the selection of Burst Mode
operation while forced continuous mode provides minimum
output ripple and constant frequency operation.
VIN
RUN
PGOOD
TRACK
LTC3601
PGNDSGND
BOOST
INTVCC
ITH
RT
MODE/SYNC
SW
VON
FB
2.2µF
10pF
0.1µF
22µF
VOUT
3.3V
1.5A
180k
40k
3601 TA01a
2.2µH
22µF
VIN
4V TO 15V
LOAD CURRENT (A)
30
EFFICIENCY (%)
POWER LOSS (mW)
90
100
20
10
80
50
70
60
40
0.001 0.1 1 10
3601 TA01b
0
1000
100
10
1
0.01
VIN = 5V
VIN = 12V
LTC3601
2
3601fc
For more information www.linear.com/LTC3601
ABSOLUTE MAXIMUM RATINGS
(Note 1)
16 15 14 13
5678
TOP VIEW
17
UD PACKAGE
16-LEAD (3mm × 3mm) PLASTIC QFN
9
10
11
12
4
3
2
1MODE/SYNC
PGOOD
SW
SW
ITH
FB
RT
SGND
VIN
VIN
RUN
TRACK
NC
BOOST
INTVCC
VON
TJMAX = 125°C, θJA = 45°C/W
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
1
2
3
4
5
6
7
8
SW
SW
PGND
PGND
BOOST
INTVCC
VON
RT
16
15
14
13
12
11
10
9
PGOOD
MODE/SYNC
VIN
VIN
RUN
TRACK
ITH
FB
TOP VIEW
17
MSE PACKAGE
16-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 38°C/W
EXPOSED PAD (PIN 17) IS SGND, MUST BE SOLDERED TO PCB
PIN CONFIGURATION
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3601EUD#PBF LTC3601EUD#TRPBF LFJC 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C
LTC3601IUD#PBF LTC3601IUD#TRPBF LFJC 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C
LTC3601EMSE#PBF LTC3601EMSE#TRPBF 3601 16-Lead Plastic MSOP –40°C to 125°C
LTC3601IMSE#PBF LTC3601IMSE#TRPBF 3601 16-Lead Plastic MSOP –40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
VIN ............................................................. –0.3V to 16V
VIN Transient Voltage ................................................18V
BOOST .....................................................–0.3V to 18.6V
BOOST-SW ................................................ –0.3V to 3.6V
INTVCC ...................................................... –0.3V to 3.6V
ITH, RT.......................................–0.3V to INTVCC + 0.3V
MODE/SYNC, FB ........................–0.3V to INTVCC + 0.3V
TRACK .......................................–0.3V to INTVCC + 0.3V
PGOOD, VON ............................................... –0.3V to 16V
SW, RUN .......................................... –0.3V to VIN + 0.3V
SW Source Current (DC) .............................................2A
Peak SW Source Current..........................................3.5A
Operating Junction Temperature Range
(Notes 2, 3) ........................................... –40°C to 125°C
Storage Temperature Range .................. –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
MSOP ............................................................... 300°C
LTC3601
3
3601fc
For more information www.linear.com/LTC3601
ELECTRICAL CHARACTERISTICS
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
VVIN Input Supply Range l4 15 V
IQInput DC Supply Current
Forced Continuous Operation
Sleep Current
Shutdown
MODE = 0V
MODE = INTVCC, VFB > 0.6V
RUN = 0V
700
300
14
1000
500
25
µA
µA
µA
VFB Feedback Reference Voltage l0.594 0.600 0.606 V
∆VLINEREG Reference Voltage Line Regulation VVIN = 4V to 15V 0.01 %/V
∆VLOADREG Output Voltage Load Regulation ITH = 0.6V to 1.6V 0.1 %
IFB Feedback Pin Input Current VFB = 0.6V ±30 nA
gm(EA) Error Amplifier Transconductance ITH = 1.2V 2.0 mS
tON(MIN) Minimum On-Time VON = 1V, VIN = 4V 20 ns
tOFF(MIN) Minimum Off-Time VIN = 6V 40 60 ns
ILIM Valley Switch Current Limit 1.7 2.2 2.8 A
Negative Valley Switch Current Limit –1.2 A
fOSC Oscillator Frequency VRT = INTVCC
RRT = 160k
RRT = 80k
1.4
1.7
3.4
2
2
4
2.6
2.3
4.6
MHz
MHz
MHz
RDS(ON) Top Switch On-Resistance 130 mΩ
Bottom Switch On-Resistance 100 mΩ
VVIN-OV VIN Overvoltage Lockout Threshold VIN Rising
VIN Falling
l
l
16.8
15.8
17.5
16.5
18
17
V
V
VINTVCC INTVCC Voltage 4V < VIN < 15V 3.15 3.3 3.45 V
∆INTVCC INTVCC Load Regulation (Note 4) IINTVCC = 0mA to 20mA 0.6 %
VUVLO INTVCC Undervoltage Lockout
Threshold
INTVCC Rising, VIN = INTVCC
INTVCC Falling, VIN = INTVCC
2.75
2.45
2.9 V
V
VRUN RUN Threshold RUN Rising
RUN Falling
l
l
1.21
0.97
1.25
1.0
1.29
1.03
V
V
IRUN(LKG) RUN Leakage Current VVIN = 15V 0 ±3 µA
VFB_GB PGOOD Good-to-Bad Threshold FB Rising
FB Falling
8
–8
10
–10
%
%
VFB_BG PGOOD Bad-to-Good Threshold FB Rising
FB Falling
–3
3
–5
5
%
%
tPGOOD Power Good Filter Time 20 40 µs
RPGOOD PGOOD Pull-Down Resistance 10mA Load 15 Ω
ISW(LKG) Switch Leakage Current VRUN = 0V 0.01 1 µA
tSS Internal Soft-Start Time VFB from 10% to 90% Full Scale 400 700 µs
VFB_TRACK TRACK Pin TRACK = 0.3V 0.28 0.3 0.315 V
ITRACK TRACK Pull-Up Current 1.4 µA
VMODE/SYNC MODE Threshold Voltage MODE VIH
MODE VIL
l
l
1.0
0.4
V
V
SYNC Threshold Voltage SYNC VIH l0.95 V
IMODE MODE Input Current MODE = 0V
MODE = INTVCC
–1.5
1.5
µA
µA
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VVIN = 12V, unless otherwise specified.
LTC3601
4
3601fc
For more information www.linear.com/LTC3601
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current
Burst Mode Operation
Efficiency vs Load Current
Forced Continuous Mode
Efficiency vs Frequency
Forced Continuous Mode
Reference Voltage vs
Temperature
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3601 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3601E is guaranteed to meet specifications from
0°C to 85°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3601I is guaranteed over the full –40°C to 125°C operating junction
temperature range. Note that the maximum ambient temperature
consistent with these specifications is determined by specific operating
Efficiency vs Input Voltage
Burst Mode Operation
conditions in conjunction with board layout, the rated package thermal
impedance and other environmental factors.
Note 3: TJ is calculated from the ambient temperature, TA, and power
dissipation, PD, according to the following formula:
TJ = TA + (PD • θJA)
where θJA = 45°C/W for the QFN package and θJA = 38°C/W for the MSOP
package.
Note 4: Maximum allowed current draw when used as a regulated output
is 5mA. This supply is only intended to provide additional DC load current
as needed and not intended to regulate large transient or AC behavior as
these waveforms may impact LTC3601 operation.
Efficiency vs Load Current
TA = 25°C, VIN = 12V, fO = 1MHz, L = 2.2µH unless otherwise noted.
LOAD CURRENT (A)
30
EFFICIENCY (%)
90
100
20
10
80
50
70
60
40
0.001 0.1 1 10
3601 G01
0
0.01
VIN = 4V
VIN = 8V
VIN = 12V
VOUT = 1.8V
LOAD CURRENT (A)
30
EFFICIENCY (%)
90
100
20
10
80
50
70
60
40
0.001 0.1 1 10
3601 G02
0
0.01
VIN = 4V
VIN = 8V
VIN = 12V
VOUT = 1.8V
LOAD CURRENT (A)
30
EFFICIENCY (%)
90
100
20
10
80
50
70
60
40
0.001 0.1 1 10
3601 G03
0
0.01
VOUT = 3.3V
VOUT = 5V
BURST
FORCED
CONTINUOUS
INPUT VOLTAGE (V)
4
EFFICIENCY (%)
75
80
85
10 14
3601 G04
70
65
60 6 8 12
90
95
100
16
ILOAD = 500mA
ILOAD = 100mA
VOUT = 1.8V
FIGURE 7 CIRCUIT
ILOAD = 1.5A
ILOAD = 10mA
FREQUENCY (MHz)
0.5
80
EFFICIENCY (%)
82
84
86
88
90
92
1.0 1.5 2.0 2.5
3601 G05
3.0
L = 2.2µH
VOUT =1.8V
ILOAD = 500mA
L = 1µH
TEMPERATURE (°C)
–50 –25
0.595
V
REF
(V)
0.599
0.605
050 75
3601 G06
0.597
0.603
0.601
25 100 125
LTC3601
5
3601fc
For more information www.linear.com/LTC3601
TYPICAL PERFORMANCE CHARACTERISTICS
Quiescent Current vs
Supply Voltage
Oscillator Frequency vs
Temperature
RDS(ON) vs Temperature Switch Leakage vs Temperature
TA = 25°C, VIN = 12V, fO = 1MHz, L = 2.2µH unless otherwise noted.
TEMPERATURE (°C)
–50 –25
0
R
DS(ON)
(mΩ)
80
200
050 75
3601 G07
40
160
120
25 100 125
TOP SWITCH
RDS(ON)
BOTTOM SWITCH
RDS(ON)
TEMPERATURE (°C)
–50
SWITCH LEAKAGE (nA)
4000
5000
6000
25 75
3601 G08
3000
2000
–25 0 50 100 125
1000
0
TOP SWITCH
BOTTOM SWITCH
VDS = 12V
SUPPLY VOLTAGE (V)
4
220
QUIESCENT CURRENT (µA)
260
300
340
380
6 8 10 12
3601 G09
14 16
TEMPERATURE (°C)
–50
FREQUENCY VARIATION (%)
1.5
25
3601 G10
0
–1.0
–25 0 50
–1.5
–2.0
2.0
1.0
0.5
–0.5
75 100 125
Bottom Switch Current Limit
vs Temperature Load Regulation
TEMPERATURE (°C)
–50
2.0
2.5
3.5
25 75
3601 G11
1.5
1.0
–25 0 50 100 125
0.5
0
3.0
ILIM (A)
ILOAD (mA)
0
1.2
1.0
0.8
0.6
0.4
0.2
0
–0.2 750 1250
3601 G21
250 500 1000 1500
∆V
OUT
/V
OUT
(%)
Burst Mode OPERATION
FORCED CONTINUOUS
Oscillator Internal Set Frequency
vs Temperature
TRACK Pull-Up Current
vs Temperature
TEMPERATURE (°C)
–50
FREQUENCY (MHz)
2.25
2.50
2.75
25 75
3601 G22
2.00
1.75
–25 0 50 100 125
1.50
1.25
RT = INTVCC
TEMPERATURE (°C)
–50
1.4
1.6
2.0
25 75
3601 G23
1.2
1.0
–25 0 50 100 125
0.8
0.6
1.8
ITRACK (µA)
LTC3601
6
3601fc
VOUT
100mV/DIV
AC COUPLED
ILOAD
1A/DIV
10µs/DIV 3601 G18
VIN = 12V
VOUT = 1.8V
ILOAD = 150mA TO 1.5A
IL
1A/DIV
VOUT
100mV/DIV
AC COUPLED
ILOAD
1A/DIV
10µs/DIV 3601 G17
VIN = 12V
VOUT = 1.8V
ILOAD = 150mA TO 1.5A
IL
1A/DIV
For more information www.linear.com/LTC3601
TYPICAL PERFORMANCE CHARACTERISTICS
Start-Up from Shutdown
Burst Mode Operation
Start-Up from Shutdown
Forced Continuous Mode
Load Step
Burst Mode Operation
Load Step
Forced Continuous Mode
Short-Circuit Waveforms
Forced Continuous Mode
TA = 25°C, VIN = 12V, fO = 1MHz, L = 2.2µH unless otherwise noted.
RUN
2V/DIV
PGOOD
5V/DIV
IL
1A/DIV
200µs/DIV 3601 G15
VIN = 12V
VOUT = 1.8V
ILOAD = 20mA
VOUT
2V/DIV
RUN
2V/DIV
IL
1A/DIV
200µs/DIV 3601 G16
VIN = 12V
VOUT = 1.8V
ILOAD = 1.5A
VOUT
1V/DIV
VOUT
1V/DIV
PGOOD
2V/DIV
100µs/DIV 3601 G19
VIN = 12V
VOUT = 1.8V
IL
2A/DIV
Start-Up Into Pre-Biased Output
(1V Pre-Bias) Burst Mode
Operation
VOUT
1V/DIV
RUN
10V/DIV
PGOOD
5V/DIV
1ms/DIV 3601 G20
VIN = 12V
VOUT = 1.8V
ILOAD = 5mA
IL
1A/DIV
Output Voltage vs Time
Burst Mode Operation
SW
5V/DIV
IL
1A/DIV
2µs/DIV 3601 G12
VIN = 12V
VOUT = 1.8V
ILOAD = 100mA
VOUT
20mV/DIV
AC COUPLED
Output Voltage vs Time
Forced Continuous Mode
SW
5V/DIV
IL
1A/DIV
2µs/DIV 3601 G13
VIN = 12V
VOUT = 1.8V
ILOAD = 100mA
VOUT
20mV/DIV
AC COUPLED
Output Tracking
2ms/DIV 3601 G14
VIN = 12V
VOUT = 1.8V
RLOAD = 36Ω
VOUT
VFB
TRACK
LTC3601
7
3601fc
For more information www.linear.com/LTC3601
PIN FUNCTIONS
(QFN/MSE)
MODE/SYNC (Pin 1/Pin 15): Mode Selection and External
Synchronization Input Pin. This pin places the LTC3601
into forced continuous operation when tied to ground.
High efficiency Burst Mode operation is enabled by either
floating this pin or by tying this pin to INTVCC. When driven
with an external clock, an internal phase-locked loop will
synchronize the phase and frequency of the internal oscil-
lator to that of the incoming clock signal. During external
clock synchronization, the LTC3601 will default to forced
continuous operation.
PGOOD (Pin 2/Pin 16): Open-Drain Power Good Output
Pin. PGOOD is pulled to ground when the voltage at the
FB pin is not within 8% (typical) of the internal 0.6V refer-
ence. PGOOD becomes high impedance once the voltage
at the FB pin returns to within ±5% (typical) of the internal
reference.
SW (Pins 3, 4/Pins 1, 2): Switch Node Output Pin. Con-
nect this pin to the SW side of the external inductor. The
normal operation voltage swing of this pin ranges from
ground to PVIN.
BOOST (Pin 6/Pin 5): Boosted Floating Driver Supply
Pin. The (+) terminal of the external bootstrap capacitor
connects to this pin while the (–) terminal connects to
the SW pin. The normal operation voltage swing of this
pin ranges from a diode voltage drop below INTVCC up
to PVIN + INTVCC.
INTVCC (Pin 7/Pin 6): Internal 3.3V Regulator Output Pin.
This pin should be decoupled to PGND with a low ESR
ceramic capacitor of 1µF or more.
VON (Pin 8/Pin 7): On-Time Voltage Input Pin. This pin
sets the voltage trip point for the on-time comparator.
Connect this pin to the regulated output to make the on-
time proportional to the output voltage when VOUT ≤ 6V. If
VOUT > 6V, switching frequency may become higher than
the set frequency. The pin impedance is normally 180kΩ.
SGND (Pin 9/Exposed Pad Pin 17): Signal Ground Pin.
This pin should have a low noise connection to reference
ground. The feedback resistor network, external compen-
sation network and RT resistor should be connected to
this ground. In the MSE package, the exposed pad must
be soldered to the PCB to provide a good thermal contact
to the PCB.
RT (Pin 10/Pin 8): Oscillator Frequency Program Pin.
Connect an external resistor, between 80k to 400k, from
this pin to SGND to program the LTC3601 switching fre-
quency from 800kHz to 4MHz. When RT is tied to INTVCC,
the switching frequency will default to 2MHz.
FB (Pin 11/Pin 9): Output Voltage Feedback Pin. Input to
the error amplifier that compares the feedback voltage to
the internal 0.6V reference voltage. Connect this pin to
the appropriate resistor divider network to program the
desired output voltage.
ITH (Pin 12/Pin 10): Error Amplifier Output and Switching
Regulator Compensation Pin. Connect this pin to appro-
priate external components to compensate the regulator
loop frequency response. Connect this pin to INTVCC to
use the default internal compensation.
TRACK (Pin 13/Pin 11): Output Voltage Tracking and Soft-
Start Input Pin. Forcing a voltage below 0.6V on this pin
overrides the internal reference input to the error amplifier.
The LTC3601 will servo the FB pin to the TRACK voltage
under this condition. Above 0.6V, the tracking function
stops and the internal reference resumes control of the
error amplifier. An internal 1.4µA pull-up current from
INTVCC allows a soft-start function to be implemented
by connecting an external capacitor between this pin and
ground. See Applications Information section for more
details.
RUN (Pin 14/Pin 12): Regulator Enable Pin. Enables chip
operation by applying a voltage above 1.25V. A voltage
below 1V on this pin places the part into shutdown. Do
not float this pin.
VIN (Pins 15, 16/Pins 13, 14): Main Power Supply Input
Pins. These pins should be closely decoupled to PGND
with a low ESR capacitor of 10µF or more.
PGND (Exposed Pad Pin 17/Pins 3, 4): Power Ground Pin.
The (–) terminal of the input bypass capacitor, CIN, and the
(–) terminal of the output capacitor, COUT
, should be tied
to this pin with a low impedance connection. In the QFN
package the exposed pad must be soldered to the PCB to
provide low impedance electrical contact to ground and
good thermal contact to the PCB.
LTC3601
8
3601fc
For more information www.linear.com/LTC3601
FUNCTIONAL BLOCK DIAGRAM
–++
–+–+
VIN
VIN
15k
Q6
RUN
SWITCH
LOGIC
AND
ANTI-
SHOOT
THROUGH
BG
ON
Q1
Q2
0.48V 1.25V
RUN
EA
INTERNAL
SOFT-START
SS
Q4
TRACK
CSS
3605 BD
SGND
R2
R1
RUN
PGND
PGOOD
INTVCC
FB
SW
TG
VIN
CIN
BOOST
SENSE+
SENSE–
–
+
–
+
OV
0.648V
–
+
0.3V
FOLDBACK
FOLDBACK
DISABLED
AT START-UP
UV
0.552V
180k
VON
VON
INTVCC
M2
M1
L1
COUT
INTVCC
ITH
RC
CC1
CVCC
CBOOST
–
+
–
+
1.4µA
ITHB
ICMP IREV
3.3V
REG
ION
CONTROLLER
OSC
PLL-SYNC
OSC
R
0.6V
REF
S Q
ION
MODE/SYNC
RRT
RT
tON = VVON
IION
AV = 1
6V
LTC3601
9
3601fc
For more information www.linear.com/LTC3601
OPERATION
The LTC3601 is a current mode, monolithic, step-down
regulator capable of providing up to 1.5A of output current.
Its unique controlled on-time architecture allows extremely
low step-down ratios while maintaining a constant switch-
ing frequency. Part operation is enabled by raising the
voltage on the RUN pin above 1.25V nominally.
Main Control Loop
In normal operation the internal top power MOSFET is
turned on for a fixed interval determined by an internal
one-shot timer (“ON” signal in the Block Diagram). When
the top power MOSFET turns off, the bottom power MOS-
FET turns on until the current comparator, ICMP
, trips,
thus restarting the one-shot timer and initiating the next
cycle. The inductor current is monitored by sensing the
voltage drop across the SW and PGND nodes of the bot-
tom power MOSFET. The voltage at the ITH pin sets the
ICMP comparator threshold corresponding to the induc-
tor valley current. The error amplifier EA adjusts this ITH
voltage by comparing an internal 0.6V reference to the
feedback signal, VFB, derived from the output voltage. If, for
example, the load current increases, the feedback voltage
will decrease relative to the internal 0.6V reference. The
ITH voltage then rises until the average inductor current
matches that of the load current.
The operating frequency is determined by the value of the
RT resistor, which programs the current for the internal
oscillator. An internal phase-locked loop servos the switch-
ing regulator on-time to track the internal oscillator edge
and force a constant switching frequency. A clock signal
can be applied to the SYNC/MODE pin to synchronize the
switching frequency to an external source. The regulator
defaults to forced continuous operation once the clock
signal is applied.
At low load currents the inductor current can drop to zero
or become negative. If the LTC3601 is configured for
Burst Mode operation, this inductor current condition is
detected by the current reversal comparator, IREV
, which
in turn shuts off the bottom power MOSFET and places
the part into a low quiescent current sleep state resulting
in discontinuous operation and increased efficiency at low
load currents. Both power MOSFETs will remain off with
the part in sleep and the output capacitor supplying the
load current until the ITH voltage rises sufficiently to initiate
another cycle. Discontinuous operation is disabled by tying
the MODE/SYNC pin to ground placing the LTC3601 into
forced continuous mode. During forced continuous mode,
continuous synchronous operation occurs regardless of
the output load current.
“Power Good” Status Output
The PGOOD open-drain output will be pulled low if the
regulator output exits a ±8% window around the regulation
point. This condition is released once regulation within
a 5% window is achieved. To prevent unwanted PGOOD
glitches during transients or dynamic VOUT changes, the
LTC3601 PGOOD falling edge includes a filter time of ap-
proximately 40µs.
VIN Overvoltage Protection
In order to protect the internal power MOSFET devices
against transient voltage spikes, the LTC3601 constantly
monitors the VIN pin for an overvoltage condition. When
VIN rises above 17.5V, the regulator suspends operation
by shutting off both power MOSFETs. Once VIN drops
below 16.5V, the regulator immediately resumes normal
operation. The regulator does not execute its soft-start
function when exiting an overvoltage condition.
Short-Circuit Protection
Foldback current limiting is provided in the event the
output is inadvertently shorted to ground. During this
condition the internal current limit (ILIM) will be lowered
to approximately one-third its normal value. This feature
reduces the heat dissipation in the LTC3601 during short-
circuit conditions and protects both the IC and the input
supply from any potential damage.
LTC3601
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APPLICATIONS INFORMATION
A general LTC3601 application circuit is shown on the first
page of this data sheet. External component selection is
largely driven by the load requirement and begins with the
selection of the inductor L. Once the inductor is chosen,
the input capacitor, CIN, the output capacitor, COUT
, the
internal regulator capacitor, CINTVCC, and the boost capaci-
tor, CBOOST, can be selected. Next, the feedback resistors
are selected to set the desired output voltage. Finally, the
remaining optional external components can be selected
for functions such as external loop compensation, track/
soft-start, externally programmed oscillator frequency
and PGOOD.
Operating Frequency
Selection of the operating frequency is a trade-off between
efficiency and component size. High frequency operation
allows the use of smaller inductor and capacitor values.
Operation at lower frequencies improves efficiency by
reducing internal gate charge losses but requires larger
inductance values and/or capacitance to maintain low
output ripple voltage.
The operating frequency, fO, of the LTC3601 is determined
by an external resistor that is connected between the RT
pin and ground. The value of the resistor sets the ramp
current that is used to charge and discharge an internal
timing capacitor within the oscillator and can be calculated
by using the following equation:
RRT =
3.2 E11
f
O
where RRT is in Ω and fO is in Hz.
Connecting the RT pin to INTVCC will default the converter
to fO = 2MHz; however, this switching frequency will be
more sensitive to process and temperature variations than
when using a resistor on RT (see Typical Performance
Characteristics).
Inductor Selection
For a given input and output voltage, the inductor value and
operating frequency determine the inductor ripple current.
More specifically, the inductor ripple current decreases
with higher inductor value or higher operating frequency
according to the following equation:
ΔIL=VOUT
f • L
1– VOUT
VIN
where ∆IL = inductor ripple current, f = operating frequency
and L = inductor value. A trade-off between component
size, efficiency and operating frequency can be seen from
this equation. Accepting larger values of ∆IL allows the use
of lower value inductors but results in greater core loss
in the inductor, greater ESR loss in the output capacitor,
and larger output ripple. Generally, highest efficiency op-
eration is obtained at low operating frequency with small
ripple current.
A reasonable starting point for setting the ripple current is
about 40% of IOUT(MAX). Note that the largest ripple current
occurs at the highest VIN. To guarantee the ripple current
does not exceed a specified maximum the inductance
should be chosen according to:
L=VOUT
f • ΔIL(MAX)
1– VOUT
VIN(MAX)
However, the inductor ripple current must not be so large
that its valley current level exceeds the negative current
limit of –1.2A (typical) when the circuit is operating in
forced continuous mode. If the inductor current trough
reaches the negative current limit while the part is in
forced continuous mode operation, VOUT may charge up
to above its target regulation voltage. In such instances,
choose a larger inductor value to reduce the ripple current.
The alternative is to reduce the inductor ripple current by
decreasing the RT resistor value, which will increase the
switching frequency.
RT (kΩ)
0
0
FREQUENCY (kHz)
1000
2000
3000
4000
6000
100 200 300 400
3601 F01
500 600
5000
Figure 1. Switching Frequency vs RT
LTC3601
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APPLICATIONS INFORMATION
Once the value for L is known the type of inductor must
be selected. Actual core loss is independent of core size
for a fixed inductor value but is very dependent on the
inductance selected. As the inductance increases, core loss
decreases. Unfortunately, increased inductance requires
more turns of wire leading to increased copper loss.
Ferrite designs exhibit very low core loss and are pre-
ferred at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core materials saturate “hard,” meaning the induc-
tance collapses abruptly when the peak design current is
exceeded. This collapse will result in an abrupt increase
in inductor ripple current, so it is important to ensure the
core will not saturate.
Different core materials and shapes will change the
size/current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy materi-
als are small and don’t radiate much energy but generally
cost more than powdered iron core inductors with similar
characteristics. The choice of which style inductor to use
mainly depends on the price versus size requirements and
any radiated field/EMI requirements. New designs for sur-
face mount inductors are available from Toko, Vishay, NEC/
Tokin, Cooper, Coilcraft, TDK and Wurth Electronik. Table
1 gives a sampling of available surface mount inductors.
CIN and COUT Selection
The input capacitance, CIN, is needed to filter the trapezoi-
dal wave current at the drain of the top power MOSFET.
To prevent large voltage transients from occurring a low
ESR input capacitor sized for the maximum RMS current
is recommended. The maximum RMS current is given by:
IRMS =IOUT(MAX)
VOUT V
IN – VOUT
(
)
V
IN
where IOUT(MAX) equals the maximum average output
current. This formula has a maximum at VIN = 2VOUT
,
where IRMS = IOUT/2. This simple worst-case condition
is commonly used for design because even significant
deviations do not offer much relief. Note that ripple cur-
rent ratings from capacitor manufacturers are often based
on only 2000 hours of life which makes it advisable to
further de-rate the capacitor or choose a capacitor rated
at a higher temperature than required.
Several capacitors may be paralleled to meet the require-
ments of the design. For low input voltage applications
sufficient bulk input capacitance is needed to minimize
transient effects during output load changes. Even though
the LTC3601 design includes an overvoltage protection
circuit, care must always be taken to ensure input voltage
transients do not pose an overvoltage hazard to the part.
The selection of COUT is primarily determined by the effec-
tive series resistance (ESR) that is required to minimize
Table 1. Inductor Selection Table
INDUCTANCE
(µH)
DCR
(mΩ)
MAX
CURRENT (A)
DIMENSIONS
(mm)
HEIGHT
(mm)
Würth Electronik WE-PD2 Typ MS Series
0.56
0.82
1.2
1.7
2.2
9.5
14
21
27
36
6.5
5.4
4.8
4
3.6
5.2 × 5.8 2
Vishay IHLP-2020BZ-01 Series
0.47
0.68
1
2.2
8.8
12.4
20
50.1
11.5
10
7
4.2
5.2 × 5.5 2
Toko DE3518C Series
0.56
1.2
1.7
24
30
35
3.3
2.4
2.1
3.5 × 3.7 1.8
Sumida CDRH2D18/HP Series
0.56
0.82
1.1
33
39
43
3.7
2.9
2.5
3.2 × 3.2 2
Cooper SD18 Series
0.47
0.82
1.2
1.5
2.2
20.1
24.7
29.4
34.5
39.8
3.58
3.24
2.97
2.73
2.55
5.5 × 5.5 1.8
Coilcraft LPS4018 Series
0.56
1
2.2
30
40
70
4.8
2.8
2.7
4 × 4 1.7
TDK VLS252012 Series
0.47
1
1.5
2.2
56
88
126
155
3.3
2.4
2
1.8
2.5 × 2 1.2
LTC3601
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APPLICATIONS INFORMATION
voltage ripple and load step transients. The output ripple,
∆VOUT, is determined by:
ΔVOUT <ΔILESR +1
8 • f • COUT
The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage. Multiple capaci-
tors placed in parallel may be needed to meet the ESR
and RMS current handling requirements. Dry tantalum,
special polymer, aluminum electrolytic, and ceramic
capacitors are all available in surface mount packages.
Special polymer capacitors offer low ESR but have lower
capacitance density than other types. Tantalum capacitors
have the highest capacitance density, but it is important
to only use types that have been surge tested for use in
switching power supplies. Aluminum electrolytic capacitors
have significantly higher ESR but can be used in cost-
sensitive applications provided that consideration is given
to ripple current ratings and long-term reliability. Ceramic
capacitors have excellent low ESR characteristics and small
footprints. Their relatively low value of bulk capacitance
may require multiple capacitors in parallel.
Using Ceramic Input and Output Capacitors
Higher value, lower cost ceramic capacitors are now
available in small case sizes. Their high voltage rating
and low ESR make them ideal for switching regulator
applications. However, due to the self-resonant and high-Q
characteristics of some types of ceramic capacitors, care
must be taken when these capacitors are used at the input
and output. When a ceramic capacitor is used at the input,
and the power is supplied by a wall adapter through long
wires, a load step at the output can induce ringing at the
VIN input. At best, this ringing can couple to the output and
be mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause a
voltage spike at VIN large enough to damage the part. For
a more detailed discussion, refer to Application Note 88.
When choosing the input and output ceramic capacitors
choose the X5R or X7R dielectric formulations. These
dielectrics provide the best temperature and voltage
characteristics for a given value and size.
INTVCC Regulator Bypass Capacitor
An internal low dropout (LDO) regulator produces a
3.3V supply voltage used to power much of the internal
LTC3601 circuitry including the power MOSFET gate
drivers. The INTVCC pin connects to the output of this
regulator and must have a minimum of 1µF of decoupling
capacitance to ground. The decoupling capacitor should
have low impedance electrical connections to the INTVCC
and PGND pins to provide the transient currents required
by the LTC3601. The user may connect a maximum load
current of 5mA to this pin but must take into account the
increased power dissipation and die temperature that
results. Furthermore, this supply is intended only to supply
additional DC load currents as desired and not intended
to regulate large transient or AC behavior this may impact
LTC3601 operation.
Boost Capacitor
The boost capacitor, CBOOST
, is used to create a voltage rail
above the applied input voltage VIN. Specifically, the boost
capacitor is charged to a voltage equal to approximately
INTVCC each time the bottom power MOSFET is turned
on. The charge on this capacitor is then used to supply
the required transient current during the remainder of the
switching cycle. When the top MOSFET is turned on, the
BOOST pin voltage will be equal to approximately VIN +
3.3V. For most applications a 0.1µF ceramic capacitor will
provide adequate performance.
Output Voltage Programming
The LTC3601 will adjust the output voltage such that VFB
equals the reference voltage of 0.6V according to:
VOUT =0.6V 1+R1
R2
The desired output voltage is set by appropriate selection of
resistors R1 and R2 as shown in Figure 2. Choosing large
values for R1 and R2 will result in improved efficiency but
may lead to undesirable noise coupling or phase margin
reduction due to stray capacitances at the FB node. Care
should be taken to route the FB line away from any noise
source, such as the SW line.
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APPLICATIONS INFORMATION
To improve the frequency response of the main control
loop a feedforward capacitor, CF
, may be used as shown
in Figure 2.
programmed value. This is an acceptable result in many
applications, so this constraint may not be of critical
importance in most cases, and high switching frequen-
cies may be used in the design without any fear of severe
consequences. As the sections on Inductor and Capacitor
Selection show, high switching frequencies allow the use
of smaller board components, thus reducing the footprint
of the application circuit.
Internal/External Loop Compensation
The LTC3601 provides the option to use a fixed internal
loop compensation network to reduce both the required
external component count and design time. The internal
loop compensation network can be selected by connect-
ing the ITH pin to the INTVCC pin. To ensure stability, it
is recommended that the output capacitance be at least
47µF when using internal compensation. Alternatively,
the user may choose specific external loop compensation
components to optimize the main control loop transient
response as desired. External loop compensation is chosen
by simply connecting the desired network to the ITH pin.
Suggested compensation component values are shown in
Figure 3. For a 2MHz application, an R-C network of 220pF
and 13kΩ provides a good starting point. The bandwidth
of the loop increases with decreasing C. If R is increased
by the same factor that C is decreased, the zero frequency
will be kept the same, thereby keeping the phase the same
in the most critical frequency range of the feedback loop.
A 10pF bypass capacitor on the ITH pin is recommended
for the purposes of filtering out high frequency coupling
from stray board capacitance. In addition, a feedforward
capacitor CF can be added to improve the high frequency
response, as previously shown in Figure 2. Capacitor CF
provides phase lead by creating a high frequency zero
with R1 which improves the phase margin.
FB
R1
R2
CF
3601 F02
VOUT
SGND
LTC3601
Figure 2. Optional Feedforward Capacitor
Figure 3. Compensation Components
ITH
RCOMP
13k
CCOMP
220pF
CBYP
3601 F03
SGND
LTC3601
Minimum Off-Time/On-Time Considerations
The minimum off-time is the smallest amount of time that
the LTC3601 can turn on the bottom power MOSFET, trip
the current comparator and turn the power MOSFET back
off. This time is typically 40ns. For the controlled on-time
current mode control architecture, the minimum off-time
limit imposes a maximum duty cycle of:
DC(MAX) =1– f • tOFF(MIN)
( )
where f is the switching frequency and tOFF(MIN) is the
minimum off-time. If the maximum duty cycle is surpassed,
due to a dropping input voltage for example, the output
will drop out of regulation. The minimum input voltage to
avoid this dropout condition is:
VIN(MIN) =
V
OUT
1−f • tOFF(MIN)
( )
Conversely, the minimum on-time is the smallest dura-
tion of time in which the top power MOSFET can be in
its “on” state. This time is typically 20ns. In continuous
mode operation, the minimum on-time limit imposes a
minimum duty cycle of:
DC(MIN) =f • tON(MIN)
( )
where tON(MIN) is the minimum on-time. As the equation
shows, reducing the operating frequency will alleviate the
minimum duty cycle constraint.
In the rare cases where the minimum duty cycle is
surpassed, the output voltage will still remain in regula-
tion, but the switching frequency will decrease from its
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Checking Transient Response
The regulator loop response can be checked by observing
the response of the system to a load step. When config-
ured for external compensation, the availability of the
ITH pin not only allows optimization of the control loop
behavior but also provides a DC coupled and AC filtered
closed-loop response test point. The DC step, rise time,
and settling behavior at this test point reflect the system’s
closed-loop response. Assuming a predominantly second
order system, the phase margin and/or damping factor can
be estimated by observing the percentage of overshoot
seen at this pin. The ITH external components shown in
Figure 3 will provide an adequate starting point for most
applications. The series R-C filter sets the pole-zero loop
compensation. The values can be modified slightly, from
approximately 0.5 to 2 times their suggested values, to
optimize transient response once the final PC layout is
done and the particular output capacitor type and value
have been determined. The output capacitors need to be
selected because their various types and values determine
the loop feedback factor gain and phase. An output cur-
rent pulse of 20% to 100% of full load current with a rise
time of 1µs to 10µs will produce output voltage and ITH
pin waveforms that will give a sense of the overall loop
stability without breaking the feedback loop
When observing the response of VOUT to a load step, the
initial output voltage step may not be within the bandwidth
of the feedback loop. As a result, the standard second
order overshoot/DC ratio cannot be used to estimate
phase margin. The output voltage settling behavior is
related to the stability of the closed-loop system and will
demonstrate the actual overall supply performance. For
a detailed explanation of optimizing the compensation
components, including a review of control loop theory,
refer to Linear Technology Application Note 76. As shown
in Figure 2 a feedforward capacitor, CF
, may be added
across feedback resistor R1 to improve the high frequency
response of the system. Capacitor CF provides phase lead
by creating a high frequency zero with R1.
APPLICATIONS INFORMATION
In some applications severe transients can be caused by
switching in loads with large (>10µF) input capacitors. The
discharged input capacitors are effectively put in parallel
with COUT
, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this output droop if the
switch connecting the load has low resistance and is driven
quickly. The solution is to limit the turn-on speed of the
load switch driver. A Hot Swap™ controller is designed
specifically for this purpose and usually incorporates cur-
rent limit, short-circuit protection and soft-start functions.
MODE/SYNC Operation
The MODE/SYNC pin is a multipurpose pin allowing both
mode selection and operating frequency synchronization.
Connecting this pin to INTVCC enables Burst Mode operation
for superior efficiency at low load currents at the expense
of slightly higher output voltage ripple. When the MODE/
SYNC pin is pulled to ground, forced continuous mode
operation is selected creating the lowest fixed output ripple
at the expense of light load efficiency.
The LTC3601 will detect the presence of the external clock
signal on the MODE/SYNC pin and synchronize the internal
oscillator to the phase and frequency of the incoming clock.
The presence of an external clock will place the LTC3601
into forced continuous mode operation.
Output Voltage Tracking and Soft-Start
The LTC3601 allows the user to control the output voltage
ramp rate by means of the TRACK pin. From 0V to 0.6V
the TRACK pin will override the internal reference input
to the error amplifier forcing regulation of the feedback
voltage to that seen at the TRACK pin. When the voltage
at the TRACK pin rises above 0.6V, tracking is disabled
and the feedback voltage will be regulated to the internal
reference voltage.
The voltage at the TRACK pin may be driven from an ex-
ternal source, or alternatively, the user may leverage the
internal 1.4µA pull-up current on TRACK to implement
LTC3601
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APPLICATIONS INFORMATION
a soft-start function by connecting a capacitor from the
TRACK pin to ground. The relationship between output
rise time and TRACK capacitance is given by:
tSS = 430,000 × CTRACK
A default internal soft-start timer forces a minimum soft-
start time of 400µs by overriding the TRACK pin input
during this time period. Hence, capacitance values less
than approximately 1000pF will not significantly affect
soft-start behavior.
When using the TRACK pin, the regulator defaults to
Burst Mode operation until the output exceeds 80% of
its final value (VFB > 0.48V). Once the output reaches this
voltage, the operating mode of the regulator switches to
the mode selected by the MODE/SYNC pin as described
above. During normal operation, if the output drops below
10% of its final value (as it may when tracking down, for
instance), the regulator will automatically switch to Burst
Mode operation to prevent inductor saturation and improve
TRACK pin accuracy.
Output Power Good
The PGOOD output of the LTC3601 is driven by a 15Ω
(typical) open-drain pull-down device. This device will be
turned off once the output voltage is within 5% (typical) of
the target regulation point allowing the voltage at PGOOD
to rise via an external pull-up resistor (100k typical). If
the output voltage exits a 8% (typical) regulation window
around the target regulation point the open-drain output
will pull down with 15Ω output resistance to ground, thus
dropping the PGOOD pin voltage. A filter time of 40µs
(typical) acts to prevent unwanted PGOOD output changes
during VOUT transient events. As a result, the output volt-
age must be within the target regulation window of 5%
for 40µs before the PGOOD pin is pulled high. Conversely,
the output voltage must exit the 8% regulation window for
40µs before the PGOOD pin pulls to ground (see Figure 4).
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
% Efficiency = 100% – (L1 + L2 + L3 +…)
where L1, L2, etc. are the individual loss terms as a per-
centage of input power.
Although all dissipative elements in the circuit produce
losses, three main sources account for the majority of the
losses in the LTC3601: 1) I2R loss, 2) switching losses
and quiescent current loss, 3) transition losses and other
system losses.
1. I2R loss is calculated from the DC resistances of the
internal switches, RSW
, and external inductor, RL.
In continuous mode, the average output current will
flow through inductor L but is “chopped” between the
internal top and bottom power MOSFETs. Thus, the
series resistance looking into the SW pin is a function
of both the top and bottom MOSFET’s RDS(ON) and the
duty cycle (DC) as follows:
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
PGOOD
VOLTAGE
VOUT
–8% –5% 5% 8%
3601 F04
0%
NOMINAL OUTPUT
Figure 4. PGOOD Pin Behavior
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The RDS(ON) for both the top and bottom MOSFETs can be
obtained from the Typical Performance Characteristics
curves. Thus to obtain I2R loss:
“I2R LOSS” = IOUT2 · (RSW + RL)
2. The internal LDO supplies the power to the INTVCC rail.
The total power loss here is the sum of the switching
losses and quiescent current losses from the control
circuitry.
Each time a power MOSFET gate is switched from low
to high to low again, a packet of charge dQ moves
from VIN to ground. The resulting dQ/dt is a current
out of INTVCC that is typically much larger than the DC
control bias current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges
of the internal top and bottom power MOSFETs and f
is the switching frequency. For estimation purposes,
(QT + QB) on the LTC3601 is approximately 1nC.
To calculate the total power loss from the LDO load,
simply add the gate charge current and quiescent cur-
rent and multiply by VIN:
PLDO = (IGATECHG + IQ) • VIN
3. Other “hidden” losses such as transition loss, cop-
per trace resistances, and internal load currents can
account for additional efficiency degradations in the
overall power system. Transition loss arises from the
brief amount of time the top power MOSFET spends in
the saturated region during switch node transitions. The
LTC3601 internal power devices switch quickly enough
that these losses are not significant compared to other
sources.
Other losses, including diode conduction losses during
dead time and inductor core losses, generally account
for less than 2% total additional loss.
APPLICATIONS INFORMATION
Thermal Considerations
The LTC3601 requires the exposed package backplane
metal (PGND pin on the QFN, SGND pin on the MSOP
package) to be well soldered to the PC board to provide
good thermal contact. This gives the QFN and MSOP
packages exceptional thermal properties, compared to
other packages of similar size, making it difficult in normal
operation to exceed the maximum junction temperature
of the part. In many applications, the LTC3601 does not
dissipate much heat due to its high efficiency and low
thermal resistance package backplane. However, in applica-
tions in which the LTC3601 is running at a high ambient
temperature, high input voltage, high switching frequency,
and maximum output current, the heat dissipated may
exceed the maximum junction temperature of the part. If
the junction temperature reaches approximately 150°C,
both power switches will be turned off until temperature
decreases approximately 10°C.
Thermal analysis should always be performed by the user
to ensure the LTC3601 does not exceed the maximum
junction temperature.
The temperature rise is given by:
TRISE = PDθJA
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to
the ambient temperature.
Consider the example in which an LTC3601EUD is operat-
ing with IOUT = 1.5A, VIN = 12V, f = 4MHz, VOUT = 1.8V,
and an ambient temperature of 70°C. From the Typical
Performance Characteristics section the RDS(ON) of the top
switch is found to be nominally 130mΩ while that of the
bottom switch is nominally 100mΩ yielding an equivalent
power MOSFET resistance RSW of:
RDS(ON)TOP • 1.8/12 + RDS(ON)BOT • 10.2/12 = 105mΩ.
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From the previous section, IGATECHG is ~4mA when f =
4MHz, and the spec table lists the typical IQ to be 1mA.
Therefore, the total power dissipation due to resistive
losses and LDO losses is:
PD = IOUT2 • RSW + VIN • (IGATECHG + IQ)
PD = (1.5)2 • (0.105) + 12V • 5mA = 296mW
The QFN 3mm × 3mm package junction-to-ambient thermal
resistance, θJA, is around 45°C/W. Therefore, the junction
temperature of the regulator operating in a 70°C ambient
temperature is approximately:
TJ = 0.296 • 45 + 70 = 83.3°C
which is well below the specified maximum junction
temperature of 125°C.
Board Layout Considerations
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3601.
1. Do the capacitors CIN connect to VIN and PGND as close
to the pins as possible? These capacitors provide the AC
current to the internal power MOSFETs and drivers. The
(–) plate of CIN should be closely connected to PGND
and the (–) plate of COUT.
2. The output capacitor, COUT
, and inductor L1 should
be closely connected to minimize loss. The (–) plate
of COUT should be closely connected to PGND and the
(–) plate of CIN.
3. The resistive divider, R1 and R2, must be connected
between the (+) plate of COUT and a ground line termi-
nated near SGND. The feedback signal, VFB, should be
routed away from noisy components and traces such as
the SW line, and its trace length should be minimized.
In addition, RT and the loop compensation components
should be terminated to SGND.
4. Keep sensitive components away from the SW pin. The
RRT resistor, the feedback resistors, the compensation
components, and the INTVCC bypass capacitor should
all be routed away from the SW trace and the inductor.
5. A ground plane is preferred, but if not available the
signal and power grounds should be segregated with
both connecting to a common, low noise reference
point. The point at which the ground terminals of the
VIN and VOUT bypass capacitors are connected makes a
good, low noise reference point. The connection to the
PGND pin should be made with a minimal resistance
trace from the reference point.
6. Flood all unused areas on all layers with copper in order
to reduce the temperature rise of power components.
These copper areas should be connected to the exposed
backside connection of the IC.
APPLICATIONS INFORMATION
LTC3601
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Design Example
As a design example, consider using the LTC3601 in an
application with the following specifications:
VIN = 12V, VOUT = 1.8V, IOUT(MAX) = 1.5A, IOUT(MIN) =
10mA, f = 1MHz
Because efficiency is important at both high and low load
currents, Burst Mode operation is selected.
First, the correct RRT resistor value for 1MHz switching
frequency must be chosen. Based on the equation dis-
cussed earlier, RRT should be 324k.
Next, determine the inductor value for approximately 40%
ripple current using:
L=1.8V
1MHz • 600mA
1– 1.8V
12V
=2.55µH
A standard value 2.2µH inductor will work well for this
application.
Next, COUT is selected based on the required output
transient performance and the required ESR to satisfy
the output voltage ripple. For this design, a 22µF ceramic
capacitor will be used.
CIN should be sized for a maximum current rating of:
IRMS =1.5A 1.8V 12V – 1.8V
( )
12V
=0.54A
Decoupling the VIN pins with a 22µF ceramic capacitor
should be adequate for most applications. A 0.1µF boost
capacitor should also work for most applications.
To save board space the ITH pin is connected to the INTVCC
pin to select an internal compensation network.
The PGOOD pin is connected to VIN through a 100k resistor.
Figure 7. 1.8V, 1.5A Regulator at 1MHz
APPLICATIONS INFORMATION
VIN
MODE/SYNC
RUN
INTVCC
PGOOD
ITH
RT
LTC3601
PGNDSGND
BOOST
SW
VON
FB
TRACK
2.2µF
CFWD
10pF
C1
0.1µF
COUT
47µF
VOUT
1.8V
1.5A
R3
80k
R4
40k
100k
324k CIN: TDK C3225X5R1C226M
COUT: TDK C3225X5R0J476M
L1: VISHAY IHLP2020BZER2R2M01
3601 F05
L1
2.2µH
CIN
22µF
VIN
4V TO 15V
LTC3601
21
3601fc
For more information www.linear.com/LTC3601
TYPICAL APPLICATIONS
12V Input to 1.8V Output at 4MHz Synchronized
Frequency with 6V UVLO and 4.3ms Soft-Start
Efficiency vs Load Current
VIN
RUN
INTVCC
PGOOD
ITH
RT
LTC3601
PGND EXTERNAL
CLOCK
SGND
BOOST
SW
VON
FB
TRACK
MODE/SYNC
2.2µF
270pF
10pF
0.1µF
10nF
COUT
22µF
VOUT
1.8V
1.5A
80k
40k
154k
40k
100k
80k
10k
3601 TA02a
L1
0.68µH
CIN
22µF
VIN
12V
CIN: TDK C3225X5R1C226M
COUT: TDK C3216X5R0J226M
L1: VISHAY IHLP2020BZERR68M01
LOAD CURRENT (A)
0.01
40
EFFICIENCY (%)
60
90
80
0.1 1 10
3601 TA02b
20
30
50
70
10
0
LTC3601
22
3601fc
For more information www.linear.com/LTC3601
8.4V Input to 3.3V Output at 2MHz Operating
Frequency Using Forced Continuous Mode
Efficiency vs Load Current
VIN
RUN
PGOOD
TRACK
LTC3601
PGNDSGND
INTVCC
ITH
RT
MODE/SYNC
BOOST
SW
VON
FB
0.1µF
COUT
47µF
VOUT
3.3V
1.5A
R1
90.9k
CFF
10pF
R2
20k
3601 TA03a
L1
2.2µH
C2
2.2µF
C1
47µF
VIN
8.4V
CIN: TDK C3225X5R1C476M
COUT: TDK C3216X5R0J476M
L1: VISHAY IHLP2020BZER2R2M01
LOAD CURRENT (A)
0.01
40
EFFICIENCY (%)
50
60
70
80
0.1 1 10
3601 TA03b
30
20
10
0
90
100
TYPICAL APPLICATIONS
LTC3601
23
3601fc
For more information www.linear.com/LTC3601
UD Package
16-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1691)
3.00 ± 0.10
(4 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
1.45 ± 0.05
(4 SIDES)
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-2)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
PIN 1
TOP MARK
(NOTE 6)
0.40 ± 0.10
BOTTOM VIEW—EXPOSED PAD
1.45 ± 0.10
(4-SIDES)
0.75 ± 0.05 R = 0.115
TYP
0.25 ± 0.05
1
PIN 1 NOTCH R = 0.20 TYP
OR 0.25 × 45° CHAMFER
15 16
2
0.50 BSC
0.200 REF
2.10 ± 0.05
3.50 ± 0.05
0.70 ±0.05
0.00 – 0.05
(UD16) QFN 0904
0.25 ±0.05
0.50 BSC
PACKAGE
OUTLINE
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
LTC3601
24
3601fc
For more information www.linear.com/LTC3601
MSOP (MSE16) 0213 REV F
0.53 ±0.152
(.021 ±.006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
0.86
(.034)
REF
0.50
(.0197)
BSC
16
161514131211 10
12345678
9
9
18
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.10
(.201)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ±0.127
(.035 ±.005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 ±0.038
(.0120 ±.0015)
TYP
0.50
(.0197)
BSC
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
2.845 ±0.102
(.112 ±.004)
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102
(.065 ±.004)
0.1016 ±0.0508
(.004 ±.002)
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
0.280 ±0.076
(.011 ±.003)
REF
4.90 ±0.152
(.193 ±.006)
DETAIL “B”
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
0.12 REF
0.35
REF
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev F)
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
LTC3601
25
3601fc
For more information www.linear.com/LTC3601
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER
A 4/10 Changed Temperature Range for E-Grade to –40°C to 125°C in Order Information Sections
Updated Note 2
Updated Pin Functions
Updated Functional Block Diagram
Updated Equation in Applications Information Section
Updated Related Parts
2
3
7
8
11, 15
26
B 11/11 Changed Units from mV to V on the VF_TRACK specification
Updated Axis labels on graphs G17 and G18
3
6
C 08/15 Added Negative Valley Switch Current Limit
Modified Inductor Selection section
3
10
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LTC3601
26
3601fc
LINEAR TECHNOLOGY CORPORATION 2009
LT 0815 REV C • PRINTED IN USA
For more information www.linear.com/LTC3601
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
LTC3602 10V, 2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 10V, VOUT(MIN) = 0.6V, IQ = 75µA,
ISD < 1µA, 4mm × 4mm QFN-20, TSSOP-16E
LTC3603 15V, 2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 15V, VOUT(MIN) = 0.6V, IQ = 75µA,
ISD < 1µA, 4mm × 4mm QFN-20
LTC3604 15V, 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 3.6V to 15V, VOUT(MIN) = 0.6V, IQ = 300µA,
ISD < 14mA, 3mm × 3mm QFN-16 and MSOP-16E Packages
LTC3605 15V, 5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 15V, VOUT(MIN) = 0.6V, IQ = 2mA,
ISD < 15µA, 4mm × 4mm QFN-24
LTC3610 24V, 12A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 24V, VOUT(MIN) = 0.6V, IQ = 900µA,
ISD < 15µA, 9mm × 9mm QFN-64
LTC3611 32V, 10A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 32V, VOUT(MIN) = 0.6V, IQ = 900µA,
ISD < 15µA, 9mm × 9mm QFN-64
TYPICAL APPLICATION
1.2V Output at 2MHz Operating Frequency
VIN
RUN
PGOOD
TRACK
LTC3601
PGNDSGND
INTVCC
ITH
RT
MODE/SYNC
BOOST
SW
VON
FB
0.1µF
COUT
47µF
VOUT
1.2V
1.5A
R1
20k
CFF
10pF
R2
20k
3601 TA04
L1
1µH
C2
2.2µF
C1
22µF
VIN
12V
CIN: TDK C3225X5R1C226M
COUT: TDK C3225X5R0J476M
L1: VISHAY IHLP2020BZER1R0M01
Efficiency vs Load Current
LOAD CURRENT (A)
0.01
40
EFFICIENCY (%)
50
60
70
80
0.1 1 10
3601 TA04b
30
20
10
0
90
100
VIN = 8V
VIN = 15V
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LTC3601
