LT3748
1
3748fb
For more information www.linear.com/LT3748
Features
applications
Description
100V Isolated
Flyback Controller
The LT
®
3748 is a switching regulator controller specifically
designed for the isolated flyback topology and capable of
high power. It drives a low side external N-channel power
MOSFET from an internally regulated 7V supply. No third
winding or opto-isolator is required for regulation as the
part senses the isolated output voltage directly from the
primary-side flyback waveform.
The LT3748 utilizes boundary mode to provide a small
magnetic solution without compromising load regulation.
Operating frequency is set by load current and transformer
magnetizing inductance. The gate drive of the LT3748
combined with a suitable external MOSFET allow it to
deliver load power up to several tens of watts from input
voltages as high as 100V.
The LT3748 is available in a high voltage 16-lead MSOP
package with four leads removed.
25W, 12V Output, Isolated Telecom Supply
n 5V to 100V Input Voltage Range
n 1.9A Average Gate Drive Source and Sink Current
n Boundary Mode Operation
n No Transformer Third Winding or Opto-Isolator
Required for Regulation
n Primary-Side Winding Feedback Load Regulation
n VOUT Set with Two External Resistors
n INTVCC Pin for Control of Gate Driver Voltage
n Programmable Soft Start
n Programmable Undervoltage Lockout
n Available in MSOP Package
n Isolated Telecom Converters
n High Power Automotive Supplies
n Isolated Industrial Power Supplies
n Military and High Temperature Applications
Output Load and Line Regulation
typical application
L, LT, LTC, LT M, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5438499 and 7471522.
EN/UVLO
TC
SS
RFB
RREF
VCGND INTVCC
LT3748
3748 TA01a
56.2k 2nF
VIN
36V TO 72V
VOUT
+
12V
2A
VOUT–
VIN
4:1
412k
15.4k
10µF3.8µH60.8µH100µF
GATE
SENSE
4700pF 4.7µF
6.04k
243k
0.033Ω
10k LOAD CURRENT (A)
0
11.4
VOUT (V)
11.6
11.8
12.0
12.2
12.4
12.6
0.5 1.0 1.5 2.0
3748 TA01b
VIN = 72V
VIN = 48V
VIN = 36V
LT3748
2
3748fb
For more information www.linear.com/LT3748
absolute MaxiMuM ratings
VIN, RFB ................................................................... 100V
VIN to RFB ..................................................................±5V
EN/UVLO ......................................................–0.3V, 100V
INTVCC ....................................................VIN + 0.3V, 20V
SS, VC, TC, RREF..........................................................6V
SENSE ......................................................................0.4V
Operating Junction Temperature Range (Note 2)
LT3748E/LT3748I ...............................–40°C to 125°C
LT3748H ............................................–40°C to 150°C
LT3748MP ......................................... –55°C to 150°C
Storage Temperature Range ..................–65°C to 150°C
orDer inForMation
electrical characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, unless otherwise noted.
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT3748EMS#PBF LT3748EMS#TRPBF 3748 16-Lead Plastic MSOP –40°C to 125°C
LT3748IMS#PBF LT3748IMS#TRPBF 3748 16-Lead Plastic MSOP –40°C to 125°C
LT3748HMS#PBF LT3748HMS#TRPBF 3748 16-Lead Plastic MSOP –40°C to 150°C
LT3748MPMS#PBF LT3748MPMS#TRPBF 3748 16-Lead Plastic MSOP –55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
PARAMETER CONDITIONS MIN TYP MAX UNITS
Input Voltage Range l5 100 V
Quiescent Current Not Switching
VEN/UVLO = 0.2V 1.3
01.75
1mA
µA
VIN Quiescent Current, INTVCC Overdriven VINTVCC = 10V 300 450 µA
INTVCC Voltage Range l4.5 20 V
INTVCC Pin Regulation Voltage 6.8 7 7.2 V
INTVCC Dropout (VIN – VINTVCC), IINTVCC = 10mA, VIN = 5V 0.7 V
INTVCC Undervoltage Lockout Falling Threshold l3.45 3.6 3.75 V
EN/UVLO Pin Threshold EN/UVLO Pin Voltage Rising l1.19 1.223 1.25 V
EN/UVLO Pin Hysteresis Current EN/UVLO = 1V 1.9 2.4 2.9 µA
Soft-Start Current VSS = 0.4V (Note 3) 5 µA
Soft-Start Threshold 0.65 V
Soft-Start Reset Current 3 mA
(Note 1)
pin conFiguration
1
3
5
6
7
8
VIN
EN/UVLO
INTVCC
GATE
SENSE
GND
16
14
12
11
10
9
RFB
RREF
TC
VC
SS
GND
TOP VIEW
MS PACKAGE
16 (12)-LEAD PLASTIC MSOP
TJMAX = 150°C, θJA = 90°C/W
LT3748
3
3748fb
For more information www.linear.com/LT3748
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3748E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design
characterization and correlation with statistical process controls. The
LT3748I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT3748H is guaranteed over the full –40°C to
150°C operating junction temperature range. The LT3748MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperatures greater than 125°C.
Note 3: Current flows out of the pin.
PARAMETER CONDITIONS MIN TYP MAX UNITS
Maximum SENSE Current Limit Threshold VC = 2.2V
l
95
90 100
100 105
110 mV
mV
Minimum SENSE Current Limit Threshold VC = 0V 15 mV
Maximum to Minimum SENSE Threshold
Ratio
l5.2 6.6 8.2 mV/mV
SENSE Overcurrent Threshold VC = 2.2V 115 130 145 mV
SENSE Input Bias Current VSENSE = 10mV (Note 3) 10 15 20 µA
RREF Voltage VC = 1.1V
l
1.20
1.195 1.223 1.24
1.245 V
V
RREF Voltage Line Regulation 5V < VIN < 100V 0.005 0.025 %/V
RREF Pin Bias Current (Note 3) l35 500 nA
TC Current into RREF RTC = 20k 27.5 µA
Error Amplifier Voltage Gain 115 V/V
Error Amplifier Transconductance ∆I = 10µA 155 µmhos
VC Source Current VC = 1.1V, VRREF = 0.5V –45 µA
VC Sink Current VC = 1.1V, VRREF = 2V 48 µA
Flyback Comparator Trip Current Current into RFB Pin, RREF = 6.04k 10 µA
Minimum GATE Off-Time 700 ns
Minimum GATE On-Time 250 ns
Maximum Discontinuous Off-Time VC = 0V 24 µs
Maximum GATE Off-Time VRREF = 0.5V 55 µs
Maximum GATE On-Time VSENSE = 0V 55 µs
GATE Output Rise Time CL = 3300pF, 10% to 90% 16 ns
GATE Output Fall Time CL = 3300pF, 10% to 90% 16 ns
GATE Output Low (VOL) 0.05 V
GATE Output High (VOH) VINTVCC – 0.05 V
electrical characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, unless otherwise noted.
LT3748
4
3748fb
For more information www.linear.com/LT3748
typical perForMance characteristics
INTVCC Undervoltage Lockout
vs Temperature
INTVCC Voltage vs VIN Voltage
INTVCC Regulator Dropout
vs INTVCC Current Soft-Start Current vs Temperature
Output Regulation vs Temperature Quiescent Current vs Temperature
INTVCC Voltage vs Temperature
TA = 25°C, unless otherwise noted.
Quiescent Current vs VIN Voltage
INTVCC Dropout vs Temperature
TEMPERATURE (°C)
–55
14.4
VOUT (V)
14.6
14.8
15.0
15.2
0 50 100 150
3748 G01
15.4
15.6
–25 25 75 125
FIGURE 16 CIRCUIT
IOUT = 150mA ON EACH OUTPUT
VIN = 12V
TEMPERATURE (°C)
–55
QUIESCENT CURRENT (mA)
1.3
1.4
1.5
150
3748 G02
1.2
1.1
0.8 050 100
–25 25 75 125
1.0
0.9
1.7
1.6
VSS = 0V
INTVCC = OPEN
VIN = 72V
VIN = 36V
VIN = 12V
VIN = 6V
VIN (V)
0
QUIESCENT CURRENT (mA)
0.6
0.8
1.0
60 100
3748 G03
0.4
0.2
020 40 80
1.2
1.4
1.6 VSS = 0V
INTVCC = OPEN
TEMPERATURE (°C)
–55
6.5
INTVCC VOLTAGE (V)
6.6
6.8
6.9
7.0
7.5
7.2
050 75
3748 G04
6.7
7.3
7.4
7.1
–25 25 100 125 150
IINTVCC = 0mA
IINTVCC = 10mA
VIN VOLTAGE (V)
4 6
V
INTVCC
(V)
6.0
6.5
7.0
60
3748 G05
5.5
5.0
8 20 80
10 40 100
4.5
4.0
7.5 IINTVCC = 0mA
IINTVCC = 10mA
TEMPERATURE (°C)
–55
INTVCC UVLO (V)
3.7
3.8
3.9
25 50 75 100 125
3748 G06
3.6
3.5
–25 0 150
3.4
3.3
4.0
RISING THRESHOLD
FALLING THRESHOLD
INTVCC CURRENT (mA)
0
0
INTV
CC
REGULATOR DROPOUT (V)
0.5
1.0
1.5
2.0
2.5
3.0
VIN = 5V
10 20 30 40
3748 G07
150°C
100°C
25°C
–50°C
TEMPERATURE (°C)
–55
0
INTVCC DROPOUT (V)
0.5
1.0
1.5
2.0
0 50 100 150
3748 G08
2.5
3.0
–25 25 75 125
VIN = 5V
IINTVCC = 20mA
IINTVCC = 10mA
IINTVCC = 5mA
TEMPERATURE (°C)
–55
0
SOFT-START CURRENT (µA)
1
2
3
4
0 50 100 150
3748 G09
5
6
–25 25 75 125
LT3748
5
3748fb
For more information www.linear.com/LT3748
typical perForMance characteristics
Error Amplifier Transconductance
vs Temperature
Error Amplifier Output Current
vs RREF Pin Voltage
SENSE Pin Threshold
vs Temperature
EN/UVLO Current vs Temperature
EN/UVLO Threshold
vs Temperature TC Pin Voltage vs Temperature
TA = 25°C, unless otherwise noted.
Maximum Discontinuous Off-Time
vs Temperature GATE Rise and Fall Time vs Charge
GATE Rise and Fall Time
vs INTVCC Voltage
TEMPERATURE (°C)
–55
0
EN/UVLO CURRENT (µA)
0.5
1.0
1.5
2.0
0 50 100 150
3748 G10
2.5
3.0
–25 25 75 125
VEN/UVLO = 1.1V
VEN/UVLO = 0.9V
VEN/UVLO = 1.3V
TEMPERATURE (°C)
–55
EN/UVLO THRESHOLD (V)
1.20
1.30
150
3748 G11
1.10
1.00 050 100
–25 25 75 125
1.40
1.15
1.25
1.05
1.35
TEMPERATURE (°C)
–55
TC VOLTAGE (V)
0.5
0.6
0.7
150
3748 G12
0.4
0.3
0050 100
–25 25 75 125
0.2
0.1
0.9
0.8
TEMPERATURE (°C)
–55
100
TRANSCONDUCTANCE (µmhos)
110
130
140
150
200
170
050 75
3748 G13
120
180
190
160
–25 25 100 125 150
VIN = 100V
VIN = 6V
VREF (V)
0
I
VC
(µA)
0
20
40
60
2.0
3748 G14
–20
–40
–10
10
30
50
–30
–50
–60 0.5 1.0 1.5 2.5
150°C
100°C
25°C
–50°C
TEMPERATURE (°C)
–55
SENSE THRESHOLD (mV)
80
120
150
3748 G15
40
0050 100
–25 25 75 125
160
60
100
20
140 OVERCURRENT
VC = 2.2V
VC = 0.2V
TEMPERATURE (°C)
–55
20
MAXIMUM DISCONTINUOUS OFF-TIME (µs)
21
23
24
25
30
27
050 75
3748 G16
22
28
29
26
–25 25 100 125 150 TOTAL GATE CHARGE (nC)
0
0
GATE RISE AND FALL TIME (ns)
AVERAGE GATE SOURCE, SINK CURRENT (A)
10
20
30
40
20 40 60 80
3748 G17
100 120
50
0
0.5
1.0
1.5
2.0
RISE TIME
FALL TIME
Q = C • V
VINTVCC = 7V
tr, tf 10% TO 90%
AVERAGE
CURRENT
VINTVCC (V)
0
GATE RISE AND FALL TIME (ns)
10
15
20
3748 G18
5
0510 15
25
20 FALLING
RISING
CGATE = 3.3nF
tr, tf 10% TO 90%
LT3748
6
3748fb
For more information www.linear.com/LT3748
pin Functions
VIN (Pin 1) Input Voltage. This pin supplies current to the
internal start-up circuitry and is the reference voltage for
the feedback circuitry connected to the RFB pin. This pin
must be locally bypassed with a capacitor.
EN/UVLO (Pin 3): Enable/Undervoltage Lockout. A resistor
divider connected to VIN is tied to this pin to program the
minimum input voltage at which the LT3748 will operate.
At a voltage below ~0.5V, the part draws less than 1µA
quiescent current. When below 1.223V but above ~0.5V,
the part will draw quiescent current but will not regulate
the INTVCC supply or power the gate drive circuitry. Above
1.223V, all internal circuitry will start and the SS pin will
source 5μA. When EN/UVLO falls below 1.223V, 2.4μA is
sunk from the pin to provide programmable hysteresis
for undervoltage lockout.
INTVCC (Pin 5): Gate Driver Bias Voltage. This pin supplies
current to the internal gate driver circuitry of the LT3748.
The INTVCC pin must be locally bypassed with a capacitor.
This pin may also be connected to VIN if a third winding
is not used and if VIN ≤ 20V. If a third winding is used,
the INTVCC voltage should be lower than the input voltage
for proper operation.
GATE (Pin 6): N-Channel MOSFET Gate Driver Output.
Switches between INTVCC and GND.
SENSE (Pin 7): The Current Sense Input for the Control
Loop. Kelvin connect this pin to the positive terminal of
the switch current sense resistor, RSENSE, in the source
of the N-channel MOSFET. The negative terminal of the
current sense resistor should be connected to the GND
plane close to the IC.
GND (Pins 8, 9): Ground.
SS (Pin 10): Soft-Start Pin. This pin delays start-up and
clamps VC pin voltage. Soft-start timing is set by the size
of the external capacitor at the pin. Switching starts when
VSS reaches ~0.65V.
VC (Pin 11): Compensation Pin for the Internal Error
Amplifier. Connect a series RC from this pin to ground to
compensate the switching regulator. A 100pF capacitor
in parallel helps eliminate noise.
TC (Pin 12): Output Voltage Temperature Compensation.
Connect a resistor to ground to produce a current pro-
portional to absolute temperature to be sourced into the
RREF node. ITC = 0.55V/RTC.
RREF (Pin 14): Input Pin for the External Ground-Referred
Reference Resistor. The resistor at this pin should be 6.04k,
but for convenience in selecting a resistor divider ratio,
the value may range from 5.76k to 6.34k. The resistor
should be as close to the LT3748 as possible.
RFB (Pin 16): Input Pin for the External Feedback Resistor.
This pin is connected to the transformer primary at the
external MOSFET power switch. The ratio of this resistor
to the RREF resistor, times the internal bandgap reference,
determines the output voltage (plus the effect of any
non-unity transformer turns ratio). The average current
through this resistor during the flyback period should be
approximately 200μA. The resistor should be as close
to the LT3748 as possible.
LT3748
7
3748fb
For more information www.linear.com/LT3748
block DiagraM
MASTER
LATCH
BOUNDARY
MODE DETECT
VARIABLE
DELAY TIMER
INTVCC
CBIAS
R1
R2
CSS
RTC
VOUT+
VOUT–
VIN
TC
SS
VIN
SENSE
VC
T1
NPS:1
5µA
20µA
COUT
DOUT
CIN LPRI LSEC
RFB
RREF
RSENSE
RC
CC
+
–
INTERNAL
REFERENCE
AND
REGULATORS
CURRENT
LIMIT
50µs MAX
ON TIMER
50µs MAX
OFF TIMER
TC
CURRENT
RQ
S
R
S
A1
ERROR AMP
100mV
+
–
+
–A3
+
–
+––
A4
A4
A2
2.4µA
GND
8, 9
3748 BD
Q2Q1
1.223V
1.223V
1.223V
EN/UVLO
RFB
RREF
GATE NMOS
gm
6.04k
11
7
6
5
161
12
14
3
10
LT3748
8
3748fb
For more information www.linear.com/LT3748
operation
The LT3748 is a current mode switching regulator con-
troller designed specifically for the isolated flyback topol-
ogy. The special problem normally encountered in such
circuits is that information relating to the output voltage
on the isolated secondary side of the transformer must
be communicated to the primary side in order to maintain
regulation. Historically, this has been done with opto-
isolators or extra transformer windings. Opto-isolator
circuits waste output power and the extra components
increase the cost and physical size of the power supply.
Opto-isolators can also exhibit trouble due to limited
dynamic response, nonlinearity, unit-to-unit variation
and aging over life. Circuits employing extra transformer
windings also exhibit deficiencies. Using an extra wind-
ing adds to the transformer’s physical size and cost, and
dynamic response is often mediocre.
The LT3748 derives its information about the isolated
output voltage by examining the primary-side flyback
pulse waveform. In this manner, no opto-isolator nor
extra transformer winding is required for regulation. The
output voltage is easily programmed with two resistors.
The LT3748 features a boundary mode control method,
(also called critical conduction mode) where the part
operates at the boundary between continuous conduc-
tion mode and discontinuous conduction mode. Due to
the boundary control mode operation, the output voltage
can be calculated from the transformer primary voltage
when the secondary current is almost zero. This method
improves load regulation without external resistors and
capacitors.
The Block Diagram shows an overall view of the system.
Many of the blocks are similar to those found in traditional
switching regulators, including current comparators, inter-
nal reference and regulators, logic, timers and an N-channel
MOSFET gate driver. The novel sections include a special
sampling error amplifier and a temperature compensation
circuit.
Boundary Mode Operation
Boundary mode is a variable frequency, current mode
switching scheme. The external N-channel MOSFET turns
on and the inductor current increases until it reaches the VC
pin-controlled current limit. After the external MOSFET is
turned off, the voltage on the drain of the MOSFET rises to
the output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the voltage on the drain of the MOSFET falls below VIN. A
boundary mode detection comparator detects this event
and turns the external MOSFET back on.
Boundary mode returns the secondary current to zero every
cycle, so the parasitic resistive voltage drops do not cause
load regulation errors. Boundary mode also allows the use
of a smaller transformer compared to continuous conduc-
tion mode and does not exhibit subharmonic oscillation.
At low output currents the LT3748 delays turning on the
external MOSFET and thus operates in discontinuous mode.
Unlike traditional flyback converters, the external MOSFET
has to turn on to update the output voltage information.
Below 0.6V on the VC pin, the current comparator level
decreases to its minimum value and a variable delay timer
waits to reset before turning on the external MOSFET. With
the addition of delay before turning the MOSFET back
on, the part starts to operate in discontinuous mode. The
average output current is able to decrease while still allow-
ing a minimum off-time for the error amplifier sampling
circuitry. The typical maximum discontinuous off-time
with VC equal to 0V is 24µs.
LT3748
9
3748fb
For more information www.linear.com/LT3748
Pseudo-DC Theory of Operation
The RREF and RFB resistors as depicted in the Block Diagram
are external resistors used to program the output voltage.
The LT3748 operates much the same way as traditional
current mode switchers with the exception of the unique
error amplifier which derives its feedback information
from the flyback pulse.
Operation is as follows: when the NMOS output switch
turns off, its drain voltage rises above VIN. The amplitude
of this flyback pulse (i.e., the difference between it and
VIN) is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = DOUT forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is converted to a current by RFB and
Q2. Nearly all of this current flows through resistor RREF to
form a ground-referred voltage. This voltage is fed into the
flyback error amplifier. The flyback error amplifier samples
this output voltage information when the secondary-side
winding current reaches zero. The error amplifier uses a
bandgap voltage, 1.223V, as the reference voltage.
The relatively high gain in the overall loop will then cause
the voltage at the RREF resistor to be nearly equal to the
bandgap reference voltage, VBG. The relationship between
VFLBK and VBG may then be expressed as:
VFLBK
RFB
=VBG
RREF
or
VFLBK =VBG RFB
RREF
VBG = Internal bandgap reference
Combining with the previous VFLBK expression yields an
expression for VOUT, in terms of the internal reference,
programming resistors, transformer turns ratio and diode
forward voltage drop:
VOUT =VBG RFB
RREF
1
NPS
−VF−ISEC (ESR)
Additionally, it includes the effect of nonzero secondary
output impedance (ESR). This term can be assumed to
be zero in boundary control mode.
Temperature Compensation
The first term in the VOUT equation does not have a tem-
perature dependence, but the diode forward drop, VF
, has a
significant negative temperature coefficient. To compensate
for this, a positive temperature coefficient current source
is internally connected to the RREF pin. The current is set
by resistor RTC to ground connected between the TC pin
and ground. To cancel the temperature coefficient, the
following equation is used:
dV
F
dT= −
R
FB
RTC
•1
NPS
•dVTC
dTor,
RTC =−RFB
NPS
•1
dVF/dT•dVTC
dT≈RFB
NPS
(dVF/dT) = Diode’s forward voltage temperature coef-
ficient
(dVTC/dT) = 1.85mV/°C
VTC = 0.55V
The resistor value given by this equation should also
be verified experimentally and adjusted, if necessary, to
achieve optimal regulation over temperature.
The revised output voltage is as follows:
VOUT =VBG RFB
RREF
1
NPS
−VF
−VTC
RTC
•RFB
NPS
– ISEC (ESR)
applications inForMation
LT3748
10
3748fb
For more information www.linear.com/LT3748
applications inForMation
Selecting Actual RREF
, RFB and RTC Resistor Values
The preceding equations define how the LT3748 would
regulate the output voltage if the system had no time delays
and no error sources. However, there are a number of
repeatable delays and parasitics in each application which
will affect the output voltage and force a re-evaluation of
the RFB and RTC component values. The following approach
is the best method for selecting the correct values.
The expression for VOUT, developed in the Operation sec-
tion, can be rearranged to yield the following expression
for RFB:
RFB =RREF •NPS VOUT +VF
( )
+VTC
VBG
where:
VOUT = Output voltage
VF = Output diode forward voltage
NPS = Effective primary-to-secondary turns ratio
VTC = 0.55V
The equation assumes the temperature coefficients of the
output diode and VTC are equal and substitutes RFB/NPS for
the value of RTC. This is a good first order approximation
but will be revisited later.
First, the value of RREF should be approximately 6.04k
since the LT3748 is trimmed and specified using this
value. If the impedance of RREF varies considerably from
6.04k, additional errors will result. However, a variation in
RREF of several percent is acceptable. This yields a bit of
freedom in selecting standard 1% resistor values to yield
nominal RFB/RREF ratios.
With starting values for RFB and RTC, an initial iteration
of the application should be built with final selections of
all external components (transformer, diode, MOSFET,
etc.). The resulting VOUT should be measured and used
to re-evaluate the value of RFB due to non-idealities in the
sampling system:
RFB(NEW) =
V
OUT(DESIRED)
VOUT(MEASURED)
•RFB(OLD)
With a new value of RFB selected, the temperature co-
efficient of the output diode in the application can be
tested to verify the nominal RTC value. The RTC resistor
should be removed from the circuit under test (this will
cause VOUT to increase for this step) and VOUT should
be measured over temperature at a desired target output
load. It is very important for this evaluation that uniform
temperature be applied to both the output diode and the
LT3748—if freeze spray or a heat gun is used there can
be a significant mismatch in temperature between the
two devices that causes significant error. Attempting to
extrapolate the data from a diode datasheet or assuming
the nominal RTC value may yield a better result if there is
no method to apply uniform heat or cooling such as an
oven. With at least two data points (although more data
points from hot to cold are recommended), the change
in V/°C can be determined by:
∆V
OUT
∆TEMP
=
V
OUT1
– V
OUT2
TEMP1– TEMP2
Using the measured VOUT temperature coefficient, an exact
RTC value can be selected using the following equation:
RTC =
R
FB
NPS
•
1.85mV/°C
∆VOUT
∆TEMP
If the value of RTC has changed significantly, which can
happen with the use of some output diodes that have
a very low forward drop, the RFB value may need to be
changed to restore VOUT to the desired value. As in the
previous iteration, after measuring VOUT
, a new RFB can
once again be selected using:
RFB(NEW) =
V
OUT(DESIRED)
VOUT(MEASURED)
•RFB(OLD)
Once the values of RFB and RTC are selected, the regulation
accuracy from board to board for a given application will be
very consistent, typically under ±5% when including device
variation of all the components in the system (assuming
resistor tolerances and transformer windings matching
of 1% or better). However, if the transformer, the output
diode or MOSFET switch are changed or the layout is
dramatically altered, there may be some change in VOUT
.
LT3748
11
3748fb
For more information www.linear.com/LT3748
applications inForMation
Minimum Primary Inductance Requirements
The LT3748 obtains output voltage information from the
external MOSFET drain voltage when the secondary winding
conducts current. The sampling circuitry needs a minimum
of 400ns to settle and sample the output voltage while the
MOSFET switch is off. This required settle and sample
time is controlled by external components independent of
the minimum off-time of the GATE pin as specified in the
Electrical Characteristics table. The electrical specification
minimum off-time is based on an internal timer and acts
as a maximum frequency clamp. The following equation
gives the minimum value for primary-side magnetizing
inductance:
LPRI ≥VOUT +VF(DIODE)
( )
•RSENSE •tSETTLE(MIN) •NPS
V
SENSE(MIN)
VSENSE(MIN) = 15mV
tSETTLE(MIN) = 400ns
NPS = Ratio of primary windings to secondary windings
In addition to the primary inductance requirement for
minimum settling and sampling time, the LT3748 has
internal circuit constraints that prevent it from setting the
GATE node high for shorter than approximately 250ns.
If the inductor current exceeds the desired current limit
during that time oscillation may occur at the output as
the current control loop will lose its ability to regulate.
Therefore, the following equation relating to maximum
input voltage must also be followed in selecting primary-
side magnetizing inductance:
LPRI ≥
V
IN(MAX)
•R
SENSE
•t
ON(MIN)
VSENSE(MIN)
tON(MIN) = 250ns
The last constraint on minimum inductance value would
relate to minimum full-load operating frequency, fSW(MIN),
and is derived from fSW = 1/(tON + tOFF):
LPRI ≤ VIN(MIN) • ( VOUT + VF(DIODE)) • NPS/(fSW(MIN) • ILIM •
((VOUT + VF(DIODE)) • NPS + VIN(MIN)))
The minimum operating frequency may be lower than
the calculated number due to delays in detecting current
limit and detecting boundary mode that are specific to
each application.
Output Power
Because the MOSFET power switch is located outside the
LT3748, the maximum output power is primarily limited
by external components. Output power limitations can
be separated into three categories—voltage limitations,
current limitations and thermal limitations.
The voltage limitations in a flyback design are primar-
ily the MOSFET switch VDS(MAX) and the output diode
reverse-bias rating. Increasing the voltage rating of either
component will typically decrease application efficiency if
all else is equal and the voltage requirements on each of
those components will be directly related to the windings
ratio of the transformer, the input and output voltages
and the use of any additional snubbing components.
The MOSFET VDS(MAX) must theoretically be higher than
VIN(MAX) + (VOUT • NPS) and the output diode reverse bias
must be higher than VOUT + (VIN(MAX)/NPS), though leak-
age inductance spikes on both the drain of the MOSFET
and the anode of the output diode may more than double
that requirement (see section on leakage inductance for
more details on snubbers). Figure 1 illustrates the effect
on available output power for several MOSFET voltage
ratings while continuously maximizing windings ratio
for input voltage with a fixed MOSFET current limit and
output voltage. Increasing the MOSFET rating increases
the possible windings ratio and or maximum input voltage
and can increase the available output power for a given
application. Both figures assume no leakage inductance
and high efficiency.
INPUT VOLTAGE (V)
0
MAXIMUM OUTPUT POWER (W)
30
40
50
80
3748 F01
20
10
020 40 60 100
VDS = 200V
VDS = 150V
VDS = 100V
Figure 1. Maximum Output Power at 12VOUT with a
3A ILIM and Maximum VDS = 100V, 150V, 200V
LT3748
12
3748fb
For more information www.linear.com/LT3748
The current limitation on output power delivery is gen-
erally constrained by transformer saturation current in
higher power applications, although the MOSFET switch
and output diode will need to be rated for the desired
currents, as well. Increasing the peak current on the pri-
mary side of the flyback by reducing the RSENSE resistor
is the primary way to increase output power, and power
delivered increases fairly linearly with current limit as
shown in Figure 2, until parasitic losses begin to dominate.
However, once the saturation current of the transformer
is exceeded the energy coupling between the primary and
the secondary will be reduced and incremental power will
not be delivered to the output. In addition, the primary
inductance will drop, the SENSE pin overcurrent threshold
may trip due to a corresponding rapid rise in current, and
the transformer will have to absorb the energy that is not
transferred through the saturated core, leading to heating.
Some manufacturers may not specify the rated saturation
current but it is a necessary specification when trying to
minimize transformer size and maximize output power
and efficiency. Also necessary for proper design is data
on saturation current over temperature —the saturation
of typical power ferrites may reduce by over 20% from
25°C to 100°C.
The thermal limitation in flyback applications for lower
output voltages will be dominated by losses in the output
diode, with resistive and leakage losses in the transformer
increasing as a percentage basis of loss as the output
voltage is increased. As power levels increase the output
diode and transformer may exceed their rated temperature
specifications. Minimizing RMS output diode current,
selecting a diode with minimal forward drop at expected
currents and minimizing parasitic resistances and leakage
inductance in the transformer will keep those components
below their maximum temperatures while maximizing
efficiency. The following section discussing transformer
selection will further help focus on how to minimize losses
in the output diode.
While quiescent current in the LT3748 itself is low (ap-
proximately 300µA from VIN and 1mA from INTVCC), the
current required to drive the external MOSFET (fSW • QG),
if drawn from VIN through the LT3748 INTVCC LDO, dis-
sipates (VIN – INTVCC) • fSW • QG. If that power is high
enough to cause significant heating of the LT3748 the
current may need to be drawn from a third winding. Doing
so will push all thermal limitations outside of the LT3748.
Selecting a Transformer
Transformer specification and design is perhaps the most
critical part of successfully applying the LT3748. In addi-
tion to the usual list of caveats dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
First and most importantly, since the voltage on the sec-
ondary side of the transformer is inferred by the voltage
sampled on the primary, the transformer turns ratio must
be tightly controlled to ensure a consistent output volt-
age. A tolerance of ±5% in turns ratio from transformer
to transformer could result in a variation of more than
±5% in output regulation. Fortunately, most magnetic
component manufacturers are capable of guaranteeing a
turns ratio tolerance of 1% or better.
Linear Technology has worked with several leading mag-
netic component manufacturers to produce predesigned
flyback transformers for use with the LT3748. Table 1
shows the details of several of these transformers.
applications inForMation
INPUT VOLTAGE (V)
0
MAXIMUM OUTPUT POWER (W)
30
40
50
80
3748 F02
20
10
020 40 60 100
ILIM = 3A
ILIM = 2A
ILIM = 1A
Figure 2. Maximum Output Power at 12VOUT
with 150V VDS(MAX) and ILIM = 1A, 2A, 3A
LT3748
13
3748fb
For more information www.linear.com/LT3748
applications inForMation
Table 1. Pre-Designed Transformers—Typical Specifications Unless Otherwise Noted
TRANSFORMER
PART NUMBER Size (W x L x H) mm
LPRI
(µH)
LLEAK
(nH)
NPS
(NP:NS)
ISAT
(A)
RPRI
(mΩ)
RSEC
(mΩ) MANUFACTURER
TARGET APPLICATION†
INPUT (V) OUTPUT
750311424 17.7 × 14.0 × 12.7 100 844 3:1 3 180 29 Würth Electronics 40 to 75 12V/1A
750311456* 17.7 × 14.0 × 12.7 100 900 3:1 2.4 225 31 Würth Electronics 40 to 75 12V/1A
750311439 17.7 × 14.0 × 12.7 37 750 2:1 2.8 89 28 Würth Electronics 30 to 75 12V/1A
750311423 17.7 × 14.0 × 12.7 50 570 4:1 4 90 12 Würth Electronics 30 to 75 5V/3A
750311457 17.7 × 14.0 × 12.7 50 600 4:1 3.7 115 12 Würth Electronics 30 to 75 5V/3A
750311689 17.7 × 14.0 × 12.7 50 600 4:1 3.7 115 12 Würth Electronics 30 to 75 5V/3A
750311458* 17.7 × 14.0 × 12.7 15 175 3:1 5 35 6 Würth Electronics 10 to 40 5V/2.5A
750311564 17.7 × 14.0 × 12.7 9 120 3:1 8 36 7 Würth Electronics 10 to 40 5V/3A
750311624 17.7 × 14.0 × 12.7 9 150 1.5:1 8 34 21 Würth Electronics 10 to 40 15V/1A
750311604 29.08 × 23.11 × 11.43 8 300 1:1 9.5 30 12 Würth Electronics 10 to 40 24V/1.3A
750311599 29.08 × 23.11 × 11.43 8 500 1.5:1 12 30 12 Würth Electronics 10 to 40 15V/2A
750311600 29.08 × 23.11 × 11.43 12 500 3:1 11 30 40 Würth Electronics 20 to 75 15V/2A
750311608 29.08 × 23.11 × 11.43 12 500 1.5:1 9 30 20 Würth Electronics 20 to 75 24V/1.3A
750311607 29.08 × 23.11 × 11.43 14 500 2.5:1 9.5 40 10 Würth Electronics 20 to 75 12V/2.5A
750311590 32.31 × 27.03 × 13.69 8 200 2:1 18 15 8 Würth Electronics 10 to 40 12V/3.8A
750311591 32.31 × 27.03 × 13.69 8 200 1.5:1 20 15 12 Würth Electronics 10 to 40 15V/3A
750311592 32.31 × 27.03 × 13.69 8 200 1:1 18 15 20 Würth Electronics 10 to 40 24V/1.9A
750311594 32.31 × 27.03 × 13.69 15 400 2.33:1 18 35 15 Würth Electronics 20 to 75 12V/3.8A
750311595 32.31 × 27.03 × 13.69 12 200 3:1 18 15 12 Würth Electronics 20 to 70 15V/3A
750311596 32.31 × 27.03 × 13.69 12 200 1.5:1 16 30 30 Würth Electronics 20 to 70 24V/1.9A
PA2367NL 17.7 × 14.0 × 12.7 85 750 2.7:1 1.7 325 26 Pulse Engineering 20 to 75 12V/1A
PA1276NL 17.7 × 14.0 × 12.7 77.4 800 1.47:1 1.6 100 75 Pulse Engineering 20 to 75 12V/1A
PA2467NL 17.7 × 14.0 × 12.7 37 750 2:1 2.9 89 28 Pulse Engineering 20 to 75 12V/1A
PA1260NL 17.7 × 14.0 × 12.7 77.4 800 3.67:1 1.5 220 18 Pulse Engineering 20 to 75 5V/2A
PA3177NL 29.21 × 21.84 × 11.43 8.3 100 2:1 8.6 10 7 Pulse Engineering 10 to 40 10V/2.5A
*2.5k isolation, others are rated for 1.5kV isolation.
†TARGET APPLICATION, NOT GUARANTEED.
Turns Ratio and RMS Diode Current
Note that when using an RFB/RREF resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, simpler ratios of small integers (e.g., 1:1, 2:1,
3:2, etc.) can be employed to provide more freedom in
setting total turns and mutual inductance.
While the turns ratio can be selected to maximize output
power for a given current limit, minimizing the turns
ratio and increasing the current limit will often increase
efficiency and better utilize the saturation current of a
given transformer. Figure 3 shows the maximum output
power using three transformers with different windings
ratios that have the same output inductance and peak
output current, illustrating that increasing current while
decreasing turns ratio can deliver more power.
There are two significant constraints on the turns ratio.
First, as described in the previous section on limitations
to output power, the drain of the MOSFET switch will
see a voltage equal to the maximum input supply plus
LT3748
14
3748fb
For more information www.linear.com/LT3748
applications inForMation
the output voltage multiplied by the windings ratio plus
some amount of overshoot caused by leakage inductance.
Second, increasing the turns ratio will increase the peak
current seen on the output diode generally increasing the
RMS diode current thereby lowering the efficiency. This
efficiency limitation is worse at lower output voltages when
the diode forward voltage is significant compared to the
output voltage. In a typical application such as the 5V, 2A
output shown on the back page, the diode losses dominate
all the other losses, as shown in Figure 4. To calculate
RMS diode current, two equations are needed—the first
for calculating duty cycle, D, and the second to calculate
the RMS current of a triangle waveform:
D=VOUT +VF(DODE)
( )
•NPS
VIN +VOUT +VF(DIODE)
( )
•NPS
IDIODE(RMS) =ILIM •NPS
( )
2•1– D
( )
3
For a more general analysis, Figure 5 illustrates a sweep
of windings ratio on the x-axis while comparing output
power and estimated efficiency for a 5V output using a
48V input. If the desired application required 20W, the
maximum power curve indicates that a winding ratio of
12:1 would be sufficient at a current limit of 2A (RSENSE =
0.05Ω), while a winding ratio of 5:1 would deliver the same
power at 3A. However, when examining the corresponding
efficiency at max load for those two windings ratios and
current limits, the 5:1, 3A selection is clearly the superior
solution with an estimated efficiency of 85% compared to
78% for the 12:1, 2A application.
There are several caveats to this evaluation. First, as the
diode forward voltage becomes a smaller percentage of
total loss at higher output voltages (>12V) the RMS current
becomes less of a concern and minimizing it will have a
much smaller impact on efficiency. More significantly, if
a lower turns ratio forces the use of a diode with a larger
forward drop to obtain a higher reverse voltage rating,
any gains from minimizing current might be lost. For low
output voltages (3.3V or 5V) or high input voltages (>48V),
a turns ratio greater than one can be used with multiple
primary windings relative to the secondary to maximize
the transformer’s current gain.
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
15
20
25
80
3748 F03
10
5
020 40 60 100
NPS = 2:1
ILIM = 3A
NPS = 3:1
ILIM = 2A
NPS = 6:1
ILIM = 1A
IOUT (A)
0.2A MIN
70
EFFICIENCY LOSS (%)
75
80
85
90
100
2A MAX
3748 F03
95
VIN = 12V
DOUT
fSW • QG + IQ
TRANSFORMER I • R + LEAKAGE
FET RDS(ON)
NPS
0
ESTIMATED MAX LOAD EFFICIENCY (%)
MAXIMUM OUTPUT POWER (W)
75
80
85
915
3748 F05
70
65
60 3 6 12
90
95
100
12
16
20
8
4
0
24
28
32
18
ILIM = 3A
ILIM = 2A
EFFICIENCY
OUTPUT
POWER
Figure 3. Maximum Output Power at 12V Out Using Three
Transformers with Equal Peak Output Current and Secondary
Inductance
Figure 4. Sources of Loss In 5V, 2A Out Typical Application
Figure 5. Estimated Efficiency and Output Power at 5VOUT from
48VIN vs Windings Ratio, NPS, at 2A and 3A Current Limits
LT3748
15
3748fb
For more information www.linear.com/LT3748
applications inForMation
Saturation Current
As discussed earlier in the Maximum Output Power sec-
tion, because the core of the transformer is being used for
energy storage in a flyback, the current in the transformer
windings should not exceed their rated saturation current
as energy injected once the core is saturated will not be
transferred to the secondary and will instead be dissipated
in the core. Information on saturation current should be
provided by the transformer manufacturers and Table 1
lists the saturation current of the transformers designed
for use with the LT3748.
Leakage Inductance and Snubbers
Transformer leakage inductance (on either the primary
or secondary) causes a voltage spike to appear at the
primary after the MOSFET switch turns off. This spike is
increasingly prominent at higher load currents where more
stored energy must be dissipated. Transformer leakage
inductance should be minimized.
In most cases, proper selection of the external MOSFET
and a well designed transformer will eliminate the need for
snubber circuitry, but in some cases the optimal MOSFET
may require protection from this leakage spike. An RC
(resistor capacitor) snubber may be sufficient in applica-
tions where the MOSFET has significant margin beyond
the predicted DC drain voltage applied in flyback while a
clamp using an RCD (resistor capacitor diode) or a Zener
might be a better option when using a MOSFET with very
little margin for leakage inductance spiking.
The recommended approach for designing an RC snubber
is to measure the period of the ringing at the MOSFET
drain when the MOSFET turns off without the snubber
and then add capacitance—starting with something in
the range of 100pF—until the period of the ringing is 1.5
to 2 times longer. The change in period will determine
the value of the parasitic capacitance, from which the
parasitic inductance can be determined from the initial
period, as well. Similarly, initial values can be estimating
using stated switch capacitance and transformer leakage
inductance. Once the value of the drain node capacitance
and inductance is known, a series resistor can be added to
the snubber capacitance to dissipate power and critically
dampen the ringing. The equation for deriving the optimal
series resistance using the observed periods (tPERIOD, and
tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is
below, and the resultant waveforms are shown in Figure 6.
CPAR =
C
SNUBBER
tPERIOD(SNUBBED)
tPERIOD
2
– 1
LPAR =tPERIOD2
CPAR •4π2
RSNUBBER =LPAR
CPAR
TIME (µs)
0
0
V
DRAIN
(V)
10
30
40
50
0.20
90
3748 F06
20
0.10
0.05 0.25
0.15 0.30
60
70
80
NO SNUBBER
WITH SNUBBER
CAPACITOR
WITH RESISTOR
AND CAPACITOR
Figure 6. Observed Waveforms at MOSFET Drain when
Iteratively Implementing an RC Snubber
Note that energy absorbed by a snubber will be converted
to heat and will not be delivered to the load. In high volt-
age or high current applications, the snubber may need to
be sized for thermal dissipation. To determine the power
dissipated in the snubber resistor from capacitive losses,
measure the drain voltage immediately before the MOSFET
turns on and use the following equation relating that volt-
age and the MOSFET switching frequency to determine
the expected power dissipation:
PSNUBBER = fSW • CSNUBBER • VDRAIN2/2
Decreasing the value of the capacitor will reduce the dis-
sipated power in the snubber at the expense of increased
peak voltage on the MOSFET drain, while increasing the
value of the capacitance will decrease the overshoot.
LT3748
16
3748fb
For more information www.linear.com/LT3748
applications inForMation
ring beyond that expected reverse voltage. An RC snubber
or RCD clamp may be implemented to reduce the voltage
spike if it is desirable to use a lower reverse voltage diode.
Secondary Leakage Inductance
In addition to the previously described effects of leakage
inductance in general, leakage inductance on the sec-
ondary in particular exhibits an additional phenomena. It
forms an inductive divider on the transformer secondary
that effectively reduces the size of the primary-referred
flyback pulse used for feedback. This will increase the
output voltage target by a similar percentage. Note that,
unlike leakage spike behavior, this phenomena is load
independent. To the extent that the secondary leakage
inductance is a constant percentage of mutual inductance
Although it typically does not decrease efficiency, leakage
inductance energy that would normally have been dis-
sipated in the switch or transformer is also dissipated in
the RC snubber resistor and can be calculated as:
PSNUBBER = fSW • LLEAK • ILIM2/2
An RCD clamp, shown in Figure 7, also prevents the
leakage inductance spike from exceeding the breakdown
voltage of the MOSFET switch. In most applications, there
will be a very fast voltage spike caused by a slow clamp
diode. Once the diode clamps, the leakage inductance
current is absorbed by the clamp capacitor. This period
should not last longer than 200ns so as not to interfere
with the output regulation. The clamp diode turns off after
the leakage inductance energy is absorbed and the switch
voltage is then equal to:
VDS = VIN + NPS • (VOUT + VF(DIODE))
Schottky diodes are typically the best choice for use in a
snubber, but some PN diodes can be used if they turn on
fast enough to limit the leakage inductance spike. Figures 8
and 9 show the waveform at the drain of the MOSFET
switch for the 48V output application shown in Figure 17
at maximum rated load and maximum input voltage with
an RC snubber and RCD clamp, respectively. Both solu-
tions limit the leakage spike to less than 190V, below the
200V VDS(MAX) rating of the Si7464DP MOSFET.
Figure 7. RCD Clamp
Figure 8. Waveform of MOSFET Drain During Normal Operation
of Figure 19 with RC Snubber (as Drawn)
3748 F07
LLEAK
VIN VOUT+
VOUT–
GATE NMOS
D
R
C
+
Figure 9. Waveform of MOSFET Drain During Normal Operation
of Figure 19 Using RCD Clamp with Central Semiconductor
CMR1U-02M-LT C Instead of RC Snubber
TIME (ns)
0
0
DRAIN VOLTAGE (V)
40
80
120
50 100 150 200
3748 F08
250
160
200
20
60
100
140
180
300
VIN = 96V
VOUT = 48V
IOUT = 0.5A
R = 66Ω
C = 150pF
TIME (ns)
0
0
DRAIN VOLTAGE (V)
40
80
120
50 100 150 200
3748 F08
250
160
200
20
60
100
140
180
300
VIN = 96V
VOUT = 48V
IOUT = 0.5A
R = 4.99k
C = TDK 0.22µF 250V
D = CMR1U-02M-LTC
Leakage Inductance and Output Diode Stress
The output diode may also see increased reverse voltage
stresses from leakage inductance. While it nominally sees
a reverse voltage of the input voltage divided by the wind-
ings ratio plus the output voltage when the MOSFET power
switch turns on, the capacitance on the output diode and
the leakage inductance will cause an LC tank which may
LT3748
17
3748fb
For more information www.linear.com/LT3748
applications inForMation
(over manufacturing variations), this can be accommodated
by adjusting the RFB/RREF resistor ratio.
Winding Resistance Effects
Resistance in either the primary or secondary will reduce
overall efficiency (POUT/PIN). Good output voltage regula-
tion will be maintained independent of winding resistance
due to the boundary mode operation of the LT3748.
Bifilar Winding
A bifilar, or similar winding technique, is a good way to
minimize troublesome leakage inductances. However, re-
member that this will also increase primary-to-secondary
capacitance and limit the primary-to-secondary breakdown
voltage, so, bifilar winding is not always practical. The
Linear Technology Applications group is available and
extremely qualified to assist in the selection and/or design
of the transformer.
Selecting a Current Sense Resistor
The external current sense resistor allows the user to
optimize the current limit behavior for the particular ap-
plication under consideration. As the current sense resistor
is varied from several ohms down to tens of milliohms,
peak switch current goes from a fraction of an ampere
to tens of amperes. Care must be taken to ensure proper
circuit operation, especially with small current sense
resistor values.
For example, a peak MOSFET switch current of 4A requires
a sense resistor of 0.025Ω. Note that the instantaneous
peak power in the sense resistor is 1W, and it must be
rated accordingly. The LT3748 has only a single sense line
to this resistor. Therefore, any parasitic resistance in the
ground side connection of the sense resistor will increase
its apparent value. In the case of a 0.025Ω sense resistor,
1mΩ of parasitic resistance will cause a 4% reduction in
peak switch current. Therefore, resistance of printed circuit
copper traces and vias cannot necessarily be ignored.
Another issue for proper operation of the current sense
circuitry is avoiding prematurely tripping the SENSE
threshold while slewing the MOSFET drain when the GATE
pin goes high. The LT3748 does not begin to compare
the SENSE pin voltage with the target threshold until the
GATE pin is near its final value, or until at least 150ns
has passed, whichever occurs more slowly. This should
be entirely sufficient for most applications but premature
tripping of the SENSE comparator may occur in cases
where a MOSFET with very high QG is used with a series
resistor at the GATE pin.
Output Short Circuits and SENSE Pin Over Current
The LT3748 has an internal threshold to detect when
primary inductor current exceeds the programmed range.
This can result from an inductive output short-circuit and
an output voltage below zero, reflecting a voltage back to
the primary side of the transformer which, in turn, causes
the LT3748 to turn the external MOSFET on before the
secondary current has discharged. When the voltage at
the SENSE pin exceeds approximately 130mV—equiva-
lent to 30% higher than the programmed ILIM(MAX) in
the RSENSE resistor—the SS pin will be reset, stopping
switching. Once the soft-start capacitor is recharged and
the soft-start threshold is reached, switching will resume
at the minimum current limit.
High Drain Capacitance and Low Current Operation
When designing applications with some combination of a
low current limit (ILIM < 1A), a high secondary-to-primary
turns ratio (NPS << 1), multiple output windings, or very
capacitive output diodes, it is important to minimize the
capacitance reflected onto the primary winding and on the
drain of the external MOSFET. After the MOSFET turns off
during each switching cycle, the primary current charges
that capacitance to slew the MOSFET drain until the second-
ary begins to deliver power, and if the drain node does not
slew and remain above VIN within approximately 200ns
once the GATE pin goes low and the MOSFET turns off,
the LT3748 may detect that the current in the secondary
is zero and turn the MOSFET back on prematurely, caus-
ing the LT3748 to switch continuously while delivering
very little power to the output. The result will be droop of
the output voltage at lighter loads and oscillation at the
VC node. This problem can be prevented by maximizing
NPS (minimizing ratio of secondary windings to primary
windings), increasing the peak drain current (minimizing
RSENSE), and minimizing the output diode and transformer
capacitance.
LT3748
18
3748fb
For more information www.linear.com/LT3748
applications inForMation
Soft-Start
The LT3748 contains an optional soft-start function that
is enabled by connecting an explicit external capacitor
between the SS pin and ground. Internal circuitry prevents
the control voltage at the VC pin from exceeding that on
the SS pin.
The soft-start function is engaged whenever power at VIN
is removed, or as a result of either undervoltage lockout,
overcurrent in the sense resistor or thermal (overtempera-
ture) shutdown. The SS node is then discharged to roughly
600mV. When this condition is removed, a nominal 5µA
current acts to charge up the SS node towards roughly
2.2V. For example, a 0.1µF soft-start capacitor will place
a 0.05V/ms limit on the turn-on ramp rate at the VC node.
ENABLE and Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin implements
undervoltage lockout (UVLO). The EN/UVLO pin threshold
is set at 1.223V. In addition, the EN/UVLO pin draws 2.4µA
when the voltage at the pin is below 1.223V. This current
provides user programmable hysteresis based on the value
of R1. The effective UVLO thresholds are:
VIN(UVLO,RISING) =
1.223V •(R1+R2)
R2 +2.4µA •R1
VIN(UVLO,FALLING) =1.223V •(R1+R2)
R2
Figure 10 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT3748 in shutdown with a quiescent current draw of
less than 1µA.
Minimum Load Requirement
The LT3748 recovers output voltage information using
the flyback pulse that occurs once the external MOSFET
turns off and the secondary winding conducts current. In
order to regulate the output voltage, the LT3748 needs to
sample the flyback pulse. The LT3748 delivers a minimum
amount of energy even during light load conditions to
ensure accurate output voltage information. The minimum
delivery of energy creates a minimum load requirement
on the output of approximately 2% of maximum load.
The minimum operating frequency at minimum load is
approximately 42kHz.
Alternatively, a Zener diode sufficiently rated to handle the
minimum load power can be used to provide a minimum
load without decreasing efficiency in normal operation.
In selecting a Zener diode for this purpose, the Zener
voltage should be high enough that the diode does not
become the load path during transient conditions but the
voltage must still be low enough that the MOSFET and
output voltage ratings are not exceeded when the Zener
functions as the minimum load.
INTVCC Pin Considerations
The INTVCC pin powers the internal circuitry and gate
driver of the LT3748. Three unique configurations exist
for regulation of the INTVCC pin as shown in Figure 11. In
the first configuration, the internal LDO drives the INTVCC
pin internally from the VIN supply. In the second configura-
tion, the VIN supply directly drives the INTVCC pin through
a direct connection bypassing the internal LDO. Use this
optional configuration for voltages lower than 20V. In the
third configuration, an external supply or third winding
drives the INTVCC pin. Use this option when a voltage
supply exists lower than the input supply but higher than
the regulated INTVCC voltage. Using a lower voltage sup-
ply provides a more efficient source of power for internal
circuitry and reduces power dissipation in the LT3748.
When calculating the minimum input voltage required for
a valid INTVCC, or the power dissipated in the LT3748, it is
useful to know how much current will be drawn from the
INTVCC LDO during normal operation. The easiest way to
calculate this current is to use the gate charge (QG) for the
selected MOSFET switch at the expected VIN and INTVCC
LT3748
EN/UVLO
GND
R2
R1
V
IN
3748 F10
RUN/STOP
CONTROL
(OPTIONAL)
Figure 10. Undervoltage Lockout (UVLO)
LT3748
19
3748fb
For more information www.linear.com/LT3748
applications inForMation
voltages and multiply that charge required with each
turn-on event by the maximum operating frequency. The
maximum operating frequency in a given application can
be approximated from the primary transformer inductance,
the windings ratio (NPS), the nominal output voltage and
the maximum input voltage. Unless the part is limited by
minimum on- or off-times, this maximum frequency will
occur when the part is regulating in boundary mode at the
minimum peak switch current, and can be derived from:
fSW(MAX) ≈VIN(MAX) •VOUT +VF(DIODE)
( )
•NPS
LPRI •ILIM(MIN) •VOUT +VF(DIODE)
( )
•NPS +VIN(MAX)
( )
With the maximum INTVCC current calculated, the expected
dropout when VIN drops below 7V can be extracted from
the curves in the Typical Performance Characteristics
section. The LT3748 is tested as low as VIN = 5V but
the hard limit on minimum VIN operation is the INTVCC
regulator dropout and the 3.6V under voltage lockout.
Figure 12 illustrates an example where operation with VIN
= 5V and IINTVCC = 20mA might be fully functional at room
temperature, but when the dropout for the same current
exceeds 1.4V and trips the UVLO at higher temperatures
the LT3748 will stop switching.
Overdriving INTVCC with a Third Winding
The LT3748 provides excellent output voltage regulation
without the need for an opto-coupler or third winding, but for
some applications with input voltages greater than 20V, an
additional winding may improve overall system efficiency.
The third winding should be designed to output a voltage be-
tween 7.2V and 20V. A resistor in series with the rectifier is
recommended to absorb leakage spikes. For a typical 48VIN,
10W application, overdriving the INTVCC pin may improve
efficiency by several percent at maximum load and as
much as 30% at light loads.
Loop Compensation
The LT3748 is compensated using an external resistor-
capacitor network on the VC pin. Typical values are in the
range of RC = 50k and CC = 1nF (see the numerous sche-
matics in the Typical Applications section for other possible
values). If too large of an RC value is used, the part will be
more susceptible to high frequency noise and jitter. If too
small of an RC value is used, the transient performance will
suffer. The value choice for CC is somewhat the inverse
of the RC choice: if too small a CC value is used, the loop
may be unstable and if too large a CC value is used, the
transient performance will also suffer. Transient response
plays an important role for any DC/DC converter.
LT3748
3.6V < BIAS < 20V,
VIN > BIAS
5V TO 100V
INTVCC
VIN
3748 F09
EXTERNAL SUPPLY
OR THIRD WINDING
LDO
LT3748
(VIN – DROPOUT) TO 7V
5V TO 100V
INTVCC
VIN
LDO
LT3748 5V TO 20V
INTVCC
VIN
OPTIONAL
LDO
Figure 11. INTVCC Pin Configurations
TEMPERATURE (°C)
–50
0
INTV
CC
DROPOUT (V)
0.5
1.0
1.5
2.0
0 50 100 150
3748 F12
2.5
3.0
–25 25 75 125
VIN = 5V
INTVCC UVLO = 3.6V
IINTVCC = 20mA
Figure 12. INTVCC Current at Low VIN Can Cause the LT3748
to Stop Switching Due to INTVCC Undervoltage Lockout
LT3748
20
3748fb
For more information www.linear.com/LT3748
applications inForMation
Synchronous Secondary Applications
Using a synchronous secondary controller such as the
LT8309 with the LT3748 is an excellent method to boost
converter efficiency and minimize heat, especially for lower
output voltages and higher output currents. However,
there are some important details to understand when
designing a synchronous application. First, although the
LT8309 controls a synchronous MOSFET in place of the
standard output rectifier, when properly configured that
synchronous MOSFET must turn off before the end of
the secondary conduction time. This ensures that there
is no reverse current sending power back to the primary
side of the transformer and no cross conduction once the
LT3748 GATE pin goes high on the next switching cycle.
As a result, the forward voltage drop of the secondary
MOSFET body diode is reflected back to the primary side
and sampled by the LT3748. In order to guarantee an
accurate sample and to maintain excellent line and load
regulation, the RDRAIN resistor of the LT8309 must be
optimized to allow the body diode to conduct long enough
to provide an accurate reflected voltage. To ensure ac-
curate output regulation the secondary MOSFET should
turn off at least 180ns before the secondary current goes
to zero. Figure 13 illustrates the expected waveform at
the primary side drain node and the LT8309 GATE pin us-
ing the circuit from Figure21 with sufficient body diode
conduction time marked.
Because the body diode is conducting at the sampling
point for the LT3748 when the secondary current goes
to zero, the temperature coefficient of this body diode
should be compensated using the TC pin using the same
procedures outlined when a normal rectifier is used on the
secondary. The silicon junction of the body diode has a
negative temperature coefficient comparable to a standard
or Schottky diode and standard values specified earlier in
the applications section should be a good starting point.
TIME (µs)
0
VOLTAGE (V)
100
80
60
40
20
053 7
LT3748 F13
842 61
BODY DIODE
CONDUCTION
PRIMARY SIDE
DRAIN VOLTAGE
LT8309
VGATE
Figure 13. Waveforms at LT3748 Primary Side MOSFET
Drain and LT8309 GATE Pin During Operation Illustrating
Optimum Body Diode Conduction Time
LT3748
21
3748fb
For more information www.linear.com/LT3748
DESIGN EXAMPLE: 12VIN TO 5V, 2A OUT
The first example is an automotive application shown on
the back page of this data sheet—a nominal 12VIN, 5VOUT
at 2A with an operating input voltage range of 6V to 45V
with a design focus of maximizing efficiency.
1. Select Transformer Turns Ratio
Transformer turns ratio will affect the requirements for the
MOSFET switch VDS rating, the output diode reverse bias
rating, the output power capability, and the efficiency of
the overall converter. Because the output voltage is low
compared to the forward drop on the output diode and
the currents are high in this application, efficiency can be
optimized by minimizing the RMS diode current. Typical
efficiency in a variety of applications will be 85% to 90%
and due to compromises made for the wide input voltage
range and the low output voltage in this specific applica-
tion, an efficiency of 85% is assumed for calculating
output power. This assumption can be revised once the
application is tested. Equations for evaluating each of the
important criteria are:
NPS = NP/NS
VDS(MAX) ≥ VIN(MAX) + VOUT • NPS
VR(DIODE) ≥ VIN(MAX)/NPS + VOUT
IOUT(MAX) ≈ 0.85 • (1 – D) • NPS • ILIM/2
D = (VOUT + VF(DIODE)) • NPS/(VIN + (VOUT + VF(DIODE))
• NPS)
IDIODE(RMS) = √(ILIM • NPS)2 • (1 – D)/3
The equation for output power can be rearranged to solve
for the current limit, ILIM, which can be solved at the
nominal or the minimum VIN depending on application
requirements. In this application the 2A load requirement
will be set at VIN = 7.5V to reduce operating stresses at
higher input voltages. The results of the aforementioned
equations in this application are found in Table 2.
applications inForMation
Evaluating the results of the table, the 1:2 turns ratio looks
demanding in terms of diode reverse-voltage requirements
(a diode with higher reverse bias capability generally will
have a larger forward drop and therefore lower application
efficiency) and primary side currents and only decreases
the output diode RMS current by 13% from the 1:1 case.
However, on evaluating the minimum and maximum in-
ductance requirements in Step 3, even the 1:1 case does
not allow for enough on-time from maximum VIN for the
range of inductance that provides sufficient off-time.
For that reason, a 2:1 turns ratio is selected, easing the
requirement on the output diode reverse voltage rating
in the process.
2. Calculate Sense Resistor Value
The sense resistor can be calculated by the following
equation:
RSENSE =
100mV
I
LIM
The desired 5.8A current limit leads to an unusual value of
0.0172Ω, so the current limit is increased to use a more
standard 0.016Ω value and ILIM of 6.25A.
3. Select a Transformer Based on Inductance and
Saturation Current Requirements
The transformer in this application will be selected to
optimize efficiency at a 80kHz minimum switching fre-
quency at maximum load from the nominal input voltage.
In applications where transformer size is the primary
requirement, reducing the current limit or increasing the
switching frequency may be required. The following equa-
tions select the inductance required for a given switching
frequency at max load and then verify that the inductance
is large enough to satisfy the minimum on and minimum
sampling times of the LT3748.
Table 2. Voltage Stresses, Output Capability and Diode Current vs Turns Ratio in 12VIN to 5V, 2A Application
NPS VDS(MAX) VR(DIODE) D (VIN = 12V) D (VIN = 7.5V) ILIM (2A OUT AT VIN = 7.5V) IDIODE(RMS) (VIN = 12V)
0.5 47.5 95 0.19 0.27 12.9 3.3
1 50 50 0.31 0.42 8.2 3.9
2 55 27.5 0.48 0.59 5.8 4.8
3 60 20 0.58 0.69 5.0 5.6
LT3748
22
3748fb
For more information www.linear.com/LT3748
LPRI ≤ VIN(MIN) • ( VOUT + VF(DIODE)) • NPS/(fSW(MIN) • ILIM •
(VOUT + VF(DIODE)) • NPS + VIN(MIN)))
LPRI ≥ (VOUT + VF(DIODE)) • RSENSE • 400ns • NPS/15mV
LPRI ≥ VIN(MAX) • RSENSE • 200ns/15mV
For this application, the primary inductance with a 2:1
transformer and a 0.016Ω sense resistor for an 6.25A
current limit is bounded by the minimum desired switch-
ing frequency and the minimum off time requirement to
be between 9.6µH and 11.5µH. Looking at Table 1, there
are no transformers that fit that exact requirement. For the
sake of prototyping, a transformer with slightly less than the
desired primary inductance is selected with the PA3177NL.
The application will need to be tested thoroughly for sta-
bility at higher input voltages and when the current limit
is at a minimum (in the middle of the output load range).
The easiest solution to ease the requirement on minimum
on-time is to reduce the maximum VIN voltage although
alternatively NPS could be increased at the expense of ef-
ficiency (and requiring a more thorough redesign).
4. Select a MOSFET Switch
The selected 2:1 transformer requires a nominal 55V rating
on the MOSFET switch, assuming no leakage inductance.
However, even a small amount of leakage inductance may
cause the drain to ring to double the anticipated voltage,
and generally this needs to be verified in the final design.
However, at currents below 10A it is fairly easy to find a
MOSFET with sufficiently low RDS(ON) to be a very small
contributor to maximum load efficiency losses while
similarly having a low enough QG to require minimum
current and minimal losses when driving the MOSFET at
lighter loads. Also, while considering the efficiency gains
and losses with a given MOSFET, it is important to real-
ize that a trade-off in RDS(ON) for VDS(MAX) may backfire
if a snubber needs to be added to the circuit to meet the
voltage requirements and dissipates more energy than the
difference in switch resistance. For that reason, a Vishay
Si7738 is selected to give lots of margin with its 150V
rating. The RMS current in the MOSFET can be calculated,
squared and multiplied by the RDS(ON) to calculate losses
and the current required to drive the FET at frequency can
be determined, by the following equations:
IMOSFET(RMS) = √ILIM2 • D/3
IINTVCC = fSW • QG
PINTVCC = IINTVCC • (VIN – VINTVCC)
In this application the MOSFET RMS current at maximum
load is about 2.7A, which into the 0.038Ω RDS(ON) will be
0.28W, or on the order of 2% loss in efficiency. Assum-
ing that the maximum operating frequency is around four
times higher than the maximum load frequency (at about a
quarter the output load) and reading the approximate QG at
7V operation from the Vishay data sheet, the approximate
INTVCC current is likely close to 8mA, dissipating 0.04W
when the load is on the order of 2.5W, or less than 2%,
and much less at maximum load.
5. Select the Output Diode
The output diode reverse voltage, as calculated earlier, is
the first important specification for the output diode. As with
the MOSFET, choosing a diode with enough margin should
preclude the use of a snubber. The second criterion is the
power requirement of the diode which is more difficult to
correctly ascertain—some manufacturers give direct data
about power dissipation versus duty cycle, which can be
used with the data from the table to determine. To avoid
using a snubber, a diode with a 60V reverse-bias capabil-
ity and minimal forward drop was selected—in this case,
the Diodes Inc. SBR 8U60P5. In this particular application
where maximizing efficiency is the goal, minimizing the
maximum voltage requirement on VIN may allow the use
of a diode with a lower reverse bias rating and a lower
forward drop which could further increase efficiency. Al-
ternatively, if no efficient diode is available for a particular
reverse bias rating, it may be more beneficial to increase
the windings ratio until a diode with low forward drop can
be selected and then reevaluate whether that solution with
higher RMS diode current is beneficial.
applications inForMation
LT3748
23
3748fb
For more information www.linear.com/LT3748
6. Select the Feedback Resistor for Proper Output
Voltage
Using the iterative process laid out earlier in the Applications
Information section, select the feedback resistor RFB and
program the output voltage to 5V. Adjust the RTC resistor
for temperature compensation of the output voltage. RREF
is selected as 6.04k.
7. Select the Output Capacitor
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in
size and cost of a larger capacitor. The following equation
calculates the output voltage ripple:
∆VMAX =LPRI •ILIM2
2•C
OUT
•V
OUT
8. Add Snubber Circuitry as Necessary
With the primary components selected, the application
should be constructed to evaluate ringing at the drain of the
MOSFET switch and to evaluate step response to optimize
the compensation network. If using an RC snubber, the
equations from the Applications Information section can
be used or a rough estimate of component values may
come from using the published leakage inductance of the
transformer and selecting a snubber capacitor ranging
from 1 to 3 times larger than the published MOSFET output
capacitance. In this application, the peak MOSFET drain
voltage was measured at maximum load from minimum VIN
and exceeded the 150V rating of the Si7738. A DZ clamp
applications inForMation
was considered in order to maximize efficiency but was
unable to turn on fast enough to sufficiently clamp the very
fast leakage spike. The final solution is an RC snubber,
implemented iteratively, that decreases efficiency by less
than 1% across the majority of the output load range while
reducing the worst-case drain voltage spike to just 80V.
Similarly, the anode of the output diode is probed to look
at potential ringing when the MOSFET switch turns on and
a peak of 45V is measured across the diode. Therefore,
no snubber circuitry is required.
9. Optimize the Compensation Network
To set the compensation, the application is first configured
with a 22nF capacitor and 10k resistor as a starting point. A
load step is applied at both light and heavy loads at the 60V
maximum input voltage and the capacitance is decreased
until damping decreases to the desired limit, in this case
with a compensation capacitance of 2.2nF and a response
implying about 60˚ of phase margin. After verifying stability
at the minimum input voltage, as well, the compensation
capacitance is doubled for safety margin. The series resis-
tance is varied from 5k to 50k but the optimal response is
observed with 24.7k. For best ripple performance, select
a compensation capacitor not less than 1nF, and select a
compensation resistor not greater than 50k.
10. Soft-Start Capacitor and UVLO Resistor Divider
A soft-start capacitor helps during the start-up of the
flyback converter. Select the UVLO resistor divider for
the intended input operation range. These equations are
aforementioned.
LT3748
24
3748fb
For more information www.linear.com/LT3748
applications inForMation
Table 3. Voltage Stresses, Output Capability and Diode Current vs Turns Ratio in 48VIN to 12V, 2A Application
NPS VDS(MAX) VR(DIODE) D (VIN = 48V) D (VIN = 36V) ILIM (2A OUT AT VIN = 36V) IDIODE(RMS) (VIN = 48V)
1 84 84 0.21 0.26 6 3.3
2 96 48 0.34 0.41 4 3.7
4 120 30 0.51 0.58 3 4.6
6 144 24 0.61 0.68 2 5.2
DESIGN EXAMPLE: 48VIN TO 12V, 2A OUT
The second example is a telecom application shown on
the front page of the datasheet. The focus of this applica-
tion is a cheap, small and simple solution. Table 3 shows
the results of the initial step for selecting the turns ratio.
In this example, the output diode is a much smaller ef-
ficiency loss due to the smaller voltage drop across it in
ratio to VOUT so minimizing output diode current is not
as important. Of greater importance is minimizing the
stresses on the MOSFET and output diode and the 4:1
case seems to be the best compromise for that to avoid
using a snubber on either device.
20µH of primary inductance is required for minimum
off-time while selecting the transformer, but in order to
minimize output ripple at maximum load a 60.8µH trans-
former is selected. To meet the saturation current (12A,
peak, on the secondary windings), a Versa-Pak VP4-0047-R
provides a compact and efficient solution.
For the MOSFET switch, since the input voltage is so high,
resistive losses on the primary side will be very low so
minimizing RDS(ON) is of minimum benefit. However, since
the current for the gate drive is pulled from a high VIN,
minimizing both QG and operating frequency is essential
unless a third winding is added. The Vishay Si7464DP, with
a 200V VDS(MAX) and low gate charge, keeps the INTVCC
current to just over 3mA, worst-case, which when added
to quiescent current will keep power dissipation in the
LT3748 to just over 1/4W at 72V VIN.
The output diode only nominally has 30V of reverse bias
but a B360 diode is selected to ensure enough margin that
a snubber will not be required. A more expensive diode
with lower forward drop might recover several percent
efficiency and if high temperature operation is required
a diode rated for more average current at temperature
might be needed, but the B360 is small and inexpensive.
The rest of the design and component selection is straight-
forward.
Suggested Layout
See Figures 14 and 15 for the DC1557A demo board lay-
out. Note the proximity of the RREF and RFB resistors (R9,
R5) to the LT3748 for optimal regulation. The location of
these two resistors as close to the physical pins of the
LT3748 is critical for accurate regulation. In addition, the
high frequency current path from the VIN bypass capacitor
(C2) through the primary-side winding, the MOSFET switch
and sense resistor (R10) is a very tight loop. Similarly,
the high frequency current path for the MOSFET gate
switching from the INTVCC capacitor through the source of
the MOSFET and sense resistor is similarly small in area.
For improved regulation it is recommended that the user
ensure that the high current ground is kept separate or
at least physically isolated from the small-signal ground
used by the other ground-referenced pins.
LT3748
26
3748fb
For more information www.linear.com/LT3748
Figure 16. Automotive IGBT Controller Supply
EN/UVLO
TC
SS
RFB
RREF
VCGND INTVCC
LT3748
3748 F16
133k 2nF
VIN
12V TYP
VIN
VO1
VO2
VO3
VO4
T1
1:1:1:1:1
D1
15V
300mA
825k
150k
10µF6µHC1
GATE
SENSE
1µF
4700pF 4.7µF
6.04k
71.5k
0.0125Ω
M1
10k
IGBT
DRIVER
D2
15V
300mA
C2
IGBT
DRIVER
D3
15V
300mA
C3
IGBT
DRIVER
D4
15V
300mA
C1-C4: 22µH 25V X7R ×2
D1-D4: DIODES INC. PDS3100
M1: VISHAY Si7898DP
T1: COILTRONICS VERSA-PAC VP4-0075-R
Z1: DIODES INC. DFLZ18-7
C4
Z1
Z1
Z1
49.9k
TL431ACD
ANODE
CATHODE
REF
320V
0V
3-PHASE
MOTOR
IGBT
DRIVER
9.09k
typical applications
LOAD CURRENT (mA)
0
OUTPUT VOLTAGE (V)
17.0
16.5
16.0
15.5
15.0
14.5
14.0
LT3748 F17
800400200 600
VO4 (NO LOAD)
VO3(100mA)
VO1 (300mA) VO2 (SWEPT)
Figure 17. Cross Regulation Performance of the Supply in Figure 16 with VO1 and VO3 Loaded with VO2 Swept
LT3748
27
3748fb
For more information www.linear.com/LT3748
typical applications
Figure 18. ±300V Isolated Flyback Converter
EN/UVLO
TC
SS
RFB
RREF
VCGND INTVCC
LT3748
3748 F18
R6
24.9k
VIN
7V TO 15V
VOUT+
300V
8mA
VOUT+
300V
8mA
VOUT–
VIN
T1
1:10:10
R1
357k
R2
93.1k
C1
10µFC5
GATE
SENSE
C2
1µF
C7
0.1µF
C4
2.2nF
C9
100pF C3
4.7µF
R4
6.04k
R3
140k
R5
10k R7
600k
R8
600k
D1
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
D3
50mΩ
M1 VOUT–
C6
C8
0.22µF
50V
D2
C5, C6: 0.1µF 600V ×2
D1, D2: CENTRAL SEMICONDUCTOR CMR1U-06M LTC
M1: FAIRCHILD FDM3622
T1: WÜRTH ELEKTRONIK 750311486
D3: CENTRAL SEMICONDUCTOR CMMR1U-02
LT3748
28
3748fb
For more information www.linear.com/LT3748
typical applications
Figure 19. 48V, 0.5A Supply from 24V to 96V Input
OUTPUT CURRENT (A)
0
EFFICIENCY (%)
75
80
85
0.3 0.5
3748 F20
70
65
60 0.1 0.2 0.4
90
95
100
VIN = 24V
VIN = 48V
VIN = 96V
Figure 20. Efficiency of 48V Supply of Figure 17
EN/UVLO
TC
SS
RFB
RREF
VCGND INTVCC
LT3748
3748 F19
2nF
VIN
48V TYP
VOUT+
48V
0.5A
VOUT–
VIN
T1
1:1
825k
49.9k
4.7µF44.1µH
4.7µF
100V
×3
GATE
SENSE
0.22µF
4700pF 4.7µF
6.04k
226k
66Ω
150pF
D1
0.030Ω
D1: CENTRAL SEMICONDUCTOR CMR5U-02-LTC
M1: VISHAY Si7464DP
T1: COILTRONICS VERSA-PAC VP4-0060-R
M1
10k
LT3748
29
3748fb
For more information www.linear.com/LT3748
typical applications
Figure 21. 5V, 8A Isolated Supply with Synchronous Secondary-Side Rectification Using LT8309
EFFICIENCY (%)
I
LOAD
(A)
90
92
76
12345678
84
82
80
78
88
86
90
VIN = 36V
VIN = 48V
VIN = 72V
LT8309 & MOSFET
PDS760
DIODE
Figure 22. Efficiency of the Supply in Figure 21 as well as Performance Using a Conventional PDS760 Schottky Rectifier
INTVCC
VCC
VOUT+
5V, 8A
VOUT–
D5
VIN
36V TO
72V
GATE
3784 TA21
DRAIN
GND
120pF
100Ω
6.04k
147k LT8309
•
•
0.22µF
28k
12.1k
15nF
0.012Ω
GATE
SENSE
RREF
INTVCC
VC
SS
TC
EN/UVLO
RFB
GND
LT3748
5.33:1:2.67
PA1735NL
3Ω
2.15k
4.7µF
M1 M2
4.7µF
910µF
470pF
1.2M
51k
62µF
1µF D6
4.7nF
D1
D2 D4
68Ω
D3
VIN
D1: SMBJ85A-13-F
D2: CMMRIU-02
D3: BAV20W-7-F
D4: BAV20W-7-F
D5: CMZ5919B
D6: CMHZ5258B
M1: BSC320N20NS3G
M2: BSC028N06NS
•
LT3748
30
3748fb
For more information www.linear.com/LT3748
typical applications
Figure 23 Thermal Image of the Supply in Figure 21 Using a PDS760 Instead of the LT8309 and Synchronous Switch at 5V/5A Output
Figure 24 Thermal Image of the Supply in Figure 21 with Synchronous Secondary-Side at 5V/5A Output with Much Lower Temperatures
LT3748
31
3748fb
For more information www.linear.com/LT3748
typical applications
Figure 25. 3.3V, 10A Isolated, Synchronous Flyback Converter
LOAD CURRENT (A)
0
EFFICIENCY (%)
100
95
90
85
80
75
70
LT3748 F26
10800400200 600
VIN = 36V
VIN = 48V
VIN = 72V
Figure 26. Efficiency of the Supply in Figure 25
INTVCC
VCC
VOUT+
3.3V, 10A
VOUT–
D5
VIN
36V TO
72V
GATE
3748 F25
DRAIN
GND
120pF
100Ω
6.04k
158k LT8309
0.22µF
19.1k
470pF
•
•
15k
22nF
0.015Ω
GATE
SENSE
RREF
INTVCC
VC
SS
TC
EN/UVLO
RFB
GND
LT3748
8:1.4
PA1477NL
3Ω
2k
4.7µF
M1 M2
4.7µF
1500µF
1.2M
51k
62µF
1µF D6
VIN
4.7nF
•
68Ω
D1: SMBJ85A-13-F
D2: CMMRIU-02
D3: BAV20W-7-F
D4: BAV20W-7-F
D5: CMZ5914 B
D6: CMHZ5258B
M1: BSC320N20NS3G
M2: BSC016N04LS
D3
D1
D2 D4
LT3748
32
3748fb
For more information www.linear.com/LT3748
package Description
MSOP (MS12) 0510 REV A
0.53 ± 0.152
(.021 ± .006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
0.86
(.034)
REF
1.0
(.0394)
BSC
0.50
(.0197)
BSC
16 14 121110
1 3 5 6 7 8
9
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ± 0.127
(.035 ± .005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 ± 0.038
(.0120 ± .0015)
TYP
0.50
(.0197)
BSC
1.0
(.0394)
BSC
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
0.1016 ± 0.0508
(.004 ± .002)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
0.280 ± 0.076
(.011 ± .003)
REF
4.90 ± 0.152
(.193 ± .006)
MS Package
Varitation: MS16 (12)
16-Lead Plastic MSOP with 4 Pins Removed
(Reference LTC DWG # 05-08-1847 Rev A)
LT3748
33
3748fb
For more information www.linear.com/LT3748
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision history
REV DATE DESCRIPTION PAGE NUMBER
A 10/10 Added H-grade information to Absolute Maximum Ratings, Pin Configuration, Order Information, and Electrical
Characteristics sections.
Revised text and Table 2 in the Applications Information section.
Revised Figures 10 and 17 in the Applications Information section.
Revised Typical Application drawing.
2, 3
15, 16, 20, 22
26, 27
30
B 2/15 Added MP-grade device.
Added Synchronous Secondary Applications paragraphs
Added Figures 21, 22, 23, 24, 25 and 26
2, 3
20
29, 30, 31
LT3748
34
3748fb
For more information www.linear.com/LT3748
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com
LINEAR TECHNOLOGY CORPORATION 2010
LT 0215 REV B • PRINTED IN USA
relateD parts
typical application
5V, 2A Output from Automotive Input with Continuous Operation from 6V to 45V
EN/UVLO
TC
SS
RFB
RREF
VCGND INTVCC
LT3748
3748 TA02
86.6k 47nF
VIN
12V TYP
VOUT+
5V, 2A
VOUT–
VIN
T1
2:1
D1
D2
825k
215k
10µF
8.3µH
GATE
SENSE
100µF
10V
18.2Ω
2.2nF 4.7µF
6.04k
330pF
48.7k
0.016Ω
D1: DIODES INC. SBR8U60P5
D2: DIODES INC. BZT52C5V6
M1: Si7738DP
T1: PULSE PA3177NL
M1
24.7k
PART NUMBER DESCRIPTION COMMENTS
LT8300 100VIN Micropower Isolated Flyback Converter with
150V/260mA Switch Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23
LT8301 42VIN Micropower Isolated Flyback Converter with
65V/1.2A Switch Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23
LT8302 42VIN Micropower Isolated Flyback Converter with
65V/3.6A Switch Low IQ Monolithic No-Opto Flybacks, 8-Lead SO-8E
LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23
LT3573 40V Isolated Flyback Converter Monolithic No-Opto Flyback with Integrated 1.25A, 60V Switch
LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 0.65A / 2.5A 60V Switch
LT3757/LT3758 40V/100V Flyback, Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958 40V/100V Flyback, Boost Converters Monolithic with Integrated 5A/3.3A Switch
LT1725 20V Isolated Flyback Controller Controller with Load Compensation Circuitry
LT1737 20V Isolated Flyback Controller No Opto-Isolator or Third Winding Required, Up to 50W Output
LTC
®
3803/LTC3803-3
LTC3803-5 200kHz/300kHz Flyback DC/DC Controllers VIN and VOUT Limited Only by External Components
LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited Only by External Components
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT3748
