ISL6537ACR - Intersil Corporation

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

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All other trademarks mentioned are the property of their respective owners.

ISL6537A

ACPI Regulator/Controller for

Dual Channel DDR Memory Systems

The ISL6537A provides a complete ACPI compliant power

solution for up to 4 DIMM dual channel DDR/DDR2 Memory

systems. Included are both a synchronous buck controller to

supply VDDQ during S0/S1 and S3 states. During S0/S1 state,

a fully integrated sink-source regulator generates an accurate

(VDDQ/2) high current VTT voltage without the need for a

negative supply. A buf fered version of the VDDQ/2 reference is

provided as VREF. A second PWM controller, which requires

external MOSFET drivers, is available for regulation of the

GMCH Core voltage. An LDO controller is also integrated for

the CPU VTT termination voltage regulation and the DAC.

The switching PWM controller drives two N-Channel

MOSFETs in a synchronous-rectified buck converter

topology. The synchronous buck converter uses voltage-

mode control with fast transient response. The switching

regulator prov i d es a ma xi mu m static regula ti on to le ra n c e of

±2% over line, load, and temperatur e ranges. The output is

user-adjustable by means of external resistors down to 0.8V.

An integrated soft-start feature brings all outputs into

regulation in a controlled manner when returning to S0/S1

state from any sleep state. During S0 the VIDPGD signal

indicates that the GMCH and CPU VTT termination voltage

is within spec and operational.

All outputs, except VDAC, have un dervoltage protection.

The switching regulator also has overvoltage and

overcurrent protection. Thermal shutdown is integrated.

Pinout ISL6537A (6X6 QFN)

TOP VIEW

Features

• Generates 5 Regulated Voltages

- Synchronous Buck PWM Controller for DDR VDDQ

- 3A Inte grated Sink/Source Linear Regulator with

Accurate VDDQ/2 Divider Reference for DDR VTT

- PWM Regulator for GMCH Core

- LDO Regulator for CPU/GMCH VTT Termination

- LDO Regulator for DAC

• ACPI Compliant Sleep State Control

• Glitch-Free Transitions During State Changes

• Integrated VREF Buffer

•V

DDQ PWM Controller Drives Low Cost N-Channel

MOSFETs

• 250kHz Constant Frequency Operation

- Both PWM Controllers are Phase Shifted 180°

• Tight Output Voltage Regulation

- All Outputs: ±2% Over Temperature

• Fully-Adjustable Outputs with Wide Voltage Range: Down

to 0.8V Supports DDR and DDR2 Specifications

• Simple Single-Loop Voltage-Mode PWM Control Design

• Fast PWM Converter Transient Response

• Under and Overvoltage Monitoring

• OCP on th e VDDQ Switching Regulator

• Integrated Thermal Shutdown Protection

• Pb-Free Plus Anneal Available (RoHS Compliant)

Applications

•Single and Dual Channel DDR Memory Power Systems in

ACPI Compliant PCs

•Graphics Cards - GPU and Memory Supp lies

• ASIC Power Supplies

• Embedded Processor and I/O Supplies

• DSP Supplies

5VSBY

S3#

P12V

GND

DDR_VTT

VDDQ

DRIVE3

FB3

PWM4

FB4

COMP4

COMP

LGATE

GND

UGATE

BOOT

PHASE

S5#

OCSET

VDDQ

DDR_VTTSNS

DRIVE2

FB2

VIDPGD

VREF_OUT

VREF_IN

28 27 26 25 24 23 22

GND

10 11 12 13 14

Data Sheet July 18, 2007 FN9143.5

2FN9143.5

Ordering Information

PART NUMBER PART MARKING TEMPERATURE

RANGE (°C) PACKAGE PKG.

DWG. #

ISL6537ACR ISL6537ACR 0 to +70 28 Ld 6x6 QFN L28.6x6

ISL6537ACRZ (Note) ISL6537ACRZ 0 to +70 28 Ld 6x6 QFN (Pb-free) L28.6x6

ISL6537ACRZA (Note) ISL6537ACRZ 0 to +70 28 Ld 6x6 QFN (Pb-free) L28.6x6

*Add “-T” suffix to part number for tape and reel packaging.

NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate

termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL

classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.

ISL6537A

3FN9143.5

Block Diagram

EA1

PWM

UGATE

VTT(2)

VTTSNS

250kHz

FB COMP

VTT

LGATE

P12V

VREF_IN

VDDQ(2)

VIDPGD

REG

S5#S3#

5VSBY

BOOT

VREF_OUT

EA3

PHASE

20µA

OCSET

COMP

GND(2)

P12V

PWM4

P12V

FB4

COMP4

DRIVE2

FB2

DRIVE3

FB3

EA2

EA4

5VSBY

UV/OV

FAULT

GND PAD

OSCILLATOR

POR

180°

PHASE

SHIFT

VOLTAGE

REFERENCE

0.800V

0.680V (-15%)

0.920V (+15%)

SOFTSTART & ENABLE A

SOFTSTART & ENABLE B

SOFTSTART & ENABLE C

MONITOR AND CONTROL

ENABLE DDR_VTT

EA1 ACTIVE

IN S3

ENABLE VIDPGD

ISL6537A

4FN9143.5

Simplified Power System Diagram

Typical Application

PWM

5VSBY

VTT

ISL6537A

CONTROLLER

REGULATOR

12V

VREF

VTT

SLEEP

STATE

LOGIC

SLP_S3

SLP_S5

3V3ATX

VDDQ

5VDUAL

Q2 +

VTT_GMCH/CPU

LINEAR

CONTROLLER

VGMCH

LINEAR

CONTROLLER

+VDAC

INTERSIL

FET DRIVER

PWM

CONTROLLER

3V3ATX

5VSBY

ISL6537A

FB4

GND

VGMCH

5VSBY

P12V

SLP_S5

SLP_S3

S5#

S3#

12V

ATX3V3

DRIVE2

FB2

VTT_GMCH/CPU

VIDPGD

PWM4

COMP4

INTERSIL

FET DRIVER

DRIVE3

FB3

VDAC

UGATE

COMP

LGATE

DDR_VTT(x2) VTT_DDR

VREF_IN

VREF_OUT

DDR_VTTSNS

VDDQ_DDR

5VDUAL

OCSET

PHASE

VREF

DDR_VDDQ(x2)

BOOT

ROCSET

C2 R3 C3

C7 R8

R9 R10

R12

R11

3VDUAL

DBOOT

CBOOT

ATX3V3

ISL6537A

5FN9143.5

Absolute Maximum Ratings Thermal Information

5VSBY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .GND - 0.3V to +7V

P12V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .GND - 0.3V to +14V

Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +15.0V

Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . .7.0V (DC)

8.0V (<10ns Pulse Width, 10μJ)

All other Pins . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5VCC + 0.3V

ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .Class 2

Recommended Operating Conditions

Supply Voltage on 5VSBY . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%

Supply Voltage on P12V . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%

Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . 0°C to +70°C

Junction Temperature Range. . . . . . . . . . . . . . . . . . . 0°C to +125°C

Thermal Resistance (Typical, Notes 1, 2) θJA (°C/W) θJC (°C/W)

QFN Package . . . . . . . . . . . . . . . . . . . 32 5

Maximum Junction Temperature (Plastic Package) . . . . . . +150°C

Maximum Storage Temperature Range. . . . . . . . . -65°C to +150°C

Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below

http://www.intersil.com/pbfree/Pb-FreeReflow.asp

CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and

result in failures not covered by warranty.

NOTES:

1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See

Tech Brief TB379.

2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.

Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System

Diagrams and Typical Application Schematics

PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS

5VSBY SUPPLY CURRENT

Nominal Supply Current ICC_S0 S3# and S5# HIGH, UGATE/LGATE Open 5.5 7.0 8.0 mA

ICC_S5 S5# LOW , S3# Don’t Care, UGATE/LGATE Open - 700 850 μA

POWER-ON RESET

Rising 5VSBY POR Threshold 4.10 - 4.45 V

Falling 5VSBY POR Threshold 3.60 - 3.95 V

Rising P12V POR Threshold 10.0 - 10.5 V

Falling P12V POR Threshold 8.80 - 9.75 V

OSCILLATOR AND SOFT-START

PWM Frequency fOSC 220 250 280 kHz

Ramp Amplitude ΔVOSC -1.5- V

Soft-Start Interval tSS 6.5 8.2 9.5 ms

REFERENCE VOLTAGE

Reference Vo ltage VREF -0.800- V

System Accuracy -2.0 - +2.0 %

VDDQ AND VGMCH PWM CONTROLLER ERROR AMPLIFIERS

DC Gain (Note 3) - 80 - dB

Gain-Bandwidth Product GBWP (Note 3) 15 - - MHz

Slew Rate SR (Note 3) - 6 - V/μs

CONTROL I/O (S3#, S5#)

LOW Level Input Threshold 0.75 - - V

HIGH Level Input Threshold --2.2V

ISL6537A

6FN9143.5

Functional Pin Description

5VSBY (Pin 1)

5VSBY is the bias supply of the ISL6537A. It is typically

connected to the 5V standby rail of an ATX power supply.

During S4/S5 sleep states the ISL6537A enters a reduced

power mode and draws less than 1mA (ICC_S5) from the

5VSBY supply. The supply to 5VSBY should be locally

bypassed using a 0.1μF capacitor.

P12V (Pin 3)

The VTT regulation circuit and the Linear Drivers are

operation. P12V is typically connected to the +12V rail of an

ATX power supply.

GND (Pins 4, 27, 29)

The GND terminals of the ISL6537A provide the return path

for the VTT LDO, and switching MOSFET gate drivers. High

PWM CONTROLLER GATE DRIVERS

UGATE and LGATE Source IGATE --0.8- A

UGATE and LGATE Sink IGATE -0.8- A

VTT REGULATOR

Upper Divider Impedance RU-2.5- kΩ

Lower Divider Impedance RL-2.5- kΩ

VREF_OUT Buffer Source Current IVREF_OUT --2mA

Maximum VTT Load Current IVTT_MAX Periodic load applied with 30% duty cycle and

10ms period using ISL6537A_6506EVAL1

evaluation board (see Application Note AN1 124)

-3 - 3 A

LINEAR REGULATORS

DC Gain (Note 3) - 80 - dB

Gain Bandwidth Product GBWP (Note 3) 15 - - MHz

Slew Rate SR (Note 3) - 6 - V/μs

DRIVEn High Output Voltage DRIVEn Unloaded 9.75 10.0 - V

DRIVEn Low Output Voltage -0.160.50 V

DRIVEn High Output Source Current VFB = 770mV, VDRIVEn = 0V - 1.7 - mA

DRIVEn Low Output Sink Current VFB = 830mV, VDRIVEn = 10V - 1.20 - mA

VIDPGD

VTT_GMCH/CPU Rising Threshold S0 .725 .740 - V

VTT_GMCH/CPU Falling Threshold S0 - 0.700 0.715 V

PROTECTION

OCSET Current Source IOCSET 18 20 22 μA

VTT_DDR Current Limit (Note 3) -3.3 - 3.3 A

VDDQ OV Level VFB/VREF S0/S3 - 115 - %

VDDQ UV Level VFB/VREF S0/S3 - 75 - %

VTT_DDR OV Level VTT/VVREF_IN S0 - 115 - %

VTT_DDR UV Level VTT/VVREF_IN S0 - 85 - %

VGMCH UV Level VFB4/VREF S0 - 75 - %

VTT_GMCH/CPU UV Level VFB2/VREF S0 - 75 - %

Thermal Shutdown Limit TSD (Note 3) - 140 - °C

NOTE:

3. Limits should be considered typical and are not production tested

Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System

Diagrams and Typical Application Schematics (Continued)

PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS

ISL6537A

7FN9143.5

ground currents are conducted directly through the exposed

paddle of the QFN package which must be electrically

connected to the ground plane through a path as low in

inductance as possible.

UGATE (Pin 26)

Connect this pin to the upper MOSFET’s gate. This pin

provides the PWM-controlled gate drive for the upper

MOSFET. This pin is also monitored by the adaptive shoot-

through protection circuitry to determine when the upper

MOSFET has turned off. Do not insert any circuitry between

this pin and the gate of the upper MOSFET, as it may

interfere with the internal adaptive shoot-th r ough protection

circuitry and render it ineffective.

LGATE (Pin 28)

Connect this pin to the lower MOSFET’s gate. This pin

provides the PWM-controlled gate drive for the lower

MOSFET. This pin is also monitored by the adaptive shoot-

through protection circuitry to determine when the lower

MOSFET has turned off. Do not insert any circuitry between

this pin and the gate of the lower MOSFET, as it may

interfere with the internal adaptive shoot-th r ough protection

circuitry and render it ineffective.

FB (Pin 15) and COMP (Pin 16)

The VDDQ switching regulator employs a single voltage

control loop. FB is the negative input to the voltage loop error

amplifier. The VDDQ output voltage is set by an external

resistor divider connected to FB. With a properly selected

divider, VDDQ can be set to any vo ltage between the power

rail (reduced by converter losses) and the 0.8V reference.

Loop compensation is achieved by connecting an AC

network across COMP and FB.

The FB pin is also monitored for under and overvoltage

events.

PHASE (Pin 24)

Connect this pin to the upper MOSFET’s source. This pin is

used to monitor the voltage drop across the upper MOSFET

for overcurrent protection.

OCSET (Pin 22)

Connect a resistor (ROCSET) from this pin to the drain of the

upper MOSFET. ROCSET, an internal 20μA current source

(IOCSET), and the upper MOSFET on-resistance (rDS(ON))

set the conver ter overcurren t (OC) trip point according to t he

following equation:

An overcurrent trip cycles the soft-start function.

VDDQ (Pins 7, 8)

The VDDQ pins should be connected externally together to

the regulated VDDQ outpu t. During S0/S1 states, the VDDQ

pins serve as inputs to the VTT regulator and to the VTT

Reference precision divider.

DDR_VTT (Pins 5, 6)

The DDR_VTT pins should be connected externally

together. During S0/S1 states, the DDR_VTT pins se rve as

the outputs of the VTT linear regulator. During S3 state, the

VTT regulator is disabled.

DDR_VTTSNS (Pin 9)

VTTSNS is used as the feedback for control of the VTT linear

regulator. Connect th is pin to the VTT output at the physical

point of desired regulation.

VREF_OUT (Pin 13)

VREF_OUT is a buff ered version of VTT and also acts as the

reference vol tage for the VTT linear regulator. It is

recommended that a minimum capacitance of 0.1μF is

connected between VDDQ and VREF_OUT and also

between VREF_OUT and ground for proper operation.

VREF_IN (Pin 14)

A capacitor, CSS, connected between VREF_IN and ground

is required. This capacitor and the parallel combination of

the Upper and Lower Divider Impeda nce (RU||RL), sets the

time constant for the start up ramp when transitioning from

S3/S4/S5 to S0/S1/S2.

The minimum value for CSS can be fo und through the

following equation:

The calculated capacitance, CSS, will charge the output

capacitor bank on the VTT rail in a controlled manner without

reaching the current limit of the VTT LDO.

BOOT (Pin 25)

This pin provides ground referenced bias voltage to the

upper MOSFET driver . A bootstrap circuit is used to create a

voltage suitable to drive a logic-level N-channel MOSFET.

PWM4 (Pin 19)

This pin provides the PWM output for the GMCH core

switching regulator. Connect this pin to the PWM input of an

Intersil MOSFET driver.

FB4 (Pin 19) and COMP4 (Pin 17)

The GMCH core switching regulator employs a single

voltage control loop. FB4 is the negative input to the voltage

loop error amplifier. The GMCH core output voltage is set by

an external resistor divider connected to FB4. With a

properly selected divider , VGMCH can be set to any voltage

between the power rail (reduced by converter losses) and

the 0.8V reference. Loop compensation is achieved by

connecting an AC network across COMP4 and FB4.

The FB4 pin is also monitored for undervoltage events.

IPEAK IOCSETxROCSET

rDS ON()

-------------------------------------------------=(EQ. 1)

CSS CVTTOUT VDDQ

⋅

10 2A RURL

⋅⋅

------------------------------------------------

>(EQ. 2)

ISL6537A

8FN9143.5

FB2 (Pin 18)

Connect the output of the VTT_GMCH/CPU linear regulator to

this pin through a properly sized resistor divider . The voltage

at this pin is regulated to 0.8V. This pin is monitored for

undervoltage events.

DRIVE2 (Pin 10)

This pin provides the gate voltage for the VTT_GMCH/CPU

linear regulator pass transistor. Connect this pin to the gate

terminal of an external N-Channel MOSFET transistor.

FB3 (Pin 18)

Connect the output of the DAC linear regulator to this pin

through a properly sized resistor divider. The voltage at this

pin is regul a ted to 0.8V.

DRIVE3 (Pin 10)

This pin provides the gate voltage for the DAC linear

regulator pass transistor. Connect this pin to the gate

terminal of an external N-Channel MOSFET transistor.

VIDPGD (Pin 12)

The VIDPGD pin is an open-drain logic output that changes

to a logic lo w if the VTT_GMCH/CPU linear regul ator is out of

regulation in S0/S1/S2 state. VIDPGD will always be low in

any state other than S0/S1/S2.

SLP_S5# (Pin 23)

This pin accepts the SLP_S5# sleep state signal.

SLP_S3# (Pin 2)

This pin accepts the SLP_S3# sleep state signal.

Functional Description

Overview

The ISL6537A provides complete control, drive, protection

and ACPI compliance for regulator powering DDR memory

systems and the GMCH core and GMCH/CPU termination

rails. It is primarily designed for computer applications

powered from an ATX power supply.

A 250kHz Synchronous Buck Regulator with a precision

0.8V reference provides the proper Core voltage to the

system memory of the computer. An internal LDO regulator

with the ability to both sink and source current and an

externally available buffered reference that tracks the VDDQ

output by 50% provides the VTT termination voltage.

A second 250kHz PWM Buck regulator, which requires an

external MOSFET driver, provides the GMCH core voltage.

This PWM regulator is +180° out of phase with the PWM

regulator used for the Memory core. Two additional LDO

controllers are included, one for the regulation of the

GMCH/CPU termination rail and the second for the DAC.

ACPI compliance is realized through the SL P_S3 and

SLP_S5 sleep signals and through monitorin g of the 12 V

ATX bus.

Initialization

The ISL6537A automati cally initializes upon receipt of input

power. Special sequencing of the input supplies is not

necessary. The Power-On Reset (POR) function continually

monitors the input bias supply voltages. The POR monitors

the bias voltage at the 5VSBY and P12V pins. The POR

function initiates soft-start operation after the bias supply

voltages exceed their POR thresholds.

ACPI State Transitio ns

Figure 1 shows how the individual regulators are controlled

during all state transitions. All references to timing in this

section are in reference to Figure 1.

Cold Start (S4/S5 to S0 Transition)

At the onset of a mechanical start, time t0 in Figure 1, the

ISL6537A receives its bias voltage from the 5V Standby bus

(5VSBY). Once the 5VSBY rail has exceeded the POR

threshold, the ISL6537A will remain in an internal S5 state

until both the SLP_S3 and SLP_S5 signal have transitioned

high and the 12V POR threshold has been exceeded by the

+12V rail from the ATX, which occurs at time t1.

Once all of these conditions are met, the PWM error

amplifiers will first be reset by internally shorting the COMP

pins to the respective FB pins. This reset lasts for three soft-

start cycles, which is typically 24ms (one soft-start cycle is

typically 8.2ms). The digital soft-start sequence will then

begin. Each regulator is enabled and soft-started according

to a preset sequence.

At time t2, the 3 soft-start cycle reset has ended and the

VDDQ_DDR rail is digitally soft-started.

The digital soft-start for both PWM regulators is accomplished

by clamping the error amplifier reference input to a level

proportional to the internal digital soft-start voltage. As the sof t-

start voltage slews up, the PWM comparator generates PHASE

pulses of increasing width that charge the output capacitor(s).

This method provides a rapid and controlled output voltage rise.

The linear regulators, with the exception of the internal

VTT_DDR LDO, are soft-started in a similar manner. The

error amplifier reference is clamped to the internal digital

soft-start voltage. As the soft-start voltage ramps up, the

respective DRIVE pin voltages increase, thus enhancing the

N-MOSFETs and charging the output capacitors in a

controlled manner.

At time t3, the VDDQ_DDR rail is in regulation and th e

VGMCH rail is soft-started. At time t4, the VGMCH rail is in

regulation and the VTT_GMCH/CPU and the DAC linear

regulators are soft-started. At time t5, the VTT_GMCH/CPU

rail and DAC rails are in regulation and the VTT_DDR internal

regulator is soft-started.

The VTT_DDR LDO soft-starts in a manner unlike the other

regulators. When the VTT_DDR regulator is disabled, the

reference is internally shorted to the VTT_DDR output. This

ISL6537A

9FN9143.5

FIGURE 1. ISL6537A TIMING DIAGRAM

SLP_S3#

SLP_S5#

12V

VDDQ_DDR

12V

POR

VDAC

VGMCH

VTT_GMCH/CPU

VIDPGD

t1t2t3t4t5t6t7t8t9t11 t12 t13 t14 t15

t10

VTT_DDR

VDDQ_DDR

(3 SOFTSTART CYCLES) (3 SOFTSTART CYCLES)

VTT_DDR FLOATING

VTT_DDR Soft-Start Rise Time Dependent Upon Capacitor On VREF_IN Pin

ISL6537A

10 FN9143.5

allows the termination voltage to float during the S3 sleep

state. When the ISL6537A enables the VTT_DDR regulator

or enters S0 state from a sleep state, this short is released

and the internal divide down resistors which set the

VTT_DDR voltage to 50% of VDDQ_DDR will provide a

controlled voltage rise on the capacitor that is tied to the

VREF_IN pin. The voltage on this capacitor is the reference

for the VTT_DDR regulator and the output will track it as it

settles to 50% of the VDDQ voltage. The combination of the

internal resistors and the VREF_IN cap acitor will determine

the rise time of th e VTT_DDR regulator (see the Functional

Pin Description section for proper sizing of the VREF_IN

capacitor).

At time t6, a full sof t-start cycle has passed from the time that

the VTT_DDR regulator was enabled. At this time the

VIDPGD comparator is enabled. Once enabled if the

VTT_GMCH/CPU output is within regulation, the VIDPGD pin

will be forced to a high impedance state.

Active to Sleep (S0 to S3 Transition)

When SLP_S3 goes LOW with SLP_S5 still HIGH, the

ISL6537A will disable all the regulators except for the VDDQ

regulator, which is continually supplied by the 5VDUAL rail.

VIDPGD will also transition LOW . When VTT is disabled, the

internal reference for the VTT regulator is internally shorted

to the VTT rail. This allows the VTT rail to float. When

floating, the voltage on the VTT rail will depend on the

leakage characteristics of the memory and MCH I/O pins. It

is important to note that the VTT rail may not bleed down to

0V. Figure 1 shows how the individual regulators are

affected by the S3 state at time t7.

Sleep to Active (S3 to S0 Transition)

When SLP_S3 transitions from LOW to HIGH with SLP_S5

held HIGH and after the 12V rail exceeds POR, the

ISL6537A will initiate the soft-start sequence. This sequence

is very similar to the mechanical start soft-start sequencing.

The transition from S3 to S0 is repr esented in Figure 1

between times t8 and t14.

At time t8, the SLP_S3 signal transitions HIGH. This enables

the ATX, which brings up the 12V rail. At time t9, the 12V rail

has exceeded the POR threshold and the ISL6537A enters a

reset mode that lasts for 3 sof t-sta rt cycles. At time t10, t he 3

soft-start cycle reset is ended and the ind ividual regulators

are enabled and soft-started in the same sequence as the

mechanical cold start sequence, with the exception that the

VDDQ regulator is already enabled and in regulation.

Active to Shutdown (S0 to S5 Transition)

When the system transitions from active, S0, state to

shutdown, S4/S5, state, the ISL6537A IC disables all

regulators and forces the VIDPGD pin LOW. This transition

is represented on Figure 1 at time t 15.

Fault Protection

The ISL6537A monitors the VDDQ regulator for under and

overvoltage eve nts. The V DDQ regulator also has overcurre nt

protection. The internal VTT_DDR LDO regulator is monitored

for under and overvoltage event s. All other regulators, with the

exception of the DAC LDO, are monitored for undervolt age

events.

An overvoltage event on either the VDDQ or VTT_DDR

regulator will cause an immediate shutdown of all regulators.

This can only be cleared by toggling the SLP_S5 signal such

that the system enters the S5 sleep state and then

transitions back to the active, S0, state.

If a regulator experiences any other fault condition (an

undervoltage or an overcurrent on VDDQ), then that

regulator, and only that regulator, will be disabled and an

internal fault counter will be incremented by 1. If the disabled

regulator is used as the input for another regulator, then that

cascoded regulator will also experience a fault condition due

to a loss of input. The cascoded regulator will be disabled

and the fault counter incremented by 1.

At every fault occurrence, the internal fault counter is

incremented by 1 and an internal F ault Reset Counter is

cleared to zero. The Fault Reset Counter will increment once

for every clock cycle (1 clock cycle is typically 1/250kHz, or

4μs). If the Fault Reset Counter reaches a count of 16384

before another fault occurs, then the Fault Counter is

cleared to 0. If a fault occurs prior to the Fault Reset Counter

reaching a count of 16384, then the Fault Reset Counter is

set back to zero.

The ISL6537A will immediately shut down whe n the Fault

Counter reaches a count of 4 when the system is restarting

from an S5 state into the active, or S0, state. The ISL6537A

will immediately shut down when the Fault Coun ter reaches

a count of 5 at any other time.

The 16384 counts that are required to reset the Fault Reset

Counter represent 8 soft-st art cycles, as one sof t-st art cycle is

2048 clock cycles. This allows the ISL6537A to attempt at least

one full soft-start sequence to restart the faulted regulators.

When attempting to restart a faulted regulator, the ISL6537A

will follow the preset start up sequencing. If a regulator is

already in regulation, then it will not be affected by the start

up sequencing.

VDDQ Overcurrent Protection

The overcurrent function protects the sw itching converter from

a shorted output by using the upper MOSFET on-resistance,

rDS(ON), to monito r the current. Thi s method enhan ces the

converter’s efficiency and reduces cost by eliminating a

current sensing resistor.

The overcurrent function cycles the sof t-st art fu nction in a

hiccup mode to provide fault protection. A resistor (ROCSET)

programs the overcurrent trip level (see Typical Application

ISL6537A

11 FN9143.5

diagrams on page 4). An internal 20 μA (typical) current sink

develops a voltage across ROCSET that is referenced to the

converter input volt age. When the vo lt age across the upper

MOSFET (also referenced to the converter input volt age)

exceeds the voltage across ROCSET, the overcurrent function

initiates a soft-st art seque nce. The initia ti on of sof t-st art may

affect other regu lators. The V TT_DDR regulator is directly

affected as it receives it s reference and input fro m VDDQ.

The overcurrent function will trip at a peak inductor current

(IPEAK) determined by:

where IOCSET is the internal OCSET current source (20μA

typical). The OC trip point varies mainly due to the MOSFET

rDS(ON) variations. To avoid ov ercurrent tripping i n th e

normal operating load range, find the ROCSET resistor from

the equation above with:

1. The maximum rDS(ON) at the highest junction

temperature.

2. The minimum IOCSET from the specification table.

3. Determine IPEAK for ,

where ΔI is the output inductor ripple current.

For an equation for the ripple current see the section under

component guidelines titled ‘Output Inductor Selection’.

A small ceramic capacitor should be placed in parallel with

OCSET

to smooth the voltage across

OCSET

in the

presence of switching noise on the input voltage.

Thermal Protection (S0/S3 State)

If the ISL6537A IC junction temp erature reaches a nominal

temperature of +140°C, all regulators will be disabled. The

ISL6537A will not re-enable the outp uts until the junction

temperature drops below +110°C a n d either the bias volt age

is toggled in order to initiate a POR or the SLP_S5 signal is

forced LOW and then back to HIGH.

Shoot-Through Protec tion

A shoot-throu g h co nd i ti o n occurs when both the upper and

lower MOSFETs are turned on simultaneously, effectively

shorting the input voltage to ground. To protect from a shoot-

through condition, the ISL6537A incorporates specialized

circuitry on the VDDQ regulator which insures that

complementary MOSFETs are not ON simultaneously.

The adaptive shoot-through protection utilized by the VDDQ

regulator looks at the lower gate drive pin, LGATE, and the

upper gate drive pin, UGATE, to determine whether a

MOSFET is ON or OFF. If the voltage from UGATE or from

LGATE to GND is less than 0.8V, then the respective

MOSFET is defined as being OFF and the other MOSFET is

allowed to turned ON. This meth od allows the VDDQ

regulator to both source and sink current.

Since the voltage of the MOSFET gates are being measured

to determine the state of the MOSFET, the designer is

encouraged to consider the repercussions of intro ducing

external components between the gate drivers and their

respective MOSFET gates before actually imple m enting

such measures. Doing so may interfere with the shoot-

through protection.

Application Guidelines

Layout Considerations

Layout is very important in high frequency switching

converter design. With power devices switching efficiently at

250kHz, the resulting current transitions from one device to

another cause voltage spikes across the interconne cting

impedances and parasitic circuit elements. These voltage

spikes can degrade efficiency, radiate noise into the circuit,

and lead to device overvoltage stress. Careful component

layout and printed circuit bo ard design minimizes these

voltage spikes.

As an example, consider the turn-off transition of the control

MOSFET. Prior to turn-off, the MOSFET is carrying the full

load current. During turn-off, current stops flowing in the

MOSFET and is picked up by the lowe r MOSFET. Any

parasitic inductance in the switched current path generates a

large voltage spike during the switching interval. Careful

component selection, tight layout of the critical components,

and short, wide traces minimizes the magnitude of voltage

spikes.

There are two sets of critical components in the ISL6537A

switching converter . The switching components are the most

critical because they switch large amounts of energy, and

therefore tend to generate large amounts of noise. Next are

the small signal components which connect to sensitive

nodes or supply critical bypass current and signal coupling.

A multi-layer printed circuit board is recommended. Figure 2

shows the connections of the critical components in the

converter. Note that capacitors CIN and COUT could each

represent numerous physical capacitors. Dedicate one solid

layer, usually a middle layer of the PC board, for a ground

plane and make all critical component ground connections

with vias to this layer. Dedicate another solid layer as a

power plane and break this plane into smaller islands of

common voltage levels. Keep the metal runs from the

PHASE terminals to the output inductor short. The power

plane should support the input power and output power

nodes. Use copper filled polygons on the top and bottom

circuit layers for the phase nodes. Use the remaining printed

circuit layers for small signal wiring. The wiring traces from

the GATE pins to the MOSFET gates should be kept short

and wide enough to easily handle the 1A of drive current.

In order to dissipate heat generated by the internal VTT

LDO, the ground pad, pin 29, should be connected to the

internal ground plane through at least four vias. This allows

IPEAK IOCSET x ROCSET

rDS ON()

-----------------------------------------------------= (EQ. 3)

IPEAK IOUT MAX()

ΔI()

----------

ISL6537A

12 FN9143.5

the heat to move away from the IC and also ties the pad to

the ground plane through a low imped ance path.

The switching components should be placed close to the

ISL6537A first. Minimize the length of the connections

between the input capacitors, CIN, and the power switches

by placing them nearby. Position both the ceramic and bulk

input capacitors as close to the upper MOSFET drain as

possible. Position the output inductor and output capacitors

between the upper and low er MOSFETs and the load.

The critical small sign al components include any bypass

capacitors, feedback components, and compensation

components. Place the PWM converter compensation

components close to the FB and COMP pins. The feedback

resistors should be located as close as po ssible to the FB

pin with vias tied straight to the ground pla ne as required.

Feedback Compensation - PWM Buck Converters

Figure 3 highlights the voltage-mode control loop for a

synchronous-rectified buck converter. The output voltage

(VOUT) is regulated to the Reference voltage level. The error

amplifier output (VE/A) is compared with the oscillator (OSC)

triangular wave to provide a pulse-w idth modulated (PWM)

wave with an amplitude of VIN at the PHASE node. The

PWM wave is smoothed by the output filter (LO and CO).

VDDQ

5VSBY

ISLAND ON POWER PLANE LAYER

ISLAND ON CIRCUIT AND/OR POWER PLANE LAYER

COUT1

CIN

5VDUAL

KEY

COMP

ISL6537A

UGATE

CBP

DRIVE3

5VSBY

FIGURE 2. PRINTED CIRCUIT BOARD POWER PLANES

AND ISLANDS

VDAC

FB3

VIA CONNECTION TO GROUND PLANE

COUT5

LOAD

R11

R12

PHASE

12VATX

CBP

GNDP

P12V

LGATE

VDDQ(2)

VTT(2)

COUT2

LOAD

VDDQ

VTT

GND PAD

VGMCH

COUT3

CIN

3.3VATX

COMP4

PWM4

FB4

LOAD

MOSFET

DRIVER

DRIVE2 VTT_GMCH/CPU

FB2

COUT4

LOAD

R10

3.3VATX

FIGURE 3. VOL T AGE-MODE BUCK CONVERTER

COMPENSATION DESIGN AND OUTPUT

VOLTAGE SELECTION

VDDQ

REFERENCE

ESR

VIN

ΔVOSC

ERROR

AMP

PWM

DRIVER

(PARASITIC)

ZFB

REFERENCE

R2C3

COMP

VDDQ

ZFB

ISL6537A

ZIN

COMPARATOR

DRIVER

DETAILED COMPENSATION COMPONENTS

PHASE

VE/A

-ZIN

OSC

VDDQ 0.8 1 R1

-------+

⎝⎠

⎜⎟

⎛⎞

×=

ISL6537A

13 FN9143.5

The modulator transfer function is the small-signal transfe r

function of VOUT/VE/A. This function is dominated by a DC

Gain and the output filter (LO and CO), with a double pole

break frequency at FLC and a zero at FESR. The DC Gain of

the modulator is simply the input voltage (VIN) divided by the

peak-to-peak oscillator voltage ΔVOSC.

Modulator Break Frequency Equations

The compensation network consists of the error amplifier

(internal to the ISL6537A) and the impedance networks ZIN

and ZFB. The goal of the compensation network is to provide

a closed loop transfer function with the highest 0dB crossing

frequency (f0dB) and adequate phase margin. Phase margin

is the difference between the closed loop phase at f0dB and

180 degrees. The equations below relate the compensation

network’s poles, zeros and gain to the components (R1, R2,

R3, C1, C2, and C3) in Figure 3. Use these guidelines for

locating the poles and zeros of the compensation network:

1. Pick Gain (R2/R1) for desired converter bandwidth.

2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).

3. Place 2ND Zero at Filter’s Double Pole.

4. Place 1ST Pole at the ESR Zero.

5. Place 2ND Pole at Half the Switching Frequency.

6. Check Gain against Error Amplifier’s Open-Loop Gain.

7. Estimate Phase Margin - Repeat if Necessary.

Compensation Break Frequen cy Equations

Figure 4 shows an asymptotic plot of the DC/DC converter’s

gain vs. frequency. The actual Modulator Gain ha s a high gain

peak due to the high Q factor of the output filter and is not

shown in Figure 4. Using the above gu idelines sho uld give a

Compensation Gain simi lar to the curve plotted . The open

loop error amplifier gain bo unds the comp ensation gain.

Check the compensation gain at FP2 with the cap abilities of

the error amplifier. The Closed Loop Gain is constructed on

the graph of Figure 4 by adding the Modulator Gain (i n dB) to

the Compensation Gain (in dB). This is equivalent to

multiplying the modulator transfer function to the

compensation transfer function and plotting the gain.

The compensation gain uses external impedance networks

ZFB and ZIN to provide a stable, high bandwidth (BW) overall

loop. A stable control loop has a gain crossing with

-20dB/decade slope and a phase margin greater than 45

degrees. Include worst case component variations when

determining phase margin.

Output Voltage Selection

The output volt age of all the exte rn al voltage regulators can

be programmed to any level between their individual inp ut

voltage and the in tern al re ference , 0.8V. An external resistor

divider is used to scale the output volt ag e relative to the

reference voltage and feed it ba ck to th e inverting input of the

error amplifier, refer to the Typical Application on page 4.

The output voltage programming resistor will depend on the

value chosen for the feedback resistor and the desired

output voltage of the particular regulator.

If the output voltage desired is 0.8V, simply route the output

voltage back to the respective FB pin through the feedback

resistor and do not populate the output voltage programming

resistor.

The output voltage for the internal VTT_DDR linear regulator

is set internal to the ISL6537A to track the VDDQ voltage by

50%. There is no need for external programming re sistors.

FLC 1

2π x LO x CO

-------------------------------------------= FESR 1

2π x ESR x CO

--------------------------------------------= (EQ. 4)

FZ1 1

2π x R2 x C1

------------------------------------=

FZ2 1

2π x R1R3

+() x C3

-------------------------------------------------------=

FP1 1

2π x R2 x C1 x C2

C1C2

----------------------

⎝⎠

⎜⎟

⎛⎞

---------------------------------------------------------=

FP2 1

2π x R3 x C3

------------------------------------=(EQ. 5)

100

-20

-40

-60

FP1

FZ2

10M1M100K10K1K10010

OPEN LOOP

ERROR AMP GAIN

FZ1 FP2

20LOG

FLC FESR

COMPENSATION

GAIN (dB)

FREQUENCY (Hz)

GAIN

20LOG

(VIN/ΔVOSC)

MODULATOR

GAIN

(R2/R1)

FIGURE 4. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN

CLOSED LOOP

GAIN

R4 R1 0.8V×

VDDQ 0.8V–

-----------------------------------=

R8 R5 0.8V×

VGMCH 0.8V–

----------------------------------------=

R10 R9 0.8V×

Vxxxxxxxxxxxx 0.8V–

-----------------------------------------------------------=

R12 R11 0.8V×

VDAC 0.8V–

----------------------------------=

TT_GMCH/CPU

(EQ. 6)

ISL6537A

14 FN9143.5

Component Selection Guidelines

Output Capacitor Selection - PWM Buck Converter

An output capacitor is required to filter the inductor current

and supply the load transient current. The filtering

requirements are a function of the switching frequency and

the ripple current. The load transient requirements are a

function of the slew rate (di/dt) and the magnitude of the

transient load current. These requirements are generally met

with a mix of capacitors and careful layou t.

DDR memory systems are capable of producing transient

load rates above 1A/ns. High frequency capacitors initially

supply the transient and slow the current load rate seen by th e

bulk capaci tors. Th e bulk filter cap acitor val ues are generall y

determined by the ESR (Effective Series Resistance) and

voltage ra ting requirement s ra th er than actual capacitance

requirements.

High frequency decoupling capacitors sho uld be placed as

close to the power pins of the load as physically possible. Be

careful not to add inductance in the circuit board wiring that

could cancel the usefulness of these low inductance

components. Consult with the manufact urer of the load on

specific decoupling requirements.

Use only specialized low-ESR cap acitors intended for

switching-regulator applications fo r the bulk capacitors. The

bulk capa cito r’s ESR will determine the outpu t ripp le voltage

and the initial volt age drop af ter a high slew-rate tra nsient. An

aluminum electrolytic capa citor’s ESR value is related to the

case size with lower ESR available in larger case sizes.

However, the Equivalent Series Inductance (ESL) of these

capacitors increases with case size and can reduce the

usefulness of the cap acitor to high slew-rate transient loadi ng.

Unfortunately, ESL is not a specified pa rameter. Work with

your capacitor supplier and measure the capacitor’s

impedance with frequency to select a suitable component. In

most cases, multiple electrolytic capacitors of smal l case siz e

perform better than a single larg e ca se cap acitor.

Output Capacitor Selection - LDO Regulators

The output capacitors used in LDO regulators are used to

provide dynamic load current. The amount of capacitance

and type of capacitor should be chosen with this criteria in

mind.

Output Inductor Selection

The output inductor is selected to meet the output voltage

ripple requirements and minimize the converter’s response

time to the load transient. The inductor value determines the

converter’s ripple current and the ripple voltage is a function

of the ripple current. The ripple voltage and current are

approximated by the following equations:

Increasing the value of inductance reduces the ripple current

and voltage. However, the large inductance values reduce

the converter’s response time to a load transient.

One of the parameters limiting the converter ’s response to

a load transient is the time required to change the inductor

current. Given a sufficiently fast control loop design, the

ISL6537A will p r o v i de e ithe r 0 % or 100% d u t y cy c l e in

response to a load transient. The response time is the time

required to slew the inductor current from an initial current

value to the transient current level. During this interval the

difference between the inductor current and the transient

current level must be supplied by the output capacitor.

Minimizing the response time can minimize the output

capacitance required.

The response time to a transient is different for the

application of load and the removal of load. The following

equations give the approximate response time interval for

application and removal of a transie nt load:

where: ITRAN is the transient load current ste p, tRISE is the

response time to the application of load, and tFALL is the

response time to the removal of load. The worst case

response time can be either at the application or removal of

load. Be sure to check both of these equations at the

minimum and ma ximum output levels for the worst case

response time.

Input Capacitor Selection - PWM Buck Converter

Use a mix of input bypass capacitors to control the voltage

overshoot across the MOSFETs. Use small ceramic

capacitors for high frequency decoupling and bulk capacitors

to supply the current needed ea ch time the upper MOSFET

turns on. Place the small ceramic cap acitors physically close

to the MOSFETs and between the drain of upper MOSFET

and the source of lower MOSFET.

The important parameters for the bulk input capacitance are

the voltage rating and the RMS current rating. For reliable

operation, select bulk capacitors with voltage and current

ratings above the maximum input voltage and largest RMS

current required by the circuit. Their voltage rating should be

at least 1.25 times greater than the maximum in put voltage,

while a voltage rating of 1.5 times is a conservative

guideline. For most cases, the RMS current rating

requirement for the input capacitor of a buck regulat or is

approximately 1/2 the DC load current.

The maximum RMS current required by the regulator may be

closely approximated through the following equatio n:

ΔI=VIN - VOUT

Fs x L

VOUT

VIN

ΔVOUT =ΔI x ESR

x(EQ. 7)

tRISE = L x ITRAN

VIN - VOUT

tFALL = L x ITRAN

VOUT

(EQ. 8)

IRMSMAX

VOUT

VIN

--------------IOUTMAX21

------ VIN VOUT

–

----------------------------- VOUT

VIN

--------------

⎝⎠

⎛⎞

×+

⎝⎠

⎛⎞

×=

(EQ. 9)

ISL6537A

15 FN9143.5

For a through hole design, several electrolytic capacitors

may be needed. For surface mount designs, solid tantalum

capacitors can be used, but caution must be exercised with

regard to the capacitor surge current rating. These

capacitors must be capable of handling the surge-current at

power-up. Some capacitor series available from reputable

manufacturers are surge current tested.

MOSFET Selection - PWM Buck Converter

The ISL6537A requires 2 N-Chan nel power MOSFETs for

switching power and a third MOSFET to block backfeed from

VDDQ to the Input in S3 Mode. These should be selected

based upon rDS(ON), gate supply requirements, and thermal

management requirements.

In high-current applications, the MOSFET power dissipation,

package selection and heatsink are the dominant design

factors. The power dissipation includes two loss

components; conduction loss and switching loss. The

conduction losses are the largest componen t of power

dissipation for both the upper and the lower MOSFETs.

These losses are distributed betwee n the two MOSFETs

according to duty factor. The switching losses seen when

sourcing current will be different from the switching losses

seen when sinking current. When sourcing current, the

upper MOSFET realizes most of the switching losses. The

lower switch realizes most of the switching losses when the

converter is sinking current (see the equations below).

These equations assume linear voltage-current transitions

and do not adequately model power loss due the reverse-

recovery of the upper and lower MOSFET’s body diode. The

gate-charge losses are dissipated in part by the ISL6537A

and do not significantly heat the MOSFETs. However, large

gate-charge increases the switching interval, tSW which

increases the MOSFET switching losses. Ensure that both

MOSFETs are within their maximum junction temperature at

high ambient temperature by calculating the temperature

rise according to package thermal-resistance specifications.

A separate heatsink may be necessary depending upon

MOSFET power , package type, ambient temperature and air

flow.

MOSFET Selection - LDO

The main criteria for selection of the linear regulator pass

transistor is package selection for efficient removal of heat.

Select a package and heatsink that maintains the junction

temperature below the rating with a maximum expected

ambient temperature.

The power dissipated in the linear regulato r is:

where IO is the maximum output current and VOUT is the

nominal output voltage of the linear regulator.

PLOWER = Io2 x rDS(ON) x (1 - D)

Where: D is the duty cycle = VOUT/VIN,

tSW is the combined switch ON and OFF time, and

fs is the switching frequency.

Approximate Losses while Sourcing current

Approximate Losses while Sinking current

PLOWER Io2rDS ON()

×1D–()×1

---Io⋅VIN

×tSW fs

××+=

PUPPER Io2rDS ON()

×D×1

---Io⋅VIN

×tSW fs

××+=

PUPPER = Io2 x rDS(ON) x D (EQ. 10)

PLINEAR IOVIN VOUT

–()×≅ (EQ. 11)

ISL6537A

All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.

Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality

Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without

notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and

reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result

from its use. No lice nse is gran t ed by i mpli catio n or other wise u nder an y p a tent or patent rights of Intersil or its subsidiari es.

For information regarding Intersil Corporation and its products, see www.intersil.com

FN9143.5

ISL6537A

Quad Flat No-Lead Plastic Package (QFN)

Micro Lead Frame Plastic Package (MLFP)

L28.6x6

28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE

(COMPLIANT TO JEDEC MO-220VJJC ISSUE C)

SYMBOL

MILLIMETERS

NOTESMIN NOMINAL MAX

A 0.80 0.90 1.00 -

A1 - - 0.05 -

A2 - - 1.00 9

A3 0.20 REF 9

b 0.23 0.28 0.35 5, 8

D 6.00 BSC -

D1 5.75 BSC 9

D2 3.95 4.10 4.25 7, 8

E 6.00 BSC -

E1 5.75 BSC 9

E2 3.95 4.10 4.25 7, 8

e 0.65 BSC -

k0.25 - - -

L 0.35 0.60 0.75 8

L1 - - 0.15 10

N282

Nd 7 3

Ne 7 3

P- -0.609

θ--129

Rev. 1 10/02

NOTES:

1. Dimensioning and tolerancing conform to ASME Y14.5-1994.

2. N is the number of terminals.

3. Nd and Ne refer to the number of terminals on each D and E.

4. All dimensions are in millimeters. Angles are in degrees.

5. Dimension b applies to the metallized terminal and is measured

between 0.15mm and 0.30mm from the terminal tip.

6. The configuration of the pin #1 identifier is optional, but must be

located within the zone indicated. The pin #1 identifier may be

either a mold or mark feature.

7. Dimensions D2 and E2 are for the exposed pads which provide

improved electrical and thermal performance.

8. Nominal dimensions are provided to assist with PCB Land Pattern

Design efforts, see Intersil Technical Brief TB389.

9. Features and dimensions A2, A3, D1, E1, P & θ are present when

Anvil singulation method is used and not present for saw

singulation.

10. Depending on the method of lead termination at the edge of the

package, a maximum 0.15mm pull back (L1) maybe present. L

minus L1 to be equal to or greater than 0.3mm.