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DATA SHEET
AAT1149: 3 MHz Fast Transient 400 mA Step-Down Converter
Applications
• Cellular phones
• Digital cameras
• Handheld instruments
• Microprocessor/DSP core/IO power
• PDAs and handheld computers
• USB devices
Features
• Ultra-small 0603 inductor (height = 1 mm)
• VIN range: 2.7 V to 5.5 V
• VOUT adjustable from 1.0 V to VIN
• Max output current: 400 mA
• Up to 98% efficiency
• 45 µA no-load quiescent current
• 3.0 MHz switching frequency
• 70 µs soft start
• Fast load transient
• Over-temperature protection
• Current limit protection
• 100% duty cycle low-dropout operation
• Shutdown current: <1 µA
• Temperature range: −40 °C to +85 °C
• SC70JW (8-pin, 2.2 mm × 2 mm) package (MSL1, 260 °C per
JEDEC-J-STD-020)
Description
The AAT1149 SwitchRegTM is a 3.0 MHz step-down converter with
an input voltage range of 2.7 V to 5.5 V and output voltage as low
as 1.0 V. It is optimized to react quickly to load variations and
operate with a tiny 0603 inductor that is only 1 mm tall.
The AAT1149 output voltage is programmable using external
feedback resistors. It can deliver 400 mA of load current while
maintaining a low 45 µA no-load quiescent current. The 3.0 MHz
switching frequency minimizes the size of external components
while keeping switching losses low.
The AAT1149 maintains high efficiency throughout the operating
range, which is critical for portable applications.
The AAT1149 is available in a Pb-free, space-saving 8-pin,
2.2 mm × 2.0 mm SC70JW package, and is rated over a −40 °C
to +85 °C temperature range.
A typical application circuit is shown in Figure 1. The pin
configuration is shown in Figure 2. Signal pin assignments and
functional pin descriptions are provided in Table 1.
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Figure 1. AAT1149 Typical Application Circuit
Figure 2. AAT1149 8-Pin SC70JW
(Top View)
Table 1. AAT1149 Signal Descriptions
Pin # Name Description
1 EN Enable pin.
2 FB Feedback input pin. This pin is connected to an external resistive divider for an adjustable output.
3 IN Input supply voltage for the converter.
4 LX Switching node. Connect the inductor to this pin. It is internally connected to the drain of both high- and low-side MOSFETs.
5 AGND Non-power signal ground pin.
6 PGND Main power ground return pins. Connect to the output and input capacitor return.
7 PGND Main power ground return pins. Connect to the output and input capacitor return.
8 PGND Main power ground return pins. Connect to the output and input capacitor return.
Electrical and Mechanical Specifications
The absolute maximum ratings of the AAT1149 are provided in
Table 2 and the electrical specifications are provided in Table 3.
Typical performance characteristics of the AAT1149 are illustrated
in Figures 3 through 28.
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Table 2. AAT1149 Absolute Maximum Ratings (Note 1)
Parameter Symbol Minimum Typical Maximum Units
Input voltage to GND VIN 6.0 V
LX to GND VLX −0.3 Vin + 0.3 V
FB to GND VFB −0.3 Vin + 0.3 V
EN to GND VEN −0.3 +6.0 V
Operating junction temperature TJ −40 +150 °C
Maximum soldering temperature (at leads, 10 seconds) TLEAD 300 °C
Maximum power dissipation (Note 2) PD 625 mW
Thermal resistance θJA 160 °C/W
Note 1: Exposure to maximum rating conditions for extended periods may reduce device reliability. There is no damage to device with only one parameter set at the limit and all other
parameters set at or below their nominal value. Exceeding any of the limits listed may result in permanent damage to the device.
Note 2: Derate 6.25 mW/°C above 25 °C.
CAUTION: Although this device is designed to be as robust as possible, Electrostatic Discharge (ESD) can damage this device. This device
must be protected at all times from ESD. Static charges may easily produce potentials of several kilovolts on the human body
or equipment, which can discharge without detection. Industry-standard ESD precautions should be used at all times.
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Table 3. AAT1149 Electrical Specifications (Note 1)
(VIN = 3.6 V, TA = −40 °C to 85 °C, Unless Otherwise Noted. Typical Values are at TA = 25 °C)
Parameter Symbol Test Condition Min Typical Max Units
Step-Down Converter
Input voltage VIN 2.7 5.5 V
UVLO threshold VUVLO
VIN rising 2.7 V
Hysteresis 100 mV
VIN falling 1.8 V
Output voltage tolerance VOUT IOUT = 0 to 400 mA, VIN = 2.7 V to 5.5 V −3.0 3.0 %
Adjustable output voltage range VOUT 1.0 VIN V
Quiescent current IQ No load 45 70 µA
Shutdown current ISHDN VEN = GND 1.0 µA
P-channel current limit ILIM 600 mA
High side switch On resistance RDS(ON)H 0.45 Ω
Low side switch On resistance RDS(ON)L 0.40 Ω
LX leakage current ILXLEAK VIN = 5.5 V, VLX = 0 to VIN, VIN = GND 1 µA
Line regulation ∆VLINEREG VIN = 2.7 V to 5.5 V 0.1 %/V
Out threshold voltage accuracy VOUT 0.6 V output, no Load, TA = 25 °C 591 600 609 mV
Out leakage current IOUT 0.6 V output 0.2 µA
Start-up time tS From enable to output regulation 70 µs
Oscillator frequency fOSC TA = 25 °C 3.0 MHz
Over-temperature shutdown threshold TSD 140 °C
Over-temperature shutdown hysteresis THYS 15 °C
EN
Enable threshold low VEN(L) 0.6 V
Enable threshold high VEN(H) 1.4 V
Input low current IEN VIN = VOUT = 5.5 V −1.0 1.0 µA
Note 1: Performance is guaranteed only under the conditions listed in this Table.
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Typical Performance Characteristics
Figure 3. Efficiency vs Load Current (VOUT = 3 V, L = 3 µH)
Figure 5. Efficiency vs Load Current (VOUT = 1.8 V, L = 2.2 µH)
Figure 7. No Load Quiescent Current vs Input Voltage
Figure 4. Load Regulation (VOUT = 3 V, L = 3 µH)
Figure 6. Load Regulation (VOUT = 1.8 V, L = 2.2 µH)
Figure 8. Switching Frequency vs Input Voltage
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Figure 9. Switching Frequency Variation vs Temperature
Figure 11. Line Regulation (VOUT = 3 V)
Figure 13. Line Regulation (VOUT = 1.1 V)
Figure 10. Output Voltage Error vs Temperature
(VIN = 3.6 V, VOUT = 1.8 V, IOUT = 400 mA)
Figure 12. Line Regulation (VOUT = 1.8 V)
Figure 14. Line Transient
(VOUT = 1.8 V; 400 mA Load; No Feed Forward Capacitor)
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Figure 15. Line Transient
(VOUT = 1.8 V; No Feed Forward Capacitor)
Figure 17. N-Channel RDS(ON) vs Input Voltage
Figure 19. Load Transient
(VOUT = 1.1 V; No Feed Forward Capacitor)
Figure 16. Line Transient (VOUT = 1.8 V; CFF = 100 pF)
Figure 18. P-Channel RDS(ON) vs Input Voltage
Figure 20. Load Transient (Vout = 1.1 V; CFF = 100 pF)
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Figure 21. Load Transient
(VOUT = 1.8 V; No Feed Forward Capacitor)
Figure 23. Load Transient
(VOUT = 1.8 V; No Feed Forward Capacitor)
Figure 25. Soft Start
(VOUT = 1.8 V; No Feed Forward Capacitor)
Figure 22. Load Transient (VOUT = 1.8 V; CFF = 100 pF)
Figure 24. Load Transient (VOUT = 1.8 V; CFF = 100 pF)
Figure 26. Soft Start
(VOUT = 1.8 V; CFF = 100 pF)
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Figure 27. Soft Start
(VOUT = 3 V; No Feed-Forward Capacitor)
Figure 28. Load Transient
(VOUT = 1.1 V; No Feed-Forward Capacitor)
Figure 29. AAT1149 Functional Block Diagram
Functional Description
The AAT1149 is a high performance 400 mA, 3.0 MHz
monolithic step-down converter. It minimizes external
component size, enabling the use of a tiny 0603 inductor that is
only 1 mm tall, and optimizes efficiency over the complete load
range. Apart from the small bypass input capacitor, only a small
L-C filter is required at the output. Typically, a 1.8 µH inductor
and a 4.7 µF ceramic capacitor are recommended (see Tables
of values). A functional block diagram is provided in Figure 29.
Only three external power components (CIN, COUT, and L) are
required. Output voltage is programmed with external feedback
resistors, ranging from 1.0 V to the input voltage. An additional
feed-forward capacitor can also be added to the external
feedback to provide improved transient response (see
Figure 31).
At dropout, the converter duty cycle increases to 100% and the
output voltage tracks the input voltage minus the RDS(ON) drop
of the P-channel high-side MOSFET.
The input voltage range is 2.7 V to 5.5 V. The converter
efficiency has been optimized for all load conditions, ranging
from no load to 400 mA.
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The internal error amplifier and compensation provides
excellent transient response, load, and line regulation. Soft start
eliminates any output voltage overshoot when the enable or the
input voltage is applied.
Control Loop
The AAT1149 is a peak current mode step-down converter. The
current through the P-channel MOSFET (high side) is sensed for
current loop control, as well as short circuit and overload
protection. A fixed slope compensation signal is added to the
sensed current to maintain stability for duty cycles greater than
50%. The peak current mode loop appears as a voltage-
programmed current source in parallel with the output
capacitor.
The output of the voltage error amplifier programs the current
mode loop for the necessary peak switch current to force a
constant output voltage for all load and line conditions. Internal
loop compensation terminates the transconductance voltage
error amplifier output. For the adjustable output, the error
amplifier reference is fixed at 0.6 V.
Soft Start/Enable
Soft start limits the current surge seen at the input and
eliminates output voltage overshoot. When the EN pin is pulled
low, it forces the AAT1149 into a low-power, non-switching
state. The total input current during shutdown is less than 1 µA.
Current Limit and Over-Temperature Protection
For overload conditions, the peak input current is limited. To
minimize power dissipation and stresses under current limit and
short-circuit conditions, switching is terminated after entering
current limit for a series of pulses. Switching is terminated for
seven consecutive clock cycles after a current limit has been
sensed for a series of four consecutive clock cycles.
Thermal protection completely disables switching when internal
dissipation becomes excessive. The junction over-temperature
threshold is 140 °C with 15 °C of hysteresis. Once an over-
temperature or over-current fault condition is removed, the
output voltage automatically recovers.
Under-Voltage Lockout
Internal bias of all circuits is controlled using the IN input.
Under-Voltage Lockout (UVLO) guarantees sufficient VIN bias
and proper operation of all internal circuitry before activation.
Applications Information
Inductor Selection
The step-down converter uses peak current mode control with
slope compensation to maintain stability for duty cycles greater
than 50%. The output inductor value must be selected so the
inductor current down slope meets the internal slope
compensation requirements. Table 4 displays suggested
inductor values for various output voltages.
Manufacturer’s specifications list both the inductor DC current
rating, which is a thermal limitation, and the peak current
rating, which is determined by the saturation characteristics.
The inductor should not show any appreciable saturation under
normal load conditions. Some inductors may meet the peak and
average current ratings yet result in excessive losses due to a
high Direct Current Resistance (DCR). Always consider the
losses associated with the DCR and its effect on the total
converter efficiency when selecting an inductor.
The 1.8 µH CDRH2D09 series inductor from Sumida has a
131 mΩ DCR and a 400 mA saturation current rating. At full
load, the inductor DC loss is 21 mW which gives a 2.8% loss in
efficiency for a 400 mA, 1.8 V output.
Table 4. AAT1149 Suggested Inductor Values For Various Output Voltages
Output Voltage
(V)
Typical Inductor Value
(
µ
H)
1.0 and 1.2 1.0 to 1.2
1.5 and 1.8 1.5 to 1.8
2.5 2.2 to 2.7
3.3 3.3
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Input Capacitor
Select a 4.7 µF to 10 µF X7R or X5R ceramic capacitor for the
input. To estimate the required input capacitor size, determine
the acceptable input ripple level (VPP) and solve for C. The
calculated value varies with input voltage and is a maximum
when VIN is double the output voltage.
S
OUT
PP
IN
OUT
IN
OUT
IN fESR
I
V
V
V
1
V
V
C
×
−
−×
=
4
1
V
V
1
V
V
IN
OUT
IN
OUT =
−×
for
OUTIN
V2V ×=
S
OUT
PP
)MIN(IN f4ESR
I
V
1
C
××
−
=
Where fS is the switching frequency. Always examine the
ceramic capacitor DC voltage coefficient characteristics when
selecting the proper value. For example, the capacitance of a
10 µF, 6.3 V, X5R ceramic capacitor with 5.0 VDC applied is
actually about 6 µF.
The maximum input capacitor RMS current is:
−××=
IN
OUT
IN
OUT
OUTRMS V
V
1
V
V
II
The input capacitor RMS ripple current varies with the input and
output voltage and always is less than or equal to half of the
total DC load current.
( )
2
1
5.0D1D
V
V
1
V
V2
IN
OUT
IN
OUT ==−×=
−×
for VIN = 2 × VOUT
2
I
IOUT
MAXRMS =
）（
The term
−×
IN
OUT
IN
OUT
V
V
1
V
V
appears in both the input voltage
ripple and input capacitor RMS current equations and is a
maximum when VOUT is twice VIN. This is why the input voltage
ripple and the input capacitor RMS current ripple are a
maximum at 50% duty cycle.
The input capacitor provides a low impedance loop for the
edges of pulsed current drawn by the AAT1149. Low Equivalent
Series Resistance/Equivalent Series Inductance (ESR/ESL) X7R
and X5R ceramic capacitors are ideal for this function. To
minimize stray inductance, the capacitor should be placed as
closely as possible to the AAT1149. This keeps the high
frequency content of the input current localized, minimizing EMI
and input voltage ripple.
The proper placement of the input capacitor (C2) can be seen in
the Evaluation Board layout in Figure 32.
A laboratory test set-up typically consists of two long wires
running from the bench power supply to the Evaluation Board
input voltage pins. The inductance of these wires, along with
the low ESR ceramic input capacitor, can create a high-Q
network that may affect converter performance. This problem
often becomes apparent in the form of excessive ringing in the
output voltage during load transients. Errors in the loop phase
and gain measurements can also result.
Since the inductance of a short PCB trace feeding the input
voltage is significantly lower than the power leads from the
bench power supply, most applications do not exhibit this
problem.
In applications where the input power source lead inductance
cannot be reduced to a level that does not affect the converter
performance, a high ESR tantalum or aluminum electrolytic
should be placed in parallel with the low ESR, ESL bypass
ceramic. This dampens the high-Q network and stabilizes the
system.
Output Capacitor
The output capacitor limits the output ripple and provides
holdup during large load transitions. A 4.7 µF to 10 µF X5R or
X7R ceramic capacitor typically provides sufficient bulk
capacitance to stabilize the output during large load transitions
and has the ESR and ESL characteristics necessary for low
output ripple.
The output voltage droop due to a load transient (∆ILOAD) is
dominated by the capacitance of the ceramic output capacitor.
During a step increase in load current, the ceramic output
capacitor alone supplies the load current until the loop
responds. Within two or three switching cycles, the loop
responds and the inductor current increases to match the load
current demand. The relationship of the output voltage droop
during the three switching cycles to the output capacitance can
be estimated by:
SDROOP
LOAD
OUT
fV I3
C×
×
=
∆
Once the average inductor current increases to the DC load
level, the output voltage recovers. The above equation
establishes a limit on the minimum value for the output
capacitor with respect to load transients.
The internal voltage loop compensation also limits the minimum
output capacitor value to 4.7 µF. This is due to its effect on the
loop crossover frequency (bandwidth), phase margin, and gain
margin. Increased output capacitance reduces the crossover
frequency with greater phase margin.
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The maximum output capacitor RMS ripple current is given by:
( )
)MAX(INS
OUT)MAX(INOUT
MAXRMS VfL
VVV
32
1
I××
−×
×=
）（
Dissipation due to the RMS current in the ceramic output
capacitor ESR is typically minimal, resulting in less than a few
degrees rise in hot-spot temperature.
Feedback Resistor Selection
Resistors R1 and R2 in Figure 31 program the output to regulate
at a voltage higher than 0.6 V. To limit the bias current required
for the external feedback resistor string while maintaining good
noise immunity, the minimum suggested value for R2 is 59 kΩ.
Although a larger value can further reduce quiescent current, it
also increases the impedance of the feedback node, making it
more sensitive to external noise and interference.
Table 5 summarizes the resistor values for various output
voltages with R2 set to either 59 kΩ for good noise immunity or
121 kΩ for reduced no-load input current.
ΩΩ
k5.88k591
V6.0 V5.1
2R1
V
V
R1
REF
OUT
=×
−=×
−=
The AAT1149, combined with an external feed-forward
capacitor (C3 in Figure 31), delivers enhanced transient
response for extreme pulsed load applications. The addition of
the feed-forward capacitor typically requires a larger output
capacitor C1 for stability.
Table 5. Feedback Resistor Values
VOUT (V)
R1 (k
Ω
)
(R2 = 59 k
Ω
)
R1 (k
Ω
)
(R2 = 121 k
Ω
)
1.00 39.2 80.6
1.10 49.9 100
1.20 59.0 121
1.30 68.1 140
1.40 78.7 162
1.50 88.7 182
1.80 118 243
1.85 124 255
2.00 137 280
2.50 187 383
3.30 267 549
Thermal Calculations
There are three types of losses associated with the AAT1149
step-down converter: conduction losses, switching losses, and
quiescent current losses. Conduction losses are associated with
the RDS(ON) characteristics of the power output switching
devices. Switching losses are dominated by the gate charge of
the power output switching devices. At full load, assuming
Continuous Conduction Mode (CCM), a simplified form of the
losses is given by:
[ ]
( )
()
INQOUTSSW
IN
OUTINL
)ON(DSOUTH)ON(DS
2
OUT
TOTAL
VIIft V
V
VRVR
I
P
×+××+
−×+××
=
IQ is the step-down converter quiescent current. The term tSW is
used to estimate the full load step-down converter switching
losses.
For the condition where the step-down converter is in dropout
at 100% duty cycle, the total device dissipation reduces to:
INQH)ON(DS
2
OUTTOTAL
VIRIP ×+×=
Since RDS(ON), quiescent current, and switching losses all vary
with input voltage, the total losses should be investigated over
the complete input voltage range.
Given the total losses, the maximum junction temperature can
be derived from the θJA for the SC70JW-8 package, which is
160 °C/W.
AJATOTALJ(MAX)
TPT +×=
θ
Layout
The suggested PCB layout for the AAT1149 is shown in
Figure 32. The following guidelines should be used to help
ensure a proper layout.
1. The input capacitor (C2) should connect as closely as
possible to IN (pin 3) and PGND (pins 6, 7, and 8).
2. C1 and L1 should be connected as closely as possible. The
connection of L1 to the LX pin should be as short as
possible.
3. The feedback trace or FB pin (pin 2) should be separate from
any power trace and connect as closely as possible to the
load point. Sensing along a high-current load trace degrades
DC load regulation. If external feedback resistors are used,
they should be placed as closely as possible to the FB pin
(pin 2) to minimize the length of the high impedance
feedback trace.
4. The resistance of the trace from the load return to PGND
(pins 6, 7, and 8) should be kept to a minimum. This helps to
minimize any error in DC regulation due to differences in the
potential of the internal signal ground and the power ground.
5. A pad thickness of less than 1 mm is recommended to
achieve higher stand-off. A high density, small footprint
layout can be achieved using an inexpensive, miniature, non-
shielded, high DCR inductor, as shown in Figure 30.
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Evaluation Board Description
The AAT1149 Evaluation Board schematic diagram is provided
in Figure 31. The PCB layer details are shown in Figure 32.
Figure 30. Minimum Evaluation Board Footprint
Using 2.0 × 1.25 × 1.0 mm Inductor
Figure 31. AAT1149 Evaluation Board Schematic
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Figure 32. AAT1149 Evaluation Board Layer Details
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Step-Down Converter Design Example
Specifications
VOUT = 1.8 V @ 400 mA, pulsed load ∆ILOAD = 300 mA
VIN = 2.7 V to 4.2 V (3.6 V nominal)
fS = 3.0 MHz
TA = 85 °C
1.8 V Output Inductor
H8.1V8.1
A
s
1V
A
s
1L1
OUT
µ
µµ
=×=×=
For Taiyo Yuden inductor CBC2518T2R2M, 2.2 µH, DCR = 130 mΩ.
mA156
V2.4 V8.1
1
MHz0.3H2.2 V8.1
V
V
1
f1LV
I
IN
OUT
S
OUT
L1
=
−×
×
=
−×
×
=
µ
∆
A478.0A078.0V4.0
2
I
II
1L
OUTPKL1
=+=+=
∆
Where, IPKL1 is the peak current on L1.
mW21m130A4.0DCRIP
22
OUTL1
=×=×=
Ω
1.8V Output Capacitor
VDROOP = 0.1 V
F7.4use,H0.3
MHz0.3V1.0 A3.03
fV I3
C
SDROOP
LOAD
OUT
µµ
∆
=
×
×
=
×
×
==
( )
( )
mArms45
V2.4MHz0.3H
2.2 V8.1V2.4V8.1
32
1
Vf1L
VVV
3
2
1
I
)MAX(INS
OUT)MAX(INOUT
RMS =
××
−×
×=
××
−×
×=
µ
W10)mA45(m5IESRP22
RMSESR
µΩ
=×=×=
Input Capacitor
Input Ripple VPP = 25 mV
F2.2useF45.1
MHz0.3m5
0.4A
mV25
4
1
fESR
I
V
4
1
C
S
OUT
PP
IN
µµ
Ω
，
=
×
−×
=
×
−×
=
Arms2.0
2
I
I
OUT
RMS
==
mW2.0)A2.0(m5IESRP22
RMS =×=×=
Ω
AAT1149 Losses
[ ]
( ) ( )
INQOUTSSW
IN
OUTINL)ON(DSOUTH)ON(DS
2
OUT
TOTAL
VIIft
V
VVRV
RI
P×+××+
−×+××
=
[ ]
( )
( )
mW140V2.4A70A4.0MHz3ns5
V2.4 V8.1V2.47.0V8.1725.04.0 2=×+××+
−×+××
=
µ
ΩΩ
( )
C107C85mW140W/C160TPT
AJALOSSJ(MAX)
°=°+×°=+×=
θ
PRELIMINARY DATA SHEET • AAT1149 3 MHZ FAST TRANSIENT 400 MA STEP-DOWN CONVERTER
Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • sales@skyworksinc.com • www.skyworksinc.com
16 July 1, 2013 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice • 201987C
Table 6 summarizes the feedback resistor values for various
output voltages. Tables 7 and 8 list the typical surface mount
inductors and surface mount capacitors.
Table 6. Feedback Resistor Values
VOUT (V) R1 (k
Ω
)
(R2 = 59 k
Ω
)
R1 (k
Ω
)
(R2 = 121 k
Ω
)
L1 (
µ
H)
1.0 39.2 80.6 1.0
1.2 59.0 121 1.2
1.5 88.7 182 1.5
1.8 118 243 1.8
2.5 187 383 2.2
3.3 267 549 3.3
Table 7. Typical Surface Mount Inductors
Manufacturer Part Number/Type Inductance (
µ
H) Rated Current (mA) DCR (
Ω
) Size (mm) L
×
W
×
H
Taiyo Yuden
BRC1608 1.0 520 180 0603
(height = 1 mm)
1.5 410 300
BRL2012
1.5 600 200
0805
(height = 1 mm)
2.2 550 250
3.3 450 350
CBC2518
Wire wound chip
1.0 1000 80
2.5×1.8×1.8
2.2 890 130
Sumida CDRH2D09
Shielded
1.2 590 97.5
3.2×3.2×1.0
1.5 520 110
1.8 480 131
2.5 440 150
3.0 400 195
Murata LQH2MCN4R7M02
Unshielded
1.0 485 300
2.0×1.6×0.95
1.5 445 400
2.2 425 480
3.3 375 600
Coiltronics SD3118
Shielded
1.2 720 75
3.15×3.15×1.2
1.5 630 104
2.2 510 116
3.3 430 139
Table 8. Typical Surface Mount Capacitors
Manufacturer Part Number Value (
µ
F) Voltage (V) Temperature Coefficient Case
Murata GRM219R61A475KE19 4.7 10 X5R 0805
Murata GRM21BR60J106KE19 10 6.3 X5R 0805
Murata GRM185R60J475M 4.7 6.3 X5R 0603
PRELIMINARY DATA SHEET • AAT1149 3 MHZ FAST TRANSIENT 400 MA STEP-DOWN CONVERTER
Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • sales@skyworksinc.com • www.skyworksinc.com
201987C • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice • July 1, 2013 17
Package Information
Package dimensions and shown in Figure 33, and tape and reel
dimensions are provided in Figure 34.
Figure 33. AAT1149 8-pin SC70JW Package Dimensions
Figure 34. AAT1149 Carrier Tape Dimensions
PRELIMINARY DATA SHEET • AAT1149 3 MHZ FAST TRANSIENT 400 MA STEP-DOWN CONVERTER
Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • sales@skyworksinc.com • www.skyworksinc.com
18 July 1, 2013 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice • 201987C
Ordering Information
Model Name Manufacturing Part Number (Note 1) Evaluation Board Part Number
AAT1149 Fast Transient Step-Down Converter AAT1149IJS-0.6-T1 AAT1149IJS-0.6-EVB
Note 1: Sample stock is generally held on the part number listed in BOLD.
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