AAT2550178
Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
2550.2008.02.1.3 1
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General Description
The AAT2550 is a fully integrated total power solution
with two step-down converters plus a single-cell lithium-
ion / polymer battery charger. The step-down converter
input voltage range spans 2.7V to 5.5V, making the
AAT2550 ideal for systems powered by single-cell lithi-
um-ion/polymer batteries.
The battery charger is a complete constant current/ con-
stant voltage linear charger. It offers an integrated pass
device, reverse blocking protection, high current accu-
racy and voltage regulation, charge status, and charge
termination. The charging current is programmable via
external resistor from 100mA to 1A. In addition to these
standard features, the device offers over-voltage, over-
current, and thermal protection.
The two step-down converters are highly integrated,
operating at a switching frequency of 1.4MHz, minimiz-
ing the size of external components while keeping
switching losses low. Each converter has independent
input, enable, and feedback pins. The output voltage
ranges from 0.6V to VIN. Each converter is capable of
delivering up to 600mA of load current.
The AAT2550 is available in a Pb-free, space-saving,
thermally-enhanced QFN44-24 package and is rated
over the -40°C to +85°C temperature range.
Features
• Two Step-Down Converters:
▪ 600mA Output Current per Converter
▪ V
IN Range: 2.7V to 5.5V
▪ 1.4MHz Switching Frequency
▪ Low RDS(ON) 0.4Ω Integrated Power Switches
▪ Internal Soft Start
▪ 27μA Quiescent Current per Converter
• Highly Integrated Battery Charger:
▪
Programmable Charging Current from 100mA to 1A
▪ Pass Device
▪ Reverse Blocking Diodes
▪ Current Sensing Resistor
▪ Digital Thermal Regulation
• Short-Circuit, Over-Temperature, and Current Limit
Protection
• QFN44-24 Package
• -40°C to +85°C Temperature Range
Applications
• Cellular Telephones
• Digital Cameras
• Handheld Instruments
• MP3, Portable Music, and Portable Media Players
• PDAs and Handheld Computers
Typical Application
AAT2550
Adapter
Serial Interface
STAT1
STAT2
ADPSET
DATA
ENBAT
INA
INB
Li-Ion Battery or
Adapter
BAT
TS
CT
Batt+
Batt-
Temp
Battery Pack
LXA
C
OUTA
FBA
V
OUTA
R
SET
LXB
C
OUTB
FBB
V
OUTB
ENA
ENB
GND
ADP
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Total Power Solution for Portable ApplicationsSystemPowerTM
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Pin Descriptions
Pin # Symbol Function
1 ENA Enable pin for Converter A. When connected to logic low, it disables the step-down converter and con-
sumes less than 1μA of current. When connected to logic high, the converter operates normally.
2 LXA Power switching node for Converter A. Connect the inductor to this pin. Internally, it is connected to the
drain of both high- and low-side MOSFETs.
3, 17 PGND Power ground. Connect the PGND pins together as close to the IC as possible. Connect AGND to PGND at a
single point as close to the IC as possible.
4 DATA Status report to the microcontroller via serial interface (open drain).
5, 7 N/C Not connected.
6 ADPSET Charge current set point. Connect a resistor from this pin to ground. Refer to Typical Characteristics curves
for resistor selection.
8 BAT Battery charging and sensing. Connect the positive terminal of the battery to BAT.
9 ADP Input for adapter charger.
10, 11, 22 AGND Analog signal ground. Connect AGND to PGND at a single point as close to the IC as possible.
12 ENBAT Enable pin for the battery charger. When connected to logic low, the battery charger is disabled and con-
sumes less than 1μA of current. When connected to logic high, the charger operates normally.
13 TS Temperature sense input. Connect to a 10kΩ NTC thermistor.
14 STAT2 Battery charge status indicator pin to drive an LED. It is an open drain input.
15 STAT1 Battery charge status indicator pin to drive an LED. It is an open drain input.
16 CT Timing capacitor to adjust internal watchdog timer. Sets maximum charge time for adapter powered
trickle, constant current, and constant voltage charge modes.
18 LXB Power switching node for Converter B. Connect the inductor to this pin. Internally, it is connected to the
drain of both high- and low-side MOSFETs.
19 ENB Enable pin for Converter B. When connected to logic low, it disables the step-down converter and con-
sumes less than 1μA of current. When connected to logic high, the converter operates normally.
20 INB Input voltage for Converter B.
21 FBB Output voltage feedback input for Converter B. FBB senses the output voltage B for regulation control. The
FBB regulation threshold is 0.6V. A resistive voltage divider is connected to the output B, FBB, and AGND.
23 FBA Output voltage feedback input for Converter A. FBA senses the output voltage A for regulation control. The
FBA regulation threshold is 0.6V. A resistive voltage divider is connected to the output A, FBA, and AGND.
24 INA Input voltage for Converter A.
EP Exposed paddle; connect to ground directly beneath the package.
Pin Configuration
QFN44-24
(Top View)
1
3
2
5
4
6
18
16
17
14
15
13
10
11
12
8
9
7
19
21
20
23
22
24
ENA
LXA
PGND
DATA
N/C
A
DPSET
N/C
BAT
ADP
AGND
AGND
ENBA
T
STAT1
CT
STAT2
TS
PGND
LXB
ENB
INB
FBB
A
GND
FBA
INA
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Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
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Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
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Absolute Maximum Ratings1
Symbol Description Value Units
VINA/B, VADP INA, INB, and ADP Voltages to GND -0.3 to 6.0 V
VLXA/B, VFBA/B VLXA, VLXB, VFBA, and VFBB to GND -0.3 to VINA/B, VADP + 0.3 V
VXVoltage on All Other Pins to GND -0.3 to 6.0 V
TJOperating Junction Temperature Range -40 to 150 °C
TLEAD Maximum Soldering Temperature (at leads, 10 sec) 300 °C
Thermal Information
Symbol Description Value Units
PD Maximum Power Dissipation 2.0 W
θJA Thermal Resistance250 °C/W
1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at conditions other than the operating conditions
specified is not implied. Only one Absolute Maximum Rating should be applied at any one time.
2. Mounted on an FR4 printed circuit board.
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Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
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Total Power Solution for Portable ApplicationsSystemPowerTM
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Electrical Characteristics1
VIN = 3.6V; TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = 25°C.
Symbol Description Conditions Min Typ Max Units
Step-Down Converters A and B
VIN Input Voltage 2.7 5.5 V
VUVLO Under-Voltage Lockout Threshold
VIN Rising 2.7 V
Hysteresis 100 mV
VIN Falling 1.8 V
VOUT Output Voltage Tolerance IOUT = 0 to 600mA, VIN = 2.7V to 5.5V -3.0 3.0 %
VOUT Output Voltage Range 0.6 VIN V
IOUT Output Current Per Converter 600 mA
IQQuiescent Current Each Converter 27 70 μA
ISHDN Shutdown Current VENA = VENB = GND 1.0 μA
ILIM P-Channel Current Limit Each Converter 0.8 1.0 A
ILX_LEAK LX Leakage Current VIN = 5.5V, VLX = 0 to VIN, VENA = VENB = GND 1.0 μA
IFB_LEAK Feedback Leakage VFB = 0.6V 0.2 μA
RFB FB Impedance VOUT > 0.6V 250 kΩ
VFB
Feedback Threshold Voltage Accuracy
(0.6V Adjustable Version) No Load, TA = 25°C 0.591 0.6 0.609 V
RDS(ON)H High-Side Switch On Resistance 0.45 Ω
RDS(ON)L Low-Side Switch On Resistance 0.40 Ω
ΔVLineReg Line Regulation VIN = 2.7V to 5.5V 0.1 %/V
FOSC Switching Frequency 1.4 MHz
TSD Over-Temperature Shutdown Threshold 140 °C
THYS Over-Temperature Shutdown Hysteresis 15 °C
VEN(L) Enable Threshold Low 0.6 V
VEN(H) Enable Threshold High 1.4 V
IEN Input Low Current VIN = VFB = 5.5V -1.0 1.0 μA
1. The AAT2550 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured by design, characterization, and correla-
tion with statistical process controls.
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Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
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Total Power Solution for Portable ApplicationsSystemPowerTM
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Electrical Characteristics1 (continued)
VADP = 5V; TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = 25°C.
Symbol Description Conditions Min Typ Max Units
Battery Charger
VADP Adapter Voltage Range 4.0 5.5 V
VUVLO
Under-Voltage Lockout Rising Edge 3.0 V
UVLO Hysteresis 150 mV
IQQuiescent Current ICHARGE = 100mA 0.75 3.0 mA
ISLEEP Sleep Mode Current VBAT = 4.25V 0.3 1.0 μA
ILEAKAGE Reverse Leakage Current VBAT = 4V, ADP Pin Open 1.0 μA
ISHDN Shutdown Current VEN = GND 1.0 μA
VBAT_EOC2End of Charge Voltage Accuracy 4.158 4.2 4.242 V
ΔVCH/VCH Output Charge Voltage Tolerance 0.5 %
VMIN Preconditioning Voltage Threshold 2.80 3.0 3.15 V
VRCH Battery Recharge Voltage Threshold VBAT_EOC - 0.1 V
ICH Charge Current 100 1000 mA
ΔICH/ICH Charge Current Regulation Tolerance 10 %
VADPSET ADPSET Pin Voltage Constant Current Mode 2.0 V
KIA Current Set Factor: ICH/IADPSET 4000
RDS(ON) Charger Pass Device VIN = 5.5V 0.20 0.25 0.35 Ω
TCConstant Current Mode Time-Out CT = 100nF, VADP = 5.5V 3.0 Hour
TPPreconditioning Time-Out CT = 100nF, VADP = 5.5V 25 Minute
TVConstant Voltage Mode Time-Out CT = 100nF, VADP = 5.5V 3.0 Hour
VSTAT Output Low Voltage ISINK = 4mA 0.4 V
ISTAT STAT Sink Current 8.0 mA
VOVP Over-Voltage Protection 4.4 V
ITK/ICH Preconditioning (Trickle Charge) Current 10 %
ITERM/ICH Charge Termination Threshold Current 7.5 %
ITS Current Source from TS Pin 70 80 90 μA
TS1TS Hot Temperature Fault Threshold 310 330 350 mV
Hysteresis 15
TS2TS Cold Temperature Fault Threshold 2.2 2.3 2.4 V
Hysteresis 10 mV
IDATA DATA Pin Sink Current DATA Pin is Active Low 3.0 mA
VDATA(H) Input High Threshold 1.6 V
VDATA(L) Input Low Threshold 0.4 V
SQPULSE Status Request Pulse Width 200 ns
TPeriod System Clock Period 50 μs
FDATA Data Output Frequency 20 kHz
TREG Thermal Loop Regulation 90 °C
TLOOP_IN Thermal Loop Entering Threshold 110 °C
TLOOP_OUT Thermal Loop Exiting Threshold 85 °C
TSD Over-Temperature Shutdown Threshold 145 °C
1. The AAT2550 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured by design, characterization, and correla-
tion with statistical process controls.
2. End of Charge Voltage Accuracy is specified over the 0° to 70°C ambient temperature range.
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Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
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Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
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Typical Characteristics — Step-Down Converter
Efficiency vs. Load
(VOUT = 1.8V; L = 4.7μ
μ
H)
Output Current (mA)
Efficiency (%)
50
60
70
80
90
100
0.1 1 10 100 100
0
VIN = 2.7V
VIN = 3.6V VIN = 4.2V
DC Regulation
(VOUT = 1.8V)
Output Current (mA)
Output Error (%)
-1.0
-0.5
0.0
0.5
1.0
0.1 1 10 100 1000
VIN = 4.2V
VIN = 3.6V
VIN = 2.7V
Efficiency vs. Load
(VOUT = 2.5V; L = 6.8μ
μ
H)
Output Current (mA)
Efficiency (%)
50
60
70
80
90
100
0.1 1 10 100 1000
VIN = 5.0V
VIN = 3.6V
VIN = 4.2V
VIN = 2.7V
DC Regulation
(VOUT = 2.5V)
Output Current (mA)
Output Error (%)
-1.0
-0.5
0.0
0.5
1.0
0.1 1 10 100 1000
VIN = 5.0V
VIN = 3.6V
VIN = 3.0V
VIN = 4.2V
Efficiency vs. Load
(VOUT = 3.3V; L = 6.8μ
μ
H)
Output Current (mA)
Efficiency (%)
50
60
70
80
90
100
0.1 1 10 100 1000
VIN = 3.6V
VIN = 4.2V
VIN = 5.0V
DC Regulation
(VOUT = 3.3V; L = 6.8µH)
Output Current (mA)
Output Error (%)
-1.0
-0.5
0.0
0.5
1.0
0.1 1 10 100 1000
VIN = 5.0V
VIN = 4.2V
VIN = 3.6V
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Total Power Solution for Portable ApplicationsSystemPowerTM
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Typical Characteristics — Step-Down Converter (continued)
Soft Start
(VIN = 3.6V; VOUT = 1.8V; IOUT = 400mA)
Time (100μ
μ
s/div)
Enable and Output Voltage
(top) (V)
Inductor Current
(bottom) (A)
0.0
1.0
2.0
3.0
4.0
5.0
-0.4
-0.2
0.0
0.2
0.4
0.6
VEN
IL
VO
Line Regulation
(VOUT = 1.8V)
Input Voltage (V)
Accuracy (%)
-0.40
-0.30
-0.20
-0.10
0.00
0.10
0.20
0.30
0.40
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0
IOUT = 10mA
IOUT = 400mA
IOUT = 1mA
Output Voltage Error vs. Temperature
(VIN = 3.6V; VO = 1.8V; IOUT = 400mA)
Temperature (°
°
C)
Output Error (%)
-2.0
-1.0
0.0
1.0
2.0
-40 -20 0 20 40 60 80 100
Switching Frequency vs. Temperature
(VIN = 3.6V; VOUT = 1.8V)
Temperature (°
°
C)
Variation (%)
-15.0
-12.0
-9.0
-6.0
-3.0
0.0
3.0
6.0
9.0
12.0
15.0
-40 -20 0 20 40 60 80 100
Frequency vs. Input Voltage
Input Voltage (V)
Frequency Variation (%)
-4.0
-3.0
-2.0
-1.0
0.0
1.0
2.0
2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5
VOUT = 1.8V
VOUT = 2.5V VOUT = 3.3V
No Load Quiescent Current vs. Input Voltage
Input Voltage (V)
Supply Current (μ
μ
A)
10
15
20
25
30
35
40
45
50
2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5
85°C25°C
-40°C
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Typical Characteristics — Step-Down Converter (continued)
P-Channel RDS(ON) vs. Input Voltage
Input Voltage (V)
RDS(ON)H (mΩ
Ω
)
300
350
400
450
500
550
600
650
700
750
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0
25°C
120°C 100°C
85°C
N-Channel RDS(ON) vs. Input Voltage
Input Voltage (V)
RDS(ON)L (mΩ
Ω
)
300
350
400
450
500
550
600
650
700
750
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0
25°C
120°C100°C
85°C
Load Transient Response
(1mA to 300mA; VIN = 3.6V; VOUT = 1.8V;
COUT = 10µF; CFF = 100pF)
Output Voltage
(top) (V)
Load and Inductor Current
(200mA/div) (bottom)
Time (50µs/div)
1.7
1.8
1.9
2.0
VO
300mA
1mA
0
IO
IL
Load Transient Response
(300mA to 400mA; VIN = 3.6V;
VOUT = 1.8V; COUT = 4.7µF)
Output Voltage
(top) (V)
Load and Inductor Current
(100mA/div) (bottom)
Time (50µs/div)
1.75
1.80
1.85
1.90
0.1
0.2
0.3
0.4
VO
IO
IL
400mA
300mA
Load Transient Response
(300mA to 400mA; VIN = 3.6V;
VOUT = 1.8V; COUT = 10µF)
Output Voltage
(top) (V)
Load and Inductor Current
(100mA/div) (bottom)
Time (50µs/div)
1.75
1.80
1.85
1.90
0.1
0.2
0.3
0.4
VO
IO
IL
400mA
300mA
Load Transient Response
(300mA to 400mA; VIN = 3.6V; VOUT = 1.8V;
COUT = 10µF; CFF = 100pF)
Output Voltage
(top) (V)
Load and Inductor Current
(100mA/div) (bottom)
Time (50µs/div)
1.775
1.800
1.825
1.850
0.1
0.2
0.3
0.4
VO
IO
IL
400mA
300mA
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Typical Characteristics — Step-Down Converter (continued)
Line Response
(VOUT = 1.8V @ 400mA)
Output Voltage
(top) (V)
Input Voltage
(bottom) (V)
Time (25µs/div)
1.80
1.81
1.82
3.0
3.5
4.0
4.5
Output Ripple
(VIN = 3.6V; VOUT = 1.8V; IOUT = 1mA)
Time (10µs/div)
Output Voltage (AC coupled)
(top) (mV)
Inductor Current
(bottom) (A)
-20
0
20
40
-0.10
-0.05
0.00
0.05
0.10
0.15
VO
IL
Output Ripple
(VIN = 3.6V; VOUT = 1.8V; IOUT = 400mA)
Time (500ns/div)
Output Voltage (AC coupled)
(top) (mV)
Inductor Current
(bottom) (A)
-20
0
20
40
0.1
0.2
0.3
0.4
0.5
0.6
VO
IL
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Typical Characteristics — Battery Charger
Constant Charging Current vs. RSET
RSET (kΩ
Ω
)
ICH (mA)
10
100
1000
10000
1 10 100
Battery Voltage vs. Supply Voltage
Supply Voltage (V)
VBAT (V)
4.158
4.179
4.200
4.221
4.242
4.5 4.75 5.0 5.25 5.5
End of Charge Voltage Regulation
vs. Temperature
Temperature (°
°
C)
VBAT_EOC (V)
4.158
4.179
4.200
4.221
4.242
-50 -25 0 25 50 75 100
Preconditioning Threshold
Voltage vs. Temperature
Temperature (°
°
C)
VMIN (V)
2.95
2.96
2.97
2.98
2.99
3.00
3.01
3.02
3.03
3.04
3.05
-50 -25 0 25 50 75 100
Preconditioning Current vs. Temperature
(ADPSET = 8.06kΩ
Ω
)
Temperature (
°
C)
ITK (mA)
80
90
100
110
120
-50 -25 0 25 50 75 100
Constant Charging Current vs. Temperature
(ADPSET = 8.06kΩ
Ω
)
Temperature (
°
C)
ICH (mA)
900
920
940
960
980
1000
1020
1040
1060
1080
1100
-50 -25 0 25 50 75 100
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Typical Characteristics — Battery Charger (continued)
Charging Current vs. Battery Voltage
(ADPSET = 8.06kΩ
Ω
; VIN = 5.0V)
Battery Voltage (V)
ICH (A)
0.0
0.2
0.4
0.6
0.8
1.0
1.2
2.5 2.9 3.3 3.7 4.1 4.5
Constant Charging Current vs. Input Voltage
(ADPSET = 8.06kΩ
Ω
)
Input Voltage (V)
ICH (mA)
0
200
400
600
800
1000
1200
4.5 4.75 5.0 5.25 5.5 5.75 6.0
VBAT = 3.3V
VBAT = 3.5V
VBAT = 3.9V
VIH vs. Input Voltage
EN Pin (Rising)
Input Voltage (V)
VIH (V)
0.4
0.5
0.6
0.7
0.8
0.9
1.0
1.1
1.2
1.3
1.4
4.2 4.4 4.6 4.8 5.0 5.2 5.4 5.6 5.8 6.0
-40°C +25°C
+85°C
VIL vs. Input Voltage
EN Pin (Falling)
Input Voltage (V)
VIH (V)
0.4
0.5
0.6
0.7
0.8
0.9
1.0
1.1
1.2
1.3
1.4
4.2 4.4 4.6 4.8 5.0 5.2 5.4 5.6 5.8 6.0
-40°C +25°C
+85°C
Adapter Mode Supply Current
vs. ADPSET Resistor
ADPSET Resistor (kΩ
Ω
)
IQ (mA)
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0.80
1 10 100 1000
Pre-Conditioning
Constant Current
Counter Timeout vs. Temperature
(CT = 0.1μ
μ
F)
Temperature (
°
C)
Counter Timeout (%)
-10
-8
-6
-4
-2
0
2
4
6
8
10
-50 -25 0 25 50 75 100
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Typical Characteristics — Battery Charger (continued)
CT Pin Capacitance vs. Counter Timeout
Time (hours)
Capacitance (μ
μ
F)
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
0246810
Precondition Timeout
Precondition + Constant Current Timeout
or Constant Voltage Timeout
Temperature Sense Output Current
vs. Temperature
Temperature (°
°
C)
TS Pin Current (
μ
A)
72
74
76
78
80
82
84
86
88
-50 -25 0 25 50 75 10
0
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Functional Description
The AAT2550 is a highly integrated power management
IC comprised of a battery charger and two step-down
voltage converters. The battery charger is designed for
charging single-cell lithium-ion / polymer batteries.
Featuring an integrated pass device and reverse block-
ing, it offers a constant current / constant voltage charge
algorithm with a user-programmable charge current
level. The two step-down converters have been designed
to minimize external component size and maximize effi-
ciency over the entire load range. Each converter has
independent enable and input voltage pins and can pro-
vide 600mA of load current.
Battery Charger
The battery charger is designed to operate with standard
AC adapter input sources, while requiring a minimum
number of external components. It precisely regulates
charge voltage and current for single-cell lithium-ion /
polymer batteries.
The adapter charge input constant current level may be
programmed up to 1A for rapid charging applications.
The battery charger features thermal loop charge reduc-
tion. In the event of operating ambient temperatures
exceeding the power dissipation abilities of the device
package for a given constant current charge level, the
Functional Block Diagram
ENB
LXB
Err.
Amp.
DH
DL
PGND
FBB
Voltage
Reference
Control
Logic
Logic
ENA
LXA
Err.
Amp.
DH
DL
PGND
FBA
Voltage
Reference
Control
Logic
Logic
INB
Charge
Control
Reverse Blocking
CV/Pre-
Charge
Constant
Current
Current
Compare
ADP BAT
OTP
Charge
Status
STAT2
STAT1
4.2V
ENBAT
A
DPSET
TS
INA
Window
Comparator
80μA
Watchdog
Timer
CT
UVLO
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charge control will enter into thermal regulation. When
the system thermal regulation becomes active, the pro-
grammed constant current charge amplitude will auto-
matically decrease to a safe level for the present operat-
ing conditions. If the ambient temperature drops to a
level sufficient to allow the device to come out of thermal
regulation, then the system will automatically resume
charging at the full programmed constant current level.
This intelligent thermal management system permits the
battery charger to operate and charge a battery cell
safely over a wide range of ambient conditions, while
maximizing the greatest possible charge current and
minimizing the battery charge time for a given set of
conditions.
Status monitor output pins are provided to indicate the
battery charge state by directly driving two external
LEDs. A serial interface output is also available to report
any one of 12 distinct charge states to the host system
microcontroller / microprocessor. Battery temperature
and charge state are fully monitored for fault conditions.
In the event of an over-voltage or over-temperature
condition, the device will automatically shut down, pro-
tecting the charging device, control system, and the bat-
tery under charge. In addition to internal charge control-
ler thermal protection, the charger also offers a tem-
perature sense feedback function (TS pin) from the
battery to shut down the device in the event the battery
exceeds its own thermal limit during charging. All fault
events are reported to the user either by simple status
LEDs or via the DATA pin function.
Charging Operation
As shown in Figure 1, there are three basic phases for
the battery charge cycle:
1. Pre-conditioning / trickle charge
2. Constant current / fast charge
3. Constant voltage charge
Battery Preconditioning
Before the start of charging, the charger checks several
conditions in order to assure a safe charging environ-
ment. The input supply must be above the minimum
operating voltage, or under-voltage lockout threshold
(VUVLO), for the charging sequence to begin. Also, the
battery temperature, as reported by a thermistor con-
nected to the TS pin from the battery, must be within the
proper window for safe charging. When these conditions
have been met and a battery is connected to the BAT
pin, the charger checks the state of the battery. If the
battery voltage is below the preconditioning voltage
threshold (VMIN), then the charge control begins precon-
ditioning the battery. The preconditioning trickle charge
current is equal to the fast charge constant current
divided by 10. For example, if the programmed fast
charge current is 1A, then the preconditioning mode
(trickle charge) current will be 100mA. Battery precon-
ditioning is a safety precaution for deeply discharged
batteries and also helps to limit power dissipation in the
pass transistor when the voltage across the device is at
the greatest potential.
Preconditioning
Trickle Charge
Phase
Constant Current
Charge Phase
Constant Voltage
Charge Phase
Charge Complete Voltage
Constant Current Mode
Voltage Threshold
Regulated Current
Trickle Charge and
Termination Threshold
I = CC / 10
I = Max CC
Figure 1: Typical Charge Profile.
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Fast Charge/Constant Current Charging
Battery preconditioning continues until the voltage on
the BAT pin exceeds the preconditioning voltage thresh-
old (VMIN). At this point, the charger begins the constant
current fast charging phase. The fast charge constant
current (ICH) amplitude is programmed by the user via
the RSET resistor. The charger remains in the constant
current charge mode until the battery reaches the volt-
age regulation threshold, VBAT_EOC.
Constant Voltage Charging
The system transitions to a constant voltage charging
mode when the battery voltage reaches the output
charge regulation threshold (VBAT_EOC) during the con-
stant current fast charge phase. The regulation voltage
level is factory programmed to 4.2V (±1%). The charge
current in the constant voltage mode drops as the bat-
tery under charge reaches its maximum capacity.
End of Charge Cycle
Termination and Recharge Sequence
When the charge current drops to 7.5% of the pro-
grammed fast charge current level in the constant volt-
age mode, the device terminates charging and goes into
a sleep state. The charger will remain in a sleep state
until the battery voltage decreases to a level below the
battery recharge voltage threshold (VRCH). When the
input supply is disconnected, the charger will automati-
cally transition into a power-saving sleep mode.
Consuming only an ultra-low 0.3μA in sleep mode, the
charger minimizes battery drain when it is not charging.
This feature is particularly useful in applications where
the input supply level may fall below the battery charge
or under-voltage lockout level. In such cases where the
input voltage drops, the device will enter sleep mode
and resume charging automatically once the input sup-
ply has recovered from the fault condition.
Step-Down Converters
The AAT2550 offers two high-performance, 600mA,
1.4MHz step-down converters. Both converters minimize
external component size and optimize efficiency over the
entire load range. Both converters can be programmed
with external feedback resistors to any voltage ranging
from 0.6V to the input voltage. At dropout, the con-
verter duty cycle increases to 100% and the output volt-
age tracks the input voltage minus the RDS(ON) drop of the
P-channel MOSFET.
Input voltage range is 2.7V to 5.5V and each converter’s
efficiency has been optimized for all load conditions,
ranging from no load to 600mA. The internal error
amplifier and compensation provides excellent transient
response, load regulation, and line regulation. Soft start
eliminates output voltage overshoot when the enable or
the input voltage is applied.
Soft Start / Enable
The internal soft start limits the inrush current during
start-up. This prevents possible sagging of the input
voltage and eliminates output voltage overshoot. Typical
start-up time for a 4.7μF output capacitor and load cur-
rent of 600mA is 100μs.
The AAT2550 offers independent enable pins for each
converter. When connected to logic low, the enable
input forces the respective step-down converter into a
low-power, non-switching, shutdown state. The total
input current during shutdown is less than 1μA for each
channel.
Current Limit and
Over-Temperature Protection
For overload conditions, the peak input current is limit-
ed. To minimize power dissipation and stresses under
current limit and short-circuit conditions, switching is
terminated after entering current limit for a series of
pulses. Switching is terminated for seven consecutive
clock cycles after a current limit has been sensed for a
series of four consecutive clock cycles.
Thermal protection completely disables switching when
internal dissipation becomes excessive. The junction
over-temperature threshold is 140°C with 15°C of hys-
teresis. Once an over-temperature or over-current fault
conditions is removed, the output voltage automatically
recovers.
Under-Voltage Lockout
The under-voltage lockout circuit prevents the device from
improper operation at low input voltages. Internal bias of
all circuits is controlled via the VIN input. Under-voltage
lockout (UVLO) guarantees sufficient VIN bias and proper
operation of all internal circuitry prior to activation.
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System Operation Flow Chart
Yes
Yes
Yes
Yes
Yes
Yes
No
No
No
No
No
No
No
No
No
Set
Enable
Timing
Expire
TERM
Yes
BAT_EOC
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Application Information
AC Adapter Power Charging
The adapter constant current charge levels can be pro-
grammed up to 1A. The AAT2550 will operate from the
adapter input over a 4.0V to 5.5V range.
The constant current fast charge current for the adapter
input mode is set by the RSET resistor connected between
the ADPSET and ground. Refer to Table 1 for recommend-
ed RSET values for a desired constant current charge level.
The precise charging function in the adapter mode may be
read from the DATA pin and/or status LEDs. Please refer
to the Battery Charge Status Indication discussion in this
datasheet for further details on data reporting.
Thermal Loop Control
Due to the integrated nature of the linear charging con-
trol pass device, a special thermal loop control system
has been employed to maximize charging current under
all operation conditions. The thermal management sys-
tem measures the internal circuit die temperature and
reduces the fast charge current when the device exceeds
a preset internal temperature control threshold. Once
the thermal loop control becomes active, the fast charge
current is initially reduced by a factor of 0.44.
The initial thermal loop current can be estimated by the
following equation:
ITLOOP = ICH · 0.44
The thermal loop control re-evaluates the circuit die tem-
perature every three seconds and adjusts the fast charge
current back up in small steps to the full fast charge cur-
rent level or until an equilibrium current is discovered and
maximized for the given ambient temperature condition.
The thermal loop controls the system charge level; there-
fore, the AAT2550 will always provide the highest level of
constant current possible in the fast charge mode for any
given ambient temperature condition.
Adapter Input Charge Inhibit and Resume
The AAT2550 has an under-voltage lockout and power on
reset feature so that the charger will suspend charging
and shut down if the input supply to the adapter pin
drops below the UVLO threshold. When power is re-
applied to the adapter pin or the UVLO condition recov-
ers and ADP > VBAT
, the system charge control will assess
the state of charge on the battery cell and will auto-
matically resume charging in the appropriate mode for
the condition of the battery.
ICH ADP RSET (kΩ)
100 84.5
200 43.2
300 28.0
400 21.0
500 16.9
600 13.3
700 11.5
800 10.2
900 9.09
1000 8.06
Table 1: Resistor Values.
Enable / Disable
The AAT2550 provides an enable function to control the
charger IC on and off. The enable (ENBAT) pin is active
high. When pulled to a logic low level, the AAT2550 will
be shut down and forced into the sleep state. Charging
will be halted regardless of the battery voltage or charg-
ing state. When the device is re-enabled, the charge
control circuit will automatically reset and resume charg-
ing functions with the appropriate charging mode based
on the battery charge state and measured cell voltage.
Programming Charge Current
The fast charge constant current charge level is pro-
grammed with a resistor placed between the ADPSET pin
and ground. The accuracy of the fast charge, as well as
the preconditioning trickle charge current, is dominated
by the tolerance of the set resistor used. For this reason,
1% tolerance metal film resistors are recommended for
the set resistor function.
Fast charge constant current levels from 100mA to 1A
can be set by selecting the appropriate resistor value
from Table 1. The RSET resistor should be connected
between the ADPSET pin and ground.
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RSET (kΩ
Ω
)
ICH (mA)
10
100
1000
10000
11010
0
ADP
Figure 2: Constant Charging Current vs. RSET.
Protection Circuitry
Programmable Watchdog Timer
The AAT2550 contains a watchdog timing circuit for the
adapter input charging mode. Typically, a 0.1μF ceramic
capacitor is connected between the CT pin and ground.
When a 0.1μF ceramic capacitor is used, the device will
time a shutdown condition if the trickle charge mode
exceeds 25 minutes and a combined trickle charge plus
fast charge mode of three hours. When the device tran-
sitions to the constant voltage mode, the timing counter
is reset and will time out after three hours and shut
down the charger (see Table 2).
Mode Time
Trickle Charge (TC) Time Out 25 minutes
Trickle Charge (TC) + Constant Current (CC)
Mode Time Out 3 hours
Constant Voltage (VC) Mode Time Out 3 hours
Table 2: Summary for a 0.1μF Used for the
Timing Capacitor.
The CT pin is driven by a constant current source and
will provide a linear response to increases in the timing
capacitor value. Thus, if the timing capacitor were to be
doubled from the nominal 0.1μF value, the time-out
durations would be doubled.
If the programmable watchdog timer function is not need-
ed, it can be disabled by connecting the CT pin to ground.
The CT pin should not be left floating or un-terminated, as
this will cause errors in the internal timing control circuit.
The constant current provided to charge the timing
capacitor is very small, and this pin is susceptible to
noise and changes in capacitance value. Therefore, the
timing capacitor should be physically located on the
printed circuit board layout as closely as possible to the
CT pin. Since the accuracy of the internal timer is domi-
nated by the capacitance value, 10% tolerance or better
ceramic capacitors are recommended. Ceramic capacitor
materials, such as X7R and X5R type, are a good choice
for this application.
Over-Voltage Protection
An over-voltage event is defined as a condition where
the voltage on the BAT pin exceeds the maximum bat-
tery charge voltage and is set by the over-voltage pro-
tection threshold (VOVP). If an over-voltage condition
occurs, the AAT2550 charge control will shut down the
device until voltage on the BAT pin drops below the over-
voltage protection threshold (VOVP). The AAT2550 will
resume normal charging operation after the over-voltage
condition is removed. During an over-voltage event, the
STAT LEDs will report a system fault, and the actual fault
condition may be read via the DATA pin signal.
Over-Temperature Shutdown
The AAT2550 has a thermal protection control circuit
which will shut down charging functions should the inter-
nal die temperature exceed the preset thermal limit
threshold.
Battery Temperature Fault Monitoring
In the event of a battery over-temperature condition,
the charge control will turn off the internal pass device
and report a battery temperature fault on the DATA pin
function. The STAT LEDs will also display a system fault.
After the system recovers from a temperature fault, the
device will resume charging operation.
The AAT2550 checks battery temperature before start-
ing the charge cycle, as well as during all stages of
charging. This is accomplished by monitoring the voltage
at the TS pin. This system is intended to use negative
temperature coefficient thermistors (NTC), which are
typically integrated into the battery package. Most of the
commonly used NTC thermistors in battery packs are
approximately 10kΩ at room temperature (25°C).
The TS pin has been specifically designed to source 80μA
of current to the thermistor. The voltage on the TS pin
that results from the resistive load should stay within a
window from 330mV to 2.3V. If the battery becomes too
hot during charging due to an internal fault, the thermis-
tor will heat up and reduce in value, pulling the TS pin
voltage lower than the TS1 threshold, and the AAT2550
will signal the fault condition.
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If the use of the TS pin function is not required by the
system, it should be terminated to ground with a 10kΩ
resistor.
Battery Charge Status Indication
The AAT2550 indicates the status of the battery under
charge with two different systems. First, the device has
two status LED driver outputs. These two LEDs can indi-
cate simple functions such as no battery charge activity,
battery charging, charge complete, and charge fault. The
AAT2550 also provides a bi-directional data reporting
function so that a system microcontroller can interrogate
the DATA pin and read any one of 13 system states.
Status Indicator Display
Simple system charging status states can be displayed
using one or two LEDs in conjunction with the STAT1 and
STAT2 pins on the AAT2550. These two pins are simple
switches to connect the LED cathodes to ground. It is not
necessary to use both display LEDs if a user simply
wants to have a single lamp to show “charging” or “not
charging.” This can be accomplished by using the STAT1
pin and a single LED. Using two LEDs and both STAT pins
simply gives the user more information to the charging
states. Refer to Table 3 for LED display definitions.
The LED anodes should be connected to ADP. The LEDs
should be biased with as little current as necessary to
create reasonable illumination; therefore, a ballast resis-
tor should be placed between the LED cathodes and the
STAT1/2 pins. LED current consumption will add to the
overall thermal power budget for the device package, so
it is wise to keep the LED drive current to a minimum.
2mA should be sufficient to drive most low-cost green or
red LEDs. It is not recommended to exceed 8mA for driv-
ing an individual status LED.
The required ballast resistor value can be estimated
using the following formulas:
For connection to the adapter supply:
RB(STAT1/2) = VADP - VF(LED)
ILED(STAT1/2)
Example:
RB(STAT1) = = 1.75kΩ
5.5V - 2.0V
2mA
Note: Red LED forward voltage (VF) is typically 2.0V @
2mA. Green LED forward voltage (VF) is typically 3.2V @
2mA.
The four status LED display conditions are described in
Table 3.
Event Description STAT1 STAT2
Charge Disabled or Low Supply Off Off
Charge Enabled Without Battery Flash1Flash1
Battery Charging On Off
Charge Completed Off On
Fault On On
Table 3: Status LED Display Conditions.
Digital Charge Status Reporting
The AAT2550 has a comprehensive digital data reporting
system by use of the DATA pin feature. This function can
provide detailed information regarding the status of the
charging system. The DATA pin is a bi-directional port
which will read back a series of data pulses when the
system microcontroller asserts a request pulse. This sin-
gle strobe request protocol will invoke one of 13 possible
return pulse counts which the microcontroller can look up
based on the serial report table shown in Table 4.
Number DATA Report Status
1 Chip Over-Temperature Shutdown
2 Battery Temperature Fault
3 Over-Voltage Turn Off
4 Not Used
5ADP Watchdog Time-Out in
Battery Condition Mode
6 ADP Battery Condition Mode
7ADP Watchdog Time-Out in
Constant Current Mode
8ADP Thermal Loop Regulation in
Constant Current Mode
9 ADP Constant Current Mode
10 ADP Watchdog Time-Out in
Constant Voltage Mode
11 ADP Constant Voltage Mode
12 ADP End of Charging
23 Data Report Error
Table 4: Serial Data Report Table.
1. Flashing rate depends on output capacitance.
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The DATA pin function is active low and should normally
be pulled high to VADP
. This data line may also be pulled
high to the same level as the high state for the logic I/O
port on the system microcontroller. In order for the DATA
pin control circuit to generate clean, sharp edges for the
data output and to maintain the integrity of the data tim-
ing for the system, the pull-up resistor on the data line
should be low enough in value so that the DATA signal
returns to the high state without delay. If too small a
pull-up resistor is used, the strobe pulse from the system
microcontroller could exceed the maximum pulse time
and the DATA output control could issue false status
reports. A 1.5kΩ resistor is recommended when pulling
the DATA pin high to 5.0V. If the data line is pulled high
to a voltage level less than 5.0V, the pull-up resistor can
be calculated based on a recommended minimum pull-up
current of 3mA. Use the following formula:
RPULL-UP ≤ VPULL-UP
3mA
Data Timing
The system microcontroller should assert an active low
data request pulse for minimum duration of 200ns; this is
specified by the SQPULSE. Upon sensing the rising edge of the
end of the data request pulse, the AAT2550 status data
control will reply the data word back to the system micro-
controller after a delay defined by the data report time
specification TDATA(RPT). The period of the following group of
data pulses will be defined by the TDATA specification.
IN
OUT
AAT2550
Status
Control
DATA Pin
μP GPIO
Port
OUT
IN
GPIO
RPULL_UP
1.8V to 5.0V
Figure 3: Data Pin Application Circuit.
Timing Diagram
SQ
S
QPULSE
Data
System Reset
System Start
CK
T
SYNC
T
LAT
N=1 N=2 N=3
T
OFF
T
DATA(RPT)
= T
SYNC
+ T
LAT
< 2.5 P
DATA
T
OFF
> 2 P
DATA
P
DATA
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Capacitor Selection
Input Capacitor
In general, it is good design practice to place a decou-
pling capacitor between the ADP pin and ground. An
input capacitor in the range of 1μF to 22μF is recom-
mended. If the source supply is unregulated, it may be
necessary to increase the capacitance to keep the input
voltage above the under-voltage lockout threshold during
device enable and when battery charging is initiated.
If the AAT2550 adapter input is to be used in a system
with an external power supply source, such as a typical
AC-to-DC wall adapter, then a CIN capacitor in the range
of 10μF should be used. A larger input capacitor in this
application will minimize switching or power bounce
effects when the power supply is “hot plugged.”
Output Capacitor
The AAT2550 only requires a 1μF ceramic capacitor on
the BAT pin to maintain circuit stability. This value should
be increased to 10μF or more if the battery connection is
made any distance from the charger output. If the
AAT2550 is to be used in applications where the battery
can be removed from the charger, such as in the case of
desktop charging cradles, an output capacitor greater
than 10μF may be required to prevent the device from
cycling on and off when no battery is present.
Step-Down Converter
Functional Description
The AAT2550 has two step-down converters and both
are designed with the goal of minimizing external com-
ponent size and optimizing efficiency over the complete
load range (600mA). Apart from the small bypass input
capacitor, only a small L-C filter is required at the output.
Typically, a 4.7μH inductor and a 4.7μF ceramic capacitor
are recommended (see Table 5).
Confi guration Output Voltage Inductor
0.6V Adjustable With
External Feedback
1V, 1.2V 2.2μH
1.5V, 1.8V 4.7μH
2.5V, 3.3V 6.8μH
Table 5: Inductor Values.
The two step-down converters can be programmed with
external feedback to any voltage, ranging from 0.6V to
the input voltage. An additional feed-forward capacitor
can also be added to the external feedback with a 10μF
output capacitor for improved transient response (see
C10 and C11 in Figure 4).
At dropout, the converter duty cycle increases to 100%
and the output voltage tracks the input voltage minus
the RDS(ON) drop of the P-channel high-side MOSFET.
The input voltage range is 2.7V to 5.5V. The converter
efficiency has been optimized for all load conditions,
ranging from no load to 600mA.
The internal error amplifier and compensation provides
excellent transient response, load, and line regulation.
Soft start eliminates any output voltage overshoot when
the enable or the input voltage is applied.
Control Loop
Both step-down converters are peak current mode control
converters. The current through the P-channel MOSFET
(high side) is sensed for current loop control, as well as
short-circuit and overload protection. A fixed slope com-
pensation signal is added to the sensed current to main-
tain stability for duty cycles greater than 50%. The peak
current mode loop appears as a voltage-programmed cur-
rent source in parallel with the output capacitor.
The output of the voltage error amplifier programs the
current mode loop for the necessary peak switch current
to force a constant output voltage for all load and line
conditions. Internal loop compensation terminates the
transconductance voltage error amplifier output. The
error amplifier reference is fixed at 0.6V.
Soft Start / Enable
Soft start limits the current surge seen at the input and
eliminates output voltage overshoot. When pulled low,
the enable input forces the AAT2550 into a low-power,
non-switching state. The total input current during shut-
down is less than 1μA.
Current Limit and
Over-Temperature Protection
For overload conditions, the peak input current is limit-
ed. To minimize power dissipation and stresses under
current limit and short-circuit conditions, switching is
terminated after entering current limit for a series of
pulses. Switching is terminated for seven consecutive
clock cycles after a current limit has been sensed for a
series of four consecutive clock cycles.
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Thermal protection completely disables switching when
internal dissipation becomes excessive. The junction
over-temperature threshold is 140°C with 15°C of hys-
teresis. Once an over-temperature or over-current fault
conditions is removed, the output voltage automatically
recovers.
Under-Voltage Lockout
Internal bias of all circuits is controlled via the VIN input.
Under-voltage lockout (UVLO) guarantees sufficient VIN
bias and proper operation of all internal circuitry prior to
activation.
Step-Down Converter
Applications Information
Inductor Selection
The step-down converter uses peak current mode con-
trol with slope compensation to maintain stability for
duty cycles greater than 50%. The output inductor value
must be selected so the inductor current down slope
meets the internal slope compensation requirements.
The internal slope compensation for the AAT2550 is
0.24A/μs. This equates to a slope compensation that is
75% of the inductor current down slope for a 1.5V out-
put and 4.7μH inductor.
0.75 ⋅ V
O
m = = = 0.24
L
0.75 ⋅ 1.5V
4.7μH
A
μsec
This is the internal slope compensation for the step-
down converter. When externally programming the 0.6V
version to 2.5V, the calculated inductance is 7.5μH.
0.75 ⋅ V
O
L = =
≈
3
⋅ V
O
= 3 ⋅ 2.5V = 7.5μH
m
0.75
⋅
V
O
0.24A
μsec
A
μsec
A
A
μsec
In this case, a standard 6.8μH value is selected.
For high-voltage output (≥2.5V), m = 0.48A/μs. Table 5
displays inductor values for the AAT2550 step-down con-
verters.
Manufacturer's specifications list both the inductor DC
current rating, which is a thermal limitation, and the
peak current rating, which is determined by the satura-
tion characteristics. The inductor should not show any
appreciable saturation under normal load conditions.
Some inductors may meet the peak and average current
ratings yet result in excessive losses due to a high DCR.
Always consider the losses associated with the DCR and
its effect on the total converter efficiency when selecting
an inductor.
The Sumida 4.7μH CDRH2D14 series inductor has a
135mΩ DCR and a 1A DC current rating. At full load, the
inductor DC loss is 48.6mW, which gives a 4% loss in
efficiency for a 600mA, 1.5V output.
Input Capacitor
Select a 4.7μF to 10μF X7R or X5R ceramic capacitor for
the input. To estimate the required input capacitor size,
determine the acceptable input ripple level (VPP) and
solve for C. The calculated value varies with input volt-
age and is a maximum when VIN is double the output
voltage.
⎛⎞
· 1 -
⎝⎠
VO
VIN
CIN =
VO
VIN
⎛⎞
- ESR · FS
⎝⎠
VPP
IO
⎛⎞
· 1 - = for VIN = 2 · V
O
⎝⎠
VO
VIN
VO
VIN
1
4
CIN(MIN) = 1
⎛⎞
- ESR · 4 · FS
⎝⎠
VPP
IO
Always examine the ceramic capacitor DC voltage coeffi-
cient characteristics when selecting the proper value. For
example, the capacitance of a 10μF, 6.3V, X5R ceramic
capacitor with 5.0V DC applied is actually about 6μF.
The maximum input capacitor RMS current is:
⎛⎞
IRMS = IO · · 1 -
⎝⎠
VO
VIN
VO
VIN
The input capacitor RMS ripple current varies with the
input and output voltage and will always be less than or
equal to half of the total DC load current.
⎛⎞
· 1 - = D · (1 - D) = 0.52 =
⎝⎠
VO
VIN
VO
VIN
1
2
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for VIN = 2 · VO
IO
RMS(MAX)
I2
=
The term
⎛⎞
· 1 -
⎝⎠
VO
VIN
VO
VIN appears in both the input voltage
ripple and input capacitor RMS current equations and is
a maximum when VO is twice VIN. This is why the input
voltage ripple and the input capacitor RMS current ripple
are a maximum at 50% duty cycle.
The input capacitor provides a low impedance loop for
the edges of pulsed current drawn by the AAT2550. Low
ESR/ESL X7R and X5R ceramic capacitors are ideal for
this function. To minimize stray inductance, the capacitor
should be placed as closely as possible to the IC. This
keeps the high frequency content of the input current
localized, minimizing EMI and input voltage ripple.
Proper placement of the input capacitors (C4 and C5) can
be seen in the evaluation board schematic in Figure 4.
A laboratory test set-up typically consists of two long
wires running from the bench power supply to the evalu-
ation board input voltage pins. The inductance of these
wires, along with the low-ESR ceramic input capacitor,
can create a high Q network that may affect converter
performance. This problem often becomes apparent in
the form of excessive ringing in the output voltage dur-
ing load transients. Errors in the loop phase and gain
measurements can also result.
Since the inductance of a short PCB trace feeding the
input voltage is significantly lower than the power leads
from the bench power supply, most applications do not
exhibit this problem.
In applications where the input power source lead induc-
tance cannot be reduced to a level that does not affect
the converter performance, a high ESR tantalum or alu-
minum electrolytic input capacitor should be placed in
parallel with the low ESR bypass ceramic input capacitor
(C6 of Figure 4). This dampens the high Q network and
stabilizes the system.
Output Capacitor
The output capacitor limits the output ripple and pro-
vides holdup during large load transitions. A 4.7μF to
10μF X5R or X7R ceramic capacitor typically provides
sufficient bulk capacitance to stabilize the output during
large load transitions and has the ESR and ESL charac-
teristics necessary for low output ripple.
The output voltage droop due to a load transient is
dominated by the capacitance of the ceramic output
capacitor. During a step increase in load current, the
ceramic output capacitor alone supplies the load current
until the loop responds. Within two or three switching
cycles, the loop responds and the inductor current
increases to match the load current demand. The rela-
tionship of the output voltage droop during the three
switching cycles to the output capacitance can be esti-
mated by:
COUT =
3 · ΔILOAD
VDROOP · FS
Once the average inductor current increases to the DC
load level, the output voltage recovers. The above equa-
tion establishes a limit on the minimum value for the
output capacitor with respect to load transients.
The internal voltage loop compensation also limits the
minimum output capacitor value to 4.7μF. This is due to
its effect on the loop crossover frequency (bandwidth),
phase margin, and gain margin. Increased output capac-
itance will reduce the crossover frequency with greater
phase margin.
The maximum output capacitor RMS ripple current is
given by:
1
23
VOUT · (VIN(MAX) - VOUT)
RMS(MAX)
IL · FS · VIN(MAX)
=·
·
Dissipation due to the RMS current in the ceramic output
capacitor ESR is typically minimal, resulting in less than
a few degrees rise in hot-spot temperature.
Feedback Resistor Selection
Table 6 shows all output voltages, which can be exter-
nally programmed. Resistors R7 through R10 of Figure 4
program the output to regulate at a voltage higher than
0.6V. To limit the bias current required for the external
feedback resistor string while maintaining good noise
immunity, the minimum suggested value for R7 and R9
is 59kΩ. Although a larger value will further reduce qui-
escent current, it will also increase the impedance of the
feedback node, making it more sensitive to external
noise and interference. Table 6 summarizes the resistor
values for various output voltages with R7 and R9 set to
either 59kΩ for good noise immunity or 221kΩ for
reduced no load input current.
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VOUT (V)
R7, R9 = 59kΩ
R8, R10 (kΩ)
R7, R9 = 221kΩ
R8, R10 (kΩ)
0.8 19.6 75
0.9 29.4 113
1.0 39.2 150
1.1 49.9 187
1.2 59.0 221
1.3 68.1 261
1.4 78.7 301
1.5 88.7 332
1.8 118 442
1.85 124 464
2.0 137 523
2.5 187 715
3.3 267 1000
Table 6: Adjustable Resistor Values for Use With
0.6V Step-Down Converter.
The AAT2550, combined with an external feedforward
capacitor (C10 and C11 in Figure 4), delivers enhanced
transient response for extreme pulsed load applications.
The addition of the feedforward capacitor (100pF) typi-
cally requires a larger output capacitor for stability.
⎛⎞
⎝⎠
R8 = -1 · R7 = - 1 · 59kΩ = 88.5kΩ
VOUT
VREF
⎛⎞
⎝⎠
1.5V
0.6V
Thermal Considerations
The AAT2550 is available in a 4x4mm QFN package,
which has a typical thermal resistance of 50°C/W when
the exposed paddle is soldered to a printed circuit board
(PCB) in the manner discussed in the Printed Circuit
Board Layout section of this datasheet. Thermal resis-
tance will vary with the PCB area, ground plane area,
size and number of other adjacent components, and the
heat they generate. The maximum ambient operating
temperature is limited by either the design derating cri-
teria, the over-temperature shutdown temperature, or
the thermal loop charge current reduction control. To
calculate the junction temperature, sum the step-down
converter losses with the battery charger losses. Multiply
the total losses by the package thermal resistance and
add to the ambient temperature to determine the junc-
tion temperature rise.
TJ(MAX) = (PSD + PC) · θJA + TAMB
PSD is the total loss associated with both step-down con-
verters and PC is the loss associated with the charger.
The total losses will vary considerably depending on
input voltage, load, and charging current. While charg-
ing a battery, the current capability of the step-down
converters is limited.
Step-Down Converter Losses
There are three types of losses are associated with the
AAT2550 step-down converter: switching losses (tSW ·
FS), conduction losses (I2 · RDS(ON)), and quiescent cur-
rent losses (IQ · VIN). At full load, assuming continuous
conduction mode, a simplified form of the step-down
converter losses is:
PSD =
+ (tSW · FS · (IOA + IOB) + 2 · IQ ) · VIN
IOA2 · (RDS(ON)H · VOA + RDS(ON)L · (VIN - VOA)) + IOB2 · (RDS(ON)H · VOB + RDS(ON)L · (VIN - VOB))
VIN
For the condition where one channel is in dropout at
100% duty cycle (IOA), the step-down converter dissipa-
tion is:
PSD = IOA2 · RDS(ON)H
+ (tSW · FS · IOB + 2 · IQ ) · VIN
+ IOB2 · (RDS(ON)H · VOB + RDS(ON)L · (VIN - VOB)
)
VIN
PSD = Step-Down Converter Dissipation
VIN = Converter Input Voltage
RDS(ON)H = High Side MOSFET On Resistance
RDS(ON)L = Low Side MOSFET On Resistance
VOA = Converter A Output Voltage
VOB = Converter B Output Voltage
IOA = Converter A Load Current
IOB = Converter B Load Current
IQ = Converter Quiescent Current
tSW = Switching Time Estimate
FS = Converter Switching Frequency
Always use the RDS(ON) and quiescent current value that
corresponds to the applied input voltage.
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Battery Charger Losses
The maximum battery charger loss is:
PC = (VADP - VMIN) · ICH + VADP · IQC
PC = Total Charger Dissipation
VADP = Adapter Voltage
VMIN = Preconditioning Voltage Threshold
ICH = Programmed Charge Current
IQC = Charger Quiescent Current Consumed by the
Charger
For an application where no load is applied to the step-
down converters and the charger current is set to 1A
with VADP = 5.0V, the maximum charger dissipation
occurs at the preconditioning voltage threshold VMIN.
PC = (VADP - VMIN) · ICH + VADP · IQC
= (5.0V - 3.0V) · 1A + 5.0V · 0.75m
A
= 2W
The charger thermal loop begins reducing the charge
current at a 110°C junction temperature (TLOOP_IN). The
ambient temperature at which the charger thermal loop
begins reducing the charge current is:
TA = TLOOP_IN - θJA · PC
= 110°C - (50°C/W · 2W)
= 10°C
Therefore, under the given conditions, the AAT2550 bat-
tery charger will enter the thermal loop charge current
reduction at an ambient temperature greater than 10°C.
Total Power Loss Examples
The most likely high power scenario is when the charger
and step-down converter are both operational and pow-
ered from the adapter. To examine the step-down con-
verter maximum current capability for this condition, it is
necessary to determine the step-down converter MOSFET
RDS(ON), quiescent current, and switching losses at the
adapter voltage level (5V). This example shows that with
a 600mA battery charge current, the buck converter out-
put current capability is limited 400mA. This limits the
junction temperature to 110°C and avoids the thermal
loop charge reduction at a 70°C ambient temperature.
Conditions:
VOA 2.5V @ 400mA Step-Down Converter A
VOB 1.8V @ 400mA Step-Down Converter B
IQ70μA Converter Quiescent Current
VIN = VADP 5.0V Charger and Step-Down
VMIN 3.0V Battery Preconditioning
Threshold Voltage
ICH 0.6A Battery Charge Current
IOP 0.75mA Charger Operating Current
The step-down converter load current capability is great-
est when the battery charger is disabled. The following
example demonstrates the junction temperature rise for
conditions where the battery charger is disabled and full
load is applied to both converter outputs at the nominal
battery input voltage.
PTOTAL =
+ (tSW · FS · (IOA + IOB) + 2 · IQ) · VIN + (VADP - VMIN) · ICH + VADP · IOP
+ 2 · (5ns · 1.4MHz · 0.4A + 70µA) · 5.0V + (5.0V - 3.0V) · 0.6A + 5.0V · 0.75mA = 1.38W
=
IOA2 · (RDS(ON)H · VOA + RDS(ON)L · (VIN - VOA)) + IOB2 · (RDS(ON)H · VOB + RDS(ON)L · (VIN - VOB))
VIN
0.4A2 · (0.475Ω · 2.5V + 0.45Ω · (5.0V - 2.5V)) + 0.4A2 · (0.475Ω · 1.8V + 0.45Ω · (5.0V - 1.8V))
5.0V
TJ(MAX) = TAMB + (θJA · PLOSS)
= 70°C + (50°C/W · 1.38W)
= 139°C
Conditions:
VOA 2.5V @ 600mA Step-Down Converter A
VOB 1.8V @ 600mA Step-Down Converter B
IQ70μA Converter Quiescent Current
VIN 3.6V
Charger and Step-Down Con-
verter Input Voltage
ICH = IOP 0A Charger Disabled
PTOTAL =
+ (tSW · FS · (IOA + IOB) + 2 · IQ) · VIN + (VADP - VMIN) · ICH + VADP · IOP
+ 2 · (5ns · 1.4MHz · 0.4A + 70µA) · 3.6V = 0.443W
=
IOA2 · (RDS(ON)H · VOA + RDS(ON)L · (VIN - VOA)) + IOB2 · (RDS(ON)H · VOB + RDS(ON)L · (VIN - VOB))
VIN
0.6A2 · (0.58Ω · 2.5V + 0.56Ω · (3.6V - 2.5V)) + 0.2A2 · (0.58Ω · 1.8V + 0.56Ω · (3.6V - 1.8V))
3.6V
TJ(MAX) = TAMB + (θJA · PLOSS)
= 85°C + (50°C/W · 0.443W)
= 107.15°C
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Printed Circuit Board Layout
Use the following guidelines to ensure a proper printed
circuit board layout.
1. Step-down converter bypass capacitors (C4 and C5
in Figure 4) must be placed as close as possible to
the step-down converter inputs.
2. The connections from the LXA and LXB pins of the
step-down converters to the output inductors should
be kept as short as possible. This is a switching
node, so minimizing the length will reduce the
potential of this noisy trace interfering with other
high impedance noise sensitive nodes.
3. The feedback trace should be separate from any
power trace and connected as closely as possible to
the load point. Sensing along a high current load
trace will degrade the DC load regulation. If external
feedback resistors are used, they should be placed
as closely as possible to the FB pins and AGND. This
prevents noise from being coupled into the high
impedance feedback node.
4. The resistance of the trace from the load return to
GND should be kept to a minimum. This minimizes
any error in DC regulation due to differences in the
potential of the internal signal ground and the power
ground.
5. For good thermal coupling, vias are required from
the pad for the QFN paddle to the ground plane. Via
diameters should be 0.3mm to 0.33mm and posi-
tioned on a 1.2mm grid. Avoid close placement to
other heat generating devices.
6. Minimize the trace impedance from the battery to
the BAT pin. The charger output is not remotely
sensed, so any drop in the output across the BAT
output trace feeding the battery will add to the error
in the EOC battery voltage. To minimize voltage
drops on the PCB, maintain an adequate high current
carrying trace width.
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1
3
2
EnVoA
1
2
Adapter
GND
Data Strobe
SW1
STAT1
D1
STAT2
D2
4.7μHL2
6.8μHL1
ENA
1
LXA
2
PGND
3
DATA
4
N/C
5
ADPSET
6
N/C
7
BA
8
AD
9
AGND
10
AGND
11
ENBA
12
TS 13
STAT2 14
STAT1 15
CT 16
PGND 17
LXB 18
ENB
19
VINB
20
FBB
21
AGND
22
FBA
23
VINA
24
AAT2550
U1
4.7μF
C8 4.7μF
C9
10μF
C4
10μF
C5
1
3
2
Charger Enable
1
3
2
Battery
ADP
BAT
GND
TS
10μF
C3
0.1μF
C12
118k
R10
59k
R9
59k
R7
187k
R8
(opt)
100pF
C11
(opt.)
100pF
C10
1.5k
R1
1.5k
R2
8.06k
R6
VIN
VoB
VoA
CT
GND
GND GND
EnVoB
(opt)
120μF
C6
1kR3
(open)
C14
Data
LXA
LXB
VoA, VoB (V) R8, R10 (Ω)
1.0
1.2
1.5
1.8
2.5
3.0
3.3
9.2k
59k
88.7k
118k
187k
237k
267k
2.2μH (CDRH2D14; DCR 75mΩ; 1200mA @ 20°C)
2.2μH (CDRH2D14; DCR 75mΩ; 1200mA @ 20°C)
4.7μH (CDRH2D14; DCR 135mΩ; 1000mA @ 20°C)
4.7μH (CDRH2D14; DCR 135mΩ; 1000mA @ 20°C)
6.8μH (CDRH2D14; DCR 170mΩ; 850mA @ 20°C)
6.8μH (CDRH2D14; DCR 170mΩ; 850mA @ 20°C)
6.8μH (CDRH2D14; DCR 170mΩ; 850mA @ 20°C)
L1, L2
VoA
VoB
10k
R4
Green
Red
10μF
C13
1
3
2
Figure 4: AAT2550 Evaluation Board Schematic.
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Figure 5: AAT2550 Evaluation Board Figure 6: AAT2550 Evaluation Board
Top Side Layout. Layer 2 Layout.
Figure 7: AAT2550 Evaluation Board Figure 8: AAT2550 Evaluation Board
Layer 3 Layout. Bottom Side Layout.
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Qty. Description
Reference
Designator Manufacturer Part Number
1 Conn. Term Block 2.54mm 2 POS Adapter Input Phoenix Contact
1 Conn. Term Block 2.54mm 3 POS Battery Output Phoenix Contact
3 Ceramic Capacitor 10μF 10%, 10V, X5R, 0805 C3, C4, C5, C13 Murata
2 Ceramic Capacitor 4.7μF 10%, 6.3V, X5R, 0805 C8,C9 Murata
1 Ceramic Capacitor 0.1μF 25V 10% X5R 0603 C12 Vishay
1 Tantalum Capacitor 100μF, 6.3V, Case C C6 Vishay
2 Optional Ceramic Capacitor 100pF, 0402, COG C10, C11 Vishay
2 Ferrite Shielded Inductor CDRH2D14 L1, L2 Sumida
2 1.5k, 5%, 1/16W, 0402 R1,R2 Vishay
1 1.0k, 5%, 1/16W, 0402 R3 Vishay
1 8.06k, 1%, 1/16W, 0402 R6 Vishay
2 59.0k, 1%, 1/16W, 0402 R7,R9 Vishay
1 118k, 1%, 1/16W, 0402 R10 Vishay
1 187k, 1%, 1/16W, 0402 R8 Vishay
1 10k, 5%, 1/16W, 0402 R4 Vishay
1 Red LED, 1206 D1 Chicago Miniature Lamp CMD15-21SRC/TR8
1 Green LED, 1206 D2 Chicago Miniature Lamp CMD15-21SRC/TR8
1 Switch Tact 6mm SPST H = 5.0mm SW1 ITT Industries/C&K Div CKN9012-ND
1AAT2550 Total Power Solution for Portable
Applications U1 Advanced Analogic Technologies AAT2550ISK-CAA-T1
Table 7: AAT2550 Evaluation Board Bill of Materials.
Manufacturer Part Number
Inductance
(μH)
Max DC Current
(A)
DCR
(Ω)
Size (mm)
LxWxH Type
Sumida CDRH2D14-2R2 2.2 1.20 0.075 3.2x3.2x1.55 Shielded
Sumida CDRH2D14-4R7 4.7 1.00 0.135 3.2x3.2x1.55 Shielded
Sumida CDRH2D14-6R8 6.8 0.85 0.170 3.2x3.2x1.55 Shielded
Coilcraft LPO3310-472 4.7 0.80 0.27 3.2x3.2x1.0 1mm
Coiltronics SD3118-4R7 4.7 0.98 0.122 3.1x3.1x1.85 Shielded
Coiltronics SD3118-6R8 6.8 0.82 0.175 3.1x3.1x1.85 Shielded
Coiltronics SDRC10-4R7 4.7 1.30 0.122 5.7x4.4x1.0 1mm Shielded
Table 8: Typical Surface Mount Inductors.
Manufacturer Part Number Value Voltage Temp. Co. Case
Murata GRM219R61A475KE19 4.7μF 10V X5R 0805
Murata GRM21BR60J106KE19 10μF 6.3V X5R 0805
Murata GRM21BR60J226ME39 22μF 6.3V X5R 0805
Table 9: Surface Mount Capacitors.
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Adjustable Version
(0.6V device)
VOUT (V)
R7, R9 = 59kΩ
R8, R10 (kΩ)
R7, R9 = 221kΩ1
R8, R10 (kΩ)
L1, L2
(μH)
0.8 19.6 75.0 2.2
0.9 29.4 113 2.2
1.0 39.2 150 2.2
1.1 49.9 187 2.2
1.2 59.0 221 2.2
1.3 68.1 261 2.2
1.4 78.7 301 4.7
1.5 88.7 332 4.7
1.8 118 442 4.7
1.85 124 464 4.7
2.0 137 523 6.8
2.5 187 715 6.8
3.3 267 1000 6.8
Table 10: Evaluation Board Component Values.
1. For reduced quiescent current, R7 and R9 = 221kΩ.
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Step-Down Converter Design Example
Specifications
VOA = 2.5V @ 400mA (VFBA = 0.6V), pulsed load ΔILOAD = 300mA
VOB = 1.8V @ 400mA (VFBB = 0.6V), pulsed load ΔILOAD = 300mA
VIN = 2.7V to 4.2V (3.6V nominal)
FS = 1.4MHz
TAMB = 85°C
2.5V VOA Output Inductor
L1 = 3 ⋅ V
O1
= 3 ⋅ 2.5V = 7.5μH
μsec
A
μsec
A
(see Table 5)
For Sumida inductor CDRH2D14, 6.8μH, DCR = 170mΩ.
V
O
V
OA
2.5
V
2.5V
ΔIA =
⋅ 1 - = ⋅ 1 - = 106m
A
L1 ⋅ F
S
V
IN
6.8μH ⋅ 1.4MHz
4.2V
I
PKA
= I
OA
+ ΔIA = 0.4A + 0.053A = 0.453A
2
P
LA
= I
OA
2
⋅ DCR = 0.45
2
⋅ 170mΩ = 34mW
⎛
⎝⎞
⎠
⎛
⎝⎞
⎠
1.8V VOB Output Inductor
L2 = 3 ⋅ V
O2
= 3 ⋅ 1.8V = 5.4μH
μsec
A
μsec
A
(see Table 5)
For Sumida inductor CDRH2D14, 4.7μH, DCR = 135mΩ.
V
OB
V
OB
1.8
V
1.8V
ΔIB =
⋅ 1 - = ⋅ 1 - = 156m
A
L ⋅ F
S
V
IN
4.7μH ⋅ 1.4MHz
4.2V
I
PKB
= I
OB
+ ΔIB = 0.4A + 0.078A = 0.48A
2
P
LB
= I
OB
2
⋅ DCR = 0.4A
2
⋅ 135mΩ = 21.6mW
⎛
⎝⎞
⎠
⎛
⎝⎞
⎠
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PRODUCT DATASHEET
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www.analogictech.com
AAT2550178
Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
32 2550.2008.02.1.3
www.analogictech.com
2.5V Output Capacitor
1
23
1 2.5V · (4.2V - 2.5V)
10μH · 1.4MHz · 4.2V
23
RMS(MAX)
IL · FS · VIN(MAX)
=·
·
3 · ΔILOAD
VDROOP · FS
3 · 0.3A
0.2V · 1.4MHz
COUT = = = 3.2μF
· = 21mArms
·
(VOUT) · (VIN(MAX) - VOUT)=
Pesr = esr · IRMS2 = 5mΩ · (21mA)2 = 2.2μW
1.8V Output Capacitor
1
23
1 1.8V · (4.2V - 1.8V)
4.7μH · 1.4MHz · 4.2V
23
RMS(MAX)
IL · FS · VIN(MAX)
=·
·
3 · ΔILOAD
VDROOP · FS
3 · 0.3A
0.2V · 1.4MHz
COUT = = = 3.2μF
· = 45mArms
·
(VOUT) · (VIN(MAX) - VOUT)=
Pesr = esr · IRMS2 = 5mΩ · (45mA)2 = 10μW
Input Capacitor
Input Ripple VPP = 25mV.
CIN = = = 6.8μF
1
⎛⎞
- ESR · 4 · FS
⎝⎠
VPP
IO1 + IO2
1
⎛⎞
- 5mΩ · 4 · 1.4MHz
⎝⎠
25mV
0.8A
IO1 + IO2
RMS(MAX)
I
P = esr · IRMS
2 = 5mΩ · (0.4A)2 = 0.8mW
2
= = 0.4Arms
AAT2550178
Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
2550.2008.02.1.3 33
www.analogictech.com
AAT2550178
Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
2550.2008.02.1.3 33
www.analogictech.com
Ordering Information
Voltage
Marking1Part Number (Tape and Reel)2
Package Converter 1 Converter 2
QFN44-24 0.6V 0.6V RJXYY AAT2550ISK-CAA-T1
All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor
products that are in compliance with current RoHS standards, including the requirement that lead not exceed
0.1% by weight in homogeneous materials. For more information, please visit our website at
http://www.analogictech.com/about/quality.aspx.
Legend
Voltage Code
Adjustable (0.6V) A
0.9 B
1.2 E
1.5 G
1.8 I
1.9 Y
2.5 N
2.6 O
2.7 P
2.8 Q
2.85 R
2.9 S
3.0 T
3.3 W
4.2 C
1. XYY = assembly and date code.
2. Sample stock is generally held on part numbers listed in BOLD.
AAT2550178
Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
34 2550.2008.02.1.3
www.analogictech.com
AAT2550178
Total Power Solution for Portable ApplicationsSystemPowerTM
PRODUCT DATASHEET
34 2550.2008.02.1.3
www.analogictech.com
Advanced Analogic Technologies, Inc.
3230 Scott Boulevard, Santa Clara, CA 95054
Phone (408) 737-4600
Fax (408) 737-4611
© Advanced Analogic Technologies, Inc.
AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual
property rights are implied. AnalogicTech reserves the right to make changes to their products or specifi cations or to discontinue any product or service without notice. Except as provided in AnalogicTech’s terms and
conditions of sale, AnalogicTech assumes no liability whatsoever, and AnalogicTech disclaims any express or implied warranty relating to the sale and/or use of AnalogicTech products including liability or warranties
relating to fi tness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate
design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to
support this warranty. Specifi c testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other
brand and product names appearing in this document are registered trademarks or trademarks of their respective holders.
Package Information
QFN44-24
4.000
±
0.050
2.7
±
0.05
0.300 × 45°
Pin 1 Dot By Marking
4.000
±
0.050 2.7
±
0.05
0.5 BSC 0.4
±
0.05
0.305
±
0.075
0.900
±
0.050
0.025
±
0.025
0.214
±
0.036
Pin 1 Identification
R0.030Max
1
6
712
13
18
19 24
Top View Bottom View
Side View
All dimensions in millimeters.
1. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing
process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required to ensure a proper bottom solder connection.
