A6271 Automotive High-Current LED Controller Not for New Design These parts are in production but have been determined to be NOT FOR NEW DESIGN. This classification indicates that sale of this device is currently restricted to existing customer applications. The device should not be purchased for new design applications because obsolescence in the near future is probable. Samples are no longer available. Date of status change: December 14, 2017 Recommended Substitutions: A6271-1 NOTE: For detailed information on purchasing options, contact your local Allegro field applications engineer or sales representative. Allegro MicroSystems reserves the right to make, from time to time, revisions to the anticipated product life cycle plan for a product to accommodate changes in production capabilities, alternative product availabilities, or market demand. The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems assumes no responsibility for its use; nor for any infringements of patents or other rights of third parties which may result from its use. A6271 Automotive High-Current LED Controller FEATURES AND BENEFITS DESCRIPTION The A6271 is a DC-DC converter controller, providing a programmable constant-current output for driving high-power LEDs in series. The controller is based on a programmable fixed-frequency, peak current-mode control architecture. The DC-DC converter can be configured in a myriad of different switching configurations including boost, buck-boost, and buck (ground-referenced switch). * * * * * * * Automotive AEC-Q100 qualified Constant-current LED drive 4.2 to 50V supply 53.3V maximum LED string voltage Boost, buck-boost, and buck switching converters Programmable switching frequency 70 to 700kHz PWM-controlled PMOS driver allows accurate LED current control at low duty cycles * Dimming via external PWM, internal PWM and/or analog dimming * Frequency dither scheme for effective spread spectrum to reduce EMI * Comprehensive fault protection and fault flag PACKAGE: The A6271 provides a cost-effective solution using an external logic-level MOSFET and minimum additional external components. The maximum LED current is set with a single external sense resistor and can be accurately modulated using a current reference input (analog control). External PWM dimming is possible via the PWMIN input, which also provides a shutdown mode. As an alternative, an internal PWM dimming circuit can be used by programming the PWMIN and DR pins. Either PWM scheme controls the PWMOUT output which drives an external p-channel MOSFET connected in series with the LED string. This MOSFET is also used to isolate the load during certain fault conditions, including output shorts to ground. 16-Pin eTSSOP (LP) with Exposed Thermal Pad Continued on next page... APPLICATIONS * Automotive high-power LED lighting systems * Fog lights, reversing lights, daytime running lights, position lights, headlights Not to scale L1 56 H D2 VBAT C1 4.7 F C2 4.7 F R1 2.7 k C3 1 F C7 47 nF VIN VREG LP LN R6 1.35 R5 150 PWMOUT M2 GND FAULTn DR PWM Control A6271 OVUV PWMIN IREF C4 22 nF R7 4.3 k SG SP OSC R2 73.2 k DITH R3 110 k COMP R4 39 C5 680 nF R8 12 LED1 C8 4.4 F M1 R11 240 k LED14 R9 1.3 k C6 22 pF R10 75 m GND Boost Switcher Driving 14LEDs at 150mA A6271-DS, Rev. 8 MCO-0000359 January 27, 2020 A6271 Automotive High-Current LED Controller Description (continued) The A6271 has been carefully designed to minimize electromagnetic emissions through distributed decoupling and an externally programmable frequency dither circuit configured for the EMI specification CISPR 25. It is also possible to program the fundamental switching frequency below 150kHz where most EMI standards begin. The A6271 has a comprehensive set of integrated protection features to protect the IC, the LED driver system, and the LED string against faults. Fixed-output overvoltage protection ensures no maximum voltage rating violations, even under a single point failure of the programmable-output overvoltage protection circuit. Other protection features include: LED overload (boost), output undervoltage (buck or buck-boost), input supply (VIN) undervoltage, 5Vregulator (VREG) output undervoltage, high-side supply (PWM PMOS) undervoltage, and thermal protection. SPECIFICATIONS SELECTION GUIDE Part Number Packing [1] A6271KLPTR-T 4000 pieces per 13-in. reel [1] Contact AllegroTM Package 16-pin TSSOP with exposed thermal pad for additional packing options. ABSOLUTE MAXIMUM RATINGS[2] Characteristic Rating Unit -0.3 to 55 V PWMOUT, LP, LN, OVUV -0.3 to 58 V OSC, DITH, COMP, FAULTn, SG, SP, IREF, PWMIN, DR, VREG -0.3 to 6.5 V -0.5 to 0.5 V TJ(max) 150 C Tstg -55 to 150 C VIN Notes VIN LP VLP Maximum Continuous Junction Temperature Storage Temperature Range [2] With Symbol With respect to LN respect to GND. THERMAL CHARACTERISTICS Characteristic eTSSOP Package Symbol RJA RJC [3] Additional Test Conditions[3] Value Unit 4-layer PCB based on JEDEC standard 34 C/W 2-layer PCB with 3.8in2 of copper area each side 43 C/W Junction to thermal pad 2 C/W thermal information available on the Allegro website. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 2 A6271 Automotive High-Current LED Controller PINOUT DIAGRAM AND TERMINAL LIST TABLE COMP 1 16 VIN IREF 2 15 LP FAULTn 3 14 LN OSC 4 13 PWMOUT DITH 5 DR 6 11 GND PWMIN 7 10 SP VREG 8 9 PAD 12 OVUV SG Package LP, 16-Pin eTSSOP Pinout Diagram Terminal List Table Symbol Number COMP 1 Compensation pin for output of GM error amplifier. Function IREF 2 Analog dimming input. With a capacitor connected to this pin, provides a soft-start period when coming out of sleep mode. FAULTn 3 Open drain. Logic low indicates detection of a fault. Faults include: LED overload (boost), output undervoltage (buck or buck-boost), output overvoltage, programmable overvoltage, input supply (VIN) undervoltage, 5VRegulator (VREG) output undervoltage. OSC 4 Oscillator input for setting switching frequency and for external synchronization. DITH 5 Dither frequency range set. Connect resistor from this pin to GND. Connect to VREG if not used. DR 6 A voltage applied to this pin programs the duty cycle of PWM internal mode. PWMIN 7 Used for either putting the device into sleep mode or analog dimming control. Can also be used for external or internal PWM control. VREG 8 5V regulator output. Connect filter capacitor from VREG to GND. SG 9 Switch gate drive output. SP 10 Switch current sense and slope compensation. GND 11 Ground. OVUV 12 Programmable-output overvoltage and undervoltage protection input. PWMOUT 13 PWM gate drive for external p-channel MOSFET (active low). LN 14 LED current sense -ve. LP 15 LED current sense +ve. VIN 16 Main supply. - Exposed pad of both packages provides both electrical contact to the ground and good thermal contact to the PCB. This pad must be soldered to the ground plane preferably by multiple through-hole vias. PAD Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 3 A6271 Automotive High-Current LED Controller RSL VBAT PWMOUT VREG 5V Linear Reg CA LP - + -8 V wrt LP - CB + - + - + CD LN - OVUV VIN AA Prog Output OV Output OV + VREG FAULTn Internal Linear Reg Output UV PWM Fault Block + -+ IREF - Analog DIM/ Soft Start + -CE AB Overload - CF + R SG S Q PWM SP + AC RSLOPE - PWMIN VIN UVLO VREG UVLO Prog Output OV Output OV Thermal - RSS CG + Osc DR Dither GND Internal PWM Generator COMP OSC DITH PWM On/Off Functional Block Diagram Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 4 A6271 Automotive High-Current LED Controller ELECTRICAL CHARACTERISTICS: Valid at TJ = -40C to 150C, VIN = 5 to 45V, unless noted otherwise. Characteristics Symbol Test Conditions Min. Typ. Max. Unit VIN VIN undervoltage turn off plus VIN undervoltage hysteresis cleared; PWMOUT undervoltage turnon cleared 4.2 - 50 V IINQ SG Open Circuit - 3.5 5 mA IINS PWMIN = GND > disable time - 6 20 A IREG = 0 to 2 mA, VIN 5.3 V 4.85 5.04 5.15 V IREG = 2 mA, VIN = 5 V 4.65 - - V IREG = 2mA, VIN = 9 to 45V, TJ = -40C to 125C 4.95 5.05 5.15 V 25 - - mA 30 - ns SUPPLY and REFERENCE VIN Functional Operating Range[2] VIN Quiescent Current VREG Output Voltage VREG VREG Output Voltage[3] VREG VREG Current Limit IREGCL GATE OUTPUT DRIVE Turn-On Time Turn-Off Time Minimum Off-Time tr CLOAD = 1 nF, 20% to 80% - tf CLOAD = 1 nF, 80% to 20% - 30 - ns - 135 165 ns TJ = 25C, IGHx = -100 mA - 1.7 - TJ = 150C, IGHx = -100 mA - - 3.6 TJ = 25C, IGLx = 100 mA - 0.75 - TJ = 150C, IGLx = 100 mA - - 2 toff(MIN) Pull-Up On-Resistance RDS(on)UP Pull-Down On-Resistance RDS(on)DN Output High Voltage VSGH ISG = -100 A VREG - 0.1 - VREG V Output Low Voltage VSGL ISG = 100 A - - 0.1 V VOL IOL = 1 mA, fault asserted - - 0.4 V IOH VO = 5.5 V, fault not asserted -1 - 1 A - - 0.3 V LOGIC INPUTS AND OUTPUTS FAULTn Output (Open Drain) FAULTn Output Leakage Current[1] PWMIN Low Voltage VPWMINL PWMIN High Voltage VPWMINH 2 - - V VIhys 150 180 - mV IPWMSLEEP - -1.5 - A - 500 - kHz Input Hysteresis PWMIN Sleep Pull-Up Current[1] OSCILLATOR ROSC = 51 k Oscillator Frequency fOSC 315 350 385 kHz Oscillator Frequency Range[3] fOSC 70 - 700 kHz OSC Input Low Voltage VOIL - - 0.8 V OSC Input High Voltage VOIH 2 - - V OSC Watchdog Period tOSWD 17 - - s ROSC = 73.4 k Between successive rising edges Continued on the next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 5 A6271 Automotive High-Current LED Controller ELECTRICAL CHARACTERISTICS (continued): Valid at TJ = -40C to 150C, VIN = 5 to 45V, unless noted otherwise. Characteristics Symbol Test Conditions Min. Typ. Max. Unit LED CURRENT SENSE Input Bias Current LN ILN VLP = VLN = 12 V - 5 - A Input Bias Current LP ILP VLP = VLN = 12 V - 200 - A 200 204 208 mV 5 - 53.3 V 24.8 29 35 ms VIREF = 0.5 V - 102 - mV VIREF = 0.25 V 47 51 55 mV Differential Sense Voltage VIDL Input Common-Mode Range PWMIN = high, VIDL = VLP - VLN, IREF > 1.2 V VCMLH VLP = VLN tDISAN PWMIN = low ANALOG DIMMING Disable Time Differential Sense Voltage VIDL IREF Maximum Voltage VIREFMAX Corresponds to sense voltage = 200 mV - 1 - V IREF Minimum Voltage VIREFMIN Corresponds to sense voltage = 0 mV - 0 - V PWMIN to LED Turn-On Time tDIMON CL = 2 nF between PWMOUT and LN - 270 - ns PWMIN to LED Turn-Off Time tDIMOFF CL = 2 nF between PWMOUT and LN - 210 - ns PWMOUT Low Voltage VPWMLO LED on, PWMOUT wrt LP, VIN = 10 V -9 - -6.5 V Peak Pull-Up Current[1] IPULLUP PWMIN = low, PWMOUT wrt LP = 0 V - -25 - mA Peak Pull-Down Current IPULLDOWN PWMIN = high, PWMOUT wrt LP = -8 V - 50 - mA 24.8 29 35 ms - 1000 - Hz PWM DIMMING: INTERNAL AND EXTERNAL PWM DIMMING: EXTERNAL Disable Time tDISEPWM PWMIN = low PWM DIMMING: INTERNAL Maximum PWM Dimming Frequency fPWM Minimum PWM Dimming Frequency fPWM PWM Dimming Frequency fPWM PWM Duty Cycle 200 - Hz 200 220 Hz TJ = 25C 87 90 93 % TJ = 150C - 90 - % TJ = 25C 4.5 5 5.5 % TJ = 150C - 5 - % TJ = 25C - 0.3 - % Minimum voltage on DR for 100% duty cycle - 3.6 - V 12.4 14.5 17.5 ms - -1 - A DPWM90 VDR = 3.24 V, fPWM = 200 Hz DPWM5 VDR = 180 mV, fPWM = 200 Hz DPWM0 VDR = 0 V, fPWM = 200 Hz VDRDCMAX Disable Time - 180 70 k between PWMIN and GND tDISIPWM PWMIN = low ISOURCE Coming out of sleep mode SOFT-START Startup Ramp Up Source Current[1] Ramp Up Threshold Ramp Down Threshold VRAMPUP - 1 - V VRAMPDOWN - 100 - mV -20 - - A SWITCH CURRENT SENSE AND AMPLIFIER Input Bias Current [1] Switch Current Overload Threshold Voltage Gain IBIASS Voltage[3] VSP = 300mV, RSLOPE = 1.5k VIDS 375 400 435 mV ACS - 2.25 - V/V Continued on the next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 6 A6271 Automotive High-Current LED Controller ELECTRICAL CHARACTERISTICS (continued): Valid at TJ = -40C to 150C, VIN = 5 to 45V, unless noted otherwise. Characteristics Symbol Test Conditions Min. Typ. Max. Unit -116 - -93 A AVEA - 62 - dB SLOPE COMPENSATION Peak Current [1] ISLOPE Sawtooth current waveform added to currentsense input (SP) GM AMPLIFIER Open Loop DC Gain Transconductance gmCOMP 550 750 950 A/V COMP Source/Sink Current [1] ICOMP - 50 - A COMP Leakage Current [1] ILCOMP - 200 - nA fDITH 7.9 9.6 11.2 kHz DITHER GENERATOR Dither Modulation Frequency Maximum Switching Frequency fOSCMAX ROSC = 72 k, RDITH = 110 k 348 400 452 kHz Minimum Switching Frequency fOSCMIN ROSC = 72 k, RDITH = 110 k 261 300 339 kHz - 3 - ms 3.9 - 4.2 V 250 - 380 mV 3.25 - 3.5 V - 300 - mV 260 320 380 mV PROTECTION FEATURES Fault Blank Timer[4] VIN Undervoltage Turn-Off VIN Undervoltage Hysteresis VREG Undervoltage Turn-Off VREG Undervoltage Hysteresis LED Overcurrent Threshold tFB VINUV Startup Decreasing VIN, IREG = 2 mA DVINUV VREGUV Decreasing VREG DVREGUV VOCLED LP wrt LN Fixed-Output Overvoltage Threshold VFOOV Monitored at LP pin with respect to GND 53.3 55.5 57 V Programmable-Output Overvoltage Threshold VPOOV OVUV wrt LN -1.24 -1.1 -1 V Output Undervoltage Threshold VOUV OVUV wrt LN -300 - - mV Switch Current Overload Period tSCOP Inner loop switch current - 64 - clock cycles LED Overcurrent Period tOPI - 2 - clock cycles LED Output Undervoltage Period tOPV - 30 - clock cycles Hiccup Shutdown Period tHIC 22 26.5 31.75 ms LED overcurrent, or output undervoltage, or overvoltage, or switch overload PWMOUT Undervoltage Turn-On VPWMUVON Measured at LP wrt GND - - 6 V PWMOUT Undervoltage Turn-Off VPWMUVOFF Measured at LP wrt GND 3.7 - 5.8 V 155 170 - C - 20 - C Overtemperature Shutdown Threshold TJF Temperature increasing Overtemperature Hysteresis DTJ Recovery = TJF - DTJ [1] For input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin. Function is correct, but some parameters may not meet specification. guaranteed by design and characterization. [4] Fault blank timer only enabled for either output undervoltage or switch current overload. [2] [3] Parameters Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 7 A6271 Automotive High-Current LED Controller FUNCTIONAL DESCRIPTION The A6271 is a DC-DC converter controller designed to drive series-connected high-power LEDs in automotive applications. The A6271 can be configured in a variety of switching topologies, including: boost, buck-boost, and buck (ground-referenced switch). For each switching configuration, the appropriate loop compensation (COMP) and slope compensation (SLOPE) passive components are selected for optimal performance. The A6271 integrates all the necessary control elements to provide a cost-effective solution using an external logic-level, n-channel MOSFET (switching device), p-channel MOSFET (PWM device), and minimum additional external passive components. The maximum LED current is set with a single external sense resistor and can be accurately modulated using a current reference input (analog control). Direct PWM control is possible via the PWMIN input, which also provides a shutdown mode. Circuit Operation CONVERTER The controller is based on a fixed-frequency, peak current-mode control architecture. There are two loops within the controller. The inner loop, formed by the amplifier AC (refer to Functional Block Diagram), the slope generator, the comparator, CF, and the RS bistable, controls the inductor current as measured through the switching MOSFET by the sense resistor RSS. The outer loop, formed by the amplifier AA and the integrating GM amplifier AB, controls the average LED current by providing the current demand signal for the inner loop. increases, causing the current demand signal to decrease. This reduces the amount of energy transferred to the LED load by terminating the switch current sooner and reducing the LED current. EXTERNAL PULSE-WIDTH MODULATION DIMMING The DR pin should be pulled to VREG. During PWM operation, when PWMIN is pulled low, the LED stack PWMOUT is pulled high with respect to LP, turning off the external p-channel MOSFET, isolating the LED string. In addition, the GM output (amplifier AB) is `parked' (COMP components disconnected) at the new level and the gate drive (SG) is disabled. As the output capacitance is isolated from the LED string, there is no loss of charge. When PWMIN goes high impedance, or is pulled high, the COMP components are reconnected (with the previous `parked' value'), the gate drive (SG) is enabled, PWMOUT is pulled to around -8V with respect to LP turning on the external MOSFET and allowing current to flow through the LED string. INTERNAL PULSE-WIDTH MODULATION DIMMING Where an external PWM signal is not available, the internal PWM generator can be used for controlling the LED brightness. A resistor connected between the PWMIN pin and GND sets the PWM frequency according to the following formula: RFREQ = 14,000 fPWM The LED current is measured by the sense resistor, RSL, and is averaged and amplified to a level where it is compared to the internal reference current to produce an error signal at the output of the GM amplifier, AB. This error signal is effectively the current demand signal and determines the amount of energy transferred to the LEDs on a cycle-by-cycle basis via the inner loop. where RFREQ is in k and fPWM is in Hz. The control loops work together as follows: at the beginning of each oscillator cycle, the bistable is set and the switching MOSFET is on. The switch current builds up due to the voltage developed across the inductor, and when the corresponding signal produced at the output of amplifier AC reaches the current demand level on the output of amplifier AB, the bistable is reset and the switching MOSFET is turned off. The cycle is repeated on the next oscillator cycle. The relationship between the DR voltage and the duty cycle is as follows: If the current through the LEDs increases, the output of AA The duty cycle is controlled by applying a voltage to the DR pin. The VREG can be used for the supply voltage and a potential divider can be used to set the DR voltage. An additional resistor can be added in parallel via a MOSFET switch between DR and GND to change the duty cycle between two levels. PWM Duty cycle (%) = 27.81 x DR voltage So, for example, with a DR voltage = 1.8V, the programmed duty cycle = 50%. In terms of the control of the external MOSFET via the PWMOUT pin, the control is identical as the external PWM scheme. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 8 A6271 Automotive High-Current LED Controller When using the internal PWM scheme, an n-channel MOSFET is required to open the ground connection of the resistor connected between PWMIN and GND to ensure that startup occurs. The gate of the MOSFET is connected to VREG as shown in Figures13 and14, or to an external control signal as shown in Figures9 and11. As the PWMIN input has a pull-up of only 1.1A in sleep mode, it is essential that the zero gate voltage, drain current (leakage) of the MOSFET does not exceed this number at maximum ambient temperatures. Csoft = where tsoft is the desired soft-start period. If analog dimming is applied, the equivalent current source from this circuit will add to the internal 1A source current on the IREF node. Generally speaking, when using analog dimming via VREG and a potential divider, no soft-start or negligible soft-start is provided as shown in the example below. References are taken from Figure 9 on page 23: ANALOG DIMMING The IREF pin can then be used for full analog control. The LED current can be linearly adjusted from zero to full (100%) LED current (ILED) by changing the IREF pin from 0 to 1V. This feature is useful in applications where PWM control is either not required or not available and the LEDs require some dynamic correction for brightness adjustment. Analog dimming can be used along with either pulse-widthmodulation technique, internal or external. This is useful for applications where some color correction is required along with brightness control. Soft-start can be provided via the analog dim signal when either coming out of sleep or hiccup mode. The internal 1A internal source current on the IREF node can be overridden by applying a ramp signal to IREF. The soft-start duration is controlled by the signal on IREF as it is ramped from 0 to 1V. -6 tsoft x 1 x 10 1.2 1 A R5 20 k VREG (5 V) R7 10 k C6 22 nF IREF Pin Figure 1 From the above diagram, VREG, R5, and R7 can be simplified using Norton's Theorem. The equivalent resistance can be found: RT = 20 x 10 = 6.67 k 20 + 10 The current source can be found: Isource= If no soft-start is required, the IREF pin should be connected to VREG. If no internal PWM is required, the DR pin should be connected to VREG. 5 = 750 A RT 1 A SOFT-START When the A6271 comes out of sleep mode, soft-start is required to bring the output voltage up in a controlled open-loop fashion. This minimizes the possibility of the control loop saturating during the startup phase and subsequent output voltage overshoot, which can induce high transient peak currents in the LED string prior to the loop being brought back into linear control. The soft-start period can be programmed by the selection of the appropriate capacitor between IREF pin and GND pin according to the following formula: 750 A RT 6.67 k C6 22 nF IREF Pin Figure 2 From the above schematic, it is clear that the 750A current source will dominate and almost no soft-start will be provided. In this particular case, the only option is to resize C6, or increase the values of R5 and R7, or both. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 9 A6271 Automotive High-Current LED Controller LED CURRENT-SENSE RESISTOR 5 V REGULATOR, VREG The LED current is programmed by the LED sense resistor, RSL, according to: To provide a filtered output and to ensure the regulator is stable, a 1F ceramic capacitor is required to be connected between VREG and GND. The ceramic type should be a quality type such as X5R, X7R, or X8R. ILED = VIDL RSL where the loop typically regulates VIDL to 200mV when in either internal or external PWM modes. The power loss of the resistor should be taken into account to ensure the correct package size is selected. The power loss of the LED current-sense resistor, RSL is: P = ILED2 x RSL It is advisable to insert a 150 resistor in series with the LN pin, as shown below, to protect the internal ESD structures between LN and LP under certain fault conditions. The 150 value is selected as a balance between limiting the fault current and minimizing the LED current error caused by the bias current flowing into the LN pin. The 5V regulator is sized for driving the external switching MOSFET. However, it can be used for functions that require minimal current, e.g. pulling up the FAULTn output and providing a reference for the DR, the IREF pin, or both. To check the load that the MOSFET provides, it is necessary to check the total gate charge required for a 5V drive. This can be derived from the gate charge, QG, versus gate drive voltage, VGS, from the MOSFET datasheet. Once the gate charge is found, the regulator load current can be determined: ILOAD = (QG x fSW )+ Iexternal where Iexternal is the additional circuitry added to the VREG output. The ILOAD should not exceed the VREG external current limit (IREGCL). LP 750 700 A6271 650 600 150 Figure 3 SLEEP MODE If PWMIN is held low for longer than the disable time, tDIS1 or tDIS2, then the A6271 will shut down and put the majority of the circuitry into a low-power sleep mode. When internal PWM dimming is used, the disable time, tDIS1, is 14.5ms. When either external PWM dimming or analog dimming is used, the disable time, tDIS2 is 29ms. Oscillator Frequency, FOSC (kHz) LN RSL 550 500 450 400 350 300 250 200 150 100 50 0 50 100 150 200 250 300 350 400 Oscillator Resistor Value, ROSC (k) Figure 4: ROSC Required for a Particular Oscillator Frequency Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 10 A6271 Automotive High-Current LED Controller OSCILLATOR The main oscillator may be configured as a clock source or it may be driven by an external clock signal. The oscillator is designed to run between 70 and 700kHz. When the oscillator is configured as a clock source, the frequency is programmed via an external resistor between OSC pin and GND pin. The appropriate resistor can be found: ROSC = 25,690 fOSC ROSC = ROSC = Therefore, RDITH can be found from: f = 22 x RDITH = 22 x Figure 4 shows the resulting ROSC for various frequencies. If the oscillator pin goes either open circuit or short circuits to GND, the running frequency defaults to 350kHz. ROSC RDITH 72 = 110 k 14.3 The switching frequency is modulated at a rate of 10kHz via a triangular waveform. This means in one modulation cycle, the switching frequency varies linearly from a minimum to a maximum to a minimum again. If the dither feature is not required, the DITH pin should be tied to VREG. PROTECTION FREQUENCY DITHERING To assist in minimizing EMI emissions, the main oscillator can be dithered so that the energy is spread over a defined frequency band. The defined frequency band is effectively the minimum and maximum switching frequency selected. This frequency is varied above and below the selected oscillator frequency and is set via a resistor connected between Dither pin and GND pin. The frequency band can be selected as follows: f = 22 x 25,690 = 73.4 k , say 72 k 350 f as a percentage of the delta with respect to the oscillator frequency is (50 / 350) x 100% = 14.3%. where ROSC is in k and fOSC is in kHz. When the OSC pin is driven by an external clock source, a number of A6271s can be synchronized together. If the clock period is greater than or equal to 17s, a watchdog circuit causes the running frequency to default to the internal oscillator, which runs at 350kHz. 25,690 fOSC ROSC RDITH where f is a plus/minus percentage change with respect to the oscillator frequency. For example, if an oscillator frequency of 350kHz and a dithered frequency band of 50kHz was selected, given a minimum switching frequency of 300kHz and a maximum switching frequency of 400kHz, the ROSC and RDITHER can be found: The A6271 includes a number of safety features to ensure the controller, the external power components, and the LED string are protected. The Fault Flag becomes active for any fault. When the device recovers from a fault, a soft-start is performed unless analog dimming is selected and the DR pin is tied to VREG. At initial startup, when coming out of sleep mode, or when the hiccup period terminates, a fault blank period, tFB, of 3ms is applied for two fault conditions including low-side switch current limit (inner loop) protection and LED overload protection (caused by an undervoltage), before the fault circuitry becomes active. This period allows steady-state conditions to occur before fault monitoring takes place. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 11 A6271 Automotive High-Current LED Controller Output Overvoltage Protection Two overvoltage protection circuits exist: an internal fixed circuit and an externally programmable circuit. In the majority of applications, the externally programmable circuit will provide the protection. The internal circuit is present in the event that the external feedback resistor chain of the programmable circuit goes open circuit. This feature is particularly desirable in systems that require high levels of reliability and the ability to withstand failure modes. Another advantage is the possibility, in lower voltage applications, to select reduced operating voltages for the switching MOSFET, PWM MOSFET, recirculation diode, and output filter capacitors with confidence. If an overvoltage occurs in either of the two circuits, the highside MOSFET drive (PWMOUT) and the low-side MOSFET drive (SG) are immediately disabled and FAULTn is active. After one fault mask switching cycle, the hiccup timer, tHIC, is initiated for a period of 26.5ms. At the beginning of the hiccup period, the IREF node (soft-start) capacitor is discharged immediately. After the hiccup period, an auto-restart is performed under control of the soft-start capacitor. A potential divider is set up between the LP node (output of the converter) and the cathode end of the LED stack. The output of the potential divider is monitored by a comparator referenced to the LN node. Once this voltage decreases below -1V(max), an overvoltage condition is reported. It is recommended that the impedance of the potential divider is kept relatively high, especially in high-voltage LED strings, to minimize the current draw. It should be noted that there is negligible bias current drawn by the comparator monitor circuit. Switcher Output R1 VOVUV = 1 V LED1 OVUV VLEDOV LEDn R2 Cathode: LED Stack Figure 5 R2 can be found: R2 = R1x (VLEDOV - VOVUV ) VOVUV Assume resistor R1 is selected to be 4.3k. VLEDOV is 1.15 x 45 = 52V. VOVUV is a minimum of 1V. From the above formula: R2 = 4.3 x ( 52 - 1 ) = 219 k, select 220 k 1 As an example, consider an LED string which has a maximum LED string voltage of 45V and an output overvoltage (VLEDOV) is to be reported at a minimum of 15% above this value. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 12 A6271 Automotive High-Current LED Controller Overload Protection Input Undervoltage or VREG Undervoltage Protection There are two circuits: If either condition occurs, the low-side MOSFET drive (SG) is disabled and FAULTn is active (assuming VREG is high enough, if FAULTn is pulled to VREG). In the case of the input undervoltage, the high-side MOSFET drive (PWMOUT) is also disabled. 1. LED overcurrent threshold 2. Output undervoltage threshold In the case of a LED overcurrent fault, the high-side MOSFET drive (PWMOUT) and the low-side MOSFET drive (SG) are disabled after two fault mask switching cycles, FAULTn is active, the IREF node (soft-start) capacitor is discharged, then the hiccup timer, tHIC, is initiated for a period of 26.5ms. After the hiccup period, an auto-restart is performed under control of the soft-start capacitor. In the case of an output undervoltage fault, the high-side MOSFET drive (PWMOUT) is immediately disabled and FAULTn is active. After thirty fault mask switching cycles, the low-side MOSFET drive (SG) is disabled, IREF node (soft-start) capacitor is discharged, and the hiccup timer, tHIC, is initiated for a period of 26.5ms. After the hiccup period, an auto-restart is performed under control of the soft-start capacitor. Low-Side Switch Current Limit (inner loop) At startup, a 3ms blank period is applied before the circuitry becomes active. Cycle-by-cycle current protection is provided through the low-side MOSFET. If an overcurrent occurs for longer than 64switching clock cycles, the high-side MOSFET drive (PWMOUT) and the low-side MOSFET drive (SG) are disabled, FAULTn is active, and the hiccup timer, tHIC, is initiated for a period of 26.5ms. During the hiccup period, the IREF node (soft-start) capacitor is discharged immediately. After the hiccup period, an auto-restart is performed under control of the soft-start capacitor. Both the input voltage and the VREG voltage must rise above their respective turn-on thresholds before a restart is possible. In the case of startup through the input voltage turn-on threshold, the system is brought up under control of the soft-start. In the case of startup through the VREG turn-on threshold, no soft-start is provided. PWM Output Undervoltage During startup, the output (LP node) must increase above 6V to ensure the high-side MOSFET turns on. This is generally not a problem with switching topologies that can boost the output voltage with respect to the input voltage. In the case of the buck topology, the LN/LP node is referenced to VIN; therefore, the input voltage has to be equal to or greater than 6V to guarantee a successful startup. Overtemperature Shutdown If the chip exceeds the overtemperature shutdown threshold, the low-side MOSFET drive (SG) is immediately disabled, FAULTn is active, and the IREF node (soft-start) capacitor is discharged immediately. An auto-restart is performed under control of the soft-start capacitor once the temperature drops below the overtemperature minus the hysteresis level. The table on the following page summarizes the above faults along with other pin specific faults. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 13 A6271 Automotive High-Current LED Controller Table 1: Fault Table Fault Action Low-Side Switch Current Limit When fault occurs, cycle-by-cycle current limit operates. If fault >64 counts: low-side MOSFET (SG) and PWM MOSFET (PWMOUT) off and FAULTn active, hiccup period, then auto-restart with soft-start. Note: fault blanked for 3ms during startup. LED Overcurrent Low-side MOSFET (SG) and PWM MOSFET (PWMOUT) immediately off and FAULTn active, hiccup period after 2 counts, then auto-restart with soft-start. Output Undervoltage Low-side MOSFET (SG) and PWM MOSFET (PWMOUT) immediately off. If fault > 30 counts: FAULTn active, hiccup period, then auto-restart with soft-start. Note: fault blanked for 3ms during startup. Fixed-Output Overvoltage Low-side MOSFET (SG) off and PWM MOSFET (PWMOUT) immediately turns off and FAULTn active, hiccup period after 1 count, then auto-restart with soft-start. Programmable-Output Overvoltage Low-side MOSFET (SG) off and PWM MOSFET (PWMOUT) immediately turns off and FAULTn active, hiccup period after 1 count, then auto-restart with soft-start. Input Undervoltage Low-side MOSFET (SG) and PWM MOSFET (PWMOUT) immediately turns off and FAULTn active assuming there is sufficient drive to the flag. Once input voltage is above the VIN undervoltage threshold, plus hysteresis, auto-restart with soft-start occurs. VREG Undervoltage Low-side MOSFET (SG) immediately turns off and FAULTn active assuming there is sufficient drive to the flag. Once VREG voltage is above the VREG undervoltage threshold, plus hysteresis, then auto-restart. Thermal Shutdown Low-side MOSFET (SG) immediately turns off and FAULTn active. Auto-restart with soft start occurs after the temperature drops below the overtemperature minus hysteresis level. PWMOUT Undervoltage Low-side MOSFET (SG) off and PWM MOSFET (PWMOUT) off immediately and FAULTn active. Auto-restart with soft-start occurs. OSC Pin Fault The oscillator will switch to default frequency of 350kHz. COMP Short to GND Force regulator to minimum duty cycle. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 14 A6271 Automotive High-Current LED Controller COMPONENT SELECTION Inductor The ripple current, I = 0.15 x IAVE . The main factor in selecting the inductor value is to target a certain ripple current to ensure the peak current-mode control works correctly. A reasonable figure is a peak-to-peak ripple current of around 15% of the average inductor current. The maximum inductor current occurs at minimum input voltage and maximum duty cycle. The minimum inductance can now be found: BOOST INDUCTOR SELECTION The maximum duty cycle can be found: DMAX = VLED + (Vf - VIN(MIN) ) VLED + Vf where VLED is the LED output voltage, Vf is the forward voltage drop of the recirculation diode, and VIN(MIN) is the minimum input voltage. The maximum average inductor current can be determined: IAVE = ILED (1 - DMAX ) The ripple current, I = 0.15xIAVE . The minimum inductance can now be found: L= (VLED + Vf - VIN(MIN) )x (1 - DMAX ) I x fSW where fSW is the switching frequency. The peak current in the inductor is: I ILPK = IAVE + 2 BUCK-BOOST INDUCTOR SELECTION The maximum duty cycle can be found: DMAX = VLED + Vf VLED + Vf + VIN(MIN) where VLED is the LED output voltage, Vf is the forward voltage drop of the recirculation diode, and VIN(MIN) is the minimum input voltage. The maximum average inductor current can be determined: IAVE = ILED 1 - DMAX L= VIN(MIN)x DMAX I x fSW where fSW is the switching frequency. The peak current in the inductor is: I 2 When selecting an inductor from manufacturers' datasheets, there are often two current ratings given: ILPK = IAVE + 1. Saturation current. This is the current level that causes the inductance to drop by between 10 and 40% depending on the manufacturer. The saturation current should be greater than the peak current, ILPK, with some margin to allow for overload conditions. 2. RMS or average current. This is the current level that determines a certain temperature rise in the inductor with a given ambient temperature. This is normally presented as a single figure: operating temperature. The RMS or average inductor current rating should be greater than the estimated maximum average current, IAVE. Recommended inductor manufacturers: * Coilcraft: MSS1278T or MSS1078T Range * TDK: SLF12575 type H SWITCH CURRENT SENSE The switch current sense of the `inner loop' is measured by the external sense resistor, RSS, and the switch sense amplifier, AC. As well as providing the peak current information to determine the duty cycle, it also provides pulse-by-pulse current limiting through the switching MOSFET and slope compensation to prevent subharmonic oscillations at duty cycles greater than 50%. The current limit of the inner loop is set by the input limit of the sense amplifier, VIDS, the maximum switch current that has been determined, and the effects of the slope compensation have to be taken into account. The operating duty cycle has to be calculated at maximum load and minimum operating input voltage. The amount of slope compensation can be calculated for this operating point and can then be added to the actual current-sense signal to determine the maximum signal amplitude before cycle-bycycle current limiting takes effect. Refer to Slope Compensation Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 15 A6271 Automotive High-Current LED Controller Section to find diL /dt then diSLOPE / dt. RSS = 1.2x ( 0.32 diSLOPE DMAX x ILP + FSW dt ( Buck-Boost Slope Resistor )) Note that the minimum value of VIDS is used with an additional 20% to allow for margin. The inductor down slope is: diL VLED + Vf = dt L The optimum down slope as illustrated by Ridley can be found from: ( ILP is the peak current in the inductor. The power loss of the switch current-sense resistor, RSS, can be found: Boost RSS Power Loss Using the DMAX and IAVE from the boost part of the inductor section, the power loss of RSS can be found: Ploss = IAVE2 x DMAX x RSS ) diSLOPE diL 0.18 = x 1- dt dt DMAX The slope compensation resistor can be found: diSLOPE x RSS dt RSLOPE= -6 -6 100 x 10 x 1 x 10 x fSW where RSLOPE is in ohms (). Buck-Boost RSS Power Loss CONTROL LOOP COMPENSATION Using the DMAX and IAVE from the buck-boost part of the inductor section, the power loss of RSS can be found: The recommended way of closing the control loop is to remove the influence of the right-hand plane zero (RHPZ) in both boost and buck-boost topologies. The reason for this is that the RHPZ increases the gain by 20dB/decade and at the same time introduces a 90-degree phase lag. Ploss = IAVE2 x DMAX x RSS Resistor manufacturers typically derate the devices from an ambient temperature of around 70C. The power rating including derating of the sense resistor should exceed the maximum power loss at maximum ambient temperature. SLOPE COMPENSATION Slope compensation can be added to the MOSFET current-sense signal on pin SP to prevent subharmonic oscillations where the peak-to-average control error becomes increasingly larger at duty cycles in excess of 50%. A current source is provided at the SP pin as a sawtooth from 0 to 100A. An external resistor, RSLOPE, connected between the SP pin and the source connection of the MOSFET, is used to program the appropriate voltage level to scale the slope compensation for correct use with the appropriate topology and set up conditions that have been adopted. Boost Slope Resistor The inductor down slope is: diL VLED + Vf - VIN(MIN) = dt L The minimum frequency that the RHPZ occurs at is: For boost mode: fRHPZ= VLED x (1 - DMAX ) 2 2 x x L x ILED For buck-boost mode: 2 fRHPZ= VLED x (1 - DMAX ) 2 x x L x ILED x DMAX It is recommended that the 0dB crossover point is approximately: fCROSS = fRHPZ 5 With effective peak current-mode control, it can be assumed that the second power pole is pushed high enough in the frequency domain to have no influence on the overall loop response. It is reasonable to assume the overall loop response is effectively a single pole set by the GM amplifier (COMP node). The error Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 16 A6271 Automotive High-Current LED Controller amp zero is set at the same frequency as the output power pole to ensure the loop is closed at a rate of 20dB/decade. The frequency position of the power stage pole and the GM amplifier zero is: The open-loop DC gain of the system can be found: Boost: Boost: DC Gain = ( VI ) ) + R )) 5 x 1,259 x RSL x (1 - DMAX ) x RSS x (( VI ) + (n x R LED dyn LED LED LED SL DC Gain = RSS x (( VI ) + D LED MAX LED ( VI ) LED LED ) x ((n x Rdyn ) + RSL ) where n = number of LEDs and Rdyn = LED dynamic resistance. Note that the LED dynamic resistance may be given in the LED datasheet. If it is not, it can be derived by a simple measurement. Set up a power supply with a current limit at the operating point (ILED1). Apply the current to an individual LED and measure the voltage drop (VLED1). Change the current limit by a small amount, say 5% (ILED2), and measure the voltage drop (VLED2). The dynamic resistance can be estimated: Rdyn = VLED1 - VLED2 ILED1 - I LED2 The RC constant required to achieve 0dB with a slope of 20dB/ decade at the crossover frequency, fCROSS: RC = 1 2 x x fCROSS The frequency of the first GM amplifier pole can be found: fp1 = 1 2 x x RC x DC Gain Capacitor on the output of the GM amplifier (COMP node) required to achieve the above pole position: -6 Ccomp = 750 x 10 2 x x fp1 x 1,258 VLED + ILED x ((n x Rdyn ) + RSL ) 2 x x VLED x COUT x ((n x Rdyn ) + RSL ) Buck-Boost: fp2 and fz1 = VLED + DMAX x ILED x ((n x Rdyn ) + RSL ) 2 x x VLED x COUT x ((n x Rdyn ) + RSL ) The resistor (Rcomp) in series with the compensation capacitor (Ccomp) on the COMP node can be found: Buck-Boost: 5 x 1,259 x RSL x (1 - DMAX) x fp2 and fz1 = Rcomp = 1 2 x x fp2 x Ccomp LOW-SIDE SWITCHING MOSFET A logic-level n-channel MOSFET is used as the switch for the DC-DC converter. In the boost configuration, the maximum voltage across the drain-source connection is: VDS = VLED + Vf In the buck-boost configuration, the maximum voltage across the drain-source connection is: VDS = VLED + Vf + VINMAX The actual rating of the MOSFET selected should be greater than the maximum voltage plus some margin. It is recommended that the minimum margin should be no less than 20% of the maximum voltage. In the case of buck-boost mode, the maximum rating should factor in load-dump conditions. In terms of the current rating, the MOSFET is generally selected for a low RDS rating to minimize the power dissipation. This means the current rating is well in excess of the actual maximum current used in the application. The power loss in the MOSFET is determined by the static loss and the switching losses. Static Loss Using the DMAX and IAVE from the boost or buck-boost part of the inductor section, the power loss of RDS can be found: Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 17 A6271 Automotive High-Current LED Controller Ploss = IAVE2 x DMAX x RDS Note that the RDS figures are generally presented at 25C room ambients. The actual RDS can be determined by considering the normalized RDS versus temperature graph. Another consideration of the static loss is cold-crank situations. It is important to ensure the gate-drive amplitude (derived from VREG) at the minimum input voltage provides sufficient drive that the RDS does not increase by much, therefore minimizing any increase in losses. A good quality logic-level MOSFET should have good RDS performance at drive voltages of less than 4V. The VREG load can be determined by estimating the gate losses. From the MOSFET datasheet, the total gate charge can estimated with a gate drive of 5V using the appropriate graph. In addition, any other circuitry that VREG is powering should also be factored. The current drawn from VREG due to the MOSFET drive can be determined: VREGMOSFETload = QTOTALGate x fSW Switching Losses The switching losses in the MOSFET are determined by the length of time of the Miller region. To minimize conducted and radiated EMI emissions, this region is deliberately extended by adding series resistance between the gate drive (SG) and the gate of the device. It is assumed that the turn-off loss is similar to the turn-on loss. In the case of the boost converter, the switching loss: Pswitch = (VLED + Vf ) x IAVE x tmiller x fSW In the case of the buck boost converter, the switching loss: Pswitch = (VLED + Vf + VIN(MIN) ) x IAVE x tmiller x fSW RECIRCULATION DIODE The diode should have a low forward voltage to reduce conduction losses and a low capacitance to reduce switching losses and minimize EMI. Schottky diodes can provide both features if carefully selected. The forward voltage drop is a natural advantage for Schottky diodes and reduces as the current rating increases. However, as the current rating increases, the diode capacitance also increases, so the optimum selection is usually the lowest current rating above the required maximum, in this case ILPK. In the boost configuration, the maximum reverse voltage across the diode is: VRRM = VLED + Vf In the buck-boost configuration, the maximum reverse voltage across the diode is: VRRM = VLED + Vf + VIN(MAX) The actual rating of the diode selected should be greater than the maximum voltage plus some margin. It is recommended that the minimum margin should be no less than 20% of the maximum voltage. In the case of buck-boost mode, the maximum rating should factor in load-dump conditions. HIGH-SIDE PWM MOSFET A p-channel MOSFET is used as the PWM switch for the LED stack. In both boost and buck-boost modes, the maximum voltage across the drain-source connection is VLED. The actual rating of the MOSFET selected should be greater than the maximum voltage plus some margin. It is recommended that the minimum margin should be no less than 20% of the maximum voltage. The power loss of this MOSFET is dominated by the static loss. The switching losses can largely be ignored as the PWM frequencies are relatively low. The power loss of the MOSFET RDS can be found: Ploss = ILED2 x RDS The gate drive for the PWM MOSFET is derived from the LED output rail (LP pin). In boost and buck-boost modes, this node is boosted with respect to the input voltage (VIN), so there should be sufficient negative gate drive. In other operating modes such as buck, where the output voltage is less than the input voltage, it may be necessary to use low threshold p-channel MOSFETs to ensure adequate overdrive during cold-crank situations. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 18 A6271 Automotive High-Current LED Controller OUTPUT CAPACITOR Layout There are several points to consider when selecting the output capacitor. The following layout guidelines should be followed to ensure satisfactory electrical and EMI performance. Due to the switching topology used, the ripple current for this circuit is high since the output capacitor provides the LED current when the DC-DC converter switch is active in both boost and buck-boost modes. The capacitor is then recharged each time the inductor passes energy to the output. The ripple current on the output capacitor will be equal to the peak inductor current. The corresponding output ripple can be derived from the amount of charge transferred to the output during the switch on time. Ground planes should be used on as many layers as possible. This is essential in minimizing `ground bounce' (differential voltage across the ground connection). `Ground bounce' can lead to radiated noise which can then be picked up on both input and output connections and manifest as common-mode noise. Any ground planes on different layers should be connected using multiple vias in an attempt to minimize ground impedances. The ground tab under the A6271 should also have multiple vias connecting to the ground plane or planes. To minimize heating effects and voltage ripple, the equivalent series resistance (ESR) and the equivalent series inductance (ESL) should be kept as low as possible. This can be achieved by multilayer ceramic chip (MLCC) capacitors. To reduce performance variation over temperature, low drift types such as X7R and X5R should be used. The value of the output capacitor will typically be in the range of 3.3to 10F, and it should be rated above the maximum LED stack voltage, VLED. There is an E-field effect with ceramic capacitors that causes the capacitance to fall at elevated voltages. It is therefore recommended that a good margin is selected to minimize this effect. One potential issue of ceramic capacitors is audible noise during pulse-width modulation (PWM). This is caused by the piezoelectric effect of the ceramic substrate. To minimize the effects of this, it is recommended to use multiple physically smaller capacitors. If this is still an issue, it is recommended that either low-impedance electrolytic or polymer capacitors be used. INPUT CAPACITOR The function of the input filter capacitor is to provide a lowimpedance shunt path for the current drawn by the A6271 when the switching MOSFET turns on. The objective is to minimize the ripple current reflected back into the source supply. This approach helps to minimize conducted emissions into the power source. Additional line impedance in the form of chokes can be added to improve the emissions further. In a correctly designed system, with a quality capacitor or capacitors positioned adjacent to the power train circuitry, these capacitors should supply the ripple current. The drain connection of the switching MOSFET, PWM MOSFET, and cathode terminal of the recirculation diode are used for thermal heatsinking. It is advised to use sufficient copper around these connections on the component layer of the PCB only. The areas directly under these connections on the PCB should form part of the ground plane. The reason for restricting the copper area on these nodes is because they can radiate noise due to the nature of the dv/dt and di/dt power signals that appear. The area of the switching power loops should be minimized as much as possible. In addition, the trace connections should be as wide as possible to minimize parasitic leakage inductances, but at the same time not compromising the power loop area. There are two power loops: Loop 1: formed by the input filter, main switching MOSFET, power inductor, and inner loop sense resistor. Loop 2: formed by the power inductor, recirculation diode, LED sense resistor, PWM MOSFET, and the output capacitor or capacitors. Where practical, keep input or output filter magnetics as far away from the power-switching inductor (L1) as possible. This is to avoid or at least minimize the effects of magnetic crosstalk. One of the major noise contributors is the switching MOSFET (M1). Slowing down the gate drive without compromising the thermal solution will help to minimize noise. To comply with CISPR 25, a common-mode choke is typically required as part of the input filter. The amount of capacitance required at the input is dictated by the EMI performance. This is usually distributed with series ferrite beads and either differential-mode chokes, or common-mode chokes, or both. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 19 A6271 Automotive High-Current LED Controller Reducing EMI INPUT FILTER It is essential that good layout practice as defined in the Layout Section should be adopted. The following techniques are also recommended. The selection of the components that form the input filter depends on the noise that is present in the system in terms of the frequency and whether it is common mode or differential mode. In addition, the common automotive standards that exist define onerous specification limit lines for the emissions in the AM band (approximately 530kHz to 1.7MHz) and the FM band (approximately 70 to 108MHz). Some consideration must be given to these frequency bands to understand how to filter these regions. SNUBBER Adding a low-loss R-C snubber network between the drain of the main switching MOSFET and ground helps to suppress the resonant ringing on the switching node. The process for selecting these components involves some `trial and error' on the actual printed circuit board. Step 1: Measure the voltage resonance frequency on the LX node. Step 2: Add an additional capacitance between LX and ground until the resonant frequency is halved. Note that this capacitance should be around 1nF. Step 3: Two equations with two unknowns are now obtained: FRES = 1 2 x x Lleak x Cleak 1 FRES = 2 x x Lleak x Cnew 2 where Cnew = Cleak + Cadd. Cadd = additional capacitance added. Cleak = parasitic capacitance. Lleak = parasitic inductance. Now to halve the frequency, Cleak + Cadd = 4 x Cleak Therefore, Cleak = Cadd 3 With Cleak solved, Lleak can also be solved. The characteristic impedance of the parasitic components can be found: RO = Lleak Cleak RO can be selected as the damping resistor. Typically, either an 0805 or 0603 resistor case size is adequate. At the lower frequencies, below a few 10s of MHz, the noise is generally dominated by differential noise with some common mode noise. At frequencies above a few 10s of MHz, the noise is dominated by common mode noise with some differential noise. To address the differential noise, a differntial inductor can be used along with differential capacitance to form an L-C filter. One problem in using standard differential inductors is that the self-resonance frequency (SRF) is typically in the region of a few 10s of MHz, even with a modest few microhenries (H) (note: the higher the inductance, the lower the SRF). This means that above the self-resonant frequency points, these components actually amplify the noise and make matters worse. Some differential-mode inductive filtering is always necessary. Ferrite beads can be used for this function. Although ferrite beads are designed to act as a lossy resistor at particular frequency bands, they do have an inherent inductive element which can be in the region of several H. The inductance of a ferrite bead can be extracted from the reactance information graph (refer to Figure7). At a particular frequency, the reactance can be found and then the inductance can be derived. As a single ferrite bead may not be effective enough, a two-stage ferrite bead filter approach can be taken. These components, along with input differential-mode capacitors, can form L-C filter stages. For the best result, the first L-C filter should be placed as close to the power stage as possible. At higher frequencies, the majority of the noise problems is associated with common-mode noise. This noise is induced by ground-referenced differential noise radiating through the ground plane. This noise can be picked up on the input stage forming common-mode noise on the positive and negative power supply connections to the battery. Even with excellent layout, this noise is always present. To address this problem, a common-mode inductor is required. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 20 A6271 Automotive High-Current LED Controller This inductor is selected to present a high impedance around the FM region. An example of a common-mode (CM) choke with a high impedance around the FM region is shown below in Figure 6. Further improvements to the differential-mode performance can be made by the use of frequency dithering techniques. The A6271 contains a dither circuit which changes the switching oscillator frequency on a cycle-by-cycle basis across a defined frequency band. 10000 As the noise in a switcher is typically narrow-band noise, both the peak and average signals are similar in amplitude. When a frequency dither scheme is introduced, it `spreads' the noise, converting it from narrow band into broad band. While the peak noise reduces across the majority of the spectrum, the reduction in the average measurement is a lot more effective. This effect is particularly helpful in the conducted emissions for the average measurement between 76 and 108MHz. This region is unusual, as typically the peak and average limit lines track one another, but in this area, while the peak limit line increases, the average limit line reduces (refer to Figure 7 below). Note the red limit lines are peak and the pink limit lines are average. 1000 Impedance () FREQUENCY DITHERING 100 10 90 80 1 1 10 100 1000 Limits 55025 P5 55025 A5 70 Frequency (MHz) Z (comm) 60 Z (diff) Figure 6: Common-Mode (CM) Choke Impedance Another important consideration is the relative positioning of the common-mode choke with respect to the switching inductor. Magnetic crosstalk can occur between these components which can degrade the effectiveness of the CM choke. Even the use of a magnetically-screened switching inductor is not sufficient to avoid this problem. It is important to physically separate these two components as far as possible. Another advantage of a good physical separation is that any `ground bounce' induced CM noise will couple onto the input power traces. The strength of the coupling will reduce with distance. The further the CM choke is away from the switching circuit, the more likely it will be to filter this noise. All filter capacitors should be a quality ceramic: X7R or X8R. dBV 50 Transducer UniRFPro 40 30 Traces PK+ AV 20 10 0 -10 150 kHz 1 MHz 10 MHz 108 MHz Figure 7: CISPR 25 Class 5 Limit Lines The reason that the average limit line is relatively low (making it challenging to pass) is that average weighted signals have an adverse effect on FM radio signals. Minimizing the noise at the high end of the conducted emission frequency spectrum also benefits the radiated noise at this frequency band and above. Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 21 A6271 Automotive High-Current LED Controller Note that a modulation frequency of 10kHz was chosen, since this aligns with the resolution bandwidth of the measurement receiver as defined in CISPR 25. The resolution bandwidth is effectively the `measurement window' at each measurement step. the dithering is. However, there is a trade-off with switching losses and sizing of the power inductor in terms of inductance value and the corresponding physical size. Another consideration when optimizing the frequency dithering is the depth of frequency. This is the maximum and minimum switching frequency that the converter operates, effectively the `spread range'. The wider the spread range is, the more effective Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 22 A6271 Automotive High-Current LED Controller APPLICATION CIRCUITS L1 12 H D2 VBAT C2 4.7 F C1 4.7 F C3 100 nF D1 C9 47 nF C4 100 nF R13 470 m R11 150 R14 2.7 R12 10 k VREG C5 1 F R1 2.7 k M4 IREF SG C6 22 nF OSC R8 100 k DITH R9 110 k LED1 OVUV SP R5 20 k R7 10 k A6271 PWMIN Analog Control R6 10 k PWMOUT R15 4.3 k DR R4 68 k M2 LN FAULTn R3 10 k High = internal PWM ON (5%) LP GND R2 270 k M1 VIN COMP R10 39 C7 470 nF R16 0 C10 C11 C12 4.7 F 4.7 F 4.7 F M3 LED12 R21 220 k R17 1.2 k R18 R19 R20 C8 10 k 68 m 68 m 22 pF GND Figure 8: Boost Driving 12LEDs at 500mA, Switching Frequency 250kHz Internal PWM (5%) and/or analog dimming, no soft-start, and frequency dither on. The minimum startup voltage is determined by the voltage on the output (LP) node. 6V is required at this node, so the input voltage, VIN = 6V + Vf (of recirculation diode, D2). If for example, Vf = 0.4V, then VIN = 6.4V. Once operating, the A6271 will function down to an input voltage of 4.5V. The maximum operating voltage on the input is the maximum LED string voltage, plus the diode drop of D2. Table 2: Application Circuit 1 Bill of Materials Reference Description Manufacturer/Part Number C1,C2,C10,C11,C12 4.7F, ceramic capacitor, X7R, 50V TDK, MuRata C3,C4 100nF, ceramic capacitor, X7R, 50V C5 1F, ceramic capacitor, X7R, 16V C6 22nF, ceramic capacitor, X7R, 50V C7 470nF, ceramic capacitor, X7R, 50V C8 22pF, ceramic capacitor, X7R, 50V C9 47nF, ceramic capacitor, X7R, 16V D1 200mA, 30V Schottky diode NXP, ON Semiconductor, Fairchild / BAT54 D2 10A, 60V Schottky diode Vishay / SS10P6 L1 22H, high current shielded construction Vishay / IHLP-5050FDER220M-5A M1,M2 N-channel signal MOSFET ON Semiconductor, IR / NTR4003N M3 N-channel 50A, 100V MOSFET Vishay / SQD50N10 Continued on next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 23 A6271 Automotive High-Current LED Controller Table 2: Application Circuit 1 Bill of Materials (continued) Reference Description M4 P-channel 15A, 100V MOSFET Infineon / SPD15P10PL G Heatsink for M3 (TO-252) AAVID Thermalloy / 573100D00010G R1 2.7k, 1%, 0603 or 0805 R2 270k, 1%, 0603 or 0805 R3,R6,R12,R18 10k, 1%, 0603 or 0805 R4 68k, 1%, 0603 or 0805 R5 20k, 1%, 0603 or 0805 R7 10k, potentiometer R8 100k, 1%, 0603 or 0805 R9 110k, 1%, 0603 or 0805 R10 39, 1%, 0603 or 0805 R11 150, 1%, 0603 or 0805 R13 470m, 1%, 0805 or 1206 R14 2.7, 1%, 0603 or 0805 R15 4.3k, 1%, 0603 or 0805 R16 0 , 0603 or 0805 R17 1.2k, 1%, 0603 or 0805 R19,R20 68m, 1%, 2010 R21 220k, 1%, 0603 or 0805 Manufacturer/Part Number Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 24 A6271 Automotive High-Current LED Controller L1 56 H D2 VBAT C1 4.7 F C2 4.7 F C3 100 nF D1 C9 47 nF C4 100 nF R7 2.7 R5 150 R8 2.7 R6 10 k R1 2.7 k C5 1 F VREG VIN LP LN PWMOUT M2 GND FAULTn R9 4.3 k DR External PWM Drive Signal A6271 OVUV SG PWMIN SP IREF C6 22 nF OSC R2 73.2 k DITH R3 110 k LED1 COMP R4 39 C7 680 nF R10 12 M1 C10 C11 C12 2.2 F 2.2 F 2.2 F R15 240 k R11 1.3 k R12 R13 10 k 150 m C8 22 pF LED14 R14 150 m GND Figure 9: Boost Driving 14LEDs at 150mA, Switching Frequency 350kHz External PWM, no analog dimming, soft-start, and frequency dither on. The minimum startup voltage is determined by the voltage on the output (LP) node. 6V is required at this node, so the input voltage, VIN = 6V + Vf (of recirculation diode, D2). If for example, Vf = 0.4V, then VIN = 6.4V. Once operating, the A6271 will function down to an input voltage of 4.5V. The maximum operating voltage on the input is the maximum LED string voltage, plus the diode drop of D2. Table 3: Application Circuit 2 Bill of Materials Reference Description Manufacturer/Part Number C1,C2 4.7F, ceramic capacitor, X7R, 50V TDK, MuRata C3,C4 100nF, ceramic capacitor, X7R, 50V C5 1F, ceramic capacitor, X7R, 16V C6 22nF, ceramic capacitor, X7R, 50V C7 680nF, ceramic capacitor, X7R, 50V C8 22pF, ceramic capacitor, X7R, 50V C9 47nF, ceramic capacitor, X7R, 16V C10,C11,C12 2.2F, ceramic capacitor, X7R, 100V TDK, MuRata D1 200mA, 30V Schottky diode NXP, ON Semiconductor, Fairchild / BAT54 D2 2A, 100V Schottky diode Vishay, ST / SS2H10 L1 56H, power inductor shielded construction Coilcraft / MSS1048T-563ML M1 N-channel, 30A, 100V MOSFET NXP / PSMN038-100YLX Continued on next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 25 A6271 Automotive High-Current LED Controller Table 3: Application Circuit 2 Bill of Materials (continued) Reference Description Manufacturer/Part Number M2 P-channel 15A, 100V MOSFET Infineon / SPD15P10PL G R1 2.7k, 1%, 0603 or 0805 R2 73.2k, 1%, 0603 or 0805 R3 110k, 1%, 0603 or 0805 R4 39, 1%, 0603 or 0805 R5 150, 1%, 0603 or 0805 R6,R12 10k, 1%, 0603 or 0805 R7,R8 2.7, 1%, 0805 or 1206 R9 4.3k, 1%, 0603 or 0805 R10 12, 1%, 0603 or 0805 R11 1.3k, 1%, 0603 or 0805 R13,R14 150m, 1%, 1206 R15 240k, 1%, 0603 or 0805 Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 26 A6271 Automotive High-Current LED Controller L1 12 H D2 VBAT C1 4.7 F C2 4.7 F C3 100 nF D1 C9 47 nF C4 100 nF R11 270 m R9 150 R12 270 m R10 10 k VREG C5 1 F R1 2.7 k LN PWMOUT LED1 DR R4 68 k A6271 SG PWMIN R5 10 k IREF C6 22 nF OSC R6 100 k DITH R7 110 k R13 4.3 k OVUV SP M2 M4 FAULTn R3 10 k High = internal PWM ON (5%) LP GND R2 270 k M1 VIN COMP R8 39 C7 220 nF R14 0 M3 C10 C11 C12 4.7 F 4.7 F 4.7 F LED5 R19 84.5 k R15 1 k R16 R17 R18 10 k 62 m 62 m C8 22 pF GND Figure 10: Buck-Boost Driving 5LEDs at 1.5A, Switching Frequency 250kHz Internal PWM (5%), no analog dimming, soft-start, and frequency dither on. The minimum start up voltage is determined by the voltage on the output (LP) node. 6V is required at this node, so the input voltage, VIN = 6V + Vf (of recirculation diode, D2). If for example, Vf = 0.4V, then VIN = 6.4V. Once operating, the A6271 will function down to an input voltage of 5.5V. The maximum operating voltage on the input is 36V, plus the diode drop of D2. Table 4: Application Circuit 3 Bill of Materials Reference Description Manufacturer/Part Number C1,C2,C10,C11,C12 4.7F, ceramic capacitor, X7R, 50V TDK, MuRata C3,C4 100nF, ceramic capacitor, X7R, 50V C5 1F, ceramic capacitor, X7R, 16V C6 22nF, ceramic capacitor, X7R, 50V C7 220nF, ceramic capacitor, X7R, 50V C8 22pF, ceramic capacitor, X7R, 50V C9 47nF, ceramic capacitor, X7R, 16V D1 200mA, 30 V Schottky diode NXP, ON Semiconductor, Fairchild / BAT54 D2 10A, 60V Schottky diode Vishay / SS10P6 L1 12H, high current shielded construction Vishay / IHLP-5050FDER120M-5A M1,M2 N-channel signal MOSFET ON Semiconductor, IR / NTR4003N M3 N-channel 50A, 100V MOSFET Vishay / SQD50N10 Continued on next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 27 A6271 Automotive High-Current LED Controller Table 4: Application Circuit 3 Bill of Materials (continued) Reference Description M4 P-channel 15A, 100V MOSFET Infineon / SPD15P10PL G Heatsink for M3 (TO-252) AAVID Thermalloy / 573100D00010G R1 2.7k, 1%, 0603 or 0805 R2 270k, 1%, 0603 or 0805 R3,R5,R10,R16 10k, 1%, 0603 or 0805 R4 68k, 1%, 0603 or 0805 R6 100k, 1%, 0603 or 0805 R7 110k, 1%, 0603 or 0805 R8 39, 1%, 0603 or 0805 R9 150, 1%, 0603 or 0805 R11,R12 270m, 1%, 0805 or 1206 R13 4.3k, 1%, 0603 or 0805 R14 0 , 0603 or 0805 R15 1k, 1%, 0603 or 0805 R17,R18 62m, 1%, 2010 R19 84.5k, 1%, 0603 or 0805 Manufacturer/Part Number Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 28 A6271 Automotive High-Current LED Controller L1 47 H D2 VBAT C1 4.7 F C2 4.7 F C3 100 nF D1 C4 47 nF C4 100 nF R8 1 R6 150 R9 1 R7 10 k R1 2.7 k C5 1 F VIN VREG LP LN PWMOUT M2 GND FAULTn DR External PWM Drive Signal A6271 OVUV SG PWMIN SP R2 20 k R3 10 k IREF Analog Control C6 22 nF OSC R4 100 k DITH R10 4.3 k COMP R5 27 C7 470 nF R11 24 M1 LED1 C10 C11 C12 4.7 F 4.7 F 4.7 F LED4 R16 68 k R12 1 k R13 R14 10 k 180 m C8 22 pF R15 180 m GND Figure 11: Buck-Boost Driving 4LEDs at 400mA, Switching Frequency 250kHz External PWM and/or analog dimming, no soft-start, and frequency dither off. The minimum startup voltage is determined by the voltage on the output (LP) node. 6V is required at this node, so the input voltage is VIN = 6V + Vf (of recirculation diode, D2). If for example, Vf = 0.4V, then VIN = 6.4V. Once operating, the A6271 will function down to an input voltage of 4.5V. The maximum operating voltage on the input is 40V, plus the diode drop of D2. Table 5: Application Circuit 4 Bill of Materials Reference Description Manufacturer/Part Number C1,C2,C10,C11,C12 4.7F, ceramic capacitor, X7R, 50V TDK, MuRata C3,C4 100nF, ceramic capacitor, X7R, 50V C5 1F, ceramic capacitor, X7R, 16V C6 22nF, ceramic capacitor, X7R, 50V C7 470nF, ceramic capacitor, X7R, 50V C8 22pF, ceramic capacitor, X7R, 50V C9 47nF, ceramic capacitor, X7R, 16V D1 200mA, 30V Schottky diode NXP, ON Semiconductor, Fairchild / BAT54 D2 2A, 60V Schottky diode ON Semiconductor / MBRS260T3 L1 47H, power inductor shielded construction Coilcraft / MSS1048T-473ML M1 N-channel, 30A, 100V MOSFET NXP / PSMN038-100YLX M2 P-channel 15A, 100V MOSFET Infineon / SPD15P10PL G Continued on next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 29 A6271 Automotive High-Current LED Controller Table 5: Application Circuit 4 Bill of Materials (continued) Reference Description R1 2.7k, 1%, 0603 or 0805 R2 20k, 1%, 0603 or 0805 R3 10k, potentiometer R4 100k, 1%, 0603 or 0805 R5 27, 1%, 0603 or 0805 R6 150, 1%, 0603 or 0805 R7,R13 10k, 1%, 0603 or 0805 R8,R9 1, 1%, 0805 or 1206 R10 4.3k, 1%, 0603 or 0805 R11 24, 1%, 0603 or 0805 R12 1k, 1%, 0603 or 0805 R14,R15 180m, 1%, 1206 R16 68k, 1%, 0603 or 0805 Manufacturer/Part Number Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 30 A6271 DRL D1 Park D3 Automotive High-Current LED Controller L1 22 H D2 C1 4.7 F C2 4.7 F C3 100 nF C9 47 nF C4 100 nF R13 470 m R11 150 R14 2.7 R12 C5 1 F R1 2.7 k DR R4 M2 R6 10 k A6271 10 k M4 R5 20 k IREF Analog Control SG C6 22 nF OSC R8 100 k DITH R9 110 k R15 4.3 k OVUV SP R7 D4 5.1 V PWMOUT PWMIN 68 k 10 k LN FAULTn 270 k R3 10 k M1 LP GND R2 R22 10 k VIN VREG COMP R10 39 C7 470 nF R16 0 M3 LED1 C10 C11 C12 4.7 F 4.7 F 4.7 F LED12 R21 220 k R17 1.2 k R18 10 k C8 22 pF R19 68 m R20 68 m GND Figure 12: Boost Driving 12LEDs at 500mA, Switching Frequency 250kHz Park Mode: internal PWM (5%) and analog dimming, no soft start, and frequency dither on. Daylight Running Mode (DRL): 100% LED current and analog dimming, no soft start, and frequency dither on. Battery voltage applied either at daylight running (DRL) terminal or park terminal. Note that Park Mode is dominant. The minimum startup voltage is determined by the voltage on the output (LP) node. 6V is required at this node, so the input voltage, VIN = 6V + Vf (of recirculation diode, D2). If for example, Vf = 0.4V, then VIN = 6.4V. Once operating, the A6271 will function down to an input voltage of 4.5V. The maximum operating voltage on the input is the maximum LED string voltage, plus the diode drop of D2. Table 6: Application Circuit 5 Bill of Materials Reference Description Manufacturer/Part Number C1,C2,C10,C11,C12 4.7F, ceramic capacitor, X7R, 50V TDK, MuRata C3,C4 100nF, ceramic capacitor, X7R, 50V C5 1F, ceramic capacitor, X7R, 16V C6 22nF, ceramic capacitor, X7R, 50V C7 470nF, ceramic capacitor, X7R, 50V C8 22pF, ceramic capacitor, X7R, 50V C9 47nF, ceramic capacitor, X7R, 16V D1, D3 3A, 100V Schottky diode ON Semiconductor / NRVBS3100T3G D2 10A, 60V Schottky diode Vishay / SS10P6 D4 5.1V, zener diode ON Semiconductor / SZBZX84C5V1 L1 22H, high current shielded construction Vishay / IHLP-5050FDER220M-5A M1, M2 N-channel signal MOSFET ON Semiconductor, IR / NTR4003N Continued on next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 31 A6271 Automotive High-Current LED Controller Table 6: Application Circuit 5 Bill of Materials (continued) Reference Description M3 N-channel, 50A, 100V MOSFET Vishay / SQD50N10 M4 P-channel, 15A, 100V MOSFET Infineon / SPD15P10PL G Heatsink for M3 (TO-252) AAVID Thermalloy / 573100D00010G R1 2.7k, 1%, 0603 or 0805 R2 270k, 1%, 0603 or 0805 R3,R6,R12,R18,R22 10k, 1%, 0603 or 0805 R4 68k, 1%, 0603 or 0805 R5 20k, 1%, 0603 or 0805 R7 10k, potentiometer R8 100k, 1%, 0603 or 0805 R9 110k, 1%, 0805 or 1206 R10 39, 1%, 0603 or 0805 R11 150, 1%, 0603 or 0805 R13 470m, 1%, 0805 or 1206 R14 2.7, 1%, 0603 or 0805 R15 4.3k, 1%, 0603 or 0805 R16 0, 0603 or 0805 R17 1.2, 1%, 0603 or 0805 R19, R20 68m, 1%, 2010 R21 220k, 1%, 0603 or 0805 Manufacturer/Part Number Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 32 A6271 Park D1 DRL D3 Automotive High-Current LED Controller L1 D2 C1 4.7 F C2 4.7 F C3 100 nF R22 10 k D5 R13 470 m R11 150 R14 2.7 R12 R5 100 k IREF R7 10 k R3 390 R6 10 k LN PWMOUT 10 k M4 FAULTn DR R4 M2 LP GND R2 RT 100 k VIN VREG C5 1 F R1 2.7 k M5 M1 D4 5.1 V C9 47 nF C4 100 nF 5.1 k R23 100 k 22 H A6271 OVUV SG PWMIN 68 k SP IREF C6 22 nF OSC R8 100 k DITH R9 110 k R15 4.3 k COMP R10 39 C7 470 nF R16 0 LED1 C10 C11 C12 4.7 F 4.7 F 4.7 F M3 LED12 R21 220 k R17 1.2 k R18 10 k C8 22 pF R19 68 m R20 68 m GND Figure 13: Boost Driving 12LEDs at 500mA, Switching Frequency 250kHz Park Mode: internal PWM (10%) and analog dimming at 45% of target level, no soft start, and frequency dither on. Daylight Running Mode (DRL): 100% LED current, no analog dimming, no soft start, and frequency dither on. Battery voltage applied either at daylight running (DRL) terminal or park terminal. Note that DRL Mode is dominant. The minimum startup voltage is determined by the voltage on the output (LP) node. 6V is required at this node, so the input voltage, VIN = 6V + Vf (of recirculation diode, D2). If for example, Vf = 0.4V, then VIN = 6.4V. Once operating, the A6271 will function down to an input voltage of 4.5V. The maximum operating voltage on the input is the maximum LED string voltage, plus the diode drop of D2. Table 7: Application Circuit 6 Bill of Materials Reference Description Manufacturer/Part Number C1,C2,C10,C11,C12 4.7F, ceramic capacitor, X7R, 50V TDK, MuRata C3,C4 100nF, ceramic capacitor, X7R, 50V C5 1F, ceramic capacitor, X7R, 16V C6 22nF, ceramic capacitor, X7R, 50V C7 470nF, ceramic capacitor, X7R, 50V C8 22pF, ceramic capacitor, X7R, 50V C9 47nF, ceramic capacitor, X7R, 16V D1, D3 3A, 100V Schottky diode ON Semiconductor / NRVBS3100T3G D2 10A, 60V Schottky diode Vishay / SS10P6 D4 5.1V, zener diode ON Semiconductor / SZBZX84C5V1 D5 Signal diode 1N4148WS L1 22H, high current shielded construction Vishay / IHLP-5050FDER220M-5A Continued on next page... Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 33 A6271 Automotive High-Current LED Controller Table 7: Application Circuit 6 Bill of Materials (continued) Reference Description Manufacturer/Part Number M1, M2, M5 N-channel signal MOSFET ON Semiconductor, IR / NTR4003N M3 N-channel, 50A, 100V MOSFET Vishay / SQD50N10 M4 P-channel, 15A, 100V MOSFET Infineon / SPD15P10PL G Heatsink for M3 (TO-252) AAVID Thermalloy / 573100D00010G R1 2.7k, 1%, 0603 or 0805 R2 5.1k, 1%, 0603 or 0805 R3 390, 1%, 0603 or 0805 R4 68k, 1%, 0603 or 0805 R5, R8, R23 100k, 1%, 0603 or 0805 R6,R7,R12,R18,R22 10k, 1%, 0603 or 0805 R9 110k, 1%, 0805 or 1206 R10 39, 1%, 0603 or 0805 R11 150, 1%, 0603 or 0805 R13 470m, 1%, 0805 or 1206 R14 2.7, 1%, 0603 or 0805 R15 4.3k, 1%, 0603 or 0805 R16 0, 0603 or 0805 R17 1.2, 1%, 0603 or 0805 R19, R20 68m, 1%, 2010 R21 220k, 1%, 0603 or 0805 RT 100k, NTC, Thermistor Vishay / NTCS0603E3104FXT Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 34 A6271 Automotive High-Current LED Controller PACKAGE OUTLINE DRAWING For Reference Only - Not for Tooling Use (Reference MO-153 ABT) Dimensions in millimeters. NOT TO SCALE Dimensions exclusive of mold flash, gate burrs, and dambar protrusions Exact case and lead configuration at supplier discretion within limits shown 0.65 0.45 8 0 5.00 0.10 16 16 0.20 0.09 1.70 B 3 NOM 4.40 0.10 6.40 0.20 A 1 3.00 0.60 0.15 1.00 REF 2 3 NOM 1 2 0.25 BSC Branded Face C 16X 0.10 C 0.30 0.19 6.10 3.00 SEATING PLANE GAUGE PLANE C SEATING PLANE PCB Layout Reference View 1.20 MAX 0.65 BSC NNNNNNN 0.15 0.00 YYWW LLLL A Terminal #1 mark area B Exposed thermal pad (bottom surface); dimensions may vary with device C Reference land pattern layout (reference IPC7351 SOP65P640X110-17M); All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary to meet application process requirements and PCB layout tolerances; when mounting on a multilayer PCB, thermal vias at the exposed thermal pad land can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5) D Branding scale and appearance at supplier discretion 1 D Standard Branding Reference View N = Device part number = Supplier emblem Y = Last two digits of year of manufacture W= Week of manufacture L = Characters 5-8 of lot number Figure 14: Package LP, 16-Pin eTSSOP with Exposed Thermal Pad Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 35 A6271 Automotive High-Current LED Controller Revision History Number Date - February 3, 2015 Initial Release 1 August 28, 2015 Corrected formula on top of page 15. Updated electrical characteristics (VREG Output Voltage, LED Current Sense and Analog Dimming Differential Sense Voltage). Updated internal PWM and analog dimming (page 9), LED current-sense resistor (page 10), output overvoltage protection (page 12). Application circuits 5 and 6 added (pages 31-34). Figures renumbered. 2 December 15, 2015 3 January 19, 2016 4 April 25, 2016 5 December 2, 2016 Description Corrected Functional Block Diagram (page 4); updated electrical characteristics (disable times, voltage gain, dither modulation frequency, programmable output overvoltage threshold, hiccup shutdown period, and output overvoltage threshold test condition); added PWM output undervoltage section (page 13); updated formula to calculate potential divider for programmable overvoltage protection (page 12 and 13); added comments to application schematics (page 31 and 33). Corrected electrical characteristics (PWMOUT Low Voltage, Peak Pull-Up Current, Peak Pull-Down Current) test conditions. Updated Features and Benefits (page 1), VIN Functional Operating Range test condition (page 5), PWM Dimming: External Disable Time limits (page 6), PWM Dimming Internal Disable Time limits and Duty Cycle (page6), Dither Modulation Frequency limits (page 7), Electrical Characteristics footnote 2 (page 7), Input Undervoltage or VREG Undervoltage Protection section (page 13), VREG Undervoltage action statement (page 14). Removed extraneous NC row from Terminal List table (page 3). 6 January 4, 2018 Updated product status 7 January 15, 2019 Minor editorial updates 8 January 27, 2020 Minor editorial updates Copyright 2020, Allegro MicroSystems. Allegro MicroSystems reserves the right to make, from time to time, such departures from the detail specifications as may be required to permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that the information being relied upon is current. Allegro's products are not to be used in any devices or systems, including but not limited to life support devices or systems, in which a failure of Allegro's product can reasonably be expected to cause bodily harm. The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems assumes no responsibility for its use; nor for any infringement of patents or other rights of third parties which may result from its use. Copies of this document are considered uncontrolled documents. For the latest version of this document, visit our website: www.allegromicro.com Allegro MicroSystems 955 Perimeter Road Manchester, NH 03103-3353 U.S.A. www.allegromicro.com 36