a
AN-382
APPLICATION NOTE
ONE TECHNOLOGY WAY
•
P.O. BOX 9106
•
NORWOOD, MASSACHUSETTS 02062-9106
•
617/329-4700
NOTE
1
The blank evaluation PCB is available to qualified OEMs at no charge.
It was designed to accompany this application note; however, it is not
necessary since this was written as a general tutorial on the SSM2120.
The SSM2120 is a versatile integrated circuit that can be
used for a variety of audio dynamic range processing
functions. It integrates two voltage controlled amplifiers
(VCAs) and two level detector side chains in a single
22- pin package. With this combination, the SSM2120 is
easily configurable as a stereo compressor/limiter, an
automatic gain control amplifier, an expander, a noise
gate, or simply as a dual VCA and dual level detector.
An evaluation board 1 was developed that employs suffi-
cient flexibility to configure the SSM2120 in all the
above applications, providing a demonstration of the
full capabilities of the part. This application note refer-
ences the evaluation board and should be read in con-
junction with the data sheet to develop a full
understanding of the SSM2120.
The functional circuit in Figure 1 shows the basic con-
nections for the VCA and level detector sections of the
SSM2120. The circuitry within the dotted boxes is in-
cluded in the SSM2120, and all the other components
are external. This circuit represents only half of the
SSM2120. The additional VCA and level detector are
functionally identical and differ only in the pin numbers.
A companion product to the SSM2120 is the SSM2122,
which integrates the two VCAs without the level detector
side chains for applications where only the VCAs are
needed. Any of the following discussion regarding the
VCA section of the SSM2120 equally applies to the
SSM2122.
Applications of the SSM2120 Dynamic Range Processor
by Joe Buxton
FULL
WAVE
RECTIFIER
1µF REC
IN
2V
S
IGNAL
INPUT
I
IN
V+
I
REF
Q2
Q1
LOG AV V–
CON
OUT
THRESH
1kΩ39kΩ
10µF 1.5MΩ
V–
LEVEL
DETECTOR
9
1
2
3CONTROL
VOLTAGE
OUTPUT
10kΩ
SSM2120
SIGNAL
OUTPUT
36kΩ
10pF
SIG
OUT
–V
C
+V
C
200Ω
200Ω
SIG
IN
36kΩ
2200pF
47Ω
SIGNAL
INPUT
VCA
7
84
5
SSM2120
Figure 1. SSM2120 Basic Circuitry
–2–
The VCAs are current-in, current-out devices requiring
an external amplifier on the output to convert the cur-
rent back into a voltage. Normally, 36 k Ω resistors are
used on the input and in the feedback of the output am-
plifier, resulting in unity gain with a 0.0 V control signal
applied. In all the application circuits, a series combina-
tion of a 2200 pF capacitor and a 47 Ω resistor at the sig-
nal input is required for stable operation. The SSM2120
has complementary control ports, which follow a 6 mV/
dB gain law. The minus control port (–V C) produces at-
tenuation for positive dc inputs, and the positive control
port (+VC) produces gain.
The level detectors include a full wave rectifier, a log-
ging circuit, and a unipolar drive amplifier, allowing de-
tection of signals over a 100 dB dynamic range. In
normal operation with ±15 V supplies, a 1.5 M Ω resistor
is connected between the LOGAV pin and the negative
supply. Doing so sets up a 10 µA reference current in the
transistor Q2. Meanwhile, a matched logging transistor
Q1, which is diode connected, develops a forward drop
based on the input current. The higher the input current,
the larger the diode drop will be. The voltage at the
LOGAV pin is then the difference in the forward junction
drops of these two matched transistors, which is propor-
tional to the input current. Deriving the formula for the
voltage at LOGAV results in:
VLOGAV =kT
qln |IIN|
IREF
If the input current matches the reference current of
10 µA, then the forward drop across Q1 equals the for-
ward drop across Q2, and the voltage at LOGAV will be
zero. For a typical input resistor of 10 k Ω, 10 µA rms of
current corresponds to a –20 dBV input (0 dBV = 1 V
rms), which gives a control voltage of 0.0 V. To either
side of zero, the V LOGAV follows a 3 mV/dB control law.
The LOGAV pin is buffered by an amplifier to give it a
low impedance output capable of driving a load. This
amplifier is normally configured with a noninverting
10k 10M1M100k1k FREQUENCY – Hz
REF LEVEL
0.000dB
0.0deg
/DIV
20.000dB
45.000deg
MARKER 917
MAG (UDF)
MARKER 917
PHASE (A/R)
731.294Hz
32.025dB
731.294Hz
23.893deg
80
60
40
20
0
–20
180
135
90
45
0
–45
GAIN – dB
PHASE – Degress
PHASE
GAIN
Figure 2 Open-Loop Gain and Phase of Control
Amplifier
gain of 40. As a result, the voltage at CONOUT follows a
ratio of 120 mV/dB. This amplifier was not designed to
be unity gain stable as the open-loop gain and phase
curves in Figure 2 show. In fact, the closed-loop gain
should never be less than 40. To this end, a capacitor
should not be added in parallel with the amplifier’s feed-
back resistor. The reason for this is simple: At high fre-
quencies, the capacitor’s impedance reduces the closed
loop gain below 40. Depending on the size of the ca-
pacitor, this could occur within the frequency range of
the amplifier, resulting in oscillations.
The basic schematic for the evaluation board is shown
in Figure 3. The silkscreen and layout for both sides are
shown in Figures 4 through 6. By combining the VCA
and level detector functional blocks in different man-
ners, all of the following circuits can be designed. The
demo board uses the OP275, a high performance dual
audio amplifier, as current-to-voltage converters at the
output of the SSM2120 VCAs. Several jumpers are used
on the board to provide flexibility for different configu-
rations. These jumper positions can be hard wired for
one type of circuit, or use pin sockets and wire jumpers
for flexibility. In each of the following applications, the
jumper positions are called out on the circuit diagram.
Figure 3. Evaluation Board Schematic
V+
OUT
–IN
+IN
OP275
OUT
–IN
+IN
V–
1
2
3
45
6
7
8
J1
V+
J4
J2
J3
R5
R16
V–
SIG 1
OUT C4
R6
C9
0.1µF
SIG 2
OUT
C10
R21
V–
C6
0.1µF
R7
R17
R8
C5
D1
R2
D2
R1
R3
J5
J6
V+
V–
SIG 1
IN
R9
R10
R11
R12
C3
C2
R14
R15
V+
J7
V– C8
0.1µF
C7 R19
J8
J9
J10J11
V– R28
R20
R27
R18
R30
V+
D3 D4
R32 C14
R31
C13
R26
R25
R24
C12
R29
R22
V+
C11
0.1µF
J13 J14
SIG 2
IN
J12
V+
V+
V–
C15
10µF
C16
10µF
V+
GND
V–
POWER SUPPLIES
R23
V–
C1
R4
R13
V+
V–
SSM2120
12
13
14
15
16
17
18
19
20
21
22
GND
V+
SIGOUT 2
+VC2
CFT 2
–VC2
SIGIN 2
RECIN 2
CONOUT 2
LOG AV 2
THRESH 2
THRESH 1
LOG AV 1
CONOUT 1
SIGOUT 1
+VC1
CFT 1
–VC1
SIGIN 1
RECIN 1
IREF
V–
1
2
3
4
5
6
7
8
9
10
11
V+
V–
R33
V+
–3–
Figure 4. Silkscreen Figure 6. Bottom Side Layout
Figure 5. Topside Layout
Linear Compressor
A compressor is a common audio function to reduce
wide dynamic range signals to a narrower signal range
as shown in Figure 7. Compressing a signal is helpful,
for instance, to prevent low level signals from being
masked by the system noise, such as storing audio on
an analog tape. A linear compressor “rotates” the trans-
fer function around the unity gain point, also referred to
as the threshold. Signals below the threshold are in-
creased, while those above are decreased. This type of
compressor is distinct from other types (such as the lim-
iter discussed below) that only function above or below
the threshold. Linear compressors are typically used in
encode/decode systems such as the companding noise
reduction system described below.
20
0
–20
–40
OUTPUT – dB
–60
–80
–100
–100 –80 –60 –40 –20 0 20
INPUT – dB
2:1
4:1
10:1
NO
COMPRESSION
AGC
20dBV
–80dBV
0dBV
–20dBV
–50dBV
INPUT
COMPRESSED
OUTPUT
2:1 COMPRESSION
100dB 50dB
–20dBV
a. b.
Figure 7. Linear Compressor Functionality
Figure 7a shows an example of a 2:1 compression ratio.
The compression ratio is defined as the ratio of change
in input level (in dB) to the change in output level. Thus,
for the 2:1 ratio shown, an input signal with 100 dB of
dynamic range is reduced to 50 dB. Figure 7b shows the
transfer function for different compression ratios. As
the ratio increases, the dynamic range of the output de-
creases. A linear compressor with a high ratio is gener-
ally referred to as an AGC circuit (Automatic Gain
Control) where the output level is nearly constant re-
gardless of the input. Notice that all the curves in Figure
7b pass through –20 dBV, which is unity gain. This is
not an arbitrary choice. As explained in the preceding
–4–
section, a –20 dBV input to the level detector corre-
sponds to CONOUT = 0 V. With a control voltage of zero
volts, the VCA will have a gain of unity. This unity gain
point can easily be adjusted to any input level as de-
scribed in the following section.
The SSM2120 is programmable for different compres-
sion ratios by simply adjusting one resistor. As ex-
plained above, the level detector has a sensitivity of
3 mV/dB, which is amplified by the control op amp (set
to a gain of 40) to 120 mV/dB. Thus, for every 10 dB rise
in input rms level, the dc voltage at the CONOUT pin
rises by 1.20 V. This voltage can then be scaled and fed
into the VCA’s control port to provide gain or attenua-
tion as shown in Figure 8. The series resistance of R1 +
R3 determines the compression ratio according to the
following formula:
R1+R3=R11 20 CR
CR –1
–1
where CR is the compression ratio.
1/2
OP275
C3
2200pF
R11
200Ω+15V J1 J4
–15V
R16
39kΩ
OPTIONAL
SEE TEXT
R5
100kΩ
C2
10µF
R14
10kΩ
J7
FULL
WAVE
RECTIFIER
RECIN
2V
SIGNAL
INPUT
IIN
V+
IREF
Q2
Q1 LOG AV V–
CONOUT
THRESH
R17
1kΩR7
39kΩ
C5
10µF R8
1.5MΩ
V–
9
1
2
3
SIGNAL
OUTPUT
SSM2120
SIG
OUT
–VC
+VC
SIG
IN
R12
36kΩ
R13
47Ω
5
SSM2120
D1 D2
R1
J6
R3
C1
84
7
R6
36kΩ
C4
10pF
R9
200Ω
SHORT OPEN
J6 J2
J7 J3
D1 J5
D2
J1
J4 OPTIONAL
R2
10kΩ
+15V
Figure 8. Linear Compressor Circuitry
An example of setting this compression ratio best illus-
trates the logic behind this formula. Let’s choose a com-
mon compression ratio of 2:1 as an example. In this
case when the input increases by 20 dB, the output
should only increase by 10 dB. To accomplish this, the
control voltage must be sufficient to produce
10 dB of attenuation in the VCA. Because of the
120 mV/dB relationship, a 20 dB increase in the input
results in a 2.4 V increase in the voltage at CONOUT.
Since the control input (–V C) has a 6 mV/dB control law,
60 mV should be applied to the –V C pin to achieve 10 dB
of attenuation. Now all that is left to do is choose R1 +
R3 such that they form a resistor divider with R11 that
results in a 60 mV/2.4 V attenuation ratio. Since R11 is
normally 200 Ω, R1 + R3 should be 7.8 k Ω, which is ex-
actly the result that the above formula gives.
Different compression ratios can be obtained by just
changing the value of R1 + R3. R1 and R3 could actually
be just one resistor; however, they are split in half in or-
der to insert a capacitor for filtering. Thus, R1 and R3
should each be equal to half the sum of the two. Like all
VCAs, the SSM2120 is sensitive to noise on the control
ports, which feeds through, causing excessive noise and
distortion in the audio signal. The capacitor reduces the
noise significantly, preserving the performance of the
SSM2120. The actual value of the capacitor should be as
large as possible without affecting the attack time of the
control signal (i.e., the time constant at 1/(R1 iR3 × C1)
should be less than the attack time).
For applications that require an adjustable compression
ratio, a potentiometer should be inserted in place of the
jumper labeled J6, which is in series with R1 and R3. To
determine the value of the potentiometer and of R1 + R3,
first determine the minimum and maximum compres-
sion ratios desired. Once these compression ratios are
determined, calculate the corresponding resistances ac-
cording to the above formula. R1 + R3 should be set to
the smaller of the two resistance values, which occurs at
the highest compression ratio desired. (For an AGC cir-
cuit with an infinite compression ratio, R1 + R3 = 3.8 k Ω.)
The potentiometer should then be set to the difference
in the two calculated resistances. For example, a 2:1
compression ratio requires a 7.8 k Ω resistor. Thus, the
pot should be at least 7.8 k Ω–3.8 kΩ = 4.0 k Ω. A standard
5 kΩ would easily do the job. The reason for not simply
replacing R1 and R3 with a potentiometer is the filtering.
A capacitor should be placed between the two 1.9 k Ω
resistors to reduce noise on the VCA’s control port.
Figure 9 shows the actual performance of a compressor
on the evaluation board. The first graph shows the
transfer function for different compression ratios. A few
Figure 9a. Compressor Transfer Function
(V
SY
=
±
15 V, f
IN
= 1 kHz)
–5–
Figure 9b. Compressor THD + N vs. Frequency (V
SY
=
±
15 V, Compression Ration = 2:1, V
IN
= 0 dBV, with
80 kHz Low-Pass Filter)
characteristics are worth pointing out. The bowing of
the 2:1 and 4:1 curves below –70 dBV is due to
nonlinearities in the VCA control port when trying to re-
alize significant amounts of gain above 20 dB. For the
4:1 compression curve, the transfer function shows
some flattening above +10 dBV, which is due to
nonlinearities for large amounts of attenuation.
The second graph shows the distortion performance of
the VCA with 2:1 compression. In doing this sweep,
ample time must be allowed for proper settling of the
level detector and VCA before the measurement is
made. The distinct rise in distortion below 100 Hz is due
to control feedthrough. Above this frequency, the aver-
aging capacitor (C5) filters the LOGAV voltage resulting
in a dc control signal. However, as the frequency drops
below 100 Hz the capacitor can no longer entirely filter
the signal, resulting in a low frequency, low amplitude
sine wave applied to the control port. Thus, the distor-
tion increases. This can be improved by increasing the
averaging capacitor or the filtering cap (C1) at the ex-
pense of an increased attack time.
This above section has discussed the most general type
of compressor, which is equal compression over the
entire input dynamic range. In practical applications,
many compressors only start compressing the signal
once the input level passes a certain threshold. These
types of circuits are discussed in the “compressor/
limiter” section below.
Adjusting the Unity Gain Point (Threshold)
Looking at the example in Figure 7, the compressor
curves pass through unity gain at –20 dBV. With the ad-
dition of the potentiometer, R5, shown as optional in
Figure 8, this threshold is easily adjusted. The voltage at
the wiper of the potentiometer has a one-to-one corre-
spondence with the voltage at CONOUT. This voltage is
summed with LOGAV to produce the CONOUT voltage.
To increase the unity gain point by 20 dB to 0.0 dBV, the
potentiometer needs to be adjusted to produce a volt-
age of 2.4 V at the wiper. This voltage produces a corre-
sponding –2.4 V at CONOUT. When the input signal
reaches 0 dBV, the level detector develops a voltage of
+2.4 V at CONOUT. The two voltages sum and cancel
each other out, leaving CONOUT with zero volts and the
VCA at unity gain. Further adjustment of the potentio-
meter can change the threshold across the entire dy-
namic range of the part. This same technique can also
be used in the expander circuit discussed below.
Automatic Gain Control
A small subsection of the general compressor is an AGC
circuit, where the output has constant amplitude regard-
less of the input. The basic compressor circuit shown in
Figure 8 realizes an AGC circuit with only a couple minor
changes resulting from gain limitations in the VCA. The
first change is to set R1 + R3 to 3.8 k Ω according to the
compressor formula. The second change involves prop-
erly setting the threshold control. The maximum usable
gain of the VCA should be limited to 40 dB; however, for
a –80 dBV input and a –20 dBV unity gain point, the re-
quired gain is 60 dB. To only require 40 dB of gain, the
unity gain point should be lowered to –40 dBV by adjust-
ing the threshold control. At the high end, 60 dB of at-
tenuation is required, which is fine since the SSM2120
has a 100 dB attenuation range. The graph in Figure 10
shows flat response of the entire 100 dB input range.
Notice, however, that the output level is –20 dBV and not
the –40 dBV that the threshold was adjusted to. The ex-
tra 20 dB of gain is realized by increasing the VCA’s out-
put resistor (R6) from 36 k Ω to 360 k Ω.
Figure 10. AGC Transfer Function
(V
SY
=
±
15 V, f
IN
= 1 kHz)
Linear Expander
The complement to a compressor is an expander. In-
stead of reducing the dynamic range of an audio signal,
an expander increases the dynamic range, as the name
implies. Figure 11 illustrates this process. As a continu-
ation of the example above, if the audio signal was com-
pressed by 2:1 and stored on an analog tape, expansion
recreates the original signal with its dynamic range of
100 dB. The actual circuit is identical to that for the com-
pressor except that the +V C port is controlled as op-
posed to the –V C port as shown in Figure 12. Deriving
the formula for the expander is very similar to the
thought process for the compressor above. For ex-
ample, a 4:1 expansion ratio implies that a
–6–
5 dB increase in the input signal results in a 20 dB
increase in the output signal, which means that 15 dB of
gain is needed. Going through the derivation results in
the following formula:
R1+R3=R11 20
ER –1 –1
where
ER
is the expansion ratio.
20
0
–20
–40
OUTPUT – dB
–60
–80
–100
–100 –80 –60 –40 –20 0 20
INPUT – dB
NO
EXPANSION
2:1
4:1
–50dBV
0dBV
–20dBV –20dBV
–80dBV
INPUT
EXPANDED
OUTPUT
2:1 EXPANSION
100dB50dB
20dBV
a. b.
Figure 11. Linear Expander Functionality
1/2
OP275
C3
2200pF
R9
200Ω
C2
10µF
R14
10kΩ
J7
FULL
WAVE
RECTIFIER
RECIN
2V
SIGNAL
INPUT
IIN
V+
IREF
Q2
Q1 LOG AV V–
CONOUT
THRESH
R17
1kΩR7
39kΩ
C5
10µF R8
1.5MΩ
V–
9
1
2
3
SIGNAL
OUTPUT
SSM2120
SIG
OUT
–VC
+VC
SIG
IN
R12
36kΩ
R13
47Ω
SSM2120
D1
D2
R1
J5
R3
C1
84
7
R6
36kΩ
C4
10pF
R11
200Ω
SHORT OPEN
J5 J1
J7 J2
D1 J3
D2 J4
J6
+15V
R2
10kΩ
5
Figure 12. Linear Expander Circuitry
Again, R1 and R3 should be the same value and equal to
half the sum of the two (R1 = R3 = 1/2 (R1 + R3)), and R11
should be no larger than 200 Ω. Capacitor C1 should be
chosen to provide the maximum filtering without in-
creasing the attack time. As with the case of a compres-
sor, a potentiometer can replace J5 and be inserted in
series with R1 and R3 if an adjustable expansion ratio is
needed. Picking the value of the potentiometer follows
the same process as described for the compressor.
Figure 13a shows the transfer function for different ex-
pansion ratios revealing linear response over the entire
input range. The distortion performance in Figure 13b is
very similar to the compressors performance shown
above. Again the distortion increases at low frequencies
due to control feedthrough.
Figure 13a. Expander Transfer Function (V
SY
=
±
15 V,
f
IN
= 1 kHz)
Figure 13b. Expander THD + N vs. Frequency (V
SY
=
15 V, Expansion Ration = 2:1, V
IN
= –5 dBV, with
80 kHz Low-Pass Filter)
Companding Noise Reduction System
The above two circuits can be used in conjunction to
form a companding noise reduction system. The block
diagram for such a system is shown in Figure 14, where
the blocks for compressor and expander are exactly the
circuits shown above. The purpose of such a system is
to reduce the effects of a noisy storage medium, such as
tape. As the graph in Figure 14 shows, the input signal
has a wide dynamic range with a low noise floor. The
SSM2120 is used to compress this signal such that its
minimum signal level is well above the noise floor of the
tape. For playback, the signal is passed through the ex-
pander half of the SSM2120, and the original dynamic
range is restored. Notice that the noise floor of the tape
is pushed down to below the minimum signal, greatly
reducing tape hiss. Table I lists some common com-
pression and expansion ratios and the resistor values re-
quired to achieve these. Please note, this table replaces
the one shown in the data sheet.
–7–
Table I.
Gain (dB) Compressor Expander Compression/
Input Signal (Reduction or Output Signal Output Signal Expansion R1 + R3 ΩR1 + R3 Ω
Increase (dB) Increase) Increase (dB) Increase (dB) Ratio Compressor Expander
20 6.67 13.33 20.00 1.5:1 11,800 7,800
20 10.00 10.00 20.00 2:1 7,800 3,800
20 13.33 6.67 20.00 3:1 5,800 1,800
20 15.00 5.00 20.00 4:1 5,133 1,130
20 16.00 4.00 20.00 5:1 4,800 800
20 17.33 2.67 20.00 7.5:1 4,415 415
20 18.00 2.00 20.00 10:1 4,244 244
20 20.00 0.00 N/A AGC 3,800 N/A
AUDIO
IN AUDIO
OUT
TAPE OR
OTHER
STORAGE
MEDIUM
COMPRESSOR EXPANDER
+20dBV
NOISE
–80dBV
100dB
+20dBV 0dBV
–80dBV
50dB 100dB
TAPE NOISE
NOISE
FLOOR
Figure 14. Companding Noise Reduction System
Figure 15a shows the actual performance of the overall
transfer function of this system. As expected, the input
versus output is 1:1 except at the extreme ends where
nonlinearities in the control path limit the system linear-
ity. This graph was generated for a 2:1:2 compression/
expansion ratio. Distortion is graphed versus input level
for a 1 kHz input signal in Figure 15b. For inputs below
–10 dBV, noise dominates the distortion measurement.
The overall performance is very good especially consid-
ering that the signal passes through two dynamic pro-
cessing stages.
Figure 15a. Companding Noise Reduction System
Transfer Function (V
SY
=
±
15 V, f
IN
= 1 kHz)
Figure 15b. Companding Noise Reduction THD + N
vs. Amplitude (V
SY
=
±
15 V, f
IN
= 1 kHz, with 22 kHz
Low-Pass Filter)
Downward Expander/Noise Gate
A downward expander is essentially a modification of
the basic expander circuit discussed above and is used
to reduce the noise during quiet sections of an audio sig-
nal. Instead of expanding the entire dynamic range of
the audio input, a downward expander only affects that
portion of the signal that is below a selected threshold
as shown in Figure 16. As you can see, the transfer func-
tion is unity gain until the signal falls below a certain
threshold, –40 dBV in this example. Below that point, the
signal is expanded downward, pushing the noise floor
down below an audible level. Because of this action, this
circuit is also referred to as a noise gate.
20
0
–20
–40
OUTPUT – dB
–60
–80
–100 –80 –60 –40 –20 0 20
INPUT – dB
THRESHOLD
2:1
–60dBV
–40dBV
NOISE –80dBV
INPUT EXPANDED
OUTPUT
2:1 EXPANSION
–40dBV THRESHOLD
100dB80dB
+20dBV
a. b.
Figure 16. Downward Expander/Noise Gate
–8–
Looking at the circuit in Figure 17, you can see that it is
very similar to the basic expander circuit with two no-
table exceptions. A trimming potentiometer is added to
control the threshold of expansion, and the resistor R2
has been removed. As the schematic shows, the output
CONOUT uses a single PNP transistor to sink current,
and it does not have a complementary NPN transistor
for sourcing current. The reason for this construction is
apparent from the functionality of the circuit. For the
SSM2120 VCA to be unity gain above the threshold, the
control port needs to see zero volts. With R2 in place,
the CONOUT voltage would go positive for large sig-
nals. For example, a 0 dB input produces a 2.4 V output.
However, with R2 removed, CONOUT is not able to pull
high and thus remains at zero. Only until CONOUT pulls
downward, does a negative voltage appear on +V C. The
voltage at CONOUT is graphed as a function of the input
voltage to further illustrate this point.
1/2
OP275
C3
2200pF
R9
200Ω
C2
10µF
R14
10kΩ
J7
FULL
WAVE
RECTIFIER
RECIN
2V
SIGNAL
INPUT
IIN
V+
IREF
Q2
Q1 LOG AV V–
CONOUT
THRESH
R17
1kΩR7
39kΩ
C5
10µF R8
1.5MΩ
V–
9
1
2
3
SIGNAL
OUTPUT
SSM2120
SIG
OUT
–VC
+VC
SIG
IN
R12
36kΩ
R13
47Ω
SSM2120
R1
J5
R3
C1
84
7
R6
36kΩ
C4
10pF
R11
200Ω
SHORT OPEN
J2 J1
J4 J3
J5 J6
J7
D1
D2
R5
100kΩ
5
J2 J4 V–
39kΩ
–40dBV 0dBV
0.0V
INPUT
Figure 17. Downward Expander/Noise Gate
The threshold potentiometer controls the level below
which expansion occurs. Because the trim potentio-
meter is connected between ground and the minus sup-
ply, it is always trying to force the output positive, but
the output is not able to go positive. Instead, through
superposition, the level detector voltage, LOGAV, must
be negative enough to force CONOUT negative. Re-
member that the voltage at LOGAV is multiplied by a
gain of 40 to CONOUT, and the threshold control voltage
is unity gain to the output. By comparing these two volt-
ages, the threshold point can be determined. As an ex-
ample, let’s pick a threshold of –40 dBV for the
downward expansion. An input of –40 dBV corresponds
to a voltage at LOGAV of –60 mV, which is multiplied by
40 to give –2.4 V at CONOUT. Setting the threshold con-
trol to –2.4 V produces a positive voltage at CONOUT of
+2.4 V. These two voltages cancel out, producing a net
voltage of zero. For any signals below –40 dBV, the volt-
age at CONOUT will go negative, and for inputs above
–40 dBV, the voltage at CONOUT wants to go positive.
However, since a pull-up resistor is not in place, the volt-
age remains at zero.
Figure 18a. Downward Expander Transfer Function
(V
SY
=
±
15 V, f
IN
= 1 kHz)
Figure 18b. Downward Expander THD + N vs. Ampli-
tude (V
SY
=
±
15 V, f
IN
= 1 kHz, with 22 kHz Low-Pass
Filter, 2:1 Expansion with –20 dBV Threshold)
Figure 18a shows the actual response of the circuit for
various threshold settings. R1 + R3 is set to 3.8 k Ω for
2:1 expansion below the threshold, and the control pot
is adjusted for three different threshold settings. The
distortion versus amplitude curve in Figure 18b shows
good performance. As before, noise dominates the mea-
surement below –10 dBV.
Compressor/Limiter
A compressor/limiter is similar to a linear compressor
except that it only takes effect after a certain threshold is
reached. In this case, when the threshold is exceeded,
the SSM2120 starts to compress the audio to prevent
clipping and its associated high distortion. A typical
transfer function of this type of circuit is shown in Figure
19 with different amounts of compression. The most ex-
–9–
treme case is hard limiting, which means that once the
threshold is passed, the output level does not increase
at all. The control voltage curve that realizes a limiter is
shown along with the circuit in Figure 20. Notice that
the control voltage is held at ground until the threshold
is reached. Above this point, the positive control volt-
age on the VCA’s –V C input results in attenuation. As
before, by selecting the proper scaling resistor, any
amount of compression can be obtained.
The top curve in Figure 19 shows the case of adding
“make-up gain.” When an audio signal is compressed,
the top end of the dynamic range of the audio system
may not be fully utilized. Make-up gain increases the
overall level to bring up the maximum audio signal to
just below clipping. Take the example of the 2:1 com-
pressor/limiter curve shown in Figure 19. The maximum
20
0
–20
–40
OUTPUT – dB
–60
–80
–80 –60 –40 –20 0 20
INPUT – dB
THRESHOLD
2:1
2:1 COMPRESSION WITH
20dB MAKE UP GAIN
4:1
AGC/LIMIT
–80dBV
–20dBV
–80dBV
INPUT LIMITED
OUTPUT
4:1 COMRESSION
–20dBV THRESHOLD
100dB
+20dBV
–20dBV
–10dBV
a. b.
Figure 19. Compressor/Limiter
1/2
OP275
C3
2200pF
R11
200Ω
C2
10µF
R14
10kΩ
J7
FULL
WAVE
RECTIFIER
RECIN
2V
SIGNAL
INPUT
IIN
V+
IREF
Q2
Q1 LOG AV V–
CONOUT
THRESH
R17
1kΩR7
39kΩ
C5
10µF R8
1.5MΩ
V–
9
1
2
3
SIGNAL
OUTPUT
SSM2120
SIG
OUT
–VC
+VC
SIG
IN
R12
36kΩ
R13
47Ω
SSM2120
R1
J6
R3
C1
84
7
R6
36kΩ
C4
10pF
R9
200Ω
SHORT OPEN
J1 J2
J3 J4
J6 J5
J7
R5
100kΩ
5
V+
J3 J1
R16
39kΩ
–20
0.0V
INPUT – dBV
D1
D2
R2
1kΩ
V1
Figure 20. Compressor/Limiter Circuitry
signal level is 0 dBV. If the audio equipment can handle
over 20 dBV signals, then the top 20 dB is wasted. How-
ever, by adding 20 dB of gain, the entire curve is raised
to take full advantage of the headroom. At the same
time, the low level signals are also amplified, improving
noise immunity. The fixed gain can easily be added by
adjusting either the input (R12) or output (R6) resistors.
For example, lowering R12 to 3.6 k Ω results in +20 dB of
fixed gain. Optimizing this adjustment is discussed in
the section headed “Optimizing VCA Performance.”
Notice in the circuit that two diodes have been included
as well as the pull up resistor on the output. The resistor
enables CONOUT to swing positive, and the diodes pre-
vent the control voltage from going below ground. The
threshold control works much the same as for the down-
ward expander circuit. The main difference is that now
the threshold voltage is between ground and the posi-
tive supply. Thus, this voltage tries to force CONOUT to
a negative voltage. When CONOUT is negative, diode
D1 is on, causing the voltage at V1 to follow CONOUT
plus approximately 0.6 V. Not until the voltage at
CONOUT reaches ground does the voltage at V1 rise
high enough to turn D2 on. Once this occurs, the volt-
age on the cathode of D2 follows CONOUT. The actual
diodes used are not critical and 1N914 types work fine.
Just remember that any mismatches in the diodes result
in errors in the threshold level. For best accuracy,
matched diodes should be used.
Figure 21a shows that the measured transfer function is
very close to the ideal curves of Figure 19. The distor-
tion performance in Figure 21b is very similar to the per-
formance of the expander circuit.
HARD LIMIT
THRESHOLD
2:1
20dB
COMPRESSION
MAKE-UP GAIN
Figure 21a. Compressor/Limiter Transfer Function
(V
SY
=
±
15 V, f
IN
= 1 kHz)
–10–
Figure 21b. Compressor/Limiter THD + N vs. Amplitude
(V
SY
=
±
15 V, f
IN
= 1 kHz, with 22 kHz Low-Pass Filter, 2:1
Compression Above –20 dBV Threshold)
Combining a Limiter with a Downward Expander
The above two circuits can easily be combined to result
in a circuit that utilizes both downward expansion for
low level signals and compression or limiting for high
level signals. Both level detector sections are used with
only one of the VCAs. The second VCA can either be
used for a different function or left unused. One of the
level detectors is configured as a downward expander
and the second as a compressor/limiter. The outputs of
the detectors feed into the respective complementary
control ports of the one VCA. To accomplish this on the
demo board a jumper wire needs to be connected be-
tween the output of the second level detector and the
control port of the opposite VCA.
Limiting the Threshold Adjustment Range
In many applications, the threshold for compression or
expansion may be adjustable by the end user of the au-
dio equipment. In such cases, the threshold adjustment
range can easily be limited by adding resistors in series
with the trimming potentiometer as shown in Figure
22a. The resistors are labeled RJ1, 2 and RJ3, 4 to sig-
nify that they are used in place of the jumpers, J1 or J2
and J3 or J4. As with the jumpers, the connection of the
resistors can be to either supply or ground depending
on whether the application is for a compressor or an
expander.
An example best illustrates how to determine the values
of the two resistors. Take the case of the compressor/
limiter shown in Figure 20 with an adjustable threshold
from –40 dBV to +20 dBV. First, RJ1, 2 should be con-
nected to V+ and RJ3, 4 should be connected to V–. Re-
member, the threshold corresponds to the point where
CONOUT = 0.0 V. A +20 dBV input produces +4.8 V at
CONOUT. To set the threshold at this point, the voltage
at the wiper of the pot needs to be +4.8 V to produce a
net voltage at CONOUT of 0 V. For a –40 dBV threshold,
the voltage at the wiper needs to be –2.4 V. These volt-
ages correspond to the wiper being at extreme ends of
the pot as shown in Figures 22b and 22c.
Now the resistor divider networks need to be solved for
the two resistors’ values. Remember that the THRESH-
OLD input is a virtual ground whose potential is within
±200 mV of ground at all times. For practical purposes,
the point can be assumed to be at ground at all times.
The resulting error from this assumption is typically less
than 1 dB. Solving for the resistor values by using mul-
tiple iterations in SPICE results in RJ1, 2 = 43 k Ω and
RJ3, 4 = 68 k Ω.
Optimizing the Usable Range of the Level Detector
In some situations, it may be desirable to change the us-
able voltage range of the level detector making it more
sensitive at the low end or more linear at the high end.
This is easily done by adjusting the value of the 10 k Ω
input resistor to RECIN. The level detector is actually
detecting the amount of current into its RECIN input, and
the resistor is needed as the voltage-to-current con-
verter. Changing this resistor optimizes the level detec-
tor for different input voltage ranges. The level detector
has a sensitivity that ranges from 10 nA to 1 mA. For a
10 kΩ resistor, this corresponds to a voltage range of
100 µV to 10 V. At the two ends of the range, the linear-
ity does suffer. Thus, if the application requires accu-
racy for small voltage inputs and is not concerned with
the high end of the range (such as for a downward
expander), the resistor should be reduced. For example,
to detect signals down to 10 µV, the resistor should be
changed to 1 k Ω. The opposite is true if accuracy is
needed for only high level signals (such as for a
compressor/limiter). In this case, the resistor should be
increased. In either case, the ratio is still 3 mV/dB at
VLOGAV and 120 mV/dB at V CONOUT.
Changing the value of the resistor will change the dc
output level at CONOUT for a given input voltage. In-
stead of a –20 dBV input corresponding to CONOUT =
0.0 V, a 1 k Ω input resistor results in a –40 dBV input
a. b. c.
Figure 22. Adjusting the Threshold Adjustment Range
R5
100kΩ
R16
39kΩ
RJ1, 2
RJ3, 4
GND OR V–
V+ OR GND
THRESHOLD 100kΩ39kΩ
RJ1, 2
RJ3, 4
V–
V+
THRESHOLD
–2.4V 100kΩ
39kΩ
RJ1, 2
RJ3, 4
V–
V+
THRESHOLD
+4.8V
–11–
giving CONOUT = 0.0 V. Remember, the input current
that corresponds to zero volts out is always 10 µA.
Whatever voltage, resistor combination provides 10 µA
will result in zero volts out.
Optimizing VCA Performance
Another degree of flexibility with the SSM2120 is the ad-
justment of the input and output resistors. In all the
above applications, these resistors default to 36 k Ω.
However, by increasing or decreasing these resistors,
the VCA can be optimized for the audio signal level. For
example, if the audio level has a maximum level of
0.0 dBV, the input and output resistors can be lowered to
3.6 k Ω. The main consideration to keep in mind is that
the maximum current into the VCA is 400 µA. Therefore,
just divide the peak input voltage by 400 µA to arrive at
the input resistor value. Operating the VCA close to its
peak input level ensures the maximum signal to noise
ratio.
Adjusting the resistors is also helpful in producing a
fixed gain or attenuation in the part. The VCA performs
best when its operated with a control input of zero volts,
keeping the VCA in unity gain. However, if the optimal
system condition is to have a nominal gain of 20 dB,
then this fixed gain should be produced by scaling the
input and output resistors. This could occur when the
input level is a maximum of 0 dBV and the desired out-
put level is 20 dBV. In this case, the input resistor should
be decreased to 3.6 k Ω and the output left at 36 k Ω.
One comment should be made about the two 200 Ω re-
sistors from the VCA’s control ports to ground. These
resistors are needed to divide down the control voltage
at CONOUT and scale it for the VCA. A user may be
tempted to increase these resistors depending on the
resistor divider requirements. However, these resistors
should never be increased above 200 Ω. Doing so in-
creases the distortion and noise of the VCA.
–12–
PRINTED IN U.S.A. E2060–9–9/95
