VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C1
C2
R1
R2
D1
D2
ON
OFF
LMR12007
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SNVS982 –SEPTEMBER 2013
LMR12007 Thin SOT23 750mA Load Step-Down DC-DC Regulator
Check for Samples: LMR12007
1FEATURES DESCRIPTION
The LMR12007 regulator is a monolithic, high
23• Thin SOT-6 Package frequency, PWM step-down DC/DC converter in a 6-
• 3.0V to 18V Input Voltage Range pin Thin SOT package. It provides all the active
• 1.25V to 16V Output Voltage Range functions to provide local DC/DC conversion with fast
transient response and accurate regulation in the
• 750mA Output Current smallest possible PCB area.
• 550kHz (LMR12007Y) and 1.6MHz (LMR12007X) With a minimum of external components and online
Switching Frequencies design support through WEBENCH®, the LMR12007
• 350mΩNMOS Switch is easy to use. The ability to drive 750mA loads with
• 30nA Shutdown Current an internal 350mΩNMOS switch using state-of-the-
• 1.25V, 2% Internal Voltage Reference art 0.5µm BiCMOS technology results in the best
power density available. The world class control
• Internal Soft-Start circuitry allows for on-times as low as 13ns, thus
• Current-Mode, PWM Operation supporting exceptionally high frequency conversion
• WEBENCH®Online Design Tool over the entire 3V to 18V input operating range down
to the minimum output voltage of 1.25V. Switching
• Thermal Shutdown frequency is internally set to 550kHz (LMR12007Y) or
1.6MHz (LMR12007X), allowing the use of extremely
APPLICATIONS small surface mount inductors and chip capacitors.
• Local Point of Load Regulation Even though the operating frequencies are very high,
efficiencies up to 90% are easy to achieve. External
• Core Power in HDDs shutdown is included, featuring an ultra-low stand-by
• Set-Top Boxes current of 30nA. The LMR12007 utilizes current-mode
• Battery Powered Devices control and internal compensation to provide high-
performance regulation over a wide range of
• USB Powered Devices operating conditions. Additional features include
• DSL Modems internal soft-start circuitry to reduce inrush current,
• Notebook Computers pulse-by-pulse current limit, thermal shutdown, and
output over-voltage protection.
Typical Application Circuit Efficiency vs Load Current "X"
VIN = 5V, VOUT = 3.3V
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2WEBENCH is a registered trademark of Texas Instruments.
3All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
1
2
3
6
5
4
BOOST
GND
FB
SW
VIN
EN
1
2
3
6
5
4
LMR12007
SNVS982 –SEPTEMBER 2013
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Connection Diagram
Figure 1. 6-Lead SOT Figure 2. Pin 1 Indentification
See Package Number DDC (R-PDSO-G6)
PIN DESCRIPTIONS
Pin Name Function
1 BOOST Boost voltage that drives the internal NMOS control switch. A bootstrap capacitor is connected between the
BOOST and SW pins.
2 GND Signal and Power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin
for accurate regulation.
3 FB Feedback pin. Connect FB to the external resistor divider to set output voltage.
4 EN Enable control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V.
5 VIN Input supply voltage. Connect a bypass capacitor to this pin.
6 SW Output switch. Connects to the inductor, catch diode, and bootstrap capacitor.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)
VIN -0.5V to 22V
SW Voltage -0.5V to 22V
Boost Voltage -0.5V to 28V
Boost to SW Voltage -0.5V to 6.0V
FB Voltage -0.5V to 3.0V
EN Voltage -0.5V to (VIN + 0.3V)
Junction Temperature 150°C
ESD Susceptibility(2) 2kV
Storage Temp. Range -65°C to 150°C
Infrared/Convection Reflow (15sec) 220°C
Soldering Information Wave Soldering Lead Temp. (10sec) 260°C
(1) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(2) Human body model, 1.5kΩin series with 100pF.
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Operating Ratings(1)
VIN 3V to 18V
SW Voltage -0.5V to 18V
Boost Voltage -0.5V to 23V
Boost to SW Voltage 1.6V to 5.5V
Junction Temperature Range −40°C to +125°C
Thermal Resistance θJA(2) 118°C/W
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For specific specifications and the test conditions,
see Electrical Characteristics.
(2) Thermal shutdown will occur if the junction temperature exceeds 165°C. The maximum power dissipation is a function of TJ(MAX) ,θJA
and TA. The maximum allowable power dissipation at any ambient temperature is PD= (TJ(MAX) – TA)/θJA . All numbers apply for
packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still
air, θJA = 204°C/W.
Electrical Characteristics
Specifications with standard typeface are for TJ= 25°C, and those in boldface type apply over the full Operating
Temperature Range (TJ= -40°C to 125°C). VIN = 5V, VBOOST - VSW = 5V unless otherwise specified. Datasheet min/max
specification limits are ensured by design, test, or statistical analysis.
Symbol Parameter Conditions Min(1) Typ(2) Max(1) Units
VFB Feedback Voltage 1.225 1.250 1.275 V
ΔVFB/ΔVIN Feedback Voltage Line Regulation VIN = 3V to 18V 0.01 % / V
IFB Feedback Input Bias Current Sink/Source 10 250 nA
Undervoltage Lockout VIN Rising 2.74 2.90
UVLO Undervoltage Lockout VIN Falling 2.0 2.3 V
UVLO Hysteresis 0.30 0.44 0.62
LMR12007X 1.2 1.6 1.9
FSW Switching Frequency MHz
LMR12007Y 0.40 0.55 0.66
LMR12007X 85 92
DMAX Maximum Duty Cycle %
LMR12007Y 90 96
LMR12007X 2
DMIN Minimum Duty Cycle %
LMR12007Y 1
RDS(ON) Switch ON Resistance VBOOST - VSW = 3V 350 650 mΩ
ICL Switch Current Limit VBOOST - VSW = 3V 1.0 1.5 2.3 A
IQQuiescent Current Switching 1.5 2.5 mA
Quiescent Current (shutdown) VEN = 0V 30 nA
LMR12007X (50% Duty Cycle) 2.2 3.3
IBOOST Boost Pin Current mA
LMR12007Y (50% Duty Cycle) 0.9 1.6
Shutdown Threshold Voltage VEN Falling 0.4
VEN_TH V
Enable Threshold Voltage VEN Rising 1.8
IEN Enable Pin Current Sink/Source 10 nA
ISW Switch Leakage 40 nA
(1) Specified to Texas Instruments' Average Outgoing Quality Level (AOQL).
(2) Typicals represent the most likely parametric norm.
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Typical Performance Characteristics
All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 4.7 µH ("X"), L1 = 10 µH ("Y"), and TA= 25°C, unless specified
otherwise.
Efficiency vs Load Current - "X" VOUT = 5V Efficiency vs Load Current - "Y" VOUT = 5V
Figure 3. Figure 4.
Efficiency vs Load Current - "X" VOUT = 3.3V Efficiency vs Load Current - "Y" VOUT = 3.3V
Figure 5. Figure 6.
Efficiency vs Load Current - "X" VOUT = 1.5V Efficiency vs Load Current - "Y" VOUT = 1.5V
Figure 7. Figure 8.
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Typical Performance Characteristics (continued)
All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 4.7 µH ("X"), L1 = 10 µH ("Y"), and TA= 25°C, unless specified
otherwise. Oscillator Frequency vs Temperature - "X" Oscillator Frequency vs Temperature - "Y"
Figure 9. Figure 10.
Current Limit vs Temperature VIN = 18V, VIN = 5V VFB vs Temperature
Figure 11. Figure 12.
RDSON vs Temperature IQSwitching vs Temperature
Figure 13. Figure 14.
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Typical Performance Characteristics (continued)
All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 4.7 µH ("X"), L1 = 10 µH ("Y"), and TA= 25°C, unless specified
otherwise. Line Regulation - "X" VOUT = 1.5V, IOUT = 500mA Line Regulation - "Y" VOUT = 1.5V, IOUT = 500mA
Figure 15. Figure 16.
Line Regulation - "X" VOUT = 3.3V, IOUT = 500mA Line Regulation - "Y" VOUT = 3.3V, IOUT = 500mA
Figure 17. Figure 18.
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Product Folder Links: LMR12007
L
R1
R2
D
1
D2
BOOST
Output
Control
Logic
Current
Limit
Thermal
Shutdown
Under
Voltage
Lockout
Corrective Ramp
Reset
Pulse
PWM
Comparator
Current-Sense Amplifier RSENSE
+
+
Internal
Regulator
and
Enable
Circuit
Oscillator
Driver 0.3:
Switch
Internal
Compensation
SW
EN
FB
GND
Error Amplifier -
+VREF
1.25V
COUT
ON
OFF
VBOOST
VSW
+
-
CBOOST
VOUT
CIN
VIN
VIN
ISENSE
+
-
+
-
+
-1.375V
OVP
Comparator
Error
Signal
-
+
IL
LMR12007
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SNVS982 –SEPTEMBER 2013
Block Diagram
Figure 19.
APPLICATION INFORMATION
THEORY OF OPERATION
The LMR12007 is a constant frequency PWM buck regulator IC that delivers a 750mA load current. The
regulator has a preset switching frequency of either 550kHz (LMR12007Y) or 1.6MHz (LMR12007X). These high
frequencies allow the LMR12007 to operate with small surface mount capacitors and inductors, resulting in
DC/DC converters that require a minimum amount of board space. The LMR12007 is internally compensated, so
it is simple to use, and requires few external components. The LMR12007 uses current-mode control to regulate
the output voltage.
The following operating description of the LMR12007 will refer to the Simplified Block Diagram (Figure 19) and to
the waveforms in Figure 20. The LMR12007 supplies a regulated output voltage by switching the internal NMOS
control switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the
reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the
internal NMOS control switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and
the inductor current (IL) increases with a linear slope. ILis measured by the current-sense amplifier, which
generates an output proportional to the switch current. The sense signal is summed with the regulator’s
corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the
feedback voltage and VREF. When the PWM comparator output goes high, the output switch turns off until the
next switching cycle begins. During the switch off-time, inductor current discharges through Schottky diode D1,
which forces the SW pin to swing below ground by the forward voltage (VD) of the catch diode. The regulator
loop adjusts the duty cycle (D) to maintain a constant output voltage.
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BOOST
SW
GND
L
D1
D2
COUT
CBOOST
VOUT
CIN
VIN
VIN
VBOOST
0
0
VIN
VD
TON
t
t
Inductor
Current
D = TON/TSW
VSW
TOFF
TSW
IL
IPK
SW
Voltage
LMR12007
SNVS982 –SEPTEMBER 2013
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Figure 20. LMR12007 Waveforms of SW Pin Voltage and Inductor Current
BOOST FUNCTION
Capacitor CBOOST and diode D2 in Figure 21 are used to generate a voltage VBOOST. VBOOST - VSW is the gate
drive voltage to the internal NMOS control switch. To properly drive the internal NMOS switch during its on-time,
VBOOST needs to be at least 1.6V greater than VSW. Although the LMR12007 will operate with this minimum
voltage, it may not have sufficient gate drive to supply large values of output current. Therefore, it is
recommended that VBOOST be greater than 2.5V above VSW for best efficiency. VBOOST – VSW should not exceed
the maximum operating limit of 5.5V.
5.5V > VBOOST – VSW > 2.5V for best performance.
Figure 21. VOUT Charges CBOOST
When the LMR12007 starts up, internal circuitry from the BOOST pin supplies a maximum of 20mA to CBOOST.
This current charges CBOOST to a voltage sufficient to turn the switch on. The BOOST pin will continue to source
current to CBOOST until the voltage at the feedback pin is greater than 1.18V.
There are various methods to derive VBOOST:
1. From the input voltage (VIN)
2. From the output voltage (VOUT)
3. From an external distributed voltage rail (VEXT)
4. From a shunt or series zener diode
In the Simplifed Block Diagram of Figure 19, capacitor CBOOST and diode D2 supply the gate-drive current for the
NMOS switch. Capacitor CBOOST is charged via diode D2 by VIN. During a normal switching cycle, when the
internal NMOS control switch is off (TOFF) (refer to Figure 20), VBOOST equals VIN minus the forward voltage of D2
(VFD2), during which the current in the inductor (L) forward biases the Schottky diode D1 (VFD1). Therefore the
voltage stored across CBOOST is
VBOOST - VSW = VIN - VFD2 + VFD1 (1)
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VIN BOOST
SW
GND
CBOOST
L
D1
D2
D3
VBOOST
VIN
CIN
COUT
VOUT
LMR12007
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SNVS982 –SEPTEMBER 2013
When the NMOS switch turns on (TON), the switch pin rises to
VSW = VIN – (RDSON x IL), (2)
forcing VBOOST to rise thus reverse biasing D2. The voltage at VBOOST is then
VBOOST = 2VIN – (RDSON x IL) – VFD2 + VFD1 (3)
which is approximately
2VIN - 0.4V (4)
for many applications. Thus the gate-drive voltage of the NMOS switch is approximately
VIN - 0.2V (5)
An alternate method for charging CBOOST is to connect D2 to the output as shown in Figure 21. The output
voltage should be between 2.5V and 5.5V, so that proper gate voltage will be applied to the internal switch. In
this circuit, CBOOST provides a gate drive voltage that is slightly less than VOUT.
In applications where both VIN and VOUT are greater than 5.5V, or less than 3V, CBOOST cannot be charged
directly from these voltages. If VIN and VOUT are greater than 5.5V, CBOOST can be charged from VIN or VOUT
minus a zener voltage by placing a zener diode D3 in series with D2, as shown in Figure 22. When using a
series zener diode from the input, ensure that the regulation of the input supply doesn’t create a voltage that falls
outside the recommended VBOOST voltage.
(VINMAX – VD3) < 5.5V
(VINMIN – VD3) > 1.6V
Figure 22. Zener Reduces Boost Voltage from VIN
An alternative method is to place the zener diode D3 in a shunt configuration as shown in Figure 23. A small
350mW to 500mW 5.1V zener in a SOT or SOD package can be used for this purpose. A small ceramic
capacitor such as a 6.3V, 0.1µF capacitor (C4) should be placed in parallel with the zener diode. When the
internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The
0.1 µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time.
Resistor R3 should be chosen to provide enough RMS current to the zener diode (D3) and to the BOOST pin. A
recommended choice for the zener current (IZENER) is 1 mA. The current IBOOST into the BOOST pin supplies the
gate current of the NMOS control switch and varies typically according to the following formula for the X -
version:
IBOOST = 0.49 x (D + 0.54) x (VZENER – VD2) mA (6)
IBOOST can be calculated for the Y version using the following:
IBOOST = 0.20 x (D + 0.54) x (VZENER - VD2) µA (7)
where D is the duty cycle, VZENER and VD2 are in volts, and IBOOST is in milliamps. VZENER is the voltage applied to
the anode of the boost diode (D2), and VD2 is the average forward voltage across D2. Note that this formula for
IBOOST gives typical current. For the worst case IBOOST, increase the current by 40%. In that case, the worst case
boost current will be
IBOOST-MAX = 1.4 x IBOOST (8)
R3 will then be given by
R3 = (VIN - VZENER) / (1.4 x IBOOST + IZENER) (9)
For example, using the X-version let VIN = 10V, VZENER = 5V, VD2 = 0.7V, IZENER = 1mA, and duty cycle D = 50%.
Then
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VIN BOOST
SW
GND
L
D1
D2
D3
R3
C4
VBOOST
CBOOST
VZ
VIN
CIN
COUT
VOUT
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
IBOOST = 0.49 x (0.5 + 0.54) x (5 - 0.7) mA = 2.19mA (10)
R3 = (10V - 5V) / (1.4 x 2.19mA + 1mA) = 1.23kΩ(11)
Figure 23. Boost Voltage Supplied from the Shunt Zener on VIN
ENABLE PIN / SHUTDOWN MODE
The LMR12007 has a shutdown mode that is controlled by the enable pin (EN). When a logic low voltage is
applied to EN, the part is in shutdown mode and its quiescent current drops to typically 30nA. Switch leakage
adds another 40nA from the input supply. The voltage at this pin should never exceed VIN + 0.3V.
SOFT-START
This function forces VOUT to increase at a controlled rate during start up. During soft-start, the error amplifier’s
reference voltage ramps from 0V to its nominal value of 1.25V in approximately 200µs. This forces the regulator
output to ramp up in a more linear and controlled fashion, which helps reduce inrush current.
OUTPUT OVERVOLTAGE PROTECTION
The overvoltage comparator compares the FB pin voltage to a voltage that is 10% higher than the internal
reference Vref. Once the FB pin voltage goes 10% above the internal reference, the internal NMOS control
switch is turned off, which allows the output voltage to decrease toward regulation.
UNDERVOLTAGE LOCKOUT
Undervoltage lockout (UVLO) prevents the LMR12007 from operating until the input voltage exceeds 2.74V(typ).
The UVLO threshold has approximately 440mV of hysteresis, so the part will operate until VIN drops below
2.3V(typ). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
CURRENT LIMIT
The LMR12007 uses cycle-by-cycle current limiting to protect the output switch. During each switching cycle, a
current limit comparator detects if the output switch current exceeds 1.5A (typ), and turns off the switch until the
next switching cycle begins.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature
exceeds 165°C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction temperature
drops to approximately 150°C.
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L = VO + VD
IO x r x fSx (1-D)
r = 'iL
lO
D = VO + VD
VIN + VD - VSW
D = VO
VIN
LMR12007
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SNVS982 –SEPTEMBER 2013
Design Guide
INDUCTOR SELECTION
The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VO) to input voltage (VIN):
(12)
The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to
calculate a more accurate duty cycle. Calculate D by using the following formula:
(13)
VSW can be approximated by:
VSW = IOx RDS(ON) (14)
The diode forward drop (VD) can range from 0.3V to 0.7V depending on the quality of the diode. The lower VDis,
the higher the operating efficiency of the converter.
The inductor value determines the output ripple current. Lower inductor values decrease the size of the inductor,
but increase the output ripple current. An increase in the inductor value will decrease the output ripple current.
The ratio of ripple current (ΔiL) to output current (IO) is optimized when it is set between 0.3 and 0.4 at 750mA.
The ratio r is defined as:
(15)
One must also ensure that the minimum current limit (1.0A) is not exceeded, so the peak current in the inductor
must be calculated. The peak current (ILPK) in the inductor is calculated by:
ILPK = IO+ΔIL/2 (16)
If r = 0.7 at an output of 750mA, the peak current in the inductor will be 1.0125A. The minimum ensured current
limit over all operating conditions is 1.0A. One can either reduce r to 0.6 resulting in a 975mA peak current, or
make the engineering judgement that 12.5mA over will be safe enough with a 1.5A typical current limit and 6
sigma limits. When the designed maximum output current is reduced, the ratio r can be increased. At a current of
0.1A, r can be made as high as 0.9. The ripple ratio can be increased at lighter loads because the net ripple is
actually quite low, and if r remains constant the inductor value can be made quite large. An equation empirically
developed for the maximum ripple ratio at any current below 2A is:
r = 0.387 x IOUT-0.3667 (17)
Note that this is just a guideline.
The LMR12007 operates at frequencies allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.
See the OUTPUT CAPACITOR section for more details on calculating output voltage ripple.
Now that the ripple current or ripple ratio is determined, the inductance is calculated by:
(18)
where fsis the switching frequency and IOis the output current. When selecting an inductor, make sure that it is
capable of supporting the peak output current without saturating. Inductor saturation will result in a sudden
reduction in inductance and prevent the regulator from operating correctly. Because of the speed of the internal
current limit, the peak current of the inductor need only be specified for the required maximum output current. For
example, if the designed maximum output current is 0.5A and the peak current is 0.7A, then the inductor should
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IRMS-OUT = IO x r
12
'VO = 'iL x (RESR + 1
8 x fS x CO)
IRMS-IN = IO x D x r2
12
1-D +
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
be specified with a saturation current limit of >0.7A. There is no need to specify the saturation or peak current of
the inductor at the 1.5A typical switch current limit. The difference in inductor size is a factor of 5. Because of the
operating frequency of the LMR12007, ferrite based inductors are preferred to minimize core losses. This
presents little restriction since the variety of ferrite based inductors is huge. Lastly, inductors with lower series
resistance (DCR) will provide better operating efficiency. For recommended inductors see Example Circuits.
INPUT CAPACITOR
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and ESL (Equivalent
Series Inductance). The recommended input capacitance is 10µF, although 4.7µF works well for input voltages
below 6V. The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any
recommended deratings and also verify if there is any significant change in capacitance at the operating input
voltage and the operating temperature. The input capacitor maximum RMS input current rating (IRMS-IN) must be
greater than:
(19)
It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always
calculate the RMS at the point where the duty cycle, D, is closest to 0.5. The ESL of an input capacitor is usually
determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL
and a 0805 ceramic chip capacitor will have very low ESL. At the operating frequencies of the LMR12007,
certain capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than that required
to provide stable operation. As a result, surface mount capacitors are strongly recommended. Sanyo POSCAP,
Tantalum or Niobium, Panasonic SP or Cornell Dubilier ESR, and multilayer ceramic capacitors (MLCC) are all
good choices for both input and output capacitors and have very low ESL. For MLCCs it is recommended to use
X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over
operating conditions.
OUTPUT CAPACITOR
The output capacitor is selected based upon the desired output ripple and transient response. The initial current
of a load transient is provided mainly by the output capacitor. The output ripple of the converter is:
(20)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple will be approximately sinusoidal and 90° phase shifted from the switching action. Given the
availability and quality of MLCCs and the expected output voltage of designs using the LMR12007, there is really
no need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability to
bypass high frequency noise. A certain amount of switching edge noise will couple through parasitic
capacitances in the inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not.
Since the output capacitor is one of the two external components that control the stability of the regulator control
loop, most applications will require a minimum at 10 µF of output capacitance. Capacitance can be increased
significantly with little detriment to the regulator stability. Like the input capacitor, recommended multilayer
ceramic capacitors are X7R or X5R. Again, verify actual capacitance at the desired operating voltage and
temperature.
Check the RMS current rating of the capacitor. The RMS current rating of the capacitor chosen must also meet
the following condition:
(21)
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R1 =VO- 1
VREF x R2
LMR12007
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CATCH DIODE
The catch diode (D1) conducts during the switch off-time. A Schottky diode is recommended for its fast switching
times and low forward voltage drop. The catch diode should be chosen so that its current rating is greater than:
ID1 = IOx (1-D) (22)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency choose a Schottky diode with a low forward voltage drop.
BOOST DIODE
A standard diode such as the 1N4148 type is recommended. For VBOOST circuits derived from voltages less than
3.3V, a small-signal Schottky diode is recommended for greater efficiency. A good choice is the BAT54 small
signal diode.
BOOST CAPACITOR
A ceramic 0.01µF capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide
the best performance.
OUTPUT VOLTAGE
The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and
R1 is connected between VOand the FB pin. A good value for R2 is 10kΩ.
(23)
PCB Layout Considerations
When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The
most important consideration when completing the layout is the close coupling of the GND connections of the CIN
capacitor and the catch diode D1. These ground ends should be close to one another and be connected to the
GND plane with at least two through-holes. Place these components as close to the IC as possible. Next in
importance is the location of the GND connection of the COUT capacitor, which should be near the GND
connections of CIN and D1.
There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching
node island.
The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup
and inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with the GND
of R2 placed as close as possible to the GND of the IC. The VOUT trace to R1 should be routed away from the
inductor and any other traces that are switching.
High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible.
However, making the traces wide increases radiated noise, so the designer must make this trade-off. Radiated
noise can be decreased by choosing a shielded inductor.
The remaining components should also be placed as close as possible to the IC. Please see Application Note
AN-1229 SNVA054 for further considerations and the LMR12007 demo board as an example of a four-layer
layout.
Copyright © 2013, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
D2
ON
OFF
C1 R3
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
LMR12007X Circuit Examples
Figure 24. LMR12007X (1.6MHz)
VBOOST Derived from VIN
5V to 1.5V/750mA
Table 1. Bill of Materials for Figure 24
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007X Texas Instruments
C1, Input Cap 10µF, 6.3V, X5R C3216X5ROJ106M TDK
C2, Output Cap 10µF, 6.3V, X5R C3216X5ROJ106M TDK
C3, Boost Cap 0.01uF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.3VFSchottky 1A, 10VR MBRM110L ON Semi
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
L1 4.7µH, 1.7A, VLCF4020T- 4R7N1R2 TDK
R1 2kΩ, 1% CRCW06032001F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
14 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
D2
ON
OFF
VIN
C1 R3
LMR12007
www.ti.com
SNVS982 –SEPTEMBER 2013
Figure 25. LMR12007X (1.6MHz)
VBOOST Derived from VOUT
12V to 3.3V/750mA
Table 2. Bill of Materials for Figure 25
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007X Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.34VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 30V, 200 mA Schottky BAT54 Diodes Inc.
L1 4.7µH, 1.7A, VLCF4020T- 4R7N1R2 TDK
R1 16.5kΩ, 1% CRCW06031652F Vishay
R2 10.0 kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
Copyright © 2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
D2
ON
OFF
D3
C4
R4
C1 R3
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
Figure 26. LMR12007X (1.6MHz)
VBOOST Derived from VSHUNT
18V to 1.5V/750mA
Table 3. Bill of Materials for Figure 26
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007X Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
C4, Shunt Cap 0.1µF, 6.3V, X5R C1005X5R0J104K TDK
D1, Catch Diode 0.4VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
D3, Zener Diode 5.1V 250Mw SOT BZX84C5V1 Vishay
L1 6.8µH, 1.6A, SLF7032T-6R8M1R6 TDK
R1 2kΩ, 1% CRCW06032001F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
R4 4.12kΩ, 1% CRCW06034121F Vishay
16 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
ON
OFF
D2D3
C1 R3
LMR12007
www.ti.com
SNVS982 –SEPTEMBER 2013
Figure 27. LMR12007X (1.6MHz)
VBOOST Derived from Series Zener Diode (VIN)
15V to 1.5V/750mA
Table 4. Bill of Materials for Figure 27
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007X Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.4VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
D3, Zener Diode 11V 350Mw SOT BZX84C11T Diodes, Inc.
L1 6.8µH, 1.6A, SLF7032T-6R8M1R6 TDK
R1 2kΩ, 1% CRCW06032001F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
Copyright © 2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
ON
OFF
D2 D3
C1 R3
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
Figure 28. LMR12007X (1.6MHz)
VBOOST Derived from Series Zener Diode (VOUT)
15V to 9V/750mA
Table 5. Bill of Materials for Figure 28
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007X Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 16V, X5R C3216X5R1C226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.4VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
D3, Zener Diode 4.3V 350mw SOT BZX84C4V3 Diodes, Inc.
L1 6.8µH, 1.6A, SLF7032T-6R8M1R6 TDK
R1 61.9kΩ, 1% CRCW06036192F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
18 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
D2
ON
OFF
C1 R3
LMR12007
www.ti.com
SNVS982 –SEPTEMBER 2013
LMR12007Y Circuit Examples
Figure 29. LMR12007Y (550kHz)
VBOOST Derived from VIN
5V to 1.5V/750mA
Table 6. Bill of Materials for Figure 29
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007Y Texas Instruments
C1, Input Cap 10µF, 6.3V, X5R C3216X5ROJ106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.3VFSchottky 1A, 10VR MBRM110L ON Semi
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
L1 10µH, 1.6A, SLF7032T-100M1R4 TDK
R1 2kΩ, 1% CRCW06032001F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
Copyright © 2013, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LMR12007
VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
D2
ON
OFF
VIN
C1 R3
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
Figure 30. LMR12007Y (550kHz)
VBOOST Derived from VOUT
12V to 3.3V/750mA
Table 7. Bill of Materials for Figure 30
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007Y Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.34VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 30V, 200 mA Schottky BAT54 Diodes Inc.
L1 10µH, 1.6A, SLF7032T-100M1R4 TDK
R1 16.5kΩ, 1% CRCW06031652F Vishay
R2 10.0 kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
20 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
D2
ON
OFF
D3
C4
R4
C1 R3
LMR12007
www.ti.com
SNVS982 –SEPTEMBER 2013
Figure 31. LMR12007Y (550kHz)
VBOOST Derived from VSHUNT
18V to 1.5V/750mA
Table 8. Bill of Materials for Figure 31
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007Y Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
C4, Shunt Cap 0.1µF, 6.3V, X5R C1005X5R0J104K TDK
D1, Catch Diode 0.4VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
D3, Zener Diode 5.1V 250Mw SOT BZX84C5V1 Vishay
L1 15µH, 1.5A SLF7045T-150M1R5 TDK
R1 2kΩ, 1% CRCW06032001F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
R4 4.12kΩ, 1% CRCW06034121F Vishay
Copyright © 2013, Texas Instruments Incorporated Submit Documentation Feedback 21
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
ON
OFF
D2D3
C1 R3
LMR12007
SNVS982 –SEPTEMBER 2013
www.ti.com
Figure 32. LMR12007Y (550kHz)
VBOOST Derived from Series Zener Diode (VIN)
15V to 1.5V/750mA
Table 9. Bill of Materials for Figure 32
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007Y Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.4VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
D3, Zener Diode 11V 350Mw SOT BZX84C11T Diodes, Inc.
L1 15µH, 1.5A, SLF7045T-150M1R5 TDK
R1 2kΩ, 1% CRCW06032001F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
22 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
VIN VIN
EN
BOOST
SW
FB
GND
VOUT
C3 L1
C2
R1
R2
D1
ON
OFF
D2 D3
C1 R3
LMR12007
www.ti.com
SNVS982 –SEPTEMBER 2013
Figure 33. LMR12007Y (550kHz)
VBOOST Derived from Series Zener Diode (VOUT)
15V to 9V/750mA
Table 10. Bill of Materials for Figure 33
Part ID Part Value Part Number Manufacturer
U1 750mA Buck Regulator LMR12007Y Texas Instruments
C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK
C2, Output Cap 22µF, 16V, X5R C3216X5R1C226M TDK
C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK
D1, Catch Diode 0.4VFSchottky 1A, 30VR SS1P3L Vishay
D2, Boost Diode 1VF@ 50mA Diode 1N4148W Diodes, Inc.
D3, Zener Diode 4.3V 350mw SOT BZX84C4V3 Diodes, Inc.
L1 22µH, 1.4A, SLF7045T-220M1R3-1PF TDK
R1 61.9kΩ, 1% CRCW06036192F Vishay
R2 10kΩ, 1% CRCW06031002F Vishay
R3 100kΩ, 1% CRCW06031003F Vishay
Copyright © 2013, Texas Instruments Incorporated Submit Documentation Feedback 23
Product Folder Links: LMR12007
PACKAGE OPTION ADDENDUM
www.ti.com 28-Feb-2017
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LMR12007XMK ACTIVE SOT-23-THIN DDC 6 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SP1B
LMR12007XMKX ACTIVE SOT-23-THIN DDC 6 3000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SP1B
LMR12007YMK ACTIVE SOT-23-THIN DDC 6 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SP2B
LMR12007YMKX ACTIVE SOT-23-THIN DDC 6 3000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SP2B
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
PACKAGE OPTION ADDENDUM
www.ti.com 28-Feb-2017
Addendum-Page 2
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LMR12007XMK SOT-
23-THIN DDC 6 1000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMR12007XMKX SOT-
23-THIN DDC 6 3000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMR12007YMK SOT-
23-THIN DDC 6 1000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
LMR12007YMKX SOT-
23-THIN DDC 6 3000 178.0 8.4 3.2 3.2 1.4 4.0 8.0 Q3
PACKAGE MATERIALS INFORMATION
www.ti.com 12-Jun-2018
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LMR12007XMK SOT-23-THIN DDC 6 1000 210.0 185.0 35.0
LMR12007XMKX SOT-23-THIN DDC 6 3000 210.0 185.0 35.0
LMR12007YMK SOT-23-THIN DDC 6 1000 210.0 185.0 35.0
LMR12007YMKX SOT-23-THIN DDC 6 3000 210.0 185.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 12-Jun-2018
Pack Materials-Page 2
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Life-critical medical equipment is medical equipment where failure of such equipment would cause serious bodily injury or death (e.g., life
support, pacemakers, defibrillators, heart pumps, neurostimulators, and implantables). Such equipment includes, without limitation, all
medical devices identified by the U.S. Food and Drug Administration as Class III devices and equivalent classifications outside the U.S.
TI may expressly designate certain products as completing a particular qualification (e.g., Q100, Military Grade, or Enhanced Product).
Designers agree that it has the necessary expertise to select the product with the appropriate qualification designation for their applications
and that proper product selection is at Designers’ own risk. Designers are solely responsible for compliance with all legal and regulatory
requirements in connection with such selection.
Designer will fully indemnify TI and its representatives against any damages, costs, losses, and/or liabilities arising out of Designer’s non-
compliance with the terms and provisions of this Notice.
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